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HT4817 pin to pin SN4088
Dual 2.2W AUDIO AMPLIFIER Plus Stereo Headphone
General Description
Key Specifications
The HT4818/7 is a dual bridge-connected audio
power amplifier which, when connected to a 5V
supply, will deliver 2.2W to a 4Ω
load with less
than 1.0% THD+N. In addition, the headphone
input pin allows the amplifiers to operate in
single-ended mode when driving stereo
Boomer audio power amplifiers were designed
specifically to provide high quality output power
from a surface mount package while requiring
few external components. To simplify audio
system design, the HT4818/7 combines dual
bridge speaker amplifiers and stereo headphone
amplifiers on one chip.
The HT4818/7 features an externally controlled,
low-power consumption shutdown mode, a
stereo headphone amplifier mode, and thermal
shutdown protection. It also utilizes circuitry to
reduce “clicks and pops”during device turn-on.
◆ PO at 1% THD+N
loads 2.2W(typ), 8Ω
load 1.1W(typ)
◆ Single-ended mode THD+N at 75mW into 32
◆ Shutdown current 0.7μA(typ)
◆ Supply voltage range 2.5 V to 5.5V
Stereo headphone amplifier mode
“Click and pop”suppression circuitry
Unity-gain stable
Thermal shutdown protection circuitry
QFN24/QFN16 packages
◆Cell phones
◆Multimedia monitors
◆Portable and desktop computers
◆Portable audio systems
Connection Diagrams
Typical Application
Absolute Maximum Ratings
Supply Voltage
Storage Temperature
Input Voltage
Power Dissipation
ESD Susceptibility
ESD Susceptibility
Junction Temperature
−65°C to +150°C
−0.3V to VDD+0.3V
Internally limited
Solder Information
Small Outline Package
Vapor Phase (60 sec.)
Infrared (15 sec.)
Thermal Resistance
θJC (typ)— SQA24B
θJ A (typ)— SQA24B
Operating Ratings
Temperature Range
TMIN ≤T A ≤
Supply Voltage
−40°C ≤
TA ≤
2.7V ≤
VDD ≤
Electrical Characteristics (5V)
The following specifications apply for VDD = 5V unless otherwise noted. Limits apply for TA = 25°C.
Electrical Characteristics for Single-Ended Operation (5V)
The following specifications apply for VDD = 5V unless otherwise specified. Limits apply for T A = 25°C.
Electrical Characteristics for Bridged-Mode Operation (5V)
The following specifications apply for VDD = 5V unless otherwise specified. Limits apply for T A = 25°C.
Electrical Characteristics (3V)
The following specifications apply for VDD = 3V unless otherwise noted. Limits apply for TA = 25°C.
Electrical Characteristics for Bridged-Mode Operation (3V)
The following specifications apply for VDD = 3V unless otherwise specified. Limits apply for T A = 25°C.
Electrical Characteristics for Single-Ended Operation (3V)
The following specifications apply for VDD = 3V unless otherwise specified. Limits apply for T A = 25°C.
Typical Performance Characteristics
Application Information
The HT4818/7’
s SQ exposed-DAP (die attach paddle) package provides a low thermal resistance between the die and
the PCB to which the part is mounted and soldered. This allows rapid heat transfer from the die to the surrounding PCB
copper traces, ground plane and, finally, surrounding air. The result is a low voltage audio power amplifier that
produces 2.1W at
1% THD with a 4Ω
load. This high power is achieved through careful consideration of necessary
thermal design. Failing to optimize thermal design may compromise the HT4818/7’
s high power performance and
activate unwanted, though necessary, thermal shutdown protection. The SQ package must have its DAP soldered to a
copper pad on the PCB. The DAP’
s PCB copper pad is connected to a large plane of continuous unbroken copper.
This plane forms a thermal mass and heat sink and radiation area. Place the heat sink area on either outside plane in
the case of a two-sided PCB, or on an inner layer of a board with more than two layers. Connect the DAP copper pad to
the inner layer or backside copper heat sink area with 6 (3x2) SQ vias. The via diameter should be 0.012in–0.013in
with a 1.27mm pitch. Ensure efficient thermal conductivity by platingthrough and solder-filling the vias.
Best thermal performance is achieved with the largest practical copper heat sink area. If the heatsink and amplifier
share the same PCB layer, a nominal 2.5in2 (min) area is necessary for 5V operation with a 4Ω
load. Heatsink areas
not placed on the same PCB layer as the HT4818/7 should be 5in2 (min) for the same supply voltage and load
resistance. The last two area recommendations apply for 25°C ambient temperature. Increase the area to compensate
for ambient temperatures above 25°C. In all circumstances and conditions, the junction temperature must be held
below 150°C to prevent activating the HT4818/7’
s thermal shutdown protection. The HT4818/7’
s power de-rating curve
in the Typical Performance Characteristics shows the maximum power dissipation versus temperature. Example
PCB layouts for the exposed-Dap SQ package is shown in the Demonstration Board Layout section.
Power dissipated by a load is a function of the voltage swing across the load and the load’
s impedance. As load
impedance decreases, load dissipation becomes increasingly dependent on the interconnect (PCB trace and wire)
resistance between the amplifier output pins and the load’
s connections. Residual trace resistance causes a voltage
drop, which results in power dissipated in the trace and not in the load as desired. For example, 0.1Ω
trace resistance
reduces the output power dissipated by a 4Ω
load from 2.1W to 2.0W. This problem of decreased load dissipation is
exacerbated as load impedance decreases. Therefore, to maintain the highest load dissipation and widest output
voltage swing, PCB traces that connect the output pins to a load must be as wide as possible.
Poor power supply regulation adversely affects maximum output power. A poorly regulated supply’
s output voltage
decreases with increasing load current. Reduced supply voltage causes decreased headroom, output signal clipping,
and reduced output power. Even with tightly regulated supplies, trace resistance creates the same effects as poor
supply regulation. Therefore, making the power supply traces as wide as possible helps maintain full output voltage
As shown in Figure 1, the HT4818/7 consists of two pairs of operational amplifiers, forming a two-channel (channel A
and channel B) stereo amplifier. External feedback resistors R2 (or R3, R4) and R8 (or R6, R7) and input resistors R1
and R9 set the closed-loop gain of Amp A (-out) and Amp B (-out) whereas two internal 20kΩ
resistors set Amp A’
(+out) and Amp B’
s (+out) gain at 1. The HT4818/7 drives a load, such as a speaker, connected between the two
amplifier outputs, −OUTA and +OUTA.
Figure 1 shows that Amp A’
s (-out) output serves as Amp A’
s (+out) input. This results in both amplifiers producing
signals identical in magnitude, but 180°out of phase. Taking advantage of this phase difference, a load is placed
between −OUTA and +OUTA and driven differentially (commonly referred to as “bridge mode”). This results in a
differential gain of
AVD = 2 * (Rf /R i )
AVD = 2 * (R2 /R 1)
Bridge mode amplifiers are different from single-ended amplifiers that drive loads connected between a single
s output and ground. For a given supply voltage, bridge mode has a distinct advantage over the single-ended
configuration: its differential output doubles the voltage swing across the load. This produces four times the output
power when compared to a single-ended amplifier under the same conditions. This increase in attainable output power
assumes that the amplifier is not current limited or that the output signal is not clipped. To ensure minimum output
signal clipping when choosing an amplifier’
s closed-loop gain, refer to the Audio Power Amplifier Design section.
Another advantage of the differential bridge output is no net DC voltage across the load. This is accomplished by
biasing channel A’
s and channel B’
s outputs at half-supply. This eliminates the coupling capacitor that single supply,
singleended amplifiers require. Eliminating an output coupling capacitor in a single-ended configuration forces a
single-supply amplifier’
s half-supply bias voltage across the load. This increases internal IC power dissipation and may
permanently damage loads such as speakers.
Power dissipation is a major concern when designing a successful single-ended or bridged amplifier. Equation (2)
states the maximum power dissipation point for a singleended amplifier operating at a given supply voltage and driving
a specified output load.
PDMAX = (VDD) /(2π2 R L)
However, a direct consequence of the increased power delivered to the load by a bridge amplifier is higher internal
power dissipation for the same conditions.
The HT4818/7 has two operational amplifiers per channel. The maximum internal power dissipation per channel
operating in the bridge mode is four times that of a single-ended amplifier. From Equation (3), assuming a 5V power
supply and a 4Ω
load, the maximum single channel power dissipation is 1.27W or 2.54W for stereo operation.
PDMAX = 4 * (VDD) /(2π2R L)
Bridge Mode
The HT4818/7’
s power dissipation is twice that given by Equation (2) or Equation (3) when operating in the
single-ended mode or bridge mode, respectively. Twice the maximum power dissipation point given by Equation (3)
must not exceed the power dissipation given by Equation (4):
The HT4818/7’
s TJMAX = 150°C. In the SQ package soldered to a DAP pad that expands to a copper area of 5in 2 on a
PCB, the HT4818/7’
s θJA is 20°C/W. At any given ambient temperature TA, use Equation (4) to find the maximum
internal power dissipation supported by the IC packaging. Rearranging Equation (4) and substituting PDMAX for PDMAX '
results in Equation (5). This equation gives the maximum ambient temperature that still allows maximum stereo power
dissipation without violating the HT4818/7’
s maximum junction temperature.
For a typical application with a 5V power supply and an 4
load, the maximum ambient temperature that allows
maximum stereo power dissipation without exceeding the maximum junction temperature is approximately 99°C for the
SQ package.
Equation (6) gives the maximum junction temperature TJMAX. If the result violates the HT4818/7’
s 150°C, reduce the
maximum junction temperature by reducing the power supply voltage or increasing the load resistance. Further
allowance should be made for increased ambient temperatures. The above examples assume that a device is a
surface mount part operating around the maximum power dissipation point. Since internal power dissipation is a
function of output power, higher ambient temperatures are allowed as output power or duty cycle decreases.
If the result of Equation (2) is greater than that of Equation (3), then decrease the supply voltage, increase the load
impedance, or reduce the ambient temperature. If these measures are insufficient, a heat sink can be added to reduce
JA . The heat sink can be created using additional copper area around the package, with connections to the ground
pin(s), supply pin and amplifier output pins. External, solder attached SMT heatsinks such as the Thermalloy 7106D
can also improve power dissipation. When adding a heat sink, the 
JA is the sum of 
JC , 
CS , and 
SA . (
JC is the
junction-to-case thermal impedance, 
CS is the case-to-sink thermal impedance, and 
SA is the sink-to-ambient thermal
impedance.) Refer to the Typical Performance Characteristics curves for power dissipation information at lower
output power levels.
As with any power amplifier, proper supply bypassing is critical for low noise performance and high power supply
rejection. Applications that employ a 5V regulator typically use a 10 µF in parallel with a 0.1 µF filter capacitor to
stabilize the regulator’
s output, reduce noise on the supply line, and improve the supply’
s transient response. However,
their presence does not eliminate the need for a local 1.0 µF tantalum bypass capacitance connected between the
s supply pins and ground. Do not substitute a ceramic capacitor for the tantalum. Doing so may cause
oscillation. Keep the length of leads and traces that connect capacitors between the HT4818/7’
s power supply pin and
ground as short as possible.
The voltage applied to the SHUTDOWN pin controls the HT4818/7’
s shutdown function. Activate micro-power
shutdown by applying GND to the SHUTDOWN pin. When active, the HT4818/7’
s micro-power shutdown feature turns
off the amplifier’
s bias circuitry, reducing the supply current. The low 0.04 µA typical shutdown current is achieved by
applying a voltage that is as near as GND as possible to the SHUTDOWN pin. A voltage that is more than GND may
increase the shutdown current. Table 1 shows the logic signal levels that activate and deactivate micro-power
shutdown and headphone amplifier operation.
There are a few ways to control the micro-power shutdown. These include using a single-pole, single-throw switch, a
microprocessor, or a microcontroller. When using a switch, connect an external 100k resistor between the
SHUTDOWN pin and Ground. Connect the switch between the SHUTDOWN pin VDD. Select normal amplifier
operation by closing the switch. Opening the switch sets the SHUTDOWN pin to ground through the 100k resistor,
which activates the micropower shutdown. The switch and resistor guarantee that the SHUTDOWN pin will not float.
This prevents unwanted state changes. In a system with a microprocessor or a microcontroller, use a digital output to
apply the control voltage to the SHUTDOWN pin. Driving the SHUTDOWN pin with active circuitry eliminates the pull
up resistor.
Applying a logic level to the HT4818/7’
s HP Sense headphone control pin turns off Amp A (+out) and Amp B (+out)
muting a bridged-connected load. Quiescent current consumption is reduced when the IC is in this single-ended mode.
Figure 2 shows the implementation of the HT4818/7’
s headphone control function. With no headphones connected to
the headphone jack, the R11-R13 voltage divider sets the voltage applied to the HP Sense pin (pin 20) at
approximately 50mV. This 50mV enables Amp A (+out) and Amp B (+out) placing the HT4818/7 in bridged mode
While the HT4818/7 operates in bridged mode, the DC potential across the load is essentially 0V. Therefore, even in an
ideal situation, the output swing cannot cause a false singleended trigger. Connecting headphones to the headphone
jack disconnects the headphone jack contact pin from −OUTA and allows R13 to pull the HP Sense pin up to V DD. This
enables the headphone function, turns off Amp A (+out) and Amp B (+out) which mutes the bridged speaker. The
amplifier then drives the headphones, whose impedance is in parallel with resistors R10 and R11. These resistors have
negligible effect on the HT4818/7’
s output drive capability since the typical impedance of headphones is 32.
Figure 2 also shows the suggested headphone jack electrical connections. The jack is designed to mate with a
threewire plug. The plug’
s tip and ring should each carry one of the two stereo output signals, whereas the sleeve
should carry the ground return. A headphone jack with one control pin contact is sufficient to drive the HP Sense pin
when connecting headphones.
There is also a second input circuit that can control the choice of either BTL or SE modes. This input control pin is
called the HP (Headphone) Logic Input. When the HP Logic input is high, HT4818/7 operates in SE mode. When HP
Logic is low (& the HP Sense pin is low), the HT4818/7 operates in the BTL mode. In the BTL mode (HP Logic low and
HP Sense Low) if the Headphones are connected directly to the Single Ended outputs (not using the HP Sense pin on
the HP Jack) then both the Speaker (BTL) and Headphone (SE) will be functional. In this case the inverted op amp
outputs drive the Speaker as well as the HP load, i.e. 8 ohms in parallel with 32 ohms. As the HT4818/7 is capable of
driving up to a 3 ohm load driving the Speakers and the Headphones at the same time will not be a problem as long as
the parallel resistance of each Speaker and each Headphone driver are more than 3 ohms.
As outlined above driving the Speaker (BTL) and Headphone (SE) loads simultaneously using HT4818/7 is simple and
easy. However this configuration will only work if the HP Logic pin is used to control the BTL/SE operation and HP
Sense pin is connected to GND.
Optimizing the HT4818/7’
s performance requires properly selecting external components. Though the HT4818/7
operates well when using external components with wide tolerances, best performance is achieved by optimizing
component values. The HT4818/7 is unity-gain stable, giving a designer maximum design flexibility. The gain should be
set to no more than a given application requires. This allows the amplifier to achieve minimum THD+N and maximum
signal-to-noise ratio. These parameters are compromised as the closed-loop gain increases. However, low gain
demands input signals with greater voltage swings to achieve maximum output power. Fortunately, many signal
sources such as audio CODECs have outputs of 1VRMS (2.83VP-P). Please refer to the Audio Power Amplifier Design
section for more information on selecting the proper gain.
Input Capacitor Value Selection
Amplifying the lowest audio frequencies requires high value input coupling capacitors (C1 and C2) in Figure 1. A high
value capacitor can be expensive and may compromise space efficiency in portable designs. In many cases, however,
the speakers used in portable systems, whether internal or external, have little ability to reproduce signals below 150
Hz. Applications using speakers with this limited frequency response reap little improvement by using large input
Besides effecting system cost and size, C1 and C2 have an effect on the HT4818/7’
s click and pop performance. When
the supply voltage is first applied, a transient (pop) is created as the charge on the input capacitor changes from zero to
a quiescent state. The magnitude of the pop is directly proportional to the input capacitor’
s size. Higher value
capacitors need more time to reach a quiescent DC voltage (usually VDD/2) when charged with a fixed current. The
s output charges the input capacitor through the feedback resistors, R2 and R8. Thus, pops can be minimized
by selecting an input capacitor value that is no higher than necessary to meet the desired −3dB frequency.
A shown in Figure 1, the input resistors (R1,4,5, and 6) and the input capacitors, C1 and C2 produce a −3dB high pass
filter cutoff frequency that is found using Equation (7). (7) As an example when using a speaker with a low frequency
limit of 150Hz, C1, using Equation (7) is 0.053µF. The .33µF C1 shown in Figure 1 allows the HT4818/7 to drive high
efficiency, full range speaker whose response extends below 30Hz.
Bypass Capacitor Value Selection
Besides minimizing the input capacitor size, careful consideration should be paid to value of C6 , the capacitor
connected to the BYPASS pin. Since C 6 determines how fast the HT4818/7 settles to quiescent operation, its value is
critical when minimizing turn-on pops. The slower the HT4818/7’
s outputs ramp to their quiescent DC voltage
(nominally 1/2 VDD), the smaller the turn-on pop. Choosing C6 equal to 1.0 µF along with a small value of C1 (in the
range of 0.1 µF to 0.39 µF), produces a click-less and pop-less shutdown function. As discussed above, choosing C1
no larger than necessary for the desired bandwith helps minimize clicks and pops. Connecting a 1µF capacitor, C6,
between the BYPASS pin and ground improves the internal bias voltage’
s stability and improves the amplifier’
The HT4818/7 contains circuitry that minimizes turn-on and shutdown transients or “clicks and pop”. For this discussion,
turn-on refers to either applying the power supply voltage or when the shutdown mode is deactivated. When the part is
turned on, an internal current source changes the voltage of the BYPASS pin in a controlled, linear manner. Ideally, the
input and outputs track the voltage applied to the BYPASS pin. The gain of the internal amplifiers remains unity until the
voltage on the bypass pin reaches 1/2 VDD. As soon as the voltage on the bypass pin is stable, the device becomes
fully operational. Although the BYPASS pin current cannot be modified, changing the size of C 6 alters the device’
turn-on time and the magnitude of “clicks and pops”. Increasing the value of C6 reduces the magnitude of turn-on pops.
However, this presents a tradeoff: as the size of C6 increases, the turn -on time increases. There is a linear relationship
between the size of C 6 and the turn-on time. Here are some typical turn-on times for various values of C6:
In order eliminate “clicks and pops”, all capacitors must be discharged before turn-on. Rapidly switching V DD on and off
may not allow the capacitors to fully discharge, which may cause “clicks and pops”.
Audio Amplifier Design: Driving 1W into an 8Ω
The following are the desired operational parameters:
Power Output:
Load Impedance:
Input Level:
Input Impedance:
100Hz−20kHz ± 0.25dB
The design begins by specifying the minimum supply voltage necessary to obtain the specified output power. One way
to find the minimum supply voltage is to use the Output Power vs Supply Voltage curve in the Typical Performance
Characteristics section. Another way, using Equation (8), is to calculate the peak output voltage necessary to achieve
the desired output power for a given load impedance. To account for the amplifier’
s dropout voltage, two additional
voltages, based on the Dropout Voltage vs Supply Voltage in the Typical Performance Characteristics curves, must
be added to the result obtained by Equation (8). The result in Equation (9).
The Output Power vs Supply Voltage graph for an 8Ω
load indicates a minimum supply voltage of 4.35V for a 1W
output at 1% THD+N. This is easily met by the commonly used 5V supply voltage. The additional voltage creates the
benefit of headroom, allowing the HT4818/7 to produce peak output power in excess of 1.3W at 5V of VDD and 1%
THD+N without clipping or other audible distortion. The choice of supply voltage must also not create a situation that
violates maximum power dissipation as explained above in the Power Dissipation section.
After satisfying the HT4818/7’
s power dissipation requirements, the minimum diffe rential gain needed to achieve 1W
dissipation in an 8Ω
load is found using Equation (10).
Thus, a minimum gain of 2.83 allows the HT4818/7’
s to reach full output swing and maintain low noise and THD+N
performance. For this example, let A VD = 3.
The amplifier’
s overall gain is set using the input (R1 and R9) and feedback resistors R2 and R8. With the desired input
impedance set at 20kΩ, the feedback resistor is found using Equation (11).
R2/R 1 = AVD/2
The value of Rf is 30kΩ.
The last step in this design example is setting the amplifier’
s −3dB frequency bandwidth. To achieve the desired
±0.25dB pass band magnitude variation limit, the low frequency response must extend to at least one-fifth the lower
bandwidth limit and the high frequency response must extend to at least five times the upper bandwidth limit. The gain
variation for both response limits is 0.17dB, well within the ±0.25dB desired limit. The results are an
fL = 100Hz/5 = 20Hz
and an
fH = 20kHz*5 = 100kHz.
As mentioned in the External Components section, R1 and C1 create a highpass filter that sets the amplifier’
s lower
bandpass frequency limit. Find the coupling capacitor’
s value using Equation (12).
C1 ≥1/(2πR 1f L)
The result is
1/(2π*20kΩ*20Hz) = 0.398µF.
Use a 0.39µF capacitor, the closest standard value. The product of the desired high frequency cutoff (100kHz in this
example) and the differential gain, AVD, determines the upper passband response limit. With AVD = 3 and fH =100kHz,
the closed-loop gain bandwidth product (GBWP) is 300kHz. This is less than the HT4818/7’
s 3.5MHz GBWP. With this
margin, the amplifier can be used in designs that require more differential gain while avoiding performance-restricting
bandwidth limitations.
Figures 3 through 6 show the recommended two-layer PC board layout that is optimized for the 24-pin SQ package.
These circuits are designed for use with an external 5V supply and 8Ω, 4Ω, 3Ω
speakers. These circuit boards are
easy to use. Apply power and ground to the board’
s VDD and GND pads, respectively. Connect the speakers between
the board’
s −OUTA and +OUTA and OUTB and +OUTB pads.
Physical Dimensions inches (millimeters) unless otherwise noted
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