LT Journal - March 2005
LINEAR TECHNOLOGY
MARCH 2005
COVER ARTICLE
Smart Batteries: Not Just for
Notebooks Anymore ....................... 1
Mark Gurries
Issue Highlights ............................ 2
LTC in the News… .......................... 2
DESIGN FEATURES
Dual Monolithic Ideal Diodes
Provide a Single-Chip Power
Management Solution .................... 7
Andy Bishop
Simplify PoE Implementation
with Complete PD Interface and
Integrated Switching Regulator
.................................................... 11
Kirk Su
High Accuracy Clock
up to 170MHz in a SOT-23 ........... 14
Albert Huntington
Simple, Precise Instrumentation
Amplifier Features Digitally
Programmable Gains from 1 to 4096
.................................................... 16
Michael Kultgen
Flexible Power Supply
Sequencing and Monitoring ......... 20
Jeff Heath and Akin Kestelli
Smart Batteries: Not Just
for Notebooks Anymore
by Mark Gurries
Introduction
With the backing of Intel and Microsoft, Smart Batteries have become the
dominant battery pack solution for
products that require an accurate gas
gauge to predict battery life. The Smart
Battery System (SBS) has simplified
the design of standalone battery systems so much that it is showing up in
applications outside its usual realm of
notebook computers. For instance, the
SBS is gaining popularity in backup
power systems for mission critical high
reliability applications.
The attraction of the SBS is that its
modular nature makes easy to design
a closed loop battery-charge system,
and upgrade components as needed.
All of the safety features are taken
into account within the battery. This
minimizes NRE costs and makes for
robust systems, especially important
to high reliability battery-backup applications. There is no need to become
a battery expert to take advantage of
the features of the SBS.
The first part of this article offers an
overview of the SBS; the second part
describes two of Linear Technology’s
Smart Battery Chargers.
continued on page 3
SMART BATTERY
UNREGULATED
POWER
(WALL ADAPTER)
IN THIS ISSUE…
VOLUME XV NUMBER 1
DISCHARGE CHARGE
SAFETY
SIGNAL
High Voltage Step-Down
Synchronous Controller Offers
Single-Supply Operation, Current
Mode Control, and 100µA Burst
Mode® Operation .......................... 25
SMART
BATTERY
CHARGER
CLK
CLK
DATA
DATA
PROTECTION
CIRCUIT
BATTERY
CELLS
GAS GAUGE
Jay Celani
DESIGN IDEAS
............................................... 29–36
CURRENT
SENSE
(complete list on page 29)
New Device Cameos...................... 37
Design Tools ................................ 39
Sales Offices................................ 40
SMBUS
CLK
DATA
Figure 1. Simplified schematic of a Smart Battery and Smart Battery Charger. Smart Batteries
have an integrated gas gauge, which communicates the condition of the battery, and requests
charge (voltage and current) over the SMBus. Charge requests are satisfied by the Smart Battery
Charger, which applies the requested voltage and current to the battery terminals. The beauty of
the system is that the charger does not need to know the chemistry of the battery. It is up to the
gas gauge to maintain the charge algorithms. The modular nature of the SBS allows for any Smart
Battery Charger to charge any Smart Battery.
, LTC, LT, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology Corporation. Adaptive Power, C-Load, DirectSense, FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD, Multimode
Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter, PanelProtect, PowerPath, PowerSOT,
SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, UltraFast and VLDO are trademarks of Linear Technology
Corporation. Other product names may be trademarks of the companies that manufacture the products.
EDITOR’S PAGE
Issue Highlights
W
ith the backing of Intel and
Microsoft, Smart Batteries have
become the dominant battery
pack solution for products that require
an accurate gas gauge to predict battery life. The Smart Battery System
(SBS) has simplified the design of
standalone battery systems so much
that it is showing up in applications
outside its usual realm of notebook
computers.
Our cover article shows how Linear
Technology’s Smart Battery Charger
circuits simplify the design of high performance Smart Battery Systems.
Featured Devices
Below is a summary of the other devices featured in this issue.
Complete PowerPath™ Management
Squeezed into 9mm2
The LTC®4413 dual monolithic ideal
diode makes it possible to build an
entire power management solution in
a 3mm × 3mm footprint. (Page 7)
Power over Ethernet Simplified
The LTC4267 reduces the complexity
and size of a PD by combining an IEEE
802.3af-compliant PD interface with
a current mode switching regulator
in a space-saving, low-profile 16-pin
SSOP or DFN package. (Page 11)
Tougher than a Crystal Oscillator:
An All Silicon Clock
The LTC6905 is an all silicon clock
that avoids the pitfalls of sensitive
and power hungry crystal oscillators.
(Page 14)
Wide Range, Digitally
Programmable Voltage Gain
The LTC6915 is the simplest, most
precise way to obtain digitally programmable voltage gain. (Page 16)
Combo Power Supply
Sequencer and Supervisor
The LTC2924 is a complete power
supply sequencer and supervisor
2
solution for multivoltage-rail systems, such as telecommunications
equipment, memory modules, optical
systems, networking equipment, servers, and base stations. (Page 20)
High Voltage DC/DC Conversion
The LT®3800 is a feature-packed high
voltage synchronous step-down
controller that helps meet the high-
power point-of-load requirements of
distributed power systems. (Page 25)
Design Ideas and Cameos
The Design Ideas start on page 29,
including a low harmonic distortion,
amplitude stable sine wave generator
using the LT1968 RMS-to-DC converter. Five New Device Cameos appear
on page 37.
Linear Technology in the News…
Linear Technology Offers All Products in Lead-Free Versions
as Part of Broader Commitment to Protecting the Environment
All of Linear Technology’s products are now available in lead-free versions as an ordering option. This is part of the company’s commitment to
meeting the European Community’s Reduction of Hazardous Substances
(RoHS) guidelines, and represents the latest step in Linear Technology’s
ongoing commitment to excellence and leadership in protecting the environment.
After evaluating various lead-free alternatives, Linear Technology selected Matte Tin as the optimal plating for lead-free products. The company
believes that this provides the best drop-in replacement since it has the
lowest reflow temperature of the lead-free alternatives, has excellent solderability performance, and provides excellent quality and reliability. In
order to ensure a smooth transition for customers, Linear Technology will
continue to offer solder plated products as well, with unique part numbers
for both the lead-free and solder plated versions.
In addition to offering lead-free plating, Linear Technology will also
provide by mid-2005 an improved plastic mold compound that eliminates
antimony trioxide and elemental bromine for a more environmentally
friendly packaging alternative. The company’s goal is to replace these
flame retardants with environmentally friendly alternatives that meet
flammability standards, while improving the product reliability.
In early 2002, Linear Technology was certified to ISO 14001 compliance
by implementing an environmental management system. The company is
dedicated to making an ongoing contribution to environmental excellence
in all of its operations.
Electronic Products Magazine Awards Linear Technology
“Product of the Year” for Compact Photoflash Capacitor Charger
From the thousands of products introduced in 2004, the editors of Electronic
Products have chosen what they feel are among the most outstanding—
based on significant advances in technology or its application, a decided
innovation in design, or a substantial gain in price-performance.
The LT3468 photoflash capacitor charger is one of these products (see
Electronic Products, Jan. 2004, p. 84 and Linear Technology Magazine,
Dec. 2003, p. 1). It enables the use of a Xenon flash lamp in a 3-Mpixel
camera phone—providing excellent photographic performance in a small
form factor and at a reasonable cost.
Linear Technology Magazine • March 2005
DESIGN FEATURES
Smart Batteries, continued from page 1
More Information about Smart Batteries…
❑ SBS gas gauge ...................www.sbs-forum.org/specs/sbdat110.pdf
❑ Smart Batter Charger ........www.sbs-forum.org/specs/sbc110.pdf
❑ Smart Battery System Manager
(SBSM) ..............................www.sbs-forum.org/specs/sbsm100b.pdf
❑ SMBus ..............................www.smbus.org/specs
About the Smart
Battery System
Figure 1 shows a simplified block
diagram of a Smart Battery. The
biggest benefit of the Smart Battery
System is the highly accurate gas
gauge integrated into the battery pack.
The gas gauge, as the name implies,
indicates how much energy is left in
the battery.
An integrated gas gauge can monitor the battery even when the battery
is on the shelf, and the gas gauge is
calibrated to a single battery, so accuracy is assured. A host-based gas
gauge can’t match this. Gas gauge
measurement techniques have evolved
to the point that the latest gas gauges
are self-calibrating—error rates are
at 1% over the lifetime of the battery.
Because the gas gauge knows the
battery better than any other circuit
can, it is responsible for charge and
discharge management.
tery, the gas gauge must work with an
external charger. Smart Batteries are
designed to work with Smart Battery
Chargers. A Smart Battery Charger
has advantages over a fixed standalone
charger, such as:
❑ True Plug and Play operation,
independent of battery chemistry
and cell-configuration. Any Smart
Battery Pack will work with any
Smart Battery Charger. Batteries
with different chemistries, cellconfigurations, and even different charge algorithms can be
swapped with no modification to
the charger circuit.
❑ Built in safety features. The battery, because of its gas gauge,
takes care of itself.
❑ A reliable battery detection system.
❑ Automatic charge management
without the need of a host processor.
❑ Any Smart Battery and Smart
Charger form a closed loop
charge system that requires no
Smart Battery Chargers
In an SBS Li-ion battery pack, there
are built-in MOSFETS that can block
charge or discharge current (see Figure 1). The SBS gas gauge can easily
prevent over-discharge by turning off
the discharge MOSFET when the battery voltage reaches a certain point;
but when it comes to charging the bat-
host processor intervention. A
host is welcome to gather gas
gauge information if required.
To understand how all this is possible, let’s see how the Smart Battery
system actually works.
How SMBus is used for a
Closed Loop Charge Process
The Smart Battery System utilizes the
System Management Bus or SMBus
standard, which is a based on, and a
subset of, the very popular and now
recently made public domain two wire
I2C serial bus standard. In practice
SMBus devices easily coexist with I2C
devices on the same bus.
The Smart Battery Standard defines
fixed addresses for the battery and
charger, and it defines some commands that allow the gas gauge to
communicate to the Smart Battery
Charger over the SMBus. This forms a
closed loop system where the gas gauge
determines the charge state of the battery, and evaluates other conditions
(such as battery temperature) to see
DCIN
3V
TO 5.5V
1.21k
17
11
6
CHGEN
10
ACP
7
9
8
15
16
13
1.13k
14
10k
54.9k
0.1µF
13.7k
20
LTC4100
VDD
DCIN
DCDIV
INFET
CHGEN
CLP
ACP
CLN
SMBALERT
TGATE
SCL
BGATE
SDA
PGND
THB
CSP
THA
BAT
ILIM
VSET
VLIM
ITH
IDC
GND
0.082µF
0.1µF
5
4
0.033Ω
4.99k
24
23
20µF
1
3
SMART BATTERY
0.025Ω
10µH
2
20µF
21
22
18
19
12
0.01µF
6.04k
0.12µF
100Ω
0.0015µF
0.1µF
SafetySignal
SMBALERT#
SMBCLK
SMBCLK
SMBDAT
SMBDAT
Figure 2. The LTC4100 in a feature-rich, simple and compact 4A Smart Battery charger
Linear Technology Magazine • March 2005
3
DESIGN FEATURES
if a safe charge can be performed. The
gas gauge requests a charge current
and voltage from the Smart Charger via
the SMBus. Because the gas gauge is
in charge, the charger is not burdened
with algorithms involving the battery
cell configuration or chemistry.
When the gas gauge requests
charge, the Smart Charger evaluates
the requested charge parameters and
does the best job it can to comply
with them while at the same time
independently evaluating various
safety conditions. In order for charge
process to happen, both the battery
and the charger must agree it is safe
to proceed. This keeps the loop simple
and safe.
Flexible Communications
Although the SMBus standard only
allows a single battery and a single
charger to exist on a bus, multiple
bus masters can coexist on the same
bus. This allows the “Smarts” of the
Smart Battery System to be distributed
or augmented.
If you read Smart Battery Specifications, you might come across the
terms called levels such as level 2
and level 3. This has nothing to do
with sophistication or revision levels.
It simply is a way of defining who is
primarily responsible for the communication of charge request between
the battery and the charger. A level 2
system means the Smart Battery is
an SMBus bus master and transmits
its charge commands to the charger
directly. This is the simplest loop. A
level 3 system means any device other
than the Smart Battery itself, such as
a host, is responsible for sending the
commands to the charger. The latter
configuration allows for other devices
to take more control of the process
to implement other unique features
beyond those available by default with
a level 2 system. Linear technology
takes advantage of that capability in
the LTC1760 smart charger by building the bus master into the charger.
This allows for parallel charge and
discharge of dual batteries, which
has numerous advantages, explained
in the “LTC1760 Dual Smart Battery
Charger” section, below.
4
Safety and Reliability
in the SMBus
The SMBus standard incorporates failsafe mechanisms for SMBus crashes
or hangs that allow bus recovery.
The charger has watchdog timers
that monitor the frequency of charge
request commands, and can detect a
loss of communication or closed loop
operation so that it can pause the
charge process and prevent accidental
overcharge. For Li-ion batteries this is
critically important.
allows the charger to make its own
determination, independent of the
gas gauge.
Alarms Warn of
Impending Problems
An active safety feature called alarms is
available to the Smart Battery. Alarms
cover temperature, overcharge and
over-discharge fault conditions, and
are sent when the battery is close to
taking direct action to stop the condition of concern. The gas gauge can send
alarms to both the host and the Smart
Battery Charger via the SMBus.
Battery Detection
via the Safety Signal
Reliable battery detection and additional safety is achieved by use of the
Safety Signal, formerly known as the
thermistor signal. The Safety Signal
is produced by applying a voltage to
a resistor, or thermistor in the Smart
Battery via the dedicated SS pin (see
Figure 1). A Smart Charger can measure the value of the resistance of
the SS pin to ground, and from the
resulting value, know if the battery
is physically present, and if an NTC
thermistor is present, determine if
the temperature range is acceptable
to allow charge.
The thermistor on the Safety Signal
is not the same thermistor the gas
gauge uses to evaluate temperature.
It is a redundant system check that
Over-Discharge Recovery
Over-discharge presents a special
challenge that is fully addressed by the
Smart Battery System. What happens
if the battery does not have enough
charge to support SMBus communications? One cannot attempt to detect
the battery via its terminals, since the
gas gauge will have already turned off
the discharge MOSFET to protect the
battery cells. A Smart Battery Charger,
though, can use the safety signal to
check for the battery pack, regardless
of its state of charge.
A Smart Battery Charger, upon
detection of a new battery, applies a
constant current charge up to 100mA
to the battery terminals. This low current quickly charges the cells enough
LTC1760
LEVEL 3
POWER MANAGER
POWER
DC/DC
CONVERTER
SYSTEM POWER
CONTROL
SMBUS 1 & SAFETY SIGNAL
LOGIC SUPPLY
SWITCH MATRIX
CIRCUITRY FOR
POWERPATH CONTROL
AND SMBUS
3
WALL WART
POWER
SYSTEM
HOST
SMBUS
CHARGE VOLTAGE
AND CURRENT
2
BATTERY 1
BATTERY 2
SMBUS 2 & SAFETY SIGNAL
3
CHARGER
4A MAX
Figure 5. Simplified schematic of a dual battery system using the LTC1760. The LTC1760 acts as
a Level 3 bus master, and handles both charge and discharge of the batteries. Even batteries of
different chemistries or cell configurations can be used.
Linear Technology Magazine • March 2005
Space Saving, Advanced
Smart Battery Chargers
As shown above, the Smart Battery
System offers advanced capabilities
with little required design effort. Linear
Technology’s Smart Battery Chargers
take advantage of Smart Battery System features and add a few of their
own, while maintaining compliance
to the standard.
Linear Technology Magazine • March 2005
3500
3000
2500
2000
1500
1000
500
0
3500
3000
2500
2000
1500
1000
500
0
BAT1
CURRENT
12.0
BAT2
CURRENT
SEQUENTIAL
BAT1
CURRENT
BAT2
CURRENT
DUAL
50
100
150
200
TIME (min)
250
10.0
9.0
BAT2
VOLTAGE
8.0
12.0
300
BATTERY TYPE: 10.8V Li-Ion (MOLTECH NI2020)
REQUESTED CURRENT = 3A
REQUESTED VOLTAGE = 12.3V
MAX CHARGER CURRENT = 4.1A
SEQUENTIAL
11.0
BAT1
VOLTAGE
9.0
1960 G10
DUAL
BAT2
VOLTAGE
10.0
100
MINUTES
0
BAT1
VOLTAGE
11.0
BATTERY VOLTAGE (V)
to where the gas gauge can come on
line and take over the recovery charge
process. As soon as the gas gauge
sends its first valid charge request
commands, the Smart Battery Charger
stops applying the wake up charge and
immediately implements the requested
charge values. A wake up charge is
not applied again until a new battery
physically takes its place.
As foolproof as this recovery process
sounds, there is one more safety issue
to address. Consider the case where
a new fully-charged Li-ion battery
is attached to a Smart Charger, but
permanent SMBus communication
problems prevent the battery from
communicating over the SMBus. The
charger would apply the wake up
charge indefinitely in the absence of
any requests from the gas gauge. This
would lead to a potentially dangerous
overcharge situation.
To cover this situation, the Safety
Signal comes to the rescue again. The
resistance of the safety signal can
fall into several ranges. Each range
defines the acceptable duration of the
wake up charge. Batteries thus fall
into two categories: those that have
the chemistry to receive an indefinite
wake up charge and those that can
only accept a short 3-minute wake
up charge. NiMH batteries fall into the
first category where as Li-ion batteries
fall into the latter category. In the case
of a damaged Li-ion battery, the battery will only receive a short wake up
charge, thus preventing the possibility
of accidental overcharge. The downside
of this important safety feature is that
overly discharged batteries must be
designed to allow SMBus communication to be established within those
three minutes regardless of the state
of the cells.
BATTERY CURRENT (mA)
DESIGN FEATURES
8.0
0
20
40
60
11
MINUTES
80 100 120 140 160 180
TIME (min)
1960 G12
BATTERY TYPE: 10.8V Li-Ion(MOLTECH NI2020)
LOAD CURRENT = 3A
Figure 3. Charging batteries in parallel
is almost twice as fast as charging them
sequentially.
Figure 4. Batteries can run longer when
discharged in parallel than they would when
discharged sequentially.
In an SBS, the charger resides in
the system, sharing precious PCB real
estate with other devices. Linear Technology has two products that directly
address those needs: The LTC4100
single Smart Battery Charger and the
LTC1760 dual Smart Battery Charger
are switching buck regulators that
include features defined in the Smart
Battery standard and other important
performance enhancements.
One of the most important recent
advances in DC/DC converter design
is the use of high capacitance and
voltage (high C/V) ceramic capacitors.
In switching regulator applications,
their low ESR allows them to handle
large ripple current per µF of capacitance compared to most other types
of capacitors, even while remaining
inexpensive to buy.
Ceramics have a problem, though.
They have piezo-electric properties
that can generate audible noise with
the PCB acting as a sounding board.
There are conditions where audiofrequency signals are generated by
typical switching battery chargers. The
LTC4100 and LTC1760 are designed
to operate outside of the audio range
to avoid this problem.
Another challenge is to use smaller
inductors, which usually means a
reduction in inductance value while
still support high currents. Smaller
inductance for a given switching frequency means more ripple current
and the corresponding increase in
capacitor count to handle the higher
ripple current. To keep the ripple current down, the switcher must switch
at higher frequencies than before.
The LTC4100 and LTC1760 operate
at 300kHz, which allows the use of
small inductors.
The LTC4100 Single
Smart Battery Charger
Single battery applications tend to be
systems that are smaller or have lower
power requirements. The LTC4100 is a
Level 2 (slave) Smart Battery Charger
specifically designed to reduce PCB
space. It is compliant with both the
V1.1 of the Smart Battery Charger
and SMBus V1.1 standards. Figure 2
shows a typical application circuit.
The LTC4100 includes a host of
features to improve charge times in a
variety of applications:
❑ It can charge batteries up to 4A
and switch continuously down
to zero load current, so as to not
make audible noise under any
conditions and take full advantage of ceramic capacitors capabilities.
❑ The high 300kHz switching frequency allows the use of small,
common, low cost 10µH inductors and ceramic capacitors for
bulk C filtering.
❑ Input voltage range is 6V to 32V
while output charge voltage range
is from 6.4V to 26V.
❑ Precision charge capabilities are
assured by the 10-bit current
DAC and an 11-bit voltage DAC
5
DESIGN FEATURES
with accuracies of 5% and accuracies of 0.8%, respectively.
❑ A topside P-channel MOSFET allows 98% maximum duty cycles,
dramatically reducing total part
count and IC pin count while
providing efficiency greater than
95%.
❑ SMBus accelerators keep the data
moving along in high capacitance
traces while preventing bus noise
from corrupting data. (More information about SMBus accelerators is available in the LTC1694
datasheet).
❑ A user adjustable AC present
signal with precision 3% accurate
user adjustable trip points.
The LTC4100 also includes important
protection features:
❑ A safety signal circuit that rejects
false thermistor tripping due to
ground bounce caused by the
sudden presence of high charge
currents
❑ A DC input FET DIODE circuit
that prevents battery current
from flowing backwards into the
wall adapter or DC power source
❑ An ultra fast overvoltage comparator circuit prevents voltage
overshoot when the battery is
suddenly removed or disconnects
itself during charge
❑ An input current limit sensing
circuit that is used to reduce
charge current to prevent wall
adapter overload as the system
power increases.
❑ Many unique features, such as a
special current limit and voltage limit system, which prevents
SMBus data corruption errors
from generating false charge values that would potentially harm
the battery.
The LTC1760
Dual Smart Battery Charger
The LTC1760 complies to the Smart
Battery System Manager (SBSM)
specification V1.0.
The LTC1760 has all the same
basic electrical specifications as the
continued on page 35
PowerPath MUX
VIN
C8, 1µF
RCL
0.03
R1
4.99k
R4
12.7k
R5
1.21k
R7
49.9k
C3
0.012µF
C11
1800pF
VDDS
SMBALERT
RPU
SCL
RPU
SDA
VDDS
C1
0.1µF
R10, 100
C1
0.1µF
RVLIMIT
10k
C9, 0.1µF
36
CLP
41
DCIN
3
BAT1
2
BAT2
LTC1760
VPLUS
GDCI
GDCO
GB1I
GB1O
GB2I
GB2O
SCP
16
SCN
DCDIV
37
LOPWR
COMP1
47
CSN
GCH2
48
CSP
SCH2
46
ITH
GCH1
45
ISET
SCH1
13
SW
VSET
40
BOOST
VCC
24
TGATE
VSS
BGATE
PGND
25
TH2A
VCC2
29
TH2B
SMBALERT
18
SCL2
SCL
22
SDA2
SDA
20
TH1A
VDDS
33
TH1B
VLIMIT
32
SCL1
ILIMIT
26
SDA1
MODE
BAT2
D2
R11
1k
D1: MBR130T3
D2: IN4148 TYPE
Q1, Q2, Q5, Q6, Q7, Q8: Si4925DY
Q3, Q4, Q9, Q10, QTG, QBG: FDS6912A
Q6
Q7
Q2
Q5
Q8
RSC
0.02
LOAD
R2
280k
C7
0.1µF
R9
3.3k C12
1000pF
C5
0.15µF
R1B, 54.9k
QTG
CB1, 0.1µF
C6
4.7µF
R6
100
R2A, 1.13k SAFETY 2
BAT1
TH
SCL
SDA
CIN
20µF
L1
10µH
RSENSE
0.025
C4, 0.22µF
QBG
BAT2
TH
SCL
SDA
R1A, 1.13k SAFETY 1
D4
C13
0.1µF
CL
20µF
R3
49.9k
R2B, 54.9k
BAT1
D3
I5
I6
CB2
0.47µF
1
7
6
9
8
11
10
5
4
12
34
35
14
15
42
43
44
39
38
28
27
17
21
30
31
19
23
Q1
D1
COUT
20µF
CHARGE
MUX
Q4
Si6928
Q9
Si6928
Q3
Si6928
Q10
Si6928
Figure 6. Dual battery charger/controller safely charges and discharges two Smart Batteries in parallel, even two batteries with different
chemistries. Parallel charge and discharge is far more efficient and faster than serial charging, and some high power applications require parallel
discharge of batteries to supply higher currents than a single battery can offer.
6
Linear Technology Magazine • March 2005
DESIGN FEATURES
Dual Monolithic Ideal Diodes
Provide a Single-Chip Power
Management Solution
Introduction
The LTC4413 dual monolithic ideal
diode helps reduce the size and improve the performance of handheld
and battery operated devices. It packs
so many features into a tiny package
that it is possible to build an entire
power management solution in a
3mm × 3mm footprint. Figure 1 shows
how simple it is to build a complete
battery-wall-adapter PowerPath™
manager.
Despite its compact size, the
LTC4413 includes features that are
necessary in demanding applications,
including thermal management, short
circuit protection, and system-level
power management and control.
Two isolated p-channel MOSFET
transistors serve as low voltage (2.5V
to 5.5V) monolithic ideal diodes. Each
ideal diode channel provides a low forward voltage drop (typically as low as
40mV when conducting 10mA) and a
low RDS(ON) (below 100mΩ)—important
in battery-powered applications.
Furthermore, each channel is
capable of providing 2.6A of continuous current from a small 10-pin DFN
package. If the load attempts to draw
more than 2.6A, the internal current
limit threshold is reached. At this point
the LTC4413 fixes the output current
at the over-current maximum. This
causes the output voltage to collapse
and the power dissipation within the
chip to increase. Current limit protects the internal p-channel MOSFET
diodes against shorts and overloads.
Sustained overloads that result in
excessive die heating are mitigated by
thermal shutdown.
System-level power management
and control are available through a
status signal pin to indicate conduction status, and two active-high disable
input pins, which independently
control the operation of each of the
PowerPath ideal diodes.
Linear Technology Magazine • March 2005
by Andy Bishop
How it Works
The low forward voltage drop,
low RDS(ON), and low reverse leakage
current of the LTC4413 offer several
additional benefits. The tiny forward
voltage drop directly results in extended battery life. The low RDS(ON)
reduces power dissipation, further
enhancing battery performance. The
very low reverse leakage current, when
compared with a Schottky diode, is
“The LTC4413 packs so
many features into a tiny
package that it is possible
to build an entire power
management solution in a
3mm × 3mm footprint.”
also beneficial in many applications
particularly where leakage current
into a battery from a reverse biased
Schottky diode could cause damage
or failure.
The LTC4413 can be used as a
replacement for two LTC4411 monolithic ideal diodes, or it can be used
in applications that may have used
one LTC4411 along with a Schottky
diode, thereby providing an improvement in terms of space and power
consumption.
Figure 1 shows an application where
the LTC4413 is configured as an
automatic power switch between a
battery and a wall adapter (or other
auxiliary power source) to supply
continuous power to the load attached
to the output.
The operation of this circuit is
shown in Figure 2, where the inputs
are ramped slowly to illustrate how
the LTC4413 functions.
First the battery input at INA is
ramped up from 0V while the auxiliary input at INB is left floating (A0).
Once the battery voltage exceeds the
under voltage lock-out (UVLO) rising
threshold of 2.2V (A1), the LTC4413
begins to conduct in forward regulation
mode, pulling the output voltage up
to within 20mV of the battery voltage
(the voltage drop across the LTC4413
depends on the load current). As the
battery voltage continues to increase
(time interval A1–A2) up to 3.5V, the
output voltage follows the battery voltage minus the small forward voltage
drop across the LTC4413. During the
forward regulation mode of operation
(from time A2 to B0), the STAT pin is
an open circuit and the 560kΩ resistor pulls the STAT pin voltage up to
VCC, indicating that the load current
is supplied by the battery connected
to INA. Alternatively, this resistor
VCC
ENBA
GND
WALL
ADAPTER
(VBATT TO 5.5V)
STAT
INB
OUTB
CIN
10µF
STAT IS HIGH WHEN
BAT IS SUPPLYING
LOAD CURRENT
IDEAL
INA
BAT
560k
LTC4413
ENBB
IDEAL
OUTA
TO LOAD
COUT
4.7µF
Figure 1. Automatic power switch between a battery and a wall adapter
7
DESIGN FEATURES
A0
A1
A2
B0 B1
B2
B3
C0
C1
Automatic Dual
Battery Load Sharing
D0
3.5V
3.48V
2.2V (UVLO RISING
THRESHOLD)
1.9V (UVLO FALLING
THRESHOLD)
BATTERY
CELL
A dual battery load sharing circuit is
shown in Figure 3. In this schematic
an LTC4413 is used to isolate two batteries, perhaps a main and a backup
battery, from the load. This circuit
takes advantage of the fact that it is
more efficient to discharge the batteries in parallel than it is to discharge
them sequentially.
Whichever battery has the higher
voltage provides the load current until
it has discharged to the voltage of the
other battery. The load is then shared
between the two batteries according to
the capacity of each battery. The higher
capacity battery provides proportionally higher current to the load.
As the LTC4413 only allows current
to flow in one direction, each battery
is isolated from the other so that no
reverse current can flow from one
battery into the other. This eliminates
the possibility of a potentially hazardous situation where one battery may
uncontrollably discharge curent into
the other. The STAT pin may be used
to indicate whether the backup battery
attached to INA is conducting, thus
providing an automatic monitor to
indicate when the backup battery is
supplying all of the load current.
0V
5.5V
3.48V – VRTO
AUXILIARY
SUPPLY
0V
5.48V
3.48V
2.2V (UVLO RISING
THRESHOLD)
1.9V (UVLO FALLING
THRESHOLD)
OUTPUT
TO LOAD
0V
VCC
STAT
0V
Figure 2. Operation waveforms of the LTC4413.
may be tied to the output as shown
in Figures 6, 7 and 8.
Consider next a wall adapter, or
other auxiliary supply voltage, applied
to pin INB (at time B0). The voltage at
INB is then ramped upwards from 0V
(starting at time B1). The LTC4413
automatically senses when the voltage
at INB is greater than the voltage at
the output (at time B2) and reverts to
supplying load current from the input
applied at pin INB; disconnecting the
battery from the load as the voltage
at the output rises above the battery
voltage at INA. At this point, the STAT
pin begins to sink 9µA causing the
STAT pin voltage to fall, indicating
that the wall adapter at INB is now
supplying load current. As the auxiliary voltage continues to rise to 5.5V
(B3) the output voltage follows the
auxiliary voltage.
When the wall adapter, or auxiliary
voltage, is removed (at time C0) and
the voltage at INB drops to zero, the
output voltage begins to ramp down as
COUT discharges; at a rate depending
on the load current. Once the output
8
voltage drops below the battery voltage
(C1) the LTC4413 reverts to supplying
load current from the battery. At this
time the STAT pin becomes an open
circuit, and the 560k resistor pulls
the STAT pin voltage to VCC to indicate
that the battery is now supplying load
current.
As the battery voltage continues
to discharge below the under voltage
lock-out threshold of 1.9V (at time D0),
the LTC4413 turns itself off, and the
battery is disconnected from the load.
The output voltage then collapses as
the load discharges capacitor COUT.
Multiple Battery Charging
Figure 4 illustrates an application of
multiple battery charging using the
LTC4413. In this example, one or both
of the batteries can be charged from
a single battery charger (not shown),
regardless of the state of charge of
the other battery. This circuit takes
advantage of the fact that charging
batteries in parallel is more efficient
than charging them sequentially.
VCC
2
4
3,11
BACKUP
BATTERY
MAIN
BATTERY
560k
ENBA
ENBB
STAT
9
STAT IS HIGH WHEN
BACKUP BATTERY IS
SUPPLYING ALL OF THE
LOAD CURRENT
GND
LTC4413
IDEAL
OUTA 10
1 INA
5 INB
TO
LOAD
IDEAL
OUTB 6
COUT
4.7F
Figure 3. Automatic dual battery load sharing with secondary battery monitor
Linear Technology Magazine • March 2005
DESIGN FEATURES
VCC
BATTERY
CHARGER
INPUT
LTC4413
9
STAT
IDEAL
OUTA 10
1 INA
4
3,11
ENBA
4
ENBB
3,11
VIN
LOAD2
BAT2
ENBA
560k
ENBA
ENBB
STAT
9
GND
LTC4413
IDEAL
OUTA 10
1 INA
LOAD1
BAT1
IDEAL
5 INB
OUTB 6
2
560k
STAT IS HIGH
WHEN BAT1
IS CHARGING
2
CIN
10F
COUT
4.7µF
IDEAL
5 INB
OUTB 6
ENBB
COUT
4.7µF
GND
When the ENBA and/or ENBB pins
are at a logical high, the LTC4413
turns off the corresponding diode
and removes power to that load. If the
load at OUTA is powered from another
(higher voltage) source, the supply connected to INA remains disconnected
from that load; the load connected to
OUTB may remain connected to INB
independent of the voltage at OUTA
and vice versa.
The STAT pin can be used to indicate
the conduction STATUS of diode A (if
either ENBA is low, or both enable pins
are low). Alternatively, the STAT pin
can be used to indicate if diode B is
conducting (if ENBA is at logic HIGH
and ENBB is at logic LOW). If both
ENBA and ENBB are logic HIGH, the
STAT pin is logic LOW.
Whichever battery has the lowest
voltage receives the full charging current until both battery voltages are the
same. Then both batteries are charged
simultaneously. One advantage
charging multiple batteries in parallel—rather than sequentially—with
the LTC4413 is that both batteries
are always charged up to the same
relative percentage of the cell capacity.
So, if the battery charger is suddenly
removed in the middle of charging,
both batteries are partially charged
to the same percentage charge. The
enable pins and STAT pin can be used
to independently control which of the
batteries is charged and monitor if the
enabled battery is charging.
Dual High Side Power Switch
Automatic Switchover
from a 4.2V Li-Ion Battery
to a Wall Adapter
and a Battery Charger
Figure 5 illustrates the LTC4413
in use as a dual high side power
switch.
When the ENBA pin is a logical
low, the LTC4413 turns on ideal
diode A, supplying current from INA
to the load attached to OUTA. When
the ENBB pin is a logical low, the
LTC4413 turns on ideal diode B,
supplying current from INB to the
load attached to OUTB.
Figure 6 illustrates an application
where the LTC4413 performs the
function of automatically switching
a load over from a battery to a wall
adapter, while controlling a LTC4059
battery charger.
Dual Battery Load Share
with Automatic Switchover
to a Wall Adapter
Figure 7 illustrates how to use the
LTC4413 to implement a circuit that
automatically switches over from
a dual battery load share to a wall
adapter. As described earlier, with
Figure 3, the LTC4413 performs a load
sharing function for BATA and BATB,
MP1 FDR8508
5V
WALL
ADAPTER
LTC4413
9
STAT
IDEAL
OUTA 10
1 INA
LTC4059
BAT
100k
1-CELL
Li-Ion
2
4
3,11
10µF
10F
100k
560k
ENBA
ENBB
GND
IDEAL
5 INB
OUTB 6
Li CC GND
WALL
ADAPTER
BATA
1-CELL Li-Ion
TO LOAD
BATB
1-CELL Li-Ion
2
4
3,11
ENBA
ENBB
STAT
9
GND
LTC4413
IDEAL
OUTA 10
1 INA
560k
TO
LOAD
IDEAL
5 INB
OUTB 6
4.7F
4.7F
Figure 6. Automatic switchover from a 4.2V Li-Ion battery to a wall adapter and battery charger
Linear Technology Magazine • March 2005
OUTB
When no wall adapter is present,
the LTC4413 powers the load from the
Li-Ion battery at INA, and the STAT
voltage is high, thereby disabling the
battery charger.
If a wall adapter voltage higher than
the battery voltage is connected to INB,
the LTC4413 automatically powers the
load from the wall adapter. When this
occurs, the STAT voltage falls, turning
on the LTC4059 battery charger and
beginning a charge cycle.
If the wall adapter is removed, the
voltage at INB collapses until it is
below the battery voltage. When this
occurs, the LTC4413 automatically
re-connects the battery to the load
and the STAT voltage rises, disabling
the LTC4059 battery charger.
665k
ENB PROG
OUTA
Figure 5. Dual high side power switch
Figure 4. Multiple battery charging
VCC
STAT IS HIGH WHEN
ENBA IS LOW AND
VIN > VOUTA
OR
ENBA IS HIGH AND
ENBB IS LOW AND
VIN > VOUTA
Figure 7. Dual battery load share with
automatic switchover to a wall adapter
9
DESIGN FEATURES
with the addition of an automatic
switchover whenever a wall adapter
is applied.
When the wall adapter is connected, both ENBA and ENBB voltages
are pulled higher than the turn-off
thresholds of 550mV through a user
programmable resistive divider.
When this occurs, the STAT voltage
falls, turning on MP1 so that the wall
adapter can provide load current. If
the wall adapter is disconnected, the
output voltage droops until the ENBA
and ENBB voltages fall through their
turn-on threshold of 450mV; enabling
both ideal diodes. The LTC4413 then
connects the higher of BATA or BATB to
the load. If the voltage at BATA is highest, the STAT voltage rises, otherwise
the STAT voltage remains low.
Automatic Switchover from
a Battery to an Auxiliary
Supply or to a Wall Adapter
Figure 8 shows automatic switchover
from a battery to either an auxiliary
supply or to a wall adapter using the
LTC4413. This simple circuit handles
all combinations of applied power
automatically.
Consider two scenarios. In the first,
the auxiliary supply is not present and
the battery provides load current when
the wall adapter is attached. In the
second, the auxiliary is present when
the wall adapter is attached.
In the first case (aux supply absent),
when the wall adapter is applied, the
diode in the external PFET (MP1) forward biases pulling the output voltage
above the BAT voltage and turning off
the ideal diode connected between
BAT and the output. This causes the
STAT voltage to fall, turning on MP1
and connecting the wall adapter to the
load. The load current is then provided
by the wall adapter and the battery is
disconnected from the load.
When the wall adapter is removed,
the output voltage falls until the BAT
voltage exceeds the output voltage.
When this event occurs the STAT
voltage rises, turning off the external
MP1 FDR8508
5V
WALL
ADAPTER
C1
10µF
R2
665k 4
R3
100k
BAT
2.5V–5.5V
AUX
ADAPTER
ENBB
STAT
LTC4413
9
IDEAL
OUTA 10
1 INA
560k
3,11
GND
IDEAL
5 INB
OUTB 6
R4
1000k 2
R5
470k
ENBA
C2
4.7µF
TO
LOAD
Figure 8. Automatic switchover from a battery to an auxiliary supply or a wall adapter
PFET, and the ideal diode between BAT
and the output automatically turns on
to provide power to the load.
In the second case (aux supply
present), the voltage divider (R5 and
R4) pull ENBA higher than its turn-off
threshold, disconnecting the battery
from the load, and the auxiliary supply
provides the load current.
When the wall adapter is applied,
the LTC4413 senses the presence of
the wall adapter as the ENBB pin voltage is pulled higher than its turn-off
threshold; through resistive divider
(R2 and R3). When this occurs, the
auxiliary is disconnected from the load
and the STAT voltage falls, turning
on MP1 so that the wall adapter can
provide the load current. When the wall
adapter is removed, ENBB falls until
the auxiliary is enabled and reverts to
providing power to the load.
If the auxiliary is removed while
the wall adapter is providing load current, the ENBA voltage falls, enabling
the ideal diode between BAT and the
output. However, if the wall adapter
voltage is higher than the BAT voltage,
the ideal diode between BAT and the
output is reverse biased and no current flows into the battery from the
wall adapter (through the LTC4413).
When the wall adapter is removed,
the output voltage falls until the BAT
voltage exceeds the the output voltage.
At this point, the ideal diode between
BAT and the output turns-on and the
STAT voltage rises, disabling MP1.
When the wall adapter is disconnected while the auxiliary supply is
present, the load voltage droops to just
below the auxiliary voltage at which
point the auxiliary supply begins to
source the load current. At this point
the STAT voltage rises; disabling
MP1. This causes the capacitor C1
to discharge until the ENBA turn-on
threshold is reached; this allows the
battery to source load current if the
output voltage drops below the battery voltage.
If the wall adapter is disconnected
when the auxiliary supply is not present, the load voltage drops until the
voltage at the ENBA pin (formed by
resistive divider R2 and R3) falls below
the turn-on threshold of 450mV. When
this occurs, the battery is connected
to the load and the STAT voltage is
pulled high, disabling MP1.
Conclusion
The LTC4413 dual monolithic ideal diode provides a simple and
efficient single-IC solution for low-loss
PowerPath management. This device
is ideal for battery-powered portable
devices. It extends battery life, significantly reduce self-heating, and
reduces form-factor with its 10-lead
3mm × 3mm footprint and minimal
external parts count.
For more information on parts featured in this issue, see
http://www.linear.com/designtools
10
Linear Technology Magazine • March 2005
DESIGN FEATURES
Simplify PoE Implementation
with Complete PD Interface and
Integrated Switching Regulator
Introduction
IEEE 802.3af Power over Ethernet
(PoE) is a standard for delivering power
over Cat-5 cables, eliminating the
need for AC-adapters for equipment
plugged into the Ethernet. The two
major components in a PoE system
are Power Sourcing Equipment (PSE),
which deliver power, and Powered Devices (PD), which receive and use the
power. A PSE will not deliver power to
the load unless it detects a valid signature resistance, which distinguishes a
compliant PD from a device that cannot
receive power. Once the PD receives
power, it must also convert the –48V
PoE efficiently to a suitable power
supply voltage. Typical PD designs
employ two ICs for these tasks. An
obvious way to simplify PD designs
would be to integrate the interface
and DC-DC conversion circuitry into
a single device.
VPORTP
1.237V
+
–
RCLASS
The LTC4267 reduces the complexity and size of a PD by combining an
IEEE 802.3af-compliant PD interface
with a current mode switching regulator. Figure 1 shows a block diagram of
the device. The LTC4267 includes the
25kΩ signature resistor, classification
current source, thermal overload protection, signature disable and a power
good signal, along with an under-voltage lockout (UVLO) optimized for use
with the IEEE-required diode bridge.
The precision dual-level current limit
allows the LTC4267 to charge large
load capacitors and interface with
legacy PoE systems. The current-mode
switching regulator is designed for
driving a 6V rated N-channel MOSFET
and features programmable slope
compensation, soft start, and constant
frequency operation, minimizing noise
even with light loads. The LTC4267
includes an onboard error amplifier
PD Implementation
Made Simple
Figure 2 presents a complete PD
detection and power conversion application—a testimony as to how simple
a PD implementation can be. The
LTC4267’s package size is the smallest in the industry, and many of the
circuits that are traditionally implemented with external components
have been folded into this device.
During detection, the Power Sourcing Equipment (PSE) identvifies the
presence of an IEEE 802.3af-compliant PD by applying two voltages,
measuring the corresponding current,
then performing a ΔV/ΔI calculation.
PVCC
SIGDISA
0.3µA 0.28V
25k
SIGNATURE
RESISTOR
9k
800mV
REFERENCE
VCC
SHUNT
REGULATOR
PWRGD
+
EN
SHUTDOWN
COMPARATOR
–
PVCC <
VTURNON UNDERVOLTAGE
LOCKOUT
VFB
INPUT
CURRENT
LIMIT
SHUTDOWN
SOFTSTART
CLAMP
CONTROL
CIRCUITS
375mA
+
16k
POWER GOOD
+
and voltage reference allowing its
use in both isolated and non-isolated
configurations. All this functionality is
packed into a space-saving, low-profile
16-pin SSOP or DFN package.
CLASSIFICATION
CURRENT LOAD
EN
by Kirk Su
ERROR
AMPLIFIER
–
+
CURRENT
COMPARATOR
R
S
–
PVCC
Q
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
NGATE
GATE
DRIVER
ITH/RUN
140mA
20mV
–
1.2V
VPORTN
200kHz
OSCILLATOR
SLOPE
COMP
CURRENT
RAMP
SENSE
POUT
PGND
BOLD LINE INDICATES HIGH CURRENT PATH
Figure 1. LTC4267 block diagram
Linear Technology Magazine • March 2005
11
DESIGN FEATURES
PA1133
SBM1040
3.3V
1.5A
10k
–48V
FROM
DATA PAIR
+
HD01
VPORTP
–
SMAJ58A
PWRGD
LTC4267
NGATE
0.1µF
–48V
FROM
SPARE PAIR
+
SENSE
RCLASS
68.1Ω
1%
HD01
–
PVCC
RCLASS
ITH/RUN
+
PVCC
4.7µF
PGND
VPORTN
POUT
CLOAD
5µF
MIN
•
•
320µF
MIN
CHASSIS
Si3440
10k
PVCC
0.1Ω
470
6.8k
VFB
SIGDISA
+
22nF
BA5516
100k
PS2911
LT4430
60.4k
Figure 2. A Class 2 PD with 3.3V isolated power supply
A 25kΩ signature resistance signals
the PSE that a valid PD is present and
is ready to receive power. Most PD detection solutions require one or more
external resistors to present a valid
signature during detection. One of the
benefits of the LTC4267 is an internal, temperature-compensated 25kΩ
signature resistor, which is precision
trimmed to account for the series resistance of the input diode bridges and
all parallel leakage paths. This ensures
proper PD detection without the need
to size external components.
Another unique feature is the Signature Disable function. When the
SIGDISA pin is exercised, the LTC4267
presents a 9kΩ resistance that signals
the PSE not to power the PD. This
feature simplifies the interface with
an external power source such as a
wall transformer.
Once the PSE detects the LTC4267,
the PSE may classify the PD by detecting a range of load currents. The
LTC4267 offers a simple scheme for
PD classification: the PD designer
programs the classification load current using a single RCLASS resistor or
leaves the RCLASS pin open for class
0. During classification, the LTC4267
asserts a precision load current from
the VPORTP pin through the RCLASS
resistor to notify the PSE of the PD
power requirements.
The IEEE 802.3af specifies the
classification voltage range to be
between –15.5V to –20.5V. However,
12
the LTC4267 is designed to remain in
classification mode from –15.5V to the
UVLO turn-on threshold of –36V nominal. IEEE 802.3af does not require
this extended classification range, but
the added range aids in the PSE-PD
power-up stability by maintaining a
monotonically increasing V-I characteristic up to the turn-on voltage.
The LTC4267 provides a complete
and self-contained dual-current
protection without the need for any
external components. The LTC4267’s
unique current limit method ensures
PD inter-operability with new and
legacy PSEs, and unlike competing
products, the LTC4267 does not depend on the PSE to monitor current
limit. As the LTC4267 reaches the
UVLO turn-on voltage and prepares to
delivers power, the LTC4267 limits the
inrush current to 140mA nominal until
the load capacitor (CLOAD in Figure 2)
charges up to within 1.5V of the final
value. Once this voltage threshold is
reached, the current-limit threshold
switches over to 375mA nominal for
the remainder of the PD operation.
The dual level current limit allows
legacy PSEs with limited current
sourcing capability to power up the
PD while allowing the PD to maximize
the power utilization from an IEEE
802.3af-compliant PSE. By maintaining a 375mA current limit once the
PD is powered up, the LTC4267 keeps
the PD from depending on the PSE
for current limit and avoids possible
inter-operability problems. These current-limiting features are controlled by
an onboard 100V, 400mA N-channel
Power MOSFET and an internal sense
resistor.
The LTC4267 presents a Power
Good signal once the load capacitor
(CLOAD in Figure 2) is charged to within
1.5V of the final value. The Power
Good signal may be used to signal to
the switching regulator that the PD
interface has charged up the load capacitor and is ready to apply power. A
3V hysteresis is included in the power
good circuit, allowing the LTC4267 to
operate near the current limit point
without inadvertently presenting an
invalid Power Good.
The thermal shutdown circuit monitors the die temperature, serving as an
additional means of self-protecting the
LTC4267 and other electronic circuitry
from over-current or over-heating
conditions. If such an event occurs
in either the classification sequence
(from the PSE probing exceeding the
IEEE-mandated 75ms maximum) or in
normal PD operations (from multiple
turn on events), the thermal shutdown circuit protects the LTC4267
by disconnecting power to the output
load and disabling the classification
current until the die returns to a safe
operating temperature.
Powering the LTC4267 switching
regulator in the simplest case can
be achieved with a dropping resistor
between VPORTP and PVCC. An internal
Linear Technology Magazine • March 2005
DESIGN FEATURES
shunt regulator maintains the supply at 9.4V, providing the required
voltage needed for the gate drive. The
LTC4267 can also be powered with a
pre-regulator and/or with a separate
bias winding on a flyback transformer.
Each of these methods offer improved
efficiency. The shunt regulator is
particularly important when using
a flyback methodology since it also
serves the vital function of protecting
the LTC4267 PVCC pin from seeing too
much voltage.
The LTC4267 switching regulator
features two ways to enable operation
during the initial power-up sequence.
The PVCC pin includes an UVLO circuit
with hysteresis, and the ITH/RUN pin
serves as an enable as well as the compensation point for the internal error
amplifier. Once the interface circuit
charges the load capacitor, enable the
switching regulator operation with the
ITH/RUN and PVCC pin. Note that both
pins must be enabled for operation to
begin. The switching regulator may be
disabled by either pulling the PVCC pin
below the UVLO turn-off threshold or
by pulling the ITH/RUN pin below the
0.28V nominal threshold.
Implementing a robust power-up
sequence between the PD Interface
circuit and the switching regulator
is critical in a successful PD application. The power good signal indicates
that the load capacitor is charged
and this signal can be used to enable
the switching regulator. An N-channel transistor driven by PWRGD can
be used to disable the switcher by
clamping the PVCC or ITH/RUN pins.
Disabling the switching regulator until
the load capacitor is charged up can
also be accomplished with an RC delay
on the PVCC pin as shown in Figure 2.
The flexibility of the LTC4267 provides
the PD designer with the freedom of
implementing a controlled power-up
sequence in a variety of ways.
The LTC4267 features a soft-start
feature that provides an additional
1.4ms delay once the ITH/RUN pin is released. The soft-start feature reduces
the switching regulator inrush current
and reduces output overshoots. Unlike
competing products, there is no minimum external capacitance required to
program this delay. The designer may
opt to provide additional soft-start delay by employing an external capacitor
between ITH/RUN and PGND pins.
An internal error amplifier with a
precision voltage reference is integrated into the LTC4267. This feature is
particularly desirable in non-isolated
power supply applications since the
PD designer does not need to add an
external amplifier or reference. The
internal precision reference provides
output voltage accuracy to within
±1.5% over the 0oC to 70oC temperature range. For isolated power supply
applications, the internal error amplifier and reference can be disabled by
connecting the VFB pin to PGND, and
connecting an external error amplifier and opto-isolator to the ITH/RUN
pin. Figure 2 shows an example of
an isolated power supply using an
external amplifier and Figure 3 shows
a non-isolated supply that uses the
LTC4267’s internal error amplifier.
Slope compensation is critical for
stabilizing the control loop against
sub-harmonic oscillations and is available on the LTC4267 by including an
optional resistor between the sense
resistor and the SENSE pin of the
LTC4267. The SENSE pin monitors
the voltage across the sense resistor. It also sources 5µA through the
slope compensation resistor, raising
the SENSE pin voltage above the
sense resistor voltage. This in turn
amends the duty cycle of the switching
regulator, preventing sub-harmonic
oscillation.
Conclusion
The LTC4267 is a self-contained Power
over Ethernet PD interface that combines 802.3af PD classification with
a switching regulator. It integrates
many features that are traditionally
implemented with separate components, but it retains the flexibility
that external components offer. The
result is a compact, easy-to-use, but
versatile device.
COILTRONICS
CTX-02-15242
5µF*
MIN
–48V
FROM
DATA PAIR
BAS516
VPORTP
–
+
HD01
–
9.1V
PVCC
300µF**
•
1µF
LTC4267
NGATE
0.1µF
UPS840
•
MMBTA42
+
HD01
SMAJ58A
–48V
FROM
SPARE PAIR
220k
100k
5V
1.8A
FDC2512
150pF
200V
PWRGD
RCLASS
45.3Ω
1%
10k
VFB
SIGDISA
ITH/RUN
22nF
VPORTN
*1µF CERAMIC + 4.7µF TANTALUM
** THREE 100µF CERAMICS
POUT
220Ω
SENSE
PGND
27k
0.04Ω
1%
42.2k
1%
8.06k
1%
Figure 3. A Class 3 PD with 5V non-isolated power supply
Linear Technology Magazine • March 2005
13
DESIGN FEATURES
High Accuracy Clock up to
by Albert Huntington
170MHz in a SOT-23
Introduction
Crystal based oscillators are often
the default choice for designers
looking to clock today’s high speed
microcontrollers, data converters and
programmable logic devices. Crystal
oscillators, although convenient, accurate and stable, come at a high price in
use—they occupy considerable board
space, consume significant power, and
are sensitive to environmental factors
like shock and temperature extremes.
The LTC6905 is an all silicon clock
that avoids these pitfalls, making it
an alternative to crystal oscillators in
applications up to 170MHz.
Accuracy and jitter specifications of
the LTC6905 are more than sufficient
for most applications, and its power
and size advantages allow the LTC6905
to fit in designs where a crystal oscillator could never go.
Device Description
The LTC6905 is a part of Linear
Technology’s line of resistor controlled
SOT-23 oscillators. These resistor
controlled oscillators use a single
external resistor to accurately set
the oscillator frequency, and there is
a simple linear relationship between
the resistor value and the frequency
(see Figure 1). The LTC6905 is pincompatible with the LTC1799 SOT-23
oscillator, but uses a different control
resistor range and a different formula
to set the frequency.
The LTC6905 is also available in
fixed frequency versions, where the
resistor is internal to the part and
no external components other than a
bypass capacitor are required. Preset
devices with master oscillator frequencies of 133MHz, 100MHz, 96MHz and
80 MHz and 1.5% accuracy are available. These devices have an internal
divider which makes it possible to
produce most popular frequencies
between 20MHz and 133MHz. Devices
can be customized to output any
14
17.225MHz ≤ fOSC ≤ 170MHz
5V
10k ≤ RSET ≤ 25k
1
0.1µF
2
3
+
V
OUT
LTC6905
5
5V
GND
SET
DIV
4
÷1
÷2
OPEN
÷4
Figure 1. A single resistor sets the frequency of this tiny, robust oscillator.
frequency in the range of 2.2MHz to
170Mhz.
The LTC6905 uses an internal
feedback loop to accurately match
the impedance of a switched capacitor
element to the external resistor connected to the RSET pin, thus setting
the master oscillator frequency. The
voltage level on the DIV pin engages
internal dividers to divide this master
frequency by 1, 2 or 4 before it is sent
to the OUT pin. With fixed frequency
devices, the LTC6905-XXX series of
parts, the RSET pin is replaced by an
output enable pin, which disables the
output when it is connected to GND.
The voltage on the RSET pin of
the LTC6905 is forced to a bandgap
controlled voltage of 1V below the
positive supply, independent of the
temperature or supply voltage, with a
tolerance of less than 5%. This stable
RSET voltage makes the LTC6905 ideal
for applications where an accurate
voltage or current controlled frequency
is required.
The frequency range of the master
oscillator in the LTC6905 is limited to
between 70MHz and 170MHz, which
corresponds to external frequency
setting resistor values between 10kΩ
and 25kΩ. This range is expanded
by the internal dividers to between
17MHz and 170MHz, and is limited
by the architecture of the high speed
master oscillator.
The master oscillator of the
LTC6905 is a voltage controlled ring
oscillator, and provides a unique
jitter profile where the jitter percentage remains relatively constant over
frequency. Traditional relaxation oscillators develop a larger percentage jitter
as the frequency increases. The jitter
of the LTC6905 actually decreases
with increasing operating frequency,
making it ideal for high frequency
applications.
Fixed Frequency Devices
The LTC6905 can be ordered in a
fixed frequency version where the
frequency-setting resistor is inside the
part. An output enable pin is made
available in place of the RSET pin on
these devices only. Four versions are
available: LTC6905-133, LTC6905100, LTC6905-96 and LTC6905-80.
These four versions collectively offer 12
popular frequencies through the use of
their DIV pins. Please see Table 1.
The LTC6905-XXX fixed frequency
oscillators offer several advantages
that stem from their internal resistor configuration. The parts are less
sensitive to external noise that may
couple into the RSET pin on the external
resistor version of the part. This lack
of sensitivity translates into improved
jitter of less than 1% at all frequencies
and accuracy of better than 1.5% over
commercial temperature range. The internal resistor parts are generally more
accurate because they are trimmed at
one specific frequency and do not have
any error term from nonlinearities over
the RSET resistor range.
The absence of an RSET pin on the
fixed-frequency devices has made
room for an output enable pin. This
output enable synchronously disables
the output drivers when brought low,
Linear Technology Magazine • March 2005
DESIGN FEATURES
V+
R
C
1
0.1µF
LTC6905
2
3
Figure 2. LTC6905 Suggested layout.
Note that the bypass capacitor is located
adjacent to the device and on the same
side of the PC board.
and does not produce pulse slivers.
Power dissipation is significantly
reduced because much of the power
is dedicated to driving output capacitance. The internal master oscillator
and bias networks remain active in
order to facilitate an immediate and
accurate frequency output when the
output is enabled. If the output enable pin is left floating or pulled to
the positive supply, the oscillator is
enabled.
Layout Considerations
Because the LTC6905 combines a high
frequency oscillator and output stage
with a sensitive analog control loop,
it is necessary to exercise great care
in board layout to maximize accuracy
and stability. The bypass capacitor
must be placed as close as possible to
the LTC6905, preferably on the same
side of the board. Even the small
inductance and resistance of vias in
the pc board can adversely effect part
performance. Additionally, the traces
to the bypass capacitor should be
larger than is indicated by the power
consumption of the device. Although
the average power consumption is low,
driving a capacitively loaded output
will induce spikes in the supply current which must be damped by the
bypass capacitor.
The RSET pin is the most sensitive
input pin, and attempts must be
made to shield it from noise coupling
or excessive parasitic capacitance. It
is recommended that the frequency
setting resistor be located as close as
possible to the RSET pin, and that the
frequency setting resistor be connected
to the positive supply as close as possible to the V+ pin. A recommended
layout is illustrated in Figure 2. If the
bypass capacitor must be situated on
the opposite side of the PC board from
Linear Technology Magazine • March 2005
+
V
OUT
LTC6905
CLOAD
GND
SET
V+
100Ω
5
0.1µF
V+
DIV
1
3
4
Figure 3. A series resistor on the LTC6905
output pin reduces power supply spikes
caused by load capacitance.
5 950Ω
GND
SET
50Ω CABLE
50Ω
V+
DIV
4
Figure 4. The LTC6905 can drive a 50Ω
cable with appropriate termination.
the LTC6905, it is strongly recommended that the connection between
the capacitor and the LTC6905 be as
short as possible and use multiple,
filled vias to minimize series inductance and resistance.
The LTC6905 is specified at an
output load of 5pF, which is equivalent to about two standard HC logic
inputs. Driving this load at 170MHz is
the single largest factor in the power
consumption of the LTC6905. The
power supply current needed to drive
a capacitive load may be calculated
as:
ISUPPLY = CLOAD • VSWING • FOSC
where CLOAD is the 5pF load capacitance, VSWING is the voltage swing, in
this case up to 5.5V, and FOSC is the
frequency of the oscillator. Driving a
5.5V swing into a 5pF load at 170MHz
takes 4.675mA on average.
The majority of this power is
expended during the risetime and
falltime of the output signal, not while
it is in a steady state. The 500ps rise
and fall times of the LTC6905 mean
that the instantaneous power supply
current required during the rise and
fall portions of the waveform is much
greater than the average. The instantaneous power supply current may be
calculated by a similar formula:
IPEAK = CLOAD × VSWING ×
2
V+
OUT
LTC6905
1
trf
where trf is the rise/fall time of the
signal. In this case, 55mA spikes are
generated by driving 5.5V into a 5pF
load.
Because of these power supply
spikes, and because of the tendency
for fast edges to couple into adjacent
lines, the layout of the output trace
is critical. Capacitance, trace length
and loading should be minimized. Additionally, with traces longer than a few
centimeters, transmission line effects
must be taken into consideration.
Should output loading and coupling
problems occur, there are methods to
mitigate the effects. A series resistance
in the range of 50Ω–1000Ω placed
adjacent to the output pin of the
device will increase the rise and fall
times of the signal being driven into
the output load, and therefore reduce
power supply spikes and coupling (see
Figure 3). A 50Ω cable may be driven
using a 950Ω series resistance and a
50Ω termination to ground, though
the signal will be attenuated (see
Figure 4). A high speed comparator
or inexpensive AHC series CMOS logic
gate may be placed in the signal path
directly after the LTC6905 in order
to buffer the output signal and drive
heavier loads.
Voltage and Current
Controlled Oscillators
The LTC6905 is an ideal candidate
for making a voltage or current controlled oscillator. Unlike other resistor
controlled parts, where the voltage on
RSET varies with power supply and
temperature, the LTC6905 maintains
continued on page 38
Table 1. LTC6905 family fixed frequency oscillators
DIV Setting
LTC6905-133
LTC6905-100
LTC6905-96
LTC6905-80
V+ (÷1)
133.33MHz
100 MHz
96 MHz
80 MHz
OPEN (÷2)
66.66MHz
50MHz
48MHz
40MHz
GND (÷4)
33.33MHz
25MHz
24MHz
20MHz
15
DESIGN FEATURES
Simple, Precise Instrumentation
Amplifier Features Digitally
Programmable Gains from 1 to 4096
by Michael Kultgen
Introduction
The LTC®6915 is the simplest, most
precise way to obtain digitally programmable voltage gain. Any system
which needs accurate amplification of
small differential voltages and rejection of large common mode signals will
benefit from the LTC6915.
The LTC6915 is an evolution of the
LTC2053, a precision rail-to-rail input
and output, zero-drift instrumentation
amplifier. Due to the amplifier’s very
low DC errors, very high levels of gain
can be taken in a single stage. The
LTC2053 uses external resistors to set
the gain; the LTC6915 uses a serial
port or a parallel port to select internal
resistors, and therefore select the gain.
The gain can be programmed to 0, 1, 2,
4, 8, 16, 32, 64, 128, 256, 512, 1024,
2048, or 4096. Programmable gain
increases the dynamic range of any
system. A fixed gain instrumentation
amplifier would have about 60dB of
useful range. The dynamic range of
the LTC6915 is more than 120dB.
Since the high CMRR (typically
125dB) is independent of the gain setting, microvolts of differential signal
can be extracted from volts of common
mode noise. Furthermore, the common
mode level of the differential signal can
be any value within the supply rails
of the LTC6915.
Other features of the LTC6915
include a flexible digital interface, a
Kelvin connected output stage, a wide
supply range, a shutdown mode, and
a choice of packages.
How it Works
Figure 1 shows a block diagram of the
LTC6915. A sophisticated, charge balanced sampling technique impresses
the differential input voltage on to a
1000pF internal capacitor. The differential input signal is converted to
a single-ended signal referenced to
16
IN +
3
+
CS
IN –
CH
2
–
15
14
CF
OUT
SENSE
GAIN
CONTROL
RESISTOR
ARRAY
13
PARALLEL_SERIAL
11
CS(D0)
DIN(D1)
CLK(D2)
DOUT(D3)
6
7
5
4-BIT
LATCH
MUX
1
Q0 Q1 Q2 Q3 Q4 Q5 Q6 Q7
8-BIT
SHIFT-REGISTER
8
16
10
4
9
REF
HOLD_THRU
SHDN
V+
DGND
V–
a. The LTC6915 in a GN16 Package
IN +
2
+
CS
IN –
1
CH
–
11
CF
OUT
GAIN
CONTROL
RESISTOR
ARRAY
10
PARALLEL_SERIAL
9
MUX
CS(D0)
DIN(D1)
CLK(D2)
DOUT(D3)
4
5
6
REF
4-BIT
LATCH
DGND
Q0 Q1 Q2 Q3 Q4 Q5 Q6 Q7
8-BIT
SHIFT-REGISTER
12
8
3
7
V+
DGND
V–
b. The LTC6915 in a DFN12 Package
Figure 1. The LTC6915 block diagram. The small GN16 package (a) has all
control features. The DFN12 package version (b) sacrifices the shutdown
and data latching control features for even smaller size.
the “REF” pin of the LTC6915. This
single-ended signal is then amplified
by a zero-drift op amp connected as
a non-inverting gain stage. With OUT
connected to SENSE, the gain is set by
an integrated precision resistor ladder.
For gains up to 1024V/V, the accuracy
is guaranteed to be better than ±0.6%
over temperature (Figure 2), with a
typical gain drift of less than 2ppm. A
Linear Technology Magazine • March 2005
DESIGN FEATURES
1.5
MIN/MAX OF
1000 UNITS
GAIN ERROR (%)
1
0.5
0
-0.5
MEAN OF
1000 UNITS
-1
-1.5
1
10
DATASHEET LIMITS
100
1k
GAIN SETTING (V/V)
10K
Figure 2. The gain accuracy of the
LTC6915 is typically better than 0.5%.
3kHz sampling rate means that signals
from DC to 1.5 kHz can be amplified
by the LTC6915.
The LTC6915 has the outstanding
DC precision inherent in all of Linear
Technology’s zero-drift amplifiers. The
room temperature DC offset is less
than ±10µV with less than ±50nV/°C
of drift. The 10nA maximum input bias
current means there is no additional
DC error from source impedances up
VIN
0.1µF
CS
µP
IN–
OUT
IN+
SENSE
V–
–5V
DIN
CLK
V+
0.1µF
VOUT
–5V
REF
HOLD_THRU
NC
CS(D0)
P/S
DIN(D1)
DGND
VIN
0.1µF
–5V
DOUT
CLK
(D3)
Figure 3. The LTC6915 uses a simple and
standard 3-wire serial interface.
using the falling edge of the clock
to output data, the LTC6915 is immune to the slow rise and fall times
often encountered in optically isolated
interfaces. But that does not imply
that the LTC6915 is slow. When the
DOUT timing is unimportant, data
can be input to the LTC6915 as fast
as 10MHz!
The internal shift register is eight
bits wide; the four LSBs set the gain,
and the four MSBs are ignored. This
simplifies some software designs be-
V+
IN –
OUT
IN+
SENSE
V–
REF
HOLD_THRU
NC
CS(D0)
P/S
DIN(D1)
DGND
CLK(D2) DOUT(D3)
CLK(D2) DOUT(D3)
Q0 Q1 Q2 Q3 Q4 Q5 Q6 Q7
8-BIT
SHIFT-REGISTER
DIN
LTC6915
SHDN #2
0.1µF
4-BIT
LATCH
CS
Versatile Serial Control
Connecting the PARALLEL/SERIAL
pin to V– puts the LTC6915 in serial
control mode. The chip select (CS),
clock (CLK), and DATA pins form a
simple 3-wire serial input (Figure 3).
For daisy chaining (Figure 4), there
is also a data out (DOUT) pin. By
LTC6915
SHDN #1
0.1µF
4-BIT GAIN
CONTROL CODE
to 10kΩ. The high CMRR and PSRR
make the LTC6915 immune to fluctuations in power supplies or common
mode levels.
The LTC6915 has both a parallel
and serial digital interface. Its unique
logic design makes it possible to guarantee input high and low thresholds
(VIH/VIL) of 2.0 and 0.8 volts for any
power supply voltage from 2.7V to
±5.5V. Therefore, microprocessors
or FPGAs running on a 2.5V supply
can directly interface to the LTC6915
without restriction. The LTC6915 supply can be a single 2.7V up to a split
±5.5V supply without additional level
shifting of the digital inputs.
0.1µF
VOUT
–5V
DOUT
CLK
D15
DIN
D11
D10
D9
D8
D7
D3
D2
D1
D0
GAIN CODE FOR #1
GAIN CODE FOR #2
CS/LD
GAIN CODE
0000
0001
0010
0011
0100
0101
0110
0111
1000
1001
1010
1011
1100
1101-1111
GAIN
0
1
2
4
8
16
32
64
128
256
512
1024
2048
4096
Figure 4. Two LTC6915s can be connected in a daisy chain. The binary codes for each gain setting are also shown.
Linear Technology Magazine • March 2005
17
DESIGN FEATURES
5V
0.1µF
LTC6915
SHDN
VIN1
5V
V
+
–
OUT
IN+
SENSE
V–
REF
IN
HOLD_THRU
NC
CS(D0)
P/S
DIN(D1)
DGND
2N304
INTERCONNECT
RESISTANCE
VOUT
RLOAD
5V
CLK(D2) DOUT(D3)
Figure 5. Kelvin sensing at the output maintains precision with increased load drive.
cause the LTC6915 appears to be a
“byte wide” device, though it is still an
option to load four bits at a time.
Simple Parallel Control
The gain can also be set by simple
pin strapping, or through the use of
a few spare I/O signals. Just connect
the PARALLEL/SERIAL pin to V+ and
put the LTC6915 in parallel control
mode. Four inputs (D3, D2, D1, D0)
directly set the gain of the amplifier
as also shown in Figure 4. There is
even the ability to “latch” the parallel
control bits into the LTC6915 using
the HOLD/TH
 R
 U
 pin (GN16 package).
When this pin goes high, whatever signals are present on D3…D0 are latched
into the LTC6915. Any changes in D3
to D0 are ignored until HOLD/THRU
is brought low again.
A Flexible Output
The SENSE pin allows clever improvements in the load driving or
multiplexing ability of the LTC6915.
In Figure 5 a discrete NPN transistor
is added to increase the load driving
ability. The SENSE pin is Kelvin connected to the load. Gain errors from
the VBE of the NPN and the IR drop
in the interconnect are eliminated.
A precise voltage is delivered to the
load. Pin-strapping is used to fix the
gain of the amplifier to Code 1011,
for a gain of 1024, using the parallel
interface option.
In Figure 6 two LTC6915s are connected as a multiplexer. One device
is programmed for a gain of 0 (threestated output) while the other device
is programmed for a gain ≥1. Using
18
the serial interface, the gain change
of one amplifier, and disabling of the
other occurs simultaneously when the
CS makes a low to high transition.
As a precaution for output shorting,
200Ω resistors are in series with each
output in case both amplifiers are
ever on simultaneously. The SENSE
pin eliminates any error due to the
IR drop in the 200Ω resistor during
normal operation.
In Figure 7 a software programmable current sink is created using the
SENSE pin, an external MOSFET, and
the internal gain network. This design
can sink a wide range of currents, 0µA
to 40.96mA, due to the precision of
the internal zero-drift amplifier and
the wide range of gain settings. With
only 10µV maximum offset voltage
error, the circuit operates with an
input reference voltage of only 250µV
to 500µV. The reference voltage is set
by a small regulated current—fixed
by the 400mV reference voltage of an
5V
5V
0.1µF
0.1µF
SHDN
V+
LTC6915
#1
IN–
OUT
VIN1
–5V
0.1µF
DATA
µP
SELECT
(TTL
LEVELS) CLOCK
IN+
SENSE
V–
REF
SHDN
V+
LTC6915
200
#2
IN–
OUT
200
VIN2
–5V
0.1µF
IN+
SENSE
V–
REF
HOLD_THRU
NC
CS
P/S
DGND
DIN
DGND
DOUT
CLK
DOUT
HOLD_THRU
NC
CS
P/S
DIN
CLK
–5V
VOUT
–5V
Figure 6. A robust variable gain multiplexer using the high impedance
0 gain setting allows very wide dynamic range in multichannel systems.
5V TO 11V
V+
LOAD
0 TO 40.96mA
LTC6915
IN OUT
+
LT6650
–
N VN2222L
RIN
5Ω
250µV TO
500µV
1M
FB
40µA TO 120µA
10k
ROUT
49.9Ω
BINARY
CODE
0 TO 13
4.99k
Figure 7. A wide range programmable precision current source takes
advantage of the precision of the zero-drift amplifier topology.
Linear Technology Magazine • March 2005
DESIGN FEATURES
LT6650—flowing through a 5Ω input
resistor. This small voltage is amplified by the programmed gain of the
LTC6915 and forces a current to flow
through the 50Ω output resistor via
the MOSFET. The voltage compliance
of the current sink is determined by
the maximum voltage output to the
SENSE pin. In this example, the maximum output voltage is only 2.048V,
required when the input reference is
500µV and the gain is 4096.
Each successive control code to the
LTC6915 increases the output current
by a factor of two. Any current level
between the binary weighted intervals
is achieved by adjusting the input
reference voltage. A gain of 1 provides
an output current of 5µA to 10µA,
and a gain of 4096 outputs 20.48mA
to 40.96mA, with better than ±1%
typical linearity.
Bridge Amplifier Application
The LTC6915 is ideal for current
sensing, thermocouple amplifiers,
strain gauges, and many other low frequency and close-to-DC applications.
The internal op amp gain bandwidth
product is 200kHz with a slew rate
of 0.2V/µs and the response time
to 0.1% accuracy to a step change
in gain can vary between 4ms and
15ms. Figure 8 shows the LTC6915
amplifying a bridge sensor. When in
standby the circuit draws less than
100µA. An interrupt to the processor
initiates a measurement cycle, turning on the LTC6915 and acquiring a
conversion result from the LTC2431
A/D converter.
Conclusion
The LTC6915 instrumentation amplifier combines outstanding precision
analog performance with a flexible
digital interface. The result is a software programmable gain stage which
is precise and easy to use.
V+
10k
BRIDGE
SENSOR
ZETEX
ZXM61P02F
R < 10K
V+
C1
0.1µF
SHDN
V+
LTC6915
#1
OUT
IN –
C5
0.1µF
IN+
SENSE
V–
REF
HOLD_THRU
NC
CS(D0)
P/S
DIN(D1)
DGND
C2
0.1µF
VCC
1.25V
CLK(D2) DOUT(D3)
REF
+
REF
–
LTC2431
SDO
SCK
IN
+
CS
IN
–
FO
GND
20
18
VDD
RC6/TX/CK
RC7/RX/OT
RC5/SDO
RC4/SDI/SDA
V+
RC3/SCK/SCL
D1
BAV74LT1
SOT-23
R1
1k
RC1/T1OSI/CCP2
RC0/T1OSO/T1CK1
PIC16LF73
9
10
X1
4MHz
RC2/CCP1
1
RB7
RB6
OSC1/CLKIN
RB5
RB4
OSC2/CLKOUT
RB3
RB2
MCLR/VPP
RB1
RB0/INT
RAS/AN4/SS
RA4/T0CLK1
RA3/AN3/VREF–
RA2/AN2
RA1/AN1
VSS
8
RA0/AN0
VSS
17
16 MOSI
15 MISO
14 SCLK
13 CS1
12 CS2
11
0V
28
27
26
25
24
23
MEASURE
STANDBY
V+
CONTROL SIGNAL
10k
V+
V+
C3
0.1µF
VIN
VOUT
LT1790-1.25
GND1
GND2
C3
1µF
22
21
7
6
5
4
3
2
19
Figure 8. Using the LTC6915 as a direct bridge amplifier in a measurement system with very low standby current
Linear Technology Magazine • March 2005
19
DESIGN FEATURES
Flexible Power Supply Sequencing
by Jeff Heath and Akin Kestelli
and Monitoring
Introduction
require multiple voltage rails that
must start up and shut down in a
specific order, otherwise the ICs can
be damaged. The LTC2924 is a simple
and compact solution to power supply
sequencing in a 16-pin SSOP package
(see Figure 1).
Sure, alternative sequencing solutions are available, but few, if any, can
The LTC2924 is a complete power supply sequencer and supervisor solution
for multivoltage-rail systems, such
as telecommunications equipment,
memory modules, optical systems,
networking equipment, servers, and
base stations. The FPGAs and other
digital ICs used in these applications
Q1
1V
VON = 0.93V
VOFF = 0.91V
0.1µF
Q2
3V
VON = 2.79V
VOFF = 2.73V
0.1µF
Q3
5V
VON = 4.21V
VOFF = 3.76V
0.1µF
0.1µF
Q4
5V
EARLY
VON = 3.32V
VOFF = 2.80V
0.1µF
VCC
10k
10k
OUT1
IN1
OUT2
IN2
OUT3
IN3
OUT4
IN4
ON LTC2924
SYSTEM
CONTROLLER
52.3k
45.3k
6.04k
1.62k
11.8k
7.68k
1.69k
3.09k
How it Works
DONE
FAULT
TMR
150nF
PGT
150nF
HYS/CFG
GND
49.9k
Q1-Q4: IRL3714S
ALL RESISTORS 1%
Figure 1a. Typical 4-supply sequencer using external N-channel MOSFETs for voltage control
POWER SUPPLY 2
POWER SUPPLY 3
5V EARLY*
VON 3.01V
VOFF 2.68V
3.3V
SHDN
POWER SUPPLY 1
SHDN
5V
VON 4.49V
VOFF 3.99V
1.6V
SHDN
VON 1.43V
VOFF 1.27V
VON 2.25V
VOFF 2V
2.5V
SHDN
POWER SUPPLY 4
0.1µF
24.9k
15.8k
49.9k
33.2k
9.31k
11.81k
7.87k
8.45k
VCC
OUT1
10k
10k
OUT2
LTC2924
IN1
IN2
OUT3
IN3
OUT4
IN4
ON
SYSTEM
CONTROLLER
DONE
FAULT
TMR
*5V EARLY MUST BE ON BEFORE
SEQUENCING SUPPLIES
150nF
150nF
PGT
HYS/CFG
GND
49.9k
Figure 1b. Similar application to Figure 1a, but control is via the converter shutdown pins
20
match the ease-of-use, space-saving
design, flexibility, and cost effectiveness of the LT2924. For instance,
solutions that use discrete components
incur a challenging and time-consuming design effort to interface with the
digital system, and consume a significant amount of board real estate.
Another option, an integrated power
supply sequencer, is more expensive
and consumes more board space than
any LTC2924-based solution, and may
require proprietary software and the
programming of complicated digital
registers. Neither of these options
comes close to offering the flexibility
across applications that the LTC2924
does. It can be used out-of-the-box,
with a few external components, to
sequence and supervise just about
any type of power supply, converter,
or power module.
Four power supplies can be easily
sequenced using a single LTC2924,
and multiple LTC2924s can be just
as easily cascaded to sequence any
number of power supplies. With
slightly reduced functionality, six
power supplies can be sequenced
with a single LTC2924 (see Figure 5
and “Sequencing Six Supplies with a
Single LTC2924” in this article).
The LTC2924 controls the start-up
and shutdown sequence, and ramp
rates, of four power supply channels
via output pins (OUT1–OUT4). Each
OUT pin uses a 10µA current source
connected to an internal charge pump
and a low resistance switch to GND.
This combination makes the outputs
flexible enough to connect them directly to power supply shutdown pins,
or to external N-Channel MOSFET
switches. Figure 1a illustrates a typical application for the LTC2924 where
four supplies are sequenced using
external N-channel MOSFETs, and
Figure 1b shows a comparable circuit
with four power supplies sequenced
using their shutdown pins.
Linear Technology Magazine • March 2005
DESIGN FEATURES
The LTC2924’s internal charge
pump allows the designer to use Nchannel MOSFET switches, which are
typically lower in cost and RDS(ON) than
comparable P-channel MOSFETs.
The internal charge pump provides
a gate voltage of VCC + 5V, which
fully enhances an external logic level
MOSFET. The 10µA pull up current
source on the output pin allows the
implementation of a soft-start (ramped
voltage start-up) by including an optional capacitor between the gate of
the MOSFET and ground.
The LTC2924 monitors the output
voltage of each sequenced power supply via four input pins (IN1–IN4). These
inputs use precision comparators and
a trimmed bandgap voltage reference
to provide better than 1% accuracy.
The power ON and power OFF voltage
thresholds are set using resistive dividers for each of the four channels. The
the power ON threshold and the power
OFF threshold is individually selectable on a channel by channel basis
(see “Selecting Resistors for the Turn
On and Turn Off Voltage Thresholds”
in the sidebar for details).
The LTC2924 timer pin (TMR) is
used to provide an optional delay
between the completion of start-up
of one supply, and the start-up of
the next power supply. The delay
time is selected by placing a capacitor between the TMR pin and ground
(delay = 200uS/nF), whereas floating
the TMR pin removes any delay. The
start-up delay can be different than
the shut-down delay. Figure 2 shows
a simple circuit where the shut-down
delay is half the start-up delay.
The LTC2924 also includes a power
good timer (PGT). The LTC2924 starts
the PGT as each individual power supply is enabled. If any power supply fails
to reach its nominal specified voltage
within the allotted time interval, a
Power ON fault is detected. The PGT
is enabled for the time interval set
by a capacitor between the PGT pin
and ground. The PGT is disabled by
grounding the PGT pin.
The LTC2924 DONE pin is used
to report the status of the power sequencing to a system controller. The
LTC2924 signals the completion of an
Linear Technology Magazine • March 2005
VCC
IN1
0.1µF
VCC
OUT1
IN2
IN3
IN4
ON
TMR
150nF
OUT2
LTC2924
OUT3
OUT4
PGT
VCC
FAULT
10k
GND DONE
150nF
POWER ON TIMER DELAY = 30ms
POWER OFF TIMER DELAY = 15ms
Figure 2. Power ON sequence timer delay
longer than power OFF sequence timer delay
entire 4-channel Power ON sequence
by pulling down the open-drain DONE
pin. The completion of a Power OFF
sequence is signaled by releasing the
DONE pin.
The LTC2924 open drain FAULT
pin is bi-directional. The LTC2924
signals a FAULT condition to a system
controller by pulling this pin down.
Conversely, a system controller can
trigger the immediate, simultaneous
turn off of all sequenced power supplies by pulling down the FAULT pin.
This may be used as an alternate way
of powering down a system.
Sequencing and Monitoring
with the LTC2924
The LTC2924 ON pin is used to initiate
Power ON and Power OFF sequences.
The ON pin uses the same precision
comparator circuits as the four IN pins.
The ON pin can either be controlled by
a logic level from a system controller,
or it can be used to sense the voltage
level of an un-sequenced power supply. When the voltage at the ON pin
rises above 0.61V, the LTC2924 initiates the Power ON sequence. At this
point the first timer interval occurs.
Upon completion of the timer interval,
the OUT1 pin is pulled high with its
10µA current source connected to the
internal charge pump voltage. Once
the voltage of the first power supply reaches its preset threshold—as
monitored at the IN1 pin—and after
the second delay, the OUT2 enable
signal is generated for the subsequent
power supply. The sequence repeats
until the forth channel is powered up.
At this time the DONE pin pulls low
to signal the completion of the Power
ON sequence.
The LTC2924 then enters supervisor mode. The LTC2924 continues to
monitor the power supply voltages (at
the IN1–IN4 pins). If any of the power
supplies fall below the designed OFF
voltage, the LTC2924 indicates a fault
and all of the OUT pins are pulled low.
The fault condition is communicated
to the system controller by pulling the
FAULT pin low.
The Power OFF sequence can be
initiated in one of two ways. To turn
off all of the power supplies simultaneously, the system controller can pull
down on the FAULT pin. To sequence
off the power supplies, the system
controller pulls the ON pin voltage low.
The Power OFF sequence is executed
in the reverse order of the Power ON
sequence, with the supply that was
powered up last is powered down first,
and the supply that was powered up
first is powered down last.
Figure 3 illustrates the power
supply up and down sequences for a
typical 4-supply application. After the
ON pin goes low, the timer delay occurs before the OUT4 pin is pulled low.
When the power supply goes below its
turn off voltage, there is another timer
5V
3.3V
2V/DIV
5V
3.3V
2V/DIV
1V
1V
10V/DIV
DONE
10V/DIV
DONE
2V/DIV
ON
2V/DIV
ON
2V/DIV
TMR
2V/DIV
TMR
25ms/DIV
25ms/DIV
Figure 3. Power up and power down sequences for a typical 4-supply circuit
21
DESIGN FEATURES
delay. The sequence repeats until the
final power supply has powered down.
The LTC2924 signals the end of the
Power OFF sequence by releasing the
DONE pin.
VCC
TMR
RHYS
ON
Cascading the LTC2924
for Eight Supplies and Up
Two or more LTC2924s can be cascaded to fully sequence eight or more
supplies. The smart configuration
logic in the LTC2924 makes the job
of cascading multiple LTC2924 ICs
easy. Figure 4 shows three devices
configured to sequence 12 supplies.
To set the sequence of each of the
LTC2924 ICs, the HYS/CFG, DONE,
and ON pins are connected as shown.
See the LTC2924 data sheet for operational details. To sequence more
GND
RHYS
TMR
VCC
HYS/CFG
GND
ON LTC2924 PGT
ON LTC2924 PGT
IN1
OUT1
IN1
OUT1
IN2
OUT2
IN2
OUT2
IN2
OUT2
IN3
OUT3
IN3
OUT3
IN3
OUT3
OUT4
IN4
OUT4
IN4
FAULT
10k
OUT4
DONE
DONE
FAULT
VCC
FAULT
VCC
DONE
10k
FAULT
Figure 4. Cascading LTC2924 ICs to sequence 12 power supplies
than 12 power supplies, simply add
more LTC2924 ICs in the middle, or
2nd position, in Figure 4. To sequence
up to eight power supplies remove the
LTC2924 in the middle position.
VOUT
VOUT
PS3
SHDN
VOUT
VOUT
PS5
SHDN
VOUT
PS6
SHDN
VOUT
VCC
OUT1
OUT2
LTC2924
ON
IN1
OUT3
IN2
OUT4
IN3
DONE
IN4
FAULT
Sequencing Six Supplies
with a Single LTC2924
Figure 5 shows how to sequence six
supplies with one LTC2924. When the
system controller releases the TURN
OFF node, the first power supply turns
on. The ON pin is tied to the output of
the first power supply. Once this power
supply is powered on, the LTC2924
sequentially starts up power supplies
2 through 5. When the DONE pin is
pulled low after the 5th power supply
powers ON, the inverted signal allows
the 6th power supply to turn on. This
inverter can be implemented with a
single transistor. The system controller
can power off all six power supplies
simultaneously by pulling the TURN
OFF node low.
Delayed Remote Sensing
PS4
SHDN
GND
Figure 5. A 6-supply sequencer
22
VCC
HYS/CFG
OUT1
PS2
*VCC EARLY MUST BE ON BEFORE
SEQUENCING SUPPLIES
TMR
DONE
SHDN
TURN OFF
RHYS
IN1
PS1
SYSTEM
CONTROLLER
GND
ON LTC2924 PGT
IN4
SHDN
VCC EARLY*
VCC
HYS/CFG
Remote sensing is a common configuration in high current applications.
Parasitic resistances in the power
supply path coupled with high DC
currents can result in unacceptable
DC voltage drops. The sense pin of
a power module is designed to regulate the DC voltage at a point in the
power distribution circuitry beyond
the parasitic resistance to compensate
for the I • R voltage drop. The output
voltage of the power module is raised
until the desired voltage is reached at
sense point.
The problem with this feedback
scheme is that many power modules
have unalterable maximum output
voltages, which, if exceeded, cause
the power supply to shut down. This
limits the amount of voltage correction
available to compensate for parasitic
Linear Technology Magazine • March 2005
DESIGN FEATURES
MODULE
VOUT VON 4.64V
5V
VOFF 4V
Q2
SENSE+
5V
PARASITIC
RESISTANCE
Q1
OUT+
D1
DC/DC
3.3V
VOUT VON 2.98V
3.3V VOFF 2.65V
1M
SHDN
0.1µF
10k
10k
OUT4
VCC
OUT3
IN4
OUT2
IN3
64.9k
33.2k
9.83k
8.55k
OUT1
IN2
LTC2924
ON
IN1
SYSTEM
CONTROLLER
D1: 1N5711
Q1, Q2: IRL3714S
FAULT
PGT
DONE
HYS
TMR
GND
49.9k
150nF
150nF
Figure 6. Delayed remote sensing
I • R voltage drops. Transient start-up
inrush currents, caused by charging power supply bypass capacitors,
often exceed the normal DC currents
and create a large I • R voltage drop
across the parasitic resistance. If the
sense pin is connected to the remote
sense point, the power module tries
to compensate for the additional
voltage drop by raising its output
voltage, possibly higher than its set
maximum. This, off course, causes
the power module to shut down before
it has even finished starting up. This
problem can be avoided by delaying
remote sensing until the inrush currents have diminished.
Figure 6 shows how delayed remote
sensing can be achieved with the
LTC2924. In Figure 6, Channel 1 is
a DC-DC converter that receives its
input power from the power module.
Channel 2 switches on the power
module that is being remote sensed
and Channel 3 is the remote sense
enable. As Figure 7 shows, when
the LTC2924 ON pin is pulled HIGH
the Power ON sequence is initiated.
After the time delay is executed, the
DC-DC converter connected to OUT1
is enabled. When the output voltage
level on this supply goes above the
user-configured threshold voltage the
second delay is triggered, and then Q1
is turned on. When the output voltage
reaches 4.64V, the 3rd output is enLinear Technology Magazine • March 2005
REMOTE SENSE ENABLE
5V
2V/DIV
3.3V
1V/DIV
ON
1V/DIV
TMR
25ms/DIV
Figure 7. Delayed remote
sensing power up sequence
abled after another delay. This enables
the remote sensing of the power supply
after the initial transient currents have
subsided. As Figure 7 illustrates, the
output voltage of the power supply
increases to the desired level after the
remote sensing is enabled.
Power Supply Fault
Monitoring and Reporting
The LTC2924 has the capability to
monitor the supply levels and report
any fault conditions that are detected.
If one or more of the following errors are
detected, the LTC2924 immediately
turns off all supplies and signals a
F
 A
 U
 L
 T
 condition by pulling the F
 A
 U
 L
 T

pin low. The LTC2924 can detect:
❑ Power ON and Power
OFF sequence errors: The
LTC2924 keeps track of each of
the supplies during the Power
ON sequence, during the time
the power is on, and during the
Power OFF sequence. If at any
time a power supply output goes
low when it should be high, a
fault is generated.
❑ System controller command errors: The ON pin is
the input signal provided by the
system controller to direct the
LTC2924 power sequencing. By
taking this pin HIGH, a Power ON
sequence is initiated. Until all the
power supplies are powered on,
the ON pin must remain HIGH.
During the Power OFF sequence
the voltage on this pin must
remain below 0.61V. If these conditions are not maintained during
the Power ON or Power OFF sequencing, the LTC2924 indicates
a fault condition.
❑ Power Good Timer (PGT)
Power ON timeout failures:
The PGT is enabled with a single
capacitor at the PGT pin with a
transfer function of 200µs/nF. If
a supply that is being sequenced
ON does not reach the desired
voltage level within the time set
by the PGT, a fault is generated.
❑ External faults: The FAULT
pin can also be used as an input.
Pulling the FAULT pin low causes
the LTC2924 to turn off all power
supplies and abort any sequence
in progress.
23
DESIGN FEATURES
Selecting Resistors for the
On and Off Voltage Thresholds
Each of the four channels of the LTC2924 can have its
own values of VON, the turn on voltage threshold, and
VOFF, the turn off voltage threshold. Setting the voltages
is easy—only two resistors are required at the input pin
of each channel, and choosing the resistor values is
simple, as described here.
Refer to Figure SB1. The first step is to select a hysteresis current (IHYS). This current is used by all four
channels, and is programmed by one resistor, RHYS on
the HYS/CFG pin in Figure SB1. The IHYS current is
switched in to each IN1-4 pin when each channel is ON.
Unless the LTC2924 is being used in a very low power
system use 50µA for IHYS. Calculate RHYS from IHYS by
the following:
RHYS =
0.5V
; 0.5µA ≤ IHYS ≤ 50µA
IHYS
or RHYS = 10kΩ for IHYS = 50µA
That leaves the two resistors for each channel. For each
sequenced power supply, choose VON, the voltage at which
power is considered on during a start up sequence, and
VOFF, the voltage at which power is considered off during a
IHYS
VON = 2.2V
VOFF = 1V
VPS
IHYS RB
IN
+
–
RA
IRB
0.61V
Figure SB1. Designing IHYS feedback resistors
Conclusion
The LTC2924 fits into a wide variety
of power supply sequencing and
monitoring applications. With very
few external components and a 1624
VON – VOFF
IHYS
RB • 0.61V
RA =
VONallows
– 0.61Vthe hysteresis band for each channel to
This
RB =
be individually tailored.
RA =
RB • 0.61V
VON – 0.61V
Perform this simple calculation for each channel. For
example, if:
IHYS = 50µA
0.5V
RHYS =
= 10kΩ
50µA
Place this resistor between the HYS/CFG pin and
ground.
With VON and VOFF voltages:
VON = 2.2V
VOFF = 1V
V –V
RB = ON OFF
IHYS
RB • 0.61V
RA =
VON – 0.61V
2.2V – 1V
= 24kΩ
50µA
24kΩ • 0.61V
= 9.2kΩ
RA =
2.2V – 0.61V
RB =
IFB = IRB + IHYS
If any of the conditions above are
met, the LTC2924 pulls all of the OUT
pins low causing all power supplies to
turn OFF. The F
 A
 U
 L
 T
 pin is also pulled
low to report the event to a system
controller. The TMR pin is also pulled
high if the fault condition was generated internally. The fault condition is
not reset until all of the IN pins and the
ON pins are below 0.61V.
shut down sequence. Referring to Figure SB1, RB is the
resistor connected between the sequenced power supply
and the IN pin and RA is connected between the IN pin
and ground. Each resistor can be then calculated by:
Repeat the last four calculations for the remaining
three channels.
pin narrow SSOP, an LTC2924 based
sequencing solution requires very little
board space.
The power supply enable pins require no configuration by the designer,
yet are versatile enough to directly drive
shutdown pins or external N-channel
MOSFETs. Soft start of power supplies
can be achieved simply by adding a
capacitor. If the sequencing of more
than four power supplies is required,
the LTC2924 can be cascaded to sequence a virtually unlimited number
of power supplies. With the addition
of a single capacitor, a timer can be
enabled and programmed. Adding one
more capacitor programs and enables
a Power Good Timer (PGT). Power supplies can be turned off in the reverse
order of turn on, or they can all be
turned off at the same time.
Tailoring the LTC2924 to a specific
application requires no software and
designs can be fine tuned during system integration simply by changing
resistor and capacitor values. Ease
of design, low component cost, and
a small footprint make the LTC2924
an excellent choice for power supply
sequencing and monitoring.
Linear Technology Magazine • March 2005
DESIGN FEATURES
High Voltage Step-Down Synchronous
Controller Offers Single-Supply
Operation, Current Mode Control,
and 100µA Burst Mode Operation
by Jay Celani
Introduction
As more features and functions are
packed into electronics packages, efficient step-down DC/DC conversion
circuits that can handle high input
voltages at substantial load currents
are increasingly necessary. This is
especially true for distributed power
systems that have high power point
of load requirements. The LT3800 is
a feature-packed high voltage synchronous step-down controller that
simplifies meeting these high power
requirements.
The LT3800 is the core of singlesupply DC/DC converter solutions
that require few external components
and maintain high-efficiencies over
wide load ranges. Burst Mode operation and a reverse inductor current
inhibit feature maximize efficiencies
during light-load and no-load conditions, making the LT3800 ideal
for use in applications with supply
maintenance requirements. Maintenance requirements are common in
automotive applications where a low
current standby mode is required in
addition to high power operating conditions. Both Burst Mode operation and
reverse inductor current inhibit can
be disabled if desired.
The LT3800 contains an integrated
start-up regulator that powers the IC
directly from the input supply for true
single-supply operation. The IC uses a
200kHz fixed-frequency current mode
architecture and operates with input
voltages from 4V to 60V. A precision
shutdown pin threshold allows for
easy integration of input supply under-voltage lockout (UVLO) using a
simple resistor divider, and quiescent
currents are reduced to less than 10µA
while the IC is in shutdown. Converter
output voltage is programmed using
Linear Technology Magazine • March 2005
a resistor divider, and the IC includes
a 1% accurate internal reference. The
LT3800 also incorporates a programmable ΔV/Δt soft-start that directly
controls the rising rate of the converter
output voltage at start-up.
The LT3800 employs continuous
high-side inductor current sensing using an external sense resistor. Inductor
“The LT3800 is a versatile
platform on which to build
high-voltage DC/DC converter
solutions that use few
external components and
maintain high-efficiencies
over wide load ranges.”
current is limited to the same value in
both positive and negative directions,
protecting the converter from both
source and sink over-current events,
and current limit is unaffected by
duty-cycle.
A LT3800 DC/DC converter uses
standard-level N-channel MOSFETs
for main and synchronous switches,
employing a bootstrapped supply rail
for the main switch MOSFET driver.
High current switch-drivers allow the
use of low RDS(ON) MOSFETs without
the need for gate drive buffers.
The LT3800 is available in a smallfootprint thermally-enhanced 16-lead
TSSOP package.
Onboard Start-Up Regulator
The LT3800 eliminates the need for
an external regulator or a slow-charge
hysteretic start scheme through integration of an 8V linear regulator.
This regulator generates VCC, the lo-
cal supply that runs the IC, from the
converter input supply, VIN.
The onboard regulator can operate
the IC continuously, provided the input
voltage and/or FET gate charge currents are low enough to avoid excessive
power dissipation in the part. Forcing
the VCC pin above its 8V regulated
voltage allows use of externally derived
power to minimize power dissipation
in the IC, reducing thermal considerations. Using the onboard regulator for
start-up then deriving power for VCC
from the converter output maximizes
conversion efficiencies and is common
practice.
The LT3800 has a start-up requirement of VIN ≥ 7.5V. This assures that
the onboard regulator has ample
headroom to bring the VCC pin above
its UVLO threshold of 6.25V. If VCC is
maintained using an external source,
such as the converter output, the
LT3800 can continue to operate with
VIN as low as 4V.
Burst Mode Operation
The LT3800 supports low current
Burst Mode operation to maximize
efficiency during low-load and no-load
conditions. Burst Mode operation is
enabled by shorting the BURST_EN
pin to SGND, and can be disabled
by shorting BURST_EN to either VFB
or VCC.
When the peak switch current is
below 15% of the programmed current
limit, Burst Mode function is engaged.
During the Burst interval, switching
ceases and all internal IC functions
are disabled, which reduces VIN pin
current to 20µA and reduces VCC current to 80µA. If no external drive is
provided for VCC, all VCC bias currents
originate from the VIN pin, giving a
25
DESIGN FEATURES
100
+
56µF
×2
VIN
LT3800
1nF
R1
20k
1%
SHDN
200k
R2
174k
1%
100pF
82.5k
680pF
SW
CSS
Si7850DP
D1
BAS19
15µH
BURST_EN
VCC
VFB
VC
1µF
TG
D2
1N4148
0.015
+
26
1
10
1
0
VOUT
12V AT 75W
270µF
When reverse-current inhibit is
enabled, the LT3800 sense amplifier
detects inductor currents approaching
zero and disables the synchronous
switch for the remainder of that
switch cycle, simulating the light-load
switching characteristics of a non-synchronous converter. Reverse-current
inhibit reduces losses associated with
inductor ripple currents, improving
conversion efficiencies with loads that
are less than half of the peak inductor
ripple current.
VIN supply. The resistor divider is set
such that the divider output puts
1.35V onto the SHDN pin when VIN is
at the desired UVLO rising threshold
voltage. The SHDN pin has 120mV of
input hysteresis, which allows the IC
to resist almost 10% of input supply
droop before disabling the converter.
The S
 H
 D
 N
 pin has a secondary threshold of 0.5V, below which the IC operates
in an ultralow-current shutdown mode
with IVIN < 10µA. The shutdown function can be disabled by connecting the
SHDN pin to VIN through a large value
pull-up resistor.
Continuous High-Side
Inductor Current Sensing
Precision Shutdown Threshold The LT3800 uses a wide commonThe LT3800 has a precision-threshold
shutdown feature, which allows use
of the SHDN pin for analog monitoring applications, as well as logic-level
controlled applications.
Input supply UVLO for sequencing
or start-up over-current protection is
easily achieved by driving the SHDN
pin with a resistor divider from the
10V/DIV
Figure 3. Output soft-start waveform
for the DC/DC converter in Figure 1
2
LOSS (48V)
Figure 2. Converter efficiency and power
loss for the DC/DC converter in Figure 1
Figure 1. This 20V–55V to 12V 75W DC/DC converter features
reverse current inhibit and input under-voltage lockout.
1ms/DIV
80
3
ILOAD (A)
10µF
2V/DIV
4
VIN = 48V
85
70
0.1
Si7370DP
SENSE–
SENSE+
SGND
The LT3800 contains a reverse-current inhibit feature, which maximizes
efficiency during light load conditions.
This mode of operation prevents negative inductor current, and is sometimes
called “pulse-skipping” mode. This
feature is always enabled with Burst
Mode operation when the BURST_EN
pin is connected to ground. The reverse-current inhibit feature can also
be enabled without Burst Mode by
connecting the BURST_EN pin to the
VFB pin, which is the configuration
used for the DC/DC converter shown
in Figure 1.
VIN = 55V
90
B160
PGND
Reverse Current Inhibit
5
75
1µF
BG
total VIN current of 100µA. An internal
negative-excursion clamp on the VC pin
is set 100mV below the switch disable
threshold, which limits the negative
excursion of the pin voltage during the
Burst interval. This clamp minimizes
converter output ripple during Burst
Mode operation.
VIN = 36V
95
POWER LOSS (W)
RA
RB 1M
82.5k
BOOST
6
VIN = 24V
1µF
×3
EFFICIENCY (%)
VIN
20V TO 55V
mode input range current sense
amplifier that operates from 0V to
36V. This current sense amplifier
provides continuous inductor current
sensing via an external sense resistor.
A continuous inductor current sensing scheme does not require blanking
intervals or a minimum on-time to
monitor current, common to schemes
10V/DIV
ILOAD = 2A
2µs/DIV
Figure 4. Switching waveform for the
DC/DC converter in Figure 1
ILOAD = 0.5A
2µs/DIV
Figure 5. Light load switching
waveform for the converter in Figure 1
Linear Technology Magazine • March 2005
DESIGN FEATURES
VIN
6.5V TO 55V
C2
1µF
100V
X7R ×3
C8
56µF
63V
×2
+
RA
1M
C7
1.5nF
R1
100k
1%
R2
309k
1%
R3
62k
C9
470pF
C10
100pF
VIN
BOOST
NC
TG
LT3800
SHDN
C1 1µF
16V X7R
D1
BAS19
M1
Si7850DP
×2
SW
R4 75k
L1
5.6µH
CSS
BURST_EN
VCC
VFB
BG
VC
R5
47k
C3 1µF
16V X7R
M2
Si7370DP
×2
DS3
B160
×2
PGND
SENSE–
SENSE+
SGND
RS
0.01
D2
1N4148
DS2
MBRO520L
DS1
MBRO520L
C4
1µF
C6
10µF
6.3V
X7R
M3
1/2 Si1555DL
M4
1/2 Si1555DL
+
C5
220µF
×2
VOUT
5V AT 10A
C5: SANYO POSCAP 6TP220M
L1: IHLP-5050FD-01
Figure 6. This 6.5V–55V to 5V 10A DC/DC converter features charge pump doubler VCC refresh and current limit foldback.
that sense switch current. The sense
amplifier monitors inductor current
independent of switch state, so the
main switch is not enabled unless
the inductor current is below what
corresponds to the VC pin voltage.
This turn-on decision is performed at
the start of each cycle, and individual
switch cycles will be skipped should
an over-current condition occur. This
eliminates many of the potential overcurrent dangers caused by minimum
on-time requirements, such as those
that can occur during startup, shortcircuit, or abrupt input transients.
Soft Start
The LT3800 employs an adaptive softstart scheme that directly controls
100
12
10
VIN = 24V
VIN = 13.8V
VIN = 48V
90
8
VIN = 55V
85
POWER LOSS
VIN = 48V
80
75
70
0
2
4
6
IOUT (A)
6
4
POWER LOSS
VIN = 13.8V
8
2
10
0
Figure 7. Efficiency and power loss
for the DC/DC converter in Figure 6
Linear Technology Magazine • March 2005
POWER LOSS (W)
EFFICIENCY (%)
95
the DC/DC converter output voltage
during start-up. The rising rate of
this voltage is programmed with a
capacitor connected to the converter
output. The capacitor value is chosen
such that the desired ΔV/Δt of the
output results in a 2µA charge current
through the capacitor. The soft start
function maintains this desired output
rising rate up to 95% of the regulated
output voltage, when the soft-start
circuitry is disabled. The soft-start
function is automatically re-enabled
if the converter output droops below
70% regulation, so converter recovery is graceful from a short duration
shutdown or an output short-circuit
condition.
20V–55V to 12V,
75W DC/DC Converter
Figure 1 shows a 20V–55V to 12V
75W converter, configured for reverse
current inhibit operation and input
UVLO. Power for the IC is obtained
directly from VIN through the LT3800’s
internal VCC regulator at start-up.
The main switch bootstrapped supply is refreshed via D1 from the 8V
generated on the VCC pin. When the
converter output comes up, D2 pulls
VCC above regulation, disabling the
internal regulator and providing a current path from the converter output to
the VCC pin. With the VCC pin driven
from the converter output, VIN current
is reduced to 20µA. Using outputgenerated power in high input voltage
converters results in significant reduction of IC power dissipation, which
increases overall conversion efficiency,
but is critical to reduce IC thermal
considerations. Figure 2 shows the
conversion efficiency and power loss
for this DC/DC converter.
Output voltage is programmed using R1 and R2, and the output is in
regulation when the voltage at the VFB
pin is 1.231V. VIN UVLO is programmed
via RA and RB, enabling the LT3800
at 90% of the specified low end of the
VIN range, or 18V, which corresponds
to 1.35V on the SHDN pin. The SHDN
input has 120mV of hysteresis, so the
converter will be disabled if VIN drops
below 16V.
The LT3800 soft-start function controls the rising slope of the output at
startup such that the ΔV/Δt current
through C8 is 2µA, so the converter
output will rise at a controlled rate of
2µA/1nF, or 2V/mS. Figure 3 shows
the soft-start ramp.
The BURST_EN pin is tied to the VFB
pin to disable Burst Mode operation
while keeping reverse current inhibit
operation enabled. Figure 4 shows
continuous current operation when
27
DESIGN FEATURES
VIN
9V TO 38V
+
C8
100µF
50V
×2
RA
RB 1M
187k
R3
82k
C2
330pF
C10
100pF
BOOST
NC
TG
LT3800
SHDN
C1 1nF
R1
100k
1%
VIN
R2
169k
1%
C3
100pF
R4 39k
C5 1µF
16V X7R
C9
4.7µF
50V
X7R ×3
M1
Si7884DP
SW
D1
MBR520
CSS
BURST_EN
VFB
VCC
BG
VC
C4 1µF
16V X7R
L1
3.3µH
M2
Si7884DP
DS1
SS14
×2
PGND
SENSE
–
SENSE+
SGND
RS
0.01
C7
10µF
6.3V
X7R
C6: SANYO POSCAP 4TPD470M
L1: IHLP-5050FD-01
+
C6
470µF
×2
VOUT
3.3V AT 10A
Figure 8. A 9V–38V to 3.3V 10A DC/DC converter with VIN UVLO
the load is greater than half of the
peak ripple current. With lighter loads,
during the switch off interval, as the
inductor current approaches zero, the
synchronous switch is disabled. The
resulting discontinuous switching
waveform is shown in Figure 5.
6.5V–55V to 5V,
10A DC/DC Converter
In LT3800 converter applications with
output voltages in the 9V to 20V range,
back-feeding VCC from the converter
output is trivial, accomplished by
connecting a diode from the output to
the VCC pins. Outputs lower than 9V
require step-up techniques to generate
back-feed voltages greater than the
7
VIN = 13.8V
90
6
88
5
86
4
84
3
82
2
80
1
78
0.1
1
ILOAD (A)
0
10
Figure 9. Efficiency and power loss for
the DC/DC converter in Figure 8
POWER LOSS (W)
EFFICIENCY (%)
92
VCC regulated output. The 6.5V–55V
to 5V 10A DC/DC converter shown in
Figure 6 uses an external Si1555DL
MOSFET pair (M3, M4) to create a
charge pump doubler that steps up
the output voltage. This simple doubler
uses the synchronous gate drive (BG
pin) as a control signal.
This converter also uses an external
current limit foldback scheme. The
foldback circuit consists of a single
1N4148 diode (D2) and a 47k resistor
(R5). The current limit foldback circuit
provides additional control during the
first few switch cycles of start-up,
and provides reduced short-circuit
output current. When the output
is at ground, the diode and resistor
clamp the VC pin to a value that corresponds to 25% of the programmed
maximum current. This circuit is only
active with VOUT close to ground, and
becomes completely disabled once
the output voltage rises above 10%
regulation. Figure 7 shows the conversion efficiency and power loss for
this converter.
9V–38V to 3.3V,
10A DC/DC Converter
In some DC/DC converter applications, the typical input voltage is
moderate, but the converter must
withstand or operate through intermittent high-voltage excursions. This
is typical of automotive battery-voltage applications, where high voltage
line transients, such as during a
load-dump condition, must be accommodated. The 9V–38V to 3.3V
10A DC/DC converter with VIN UVLO
shown in Figure 8 is an automotive
application that typically operates
with VIN = 13.8V, but can operate
through VIN excursions from 9V up to
38V. Because the typical line voltage
is moderate, the LT3800 can operate
directly from the internal VCC regulator
without excessive power dissipation,
eliminating the need for a step-up
scheme to regenerate VCC from the
converter output. Figure 9 shows the
conversion efficiency and power loss
for this circuit.
Conclusion
The LT3800 is a versatile platform
on which to build high voltage DC/
DC converter solutions that use few
external components and maintain
high efficiencies over wide load ranges.
The integrated start-up regulator facilitates true single-supply operation
and Burst Mode function enables
efficient solutions to power-supply
maintenance requirements.
For more information on parts featured in this issue, see
http://www.linear.com/designtools
28
Linear Technology Magazine • March 2005
DESIGN IDEAS
Buck-Boost Converter Minimizes
Output Voltage Transients from
Very Low to High Output Current
Introduction
Most handheld devices incorporate
a low power mode (Burst Mode operation) to save precious battery life
during an extended period of inactivity.
Transitions from Burst Mode operation, however, can induce transient
perturbations in the output voltage,
which drain the battery unnecessarily.
The LTC3443 Buck-Boost converter
minimizes output voltage transient
perturbations, and thus realizes the
promise of Burst Mode operation to
significantly increase battery run
time.
The LTC3443 incorporates an adaptive clamp on the VC pin—active during
Burst Mode operation—which holds
the error amp integrator capacitor
to a fixed voltage determined by the
input and output voltage. The clamp
is removed when the LTC3443 is commanded out of Burst Mode operation.
In this way, the compensation capacitors are already close to the nominal
steady state voltage at the transition
out of Burst Mode operation, so the
output voltage transient magnitude
and duration is minimized.
DESIGN IDEAS
Buck-Boost Converter Minimizes
Output Voltage Transients from
Very Low to High Output Current . 29
Mark Jordan
2-Phase Controller for High Current,
High Step-Down Ratio Applications
.................................................... 30
by Mark Jordan
L1
6µH
VIN
2.5V
TO 4.2V
Li-Ion
SW1
SW2
PVIN
VOUT
LTC3443
FB
VIN
SHDN/SS
C1
10µF
VC
MODE/SYNC GND
PGND
VOUT
3.3V
1A
220pF
340k
2.2k
15k
560pF
PGND
C2
44µF
(2 × 22µF)
200k
C1: TAIYO YUDEN JMK212BJ106MG
C2: TAIYO YUDEN JMK325BJ226MM
L1: SUMIDA CDRH6D28-6R0NC
Figure 1. Lithium-Ion to 3.3V converter at 1A utilizing all ceramic capacitors
3.3W Li-Ion to 3.3V Converter
LTC3443 Features
A typical application for the LTC3443
is illustrated in Figure 1 with a Li-Ion
battery as the input source with the
output voltage set to 3.3V at 1A max.
Peak efficiency for the application is
96% and 94% during the Buck-Boost
region (VOUT ≈ VIN) when all four switches are commutating. Figure 2 shows
the output voltage response when the
LTC3443 transitions from Burst Mode
operation to fixed frequency operation.
The output transient is within 3% of
the nominal output voltage. The output
ripple during Burst Mode operation is
typically 1%. A +1% offset is incorporated in the DC value of the output
voltage during Burst Mode operation
to better “voltage position” the output
in case of an immediate load transient.
Figure 3 shows the reverse transition:
from fixed frequency operation to Burst
Mode operation.
The LTC3443 has an internally
trimmed 600kHz oscillator, which
can be synchronized from 690kHz to
1.2MHz. The input range is 2.4V to
5.5V and the output range is specified
from 2.4V to 5.25V. The output can
operate as low at 0.4V with the addition of Schottky diode. The LTC3443
has true output disconnect and inrush
current control via a soft start function. The quiescent current in Burst
Mode is a mere 28µA, maximizing light
load efficiency. During shutdown the
supply current is less than 1µa. The
LTC3443 is designed to withstand a
short circuit by incorporating features
such as foldback current limit and
thermal shutdown. All of this power
and functionality is packed into a tiny
4mm by 3mm thermally enhanced
surface mount DFN package.
VOUT
100mV/DIV
AC COUPLED
MODE/SYNC
5V/DIV
VOUT
100mV/DIV
AC COUPLED
MODE/SYNC
5V/DIV
Xiaoyong Zhang
2A, 40V, SOT-23
Boost Converter Provides
High Power in Small Spaces ........ 31
Jeff Witt
Efficient and Reliable Drive for
Synchronous MOSFET Rectifiers .. 33
Goran Perica
Low-Distortion Sine Wave Oscillator
with Precise RMS Amplitude
Stability ...................................... 36
Cheng-Wei Pei
Linear Technology Magazine • March 2005
VOUT = 3.3V
VIN = 3.3V
IOUT = 30mA
500µs/DIV
Figure 2. Transient response of the converter
in a transition from Burst Mode operation to
fixed frequency operation
VOUT = 3.3V
VIN = 3.3V
IOUT = 30mA
500µs/DIV
Figure 3. Transient response of the converter
in a transition from fixed frequency operation
to Burst Mode operation
29
DESIGN IDEAS
2-Phase Controller for High Current,
High Step-Down Ratio Applications
by Xiaoyong Zhang
One of the biggest challenges in designing power supplies for high speed
digital systems is achieving high stepdown ratios at high load currents, all
while maintaining high efficiency and
meeting stringent transient response
and board space requirements. Designers can easily meet this challenge
by using the LTC3709 dual phase,
synchronous step-down switching
regulator.
The LTC3709 uses a constant ontime with phase locked loops (PLLs),
valley current control architecture to
deliver very low duty cycles and does
not require an output current sense
resistor. Figure 1 shows the LTC3709
in a step down circuit that features fast
transient response and high efficiency
over wide load range.
High Current
and High Efficiency
Power losses, and the resulting heat,
are significant problems in high
current systems, so power supplies
must be as efficient as possible. The
LTC3709 guarantees high efficiency
from light load to heavy load, especially important in high power portable
computers (Figure 2).
Much of the efficiency of an
LTC3709-based circuit is a result of its
2-phase architecture, which enables
supply currents over 30A. The two
channels operate out of phase, thus
minimizing the input RMS current
and the power loss along the input
supply path.
The LTC3709 also senses current
through the bottom MOSFET, so there
is no added power loss from sense
resistors, and the powerful onboard
synchronous MOSFET drivers effectively suppress conduction losses.
The LTC3709 also offers Stage
Shedding™ mode to boost the efficiency at light load. In Stage Shedding
mode, the second channel is turned off
30
1µF
5V
4.7µF
10k
0.1µF
100nF
680pF
BOOST1
PGOOD
SW1
SENSE1+
RUN/SS
BG1
20k
0.22µF
1.22µH
HAT2165H
EXTLPF
SENSE1–
INTLPF
PGND1
ITH LTC3709
10k
TG2
BOOST2
15k
HAT2168H
TG1
SGND
VFB
DIFFOUT
SW2
SENSE2+
VOS–
VOS+
SENSE2–
PGND2
VIN
4.5V TO 28V
10µF
35V
×3
VCC DRVCC ION
TRACK
VRNG
FCB
100k
3.32k
PGND2
324k
10
1µF
47.5k
PGND1
1µF
VIN
HAT2168H
0.22µF
1.22µH
HAT2165H
BG2
VOUT
1.5V
30A
+
330µF
2.5V
×4
Figure 1. This high current, 2-phase power supply is
efficient and responds quickly to load transients.
at light loads, which halves the light
load switching losses.
If the load current further drops
very low, no reverse inductor current
is allowed and the switching frequency
drops down as low as necessary to
maintain regulation, while keeping
the efficiency high.
has an inherently fast transient response. The LTC3709 responds to a
load transient immediately, without
the clock latency typical of traditional
constant frequency controllers.
Constant On-Time
Architecture and Fast
Transient Response
The two channels of the LTC3709
operate 180 degrees out of phase. A
Modern power supply designs often
require a high step down ratio (low duty
cycle) and fast operation frequency
at the same time. This means a very
short on-time feature is indispensable for an excellent controller. Unlike
traditional constant frequency controllers that have minimum on-times
of several hundreds of nanoseconds,
the LTC3709 has a minimum on-time
of only 50ns, which makes it a good
choice for high current power designs.
The constant on-time, valley current
control architecture of the LTC3709
Anti-Phase Operation
and External Clock
Synchronization
continued on page 32
100
95
VIN = 12V
90
EFFICIENCY (%)
Introduction
EFFICIENCY
85
80
75
70
65
60
55
50
0.01
0.1
1
10
LOAD CURRENT (A)
100
Figure 2. The circuit in Figure 1 has high
efficiency over a wide load current range.
Linear Technology Magazine • March 2005
DESIGN IDEAS
2A, 40V, SOT-23 Boost Converter
Provides High Power in Small Spaces
by Jeff Witt
Introduction
L1
1.8µH
VIN
2.3V TO 4.8V
C1
4.7µF
ON OFF
VIN
SOT-23 Boost with 2A Switch
Figure 1 shows the LT1935 generating
5V. Maximum load with VIN = 3.3V is
1A; from 2.5V the maximum load is
600mA. Note that the circuit efficiency
D1
SW
R1
29.4k
LT1935
SHDN
The small size eases system design
in many applications. Large digital
systems with dense layouts often
need point-of-load converters to generate secondary logic supplies. With
a minimum input voltage of 2.3V, the
LT1935 can convert power from 2.5V,
3.3V or 5V logic rails to a higher output
voltage. Even handheld electronics
such as cell phones, digital cameras
and music players require peak power
levels of several watts to drive LEDs,
audio amplifiers or large displays.
And space is always at a premium in
these products.
C3
150pF
C2
20µF
FB
R2
10k
GND
remains high even at low input voltage
and high load current. The LT1935’s
bipolar NPN power switch maintains
its low forward drop when the input
voltage is at its minimum of 2.1V (2.3V
max), unlike some MOS devices that
suffer increased RDS,ON with low gate
drive. The circuit in Figure 1 occupies
80mm2. Figure 2 shows a 12V circuit
that generates 600mA from 5V or
320mA from 3.3V. This higher power
circuit requires 100mm2 of PCB.
Soft-Start Reduces
Peak Input Current
During start-up, the input current of
an LT1935 circuit can reach 3A. This
can cause problems if the input source
is current-limited or if other circuits
are sensitive to disturbances at VIN.
The SHDN pin can be used to soft start
90
VOUT
5V
1A, VIN = 3.3V
0.6A, VIN = 2.5V
85
VIN = 3.3V
80
EFFICIENCY (%)
The LT1935 is a current mode boost
regulator in a tiny 5-lead ThinSOT
package. With its small package, high
switching frequency (1.2MHz) and
internal 2A, 40V power switch, the
LT1935 can deliver high power while
occupying very little circuit board
space. For instance, from a 5V input,
the LT1935 delivers 500mA average
and 600mA peak current at 12V (7.2W)
using only100mm2 of PCB.
The LT1935’s power switch drops
just 180mV at 2A, minimizing power
loss and temperature rise on the circuit board. Current mode control and
internal compensation allow the use
of small ceramic capacitors, resulting
in very low input and output ripple.
The input voltage range is 2.3V to
16V. Supply current is less than 1µA
in shutdown.
VIN = 2.5V
75
70
65
60
C1, C2: X5R OR X7R 6.3V
D1: ON SEMI MBRM120
L1: SUMIDA CR43-1R8
55
50
0
200
400
600
800
1000
1200
LOAD CURRENT (mA)
Figure 1. The LT1935 can deliver 1A at 5V from a 3.3V input in a circuit that occupies only 80mm2.
VIN
3V–11V
C1
4.7µF
ON OFF
VIN
90
D1
VOUT
12V
320mA, VIN = 3.3V
600mA, VIN = 5V
SW
47pF
LT1935
SHDN
R1
84.5k
FB
GND
D1: ON SEMI MBRM120
L1: SUMIDA CDRH5D28-4R2
R2
10k
C2
22µF
VIN = 5V
85
80
EFFICIENCY (%)
L1
4.2µH
VIN = 3.3V
75
70
65
60
55
50
0
100
200
300
400
500
600
700
LOAD CURRENT (mA)
Figure 2. The LT1935 delivers 600mA at 12V from a 5V input. High power density is achieved
using the internal 2A, 40V, 90mΩ switch and the high 1.2MHz operating frequency.
Linear Technology Magazine • March 2005
31
DESIGN IDEAS
the LT1935, reducing the maximum
input current during start-up.
The SHDN pin is driven through an
external RC filter to create a voltage
ramp at this pin. Figure 3 shows the
start-up waveforms with and without
the soft-start circuit. Without softstart, the input current peaks at ~3A.
With soft start, the peak current is
reduced to 1A. By choosing a large
RC time constant, the peak start-up
current can be reduced to the current
that is required to regulate the output,
with no overshoot. (The value of the
resistor should be chosen so that it
can supply 100µA when the SHDN
pin reaches 1.8V.)
ON OFF
VOUT
2V/DIV
IIN
1A/DIV
RUN
SHDN
GND
from cell phones to televisions. Power
requirements grow as well, but the
basic need for three supply voltages
C5
0.1µF
C3
1µF
D2B
D2A
16V
10mA
8V
450mA
SW
LT1935
SHDN
D1
R1
100k
C2
10µF
FB
GND
C1: X5R OR X7R 6.3V
C2, C4, C5, C6: X5R OR X7R 10V
C3: X5R OR X7R 25V
D1: MBRM120 OR EQUIVALENT
D2, D3: BAT-54S OR EQUIVALENT
L1: SUMIDA CDRH4D28-2R2
R2
18.7k
C6
0.1µF D3A
D3B
C4
1µF
–8V
10mA
Figure 4. This TFT-LCD supply produces three outputs using a single inductor.
LTC3709, continued from page 30
PLL monitors the switching of the two
channels and forces the switching
frequency of the second channel to
follow that of the first channel. The
interleaved operation of two channels
minimizes the input RMS current
and power loss along the input supply path.
A second PLL is provided for external
clock synchronization. The LTC3709
is synchronized by adjusting its ontime, indirectly adjusting its switching
frequency. When synchronized, the
LTC3709 combines the advantages
of constant frequency and constant
on-time architectures. The switch32
200µs/DIV
RUN
10k
0.22µF
SHDN
GND
Figure 3. The SHDN pin can be used to soft start the LT1935
reducing the peak input current during start up.
L1
2.2µH
VIN
VOUT
2V/DIV
20µs/DIV
TFT LCD display panels continue to
grow in size in every type of product
C1
4.7µF
RUN
5V/DIV
IIN
1A/DIV
More Power for
Larger LCD Panels
VIN
3.3V
RUN
5V/DIV
ing frequency stays constant despite
the changes of input voltage, output
voltage (if programmable) and load current. The LTC3709 can still respond to
load transients without clock latency
because of the indirect adjustment of
switching frequency during synchronization. The time constant of the PLL
is much longer than the load transient
duration, so the switching frequency of
the LTC3709 is temporarily altered to
take advantage of a constant on-time
architecture.
Other Features
The LTC3709 has a differential amplifier for remote sensing of both the high
remains. In Figure 4 the LT1935 produces three outputs using a single
inductor. From a 3.3V input, the boost
circuit produces the main output of
8V at 450mA. Two discrete charge
pumps produce the secondary outputs
of 16V and –8V.
Conclusion
By integrating a high frequency, current mode control with 2A, 90mΩ
switch in a SOT-23, the LT1935
delivers outsized power in a small
space. The 40V switch rating and the
wide input range (2.3V to 16V) allow
a wide variety input sources, output
voltages and circuit topologies, unlike
many regulators with restrictive 5V
ratings.
and low sides of the output voltage.
An output tracking function makes the
LTC3709 easy to use in multiple power
supplies applications. The LTC3709
also has a short-circuit shutdown
timer which is easily defeated.
Conclusion
The LTC3709 employs a constant ontime with PLLs and a valley current
control architecture. It has fast transient response, very short minimum
on-time and high efficiency from light
to full load. The LTC3709 is well suited
to high output current, high step-down
ratio applications.
Linear Technology Magazine • March 2005
DESIGN IDEAS
Efficient and Reliable Drive for
Synchronous MOSFET Rectifiers
by Goran Perica
Introduction
Many telecom and industrial applications require low voltage, high
efficiency isolated power converters.
Typical output voltages in these applications are between 1.8V and 12V,
thus making a synchronous forward
converter a good choice.
Synchronous forward converters
require a pair of MOSFETs that rectify
the output from the power transformer.
The synchronous MOSFET rectifiers
can be self-driven, transformer driven,
or driven by an integrated MOSFET
driver. The most efficient solution
is to use a MOSFET driver, like the
LTC3900, that is synchronized to the
primary PWM controller. The LTC3900
has other advantages, such as protection features not found in other drive
methods.
Synchronous
Telecom Bus Converter
Figure 1 shows an example of a
synchronous forward converter that
generates an isolated, semi-regulated
12V output, which in turn is used to
generate all of the non-isolated low
voltage rails on a system board. The
converter in Figure 1 regulates its output by sensing the input voltage and
adjusting its pulse width in order to
maintain constant VIN • TON product.
The constant VIN • TON product results
in a constant output voltage. The only
variations in output voltage in this
type of circuit would be due to circuit
parasitic elements like winding resistance, transformer coupling, MOSFET
resistance and ramp errors.
The VIN • TON product is generated
by the RAMP pin of the LTC3723 in
Figure 1. The input voltage develops
current in resistor R1 that charges
capacitor C1. When the voltage across
capacitor C1 reaches the threshold
of Ramp pin, the output pulse is
terminated. Figure 2 shows the dependency of output voltage on input
voltage and output current. The output
voltage variation is well within the
requirements for a bus converter. The
specification for bus converter allows
the output voltage to be proportional
The main problem with using a bus
converter is keeping the size small
while processing all of the power
required by the system board. Therefore, it is critical to obtain the highest
efficiency in order to keep the power
density high. For example, 90% efficient converter generating 100W of
output power dissipates 11W of heat,
which makes it difficult to keep the
circuit small. In comparison, a 95%
converter dissipates only 5.25W, which
simplifies thermal management, and
thus shrinks the circuit size.
The only way to obtain such high
efficiencies is with synchronous output
rectifiers, as shown in Figure 1. All
of the switching and power handling
components must be optimized in
order to achieve the highest possible
L1
PA1494.242
• •
383k
R1
620k
CIN
2.2µF
x2
49.9k
UVLO
100k
10k
SPRG
DRVB
Q1
Si7450
x2
PR GATE
CS
0.1µF
VREF
1µF
270pF
1nF
523k
1nF
47Ω
0.006Ω
DPRG
CT
1k
BAT760
VREF
SS
0.1µF
GND
4
FG
10k
CG
COUT
47µF, 16V
x4
3
GND
1
CS+
10k
VCC
CS–
2
10k
SYNC TIMER
7
RT
15k
SDRB
CT
1nF
220pF
LTC3723-1
330Ω
1µF
8
COMP
DRVA
5
6
PDZ9.1
Q3
PH4840
x2
VOUT
12V
20A
LTC3900
BCX55
FB
68pF
Q2
Si7370
x2
C1
330pF
RAMP
VCC
1N1418
Efficiency is Everything
PA0962
VIN
36V TO 63V
10V
BIAS
to the input voltage. In other words,
there is no requirement for input voltage regulation. This has also led to the
term DC-Transformer to be used for
bus-converters.
• •
VREF
560Ω
1µF
Q4470-B
Figure 1. This highly efficient bus converter provides 12V at 20A of isolated, semi-regulated power.
The 12V bus can be used by boards to provide low voltage rails via simple, non-isolated converter circuits.
Linear Technology Magazine • March 2005
33
LTC3900 Drive and
Protection Features
MOSFET switching timing in the
circuit of Figure 1 is critical in order to achieve high efficiency. The
LTC3723-2 PWM controller generates the appropriate timing delay
between the main MOSFET DRVB
and synchronous MOSFET SDRB
outputs. The synchronous MOSFET
driver output, DRVB, is pulse coupled
through a small transformer T2. Pulse
coupling of the synchronizing signal
has the benefit of not requiring the DC
levels to be restored on the secondary
side of transformer T2. The LTC3900
sync input was designed to accept
symmetrical bipolar pulses and to
convert these narrow pulses back
into appropriate square wave pulses
for driving the output synchronous
MOSFETs. Another advantage of using pulse coupling is that the coupling
transformer can be very small even at
low switching frequencies.
One of the problems with driving
synchronous MOSFETs has always
97
96
EFFICIENCY (%)
95
94
93
92
91
90
89
88
0
5
10
15
LOAD CURRENT (A)
20
Figure 3. The efficiency of 12V converter is
better than 94% over a wide range of loads.
High efficiency makes it possible to fit this
converter into 35mm by 55mm of space, since
thermal management is simplified.
34
15
15
14
14
OUTPUT VOLTAGE (V)
efficiency. Once all of the components have been optimized, the only
thing remaining is to provide the
converter with precise timing of the
synchronous output rectifiers. The
LTC3900 synchronous output rectifier
controller is not a typical controller. It
provides critical timing and protection
functions that make the converter in
Figure 1 highly reliable and efficient.
Efficiency is as high as 94.5% as shown
in Figure 3.
OUTPUT VOLTAGE (V)
DESIGN IDEAS
13
12
12
11
11
10
13
35
40
55
45
50
INPUT VOLTAGE (V)
60
65
10
0
5
10
15
LOAD CURRENT (A)
20
Figure 2. The output voltage of the circuit in Figure 1 depends on input voltage and output load.
This is a semi-regulated bus supply, intended as input for downstream converters. Even though
the output is inexact, its voltage variance remains well within the limits for a bus supply.
been the last pulse that comes from
the primary PWM controller following
converter shut-down. Depending on
the primary controller, the last edge of
last synchronizing pulse may leave one
of the output MOSFETs turned ON. In
that case, the output capacitor may
drive a huge reverse current through
the output inductor and cause a failure
of one of the MOSFETs.
The MOSFET driver circuit LTC3900
has two functions that protect synchronous MOSFETs following PWM
shut-down. Figure 4 shows a condition
where after the last PWM pulse, the
output inductor current has reversed
to –25A. The LTC3900 in this case
was programmed to turn the forward
rectifier MOSFET OFF in order to prevent further reverse current increase.
Otherwise, the current could increase
far beyond MOSFETs ratings and thus
result in MOSFET failure.
As can be seen in Figure 4, the pulse
width of the last forward gate pulse
can be quite long. The duration of
this pulse can be programmed by the
Timer function of the LTC3900. The
Timer pulse duration doesn’t have to
be any longer than the normal switching period of primary PWM controller.
However, the Timer pulse should not be
any shorter than the longest required
ON time of the synchronous MOSFETs.
If the Timer pulse is too short, the
synchronous MOSFET will be turned
off too soon and the MOSFETs body
diode will have to conduct which will
result in far greater power dissipation. A better Timer pulse duration
than the one shown in Figure 4 is
shown in Figure 5. The Timer dura-
tion programmed in Figure 5 produces
reverse current of only –10A, which is
well within the ratings of synchronous
MOSFET used in this application.
In addition to the Timer function,
the LTC3900 also has a reverse current detector. The reverse current
detector monitors catch MOSFET
Q3 in Figure 1 and terminates the
catch MOSFET conduction if current
through the MOSFET reverses. In
normal operation, the catch MOSFET
current creates a negative drain-tosource voltage. If the inductor current
reverses, the catch MOSFET drainto-source voltage becomes positive.
If the drain-to-source voltage exceeds
a 21mV threshold (10.5mV at CS+
IOUT
10A/DIV
VOUT
5V/DIV
PR GATE
5V/DIV
FG
5V/DIV
10µs/DIV
Figure 4. Output inductor reverse current
following last PWM pulse keeps increasing
until Q3 is turned off. A high reverse current
can damage the MOSFET Q3.
IOUT
10A/DIV
VOUT
5V/DIV
PR GATE
5V/DIV
FG
5V/DIV
10µs/DIV
Figure 5. The reverse inductor current is
reduced with a shorter TIMER period.
Linear Technology Magazine • March 2005
DESIGN IDEAS
input), the catch MOSFET turns OFF.
The 10.5mV current sense threshold of
CS+ input is temperature dependent
so it can follow the temperature variation of the MOSFET’s RDS(ON), as long
as it is mounted close to the MOSFET
on the PC board. To calculate the
amount of reverse current that turns
the catch MOSFET off, use RDS(ON)
at room temperature with a 10.5mV
current sense threshold. Just as with
the Timer function, the catch MOSFET
should not be turned off under normal conditions. Therefore, a catch
MOSFET with sufficiently low RDS(ON)
should be used or voltage divider on
the CS+ pin can be added.
Smart Batteries, continued from page 6
power source. Supply continuity is
paramount—changing from one power
path to another should not interrupt
power to the system. This daunting
task has historically fallen to the host
running custom application software.
The LTC1760 avoids the need for
complicated software development by
operating in a stand-alone Level 3 Bus
Master mode, thus precluding the need
for host intervention. The LTC1760
polls each battery at an accelerated
rate so that it can continuously optimize battery charging and PowerPath
switching modes between two batteries
and a wall adapter. It also has built in
crises power management hardware
to keep the power flowing even if the
SMBus is jammed with traffic.
If there is a feature that will shorten
charge times, lengthen run times or
make the system more robust, it has
been included in the LTC1760:
❑ Proprietary charge algorithms allow parallel charging even for two
batteries of different chemistries
or cell configuration.
❑ Level 3 capabilities allow the
LTC1760 to implement a servo
charge current and charge voltage
system that eliminates hardware
related losses that would extend
charge time (see Figure 7).
❑ A turbo-charge mode maximizes
the charge current for the fastest
battery charging possible.
❑ Support for full dual battery conditioning, another name for gas
gauge calibration for less sophisticated gas gauges.
As sophisticated the LTC1760 is, it
remains easy to use. There are only
four key parameter choices to make.
You can literally drop it into your system, throw some smart batteries and
an AC adapter at it, and it will start
working right away. A full schematic
is shown in Figure 6 (Figure 5 shows
a simplified schematic).
LTC4100 including the SMBus accelerators except it is designed to
work with two batteries (see Figure 5).
Traditionally, dual battery systems
are sequential-discharge systems
designed to simply increase total battery run time. Dual-battery systems
are increasingly used in paralleldischarge systems to satisfy current
requirements beyond the capability
of a sequential, battery 1 then battery 2 discharge priority system. The
LTC1760 addresses the issue by allowing the safe parallel discharge of
two batteries.
It also charges the batteries in
parallel. A parallel-charge, paralleldischarge dual-battery system can
reduce charge times and increase run
times over an equivalent sequential
system1—see Figures 3 and 4.
The key to allowing the LTC1760 to
safely control two batteries in parallel
is the utilization of the ideal diode2
concept where the power MOSFETs
are driven to act like diodes as opposed
to simple on-off switches.
It is no simple feat to safely juggle
the charge and discharge state of
multiple batteries and a DC input
16.9
3140
3120
16.8
ACTUAL
3080
3060
3040
REQUESTED
3020
ACTUAL
16.6
16.5
16.4
3000
16.3
2980
2960
REQUESTED
16.7
VOLTAGE (V)
CURRRENT (mA)
3100
0
20
40
TIME (s)
60
80
16.2
0
50
100
TIME (s)
150
200
Figure 7. The LTC1760’s servo charge current and voltage system eliminates hardware
related inefficiencies, therefore decreasing charge times. This is only one of the many unique
performance-enhancing features in the LT1760.
Linear Technology Magazine • March 2005
Conclusion
The LTC3900 MOSFET is a comprehensive solution for implementing
robust, high efficiency, high performance synchronous converters.
Conclusion
The LTC4100 offers a simple and
reliable Smart Battery System implementation that uses a single battery.
The LTC1760 represents perhaps
the most comprehensive single chip
dual battery system, providing more
control, safety, and automatic crisis
management compared to any other
solution available today. Both parts
offer minimal NRE effort needed to get
up and running as a complete battery
standalone charger system—no battery expertise required. They are also
reduce solution cost, PCB space and
part count.
Notes
1 For a more in depth description of parallel charge,
parallel discharge systems, see Linear Technology
Magazine, December 2001, page 12, “Monolithic
Dual Battery Power Manager Increases Run Time
and Decreases Charge Time”).
2 For more about ideal diodes, see Linear Technology
magazine, December 2002, page 1, “Ideal Diode
Controller Eliminates Energy Wasting Diodes in
Power OR-ing Applications”, or any materials
describing the LTC4412.
35
DESIGN IDEAS
Low-Distortion Sine Wave Oscillator
with Precise RMS Amplitude Stability
by Cheng-Wei Pei
Many applications require a frequency
and/or amplitude-stable sine wave
as a reference for calibration or measurement. Low harmonic distortion is
also required for meaningful results
in applications such as LVDT signal
conditioning, ADC testing, and, of
course, harmonic distortion testing.
Many sine wave generation techniques simply cannot achieve the low
harmonic distortion and amplitude
stability required of a precision sine
wave reference. The technique shown
here generates a sine wave with less
than 0.003% distortion and 0.1%
amplitude stability.
Figure 1 shows a simple oscillator
circuit consisting of a Wien bridge
oscillator core and an amplitude
stabilization loop. The LT1632 highspeed low-distortion amplifier and its
positive feedback RC network generate
the oscillations. The amplitude, and
amplitude stability, of the sine wave
is controlled via a negative feedback
loop comprising an LTC1968 RMS-to-
C
DC converter, an LTC2054 buffer, and
an LT1632 error amplifier.
The oscillation occurs at a frequency
of 1/(2πRC), where R and C are the
positive feedback components of the
amplifier. The attenuation of the negative feedback network is approximately
3, to match the attenuation encoun-
The LTC1968 true RMS-to-DC
converter makes it possible
to achieve 0.1% amplitude
stability in a simple circuit.
tered in the positive feedback network.
The 2N4338 JFET acts as a variable
resistor whose resistance changes
according its the gate-source voltage
bias. Changing the bias of the JFET
adjusts the gain of the oscillator, and
thus the amplitude of the resulting
sine wave signal. The turn-on and
amplitude settling time of this circuit
OUT
f ≈ 1/2πRC
FOR f ≈ 100kHz: C = 1000pF AND R = 1.62k
VOUT(RMS) ≈ 3 • VSET ≈ 3 • VOUT(DC)
VSET
(SET AMPLITUDE)
R
5V
5V
+
7
1/2 LT1632
C
R
6
–
5
VOUT(DC)
–5V
10k
750Ω
1k
10k
5V
+
100k
2N4338
10k
3
1
1k
2k
1µF
LTC1968
0.01µF
10k
4.53k
2
1/2 LT1632
–
11.5k
1µF
1k
1k
+
LTC2054
–
Figure 1. Schematic of the sine wave oscillator, consisting of a low-distortion LT1632-based Wien
bridge oscillator core and an LTC1968-controlled amplitude stabilization loop. The output of the
LTC1968 equals the RMS level of the sine wave divided by 3.
36
are dominated by the settling time of
the LTC1968, which is typically around
1 millisecond with a 0.01µF averaging
capacitor.
The LTC1968 precisely measures
the RMS amplitude of the LT1632’s
output sine wave and gives a DC output
that corresponds to the RMS level of
the sine wave divided by three. The
resistive attenuator at its input allows
the LTC1968 output to remain within
its low-error region of ≤1V for up to
3VRMS output sine waves.
The LTC2054 buffers the output of
the LTC1968 for minimal error due to
output loading, and the LT1632 error
amplifier compares the RMS level of
the sine wave with VSET, which sets the
desired RMS amplitude. The error amp
controls the gate-source voltage bias
of the JFET to modulate the amplitude
accordingly. As shown, the output
amplitude of the sine wave is
VOUT(RMS) = 3 • VSET; with 0V ≤ VSET ≤ 1V
The 10k-11.5k resistive attenuator
at the gate of the JFET compensates
for the channel modulation effects
of the JFET, which otherwise would
cause severe harmonic distortion in
the circuit.
As measured with a Hewlett-Packard 3589A Spectrum Analyzer, the
harmonic distortion of this circuit with
a 100kHz, 1VRMS sine wave output
is –92dBc (0.0025%). The amplitude
stability is better than –60dBc (0.1%).
With a 2VRMS output, the circuit yields
only slightly degraded performance,
at –80dBc (0.01%) harmonic distortion and –55dBc (0.18%) amplitude
stability.
The LTC1968 can measure the amplitude of sine waves up to 500kHz in
frequency with less than 1% absolute
error (independent of the amplitude
stability of the circuit). Producing
higher frequency sine waves using this
circuit is possible, up to the 15MHz
bandwidth of the LTC1968.
Linear Technology Magazine • March 2005
NEW DEVICE CAMEOS
New Device Cameos
16-Bit Quad DAC with
Separate Reference Inputs
The LTC2604 is the latest in Linear
Technology’s family of 16-bit voltage
output DACs that establish excellent
performance standards for output
drive, crosstalk and noise in single
supply, voltage output multiples. High
resolution, low power and small size
make the LTC2604 ideal for portable
instrumentation and industrial process control application. The device’s
guaranteed monotonic performance
is ideal for digital calibration, trim
adjust and level setting applications
in a wide variety of products.
The rail-to-rail DAC output buffers provide excellent drive capability
over a wide supply voltage range of
2.5V to 5.5V. The output can drive
capacitive loads up to 1000pF or a
maximum current load of 15mA while
maintaining excellent linearity within
millivolts of both supply rails. The low
output offset (9mV max) provides a
zero scale voltage much closer to 0V
than competitive parts. Low offset
drift (5µV/°C) and gain error drift (±5
ppm/°C) makes these parts ideal for
use in digital calibration and trim/
adjust applications. The low output
noise reduces the need for additional
output filtering and its 0.1Hz to 10Hz
noise (15µV p-p) is much lower than
competitive devices.
In addition to separate reference
inputs, the LTC2604 also has a Reference Low (REFLO) pin, which allows
the zero scale voltage of the DACs to be
set higher than 0V. Low DC crosstalk
between DACs and separate reference
inputs allow true independent control
of each of the four DAC output levels.
This makes the LTC2604 ideal for “set
and forget” type of applications, as an
update of one DAC output does not
require that the other DAC outputs
be re-adjusted.
The LTC2604 family uses a simple
SPI/MICROWIRE compatible 3-wire
interface that can be operated at a
max clock speed of 50MHz. The daisy
chain capability allows control of
multiple serial devices with a single
Linear Technology Magazine • March 2005
3-wire interface. The asynchronous
clear function required for servo and
control applications is also provided.
The LTC2604 is available in a narrow16-pin SSOP package, and is one
of many pin-compatible devices in a
family of compact DACs, making it
possible to produce a number of performance options from a single design.
This family also includes octal, dual
and single DACs that feature superior
performance in the smallest available
footprints.
Micropower Op Amp has
Precision On-Chip Resistors
The LT1996 is a precision micropower
op amp with eight on-chip precision
resistors packed into a tiny 3mm ×
3mm DFN package. It can be configured to hundreds of applications by
simply strapping its pins—no external
resistors required.
The op amp has an input offset
voltage of only 50µV and the supply
current is 100µA. The outputs swing
to within 40mV of either supply rail,
which is critical in low voltage applications. The gain bandwidth product of
the op amp is 560kHz, and the LT1996
can operate from any supply voltage
between 2.7V and 36V, adding to its
versatility.
The eight on-chip precision resistors have excellent matching of 0.05%
over temperature, and the matching
temperature coefficient is guaranteed
less than 3ppm/°C. The nominal resistor values are 450k, 50k, 16.667k
and 5.555k. When configured as a
difference amplifier, this readily allows
gains of 9, 27 and 81. Hundreds of
other gains (inverting, noninverting
or differential) can be implemented
without any external components,
and all with excellent precision. An
otherwise identical device with different resistor values is also available as
the LT1991.
Some of the resistor inputs can be
taken well beyond the supply rails,
to as high as ±60V. This allows, for
example, precise monitoring of signals
with a negative 48V common mode
while the part is powered by only 5V
and ground.
The LT1996 is specified to operate
from –40°C to 85°C, comes in standard and A grades, and is available
in a 10-lead MSOP as well as a tiny
3mm × 3mm leadless DFN package.
Synchronous Switching
Regulator Controller
Allows Inputs to 60V
The LTC3703-5 is a synchronous
switching regulator controller that can
directly step-down input voltages up
to 60V, and withstand transients up
to 80V, making it ideal for the harsh
environments seen in automotive,
telecom and industrial applications.
The ability to step-down the high
input voltage directly allows a simple
single inductor topology resulting in
a compact high performance power
supply—in contrast to the low-side
drive topologies that require bulky,
expensive transformers.
The LTC3703-5 is similar to its
predecessor, the LTC3703, except that
it is optimized for logic-level MOSFETs
instead of standard-threshold devices.
Like the LTC3703, the LTC3703-5
drives external N-channel MOSFETs
using a constant frequency, voltagemode architecture. A high bandwidth
error amplifier and patented line feed
forward compensation provide very
fast line and load transient response.
Strong 1Ω gate drivers minimize
switching losses even when multiple
MOSFETs are used for high current
applications.
The LTC3703-5 can also operate
as a synchronous step-up regulator
eliminating the bulky catch diode and
associated heat-sinking required in
high current non-synchronous boost
converters. Operating as a boost converter, the LTC3703-5 can regulate
outputs up to 60V.
Other features include an external
clock synchronization input, precise
0.8V 1% reference, programmable
current limit, programmable operating
frequency (100kHz to 600kHz), and
selectable pulse-skip mode operation.
The LTC3703-5 is available in 16pin narrow SSOP and 28-pin SSOP
packages.
37
NEW DEVICE CAMEOS
Low Noise, Synchronous Step- light loads to reduce battery drain. 1µA max shutdown current) makes
Up DC-DC Converter Connects The current mode PWM architecture the LTC2606 ideal for battery-powered
VOUT to VIN in Shutdown
of the LTC3400-1 is internally com- applications. The low output noise (15
The LTC3400-1 is an efficient and
tiny synchronous step-up DC-DC
converter that provides a direct connection from output to input when
shut down. This feature is useful in
products where the battery power
source needs to be monitored while the
converter is turned off, or to provide a
path for backup power from the main
battery.
The LTC3400-1 is pin-for -pin
compatible with the LTC3400 family
of synchronous step-up regulators.
Like the LTC3400 family, it can operate from a single-cell alkaline battery
input up to 4.5V and features power
conversion efficiency up to 95%. The
internal switch and synchronous
rectifier are rated at 600mA (min). The
output voltage can be programmed
from 2.5 to 5.0V with an external
resistor divider. The 1.2MHz fixed
frequency architecture provides very
low VOUT ripple, making it compatible
with sensitive measurement applications, and it allows the use of a tiny,
low profile inductor and ceramic input
and output capacitors.
The LTC3400-1 automatically
switches to Burst Mode operation at
pensated, reducing external parts
count. Shutdown quiescent current
is less than 1µA.
LTC6905, continued from page 15
modulation of the frequency, whether
by voltage or current, relies on modulation of the current input to the RSET
pin. Because the RSET voltage is fixed
at 1V, the frequency of the output
depends to the first degree only on
the current into the RSET pin. The
master oscillator frequency may be
approximated as:
the voltage at the RSET pin at 1V below
the positive supply. Because the frequency of oscillation is based on the
resistance, or V/I at the RSET pin, a
stable V at the RSET pin provides the
ability to generate an accurate output
frequency by injecting an accurate
current (I) at the RSET pin.
For stability reasons, it is recommended that the RSET pin be driven
by resistors as shown in Figure 5. All
V + = 3V
1
0.1µF
RSET
10k
3
RCNTRL
33.2k
+
–
2
V+
OUT
LTC6905
5
fOSC
69.8MHz TO 170MHz
GND
SET
DIV
4
V+
N=1
VCNTRL
0V TO 2V
Figure 5. The LTC6905 as a
voltage controlled oscillator
38
16-Bit DAC with I2C Interface
in a 3mm × 3mm Footprint
The LTC2606 reduces the size and
improves performance of compact
portable products by integrating a
high performance voltage output
16-bit DAC in a 3mm × 3mm 10-pin
DFN package. The LTC2606 is ideal
for space-constrained applications
optimizing board layout. The device’s
guaranteed monotonic performance is
ideal for digital calibration, trim/adjust and level setting applications in
a wide variety of products.
The LTC2606’s output buffer has
excellent drive capability over its
entire 2.7V to 5.5V supply voltage
range. The DAC output can directly
drive capacitive loads up to 1000pF
and current loads up to 15mA while
maintaining good linearity to within
millivolts of both supply rails. The
low output offset (9mV max) provides
a zero-scale voltage closer to 0V than
competitive devices. Low power consumption (270µA supply current and
FOSC =
10kΩ
× 170MHz
RSET
Substituting VRSET/IRSET for RSET,
where VRSET=1V, we get:
FOSC = 10kΩ ×IRSET × 170MHz
This indicates that a 50µA current into the RSET pin would result
in a master oscillator frequency of
85MHz. More applications circuits
and information regarding using the
LTC6905 as a VCO is available in the
data sheet.
µVP–P over 0.1Hz to 10Hz) reduces the
need for output filtering and is much
lower than competitive devices.
The LTC2606 uses a 2-wire I2C
serial digital interface that is compatible with both the standard-mode
(100kHz max clock speed) and fastmode (400kHz max clock speed) of
operation with 27 selectable I2C slave
addresses, which minimizes address
conflicts with other I2C components
in the system. The device features
an asynchronous update pin (LDAC),
which allows the DAC update to be
synchronized to a hardware signal.
It also allows simultaneous updates
and power-up of multiple DACs in
a system. A power-on reset sets the
LTC2606 to zero-scale on power-up.
The LTC2606 is one of many pin
compatible devices in a family of
compact DACs, yielding multiple
performance options from one design.
The LTC2616 and LTC2626 are pin
compatible 14-bit and 12-bit DACs;
the LTC2606-1, LTC2616-1 and the
LTC2626-1 are pin compatible 16-bit,
14-bit and 12-bit DACs which powerup at mid-scale.
The modulation bandwidth of the
LTC6905 is dictated by its internal
control loop, which is limited to between 700kHz and 2MHz, depending
on output frequency. Due to the low
modulation bandwidth in relation to
the output frequency, it is recommended that the LTC6905 be used
as a VCO only in applications where
the rate of modulation is less than the
output frequency divided by 128.
Conclusion
The LTC6905 and LTC6905-XXX are
low power, highly accurate silicon
oscillators that can replace crystals in
many applications. They offer advantages of lower cost, lower sensitivity
to temperature and shock, and ease
of frequency modulation—important
features in driving microcontrollers,
FPGAs and other complex systems.
Linear Technology Magazine • March 2005
DESIGN TOOLS
DESIGN TOOLS
Databooks
The 2004 set of eleven Linear databooks is available
and supersedes all previous Linear databooks. Each
databook contains product data sheets, selection
guides, QML/space information, package information,
appendices, and a complete index to the set.
For more information, or to obtain any of the databooks,
contact your local sales office (see the back of this
magazine), or visit www.linear.com.
Amplifiers (Book 1 of 2) —
• Operational Amplifiers
Amplifiers (Book 2 of 2) —
• Operational Amplifiers
• Instrumentation Amplifiers
• Application Specific Amplifiers
References, Filters, Comparators, Special
Functions, RF & Wireless —
• Voltage References • Special Functions
• Monolithic Filters
• RF & Wireless
• Comparators
• Optical Communications
• Oscillators
Monolithic Switching Regulators —
• Micropower Switching Regulators
• Continuous Switching Regulators
Switching Regulator Controllers (Book 1 of 2) —
• DC/DC Controllers
Switching Regulator Controllers (Book 2 of 2) —
• DC/DC Controllers
• Digital Voltage Programmers
• Off-Line AC/DC Controllers
Linear Regulators, Charge Pumps,
Battery Chargers —
• Linear Regulators
• Charge Pump DC/DC Converters
• Battery Charging & Management
Hot Swap Controllers, MOSFET Drivers, Special
Power Functions —
• Hot Swap Controllers
• Power Switching & MOSFET Drivers
• PCMCIA Power Controllers
• CCFL Backlight Converters
• Special Power Functions
Data Converters (Book 1 of 2) —
• Analog-to-Digital Converters
Data Converters (Book 2 of 2) —
• Analog-to-Digital Converters
• Digital-to-Analog Converters
• Switches & Multiplexers
Interface, System Monitoring & Control —
• Interface — RS232/562, RS485,
Mixed Protocol, SMBus/I2C
• System Monitoring & Control — Supervisors,
Margining, Sequencing & Tracking Controllers
Information furnished herein by Linear Technology Corporation is believed to be accurate and reliable. However,
no responsibility is assumed for its use. Linear Technology
Corporation makes no representation that the interconnection of its circuits, as described herein, will not infringe on
existing patent rights.
Linear Technology Magazine • March 2005
www.linear.com
Brochures
Customers can quickly and conveniently find and retrieve
product information and solutions to their applications.
Located at www.linear.com., the site quickly searches our
database of technical documents and displays weighted
results of our data sheets, application notes, design
notes, Linear Technology magazine issues and other
LTC publications. The LTC website simplifies the product selection process by providing convenient search
methods, complete application solutions and design
simulation programs for Power, Filter, Op Amp and Data
Converter applications. Search methods include a text
search for a particular part number, keyword or phrase.
And the most powerful, a parametric search engine. After
selecting a desired product category, engineers can
specify and sort by key parameters and specifications
that satisfy their design requirements.
Power Management & Wireless Solutions for Handheld
Products — The solutions in this product selection guide
solve real-life problems for cell phones, digital cameras,
PDAs and other portable devices. Circuits are shown for
Li-Ion battery chargers, battery managers, USB support,
system power regulation, display drivers, white LED
drivers, photoflash chargers, DC/DC converters, SIM
and smart card interfaces, photoflash chargers, and RF
PA power supply and control. All solutions are designed
to maximize battery run time, save space and reduce
EMI where necessary—important considerations when
designing circuits for handheld devices.
Purchase Products Online
Credit Card Purchases—Purchase online direct from
Linear Technology at www.linear.com using a credit card.
Create a personalized account to check order history,
shipment information and reorder products.
Linear Express Distribution — Get the parts you need.
Fast. Most devices are stocked for immediate delivery.
Credit terms and low minimum orders make it easy to get
you up and running. Place and track orders online. Apply
today at www.linear.com or call (866) 546-3271.
Applications Handbooks
Linear Applications Handbook, Volume I — Almost a
thousand pages of application ideas covered in depth by
40 Application Notes and 33 Design Notes. This catalog
covers a broad range of real world linear circuitry. In
addition to detailed, systems-oriented circuits, this
handbook contains broad tutorial content together with
liberal use of schematics and scope photography. A
special feature in this edition includes a 22-page section
on SPICE macromodels.
Linear Applications Handbook, Volume II — Continues
the stream of real world linear circuitry initiated by Volume
I. Similar in scope to Volume I, this book covers Application Notes 40 through 54 and Design Notes 33 through
69. References and articles from non-LTC publications
that we have found useful are also included.
Linear Applications Handbook, Volume III —
This 976-page handbook includes Application Notes 55
through 69 and Design Notes 70 through 144. Subjects
include switching regulators, measurement and control
circuits, filters, video designs, interface, data converters,
power products, battery chargers and CCFL inverters.
An extensive subject index references circuits in Linear
data sheets, design notes, application notes and Linear
Technology magazines.
CD-ROM
The March 2005 CD-ROM contains product data sheets,
application notes and Design Notes released through
February of 2005. Use your browser to view product
categories and select products from parametric tables
or simply choose products and documents from part
number, application note or design note indexes.
Automotive Electronic Solutions— This selection guide
features recommended Linear Technology solutions for
a wide range of functions commonly used in today’s
automobiles, including telematics and infotainment
systems, body electronics and engine management,
safety systems and GPS/navigation systems.
Linear Technology’s high-performance analog ICs
provide efficient, compact and dependable solutions
to solve many automotive application requirements.
Software
SwitcherCAD™ III/LTC SPICE — LTC SwitcherCAD III is
a fully functional SPICE simulator with enhancements
and models to ease the simulation of switching regulators. This SPICE is a high performance circuit simulator
and integrated waveform viewer, and also includes
schematic capture. Our enhancements to SPICE result
in much faster simulation of switching regulators than is
possible with normal SPICE simulators. SwitcherCAD III
includes SPICE, macromodels for 80% of LTC’s switching
regulators and over 200 op amp models. It also includes
models of resistors, transistors and MOSFETs. With this
SPICE simulator, most switching regulator waveforms
can be viewed in a few minutes on a high performance
PC. Circuits using op amps and transistors can also be
easily simulated. Download at www.linear.com
FilterCAD™ 3.0 — FilterCAD 3.0 is a computer aided design program for creating filters with Linear Technology’s
filter ICs. FilterCAD is designed to help users without
special expertise in filter design to design good filters
with a minimum of effort. It can also help experienced
filter designers achieve better results by playing “what if”
with the configuration and values of various components
and observing the results. With FCAD, you can design
lowpass, highpass, bandpass or notch filters with a
variety of responses, including Butterworth, Bessel,
Chebychev, elliptic and minimum Q elliptic, plus custom
responses. Download at www.linear.com
SPICE Macromodel Library — This library includes LTC
op amp SPICE macromodels. The models can be used
with any version of SPICE for analog circuit simulations.
These models run on SwitcherCAD III/LTC SPICE.
Noise Program — This PC program allows the user to
calculate circuit noise using LTC op amps, determine
the best LTC op amp for a low noise application, display
the noise data for LTC op amps, calculate resistor noise
and calculate noise using specs for any op amp.
39
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© 2005 Linear Technology Corporation/Printed in U.S.A./30K
Linear Technology Magazine • March 2005
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