ADP1610 1.2 MHz DC-to-DC Step-Up Switching
1.2 MHz DC-to-DC Step-Up Switching
Converter
ADP1610
FEATURES
GENERAL DESCRIPTION
Fully integrated 1.2 A, 0.2 Ω, power switch
Pin-selectable 700 kHz or 1.2 MHz PWM frequency
92% efficiency
Adjustable output voltage up to 12 V
3% output regulation accuracy
Adjustable soft start
Input undervoltage lockout
MSOP 8-lead package
The ADP1610 is a dc-to-dc step-up switching converter with
an integrated 1.2 A, 0.2 Ω power switch capable of providing
an output voltage as high as 12 V. With a package height of less
than 1.1 mm, the ADP1610 is optimal for space-constrained
applications such as portable devices or thin film transistor
(TFT) liquid crystal displays (LCDs).
The ADP1610 operates in pulse-width modulation (PWM) current
mode with up to 92% efficiency. Adjustable soft start prevents
inrush currents at startup. The pin-selectable switching frequency
and PWM current mode architecture allow excellent transient
response, easy noise filtering, and the use of small, cost-saving
external inductors and capacitors.
APPLICATIONS
TFT LC bias supplies
Portable applications
Industrial/instrumentation equipment
The ADP1610 is offered in the RoHS compliant 8-lead MSOP
and operates over the temperature range of −40°C to +85°C.
FUNCTIONAL BLOCK DIAGRAM
REF
COMP
IN
1
6
ERROR
AMP
ADP1610
gm
BIAS
FB 2
5
F/F
R Q
S
RAMP
GEN
COMPARATOR
SS 8
DRIVER
OSC
SOFT START
CURRENT
SENSE
AMPLIFIER
SD 3
4
GND
04472-001
RT 7
SW
Figure 1.
Rev. A
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ADP1610* PRODUCT PAGE QUICK LINKS
Last Content Update: 02/23/2017
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ADP1610
TABLE OF CONTENTS
Features .............................................................................................. 1
Setting the Output Voltage ........................................................ 10
Applications ....................................................................................... 1
Inductor Selection ...................................................................... 10
General Description ......................................................................... 1
Choosing the Input and Output Capacitors ........................... 11
Functional Block Diagram .............................................................. 1
Diode Selection........................................................................... 13
Revision History ............................................................................... 2
Loop Compensation .................................................................. 13
Specifications..................................................................................... 3
Soft Start Capacitor .................................................................... 14
Absolute Maximum Ratings............................................................ 4
Application Circuits ................................................................... 15
ESD Caution .................................................................................. 4
Pin Configuration and Function Descriptions ............................. 5
DC-to-DC Step-Up Switching Converter with True
Shutdown ..................................................................................... 15
Typical Performance Characteristics ............................................. 6
TFT LCD Bias Supply ................................................................ 15
Theory of Operation ...................................................................... 10
Sepic Power Supply .................................................................... 16
Current Mode PWM Operation ............................................... 10
Layout Procedure ........................................................................... 17
Frequency Selection ................................................................... 10
Outline Dimensions ....................................................................... 18
Soft Start ...................................................................................... 10
Ordering Guide .......................................................................... 18
On/Off Control ........................................................................... 10
REVISION HISTORY
9/08—Rev. 0 to Rev. A
Changes to Table 4 .......................................................................... 11
Changes to Table 5 .......................................................................... 14
10/04—Revision 0: Initial Version
Rev. A | Page 2 of 20
ADP1610
SPECIFICATIONS
VIN = 3.3 V, TA = −40°C to +85°C, unless otherwise noted. All limits at temperature extremes are guaranteed by correlation and
characterization using standard statistical quality control (SQC), unless otherwise noted.
Table 1.
Parameter
SUPPLY
Input Voltage
Quiescent Current
Nonswitching State
Shutdown
Symbol
IQ
IQSD
VFB = 1.3 V, RT = VIN
VSD = 0 V
Switching State 1
OUTPUT
Output Voltage
Load Regulation
Overall Regulation
REFERENCE
Feedback Voltage
Line Regulation
ERROR AMPLIFIER
Transconductance
Voltage Gain
FB Input Bias Current
SWITCH
SW On Resistance
SW Leakage Current
Peak Current Limit 2
OSCILLATOR
Oscillator Frequency
IQSW
FSW = 1.23 MHz, no load
Maximum Duty Cycle
SHUTDOWN
Shutdown Input Voltage Low
Shutdown Input Voltage High
Shutdown Input Bias Current
SOFT START
SS Charging Current
UNDERVOLTAGE LOCKOUT 3
UVLO Threshold
UVLO Hysteresis
1
2
3
Conditions
VIN
Min
VFB
VIN = 2.5 V to 5.5 V
5.5
V
390
0.01
600
10
μA
μA
1
2
mA
12
V
mV/mA
%
1.248
+0.15
V
%/V
0.05
±3
1.212
−0.15
ΔI = 1 μA
ISW = 1.0 A
VSW = 12 V
ICLSET
DMAX
RT = GND
RT = IN
COMP = open, VFB = 1 V, RT = GND
0.49
0.89
78
VIL
VIH
ISD
Nonswitching state
Switching state
VSD = 3.3 V
2.2
This parameter specifies the average current while switching internally and with SW (Pin 5) floating.
Guaranteed by design and not fully production tested.
Guaranteed by characterization.
Rev. A | Page 3 of 20
μA/V
dB
nA
200
0.01
2.0
400
20
mΩ
μA
A
0.7
1.23
83
0.885
1.6
90
MHz
MHz
%
0.6
V
V
μA
0.01
VSS = 0 V
VIN rising
1.230
100
60
10
VFB = 1.23 V
fOSC
Unit
VIN
ILOAD = 10 mA to 150 mA, VOUT = 10 V
Line, load, temperature
RON
Max
2.5
VOUT
gm
AV
Typ
1
3
2.2
2.4
220
μA
2.5
V
mV
ADP1610
ABSOLUTE MAXIMUM RATINGS
ESD CAUTION
Table 2.
Parameter
IN, COMP, SD, SS, RT, FB to GND
SW to GND
RMS SW Pin Current
Operating Ambient Temperature Range
Operating Junction Temperature Range
Storage Temperature Range
θJA, Two Layers
θJA, Four Layers
Lead Temperature (Soldering, 60 sec)
Rating
−0.3 V to +6 V
14 V
1.2 A
−40°C to +85°C
−40°C to +125°C
−65°C to +150°C
206°C/W
142°C/W
300°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
IN
RC
CC
VOUT
1
COMP
6
ERROR
AMP
R1
REF
L1
ADP1610
BIAS
gm
FB
CIN
IN
2
R2
SW
5
F/F
VIN
1.2MHz
S
DRIVER
COMPARATOR
RT
7
VOUT
COUT
R Q
RAMP
GEN
D1
OSC
700kHz
SD 3
CSS
8
SOFT START
CURRENT
SENSE
AMPLIFIER
4
GND
Figure 2. Block Diagram and Typical Application Circuit
Rev. A | Page 4 of 20
04472-002
SS
ADP1610
COMP 1
FB 2
SD 3
GND 4
ADP1610
TOP VIEW
(Not to Scale)
8
SS
7
RT
6
IN
5
SW
04472-003
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 3. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
Mnemonic
COMP
2
FB
3
4
5
SD
GND
SW
6
IN
7
RT
8
SS
Description
Compensation Input. Connect a series resistor-capacitor network from COMP to GND to compensate the
regulator.
Output Voltage Feedback Input. Connect a resistive voltage divider from the output voltage to FB to set the
regulator output voltage.
Shutdown Input. Drive SD low to shut down the regulator; drive SD high to turn it on.
Ground.
Switching Output. Connect the power inductor from the input voltage to SW and connect the external rectifier
from SW to the output voltage to complete the step-up converter.
Main Power Supply Input. IN powers the ADP1610 internal circuitry. Connect IN to the input source voltage.
Bypass IN to GND with a 10 μF or greater capacitor as close to the ADP1610 as possible.
Frequency Setting Input. RT controls the switching frequency. Connect RT to GND to program the oscillator to 700 kHz,
or connect RT to IN to program the oscillator to 1.2 MHz.
Soft Start Timing Capacitor Input. A capacitor from SS to GND brings up the output slowly at power-up.
Rev. A | Page 5 of 20
ADP1610
TYPICAL PERFORMANCE CHARACTERISTICS
100
90
VIN = 2.5V
VIN = 5.5V
VIN = 3.3V
VIN = 2.5V
80
EFFICIENCY (%)
70
VOUT = 7.5V
FSW = 1.2MHz
L = 4.7µH
90
VIN = 3.3V
80
EFFICIENCY (%)
100
VIN = 5.5V
VOUT = 10V
FSW = 700kHz
L = 10µH
60
50
40
30
70
60
50
20
1
10
100
LOAD CURRENT (mA)
04472-008
30
1k
1
Figure 4. Output Efficiency vs. Load Current
100
90
CURRENT LIMIT (A)
VIN = 2.5V
70
60
50
40
30
20
04472-006
1
10
100
LOAD CURRENT (mA)
VIN = 3.3V
1.8
VIN = 2.5V
1.6
–15
10
35
AMBIENT TEMPERATURE (°C)
60
85
Figure 8. Current Limit vs. Ambient Temperature, VOUT = 10 V
1.4
VOUT = 7.5V
FSW = 700kHz
L = 10µH
90
VIN = 5.5V
OSCILLATORY FREQUENCY (MHz)
100
VIN = 3.3V
VIN = 2.5V
80
70
60
50
40
04472-007
EFFICIENCY (%)
2.0
1.2
–40
1K
Figure 5. Output Efficiency vs. Load Current
30
VIN = 5.5V
1.4
10
0
2.2
04472-009
EFFICIENCY (%)
2.4
VIN = 3.3V
80
1k
Figure 7. Output Efficiency vs. Load Current
VIN = 5.5V
VOUT = 10V
FSW = 1.2MHz
L = 4.7µH
10
100
LOAD CURRENT (mA)
1
10
100
LOAD CURRENT (mA)
1.2
1.0
0.8
0.6
RT = GND
0.4
0.2
0
–40
1k
RT = VIN
VOUT = 10V
VIN = 3.3V
–15
04472-010
0
40
04472-005
10
10
35
AMBIENT TEMPERATURE (°C)
60
Figure 9. Oscillatory Frequency vs. Ambient Temperature
Figure 6. Output Efficiency vs. Load Current
Rev. A | Page 6 of 20
85
ADP1610
0.50
FSW = 700kHz
VFB = 1.3V
1.2
0.45
QUIESCENT CURRENT (mA)
RT = VIN
1.0
0.8
0.6
RT = GND
0.4
VOUT = 10V
3.0
3.5
4.0
4.5
SUPPLY VOLTAGE (V)
5.0
0.35
VIN = 3.3V
0.30
VIN = 2.5V
0.20
–40
5.5
Figure 10. Oscillatory Frequency vs. Supply Voltage
60
85
FSW = 1.23kHz
VFB = 1.3V
VIN = 2.5V
QUIESCENT CURRENT (mA)
0.55
250
VIN = 3.3V
200
VIN = 5.5V
150
100
0.50
VIN = 5.5V
0.45
VIN = 3.3V
0.40
VIN = 2.5V
04472-012
10
35
AMBIENT TEMPERATURE (°C)
60
0.30
–40
85
Figure 11. Switch Resistance vs. Ambient Temperature
1.24
1.8
SUPPLY CURRENT (mA)
2.0
1.235
1.23
1.225
1.22
5
20
35
50
65
80
AMBIENT TEMPERATURE (°C)
95
110
60
85
FSW = 1.23kHz
VFB = 1V
1.6
VIN = 5.5V
1.4
1.2
VIN = 3.3V
1.0
VIN = 2.5V
0.8
04472-013
–25 –10
10
35
AMBIENT TEMPERATURE (°C)
Figure 14. Quiescent Current vs. Ambient Temperature
1.245
1.215
–15
0.6
–40
125
–15
10
35
AMBIENT TEMPERATURE (°C)
60
Figure 15. Supply Current vs. Ambient Temperature
Figure 12. FB Regulation Voltage vs. Ambient Temperature
Rev. A | Page 7 of 20
04472-016
–15
04472-015
0.35
50
1.21
–40
10
35
AMBIENT TEMPERATURE (°C)
0.60
300
0
–40
–15
Figure 13. Quiescent Current vs. Ambient Temperature
350
SWITCH RESISTANCE (mΩ)
VIN = 5.5V
04472-014
0.2
0
2.5
FB REGULATION VOLTAGE (V)
0.40
0.25
04472-011
OSCILLATORY FREQUENCY (MHz)
1.4
85
ADP1610
1.4
CH1 = IL 200mA/DIV
CH2 = VSW 5V/DIV
FSW = 700kHz
VFB = 1V
1.3
VIN = 3.3V
VOUT = 10V
ILOAD = 20mA
FSW = 700kHz
L = 10µH
SUPPLY CURRENT (mA)
1.2
VIN = 5.5V
1.1
1.0
2
0.9
0.8
VIN = 3.3V
0.7
VIN = 2.5V
–15
10
35
AMBIENT TEMPERATURE (°C)
60
04472-020
0.5
0.4
–40
1
04472-017
0.6
CH1 10.0mVΩ CH2 5.00V
85
M400ns
A CH2
T
136.000ns
10.0V
Figure 19. Switching Waveform in Discontinuous Conduction
Figure 16. Supply Current vs. Ambient Temperature
3.5
VIN = 3.3V
SD = 0.4V
3.0
VIN = 3.3V, VOUT = 10V
COUT = 10μF, L = 10μH, RC = 130Ω
CC = 270pF, FSW = 700kHz
CH1 = VOUT, 200mV/DIV
CH2 = IOUT, 200mA/DIV
SUPPLY CURRENT (µA)
2.5
1
2.0
1.5
1.0
15
70
AMBIENT TEMPERATURE (°C)
125
CH1 200mV
CH2 10.0mVΩ M200μs
A CH2
7.60mV
Figure 20. Load Transient Response, 700 kHz , VOUT = 10 V
Figure 17. Supply Current in Shutdown vs. Ambient Temperature
CH1 = IL 500mA/DIV
CH2 = VSW 5V/DIV
04472-021
0
–40
2
04472-018
0.5
VIN = 3.3V
VOUT = 10V
ILOAD = 200mA
FSW = 700kHz
L = 10μH
VIN = 3.3V, VOUT = 10V
COUT = 10µF, L = 4.7µH, RC = 220kΩ
CC = 150pF, FSW = 1.2MHz
CH1 = VOUT, 200mV/DIV
CH2 = IOUT, 200mA/DIV
1
2
04472-019
CH1 10.0mVΩ CH2 5.00V
M400ns
A CH2
T
136.000ns
2
04472-022
1
10.0V
CH1 200mV
CH2 10.0mVΩ M200µs
A CH2
7.60mV
Figure 21. Load Transient Response, 1.2 MHz, VOUT = 10 V
Figure 18. Switching Waveform in Continuous Conduction
Rev. A | Page 8 of 20
ADP1610
2
2
4
4
VIN = 3.3V
VOUT = 10V
IOUT = 0.2A
CSS = 0nF
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
M100μs
T
A CH2
680mV
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
414.800μs
2
4
4
VIN = 3.3V
VOUT = 10V
IOUT = 0.2A
CSS = 10nF
3
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
M100μs
T
A CH2
A CH2
1.72V
405.600μs
CH1 = IL 1A/DIV
CH2 = SD
CH3 = VOUT
CH4 = SW, FSW = 700kHz
IOUT = 0.2A
VIN = 3.3V
VOUT = 10V
CSS = 10nF
1
04472-024
1
VIN = 3.3V
VOUT = 10V
IOUT = 0.2A
CSS = 0nF
Figure 24. Start-Up Response from Shutdown, SS = 0 nF
2
CH1 = IL 1A/DIV
CH2 = VIN
CH3 = VOUT
CH4 = SW, FSW = 700kHz
M100μs
T
Figure 22. Start-Up Response from VIN, SS = 0 nF
3
CH1 = IL 1A/DIV
CH2 = VIN
CH3 = VOUT
CH4 = SW, FSW = 700kHz
1
04472-023
1
3
04472-025
CH1 = IL 1A/DIV
CH2 = VIN
CH3 = VOUT
CH4 = SW, FSW = 700kHz
04472-026
3
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
680mV
M100μs
T
414.800μs
Figure 23. Start-Up Response from VIN, SS = 10 nF
A CH2
1.72V
405.600μs
Figure 25. Start-Up Response from Shutdown, SS = 10 nF
Rev. A | Page 9 of 20
ADP1610
THEORY OF OPERATION
The ADP1610 current mode step-up switching converter converts
a 2.5 V to 5.5 V input voltage up to an output voltage as high as
12 V. The 1.2 A internal switch allows a high output current, and
the high 1.2 MHz switching frequency allows tiny external
components. The switch current is monitored on a pulse-bypulse basis to limit it to 2 A.
CURRENT MODE PWM OPERATION
The ADP1610 uses current mode architecture to regulate the
output voltage. The output voltage is monitored at FB through
a resistive voltage divider. The voltage at FB is compared to the
internal 1.23 V reference by the internal transconductance error
amplifier to create an error current at COMP. A series resistor
capacitor at COMP converts the error current to a voltage. The
switch current is internally measured and added to the stabilizing
ramp. The resulting sum is compared to the error voltage at
COMP to control the PWM modulator. This current mode
regulation system allows fast transient response, while maintaining a stable output voltage. By selecting the proper resistor
capacitor network from COMP to GND, the regulator response
is optimized for a wide range of input voltages, output voltages,
and load conditions.
FREQUENCY SELECTION
The frequency of the ADP1610 is selectable to operate at either
700 kHz to optimize the regulator for high efficiency or at 1.2 MHz
for small external components. Connect RT to IN for 1.2 MHz
operation, or connect RT to GND for 700 kHz operation. To
achieve the maximum duty cycle, which may be required for
converting a low input voltage to a high output voltage, use the
lower 700 kHz switching frequency.
SOFT START
To prevent input inrush current at startup, connect a capacitor
from SS to GND to set the soft start period. When the ADP1610
is in shutdown (SD is at GND) or the input voltage is below the
2.4 V undervoltage lockout voltage, SS is internally shorted to
GND to discharge the soft start capacitor. When the ADP1610
is turned on, SS sources 3 μA to the soft start capacitor at
startup. As the soft start capacitor charges, it limits the voltage
at COMP. Because of the current mode regulator, the voltage at
COMP is proportional to the switch peak current, and,
therefore, the input current. By slowly charging the soft start
capacitor, the input current ramps slowly to prevent the current
from overshooting excessively at startup.
ON/OFF CONTROL
The SD input turns the ADP1610 regulator on or off. Drive SD
low to turn off the regulator and reduce the input current to 10 nA.
Drive SD high to turn on the regulator.
When the dc-to-dc step-up switching converter is turned off,
there is a dc path from the input to the output through the
inductor and output rectifier. This causes the output voltage
to remain slightly below the input voltage by the forward
voltage of the rectifier, preventing the output voltage from
dropping to zero when the regulator is shut down. Figure 28
shows the application circuit to disconnect the output voltage
from the input voltage at shutdown.
SETTING THE OUTPUT VOLTAGE
The ADP1610 features an adjustable output voltage range of VIN
to 12 V. The output voltage is set by the resistive voltage divider
(R1 and R2 in Figure 2) from the output voltage (VOUT) to the
1.230 V feedback input at FB. To calculate the output voltage
use the following equation:
VOUT = 1.23 × (1 + R1/R2)
(1)
To prevent output voltage errors due to the 10 nA FB input bias
current, an R2 resistance of 10 kΩ or less is to be used. Choose
R1 based on the following equation:
− 1.23 ⎞
⎛V
R1 = R2 × ⎜ OUT
⎟
1.23
⎝
⎠
(2)
INDUCTOR SELECTION
The inductor is an essential part of the step-up switching converter.
It stores energy during the on-time, and transfers that energy to
the output through the output rectifier during the off-time. Use
inductance in the range of 1 μH to 22 μH. In general, lower
inductance values have higher saturation current and lower series
resistance for a given physical size. However, lower inductance
results in higher peak current that can lead to reduced efficiency
and greater input and/or output ripple and noise. A peak-to-peak
inductor ripple current at approximately 30% of the maximum dc
input current typically yields an optimal compromise.
For determining the inductor ripple current, the input (VIN) and
output (VOUT) voltages determine the switch duty cycle (D) by
the following equation:
Rev. A | Page 10 of 20
D=
VOUT − VIN
VOUT
(3)
ADP1610
Table 4. Recommended Capacitor and Inductor Manufacturers
Vendor
Sumida
Part
CMD4D11-2R2MC
CMD4D11-4R7MC
CDRH4D28-100
CDRH5D18-220
CR43-4R7
CR43-100
DS1608-472
DS1608-103
D52LC-4R7M
D52LC-100M
GRM32ER61A226KE20L
Coilcraft
Toko
Murata
L (μH)
2.2
4.7
10
22
4.7
10
4.7
10
4.7
10
C (μF)/Rating (V)
22/10
GRM31CR61E106KA12L
10/25
GRM31CR61A106KA01L
10/10
GRM31CR71E475KA88L
4.7/25
D
tON =
f SW
(4)
The inductor ripple current (ΔIL) in steady state is calculated
using the following equation:
VIN × t ON
L
(5)
The inductance value (L) is calculated using the following
equation:
L=
VIN × t ON
(6)
ΔI L
The peak inductor current (the maximum input current plus
half the inductor ripple current) must be less than the rated
saturation current of the inductor. Likewise, the maximum
rated rms current of the inductor must be greater than the
maximum dc input current to the regulator.
For duty cycles greater than 50%, which occur with input
voltages greater than one-half the output voltage, slope compensation is required to maintain stability of the current-mode regulator.
For stable current-mode operation, ensure that the selected
inductance is equal to or greater than LMIN using the following
equation:
L > L MIN =
VOUT − VIN
1.8 A × f SW
Maximum
DCR (mΩ)
116
216
128
290
109
182
60
75
87
150
Height (mm)
1.2
1.2
3.0
2.0
3.5
3.5
2.9
2.9
2.0
2.0
Comments
Used only for 3.3 V
input and 5 V output
Used for 9 V, 10 V,
and 12 V outputs
Used for inputs and
outputs of 5 V
Using the duty cycle and switching frequency (FSW), the on-time
is calculated using the following equation:
ΔIL =
Maximum
DC Current
0.95
0.75
1.00
0.80
1.15
1.04
1.40
1.00
1.14
0.76
CHOOSING THE INPUT AND OUTPUT CAPACITORS
The ADP1610 requires input and output bypass capacitors to
supply transient currents while maintaining constant input and
output voltage. To prevent noise at the ADP1610 input, a low
ESR (equivalent series resistance), 10 μF or greater input capacitor
is to be used. Place the capacitor between IN and GND as close to
the ADP1610 as possible. Ceramic capacitors are preferred because
of their low ESR characteristics. Alternatively, use a high value,
medium ESR capacitor in parallel with a 0.1 μF low ESR capacitor
as close to the ADP1610 as possible.
The output capacitor maintains the output voltage and supplies
current to the load while the ADP1610 switch is on. The value
and characteristics of the output capacitor greatly affect the
output voltage ripple and stability of the regulator. Use a low
ESR output capacitor; ceramic dielectric capacitors are preferred.
The input bypass capacitor limits the high frequency impedance
of the input voltage source. The value of this capacitor may
need to be increased if the wire length from the input voltage
source to the applications board is longer than 18” to optimize
the transient performance.
(7)
Rev. A | Page 11 of 20
ADP1610
For very low ESR capacitors, such as ceramic capacitors, the ripple
current due to the capacitance is calculated using the following
equation. Because the capacitor discharges during the on-time
(tON), the charge removed from the capacitor (QC) is the load
current multiplied by the on-time. The output voltage ripple
(ΔVOUT) is
ΔVOUT
Q
I ×t
= C = L ON
COUT
COUT
(8)
where:
COUT is the output capacitance.
IL is the average inductor current.
t ON =
D
FSW
Choose the output capacitor based on the following equation:
C OUT ≥
The following are recommended capacitor manufacturers:
•
•
•
•
•
•
(9)
and
D=
VOUT − VIN
VOUT
I L × (VOUT − V IN )
FSW × VOUT × ΔVOUT
(10)
Rev. A | Page 12 of 20
AVX
Kyocera
Murata
Sanyo
Taiyo–Yuden
TDK
(11)
ADP1610
The regulator loop gain is calculated using the following
equation:
DIODE SELECTION
The output rectifier conducts the inductor current to the
output capacitor and load while the switch is off. For high
efficiency, minimize the forward voltage drop of the diode.
For this reason, Schottky rectifiers are recommended. The
following are recommended Schottky diode manufacturers:
•
•
AVL =
ON Semiconductor
Diodes, Inc.
However, for high voltage, high temperature applications, where
the Schottky rectifier reverse leakage current becomes significant
and can degrade efficiency, use an ultrafast junction diode.
Make sure that the diode is rated to handle the average output
load current. Many diode manufacturers derate the current
capability of the diode as a function of the duty cycle. Verify
that the output diode is rated to handle the average output load
current with the minimum duty cycle. The minimum duty cycle
of the ADP1610 is calculated by the following equations:
D MIN =
VOUT − VIN − MAX
VOUT
(12)
where VIN-MAX is the maximum input voltage.
LOOP COMPENSATION
The step-up converter produces an undesirable right-half plane
zero in the regulation feedback loop. This requires compensating
the regulator such that the crossover frequency occurs well below
the frequency of the right-half plane zero. The right-half plane
zero is determined by the following equation:
⎛ V
FZ (RHP ) = ⎜⎜ IN
⎝ VOUT
To determine the crossover frequency, it is important to note
that at that frequency the compensation impedance (ZCOMP) is
dominated by the resistor, and the output impedance (ZOUT) is
dominated by the impedance of the output capacitor. Therefore,
when solving for the crossover frequency, the equation (by
definition of the crossover frequency) is simplified to
V
VFB
1
× IN × G MEA × RCOMP × GCS ×
= 1 (15)
2π × f C × COUT
VOUT VOUT
where:
fC is the crossover frequency.
RCOMP is the compensation resistor.
RCOMP is calculated using the following equation:
RCOMP =
2
⎞ RLOAD
⎟ ×
⎟ 2π × L
⎠
(13)
where:
FZ(RHP) is the right-half plane zero.
RLOAD is the equivalent load resistance or the output voltage
divided by the load current.
2π × f C × C OUT × VOUT × VOUT
VFB × V IN × G MEA × GCS
(16)
VFB = 1.23, GMEA = 100 μS, and GCS = 2 S is calculated using the
following equation:
RCOMP =
To stabilize the regulator, make sure that the regulator crossover
frequency is less than or equal to one-fifth of the right-half plane
zero and less than or equal to one-fifteenth of the switching
frequency.
(14)
where:
AVL is the loop gain.
VFB is the feedback regulation voltage, 1.230 V.
VOUT is the regulated output voltage.
VIN is the input voltage.
GMEA is the error amplifier transconductance gain.
ZCOMP is the impedance of the series RC network from COMP
to GND.
GCS is the current sense transconductance gain (the inductor
current divided by the voltage at COMP), which is internally
set by the ADP1610.
ZOUT is the impedance of the load and output capacitor.
AVL =
The ADP1610 uses external components to compensate the
regulator loop, allowing optimization of the loop dynamics
for a given application.
V
VFB
× IN × G MEA × Z COMP × GCS × Z OUT
VOUT VOUT
2.55 × 10 4 × f C × COUT × VOUT × VOUT
VIN
(17)
When the compensation resistor is known, the zero formed
by the compensation capacitor and resistor is to be set to onefourth of the crossover frequency using the following equation:
CCOMP =
2
π × f C × RCOMP
where CCOMP is the compensation capacitor.
Rev. A | Page 13 of 20
(18)
ADP1610
Table 5. Recommended External Components for Popular Input/Output Voltage Conditions
VIN (V)
3.3
5
VOUT (V)
5
5
9
9
10
10
12
12
9
9
10
10
12
12
FSW (MHz)
0.700
1.23
0.700
1.23
0.700
1.23
0.700
1.23
0.700
1.23
0.700
1.23
0.700
1.23
L (μH)
4.7
2.7
10
4.7
10
4.7
10
4.7
10
4.7
10
4.7
10
4.7
COUT (μF)
22
10
10
4.7
10
4.7
10
10
10
4.7
10
4.7
10
4.7
CIN (μF)
22
22
10
10
10
10
10
10
10
10
10
10
10
10
gm
FB 2
COMP
1
RC
C2
CC
04472-004
The capacitor (C2) is chosen to cancel the zero introduced by
output capacitance ESR. C2 is calculated using the following
equation:
ESR × C OUT
RCOMP
RCOMP (kΩ)
68
100
20
47
33
39
33
75
24
47
33
51
33
75
Ccomp (pF)
150
100
680
150
470
220
470
150
470
150
330
220
330
150
IOUT_MAX (mA)
600
600
350
350
300
300
250
250
450
450
425
425
350
350
A 20 nF soft start capacitor results in negligible input current
overshoot at startup and is therefore suitable for most applications.
However, if an unusually large output capacitor is used, a longer
soft start period is required to prevent input inrush current.
Figure 26. Compensation Components
C2 =
R2 (kΩ)
10
10
10
10
10
10
10
10
10
10
10
10
10
10
The soft start capacitor limits the rate of voltage rise on the COMP
pin. This limits the peak switch current at startup. Table 6 shows a
typical soft start period (tSS) at maximum output current (IOUT_MAX)
for several conditions.
ERROR AMP
REF
R1 (kΩ)
30.9
30.9
63.4
63.4
71.5
71.5
88.7
88.7
63.4
63.4
71.5
71.5
88.7
88.7
Table 6. Typical Soft Start Period
VIN (V)
3.3
(19)
For low ESR output capacitance such as with a ceramic capacitor, C2 is optional. For optimal transient performance, the
RCOMP and CCOMP may need to be adjusted by observing the load
transient response of the ADP1610. For most applications, the
compensation resistor is to be in the range of 30 kΩ to 400 kΩ,
and the compensation capacitor is to be in the range of 100 pF
to 1.2 nF. Table 5 shows external component values for several
applications.
SOFT START CAPACITOR
The voltage at SS ramps up slowly by charging the soft start
capacitor (CSS) with an internal 3 μA current source. Table 6
listed the values for the soft start period, based on maximum
output current and maximum switching frequency.
5
VOUT (V)
5
5
9
9
12
12
9
9
12
12
COUT (μF)
10
10
10
10
10
10
10
10
10
10
CSS (nF)
20
100
20
100
20
100
20
100
20
100
tSS (ms)
0.3
2
2.5
8.2
3.5
15
0.4
1.5
0.62
2
Conversely, if fast startup is a requirement, the soft start capacitor
can be reduced or even removed, allowing the ADP1610 to start
quickly, but allowing greater peak switch current (see Figure 22
to Figure 25).
Rev. A | Page 14 of 20
ADP1610
APPLICATION CIRCUITS
TFT LCD BIAS SUPPLY
The circuit in Figure 27 shows the ADP1610 in a step-up configuration that can be used to generate a 10 V regulator with the
following specifications: VIN = 2.5 V to 5.5 V, VOUT = 10 V, and
IOUT ≤ 400 mA.
Figure 29 shows a power supply circuit for TFT LCD module
applications. This circuit has +10 V, −5 V, and +22 V outputs.
The +10 V is generated in the step-up configuration. The −5 V
and +22 V are generated by the charge-pump circuit. During
the step-up operation, the SW node switches between 10 V and
ground (neglecting forward drop of the diode and on resistance
of the switch). When the SW node is high, C5 charges up to 10 V.
C5 holds its charge and forward-biases D8 to charge C6 to −10 V.
The Zener diode (D9) clamps and regulates the output to −5 V.
4.7µH
L
6
IN
3
SD
7
RT
8
SS
D1
10V
SW 5
R1
71.3kΩ
ON
The VGH output is generated in a similar manner by the chargepump capacitors, C1, C2, and C4. The output voltage is tripled
and regulated down to 22 V by the Zener diode, D5.
FB 2
CSS
22nF
R2
10kΩ
COMP 1
COUT
10µF
RCOMP
220kΩ
CCOMP
150pF
GND
4
04472-030
CIN
10µF
R4
200Ω
VGL
–5V
D9
BZT52C5VIS
Figure 27. 3.3 V to 10 V Step-Up Regulator
R3
10kΩ
SD
CSS
22nF
7
RT
OFF
8
SS
10V
R1
71.3kΩ
CSS
22nF
R2
10kΩ
COMP 1
GND
4
RCOMP
220kΩ
CCOMP
150pF
D5
BZT52C22
BAV99
D3
IN
3
SD
7
RT
8
SS
D1
C2
1µF
10V
SW 5
R1
71.3kW
R2
10kΩ
COMP 1
GND
4
RCOMP
220kΩ
CCOMP
150pF
Figure 29. TFT LCD Bias Supply
FB 2
CIN
10µF
VGH
22V
FB 2
CIN
10µF
COUT
10mF
04472-031
Q1 B
3
D1
SW 5
6
ON
L
IN
D4
C3
10µF
C1
10nF D2
BAV99
ADP1610
3.3V
4.7µH
6
D5
L
Some battery-powered applications require very low standby
current. The ADP1610 typically consumes 10 nA from the
input, which makes it suitable for these applications. However,
the output is connected to the input through the inductor and
the rectifying diode, allowing load current draw from the input
while shut down. The circuit in Figure 28 enables the ADP1610
to achieve output load disconnect at shutdown. To shut down
the ADP1610 and disconnect the output from the input, drive
the SD pin below 0.4 V.
ADP1610
D8
4.7mH
DC-TO-DC STEP-UP SWITCHING CONVERTER
WITH TRUE SHUTDOWN
Q1 A FDC6331
C6
10µF
C5
10nF
D7
The output is set to the desired voltage using Equation 1.
To change the compensation network, see Equation 17 and
Equation 18.
3.3V
BAV99
R3
200Ω
C4
10nF
Figure 28. Step-Up Regulator with True Shutdown
Rev. A | Page 15 of 20
COUT
10µF
04472-033
ADP1610
3.3V
ADP1610
SEPIC POWER SUPPLY
The input and the output are dc-isolated by a coupling capacitor
(C1). In steady state, the average voltage of C1 is the input voltage.
When the ADP1610 switch turns on and the diode turns off, the
input voltage provides energy to L1, and C1 provides energy to
L2. When the ADP1610 switch turns off and the diode turns on,
the energy in L1 and L2 is released to charge the output capacitor
(COUT) and the coupling capacitor (C1) and to supply current to
the load.
4.7µH
L1
C1
10µF
ADP1610
2.5V–5.5V
6
IN
3
SD
SW
ON
CIN
10µF
Rev. A | Page 16 of 20
CSS
22nF
7
RT
8
SS
3.3V
5
4.7µH
L2
R1
16.8kΩ
FB 2
COMP 1
GND
4
RCOMP
60kΩ
CCOMP
1nF
Figure 30. 3.3 V DC-to-DC Converter
COUT
10µF
R2
10kΩ
04472-032
The circuit in Figure 30 shows the ADP1610 in a single-ended
primary inductance converter (SEPIC) topology. This topology
is useful for an unregulated input voltage, such as a batterypowered application in which the input voltage can vary between
2.7 V to 5 V, and the regulated output voltage falls within the input
voltage range.
ADP1610
LAYOUT PROCEDURE
To get high efficiency, good regulation, and stability, a welldesigned PCB layout is required. Where possible, use the
sample application board layout as a model.
•
•
•
•
•
Figure 32. Sample Application Board (Top Layer)
04472-029
•
The low ESR input capacitor (CIN) is to be kept close to IN
and GND.
The high current path from CIN through the inductor L1 to
SW and PGND is to be kept as short as possible.
The high current path from CIN through L1, the rectifier
D1, and the output capacitor COUT is to be kept as short as
possible.
High current traces are to be kept as short and as wide as
possible.
The feedback resistors are to be placed as close to the FB
pin as possible to prevent noise pickup.
The compensation components are to be placed as close as
possible to COMP.
To prevent radiated noise injection, high impedance traces
are not to be routed near any node connected to SW or
near the inductor.
Figure 33. Sample Application Board (Silkscreen Layer)
04472-027
•
04472-028
When designing PCBs the following guidelines are to be used
(see Figure 2):
Figure 31. Sample Application Board (Bottom Layer)
Rev. A | Page 17 of 20
ADP1610
OUTLINE DIMENSIONS
3.20
3.00
2.80
8
3.20
3.00
2.80
1
5
5.15
4.90
4.65
4
PIN 1
0.65 BSC
0.95
0.85
0.75
1.10 MAX
0.15
0.00
0.38
0.22
COPLANARITY
0.10
0.23
0.08
8°
0°
0.80
0.60
0.40
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-187-AA
Figure 34. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP1610ARMZ-R7 1
ADP1610-EVALZ1
1
Temperature Range
−40°C to +85°C
Package Description
8-Lead Mini Small Outline Package [MSOP]
Evaluation Board
Z = RoHS Compliant Part.
Rev. A | Page 18 of 20
Package Option
RM-8
Branding
P03
ADP1610
NOTES
Rev. A | Page 19 of 20
ADP1610
NOTES
©2004–2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04472-0-9/08(A)
Rev. A | Page 20 of 20
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