SiGe HBT Wideband Amplifier for Millimetre Wave Applications

SiGe HBT Wideband Amplifier for Millimetre Wave Applications
SiGe HBT Wideband Amplifier
for Millimetre Wave Applications
*M. Krčmar, *N. Noether, **B. Heinemann, **F. Korndörfer, ***Jan Hoffmann, *G. Boeck
*Technische Universität Berlin, Microwave Engineering,
Sekr. HFT 5-1, Einsteinufer 25, 10587 Berlin, Germany
**IHP, Im Technologiepark 25, 15236 Frankfurt (Oder), Germany
***MergeOptics GmbH, Am Borsigturm 17, 13507 Berlin, Germany
Abstract - A wideband amplifier up to 50 GHz has
been implemented in a 0.25 µm, 200 GHz ft SiGe
BiCMOS technology. Die size was 0.7 x 0.73 mm2.
The two-stage design achieves more than 11 dB gain
over the whole 20 to 50 GHz band. Gain maximum
was 14.2 dB at 47.5 GHz. Noise Figure was lower
than 9 dB up to 34 GHz and a current of 30 mA was
drawn from a 4 V supply. To the author’s best
knowledge this is the highest gain bandwidth
product of a monolithic SiGe HBT amplifier ever
reported.
Index Terms – Wideband amplifier, HBT,
millimetre-wave, bipolar integrated circuits.
SiGe,
I. INTRODUCTION
Up to now compound semiconductors dominate
the applications in the millimeter wave range. Rapid
technological progress in the field of SiGe HBT
technology [1] allows to re-think and to apply silicon
technologies also for millimeter wave applications.
SiGe BiCMOS technologies allow integration of
analog and digital parts, provide high integration
densities and save cost. The main purpose of this work
is the demonstration of the millimeter wave capabilities
of SiGe BiCMOS technologies. A broadband amplifier
has been chosen for this reason. The main goal was a
gain greater than 10 dB up to 50 GHz and a bandwidth
as high as possible. Compared with narrow band
designs, the difficulties grow up significantly because
of broadband matching circuits and the lossy silicon
substrate the overall performance degreases. Because
of these reasons only less comparable work has been
published so far. To our knowledge SiGe HBT
wideband amplifiers have been published just in [2]
(bandwidth =10 GHz) and [3] (1 – 15 GHz).
Elevated extrinsic base regions self-aligned to
the emitter window resulting in low base
resistance.
• Formation of the whole HBT structure in one
active area without shallow trench isolation
between emitter and collector contact
resulting in low collector resistance and small
collector-substrate junction areas.
• Device isolation without deep trenches
resulting in reduced process complexity and
improved heat dissipation.
The HBT module was fabricated in a BiCMOS
process after gate patterning and gate spacer formation.
During HBT fabrication, CMOS regions are protected
with a layer stack that is opened over HBT regions.
The HBT fabrication begins with the formation of the
collector wells by high-dose ion implantation. The
collector wells are laterally confined by shallow trench
regions. Next, the active collector region is defined by
depositing and patterning on oxide layer. A Si buffer
layer, the SiGe:C base layer, and a Si cap layer are
grown in one epitaxial step. After epitaxy, a sacrificial
layer is deposited, and emitter windows are structured.
An additional inside spacer is formed before depositing
and structuring the As-doped emitter. Next, spacers are
formed at the emitter and the sacrificial layer is
removed by wet etching, followed by the self-aligned
selective growth of the B-doped extrinsic base. In a
reference process without the elevated base, the
extrinsic base is formed by ion implantation after
emitter structuring as described in [4]. This HBT
structure provides low-capacitance isolation from the
substrate and low collector resistances.
•
II. DEVICE TECHNOLOGY
The circuit was fabricated in a commercially
available SiGe BiCMOS technology with ft = fmax =
200-GHz [4], [5]. The HBTs were designed for high
performance at low cost. The cross section of the
structure is illustrated in Fig. 1. Key features of the
device technology are:
Fig. 1: Schematic cross section of an HBT with elevated
extrinsic base.
Vcc
Vcc
Out
In
Fig. 2: Simplified schematic of the wideband amplifier.
III. CIRCUIT DESIGN
Fig. 2 shows the two-stage amplifier, implemented
with two cascode stages [6], [7]. Among different
architectures this topology has been found to be the
best trade-off with respect to gain and bandwidth. The
first stage was implemented with the goal of best noise
performances. The second one was designed for
maximum power gain. Both cascode stages are biased
with VCC = 4 V and IC = 10 mA and 20 mA,
respectively.
Fig. 3: Die photograph.
Fig. 3 shows a photogropf of the realized chip. In/output signal pads are on the left and right side,
respectively. The pad structure has been designed for
standard GSG on-wafer probing. The three top pads
provide the biasing for the whole amplifier in a groundVcc-ground structure. The identical structure at the
bottom side allows the monitoring of the bias voltage.
All DC-lines are RF-shorted to ground wherever
possible, in order to provide a low ohmic ground for
RF-signals. E.g. the relatively wide microstrip line in
the middle of the chip is a short circuited stub.
Because the input impedance of a bipolar transistor
shows a dominant real part with week frequency
dependence a simple microstrip line was adequate to
achieve acceptable 50 Ω-input matching over 2050 GHz. On the contrary, the output impedance of a
bipolar transistor is strongly capacitive and broadband
matching is complicated. The solution for this problem
was a compensation microstrip line connected in
parallel to the output transistor (see Fig. 2). It works as
a resonator with a low Q factor and a high bandwidth,
neutralizing the parasitic capacitances of the transistor
and allowing us to achieve a satisfying 50 Ω-output
matching in the 20-50 GHz band. This compensation
line not only allows for wideband output matching, but
also improves the stability of the amplifier at higher
frequencies. A short circuited stub was introduced
between the two stages in order to tune the gain
flatness, because at the beginning of the design process
the gain at the low frequency end of the band was
significantly higher. Theoretically this stub is an open
circuit for 30 < f < 50 GHz. Besides of degreasing the
gain below 20 GHz we automatically reduced also the
risk of instabilities at low frequencies. Once again,
microstrip lines were used instead of inductors [8],
which were not available, for DC-RF-decoupling. All
microstrip lines were realized on the same metal level
in order to avoid the coupling between the lines as
much as possible [8]. We chose hexagon-shaped signal
pads in order to reduce the capacitive coupling between
pads and substrate. The die size is 0.7 x 0.73 mm2. For
the biasing of the cascode transistors a fixed potential
provided by a simple voltage divider has been used
whereas the common emitter transistors were fed by a
diode/resistor combination providing a constant base
current. This solution saves chip size and provides On
the bottom side satisfying temperature stability, too.
Wherever possible, the first metal plane, used as
ground, was connected with the substrate by p-taps.
Thus, substrate coupling effects are effectively
suppressed. This p-taps behave like low ohmic
conductors at high frequencies. RF losses are
minimized by this way.
]
B -6
d
[ -7
IV. RESULTS
-8
1
2 -9
S
-10
Seven chips have been characterized from the first
wafer run so far with respect to its RF performance.
Power supply values and ambient temperature were in
all cases 4 V, 30 mA and 25°C, respectively. Fig. 4
shows the S21 graphs of the ensemble of 7 dies. Besides
of one curve we can conclude to a remarkable
uniformity and reliability of the semiconductor process
technology.
-11
-12
20
50
60
70
80
Fig. 5b: Simulated and measured input return loss.
]
B 0
d
[ -5
1 5
2
S 0
-10
2
2 -15
S
-20
-25
-10
20
30
40
50
60
70
80
freq, GHz
•
]
B 20
d
[ 15
Frequency response
30
•
40
50
60
70
80
Noise
Noise behavior was measured between 30 and 34 GHz.
The band limitation was given by the measurement
equipment. Tab. 1 shows the results. From simulations
a NF of at least 1 dB lower was expected at these
frequencies. According to our simulations best noise
values should occur between 40 and 50 GHz in
correlation with gain maximum. In that range we
expect experimental NF values in the order of 7 to
7.5 dB corresponding to simulation results of about
6 dB.
10
5
0
-5
-10
20
-30
20
freq, GHz
Fig. 5c: Simulated and measured output return loss.
Fig. 4: Measures S21 of seven Chips (VCC = 4 V).
1
2
S
40
freq, GHz
]
B 15
d
[ 10
-5
30
30
40
50
60
70
80
freq, GHz
Fig. 5a: Simulated and measured power gain (VCC = 4 V).
Figs. 5a to 5c represent both, measured (squares) and
simulated S-parameters. The maximum measured S21
achieved is 14.2 dB at 47.5 GHz and more than 11 dB
were measured between 20 and 50 GHz (Fig. 5a). Gain
flatness within the whole frequency band is about
±1.6 dB. At the same time, the reverse isolation (not
shown) is more than 40 dB from 20-50 GHz. Measured
input (Fig. 5b) and output (Fig. 5c) return losses are
lower than 6.5 dB and 10 dB, respectively, from 20 to
50 GHz.
f [GHz]
NF [dB]
30
8.9
31
8.8
33
9.0
34
8.7
Table 1: Measured Noise Figure
•
Gain compression
Gain compression was measured and biasing was
VCC = 4 V and IC = 10 mA and 20 mA, again. The
measurements were performed at 47.5 GHz, and an
input 1 dB compression point of -10 dBm (Fig. 6) was
found corresponding to +3 dBm at the output. This
value is slightly higher as predicted by the simulations.
1 dB Compression Point
15
14
gain [dB]
13
12
11
10
been achieved. To the best knowledge of the authors
this is the highest gain-bandwidth product ever
reported for a monolithic SiGe HBT amplifier. A
comparison with state of the art work is given in
Tab. 2. Nevertheless, the achieved performance should
be improvable if more reliable models for the passive
elements are available for the design process. In this
case we expect a considerable performance
enhancement especially with respect to gain, noise and
bandwidth. In case of low power applications the
power consumption should be considerably reducible,
too.
9
ACKNOWLEDGMENT
8
-40
-35
-30
-25
-20
-15
-10
-5
Input power [dBm]
Fig. 6: Gain compression at 47.5 GHz.
We would like to thank Dr. P. Heymann from
Ferdinand-Braun-Institut
Berlin
for
valuable
discussions and support concerning the noise
measurements.
V. DISCUSSION
The observed discrepancies between measured and
simulated results are mainly due by not sufficient
adequate models for the used passive structures in a
silicon environment in the millimeter wave region.
Verification work at some microstrip discontinuities,
transmission lines, vias and substrate coupling effects
based on 3D EM simulations showed us considerable
discrepancies with circuit simulator based results.
To overcome these problems we are working
towards experimental investigation of a passive
element library. Based on the characterization of these
structures we will establish a design library containing
the models of all passive elements. Using this library
we will then re-design the whole amplifier with respect
to enhanced performance. Especially with respect to
gain, noise and bandwidth we expect considerable
improvements.
this work
3-10 GHz
LNA [9]
3dB-freq.
range
20-50 GHz
3.1-17.6
GHz
Max.
gain
14.2 dB
@ 47.5 GHz
22 dB
@ 10 GHz
NF
ca. 8 dB
2.7-3.9 dB
Table 2: Comparison of state of the art work
VI. CONCLUSION
In this work a monolithic wideband amplifier
using a SiGe HBT technology has been presented. The
chip has been realized on a very low silicon area of
0.7 x 0.73 mm2. The main design goals, high bandwidth, more than 10 dB gain within this band and
acceptable noise figure and power consumption have
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