LT1513 SEPIC Constant-Current/ Constant
Final Electrical Specifications
LT1513
SEPIC Constant-Current/
Constant-Voltage
Battery Charger
May 1996
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DESCRIPTION
FEATURES
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The LT ®1513 is a 500kHz current mode switching regulator specially configured to create a constant-current/
constant-voltage battery charger. In addition to the usual
voltage feedback node, it has a current sense feedback
circuit for accurately controlling output current of a flyback
or SEPIC (Single-Ended Primary Inductance Converter)
topology charger. These topologies allow the current
sense circuit to be ground referred and completely separated from the battery itself, simplifying battery switching
and system grounding problems. In addition, these topologies allow charging even when the input voltage is
lower than the battery voltage. The LT1513 can also drive
a CCFL Royer converter with high efficiency in floating or
grounded mode.
Charger Input Voltage May Be Higher, Equal to or
Lower Than Battery Voltage
Charges Any Number of Cells Up to 20V
1% Voltage Accuracy for Rechargeable Lithium
Batteries
100mV Current Sense Voltage for High Efficiency
Battery Can Be Directly Grounded
500kHz Switching Frequency Minimizes
Inductor Size
Charging Current Easily Programmable or Shut Down
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APPLICATIONS
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Battery Charging of NiCd, NiMH, Lead-Acid
or Lithium Rechargeable Cells
Precision Current Limited Power Supply
Constant-Voltage/Constant-Current Supply
Transducer Excitation
Universal Input CCFL Driver
Maximum switch current on the LT1513 is 3A. This allows
battery charging currents up to 2A for a single lithium-ion
cell. Accuracy of 1% in constant-voltage mode is perfect
for lithium battery applications. Charging current can be
easily programmed for all battery types.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
L1A*
•
+
VIN
CHARGE
VSW
6
S/S
GND
4
VC
TAB
D1†
1.25A
L1B*
VFB
IFB
1
3
C5
0.1µF
2.4
5
LT1513
SYNC
AND/OR
SHUTDOWN SHUTDOWN
Maximum Charging Current
C2**
4.7µF
7
C3
22µF
25V
2
C4
0.22µF
SINGLE
Li-Ion CELL
(4.1V)
2.0
R1
1.8
•
R4
24Ω
INDUCTOR = 10µH
2.2
R2
+
C1
22µF
25V
×2
R3
0.08Ω
CURRENT (A)
WALL
ADAPTER
INPUT
1.6
DOUBLE Li-Ion
CELL (8.2V)
1.4
1.2
INDUCTOR = 10µH
ACTUAL PROGRAMMED
CHARGING CURRENT WILL
BE INDEPENDENT OF INPUT
VOLTAGE IF IT DOES NOT
EXCEED VALUES SHOWN
1.0
0.8
LT1513 • TA01
* L1A, L1B ARE TWO 10µH WINDINGS ON A
COMMON CORE: COILTRONICS CTX10-4
** CERAMIC MARCON THCR40EIE475Z OR TOKIN 1E475ZY5U-C304
† MBRD340 OR MBRS340T3. MBRD340 HAS 5µA TYPICAL
LEAKAGE, MBRS340T3 50µA TYPICAL
0.6
0.4
0
5
10
20
15
INPUT VOLTAGE (V)
25
30
LT1513 • TA02
Figure 1. SEPIC Charger with 1.25A Output Current
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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LT1513
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ABSOLUTE
RATI GS
Supply Voltage ....................................................... 30V
Switch Voltage ........................................................ 40V
S/S Pin Voltage ....................................................... 30V
FB Pin Voltage (Transient, 10ms) ......................... ±10V
VFB Pin Current .................................................... 10mA
IFB Pin Voltage (Transient, 10ms) ......................... ±10V
Storage Temperature Range ................ – 65°C to 150°C
Ambient Temperature Range
LT1513C .................................................. 0°C to 70°C
LT1513I .............................................. – 40°C to 85°C
Operating Junction Temperature Range
LT1513C ............................................... 0°C to 125°C
LT1513I ............................................ – 40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
VIN
S/S
VSW
GND
IFB
FB
VC
ORDER PART
NUMBER
LT1513CR
LT1513IR
ORDER PART
NUMBER
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
R PACKAGE
7-LEAD PLASTIC DD
VIN
S/S
VSW
GND
IFB
FB
VC
LT1513CT7
LT1513IT7
T7 PACKAGE
7-LEAD TO-220
TJMAX = 125°C, θJA = 30°C/ W
WITH PACKAGE SOLDERED TO 0.5INCH2 COPPER
AREA OVER BACKSIDE GROUND PLANE OR INTERNAL
POWER PLANE, θJA CAN VARY FROM 20°C/W TO
> 40°C/W DEPENDING ON MOUNTING TECHNIQUE
TJMAX = 125°C, θJA = 50°C/ W, θJC = 4°C/ W
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
VREF
FB Reference Voltage
Measured at FB Pin
VC = 0.8V
1.233
1.228
1.245
1.245
1.257
1.262
V
V
300
550
600
nA
nA
0.01
0.03
%/V
FB Input Current
●
VFB = VREF
●
VIREF
FB Reference Voltage Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
IFB Reference Voltage
Measured at IFB Pin
VFB = 0V, VC = 0.8V
●
– 107
–110
– 100
– 100
– 93
– 90
mV
mV
VIFB = VIREF (Note 2)
●
10
25
35
µA
IFB Reference Voltage Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
0.01
0.05
%/V
Error Amplifier Transconductance
∆IC = ±25µA
1500
●
1100
700
1900
2300
µmho
µmho
120
200
350
µA
1400
2400
µA
1.95
0.40
2.30
0.52
V
V
IFB Input Current
gm
AV
Error Amplifier Source Current
VFB = VREF – 150mV, VC = 1.5V
●
Error Amplifier Sink Current
VFB = VREF + 150mV, VC = 1.5V
●
Error Amplifier Clamp Voltage
High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.70
0.25
Duty Cycle = 0%
0.8
Error Amplifier Voltage Gain
VC Pin Threshold
2
500
1
V/ V
1.25
V
LT1513
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
f
Switching Frequency
2.7V ≤ VIN ≤ 25V
0°C ≤ TJ ≤ 125°C
TJ < 0°C
450
430
400
500
500
550
580
580
kHz
kHz
kHz
85
95
Maximum Switch Duty Cycle
●
Switch Current Limit Blanking Time
130
BV
Output Switch Breakdown Voltage
0°C ≤ TJ ≤ 125°C
TJ < 0°C
VSAT
Output Switch ON Resistance
ISW = 2A
●
ILIM
Switch Current Limit
Duty Cycle = 50%
Duty Cycle = 80% (Note 1)
●
●
∆IIN
∆ISW
40
35
260
ns
47
V
V
0.25
0.45
Ω
3.8
3.4
5.4
5.0
A
A
Supply Current Increase During Switch ON Time
15
25
mA/A
Control Voltage to Switch Current
Transconductance
4
Minimum Input Voltage
IQ
%
3.0
2.6
A/V
●
2.4
2.7
V
Supply Current
2.7V ≤ VIN ≤ 25V
●
4
5.5
mA
Shutdown Supply Current
2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V, TJ ≥ 0°C
TJ < 0°C
●
12
30
50
µA
µA
Shutdown Threshold
2.7V ≤ VIN ≤ 25V
●
0.6
1.3
2
V
●
5
12
25
µs
●
– 10
15
µA
●
600
800
kHz
Shutdown Delay
0V ≤ VS/S ≤ 5V
S/S Pin Input Current
Synchronization Frequency Range
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: For duty cycles (DC) between 50% and 85%, minimum
guaranteed switch current is given by ILIM = 1.33 (2.75 – DC).
Note 2: The IFB pin is servoed to its regulating state with VC = 0.8V.
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
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150°C
100°C
0.9
25°C
SWITCH CURRENT LIMIT (A)
0.8
0.7
0.6
0.5
–55°C
0.4
0.3
0.2
3.0
5
2.8
25°C AND
125°C
4
–55°C
3
2
INPUT VOLTAGE (V)
1.0
SWITCH SATURATION VOLTAGE (V)
Minimum Input Voltage
vs Temperature
2.6
2.4
2.2
2.0
1
0.1
0
0
0
0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0
SWITCH CURRENT (A)
LT1513 • G01
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
LT1513 • G02
1.8
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1513 • G03
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LT1513
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TYPICAL PERFORMANCE CHARACTERISTICS
Feedback Input Current
vs Temperature
0
800
fSYNC = 700kHz
2.5
2.0
1.5
1.0
0.5
0
–50 –25
Negative Feedback Input Current
vs Temperature
700
NEGATIVE FEEDBACK INPUT CURRENT (µA)
3.0
FEEDBACK INPUT CURRENT (nA)
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
Minimum Peak-to-Peak
Synchronization Voltage vs Temp
VFB = VREF
600
500
400
300
200
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1513 • G04
LT1513 • G05
–10
–20
–30
–40
–50
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1513 • G06
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PIN FUNCTIONS
VC: The Compensation pin is primarily used for frequency
compensation, but it can also be used for soft starting and
current limiting. It is the output of the error amplifier and
the input of the current comparator. Peak switch current
increases from 0A to 3.6A as the VC voltage varies from 1V
to 1.9V. Current out of the VC pin is about 200µA when the
pin is externally clamped below the internal 1.9V clamp
level. Loop frequency compensation is performed with a
capacitor or series RC network from the VC pin directly to
the Ground pin (avoid ground loops).
FB: The Feedback pin is used for positive output voltage
sensing. The R1/R2 voltage divider connected to FB defines Li-Ion float voltage at full charge, or acts as a voltage
limiter for NiCd or NiMH applications. FB is the inverting
input to the voltage error amplifier. Input bias current is
typically 300nA, so divider current is normally set to
100µA to swap out any output voltage errors due to bias
current. The noninverting input of this amplifier is tied
internally to a 1.245V reference. The grounded end of the
output voltage divider should be connected directly to the
LT1513 Ground pin (avoid ground loops).
IFB: The Current Feedback pin is used to sense charging
current. It is the input to a current sense amplifier that
controls charging current when the battery voltage is
below a programmed limit. During constant-current op-
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eration, the IFB pin regulates at – 100mV. Input resistance
of this pin is 5kΩ, so filter resistance (R4, Figure 1) should
be less than 50Ω. The 24Ω, 0.22µF filter shown in Figure
1 is used to convert the pulsating current in the sense
resistor to a smooth DC current feedback signal.
GND: The Ground pin is internally connected to the TAB
and both must be connected directly to a ground plane.
The TAB of the surface mount R package should be
soldered directly to the plane. It is also important that the
compensation network, the output voltage divider, the
output capacitor and the input capacitor be connected
directly to this ground plane. If the through-hole TO-220
package is mounted vertically with a heat sink, special
provisions must be made for a low impedance connection
between the heat sink and the ground plane, as outlined in
the Application Information section.
VSW: The Switch pin is the collector of the power switch,
carrying up to 3A of current with fast rise and fall times.
Keep the traces on this pin as short as possible to minimize
radiation and voltage spikes. In particular, the path in
Figure 1 which includes SW to C2, D1, C1 and around to
the LT1513 Ground pin should be as short as possible to
minimize voltage spikes at switch turn-off.
S/S: This pin can be used for shutdown and/or synchronization. It is logic level compatible, but can be tied to VIN if
LT1513
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PIN FUNCTIONS
desired. It defaults to a high ON state when floated. A logic
low state will shut down the charger to a micropower state.
Driving the S/S pin with a continuous logic signal of
600kHz to 800kHz will synchronize switching frequency to
the external signal. Shutdown is avoided in this mode with
an internal timer.
VIN (Pin 7): The Input Supply pin should be bypassed with
a low ESR capacitor located right next to the IC chip. The
grounded end of the capacitor must be connected directly
to the ground plane to which the TAB is connected. (See
special connection for the TO-220 package).
TAB: The TAB on both the surface mount R package and
the through-hole TO-220 T package is electrically connected to the Ground pin, but a low inductance connection
must be made to both the TAB and the pin for proper circuit
operation. See suggested PC layout in Figure 4 and T7
Package Layout Considerations.
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BLOCK DIAGRAM
VIN
SHUTDOWN
DELAY AND RESET
S/S
SYNC
SW
LOW DROPOUT
2.3V REG
500kHz
OSC
ANTI-SAT
LOGIC
DRIVER
SWITCH
+
IFBA
4k
–
IFB
COMP
50k
–
–
+
VFB
1.245V
REF
+
EA
IA
AV ≈ 6
VC
0.04Ω
–
GND
GND SENSE
LT1513 • BD
Figure 2
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OPERATION
The LT1513 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage or current. Referring to the
Block Diagram, the switch is turned “on” at the start of each
oscillator cycle. It is turned “off” when switch current
reaches a predetermined level. Control of output voltage
and current is obtained by using the output of a dual
feedback voltage sensing error amplifier to set switch
current trip level. This technique has the advantage of
simplified loop frequency compensation. A low dropout
internal regulator provides a 2.3V supply for all internal
circuitry on the LT1513. This low dropout design allows
input voltage to vary from 2.7V to 25V. A 500kHz oscillator
is the basic clock for all internal timing. It turns “on” the
output switch via the logic and driver circuitry. Special
adaptive antisat circuitry detects onset of saturation in the
power switch and adjusts driver current instantaneously to
limit switch saturation. This minimizes driver dissipation
and provides very rapid turn-off of the switch.
A unique error amplifier design has two inverting inputs
which allow for sensing both output voltage and current. A
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LT1513
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OPERATION
1.245V bandgap reference biases the noninverting input.
The first inverting input of the error amplifier is brought out
for positive output voltage sensing. The second inverting
input is driven by a “current” amplifier which is sensing
output current via an external current sense resistor. The
current amplifier is set to a fixed gain of – 12.5 which
provides a – 100mV current limit sense voltage.
The error signal developed at the amplifier output is
brought out externally and is used for frequency compensation. During normal regulator operation this pin sits at a
voltage between 1V (low output current) and 1.9V (high
output current). Switch duty cycle goes to zero if the VC pin
is pulled below the VC pin threshold, placing the LT1513 in
an idle mode.
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APPLICATIONS INFORMATION
The LT1513 is an IC battery charger chip specifically optimized to use the SEPIC converter topology. A complete
charger schematic is shown in the Typical Application
section. The SEPIC topology has unique advantages for
battery charging. It will operate with input voltages above,
equal to or below the battery voltage, has no path for battery
discharge when turned off, and eliminates the snubber
losses of flyback designs. It also has a current sense point
that is ground referred and need not be connected directly to
the battery. The two inductors shown are actually just two
identical windings on one inductor core, although two
separate inductors can be used.
A current sense voltage is generated with respect to ground
across R3 in Figure 1. The average current through R3 is
always identical to the current delivered to the battery. The
LT1513 current limit loop will servo the voltage across R3 to
– 100mV when the battery voltage is below the voltage limit
set by the output divider R1/R2. Constant current charging
is therefore set at 100mV/R3. R4 and C4 filter the current
signal to deliver a smooth feedback voltage to the IFB pin. R1
and R2 form a divider for battery voltage sensing and set the
battery float voltage. The suggested value for R2 is 12.4k. R1
is calculated from:
R2(VBAT – 1.245)
1.245 + R2(0.3µA)
VBAT = battery float voltage
0.3µA = typical FB pin bias current
R1 =
A value of 12.4k for R2 sets divider current at 100µA. This is
a constant drain on the battery when power to the charger is
off. If this drain is too high, R2 can be increased to 41.2k,
reducing divider current to 30µA. This introduces an addi-
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tional uncorrectable error to the constant voltage float mode
of about ±0.5% as calculated by:
VBAT Error =
±0.15µA(R1)(R2)
1.245(R1+ R2)
±0.15µA = expected variation in FB bias current around
the nominal 0.3µA typical value.
With R2 = 41.2k and R1 = 228k, (VBAT = 8.2V), the error due
to variations in bias current would be ±0.42%.
A second option is to disconnect the divider when charger
power is off. This can be done with a small FET as shown in
Figure 3. Disconnecting the divider leaves only diode leakage as a battery drain. See Diode Selection for a discussion
of diode leakage.
BATTERY
ADAPTER
INPUT
R1
VIN
Q1
VN2222
LT1513
VFB
GND
R2
LT1513 • F03
Figure 3. Eliminating Divider Current
Maximum input voltage for this circuit is partly determined
by battery voltage. A SEPIC converter has a maximum
switch voltage equal to input voltage plus output voltage.
The LT1513 has a maximum input voltage of 30V and a
maximum switch voltage of 40V, so this limits maximum
input voltage to 30V, or 40V – VBAT, whichever is less.
LT1513
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APPLICATIONS INFORMATION
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulled high or left floating for normal operation. A logic low
on the S/S pin activates shutdown, reducing input supply
current to 12µA. To synchronize switching, drive the S/S pin
between 600kHz and 800kHz.
Inductor Selection
L1A and L1B are normally just two identical windings on one
core, although two separate inductors can be used. A typical
value is 10µH, which gives about 0.5A peak-to-peak inductor current. Lower values will give higher ripple current,
which reduces maximum charging current. 5µH can be used
if charging currents are at least 20% lower than the values
shown in the maximum charging current graph. Higher
inductance values give slightly higher maximum charging
current, but are larger and more expensive. A low loss toroid
core such as KoolMµ®, Molypermalloy or Metglas® is recommended. Series resistance should be less than 0.04Ω for
each winding. “Open core” inductors, such as rods or
barrels are not recommended because they generate large
magnetic fields which may interfere with other electronics
close to the charger.
Input Capacitor
The SEPIC topology has relatively low input ripple current
compared to other topologies and higher harmonics are
especially low. RMS ripple current in the input capacitor is
less than 0.25A with L = 10µH and less than 0.5A with
L = 5µH. A low ESR 22µF, 25V solid tantalum capacitor (AVX
type TPS or Sprague type 593D) is adequate for most
applications with the following caveat. Solid tantalum capacitors can be destroyed with a very high turn-on surge
current such as would be generated if a low impedance input
source were “hot switched” to the charger input. If this
condition can occur, the input capacitor should have the
highest possible voltage rating, at least twice the surge input
voltage if possible. Consult with the capacitor manufacturer
before a final choice is made. A 4.7µF ceramic capacitor such
as the one used for the coupling capacitor can also be used.
These capacitors do not have a turn-on surge limitation. The
input capacitor must be connected directly to the VIN pin and
the ground plane close to the LT1513. See special considerations for the TO-220 through-hole package.
Output Capacitor
It is assumed as a worst case that all the switching output
ripple current from the battery charger could flow in the
output capacitor. This is a desirable situation if it is necessary
to have very low switching ripple current in the battery itself.
Ferrite beads or line chokes are often inserted in series with
the battery leads to eliminate high frequency currents that
could create EMI problems. This forces all the ripple current
into the output capacitor. Total RMS current into the capacitor has a maximum value of about 1A, and this is handled
with the two paralleled 22µF, 25V capacitors shown in Figure
1. These are AVX type TPS or Sprague type 593D surface
mount solid tantalum units intended for switching applications. Do not substitute other types without ensuring that
they have adequate ripple current ratings. See Input Capacitor section for details of surge limitation on solid tantalum
capacitors if the battery may be “hot switched” to the output
of the charger.
Coupling Capacitor
C2 in Figure 1 is the coupling capacitor that allows a SEPIC
converter topology to work with input voltages either higher
or lower than the battery voltage. DC bias on the capacitor is
equal to input voltage. RMS ripple current in the coupling
capacitor has a maximum value of about 1A at full charging
current. A conservative formula to calculate this is:
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(V + V )(1.1)
ICOUP(RMS) = CHRG IN BAT
2(VIN )
(1.1 is a fudge factor to account for inductor ripple current
and other losses)
With ICHRG = 1.2A, VIN = 15V and VBAT = 8.2V, ICOUP = 1.02A
The recommended capacitor is a 4.7µF ceramic type from
Marcon or Tokin. These capacitors have extremely low ESR
and high ripple current ratings in a small package. Solid
tantalum units can be substituted if their ripple current rating
is adequate, but typical values will increase to 22µF or more
to meet the ripple current requirements.
KoolMµ is a registered trademark of Magnetics, Inc.
Metglas is a registered trademark of AlliedSignal Inc.
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LT1513
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APPLICATIONS INFORMATION
Diode Selection
The switching diode should be a Schottky type to minimize
both forward and reverse recovery losses. Average diode
current is the same as output charging current , so this will
be under 2A. A 3A diode is recommended for most applications, although smaller devices could be used at reduced
charging current. Maximum diode reverse voltage will be
equal to input voltage plus battery voltage.
Diode reverse leakage current will be of some concern
during charger shutdown. This leakage current is a direct
drain on the battery when the charger is not powered. High
current Schottky diodes have relatively high leakage currents (5µA to 500µA) even at room temperature. The latest
very-low-forward devices have especially high leakage currents. It has been noted that surface mount versions of some
Schottky diodes have as much as ten times the leakage of
their through-hole counterparts. This may be because a low
forward voltage process is used to reduce power dissipation
in the surface mount package. In any case, check leakage
specifications carefully before making a final choice for the
switching diode. Be aware that diode manufacturers want to
specify a maximum leakage current that is ten times higher
than the typical leakage. It is very difficult to get them to
specify a low leakage current in high volume production.
This is an on going problem for all battery charger circuits
and most customers have to settle for a diode whose typical
leakage is adequate, but theoretically has a worst-case
condition of higher than desired battery drain.
Thermal Considerations
Care should be taken to ensure that worst-case conditions
do not cause excessive die temperatures. Typical thermal
resistance is 30°C/W for the R package but this number will
vary depending on the mounting technique (copper area, air
flow, etc).
Average supply current (including driver current) is:
IIN = 4mA +
(VBAT )(ICHRG )(0.024)
VIN
Switch power dissipation is given by:
(ICHRG )2 (RSW )(VBAT + VIN )(VBAT)
PSW =
(VIN )2
RSW = output switch ON resistance
Total power dissipation of the die is equal to supply current
times supply voltage, plus switch power:
PD(TOTAL) = (IIN)(VIN) + PSW
GROUND PLANE
VBAT
LT1513 TAB AND GROUND
PIN SOLDERED TO
GROUND PLANE
C1
C1
D1
C2
C5
L1A
C1,C3,C5 AND R3
TIED DIRECTLY TO
GROUND PLANE
R3
2 WINDING
INDUCTOR
L1B
C3
LT1513 • F04
VIN
Figure 4. LT1513 Suggested Layout for Critical Thermal and Electrical Paths
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LT1513
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APPLICATIONS INFORMATION
For VIN = 10V, VBAT = 8.2V, ICHRG = 1.2A, RSW = 0.3Ω
IIN = 4mA + 24mA = 28mA
PSW = 0.64W
PD = (10)(0.028) + 0.64 = 0.92W
T7 Package Layout Considerations
Electrical connection to the TAB of a T7 package is required
for proper device operation. If the TAB is tied directly to the
ground plane (like the surface mount package in Figure 4) no
other considerations are necessary. If the TAB is not connected directly to the ground plane, as in a vertically mounted
application, a separate electrical connection from the TAB to
a “floating node” is required. Ground returns for the VIN
capacitor, VC components, C4, R3 and output feedback
resistor divider are then connected to the floating node. This
is shown schematically in Figure 5. All other system ground
connections are made to Pin 4.
The electrical connection from the T7 package TAB to the
floating node must be a low resistance (<0.1Ω), low inductance (<20nH) path that can be accomplished with a jumper
wire or an electrically conductive heat sink.
Bolt the jumper wire directly to the TAB using a solder tail to
maintain low resistance. The jumper wire length should not
exceed 3/4 inch of 24 AWG gauge wire or larger to minimize
the inductance.
L1A
C2
VIN
VBAT
TO-220
T7 PACKAGE
C3
VC
D1
VSW
L1B
R1
VFB
GND GND
(TAB) (PIN 4) IFB
C1
R4
C5
C4
R2
Vertically mounted electrically conductive heat sinks are
available from many heat sink manufacturers. These heat
sinks also have tabs that solder directly to the board creating
the required low resistance, low inductance path from the
TAB to the floating node. The TAB should be bolted or
soldered directly to the heat sink to maintain low resistance.
Heat sinks are available in clip-on styles but are only
recommended if the TAB to heat sink contact resistance can
be maintained below 0.1Ω for the life of the product.
Programmed Charging Current
LT1513 charging current can be programmed with a PWM
signal from a processor as shown in Figure 6. C6 and D2
form a peak detector that converts a positive logic signal to a
negative signal. The average negative signal at the input to
R5 is equal to the processor VCC level multiplied by the
inverse PWM ratio. This assumes that the PWM signal is a
CMOS output that swings rail-to-rail with a source resistance less than a few hundred ohms. The negative voltage is
converted to a current by R5 and R6 and filtered by C7. This
current multiplied by R4 generates a voltage that subtracts
from the 100mV sense voltage of the LT1513. This is not a
high precision technique because of the errors in VCC and
the diode voltage, but it can typically be used to adjust
charging current over a 20% to 100% range with good
repeatability (full charging current accuracy is not affected).
To reduce the load on the logic signal, R4 has been increased
from 24Ω to 200Ω. This causes a known increase in fullscale charging current (PWM = 0) of 3% due to the 5k input
resistance of the IFB pin. Note that 100% duty cycle gives full
charging current and that very low duty cycles (especially
zero!) will not operate correctly. Very low duty cycle (<10%)
is a problem because the peak detector requires a finite
up-time to reset C6.
R3
LT1513
FLOATING NODE. TAB IS
TIED INTERNALLY TO GND
PIN 4. DO NOT CONNECT
THIS NODE TO GROUND.
IT IS CONNECTED TO
GROUND THROUGH
PACKAGE PIN 4 AS SHOWN
IFB
PWM
INPUT
≥1kHz
GROUND PLANE
(SYSTEM GROUND)
Figure 5
+
C6
1µF
R5
4.02k
D2
R6
4.02k
+
C7
10µF
R4
200Ω
C4
0.22µF
L1B
R3
LT1513 • F05
1513 F06
Figure 6
9
LT1513
U
W
U
U
APPLICATIONS INFORMATION
More Help
ment is always ready to lend a helping hand. The LT1371
data sheet may also be helpful. This part is identical to the
LT1513 except for the current amplifier circuitry.
Linear Technology Field Application Engineers have a CAD
spreadsheet program for detailed calculations of circuit
operating conditions. In addition, our Applications Depart-
U
TYPICAL APPLICATION
This Cold Cathode Fluorescent Lamp (CCFL) driver uses a
Royer class self-oscillating sine wave converter to drive a
high voltage lamp with an AC waveform. CCFL Royer
converters have significantly degraded efficiency if they
must operate at low input voltages, and this circuit was
designed to handle input voltages as low as 2.7V. Therefore, the LT1513 is connected to generate a negative
current through L2 that allows the Royer to operate as if it
were connected to a constant higher voltage input.
2.8V/R1. Dimming is accomplished by feeding a PWM
signal through R3 and R4 that is filtered by C2 and summed
with the bulb feedback. For more information on this
circuit, contact the LTC Applications Department. Considerable written application literature on Royer CCFL circuits
is also available from LTC Application and Design Notes.
Note: This circuit operates with one end of the bulb effectively grounded. In some situations with high stray bulb
capacitance caused by enclosures, a floating bulb drive may
be much more efficient. See CCFL Driver (Floating Lamp).
Lamp current is tightly controlled with the rectifying feedback loop through D2 and D3. Bulb current is equal to
+V
2.7V TO 26V
47µF
TANT
L1*
20µH
4.7µF
CERAMIC
L2*
20µH
+
5
VSW
S/S
1
1µF
2N3904
5nF
5nF
3
5V PWM
DIMMING
(≥1kHz)
R4
20k
470Ω
T1
2
4, TAB
R3
1N4148
10k
C2
1µF
1
R2
12k
C1
0.1µF
2
•3
D2
1N4148
R1
500Ω
T1: CTX110605 (67:1)
*COILTRONICS CTX20-4 (MUST BE SEPARATE INDUCTORS)
Figure 7. CCFL Driver (Grounded Lamp)
10
Q2
GND
2k
10k
FOR INPUT VOLTAGES GREATER THAN 20V
ADD CIRCUITRY IN DASHED LINE. THIS
CIRCUITRY PREVENTS SWITCH DAMAGE
DURING START-UP OR LOOP TRANSIENTS
WHEN PEAK SWITCH VOLTAGE MIGHT
EXCEED THE 40V LIMIT
FB
•
D1
3A
VC
IFB
ZTX849 (2 EACH)
4
5
0.1µF
WIMA
LT1513
36V
Q1
10
27pF
CCFL
6
4.7k
7
VIN
•6
LAMP
CURRENT
5.6mA
D3
1N4148
1513 TA03
LT1513
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
R Package
7-Lead Plastic DD Pak
(LTC DWG # 05-08-1462)
0.256
(6.502)
0.060
(1.524)
0.060
(1.524)
TYP
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
15° TYP
0.060
(1.524)
0.183
(4.648)
0.059
(1.499)
TYP
0.330 – 0.370
(8.382 – 9.398)
BOTTOM VIEW OF DD PAK
HATCHED AREA IS SOLDER PLATED
COPPER HEAT SINK
(
+0.008
0.004 –0.004
+0.203
0.102 –0.102
)
0.095 – 0.115
(2.413 – 2.921)
0.075
(1.905)
0.300
(7.620)
0.045 – 0.055
(1.143 – 1.397)
(
+0.012
0.143 –0.020
+0.305
3.632 –0.508
0.040 – 0.060
(1.016 – 1.524)
0.026 – 0.036
(0.660 – 0.914)
)
0.013 – 0.023
(0.330 – 0.584)
0.050 ± 0.012
(1.270 ± 0.305)
R (DD7) 0695
T7 Package
7-Lead Plastic TO-220 (Standard)
(LTC DWG # 05-08-1422)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.293 – 4.572)
0.147 – 0.155
(3.734 – 3.937)
DIA
0.045 – 0.055
(1.143 – 1.397)
0.230 – 0.270
(5.842 – 6.858)
0.460 – 0.500
(11.684 – 12.700)
0.570 – 0.620
(14.478 – 15.748)
0.330 – 0.370
(8.382 – 9.398)
0.620
(15.75)
TYP
0.700 – 0.728
(17.780 – 18.491)
0.152 – 0.202
0.260 – 0.320 (3.860 – 5.130)
(6.604 – 8.128)
0.040 – 0.060
(1.016 – 1.524)
0.095 – 0.115
(2.413 – 2.921)
0.013 – 0.023
(0.330 – 0.584)
0.026 – 0.036
(0.660 – 0.914)
0.135 – 0.165
(3.429 – 4.191)
0.155 – 0.195
(3.937 – 4.953)
T7 (TO-220) (FORMED) 0695
11
LT1513
U
TYPICAL APPLICATION
This Cold Cathode Fluorescent Lamp driver uses a Royer
class self-oscillating sine wave converter to driver a high
voltage lamp with an AC waveform. CCFL Royer converters
have significantly degraded efficiency if they must operate at
low input voltages, and this circuit was designed to handle
input voltages as low as 2.7V. Therefore, the LT1513 is
connected to generate a negative current through L2 that
allows the Royer to operate as if it were connected to a
constant higher voltage input.
The Royer output winding and the bulb are allowed to float in
this circuit. This can yield significantly higher efficiency in
situations where the stray bulb capacitance to surrounding
enclosure is high (see Figure 7). To regulate bulb current in
Figure 8, Royer input current is sensed with R1 and filtered
with R2 and C1. This negative feedback signal is applied to
the IFB pin of the LT1513. For more information on this
circuit contact the LTC Applications Department. Considerable written application literature on Royer CCFL circuits is
also available from LTC Application and Design Notes.
+V
2.7V TO 26V
47µF
TANT
L1*
20µH
4.7µF
CERAMIC
L2*
20µH
LAMP CURRENT
5.6mA
+
CCFL
6
4.7k
7
5
VIN
VSW
1µF
2N3904
FB
5nF
C6
1µF
D2
1N4148
PWM DIMMING**
(≈1kHz)
R5
2.7k
R6
2.7k
470Ω
D1
3A
3
4, TAB
•
10
GND
2k
10k
FOR INPUT VOLTAGES GREATER THAN 20V
ADD CIRCUITRY IN DASHED LINE. THIS
CIRCUITRY PREVENTS SWITCH DAMAGE
DURING START-UP OR LOOP TRANSIENTS
WHEN PEAK SWITCH VOLTAGE MIGHT
EXCEED THE 40V LIMIT
IFB
Q2
27pF
0.1µF
WIMA
VC
2
5nF
Q1
S/S
LT1513
1
36V
ZTX849 (2 EACH)
4
5
R2
300Ω
1
2
C1
0.1µF
6
T1
•3
R1
0.2Ω
C7
10µF
1513 TA04
T1: CTX110605 (67:1)
*COILTRONICS CTX20-4 (MUST BE SEPARATE INDUCTORS)
**100% DUTY CYCLE = FULL BRIGHTNESS. DO NOT USE DUTY
CYCLE <10% (BRIGHTNESS SHOULD GO TO ZERO BEFORE 10%)
Figure 8. CCFL Driver (Floating Lamp)
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LT1239
Backup Battery Management System
Charges Backup Battery and Regulates Backup Battery Output when
Main Battery Removed
LTC®1325
Microprocessor Controlled Battery Management System
Can Charge, Discharge and Gas Gauge NiCd, NiMH and Pb-Acid Batteries
with Software Charging Profiles
LT1510
1.5A Constant-Current/Constant-Voltage Battery Charger
Step-Down Charger for Li-Ion, NiCd and NiMH
LT1511
3.0A Constant-Current/Constant-Voltage Battery Charger
with Input Current Limiting
Step-Down Charger that Allows Charging During Computer Operation and
Prevents Wall-Adapter Overload
LT1512
SEPIC Constant-Current/Constant-Voltage Battery Charger Step-Up/Step-Down Charger for Up to 1A Current
12
Linear Technology Corporation
LT/GP 0596 5K • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
 LINEAR TECHNOLOGY CORPORATION 1996
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