Intel® Technology Journal
Volume 08
Issue 03
Published, August 20, 2004
ISSN 1535-864X
Intel
Technology
Journal
®
WiMAX
This issue of Intel Technology Journal (Volume 8, Issue 3) examines the technologies and standards
for WiMAX (Worldwide Interoperability for Microwave Access)—an evolving standard for point-tomultipoint wireless networking—and Intel's research and development efforts in these areas.
Inside you’ll find the following papers:
Global, Interoperable
Broadband Wireless
Networks: Extending WiMAX
Technology to Mobility
IEEE 802.16 Medium
Access Control and
Service Provisioning
RF System and Circuit
Challenges for WiMAX
Multiple-Antenna Technology
in WiMAX Systems
Scalable OFDMA Physical
Layer in IEEE 802.16
WirelessMAN
Fully Integrated CMOS
Radios from RF to Millimeter
Wave Frequencies
More information, including current and past issues of Intel Technology Journal, can be found at:
http://developer.intel.com/technology/itj/index.htm
Volume 08
Issue 03
Published, August 20, 2004
ISSN 1535-864X
Intel Technology Journal
®
WiMAX
Articles
Preface
iii
Foreword
v
Technical Reviewers
vii
Global, Interoperable Broadband Wireless Networks:
Extending WiMAX Technology to Mobility
173
RF System and Circuit Challenges for WiMAX
189
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
201
IEEE 802.16 Medium Access Control and Service Provisioning
213
Multiple-Antenna Technology in WiMAX Systems
229
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
241
WiMAX
i
Intel Technology Journal, Volume 8, Issue 3, 2004
THIS PAGE INTENTIONALLY LEFT BLANK
WiMAX
ii
Intel Technology Journal, Volume 8, Issue 3, 2004
Preface
By Lin Chao
Publisher, Intel Technology Journal
WiMAX is a technology for “wireless” broadband. Today, when you want broadband, you connect
using T1, DSL or cable modems to physical cables called landlines. WiMAX (Worldwide
Interoperability for Microwave Access), an evolving standard for point-to-multipoint wireless
networking, works for the “last mile” in the same way that Wi-Fi “hotspots” work for the last one
hundred feet of networking within a building or a home. In addition to “last mile” broadband
connections, WiMAX has a number of other applications in hotspots, cellular backhaul and in highspeed enterprise connectivity.
Generally speaking, WiMAX has a range of up to 30 miles. WiMAX covers several different
frequency ranges. The base 802.16 standard is for the 10 to 66 GHz range. 802.16a added coverage
for the 2 to 11 GHz range. WiMAX, and most commercial interests, cover these lower ranges.
The ability to provide these broadband connections wirelessly, without laying wire or cable in the
ground, greatly lowers the cost to provide these services. So, WiMAX may change the economics for
any place where the cost of laying or upgrading landlines to broadband capacity is prohibitively
expensive, as in emerging countries. In countries like India, Mexico, and China, where there is
currently insufficient wired infrastructure, WiMAX can become part of the broadband backbone.
This issue of Intel Technology Journal (Volume 8, Issue 3) examines the technologies and standards
for WiMAX, and Intel’s research and development efforts in these areas. The first paper is an
overview and examines Intel’s architecture vision for 802.16 and the Worldwide Interoperability
Microwave Access (WiMAX) certification process. It also covers the three stages of deployments that
Intel sees. The second paper discusses several RF and circuit challenges for WiMAX. WiMAX’s RF
is made more complicated by the fact that WiMAX covers both licensed and unlicensed bands.
The third paper provides a brief tutorial on the IEEE 802.16 WirelessMAN Orthogonal Frequency
Division Multiple Access (OFDMA) with an emphasis on a scalable OFDMA. OFDM is a spreadspectrum technology that bundles data over narrowband carriers transmitted in parallel at different
frequencies. The fourth paper discusses the IEEE 802.16 Medium Access Control (MAC) protocols,
which are key elements for WiMAX deployments. The fifth paper discusses the benefits of multiple
antenna systems over single antenna systems in WiMAX deployments. Currently, IEEE 802.16
supports several multiple-antenna options, including Space-Time Codes (STC), Multiple-Input
Multiple-Output (MIMO) antenna systems and Adaptive Antenna Systems (AAS).
The last paper explores fully integrated CMOS radios from RF to millimeter wave frequencies. The
paper discusses recent CMOS with capabilities for Radio Frequency (RF), microwave, and millimeter
wave circuits from 1 GHz to 100 GHz, advances in on-die isolation structures for integrating radio's
delicate circuits with noisy processors, and novel design methods for complex RF passive networks
on the substrate of the package.
WiMAX
iii
Intel Technology Journal, Volume 8, Issue 3, 2004
These papers reveal the collective efforts by Intel, standards bodies, and the wireless industry to make
WIMAX technology deployment a reality for practical applications in our everyday life.
WiMAX
iv
Intel Technology Journal, Volume 8, Issue 3, 2004
Foreword
Emerging Broadband Networks: The Case for WiMAX
By Scott G. Richardson
Broadband Wireless Division, Intel Corporation
Broadband wireless will revolutionize people's lives by enabling a high-speed connection directly to
the information they need, whenever and wherever they need it. Broadband data services, such as
delivery of rich Internet Protocol and media content, are an increasingly important component of the
services and revenue of network operators, who want to expand the reach of their broadband data
networks without expensive construction and infrastructure costs. High-speed broadband wireless data
overlays to voice network are just emerging, as service providers respond to these consumer and
enterprise demands for rich media, mobile applications and services.
Intel is, and will continue to be, a key player in this broadband wireless wave, offering silicon
products, platform solutions and helping to drive and develop the industry ecosystem. Intel believes
multiple wireless technologies will coexist, working synergistically where the user will be “best
connected” with the technology most suited to network conditions and desired services. This issue of
Intel Technology Journal (ITJ) delves deeply into one of these key wireless technologies – WiMAX.
WiMAX (Worldwide Interoperability for Microwave Access) is poised to become a key technical
underpinning of fixed, portable and mobile data networks. WiMAX is an implementation of the
emerging IEEE 802.16 standard that uses Orthogonal Frequency Division Multiplexing (OFDM) for
optimization of wireless data services. OFDM technology uses “sub-carrier optimization,” assigning
small sub-carriers (kHz) to users based on radio frequency conditions. This enhanced spectral
efficiency is a great benefit to OFDM networks and makes them very well suited to high-speed data
connections for both fixed and mobile users. Systems based on the emerging IEEE 802.16 standards
are the only standardized OFDM-based Wireless Wide Area Networks (WWAN) infrastructure
platforms today.
Service providers will operate WiMAX on licensed and unlicensed frequencies. The technology
enables long-distance wireless connections with speeds up to 75 megabits per second. (However,
network planning assumes a WiMAX base station installation will cover the same area as cellular
base stations do today.) Wireless WANs based on WiMAX technology cover a much greater distance
than Wireless Local Area Networks (WLAN), connecting buildings to one another over a broad
geographic area. WiMAX can be used for a number of applications, including "last-mile" broadband
connections, hotspot and cellular backhaul, and high-speed enterprise connectivity for businesses.
Intel sees WiMAX deploying in three phases: the first phase of WiMAX technology (based on IEEE
802.16-2004) will provide fixed wireless connections via outdoor antennas in the first half of 2005.
Outdoor fixed wireless can be used for high-throughput enterprise connections (T1/E1 class services),
hotspot and cellular network backhaul, and premium residential services.
In the second half of 2005, WiMAX will be available for indoor installation, with smaller antennas
similar to 802.11-based WLAN access points today. In this fixed indoor model, WiMAX will be
WiMAX
v
Intel Technology Journal, Volume 8, Issue 3, 2004
available for use in wide consumer residential broadband deployments, as these devices become "user
installable," lowering installation costs for carriers.
By 2006, technology based on the IEEE 802.16e standards will be integrated into portable computers
to support movement between WiMAX service areas. This allows for portable and mobile
applications and services. In the future, WiMAX capabilities will even be integrated into mobile
handsets.
In this issue of the Intel Technology Journal, we give background into the key silicon and system
design issues for WiMAX networks, including radio frequency, physical layer and media access
control technologies. We also discuss network-level architecture for WiMAX and how to create endto-end, interoperable networks based on a common set of protocols and standards. In addition, Intel
Technology Journal pays particular attention to issues of silicon integration and managing multiple
antennas, very important in an environment where cost/power are paramount and users will use
multiple wireless technologies to access the network. With the background provided in this issue of
the ITJ, the reader will be better informed of the exciting benefits of this new standard and technology,
and will be better able to profit from this new wireless wave.
WiMAX
vi
Intel Technology Journal, Volume 8, Issue 3, 2004
Technical Reviewers
Alluri, Prasad, Intel Communications Group
Andelman, Dov, Intel Communications Group
Baraa, Al-Dabagh, Intel Communications Group
Cox, Timothy F., Intel Communications Group
Foerster, Jeffrey R., Corporate Technology Group
Ho, Minnie, Corporate Technology Group
Kalluri, Sudhakar, Desktop Platforms Group
Lebizay, Gerald, Intel Communications Group
Liu, Tony, Intel Communications Group
Mitchel, Henry, Intel Communications Group
Ovadia, Shlomo, Intel Communications Group
Putzolu, David, Intel Communications Group
Salvekar, Atul A., Intel Communications Group
Talwar, Shilpa, Corporate Technology Group
Teckman, Tim, Intel Communications Group
Thomas, Rainer E., Desktop Platforms Group
WiMAX
vii
Intel Technology Journal, Volume 8, Issue 3, 2004
THIS PAGE INTENTIONALLY LEFT BLANK
WiMAX
viii
Global, Interoperable Broadband Wireless Networks:
Extending WiMAX Technology to Mobility
Ed Agis, Intel Communications Group, Intel Corporation
Henry Mitchel, Intel Communications Group, Intel Corporation
Shlomo Ovadia, Intel Communications Group, Intel Corporation
Selim Aissi, Corporate Technology Group, Intel Corporation
Sanjay Bakshi, Corporate Technology Group, Intel Corporation
Prakash Iyer, Corporate Technology Group, Intel Corporation
Masud Kibria, Corporate Technology Group, Intel Corporation
Christopher Rogers, Corporate Technology Group, Intel Corporation
James Tsai, Corporate Technology Group, Intel Corporation
Index words: 802.16, WiMAX, OFDM, OFDMA, portability, mobility, broadband wireless
architecture, PKM, WiMAX certification, interoperability
ABSTRACT
IEEE* 802.16 is an emerging global broadband wireless
access standard capable of delivering multiple megabits
of shared data throughput supporting fixed, portable, and
mobile operation. The standard offers a great deal of
design flexibility including support for licensed and
license-exempt frequency bands, channel widths ranging
from 1.25 to 20 MHz, Quality of Service (QoS)
establishment on a per-connection basis, strong security
primitives, multicast support, and low latency/low packet
loss handovers1. Mass deployments of Subscriber
Stations (SS) and Access Points2 (AP) for portable and
mobile services are expected to be based on scalable
Orthogonal Frequency Division Multiplexing with
Multiple Access (OFDMA). A broad range of network
operators are anticipated to deploy such systems in
licensed frequencies below 11 GHz. However, universal
acceptance of 802.16 for portable and mobile use is
contingent on the Industry’s development, acceptance,
*
Other brands and names are the property of their
respective owners
1
Optimization of PHY and MAC handover primitives is ongoing in the
802.16e Task Group and is expected to be completed by the end of 2004.
2
In this paper the term Access Point is synonymous with Base Station, and
an AP can be logically broken into a combination of one APC and one or
more APTs.
and conformance to two complementary aspects of the
IEEE 802.16 air interface standards work: (1)
development and adoption of an open and extensible endto-end architecture framework and specification that is
agnostic to incumbent operator backend networks; and
(2) a means for ensuring spec-compliant and vendor
interoperable equipment to support cost-effective
deployments and give users the capability to roam across
networks established by different network operators. A
common architecture framework and standardized
compliance testing mechanisms based on a suite of PHY
and
MAC profiles
will enable
multivendor
interoperability supporting different deployment and usecase scenarios. In this paper, we describe Intel’s 802.16
architecture vision and the Worldwide Interoperability
Microwave Access (WiMAX) certification process to
address these two important market needs.
INTRODUCTION
IEEE 802.16 is an emerging suite of air interface
standards for combined fixed, portable, and Mobile
Broadband Wireless Access (MBWA). Initially conceived
as a radio standard to enable cost-effective last-mile
broadband connectivity to those not served by wired
broadband such as cable or DSL, the specifications are
evolving to target a broader market opportunity for
mobile, high-speed broadband applications. The promise
of realizing a low-cost, broadly interoperable wide-area
data network that supports portable and mobile usage
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
173
Intel Technology Journal, Volume 8, Issue 3, 2004
could have significant end-user benefits. Notably, this
network can complement and extend the Wi-Fi hotspot
usage model to provide broader Internet Protocol (IP)
data service coverage and roaming that has so far eluded
current 3G systems, due to system cost and complexity.
The 802.16-2004 [1] standard to be published later this
year supersedes all previous versions as the base standard
and specifies networks for the current fixed access
market segment. The 802.16e [2] amendment and the
soon to be approved 802.16f and 802.16g task groups will
amend the base specification to enable not just fixed, but
also portable and mobile operation in frequency bands
below 6 GHz.
802.16 is optimized to deliver high, bursty data rates to
Subscriber Stations (SS) but the sophisticated Medium
Access Control (MAC) architecture can simultaneously
support real-time multimedia and isochronous
applications such as Voice Over IP (VoIP) as well. This
means that IEEE 802.16 is uniquely positioned to extend
broadband wireless beyond the limits of today’s Wi-Fi
systems, both in distance and in the ability to support
applications requiring advanced Quality of Service (QoS)
such as VoIP, streaming video, and on-line gaming.
The technology is expected to be adopted by different
incumbent operator types–for example, Wireless Internet
Service Providers (WISPs), cellular operators (CDMA
and WCDMA), and wireline broadband providers. Each
of these operators will approach the market with different
business models, each based on their current markets and
perceived opportunities for broadband wireless as well as
different requirements for integration with existing
(legacy) networks. As a result, 802.16 network
deployments face the challenging task of needing to adapt
to different network architectures while still supporting
standardized components and interfaces for multivendor
interoperability.
This paper is organized into two main sections. The first
section presents Intel’s deployment vision and
architecture framework for 802.16. The architecture and
usage is presented as a two-stage evolution: initially
combining fixed access with portability and scaling up to
evolve to full mobility. The framework is based on
several core principles:
•
Support for different Radio Access Network (RAN)
topologies.
•
Well-defined interfaces to enable 802.16 RAN
architecture independence while enabling seamless
integration and interworking with Wi-Fi, 3GPP3 and
3GPP2 networks.
•
Leverage open, Internet Engineering Task Force
(IETF)-defined IP technologies to build scalable allIP 802.16 access networks using Common Off The
Shelf (COTS) equipment.
•
Support for IPv4 and IPv6 clients and application
servers; recommending use of IPv6 in the
infrastructure.
•
Functional extensibility to support future migration
to full mobility and delivery of rich broadband
multimedia.
In the second section, the WiMAX certification process
with its key building blocks is reviewed. The WiMAX
certification process, which is being established by the
WiMAX Forum, enables multivendor interoperability of
subscriber systems and access points for this ecosystem.
BROADBAND WIRELESS DEPLOYMENT
SCENARIOS
Initial deployments of IEEE 802.16 standards-based
networks will likely target fixed access connectivity to
unserved and underserved markets where wireline
broadband services are insufficient to fulfill the market
need for high-bandwidth Internet connectivity. Prestandards implementations exist today that are beginning
to address this fixed access service environment.
Standardization will help accelerate the ramp for these
fixed access solutions by providing interoperability
amongst equipment and economies of scale resulting
from high-volume standards-based components.
As IEEE 802.16 solutions evolve to address portable and
mobile applications, the required features and
performance of the system will increase. Beyond fixed
access service, even larger market opportunities exist for
providing cost-effective broadband data services to users
on the go. Initially this includes portable connectivity for
customers who are not within reach of their existing fixed
broadband or WLAN service options. This type of service
is characterized by access that is unwired but stationary
in most cases, albeit with some limited provisions for user
mobility during the connection. In this manner, 802.16
can be seen as augmenting coverage of 802.11 for private
and public service networks and cost effectively
extending hotspot availability to wider ranges of
3
3GPP – Third Generation Partnership Project – a collaborative effort
between ARIB, CCSA, ETSI, ATIS, TTA, and TTC to develop 3G
telecommunications standards. 3GPP2 is a similar collaborative effort
between ANSI, TIA and EIA-41.
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
174
Intel Technology Journal, Volume 8, Issue 3, 2004
coverage. Based on this described capability, this phase
of deployment is referred to as “Portability with Simple
Mobility.”
The next phase of functionality, known as “Full Mobility”
provides incremental support for low latency, low packet
loss real-time handovers between APs at speeds of 120
km/hr or higher, both within a network and between
networks. This will deliver a rich end-user experience for
high-quality multimedia applications. Figure 1
summarizes Intel’s deployment evolution vision of the
802.16 standard.
Figure 2: Usage evolution
Figure 1: 802.16 standards and deployment evolution
To support the incremental functionality beyond fixed
access deployment, there are required enhancements to
both the air interface and network infrastructure. Both of
these enhancements must also be standardized before
interoperable services meeting end user demands can be
realized. To understand these requirements, we need to
examine usage models and service models for each stage
of 802.16 deployment. From these usage expectations, we
can then draw conclusions about required system
capabilities that must be driven into the end-to-end
architecture, interfaces, and network features. The usage
evolution is depicted in Figure 2.
Service and consumer usage of 802.16 for fixed access is
expected to mirror that of fixed wireline service with
many of the standards-based requirements being confined
to the air interface. Because communication takes place
via wireless links from customer premises equipment to
remote Non Line of Sight (NLOS) APs, requirements for
link security are increased beyond those needed for
wireline service. The security mechanisms within the
IEEE 802.16-2004 standard may be adequate for fixed
access service, but need to be enhanced for portable and
mobile applications.
An additional challenge for the fixed access air interface–
as well as subsequent portable and mobile service–is the
need to establish high-performance radio links capable of
data rates comparable to wired broadband service, using
equipment that can be self installed indoors by users, as is
the case for DSL and cable modems. Doing so requires
advanced Physical (PHY) layer techniques to achieve link
margins capable of supporting high throughput in NLOS
environments.
As 802.16 technology evolves to address portable and
mobile service, so do the feature requirements of the air
interface and RAN network, interoperability demands,
and interworking with other dissimilar networks like WiFi and 3G. The simple fact that mobile clients can
dynamically associate and perform handover across APs
crossing large, possibly discontiguous geographic regions
and operator domains, drives the need for a number of
network-related enhancements.
The simplest case of portable service (referred to as
Nomadicity) involves a user transporting an 802.16
modem to a different location. Provided this visited
location is served by wireless broadband service, in this
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
175
Intel Technology Journal, Volume 8, Issue 3, 2004
scenario, the user re-authenticates and manually reestablishes new IP connections and is afforded broadband
service at the visited location.
This usage enhancement over fixed access requires
enhancements to security such as strong mutual
authentication between the user/client device and the
network AP supporting a flexible choice of credential
types. Portable and mobile devices need a means for
authenticating trusted APs and detecting rogue APs.
Such mutual authentication is not present in the fixed
access standard. Also a common centralized mechanism
for user authentication is needed as users may move
between different APs within an IP prefix or subnet, or
across APs in different subnets, or even roam to other
service providers in different locales.
The next stage, portability with simple mobility, describes
a more automated management of IP connections with
session persistence or automatic reestablishment
following transitions between APs. This incremental
enhancement allows for more user transparent mobility
and is suitable for latency tolerant applications such as
TCP [13]; it does not provide adequate handover
performance for delay and packet loss sensitive real-time
applications such as VoIP.
In the fully mobile scenario, user expectations for
connectivity are comparable to those experienced in 3G
voice/data systems. Users may be moving while
simultaneously engaging in a broadband data access or
multimedia streaming session. The need to support low
latency and low packet loss handovers of data streams as
users transition from one AP to another is clearly a
challenging task. For mobile data services, users will not
easily adapt their service expectations because of
environmental limitations that are technically challenging
but not directly relevant to the user (such as being
stationary or moving). For these reasons, the network and
air interface must be designed up front to anticipate these
user expectations and deliver accordingly.
THE 802.16 RADIO–SCALING TO FULL
MOBILITY
The 802.16 standard provides an excellent framework
upon which systems can be built to satisfy the broad
spectrum of usage models described above. Of the three
PHY layers supported in the standard, scalable OFDMA
is the most versatile and the one preferred for operation
across channel widths ranging from 1.75 MHz to 20
MHz. Single Carrier Access (SCa) will likely be
considered for backhaul links while OFDM with 256point Fast Fourier Transform (FFT) is best suited for
Fixed Access in up to 10 MHz channel widths. Scalable
OFDMA supports features (enhanced over OFDM) that
are especially suited for high-speed mobile operation
such as Downlink (DL) and Uplink (UL)
subchannelization, fixed subcarrier spacing (by
maintaining constant ratio of FFT size to channel width),
and reduced overhead for Cyclic Prefix (CP) by keeping
its duration constant at 1/8th the OFDMA symbol
duration.
The 802.16 MAC is designed for Point-to-Multipoint
(PMP) applications and is based on Collision Sense
Multiple Access with Collision Avoidance (CSMA/CA).
The 802.16 AP MAC manages UL and DL resources
including Transmit and Receive scheduling. The MAC
incorporates several features suitable for a broad range of
applications at different mobility rates, such as the
following:
•
Four service classes–Unsolicited Grant Service
(UGS), real-time Polling Service (rtPS), non-realtime Polling Service (nrtPS), and Best Effort (BE).
•
Header suppression, packing, and fragmentation for
efficient use of spectrum.
•
Privacy Key Management (PKM) for MAC layer
security. PKM version 2 incorporates support for
Extensible Authentication Protocol (EAP).
•
Broadcast and Multicast support.
•
Manageability primitives.
•
High-speed handover and mobility management
primitives.
•
Three power management levels: Normal Operation,
Sleep, and Idle (with paging support).
These features combined with the inherent benefits of
scalable OFDMA make 802.16 suitable for high-speed
data and bursty or isochronous IP multimedia
applications.
REQUIREMENTS AND TENETS FOR A
GLOBAL INTEROPERABLE END-TOEND ARCHITECTURE FRAMEWORK
The architecture framework presented in this paper is
based on the following requirements:
•
Applicability: The architecture shall be applicable to
licensed and license-exempt 802.16 deployments.
•
Service Provider Categories: The architecture,
especially the RAN, shall be suitable for adoption by
all incumbent operator types, examples of which
were listed earlier.
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
176
Intel Technology Journal, Volume 8, Issue 3, 2004
•
•
Harmonization/Interworking: The architecture
shall lend itself to integration with an existing IP
operator core network (e.g., DSL, cable, or 3G) via
interfaces that are IP-based and not operator-domain
specific. This permits reuse of mobile client software
across operator domains.
Provisioning and Management: The architecture
shall accommodate a variety of online and offline
client provisioning, enrollment, and management
schemes based on open, broadly deployable Industry
standards.
•
IP Connectivity: The architecture shall support a
mix of IPv4 and IPv6 network interconnects and
communication endpoints and a variety of standard
IP context management schemes.
•
IP Services: The architecture shall support a broad
range of TCP and UDP real-time and non-real-time
applications.
•
Security: The architecture shall support Subscriber
Station (SS) authorization, strong bilateral user
authentication based on a variety of authentication
mechanisms such as username/password, X.509
certificates, Subscriber Identity Module (SIM),
Universal SIM (USIM), Removable User Identity
Module (RUIM), and provide services such as data
integrity, data replay protection, data confidentiality,
and non-repudiation using the maximum key lengths
permissible under global export regulations.
•
Mobility Management: The architecture shall scale
from fixed access to fully mobile operation scenarios
with scalable infrastructure evolution, eventually
supporting low latency (< 100 msec) and virtually
zero packet loss handovers at mobility speeds of 120
km/hr or higher.
•
IP Connectivity: The architecture shall support a
mix of IPv4 and IPv6 network interconnects and
communication endpoints and a variety of standard
IP context management schemes.
The architecture framework is based on the following
principles:
•
•
Extensive use of IETF standards for IP routing,
AAA, QoS and traffic engineering protocols in the
RAN and integration with an operator’s IP core/data
center,
enabling
multivendor
infrastructure
interoperability.
Functional decomposition that supports mixed
operation and scaling up from NLOS portable
operation to seamless mobility across RAN clouds
spanning multiple IP subnets or prefixes.
•
RAN architecture independence from an operator IP
core or other interconnected networks.
•
Loosely coupled interworking with 3G and Wi-Fi
networks.
•
An end-to-end security framework that is compatible
with Wi-Fi, supporting credential reuse and similar
consistent use of AAA protocols.
END-TO-END ARCHITECTURE
EVOLUTION
Figure 3 conceptually depicts the architecture evolution
for 802.16. A basic 802.16-2004-based Fixed Access
(indoor4 and outdoor) deployment is typically
accomplished via a static provisioning relationship
between an SS and an 802.16 AP. The collection of APs
and interconnecting routers or switches comprising the
RAN can be logically viewed as a contiguous cloud with
no inter-AP mobility requirements from an SS
perspective. The RAN(s) interconnect via a logically
centralized operator IP core network to one or more
external networks as shown. The operator IP core may
host services such as IP address management, Domain
Name Service (DNS) [12], media switching between IP
packet-switched data and Public Switched Telephony
Network (PSTN) circuit-switched data, 2.5G/3G/Wi-Fi
harmonization and interworking, and VPN services
(provider hosted or transit).
Going from Fixed access to Portability with Simple
Mobility involving the use of Mobile SSs (MSS) such as
laptops and Personal Device Assistants (PDA) introduces
network infrastructure changes such as the need to
support break-before-make micro- and macro-mobility5
handovers across APs with relaxed handover packet loss
and latency6 (less than two seconds), cross-operator
roaming, and the need to support reuse of user and MSS
credentials across logically partitioned RAN clouds.
Going from Portability to Full Mobility requires support
in the RAN for low (~zero) packet loss and low latency
(<100 msec) make-before-break handovers and
mechanisms such as Idle mode with paging for extended
low-power operation.
4
Indoor operation may require use of Beam Forming or Multiple Input
Multiple Output (MIMO) Advanced Antenna Systems (AAS) which are
supported in the 802.16 standard.
5
Micro-mobility refers to handovers between APs within the same IP
prefix or subnet domain. Macro-mobility refers to handovers across APs in
different IP prefix or subnet domains.
6
Latency may be unacceptable for real-time IP services such as VoIP
during handovers but acceptable for TCP and VPN services as well as
store-and-forward multimedia services.
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
177
Intel Technology Journal, Volume 8, Issue 3, 2004
An important design consideration is QoS. Fixed Access
and Portable usage models need only support acceptable
QoS guarantees for stationary usage scenarios. Portability
introduces the requirement to transfer the Service Level
Agreement (SLA) across APs involved in a handover,
although QoS may be relaxed during handovers. Full
Mobility requires consistent QoS in all operating modes,
including handovers. The 802.16 RAN will need to
deliver Bandwidth and/or QoS on Demand as needed to
support diverse real-time and non-real-time services over
the 802.16 RAN. Besides the traditional Best Effort
forwarding, the RAN will need to handle latency
intolerant traffic generated by applications such as VoIP
and interactive games.
The reference architecture, especially interconnectivity in
the RAN and interconnects to remote IP networks, is
based on extensive use of native IP suite of protocols that
in turn can deliver desired economies of scale. In the
sections below, we describe three logical entities: the
Radio Network Serving Node (RNSN), AP, and SS/MSS.
We also briefly describe the interoperability interfaces
identified in Figure 4.
The decoupling of the RAN from an operator IP core
network permits incremental migration to fully mobile
operation. An operator must however give due
consideration to the RAN topology (such as coverage
overlap, user capacity, and range) to ensure that the
physical network is future-proof for such an evolution.
Figure 4: 802.16 reference architecture
Radio Network Serving Node (RNSN)
Figure 3: 802.16 architecture evolution
END-TO-END REFERENCE
ARCHITECTURE
Figure 4 depicts an end-to-end reference architecture for
802.16. Various functional entities and interoperability
interfaces are identified. The network essentially
decomposes into three major functional aggregations: the
802.16 SS/MSS, the 802.16 RAN, and interconnect to
various operator IP core and application provider
networks. The IP core network a) manages the resources
of the 802.16 RAN, and b) provides core network
services such as address management, authentication,
service authorization, and provisioning for 802.16
SS/MSSs.
A Radio Network Service Node (RNSN) is a logical
network entity that interfaces the RAN with the operator
IP core network, Application Service Provider (ASP)
networks, and other service networks such as IP
Multimedia Subsystems (IMS), remote Enterprise
Intranets, PSTN, and the Internet. Each RNSN instance
manages a cloud of APs across a hybrid wireline/wireless
backhaul network and is responsible for Radio Resource
Management (RRM), data forwarding, and interconnects
to back-end networks. Functions such as QoS, mobility,
and security are cooperatively managed as a network of
managed APs. An RNSN may also host RAN-specific
centralized functions such as paging groups and macro
mobility agents, an example of which is a Mobile IP
(MIP) Foreign Agent (FA), and so on. An RNSN may be
rendered on a convenient network infrastructure platform
such as a Packet Data Gateway (PDG) [5] in 3GPP
networks or a Packet Data Serving Node (PDSN) in a
3GPP2 network or on a standalone router platform.
Access Point (AP)
An 802.16 Access Point (referred to in the 802.16
standard as a base station) is a logical entity that provides
the
necessary
over-the-air
standards-compliant
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
178
Intel Technology Journal, Volume 8, Issue 3, 2004
functionality including SS/MSS admission control and all
RRM and UL/DL scheduling.
We envision a number of AP/RAN topologies as depicted
in Figure 5.
Mobile/Fixed Subscriber Station (MSS/SS)
Mobile and Fixed SSs form the third most important
functional aggregation in the end-to-end framework. We
envision that most operator networks would, over time,
have to support a mix of SSs with varying degrees of
mobility support.
INTEROPERABILITY INTERFACES
Figure 4 identified several key interoperability interfaces
within the end-to-end framework. The functionality and
purpose of each of these interfaces is discussed below.
All interfaces are bi-directional unless noted otherwise.
I-SSAP and I-MSSAP
This is the control, data, management and service plane
interface between fixed-only or mobile SSs and 802.16
APs. The functions supported over this interface include,
but are not limited to the following:
Figure 5: 802.16 RAN topologies
An AP may form a subnet/prefix boundary as indicated
by an AP Router (APR) in the figure. An AP may be
implemented as an integrated MAC/PHY entity or may
take on a more distributed architecture involving an AP
Controller (APC) and AP Transceivers (APT) that would
render cells in groups.
A combination of an APC with one or more APT
instances may render a multisector cell. Where multiple
APTs are managed by an APC, the APC may host a
common MAC instance across all APTs or a dedicated
MAC instance for each APT. An APC would typically
localize all micromobility functions across its managed
APTs and as such would support all relevant 802.16
PHY, MAC, and Convergence Sublayer (CS) Service
Access Point (SAP) primitives. An APC may also host
optional wireless link services such as header
suppression, payload compression, and MSS paging.
An AP hosting more than one logical APC instance can
optimize control and management plane functions across
all hosted instances. Factors such as projected scalability
requirements (coverage, user density), degree of mobility,
and need for incremental network growth would drive an
operator’s choices between the different AP
configurations. However, the architecture framework is
agnostic to specific RAN topologies and can support a
mix of all possible variants simultaneously.
•
SS/MSS connectivity provisioning and admission
control
•
Over-the-air and end-to-end security
•
Mobility management
•
Device management
•
UL and DL data exchange
•
Authorization and tunneling for specialized IP
services
•
Application layer end-to-end signaling
•
Advanced functions such as power management
(paging), compression, data reliability
As noted earlier, the 802.16 standard presents a rich
selection of optional features that in turn presents
significant interoperability challenges to the Industry. We
expect the WiMAX Forum to define profiles targeting
operation in specific frequency bands, channel widths,
PHY modes, and duplexing modes to drive multivendor
interoperability. All such applicable profiles will be
incorporated in the I-SSAP and I-MSSAP interfaces.
I-CN1 and I-CN2
I-CN1 represents the control, data, and management
planes between 802.16 RANs and an operator’s core
network (with interfaces in turn to other remote
networks). I-CN2 represents control, management, and
service planes to ASP networks. Both of these interfaces
are exposed by the RNSN and enable a consistent all-IP
interface to diverse core networks. The functions
modeled over this interface may be provided by a cluster
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
179
Intel Technology Journal, Volume 8, Issue 3, 2004
of servers, for example, DHCP, DNS, IMS Core Network
components such as Proxy-Call Session Control Function
(P-CSCF), Interrogating-CSCF (I-CSCF), Serving-CSCF
(S-CSCF), Media Gateway (MGW), and so on. These
interfaces may also host IP tunnels to carry data between
provider networks.
The functions supported over this interface include, but
are not limited to the following:
•
•
•
•
•
•
Assignment of traffic engineering parameters for
provisioned QoS for both control and data plane
traffic.
User authentication via AAA intermediaries and
servers.
Services authorization, access control, and charging.
IP connectivity management and security (for
example, domain firewall).
Troubleshooting network access problems,
application-specific problems and RAN event
handling.
Data traffic and macro mobility management.
I-RNSN
This is the control, data, and management plane interface
between two RNSNs that logically may demarcate two
RAN clouds. The interface typically handles inter-RNSN
mobility management control and data plane traffic
(including temporary data tunneling between RNSNs
serving Serving and Target APs during handovers).
I-RNSNAP
This is the control, data, and management plane interface
between an AP (or any of its control plane variants) and
an RNSN. This interface demarcates the two endpoints of
the RAN across which intra-RAN micro- and macromobility functions are performed. The interface also
supports functions such as paging.
Mobility Management
The 802.16-2004 standard defines a BS as a single sector
entity supporting one frequency assignment. The 802.16e
amendment defines MAC message primitives to support
network or MSS initiated handovers. The very basic
handover scenario for a real-world multisector AP would
be an inter-sector handover. The amendment defines
handover optimization flags representing levels of
handover context information that is shared between
neighbor AP entities (sector line cards in a multisector
AP or between the sector line cards in two different APs).
The optimization flags consequently enable modeling of
all possible handover scenarios from the most basic
nomadic access scenario (where no network entry context
is shared between APs across a handover) to scenarios
involving inter-subnet, inter-frequency assignment, Idle
mode, and inter-physical AP handovers. Furthermore,
optional advanced features such as Soft handover (with
PHY layer macro diversity) and Fast Base Station
Switching are being defined to support zero packet loss,
low latency inter-sector handovers. The design goal for
mobility management is to build on these primitives to
deliver the desired handover performance. Fixed access
and nomadic access require no handover support.
Portability implies fast intra-RAN switching with
potential data loss during handovers and even more
latency and data loss during inter-subnet handovers. Full
mobility requires zero/low packet loss and low latency
handovers that are acceptable to real-time applications
such as VoIP.
The end-to-end reference architecture classifies mobility
management into macro-mobility and micro-mobility as
illustrated in Figure 6. Within the RAN, this paper
recommends the use of Multiprotocol Label Switching
(MPLS) [11] or IP-in-IP tunneling with Diffserv [10]
provisioned QoS to switch data paths across traffic
engineered backhaul links during handovers for micromobility. With MPLS, we recommend fast preprovisioned Label Switched Paths (LSP) switching
between the RNSN and AP/APC that perform the role of
Label Edge Routers (LER). Efficient MAC layer
handover triggers and limited micro-mobility signaling
would be used to initiate traffic forwarding/multiple
unicasting and switching to minimize handover latency
and data loss between RNSN and AP/APC. For macromobility this paper recommends the use of SIP mobility
for real-time low-latency interactive applications such as
VoIP, and Mobile IP for all other generic applications. In
the case of SIP mobility, an IMS can overlay on top of
the end-to-end framework via the RNSN defined above.
The RAN can leverage the IP Differentiated Services
QoS model or MPLS-based traffic engineering
technologies to provide appropriate forwarding treatment
to end-user traffic flows as they traverse between an
RNSN and APs.
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
180
Intel Technology Journal, Volume 8, Issue 3, 2004
•
Support reuse of credentials and cryptographically
strong bilateral authentication and session key
management across these networks.
•
A provisioning and access framework for advanced
IP services that is compatible with the architecture
for Wi-Fi hotspots.
•
Enable offering of multiple IP services with
attributes such as provisioned bandwidths, SLAs,
QoS, and variable tariff profiles.
The all-IP architecture framework for Wi-Fi hotspots and
802.16 permit both loosely and tightly coupled
harmonization scenarios. Figure 7 conceptually depicts
these two forms of interworking.
Figure 6: Mobility management
Harmonization and Interworking with
Public Wi-Fi and 3G Networks
As noted earlier, different incumbent operators are likely
to deploy 802.16 networks either as a data overlay
network or as a standalone broadband access network.
Integration with an existing operator network would
involve either harmonization or interworking as defined
below.
Interworking implies a technical and business
relationship between operators owning homogenous or
heterogeneous networks enabling subscribers to
authenticate/authorize to their home operator network via
the “visited” network and utilize system functions and IP
services offered by both networks.
Harmonization on the other hand is a situation where
two or more homogeneous or heterogeneous networks
owned by an operator are offered as an integrated
network to users.
The document
http://www.intel.com/technology/IWS/WLAN_study.pdf
describes Intel’s proposed interworking framework for
public Wi-Fi hotspots. We recommend adopting and
extending the same principles for inter-operator 802.16
interworking, supporting the following goals:
•
Figure 7: Loose and tight coupling of Wi-Fi and
802.16 networks
The loosely coupled framework is preferred in scenarios
involving interworking between 802.16 networks and WiFi hotspots managed by different operators. Other
technical considerations are similar to what has been
proposed for Wi-Fi roaming and inter-operator
interworking in the reference cited above.
This paper recommends a loosely coupled integration
approach for 802.16 and 2.5G/3G networks. Both 3GPP
[4, 5] and 3GPP2 have ongoing efforts to develop an
interworking architecture between Wi-Fi hotspots and
2.5G/3G networks. The loosely coupled interworking
model is consistent with the developments in these two
organizations. Figure 8 depicts the interworking model
for 3GPP and 3GPP2 networks.
An operator type–agnostic one-bill roaming (via
common, extensible RADIUS [6] and DIAMETER
[7] accounting primitives) framework across 802.16
networks–eventually leading to seamless IP services
mobility across these networks.
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
181
Intel Technology Journal, Volume 8, Issue 3, 2004
Figure 8: Reference model for 802.16 interworking
with 2.5G/3G
3GPP has defined a Public Wi-Fi IP interworking entity
called the Packet Data Gateway (PDG) to be incorporated
in Release 6. With adaptations as needed based on
functional requirements, the PDG can serve as the ingress
to the operator IP core network (the 802.16 core
network).
3GPP2 has a similar ongoing effort for Wi-Fi-3GPP2
interworking and will also identify a transport and
signaling gateway that essentially supports integration of
a 802.16 RAN into a 3GPP2 IP core network.
Note that while the RNSN is shown as a separate logical
entity in Figure 8, most if not all of its functions on the IP
core interconnect interface may be entirely subsumed by
the PDG or PDSN while functions on the RAN interface
may be subsumed by one or more APs.
End-to-End Security
Figure 9 conceptually depicts end-to-end Authentication,
Authorization, and Accounting (AAA) on 802.16
networks supporting portability and fully mobile
operations. The figure borrows terminology from Wi-Fi
and is built on the three-party protocol (PKM v2)
foundation being defined in 802.16e.
Figure 9: 802.16 security framework
As shown in this figure, over-the-air authentication and
encryption (security association) is established using the
PKM-EAP protocol. Extensible Authentication Protocol
(EAP) is carried over RADIUS or DIAMETER to the
AAA backend. The use of EAP enables support for
cryptographically strong key-deriving methods such as
EAP-AKA and EAP-MSCHAPv2. Intel also recommends
using an end-to-end tunneling protocol such as Protected
EAP (PEAP) or Tunneled TLS (TTLS) to afford mutual
authentication and 128-bit or better Transport Layer
Security (TLS) encryption to further enhance end-to-end
security (especially in situations where cryptographically
weaker EAP methods may be deployed). The AP or APC
or APR serves as the “Authenticator” and hosts a
RADIUS or DIAMETER AAA client. All AAA sessions
are terminated on an AAA server that may be in the
operator’s IP core network or an external IP network in
roaming scenarios. The RNSN is merely a conduit for the
AAA messages and does not play a significant role in the
AAA process. In some instances, the network may
employ an AAA aggregator/intermediary but the
architecture is not impacted in those cases. Additionally,
the RNSN may host a firewall to filter downstream traffic
to a RAN.
THE WIMAX FORUM
In order for the defined IEEE 802.16 broadband wireless
network architecture to become a reality, service
providers must be assured that multivendor BS/SS
interoperability is verified by an independent certification
lab. The WiMAX Forum is a non-profit consortium of
broadband wireless system vendors, service providers,
component suppliers, and operators focused on enabling
the development and deployment of interoperable
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
182
Intel Technology Journal, Volume 8, Issue 3, 2004
broadband wireless products around the world. Today,
the consortium is focused on the development of
conformance, interoperability, and certification of APs
and SSs for Non Line of Sight (NLOS) operation below
11 GHz based on the IEEE 802.16 standard.
Figure 10: WiMAX certification process
(preliminary)
The following section is divided into two main parts.
First, the WiMAX conformance and interoperability
processes are explained, and the WiMAX system profiles
for certification are discussed. Second, the challenges
facing the WiMAX certification process, including
certification lab set-up and the development of the
WiMAX protocol analyzer, are discussed.
WIMAX CERTIFICATION PROCESS
Conformance vs. Interoperability
WiMAX conformance should not be confused with
interoperability. However, the combination of these two
types of testing make up what is commonly referred to as
certification testing. WiMAX conformance testing can be
done by either the certification lab or another test lab and
is a process where BS and SS manufacturers will be
testing their pre-production or production units to ensure
that they perform in accordance with the specifications
called out in the WiMAX Protocol Implementation
Conformance Specification (PICS) document. Based on
the results of conformance testing, BS/SS vendors may
choose to modify their hardware and/or firmware and
formally re-submit these units for conformance testing.
The conformance testing process may be subject to a
vendor’s personal interpretation of the IEEE standard, but
the BS/SS units must pass all mandatory and prohibited
test conditions called out by the test plan for a specific
system profile.
On the other hand, WiMAX interoperability is a
multivendor (≥3) test process hosted by the certification
lab to test the performance of BS and/or SS from one
vendor to transmit and receive data bursts from another
vendor BS and/or SS based on the WiMAX PICS. Figure
10 shows the preliminary WiMAX certification process
with its components. First, the vendor submits BS/SS to
the certification lab for Pre-Certification Qualification
testing where a subset of the WiMAX conformance and
interoperability test cases is done. These test results are
used to determine if the vendor products are ready to start
the formal WiMAX conformance testing process. Upon
successful completion of the conformance testing, the
certification lab can start full interoperability testing.
However, if the vendor BS/SS failed some of the test
cases, the vendor must first fix or make the necessary
changes to his products (BS, SS) and provide the
upgraded BS/SS with the self-test results to the
certification lab before additional conformance and
regulatory testing can be done. If the BS/SS vendor fails
the interoperability testing, the vendor must make the
necessary firmware/software modifications and then resubmit his products with the self-test results for a partial
conformance testing depending on the type of failure and
the required modification. The end goal is to show service
providers and end users that as WiMAX Forum Certified
hardware becomes available, service providers will have
the option to mix and match different BSs and SSs from
different vendors in their network in their deployments.
Upon successful completion of the described process
flow, the WiMAX Forum would then grant and publish a
vendor’s product as WiMAX Forum Certified. It should
be pointed out that each BS/SS must also pass regulatory
testing, which is an independent parallel process to the
WiMAX certification process.
Abstract Test Suite Process
The WiMAX Forum is working on the development of
numerous process and procedural test documents under
the umbrella of the IEEE 802.16 standard. The key
WiMAX test documents are as follows:
•
Protocol Implementation Conformance Specification
(PICS) in a table format.
•
Test Purposes and Test Suite Structure (TP and
TSS).
•
Radio Conformance Test Specification (RCT).
•
Protocol Implementation eXtra Information for
Testing (IXIT) in a table format.
Figure 11 shows how these test documents are used in the
development of a standardized Abstract Test Suite (ATS).
The ATS is the culmination of test scripts written in a
Tree and Tabular Combined Notation (TTCN) language.
The end product of the ATS are test scripts for
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
183
Intel Technology Journal, Volume 8, Issue 3, 2004
conformance and interoperability testing under a number
of test conditions called out in the PICS document for a
specified WiMAX system profile. The development of
the first set of available test scripts is planned for the
fourth quarter of 2004. With available test scripts, the
manual WiMAX certification testing will eventually
become an automated process.
As previously mentioned WiMAX defines interoperable
system profiles between the BS and SS, which are
targeted for licensed and licensed-exempt frequency
bands used around the world. Table 1 lists only the first
stage of the basic system profiles that will be used for
WiMAX certification. This list is limited initially to 3.5
GHz licensed (international) and 5.8 GHz license-exempt
frequency bands. Data bursts can be transmitted using
either FDD or TDD schemes. In the TDD scheme, both
the UL and DL share the same channel, but do not
transmit simultaneously, and in the FDD scheme, the UL
and DL operate on different channels, sometimes
simultaneously. The second stage of profiles is pending
regulatory and service providers contributions. WiMAX
system profiles with 5 MHz channel bandwidth at 2.5
GHz frequency band (i.e., MMDS) using either TDD or
FDD schemes are planned to be added in the second
stage.
Protocol Implementation
Standardized
Abstract Test Suite
(in TTCN)
ATS
RCT
Profiles Conformance Statement
(tables)
PICS
1st Stage Profile
Configuration
Profile Name
3.5GHz, TDD, 7MHz
3.5T1
3.5GHz, TDD 3.5MHz 3.5T2
WiMAX System Profiles
Standardized Test
Standard Purposes (in English)
TPs
Table 1: First-stage system profiles for WiMAX
certification
Radio Conformance
Test Specification
IXIT
Protocol Implementation
eXtra
Information for Testing (tables)
Figure 11: Abstract test suite development process
3.5GHz, FDD, 3.5MHz 3.5F1
3.5GHz, FDD, 7MHz
3.5F2
5.8GHz, TDD, 10MHz 5.8T
What is Certified?
As described above, certification is a combination of
conformance and interoperability testing scripts based on
selected profiles with test conditions specified from the
PICS document. The selection of test cases for
certification is currently in development by the WiMAX
Forum. Development of the certification program is one
of the many activities under the auspices of the WiMAX
Certification Working Group (CWG). Certification
testing is intended only for complete systems such as a
BS or an SS, not individual solution components such as
radio chips or software stacks. The introduction of BS/SS
reference designs may also be considered for testing to
show that the design conforms to the IEEE 802.16
specification and is interoperable with other WiMAX
Forum Certified equipment, but will not preclude any
requirement for a system vendor using components from
the reference design from having to submit their product
for certification testing. For portable and mobile
platforms, various vendors are expected to client-based
cards introduce later on that plug into a notebook or
another portable platform. Such products will necessitate
submission of the client-based cards with a notebook for
testing similar to what has been done by a Wi-Fi
certification lab. While much work still needs to be done,
as the IEEE 802.16e standard becomes more stable, the
working groups within the WiMAX Forum will continue
to lay the framework for test integration and certification
in migrating from supporting IEEE 802.16d towards the
introduction of IEEE 802.16e-based products.
Certification Challenges
There are two main challenges facing the WiMAX
certification process. The first challenge is to establish a
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
184
Intel Technology Journal, Volume 8, Issue 3, 2004
WiMAX certification lab with all the necessary resources
and equipment. The second challenge is to have all the
necessary specialized test equipment such as a Protocol
Analyzer (PA) ready for use by the certification lab.
functions in order to aid the diagnosis of failed test
cases.
Test setup
DUT
GPIB
Certification Lab Set-up
Establishing a WiMAX certification lab presents several
unique and important requirements to be successful.
Since this new technology is based on an open standard,
the test-bed must be validated before the certification can
be started. To accomplish this, the following key issues
must be addressed:
•
Availability of BS/SS from different vendors with
different Si solutions.
•
Specialized test equipment to analyze, track, and
report test results.
•
Integration of testing methodology with the vendor
hardware, test equipment, and test scripts called out
in the test plan.
Intruder
GPIB
Signal
Analysis
PA
PA
UL Trigger
DL Trigger
Signal
Analysis
BS
BS
BS
SS
SS
SS
SS
Wireless Channel
Impairments
Traffic Generator
Controller
LAN/GPIB
DHCP
TFTP
TOD
•
Establishing a baseline of acceptable test results from
available hardware in the test bed.
Figure 12: WiMAX protocol analyzer test-bed
configuration
•
Ability to replicate the test configuration so vendors
can conduct their own pre-testing.
In the second stage of the PA development, it is expected
that the PA will be able to emulate either the BS or SS in
order to analyze the prohibited test cases.
Protocol Analyzer Development
The WiMAX Forum is facilitating the development of a
PA through a third party to help analyze the transmitted
DL and UL IP packets between a BS and SS based on the
WiMAX PICS document. Figure 12 shows, for example,
a WiMAX test-bed configuration using the PA. In this
configuration, the controller turns test scripts into test
commands, which are then issued to the traffic simulator,
PA, and Device Under Test (DUT). The PA development
challenge is the system integration of a modified BS
hardware platform with different radios for both licensed
and license-exempt frequency bands with a software
emulation tool. The key features of the PA system
include the following:
• Data packet capture and display
o Display multiple levels of information (summary,
decode tree, raw data packets, etc.).
o Ability to correlate capture data with test results.
• Display of message sequence charts.
• Ability to trigger on packet content (protocol, field
values, patterns) and on extended sequences of
events.
• Display of statistics of collected data.
• Generation of summary and detailed diagnostic test
automated alarm generation capability.
• Support of a flexible scripting interface that enables
users to create custom scripts and to control PA
In conclusion, the building blocks for the WiMAX
certification process, which include both conformance
and interoperability testing, were reviewed. The key
challenges facing the Industry today include setting up
the WiMAX certification lab with the PA to validate their
test-bed using BS/SS from different equipment vendors.
Furthermore, the participation of multiple vendors in
public plugfest events is critical to ensure the Industrywide acceptance of WiMAX certified units.
CONCLUSION
Although wireless networks and radio coverage in general
have proliferated over the years, data service offerings
continue to be either limited in range (as in 802.11) or
deficient in data speed and cost as in Wireless Wide Area
Networks (WWANs). Wireless data rates for WWANs
are limited and are of high-cost partly due to the
inherently granular physical and network layer
specifications that burden the WWAN RAN and core
switching fabric, and partly due to the limited available
bandwidth for operation. As extended battery life and
reduced size of laptops affords increased portability, so
does the need for ubiquitous connectivity with rich data
content at affordable prices become more urgent. By
delivering a combination of higher modulation schemes
within greater channel bandwidths and link budget
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
185
Intel Technology Journal, Volume 8, Issue 3, 2004
margins that are comparable to wide area wireless
systems, IEEE 802.16 is uniquely positioned to extend
broadband wireless beyond the small islands of service
afforded by Wi-Fi systems today. Incremental evolution,
from Fixed access to Portability and then to Full
Mobility, with laptops and PDAs enabled with IEEE
802.16, furthers Intel’s vision of coupling wireless
connectivity and computing in a single processor
platform. The set of ongoing activities outlined in this
article, a PHY and MAC layer specification that unites
the market behind a common set of standards, a flexible
end-to-end network architecture that is coupled with a
coherent service vision, and an efficient certification
process that enables interoperability, are key enablers for
realizing the WiMAX vision.
ACKNOWLEDGMENTS
The authors thank Gerald Lebizay, Glenn Begis, Tim
Teckman, and David Putzolu for their review of this
paper.
REFERENCES
[1] “Part 16: Air Interface for Fixed Broadband Wireless
Access Systems,” IEEETM P802.16-REVd/D5-2004.
[2] “Part 16: Air Interface for Fixed and Mobile
Broadband Wireless Access Systems,’ IEEETM
P802.16e/D3-2004.
[3] “WiMAX PICS for WirelessMAN-OFDM and
WirelessHUMAN(-OFDM) Rev.7f (2004).”
[4] “3GPP TS 22.234–Requirements on 3GPP System to
WLAN Interworking (Release 6).”
[5] “3GPP TS 23.234–3GPP system to WLAN
Interworking; System Description (Release 6).”
[6] RFC 2865, “Remote Authentication Dial In User
Service,” Related RFCs at
http://www.freeradius.org/rfc/*
[7] RFC 3588, “DIAMETER base protocol.”
[8] RFC 3344, “IP Mobility support for IPv4.”
[9] RFC 3775, “Mobility support in IPv6.”
[10] RFC 2475, “An architecture for differentiated
services.”
[11] RFC 3031, “Multiprotocol label switching
architecture.”
[12] DNS RFCs can be found at
AUTHORS’ BIOGRAPHIES
Ed Agis is a market development manager for the
Wireless Broadband Division (WBD) at Intel
Corporation. He is also the co-chair of the WiMAX
Forum Certification Working Group and a member of the
WiMAX Technical and Marketing Working Groups. Ed
holds a B.Sc. degree from the Air Force Academy
graduating Magna cum Laude as well as a Masters of
Business Administration in Management and another in
Operations/Product Marketing from Amber University.
His e-mail is ed.agis at intel.com.
Henry Mitchel is a systems architect in Intel’s Modular
Communications Platform Division within the
Communications Infrastructure Group specializing in
chip architectures, firmware, protocols, and standards,
and their impacts on systems architecture. Prior to joining
Intel he was director of R&D at DataStorm Technologies,
Inc., makers of PROCOMM PLUS*. He holds a BS
degree from the Massachusetts Institute of Technology
and an MS degree from the University of Missouri. His email is henry.mitchel at intel.com.
Shlomo Ovadia received a Ph.D. degree in Optical
Sciences from the Optical Sciences Center, University of
Arizona in 1984. He held various technical positions at
IBM, Bellcore, and General Instruments before joining
Intel in 2000 as principal architect in CTG, where he was
leading the effort on the architecture, design, and
development of optical burst switching in enterprise
networks. Currently at ICG, Shlomo is leading Intel’s
WiMAX interoperability and certification effort for IEEE
802.16d/e-based wireless products. He is the author of a
recently published book titled Broadband Cable TV
Access Networks: From Technologies to Applications
(Prentice Hall, 2001). He is a senior member of
IEEETM/LEOS/COMSOC with more than 70 technical
publications and conference presentations. He is the
holder of 35 patents, and his personal biography is
included in the Millennium edition of Who’s Who in
Science and Engineering (2000/2001). His e-mail is
shlomo.ovadia at intel.com.
Selim Aissi is lead MID and security architect in the
Virtualization and Trust Lab at Intel’s Corporate
Technology Group. He also leads standards efforts in
3GPP. Before joining Intel in 1999, he worked at the
University of Michigan, General Dynamics’ M1A2
Battlefield Tank Division, General Motors’ Embedded
Controller Excellence Center, and Applied Dynamics
International. Selim serves on the review board of several
http://www.dns.net/dnsrd/rfc/*
[13] RFC 793, “Transmission Control Protocol.”
*
Other brands and names are the property of their
respective owners.
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
186
Intel Technology Journal, Volume 8, Issue 3, 2004
publications and conferences, including ACM CCS, ACM
SWS, and he is the vice-chair of the Security and
Management (SAM) Conference. He holds a Ph.D.
degree in Aerospace Engineering from the University of
Michigan and is a senior member of the IEEE and ACM.
His e-mail is selim.aissi at intel.com.
Sanjay Bakshi is an 802.16e network architect in Intel’s
Mobile Networking Lab within the Corporate Technology
Group. Prior to joining the Mobile Networking Lab,
Sanjay was engineering manager and architect in the
Performance Networking Lab within the Corporate
Technology Group. He has led a number of projects
related to the usage of the Intel Internet Exchange
Processor in various fields such as IP routing, MPLS, 3G
wireless, and next-generation control plane architecture.
Sanjay received his B.E. degree in Computer Science
from the Regional Engineering College, Tiruchirapalli,
India. His e-mail is sanjay.bakshi at intel.com.
Prakash Iyer is a senior staff architect in Intel’s Mobile
Networking Lab within the Corporate Technology Group.
He is an active member of the IEEE 802.16 Working
Group including chair for the Handoff Adhoc group. He
leads standards efforts in the IEEE, IETF, 3GPP, and
3GPP2 on heterogeneous wireless interworking and
directs architecture, prototyping, and simulation efforts
for seamless networking–including 802.11 and 802.16.
He holds B.S. degrees in Physics and Electrical and
Computer Engineering and an M.S. degree in Computer
Science. His e-mail is prakash.iyer at intel.com.
industry specifications subgroup for mobile broadband
wireless architecture. He received his Bachelors degree in
engineering from Georgia Institute of Technology and
holds an MBA degree from Carnegie Mellon University.
His e-mail is chris.b.rogers at intel.com.
James Tsai is a wireless network and mobile platform
architect in Intel’s Mobile Networking Lab within the
Corporate Technology Group. His research work has
focused on wireless network architectures (Wi-Fi,
WiMAX, and WWAN) and next-generation mobile
platform technologies such as extended mobile access
technology and multi-radio subsystems. He received his
B.S degree in Electrical Engineering from the Chinese
Culture University in Taiwan and an M.S. degree in
Computer Science from Columbia University. His e-mail
is james.tsai at intel.com.
Copyright © Intel Corporation 2004. This publication
was downloaded from http://developer.intel.com/.
Legal notices at
http://www.intel.com/sites/corporate/tradmarx.htm.
Masud Kibria is 802.16e initiative manager within
Intel’s Communications Technology Lab focused on the
technical validation of 802.16e. Prior to joining Intel,
Masud led various strategic projects at AT&T Wireless
including WCDMA evolution to HSDPA, Interoperability
for 2.5G/3G Networks, WLAN, Wireless-PBX, emerging
technology feasibility studies, etc. Previously, he has
worked on various theoretical and practical aspects of
coverage, capacity, and interference for mobile wireless
systems and has led regional teams in WWAN design,
implementation, and operation. Masud received his BSEE
degree from the University of Maryland. His e-mail is
masud.kibria at intel.com.
Christopher B. Rogers is a WWAN technology
strategist within Intel’s Communications Technology
Labs. Prior to his current focus on 802.16 technology, he
was marketing director and co-founder of Intel’s
Ultrawideband Wireless Group and has also held business
development and marketing roles in other WPAN- and
WWAN-related businesses within Intel. Chris is a
member of IEEE 802.15 and 802.16 and chairs an
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
187
Intel Technology Journal, Volume 8, Issue 3, 2004
THIS PAGE INTENTIONALLY LEFT BLANK
Global, Interoperable Broadband Wireless Networks: Extending WiMAX Technology to Mobility
188
RF System and Circuit Challenges for WiMAX
Balvinder Bisla, Intel Communications Group, Intel Corporation
Roger Eline, Intel Communications Group, Intel Corporation
Luiz M. Franca-Neto, Intel Communications Group, Intel Corporation
Index words: WiMAX, I/Q, IF, FDD, TDD
ABSTRACT
Broadband Wireless Access has occupied a niche in the
market for about a decade, but with the signing of the
802.16d standard it could finally explode into the mass
market. Intel’s baseband transceiver chip is flexible
enough to accommodate Radio Frequency Integrated
Circuit (RFIC) architectures of today and the future.
With the emergence of this standard an ecosystem is
developing that will allow multiple vendors to produce
components that adhere to a standard specification and
hence allow large-scale deployment. One of the major
challenges of the 802.16d standard is the plethora of
options that exist; Worldwide Interoperability
Microwave Access (WiMAX) will address this issue by
limiting options and hence ensure interoperability. The
result will allow manufacturers of Radio Frequency
(RF) components and test equipment to have their
products used for mass deployment.
In this paper, we focus on the various RF challenges
that exist on a RF system-level and show how such
challenges can translate into circuit designs. The RF is
made more complicated by the fact that WiMAX indeed
addresses wireless markets across the world both in
licensed and unlicensed bands. Thus, solutions have to
be flexible enough to allow for the many RF frequency
bands and different regulations around the globe.
Several major RF architectures are discussed and the
implications for WiMAX specifications are explored, in
particular both Intermediate Frequency (IF)- and I/Qbased structures are investigated.
Part of our discussion will provide insight into the cost
and performance tradeoffs between Time Division
Duplex (TDD) and Frequency Division Duplex (FDD)
systems both in licensed and unlicensed bands. It is
generally accepted that TDD systems offer cost
advantages over their FDD counterparts; however, most
licensed bands intended for data applications operate
with FDD systems in mind. Some of the RF subsystem
RF System and Circuit Challenges for WiMAX
blocks that have stringent WiMAX specifications are
also elaborated upon: these include synthesizers, power
amplifiers, and filtering. These fundamental subsystem
blocks are where most of the transceiver costs reside;
the same blocks are also responsible for most of the RF
performance.
The industry is moving towards using Orthogonal
Frequency Division Multiplexing Access (OFDMA) and
either spatial diversity or beam forming techniques to
enhance link margins. We touch on the RF challenges
associated with these techniques. Finally, we view some
of the important WiMAX specifications for RF and the
implications for the design of RF circuits, which include
SNDR, channel bandwidths, RF bands, noise figures,
output power levels, and gain setting. Some important
differences between WiMAX and 802.11 RF
specifications are also highlighted.
INTRODUCTION
As the RF challenges mount so do the costs of the
Radio. For WiMAX to be successful the cost vs.
performance equation has to be balanced carefully. Two
extreme examples of this cost and performance equation
are a Single In Single Out (SISO) system from Hybrid
Networks (now defunct) requiring Line of Sight (LOS)
radios. LOS radios result in truck rollouts utilizing
experienced technicians to set the equipment up.
However the cost of the radio is low due to its
simplicity. In general, the SISO radio requires expensive
installation and reliability is poor; link margins are
typically 145 dB. On the other hand, Iospan Wireless
(now defunct) demonstrated a Multiple In Multiple Out
(MIMO) radio with a 3x2 system; i.e., three receive and
two transmit chains. It was able to support link margins
of 165 dB that could penetrate inside homes in
multipath environments. With this ability, the issue of
costly truck rollouts is eliminated; however, the cost of
the multiple radio chains becomes a deterrent. Still, as
Radio Frequency Integrated Circuit (RFIC) integration
189
Intel Technology Journal, Volume 8, Issue 3, 2004
improves, costs will head down. WiMAX, through the
use of integration and advanced techniques to increase
link margins, should be able to achieve reliable wireless
systems at a reasonable cost.
RF ARCHITECTURES
This section describes the plethora of tradeoffs and
challenges for RF architectures for WiMAX-related
radios. We discuss Frequency Division Duplex (FDD)
and its cousin, Half FDD (HFDD) as well as Time
Division Duplex (TDD). Intermediate Frequency (IF),
Direct Conversion or Zero Intermediate Frequency
(ZIF) as well as variants of these are presented. The
interface between the Baseband (BB) chip and the radio
must be carefully designed, so these challenges are
exposed. Methods to improve Link Margins, namely
MIMO, and beam forming can be used in WiMAX. In
addition, OFDMA, which allows for subchannelization,
improves capacity efficiency. We discuss the RF
challenges inherent in the use of these methods.
TDD/FDD and HFDD Architectures
TDD
Figure 1 shows a TDD radio. The darkened blocks are
the most costly in the radio. TDD systems utilize one
frequency band for both Transmit and Receive. This
concept requires only one Local Oscillator (LO) for the
radio. In addition only one RF filter is necessary and
this filter is shared between the Transmitter (TX) and
the Receiver (RX). The synthesizer and RF filters are
major cost drivers in radios. Having one synthesizer
saves on die area; a large part of the radio die size can
be taken up by the LO, in particular the inductor, which
is part of the resonant structure.
systems. The Medium Access Control (MAC)-level
software tends to have a more complicated scheduler
than an FDD system since it must deal with
synchronizing many users’ time slots in both TX and
RX mode. It must be noted that while the RF filtering
specifications are relaxed, this tends to imply that
subscriber stations will have to be spaced further apart
from each other to avoid interference. In essence, the
system must handle fewer users in a given area than in
FDD systems.
TDD systems are most prominent in unlicensed bands;
in these bands the regulations for output noise are more
relaxed than in licensed bands. Thus, inexpensive RF
filters can be specified. Since the unlicensed bands are
free of cost there is competition to drive for the lowest
cost architecture, TDD.
FDD
Figure 2 shows an FDD radio. A high-performance RF
front-end is required in FDD systems. Collocation
issues from a TX noise perspective are solved since the
worst-case scenario of self jamming is not possible.
FDD systems do not have to switch the RX or TX; this
alleviates settling time specifications, which results in a
simpler radio design. The MAC software is simpler
because it does not have to deal with the time
synchronization issues as in TDD systems.
The radio must be capable of data transmission while in
Receive mode without incurring any degradation in Bit
Error Rate (BER). To ease the burden on the filter there
is a gap between the TX frequency band and the RX
band; however, carriers wish to minimize this space.
Typically this is a separation of 50 MHz to 100 MHz.
The RF filter in a TDD system is not required to
attenuate its TX noise as severely as in FDD systems.
The TDD mode prevents the TX noise from self
jamming the RX since only one is on at any time. As
well as relief of the RF filter specifications, having just
one RF filter saves cost and space. It should be noted
that to ensure Transmitting radios do not interfere with
nearby Receiving radios, the specification for TX noise
cannot be eased with abandon. The Transmission noise
from Radio 1 will interfere with the Received signal of
Radio 2. Thus, although self-jamming specifications are
made easier, collocation specifications must be carefully
considered. There is a notable savings in power from the
TDD architecture, a direct result of turning the RX off
while in TX mode and vice versa.
Several disadvantages exist, however. There is a
reduction of data throughput since there is no
transmission of data while in RX mode unlike FDD
RF System and Circuit Challenges for WiMAX
190
Intel Technology Journal, Volume 8, Issue 3, 2004
Figure 1: TDD radio
Figure 2: FDD radio
Figure 3: HFDD radio
We try to specify the TX noise to be 10 dB below the
RX input noise floor, in which case the TX noise will
only degrade the RX by 0.5 dB. Unfortunately the
specifications usually tie FDD systems to using cavity
RF System and Circuit Challenges for WiMAX
filters or 4-pole ceramic filters. Cavity filters run in the
order of $35 each while ceramic filters can be in the $8
range. Most licensed bands do not have one standard
structure but are flexible; i.e., the TX and RX could be
191
Intel Technology Journal, Volume 8, Issue 3, 2004
swapped in different geographical regions. This results
in having to design several flavors of the filters,
something that does not lend itself to mass production
of the filters.
Once again the collocation issues have to be addressed
carefully. Self jamming is not a problem as in TDD but
then too much relief on the TX filter can result in
interference between users.
To give an idea of the filter requirements in FD:
There is also a capacity loss at the Subscriber Station
since the radio cannot simultaneously Transmit and
Receive.
Filter_rej (dB) = Po(dBm/Hz) – Mask (dBc)-[174+NFcochannel_rej]
For example, if power output Po = -33 dBm/Hz, in a 1
MHz signal bandwidth, output power is +27 dBm.
Mask of TX is = 60 dBc; i.e., the thermal floor of TX is
60 dB below the Po.
NF is Noise Figure of Receiver = 5 dB.
CoChannel_rej is how far in dB is the undesired signal
below the desired signal. = 10 dB i.e., the undesired
signal is 10 dB below the desired signal.
We get Filter_rej at the RX frequency of 86 dB. If the
RX is 100 MHz away from the TX, this filter is an
expensive cavity filter.
The full-duplex nature of the circuit requires a separate
TX and RX synthesizer. The RFIC die area is
significantly impacted by the inductor of a resonant
circuit; this is part of a Voltage Controlled Oscillator
(VCO) which is used in the synthesizer. Thus, two of
these have a large impact on the cost of the RFIC.
A final note on FDD systems is that they are power
hungry; this also increases the cost of the power system.
Thus, FDD is not an ideal platform to build portable or
mobile radios.
FDD systems are typically deployed in licensed bands
e.g., 5.8 GHz, 3.5 GHz, 2.5 GHz: the spectrum is
expensive. The cost of the spectrum forces the carriers
to serve as many users as possible. Capacity must be
optimized, which results in carriers favoring FDD
architecture. Clearly it is very desirable to have the Base
Station work in FDD, but to reduce costs, the Subscriber
Station could be a HFDD structure.
HFDD
Figure 3 shows a HFDD radio. The HFDD architecture
combines the benefits of the TDD systems while still
trying to allow for frequency duplexing. The Base
Station can operate in FDD and retain its capacity
advantage over TDD systems. This can lower the cost of
the radio significantly at the Subscriber Station where
the unit cost must be driven down.
The HFDD structure can be used in both licensed and
unlicensed bands. The Transmit and Receive can be at
the same frequency as in TDD systems or separated by
a frequency gap as in FDD. This type of radio is very
flexible. Its cost structure approaches that of a TDD
radio.
In summarizing the duplexing schemes, Intel’s baseband
chip can support both TDD and HFDD modes. This
takes care of most of the Subscriber Stations. In a
typical deployment the ratio between the Base Station
and Subscriber Stations is 1 to 100, due to the low
volume of the Base Station. The Physical (PHY) and
Media Access Control (MAC) layer need not be
designed as a custom chip; a Field Programmable Gate
Array (FPGA) could be cost effective. It is possible to
connect two baseband chips together to support an FDD
scenario for the Base Station.
We discuss various radio architectures in the following
sections; these include IF- and I/Q-based architectures
and some variants on these. Some of the interface
between the radio and the baseband chip is deliberated.
RF Interface
The baseband chip digitizes the analog signal and
performs signal processing. This PHY layer chip
contains the blocks for filtering, Automatic Gain
Control (AGC), demodulation of data, security, and
framing of data. The algorithms that do power
measurements, such as AGC and RF selection can be
taken care of by the lower-level MAC. As can be seen,
there are common parameters such as AGC that are
shared across the PHY, MAC, and radio.
The major blocks within a radio that need control from
the baseband IC are AGC, frequency selection,
sequencing of the TX/RX chain, monitoring of TX
power, and any calibration functions e.g., I/Q
imbalance. Each of these blocks are tightly coupled with
the PHY and/or lower-level MAC.
The cost reduction appears in the form of relief in the
RF TX filter, and since there is one synthesizer the die
area of the RFIC shrinks. Power savings are also
realized as in TDD systems.
RF System and Circuit Challenges for WiMAX
192
Intel Technology Journal, Volume 8, Issue 3, 2004
Figure 4: HFDD architecture
Figure 5: Block diagram of ZIF architecture
Figure 6: I/Q Baseband architecture 1
RF System and Circuit Challenges for WiMAX
193
Intel Technology Journal, Volume 8, Issue 3, 2004
A reasonable way to communicate with the radio is
through a Serial Peripheral Interface (SPI); it minimizes
pins on the RFIC.
Usually the SPI is used to control the synthesizer. In
order to make the interface more useful so that it can
control the digital AGC of an RFIC and help perform
measurements of power and temperature, the SPI needs
to be a dedicated time-critical element. In this way, the
SPI can respond to AGC, measurements, and frequency
commands in a timely and predictable manner. A note
of caution, however: traffic on the SPI could cause
interference to the incoming signal and put spurs on the
TX signal. Therefore all SPI communication should
only occur in the TX to RX time gaps. Other interface
blocks are General Purpose Input/Output (GPIO), Pulse
Width Modulators (PWM), DACs, and ADCs.
The AGC is split into RX AGC and TX AGC. In the RX
AGC, response times may have to be rapid to cope with
the changing RF channel in a mobile environment, in
the order of usec. However, in a fixed wireless
application, the channel change is in the order of msec.
The TX AGC can be relatively slow in steady state.
However, in powering up the TX, the AGC may need to
attain the correct power level in the usec time frame.
Typically, the AGC is controlled through single-bit
digital to analog converter, i.e., sigma delta converters.
Either of these methods have clock noise that needs to
be filtered out. The tradeoff here is that for a large slope
of the RF AGC, the clock noise must be filtered to avoid
distorting the signal. However, the filtering introduces a
delay that slows down the AGC response. To increase
the time response of the AGC, multibit DACs can be
used.
The selection of the RF is done through the SPI. For
HFDD systems there is a settling time from TX to RX
frequency, and the loading of the SPI is part of the
timing budget.
Monitoring the temperature of the radio is a slow
process; however, power measurements either from TX
or RX require synchronization with the TX/RX timing
gaps. Interfacing to the radio must take into account the
sequencing of the radio; for example, in the case of the
Transmitter we need to switch the antenna, enable the
TX and load frequency, change the TX gain, turn on the
PA, and finally ramp the modulation. Switching to the
RX requires sequencing the TX down to avoid spurious
emissions.
Two fundamental parameters drive radio design: noise
and linearity. The goal is to attain as much dynamic
range in the presence of undesired signals. This requires
a distribution of gain and filtering through the TX or RX
chain. Many architecture designers struggle with the
RF System and Circuit Challenges for WiMAX
placement of this gain and filtering. We look at some of
these radio architectures in the next sections.
HFDD Architecture
The details of the HFDD architecture are shown in
Figure 4. There is a frequency separation between the
TX and RX so separate filtering is necessary in the RF
front-end. However, the IF is shared between the TX
and RX. A Surface Acoustic Wave (SAW) filter
provides for excellent adjacent/alternate channel
rejection. There is a final frequency conversion to a
lower IF that can be handled by an AD. Much of the
AGC range is at the lower IF. An AGC range of 70 dB
is required; the absolute gain is higher to overcome
losses. For the TX AGC, a 50 dB range is required. The
AGC can be controlled through PWMs for analog AGC
or GPIO for step attenuators.
Two synthesizers are necessary for the double
conversion. The low-frequency synthesizer is fixed and
does not have to be switched during the RX to TX
change. The high frequency synthesizer is the
challenging block; it is required to settle within 100
usec. The step size could be as low as 125 KHz in the
3.5 GHz band.
Several signals are also sent to the Baseband IC: TX
power level (sometimes RX power level), temperature,
and synthesizer lock detect. The power level is most
important since power output has to be as close as
possible to the intended value and still within
regulations.
TDD Architecture
TDD is a good example of direct conversion
transceivers or ZIF. Figure 5 is a block diagram of the
ZIF architecture. The TX and RX frequencies are the
same so the RF filter can be shared. The
downconversion process is done with I/Q mixers; these
consume a small area on the die. The issue with such
mixers is they need to be matched; otherwise, distortion
is introduced. Also LO feedthrough effects tend to
increase due to dc imbalances. These effects are
significant since most of the gain is at the final
conversion. The dc offset results in a reduction in the
dynamic range of the AD since extra bits are required
for this offset. A dc calibration circuit can be
implemented to reduce the effect. In addition, I/Q
imbalance will result in distortion. The problems are
aggravated by temperature, gain changes, and
frequency. By going to dc, low-pass filters can be used
that are selective to channels. These can be
implemented on chip and can save on cost. It must be
noted that the on-chip low-pass filters do consume a
large die area. They can also introduce noise. WiMAX
194
Intel Technology Journal, Volume 8, Issue 3, 2004
has variable bandwidths ranging from 1 MHz to 14
MHz but as the cut-off frequency is reduced there are
significant challenges in the on-chip filter. For such ZIF
schemes there must be an Automatic Frequency Control
(AFC) loop whereby the Baseband IC controls the
reference oscillator of the RFIC. This ensures that any
dc leakage terms stay at dc and do not spill over into the
desired tones of the OFDM waveform.
I/Q Baseband Architecture 1
A variant of the HFDD and TDD architectures
mentioned above is a combination shown in Figure 6.
This structure has the advantage that some filtering is
done at an IF removing some of the strain on the dc
filters.
In addition, power can be saved by having the final
stage operate at lower frequencies. The issues related to
I/Q mismatch and dc leakage are lessened by having
less gain at dc and operating the mixers at an IF instead
of an RF. Savings can be realized at the TX filtering:
because the SAW can do most of the filtering there is no
need for the TX low-pass filters. This has the added
advantage that the I/Q mismatch from the low-pass
filters is removed. One drawback is that two Digital to
Analog (DA) converters and two Analog to Digital (DA)
converters are required.
I/Q Baseband Architecture 2
To address the problems of the I/Q baseband radios
another architecture is considered. Figure 7 shows an
RX where the signal is mixed to dc then mixed up to a
near Zero IF (NZIF). By going to dc the IF filter is
removed and filtering can be done on-chip. To avoid dc
and I/Q problems the signal is mixed to an IF. The
choice of IF is greater than half the channel bandwidth.
This structure allows the gain to be distributed between
the dc and IF stages. Also, as an added benefit, only one
AD is required. For the TX stage, I/Q upconversion is
used.
RF Challenges for MIMO, AAS, and OFDMA
Antenna diversity is an important technique that can
inexpensively enhance the performance of low-cost
subscriber stations. It can help mitigate the effects of
channel impairments like multipath, shadowing, and
interference that severely degrade a system’s
performance, and in some cases make it inoperable. By
using multiple antennas, a system’s link budget can be
significantly improved by reducing channel fading, and
in some implementations, by providing array gain.
There are several designs, all of which yield excellent
gains, that can be implemented, ranging from low to
high complexity. The basic designs are Selection
RF System and Circuit Challenges for WiMAX
Diversity Combining (SDC), Equal Gain Combining
(EGC), and Maximum Ratio Combining (MRC). SDC is
a scheme of sampling the receive performance of
multiple antenna branches and selecting the branch that
maximizes the receiver signal to noise ratio. To work
properly each antenna branch must have relatively
independent channel fading characteristics. To achieve
this, the antennas are either spatially separated, use
different polarization, or are a combination of both. The
spatial correlation of antennas can be approximated by
the zero order Bessel function given by the equation
ρ=J02(2пd/λ) and shown in Figure 8. From Figure 8, it is
seen that relatively uncorrelated antenna branches can
be achieved for spatial separations greater than onethird a wavelength, supporting the requirement for small
form-factor subscriber stations.
For optimal SDC performance the selection process and
data gathering must be completed within the coherence
time. The coherence time is the period over which a
propagating wave preserves a near-constant phase
relationship both temporally and spatially. After the
coherence time has elapsed the antennas should be resampled to account for expected channel variations and
to allow for re-selection of the optimal antenna. For a
TDD system, where reciprocal uplink (UL) and
downlink (DL) channel characteristics are expected, the
selected receive antenna can also be used as the transmit
antenna. Although the SDC technique sounds rather
simple, surprisingly large system gain improvements are
possible if the algorithms can be designed effectively.
There are two figures of merit for judging the gain
enhancement of an antenna diversity scheme. These are
diversity gain and array gain. Under changing channel
conditions, diversity gain is equivalent to the decrease in
gain variance of local signal strength fluctuations of a
multiantenna array system when compared to a singleantenna array system. The result of increased diversity
gain is the reduction in fading depth. This is due to each
antenna of a multiantenna system experiencing
independent fading channels over frequency and time.
The second figure of merit, array gain, is the
accumulation of antenna gain associated with increased
directivity via a multiantenna array system. In a typical
system, as the number of antenna array elements grows,
the gain increases 10*log (n), where n is the number of
antenna array elements. This means a doubling of gain
for every doubling of antenna elements.
195
Intel Technology Journal, Volume 8, Issue 3, 2004
Figure 7: I/Q Baseband architecture 2
Table 1: Performance enhancement of antenna
diversity
Correlation Coefficient
Spatial Correlation vs Antenna Spacing
Antenna
Scheme
1
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
Diversity
(4 Antenna
Branches)
0
0.2
0.4
0.6
0.8
Antenna Spacing (normalized to wavelength)
Figure 8: Bessel function approximation of the
spatial correlation coefficient
The SDC scheme exhibits no array gain, as only one
from n antennas is used at any instance. However,
through spatial or polarization diversity, the SDC
achieves stellar diversity gain, as shown in Table 1.
1
Antenna Gain
(SUI3,SUI4
model
w/100uSec
Rayleigh delay
spread)
Implementation
Complexity
Selection Diversity
Combining
8 dB
Low
Equal Gain
Combining
9 dB
Mid
Maximum Ratio
Combining (Analog)
10 dB
High
Maximum Ratio
Combining (Digital)
14 dB
High
(Analog)
Another basic antenna diversity technique using
multiple antennas is EGC. Instead of selecting one from
n antennas, as in SDC, the algorithm combines the
power of all antennas. The multiple independent signal
branches are co-phased, the gain of each branch set to
unity (equal gain), and then all branches combined. The
EGC antenna diversity technique achieves diversity
gain, while also producing array gain. Thus, EGC
provides higher antenna diversity gain then SDC, as can
be seen in Table 1. To achieve an antenna diversity
benefit closer to optimal, MRC of the antenna elements
RF System and Circuit Challenges for WiMAX
196
Intel Technology Journal, Volume 8, Issue 3, 2004
can be used. This technique is similar to EGC, with the
exception that the algorithm tries to optimally adjust
both the phase and gain of each element prior to
combining the power of all antennas. The summation of
the signals may be done in either the analog or digital
domain. When summation occurs in the digital domain,
RF hardware for each independent antenna branch is
required from RF to baseband. When MRC is realized
in the analog domain, summation may occur directly at
RF. Performance is better when processing is done in
the digital domain, as frequency selective channel
characteristics are compensated for in each branch. In
an analog MRC, only the average channel distortion
over frequency is used to compensate for the amplitude
and phase variation between array elements. In digital
MRC, discrete frequency components across the signal
bandwidth are co-phased and individually weighted
based on SNR at the receiver. MRC realizes the highest
antenna diversity gain compared to the other techniques
discussed, (refer to Table 1). Although the complexity is
high, MRC implementation costs are decreasing through
better RF integration and reduced CMOS geometries of
the baseband processor integrated circuit.
MIMO and AAS systems are used to improve link
margins. Using MIMO requires multiple RF chains with
multiple ADs. With integration, the cost of these
multiple chains should come down. Isolation between
the receive chains needs to be in the order of 20 dB,
which is easy to accomplish. There are no matching
requirements for the gain and phase between the RX
chains, which means that the radio design is simplified.
MIMO works well in TDD or FDD, and its
improvements to link margins are observed in multipath
environments.
In contrast, for AAS or beam forming systems, the TX
and RX chain need to be matched across frequency and
over gain and phase. However the subscriber station
does not have multiple chains. Such systems work well
in TDD mode since the TX frequency is the same as RX
frequency. AAS estimate the TX channel based on
information they get from the RX channel, so having the
same frequency improves these estimates.
OFDMA allows the RF channel to be split into
subchannels. As a result, the power can be boosted since
fewer tones are used. For users that do not TX much
data on the UL, a smaller bandwidth can be allocated.
Thus, more efficient use of the bandwidth can be made
on a per-user basis. This technique does pose some
challenges for the radio. Interference and noise between
subchannels must be carefully considered over the
whole transmit gain range. This problem is similar to the
FDD case except there is no frequency separation.
Therefore, noise performance and linearity must be
RF System and Circuit Challenges for WiMAX
excellent since there is no help from filtering. Another
issue with OFDMA is that the RF must be maintained to
<1% accuracy; otherwise, different users will collide
with each other within the subchannels.
We have discussed various duplex schemes:
architectures were outlined and some methods
improve link margin considered. Next, we discuss
particular circuit blocks within the RF system that
particular cost drivers.
RF
to
the
are
RF SYSTEM BLOCKS
There are three main areas of cost for a radio:
synthesizer, power amplifier, and filter.
Synthesizer
The synthesizer generates the LO that mixes with the
incoming RF to create a lower frequency signal that can
be digitized and processed by the Baseband IC. The
WiMAX specifications call for a high-performance
synthesizer. The synthesizer block takes up a large part
of the RFIC die area and is therefore a costly
component of the RFIC. The Integrated Phase Noise is
<1deg rms with an integration frequency of 1/20 of the
tone spacing to ½ the channel bandwidth. Thus, for the
smaller bandwidths of 1.75 MHz, the integration of the
phase noise can start as low as 100 Hz. For HFDD
architectures, the TX to RX frequency has to settle
within 100 usecs. The step size of the channel is 125
KHz in the 3.5 GHz band. In order to settle and
maintain this step size, fractional synthesizers must be
considered. It must be noted that as RF increases,
obtaining phase noise <1deg rms becomes a challenge.
As well as all the radio LOs, the clock for the AD must
be also viewed as an LO that adds phase noise to the
overall jitter specification.
Power Amplifier
Wideband digital modulation requires a high degree of
linearity. Linearity implies higher power consumption.
The tradeoff between efficiency and linearity is a
constant battle. For WiMAX, a power amplifier can
work at 4 to 5% efficiency for about a 6 dB backoff
from output P1 dB. Such a backoff results in about a
2.5% Error Vector Magnitude (EVM) or 32 dBc of
Signal to Noise plus Distortion (SNDR). With a class
AB Power Amplifier (PA) the efficiencies can run as
high as 15 to 18% with similar EVM numbers.
A much overlooked parameter in PA design is settling
time. When a PA is switched on from cold the power
level will overshoot (or undershoot), then settle out.
This settling time can be as poor as 100s of msec to get
within 0.1 dB of the final value. For OFDM symbols,
197
Intel Technology Journal, Volume 8, Issue 3, 2004
the RX has to estimate the power of a tone from the
beginning of a frame to the end of a frame. If there is a
droop of power from the beginning to the end of >0.1
dB across the frame, the BER for 64 Quadrature
Amplitude Modulation (QAM) will increase. The
primary cause for this power droop is that the bias
circuits and the output power Field Effect Transistor
(FET) are at thermally different points. Since this
phenomenon is thermal the effect can last 100s of msec.
To mitigate power droop the bias circuits have to be
placed as close to the output FETs as possible so they
see the same temperature. In some cases the PA may
have to be turned on ahead of the TX cycle to allow the
PA to stabilize and remove some of the droop. This
implies having a trigger signal based on when data are
to be transmitted. Having the MAC and PHY realize
this trigger is not a simple matter. The budget of 100
usec for HFDD is taken up by the synthesizer settling
and any PA turn-on issues. A possible solution is to
design the PA so that the PA settling is <5 usec.
used to reduce the image and far blockers; i.e., out of
the RF band. The RF front-end must be linear enough to
support the largest in-band blocker. In addition,
reciprocal mixing of the LO with the undesired signal
must be considered. The RF filters are typically >50
MHz wide and are constructed from various
technologies each with different Qs. The larger the Q,
the larger the size and the better filter shape. In FDD
systems cavity filters may have to be used; these are
large mechanical cavities and can cost >$30 in high
volume.
WIMAX SPECIFICATIONS
We highlight some of the WiMAX RF specifications
and contrast them with 802.11 specifications where
possible. The specifications are broken into RX and TX.
It should be noted that most designs aim to do better
than the standards, hence these numbers should be
viewed as the minimum requirements. In addition we
note the impact on the RFIC due to these specifications.
Filtering
Filtering is required to eliminate undesired signals from
adjacent or alternate channels. Any noise from these
immediate signals can leak noise into the desired band.
Filtering at the receiver does not help; only a clean
transmitted signal will prevent such degradation.
Regulatory bodies control the transmitted mask.
For the adjacent channel problem the challenge is
between linearity and filtering complexity. If the
undesired channels are filtered out then less backoff in
the radio is required and more of the AD bits are
available for fading margin. SAW filters have
depreciated in cost and are now in the <$2 range for
high volume. SAWs provide the optimum filtering. A
significant drawback is that the technology does fix the
maximum channel bandwidth that can be supported.
Another issue is that it is difficult to support a large
array of RF bands with a fixed IF. For spurious analysis,
the optimum IF depends on the RF.
Filtering on-chip requires a large die area and as the
channel bandwidth is reduced the die size increases. Onchip filters also produce more noise. A benefit is that
the filter can be adjusted to accommodate the various
bandwidths.
For I/Q-based designs, on-chip filters are necessary. The
filters can be matched much more closely if on-chip.
This minimizes the I/Q mismatch due to filtering. The
final channel selectivity is performed in the Baseband
IC using digital filters.
Filtering, like gain, must be distributed between the RF
and subsequent down conversions. The RF filtering is
RF System and Circuit Challenges for WiMAX
198
Intel Technology Journal, Volume 8, Issue 3, 2004
Table 3: TX specifications
Table 2: RX specifications
Parameter
802.11
*
WiMAX
Impact on RFIC
Parameter
802.11*
WiMAX
Impact on
RFIC
NF (dB)
10
7
The implication for
the RFIC is that it
may require an
external LNA to
meet a 5 dB NF.
Licensed
Band
Operation
No
Yes
The implication
for RFIC is that
the regulations
are tighter and
increase cost.
SNDR-64QAM
(dBc)
<29
29
The implication for
the RFIC is
excellent phase
noise for tone
spacing of 5 KHz
and linearity. For
802.11 the tone
spacing is larger;
i.e., 300 KHz thus
phase-noise
requirement is less
stringent.
AGC
Range
NA
50
The implication
for RFIC is that
linearity must
be maintained
over AGC range
for 64-QAM.
SNDR
(dBc)
<31
31
The implication
for RFIC is NF
of TX chain,
linearity and
phase noise.
OFDMA
No
Yes
Noise and
linearity must
be maintained
over the AGC
range for inchannel cases.
Smart
Antenna
No
YesOption
More RF chains
for MIMO or
matched RF
chains for beam
forming.
Power
Output
(dBm)
Restricted
in
unlicensed
bands
<24
dBm
The implication
for RFIC is PAs
require higher
efficiency, or
even smart PA
technology.
Alternate
Channel
Rejection
(dBc)
NA
30
The AD bits may be
used for allowing
the adjacent channel
through and some of
the alternate
channel. The digital
filter would perform
the bulk of the
close-in channel
filtering. Results in
increase in linearity
for RFIC.
HFDD
mode
No
Yes
More complicated
synthesizer to
support dual
frequency.
Channel
BW (MHz)
10; 20
1.25
;1.75;3.5
;
The implication for
the RFIC is that the
smaller bandwidths
result in a
complicated
synthesizer due to
the smaller step size.
Filtering for an array
of bandwidths
introduces adjacent
channel
compromises.
7;14;
5;10;20;
RF System and Circuit Challenges for WiMAX
(dB)
SUMMARY
WiMAX poses significant challenges to the RF
subsystem. Several RF architectures were discussed
both in FDD, HFDD, and TDD modes. The costperformance tradeoffs in the various architectures were
deliberated: these included IF- and Baseband-type
radios. Some of the more important RF system blocks,
synthesizers, power amplifiers, and filtering that relate
cost and specifications were discussed. Finally, some of
the WiMAX radio specifications were highlighted and
contrasted with 802.11, and the impact to RFIC
development was noted.
199
Intel Technology Journal, Volume 8, Issue 3, 2004
REFERENCES
[1] 802.16 REV d/D5- 2004.
[2] P.Vizmuller, RF Circuits and Systems, Artech
House, MA, USA, 1995.
AUTHORS’ BIOGRAPHIES
Balvinder Bisla received his B.Sc. degree at Sussex
University, England in 1984. He then worked at
Rutherford Appleton Labs in the UK before moving to
the USA to work on wireless metering and global
positioning systems. He was a principal RF engineer
with Iospan Wireless where they developed the world’s
first MIMO-OFDM system. Currently, he is working at
Intel on RF and microwave communication issues for
WiMAX products. His e-mail is Balvinder.s.Bisla at
intel.com.
how substrate noise spectrum structure can be exploited
for full integration of digital processors and RF delicate
circuits in the same die. Also in the labs, Luiz led the
research to move all RF passives from the die to the
substrate package in order to realize higher performance
RF System-on-Package and free silicon area for hosting
more digital functions and general-purpose processors.
Since February 2004, Luiz leads the WiMAX RF &
Analog IC internal development within the ICG/BWD
group in Santa Clara. His homepage is http://wwwsnow.stanford.edu/~franca*.
Copyright © Intel Corporation 2004. This publication
was downloaded from http://developer.intel.com/.
Legal notices at
http://www.intel.com/sites/corporate/tradmarx.htm.
Roger Eline received a B.S.E.E. degree from UC Davis
and an M.S.E.E from Santa Clara University in 1991.
Since then his work has focused on RF and microwave
communication system development. He currently
works for the Broadband Wireless Division of Intel,
where he manages the Platform Engineering Group. He
has been with Intel for one and a half years developing
low-cost IEEE 802.16 baseband and radio reference
platforms based on Intel’s IEEE 802.16 baseband
processor/modem ASIC. His e-mail is Roger.j.eline at
intel.com.
Luiz M. Franca-Neto earned his Electronic
Engineering degree, with distinction, from ITA/CTA,
SJC, Sao Paulo, Brazil, in 1989, and he received the
TASA award for being first in class in communications.
He received his M.Sc. and Ph.D. degrees from Stanford
University, all in Electrical Engineering, in 1995 and
1999, respectively. From 1990 to 1992, he was a design
engineer with ALCATEL/Elebra Telecom for public
telecommunications and optical line terminal
equipment. In USA from 1993-1996, he has worked for
HP-Labs, Palo Alto, CA, and Texas Instruments, Dallas,
TX. He was with Intel R&D Labs from 1999-2004,
where
he
led
research
on
CMOS
for
RF/Microwave/Millimeter wave frequencies. He created
new circuit design methods such as “backing off” for
LNAs and “optimum pump” for VCOs with
demonstrated circuits operating from 2.4 GHz to 100
GHz (a world record for CMOS). He led the
investigations for substrate noise in Pentium® 4
processors and deep nwell isolation where he articulated
®
Pentium is a registered trademark of Intel Corporation
or its subsidiaries in the United States and other
countries.
RF System and Circuit Challenges for WiMAX
200
Intel Technology Journal, Volume 8, Issue 3, 2004
Scalable OFDMA Physical Layer in IEEE 802.16
WirelessMAN
Hassan Yaghoobi, Intel Communications Group, Intel Corporation
Index words: OFDMA, Scalable, IEEE 802.16, WirelessMAN, subchannel, subcarrier
ABSTRACT
The concept of scalability was introduced to the IEEE
802.16 WirelessMAN Orthogonal Frequency Division
Multiplexing Access (OFDMA) mode by the 802.16 Task
Group e (TGe). A scalable physical layer enables
standard-based solutions to deliver optimum performance
in channel bandwidths ranging from 1.25 MHz to 20
MHz with fixed subcarrier spacing for both fixed and
portable/mobile usage models, while keeping the product
cost low. The architecture is based on a scalable
subchannelization structure with variable Fast Fourier
Transform (FFT) sizes according to the channel
bandwidth. In addition to variable FFT sizes, the
specification supports other features such as Advanced
Modulation and Coding (AMC) subchannels, Hybrid
Automatic Repeat Request (H-ARQ), high-efficiency
uplink subchannel structures, Multiple-Input-MultipleOutput (MIMO) diversity, and coverage enhancing safety
channels, as well as other OFDMA default features such
as different subcarrier allocations and diversity schemes.
The purpose of this paper is to provide a brief tutorial on
the IEEE 802.16 WirelessMAN OFDMA with an
emphasis on scalable OFDMA.
GHz NLOS applications. OFDM technology has been
recommended in other wireless standards such as Digital
Video Broadcasting (DVB) [2] and Wireless Local Area
Networking (WLAN) [3]-[4], and it has been successfully
implemented in the compliant solutions.
Amendments for PHY and Medium Access Control
(MAC) layers for mobile operation are being developed
(working drafts [5] are being debated at the time of
publication of this paper) by TGe of the 802.16 Working
Group. The task group’s responsibility is to develop
enhancement specifications to the standard to support
Subscriber Stations (SS) moving at vehicular speeds and
thereby specify a system for combined fixed and mobile
broadband wireless access. Functions to support optional
PHY
layer
structures,
mobile-specific
MAC
enhancements, higher-layer handoff between Base
Stations (BS) or sectors, and security features are among
those specified. Operation in mobile mode is limited to
licensed bands suitable for mobility between 2 and 6
GHz.
The IEEE 802.16 WirelessMAN standard [1] provides
specifications for an air interface for fixed, portable, and
mobile broadband wireless access systems. The standard
includes requirements for high data rate Line of Sight
(LOS) operation in the 10-66 GHz range for fixed
wireless networks as well as requirements for Non Line
of Sight (NLOS) fixed, portable, and mobile systems
operating in sub 11 GHz licensed and licensed-exempt
bands.
Unlike many other OFDM-based systems such as WLAN,
the 802.16 standard supports variable bandwidth sizes
between 1.25 and 20 MHz for NLOS operations. This
feature, along with the requirement for support of
combined fixed and mobile usage models, makes the need
for a scalable design of OFDM signaling inevitable. More
specifically, neither one of the two OFDM-based modes
of the 802.16 standard, WirelessMAN OFDM and
OFDMA (without scalability option), can deliver the kind
of performance required for operation in vehicular
mobility multipath fading environments for all
bandwidths in the specified range, without scalability
enhancements that guarantee fixed subcarrier spacing for
OFDM signals.
Because of its superior performance in multipath fading
wireless channels, Orthogonal Frequency Division
Multiplexing (OFDM) signaling is recommended in
OFDM and WirelessMAN OFDMA Physical (PHY) layer
modes of the 802.16 standard for operation in sub 11
The concept of scalable OFDMA is introduced to the
IEEE 802.16 WirelessMAN OFDMA mode by the 802.16
TGe and has been the subject of many contributions to
the standards committee [6]-[9]. Other features such as
AMC subchannels, Hybrid Automatic Repeat Request
INTRODUCTION
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
201
Intel Technology Journal, Volume 8, Issue 3, 2004
(H-ARQ), high-efficiency Uplink (UL) subchannel
structures, Multiple-Input-Multiple-Output (MIMO)
diversity, enhanced Advanced Antenna Systems (AAS),
and coverage enhancing safety channels were introduced
[10]-[14] simultaneously to enhance coverage and
capacity of mobile systems while providing the tools to
trade off mobility with capacity.
The rest of the paper is organized as follows. In the next
section we cover multicarrier system requirements,
drivers of scalability, and design tradeoffs. We follow
that with a discussion in the following six sections of the
OFDMA frame structure, subcarrier allocation modes,
Downlink (DL) and UL MAP messaging, diversity
options, ranging in OFDMA, and channel coding options.
Note that although the IEEE P802.16-REVd was ratified
shortly before the submission of this paper, the IEEE
P802.16e was still in draft stage at the time of
submission, and the contents of this paper therefore are
based on proposed contributions to the working group.
MULTICARRIER DESIGN
REQUIREMENTS AND TRADEOFFS
A typical early step in the design of an Orthogonal
Frequency Division Multiplexing (OFDM)-based system
is a study of subcarrier design and the size of the Fast
Fourier Transform (FFT) where optimal operational point
balancing protection against multipath, Doppler shift, and
design cost/complexity is determined. For this, we use
Wide-Sense
Stationary
Uncorrelated
Scattering
(WSSUS), a widely used method to model time varying
fading wireless channels both in time and frequency
domains using stochastic processes. Two main elements
of the WSSUS model are briefly discussed here: Doppler
spread and coherence time of channel; and multipath
delay spread and coherence bandwidth.
A maximum speed of 125 km/hr is used here in the
analysis for support of mobility. With the exception of
high-speed trains, this provides a good coverage of
vehicular speed in the US, Europe, and Asia. The
maximum Doppler shift [15] corresponding to the
operation at 3.5 GHz (selected as a middle point in the 26 GHz frequency range) is given by Equation (1).
f =
m
ν 35m / s
=
= 408Hz
λ 0.086m
Equation (1)
The coherence time of the channel, a measure of time
variation in the channel, corresponding to the Doppler
shift specified above, is calculated in Equation (2) [15].
T =
C
9
16 ⋅ π ⋅ f
2
= 1.03ms
Equation (2)
m
This means an update rate of ~1 KHz is required for
channel estimation and equalization.
The maximum delay spread for fixed broadband wireless
is specified by the Stanford University Interim (SUI)
channel model [17]. The worst-case rms delay spread
corresponding to SUI-6 (Terrain Type A: hilly terrain
with moderate-to-heavy tree densities) channel is 5.24 µs.
The International Telecommunications Union (ITU-R)
Vehicular Channel Model B [18] shows delay spread
values of up to 20 µs for mobile environments. The
subcarrier spacing design requires a flat fading
characteristic for worst-case delay spread values of 20 µs
with a guard time overhead of no more than 10% for a
target delay spread of 10 µs. The coherence bandwidth of
the channel (50% frequency correlation) corresponding to
the 20 µs delay spread, given by Equation (3) [15], is
shown to be approximately 10 KHz.
B ≈
C
1
1
=
= 10 KHz
5 ⋅σ
5 ⋅ 20 µs
Equation (3)
τ
This means that for delay spread values of up to 20 µs,
multipath fading can be considered as flat fading over a
10 KHz subcarrier width.
An OFDM system is also sensitive to phase noise and the
negative impact of impairment increases for narrower
subcarrier spacing, which makes the design more
expensive and complex.
The above rationale, based on the coherence time,
Doppler shift, and coherence bandwidth of the channel, is
the basis for the consideration of a scalable structure
where the FFT sizes scale with bandwidth to keep the
subcarrier spacing fixed.
Simulation results generated in [6] for a 2.5 MHz channel
bandwidth when the FFT size is kept at 2048 shows a
considerable amount of degradation in performance plot
(Bit Error Rate vs. Signal to Noise Ratio) which is clearly
recognizable for 64-QAM and high mobility.
The worst-case Doppler shift value for 125 km/hr (35
m/s) would be ~700 Hz for operation at the 6 GHz upper
limit specified by the standard. Using a 10 KHz
subcarrier spacing, the Inter Channel Interference (ICI)
power corresponding to the Doppler shift calculated in
Equation (1) can be shown [16] to be limited to ~-27 dB.
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
202
Intel Technology Journal, Volume 8, Issue 3, 2004
Table 1: OFDMA scalability parameters
Parameters
Table 2: Scalable OFDMA frame sizes
Values
1.25
2.5
5
10
20
1.429
2.85
7
5.714
11.429
22.857
Sample time (1/Fs,nsec)
700
350
175
88
44
FFT size (NFFT)
128
256
512
1024
2048
System bandwidth (MHz)
Sampling frequency
(Fs,MHz)
Subcarrier frequency
spacing
11.16071429 kHz
Frame Sizes
(msec)
Frame Sizes
(OFDM symbols)
2
2.5
4
5
8
10
12.5
20
19
24
39
49
79
99
124
198
In the remainder of this paper, the following items are
emphasized as the drivers of scalability and are revisited
frequently.
a.
Subcarrier spacing is independent of bandwidth.
89.6 µs
b.
The number of used subcarriers (and FFT size)
should scale with bandwidth.
Guard time (Tg=Tb/8)
11.2 µs
c.
OFDMA symbol time
(Ts=Tb+Tg)
100.8 µs
The smallest unit of bandwidth allocation, specified
based on the concept of subchannels (to be defined
later), is fixed and independent of bandwidth and
other modes of operation.
d.
The number of subchannels scales with FFT size
rather than with the capacity of subchannels.
e.
Tools are provided to trade mobility for capacity.
Useful symbol time
(Tb=1/ f)
Without scalability, performance is reduced or cost is
increased for low- and mid-size channel bandwidths.
Table 1 summarizes the main scalability parameters as
recommended for adoption in the standard.
Note that in Table 1, the over-sampling factor used is 8/7
(Fs = floor(8/7 BW/0.008)x0.008) as globally specified in
the standard for all OFDMA operations. The guard time
can attain any of the four possible values 1/4, 1/8, 1/16
and 1/32. By setting the value to 1/8 of an OFDM
symbol, a maximum of 11.2 µs delay spread can be
tolerated with an overhead of around 10%.
WirelessMAN OFDMA supports a wide range of frame
sizes (see Table 2) to flexibly address the need for
various applications and usage model requirements. With
a 2048 FFT size, the number of OFDM symbols in the
short frame size, (e.g., 2 ms), will be very small for
narrow bandwidths (less than 2 OFDM symbols for 1.25
MHz band) which makes the short frame sizes practically
unusable (due to high overhead). Another advantage of
scalability is to guarantee a lower bound on the number of
OFDM symbols per frame (particularly a problem for
small bandwidth and frame sizes).
Note that fixing the capacity of the subchannel may not
be the best choice especially for low-bandwidth systems
where typical applications are different in nature.
BASICS OF OFDMA FRAME
STRUCTURE
There are three types of OFDMA subcarriers:
1.
Data subcarriers for data transmission.
2.
Pilot subcarriers for various
synchronization purposes.
3.
Null subcarriers for no transmission at all, used for
guard bands and DC carriers.
estimation
and
Active subcarriers are divided into subsets of subcarriers
called subchannels. The subcarriers forming one
subchannel may be, but need not be, adjacent. Bandwidth
and MAP allocations are done in subchannels.
The pilot allocation is performed differently in different
subcarrier allocation modes. For DL Fully Used
Subchannelization (FUSC), the pilot tones are allocated
first and then the remaining subcarriers are divided into
data
subchannels.
For
DL
Partially
Used
Subchannelization (PUSC) and all UL modes, the set of
used subcarriers, that is, data and pilots, is first
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
203
Intel Technology Journal, Volume 8, Issue 3, 2004
partitioned into subchannels, and then the pilot
subcarriers are allocated from within each subchannel. In
FUSC, there is one set of common pilot subcarriers, but
in PUSC, each subchannel contains its own set of pilot
subcarriers.
another BS rather than a serving BS. Simultaneous ULs
can be data allocations and ranging or bandwidth
requests.
OFDMA Symbol Number
k+7
k+9
k+11
k+13
k+15
UL-MAP
FCH
UL-MAP
Preamble
DL-MAP
Sub-channel Logical Number
S+2
DL Burst SS1
k+18
DL Burst SS2
DL Burst
SS1
(from BS2)
DL Burst
Multicast
Variable Set #1
Variable Set #2
Fixed Set #1
Fixed Set #2
k+23 k+24
UL Burst SS1
DL Burst
SS3
UL Burst SS2
DL Burst
Broadcast
k+21
UL Burst SS3
DL Burst
SS4
Pilot Sets
k+5
Preamble
DL-MAP
k+3
UL-MAP
k k+1
S
S+1
UL Burst SS4
RNG/BW-REQ
S+L
0
DL Sub-frame
200
400
600
800
1000
1200
Sub-carrier Physical Index
1400
1600
1800
UL Sub-frame
TTG
RTG
Figure 2: Pilot distribution for FUSC
Figure 1: OFDMA frame structure (TDD, PUSC)
In a DL, subchannels may be intended for different
(groups of) receivers while in UL, Subscriber Stations
(SS) may be assigned one or more subchannels and
several transmitters may transmit simultaneously.
The subcarriers forming one subchannel may, but need
not be, adjacent. Figure 1 shows the OFDM frame
structure for Time Division Duplexing (TDD) mode.
Each frame is divided into DL and UL subframes
separated by Transmit/Receive and Receive/Transmit
Transition (TTG and RTG, respectively) gaps. Each DL
subframe starts with a preamble followed by the Frame
Control Header (FCH), the DL-MAP, and a UL-MAP,
respectively.
The FCH contains the DL Frame Prefix (DLFP) to
specify the burst profile and the length of the DL-MAP
immediately following the FCH. The DLFP is a data
structure transmitted at the beginning of each frame and
contains information regarding the current frame; it is
mapped to the FCH.
According to the OFDMA specifications, a DL-MAP
message, if transmitted in the current frame, shall be the
first MAC PDU in the burst following the FCH. An ULMAP message shall immediately follow either the DLMAP message (if one is transmitted) or the DLFP. If
Uplink Channel Descriptor (UCD) and Downlink
Channel Descriptor (DCD) messages are transmitted in
the frame, they shall immediately follow the DL-MAP
and UL-MAP messages.
Simultaneous DL allocations can be broadcast, multicast,
and unicast and they can also include an allocation for
SUBCARRIER ALLOCATION MODES
There are two main types of subcarrier permutations:
distributed and adjacent. In general, distributed subcarrier
permutations perform very well in mobile applications
while adjacent subcarrier permutations can be properly
used for fixed, portable, or low mobility environments.
These options enable the system designers to trade
mobility for throughput.
In the following section, various subcarrier allocation
modes are identified and their main characteristics are
summarized.
DL Distributed Subcarrier Permutations: Fully
Used Subchannelization (FUSC)
This method uses all the subchannels and employs fullchannel diversity by distributing the allocated subcarriers
to subchannels using a permutation mechanism. This
mechanism is designed to minimize the probability of hits
(probably of using the same physical subcarriers in
adjacent cells and sectors) between adjacent sectors/cells
by reusing subcarriers while frequency diversity
minimizes the performance degradation due to fast fading
characteristics of mobile environments.
Table 3 summarizes the subcarrier allocation structure
parameters. In DL FUSC, there are variable and fixed sets
of pilots. The fixed sets are used in all OFDM symbols
while the variable sets are divided into subsets that are
used in odd and even symbols alternatively. This provides
an appropriate tradeoff between allocated power and
frequency diversity on pilots for channel estimation.
Figure 2 shows the distribution of variable and fixed sets
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
204
Intel Technology Journal, Volume 8, Issue 3, 2004
of pilots in the case of 2048 FFT. Pilot sets for other FFT
sizes are subsets of those for the 2048 FFT.
Table 3: DL distributed subcarrier permutation
(FUSC)
Parameters
Values
illustrated in Figure 4, that spans over three OFDM
symbols (in time) of four subcarriers, each with total of
four pilot subcarriers.
Note that because of the DL and UL, cluster and tile
structures are composed of two and three OFDM
symbols, respectively; the DL and UL subframe size and
the granularity of the DL and UL allocations are also two
or three OFDM symbols, respectively.
System bandwidth (MHz)
1.25
2.5
5
10
20
FFT size (NFFT)
128
N/A**
512
1024
2048
Number of guard
subcarriers
22
N/A
86
173
345
Number of used subcarriers
106
N/A
426
851
1703
Parameters
Number of data subcarriers
96
N/A
384
768
1536
System bandwidth (MHz)
1.25
2.5
5
10
20
Number of pilot subcarriers
(uses both variable and
constant sets)
9*
N/A
42
83
166
FFT size (NFFT)
128
N/A
512
1024
2048
Number of guard
subcarriers
43
N/A
91
183
367
Number of subchannels
2
Number of
clusters/subchannels
6/3
N/A
30/15
60/30
120/60
Number of used
subcarriers
85
N/A
421
841
1681
Number of data
subcarriers
72
N/A
360
720
1440
Number of pilot
subcarriers
12
N/A
60
120
240
Subcarrier Permutation
N/A
8
16
32
Uses Permutation Type 1 for Tone
Distribution (Eq. 107 [20])
* variable set only
** FFT size of 256 is not supported
Odd Symbols
Table 4: DL distributed subcarrier permutation
(PUSC)
Values
Even Symbols
Pilot sub-carriers
Subcarrier permutation
Uses Permutation Type 1 for Tone
Distribution (Eq. 107 [20])
Cluster renumbering
Activated
Pilot sub-carriers
Figure 3: DL PUSC cluster structure
DL and UL Distributed Subcarrier Permutation:
Partially Used Subchannelization (PUSC)
According to the OFDMA specification, all OFDMA DL
and UL subframes shall start in DL and UL PUSC mode,
respectively. In DL PUSC, subchannels are divided and
assigned to three segments that can be allocated to
sectors of the same cell. The method employs fullchannel diversity by distributing the allocated subcarriers
to subchannels. A permutation mechanism is designed to
minimize the probability of hits between adjacent
sectors/cells by reusing subcarriers, while frequency
diversity minimizes the performance degradation due to
fast fading characteristics of mobile environments.
Table 4 summarizes the parameters of DL PUSC
subcarrier allocation. DL PUSC uses a cluster structure,
as illustrated in Figure 3, which spans over two OFDM
symbols (in time) of fourteen subcarriers, each with a
total of four pilot subcarriers per cluster.
Table 5 summarizes the parameters of UL PUSC
subcarrier allocation. UL PUSC uses a tile structure, as
Optional DL Distributed Subcarrier
Permutation: Fully Used Subchannelization
(OFUSC)
This method employs full-channel diversity by
distributing the allocated subcarriers to subchannels using
a permutation mechanism designed to minimize the
probability of hits between adjacent sectors/cells by
reusing subcarriers, while frequency diversity minimizes
the performance degradation due to fast fading
characteristics of mobile environments.
Table 6 summarizes the parameters of OFUSC subcarrier
allocation. In OFUSC, pilots are mapped as specified
below, which is different from the assignment in the
FUSC mode.
Compared to FUSC mode, the number of used subcarriers
in this method is considerably larger (1681 vs. 1729). As
a result, compliance with spectral mask requirements,
without a change in the over-sampling factor, may be a
challenge for this mode.
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
205
Intel Technology Journal, Volume 8, Issue 3, 2004
Table 5: UL distributed subcarrier permutation
(PUSC)
Table 6: DL distributed subcarrier permutation
(optional FUSC)
Parameters
Parameters
Values
Values
System bandwidth
1.25
2.5
5
10
20
System bandwidth
1.25
2.5
5
10
20
FFT size (NFFT)
128
N/A
512
1024
2048
FFT size (NFFT)
128
N/A
512
1024
2048
Number of guard
subcarriers
31
N/A
103
183
367
Number of guard
subcarriers
19
N/A
79
159
319
Number of tiles
24
N/A
102
210
552
109
N/A
433
865
1729
Number of subchannels
4
N/A
17
35
92
Number of used
subcarriers
Number of subcarriers per
tile
N/A
384
768
1536
N/A
4
4
3
Number of data
subcarriers
96
4
Number of used
subcarriers
N/A
48
96
192
N/A
409
841
1681
Number of pilot
subcarriers (Npilots)
12
97
Number of data
subcarriers per subchannel
48
N/A
48
48
48
Number of subchannels
2
N/A
8
16
32
Tile permutation
Uses Permutation Type 2 for Tile
Distribution (Eq. 109 [20])
Subcarrier permutation
Uses Permutation Type 3 for Subcarrier
Distribution (Eq. 110 [20])
Subcarrier permutation
Optional UL Distributed Subcarrier
Permutation: Partially Used Subchannelization
(OPUSC)
This method employs full-channel diversity by
distributing the allocated subcarriers to subchannels using
a permutation mechanism designed to minimize the
probability of hits between adjacent sectors/cells by
reusing subcarriers, while frequency diversity minimizes
the performance degradation due to fast fading
characteristics of mobile environments.
Symbol 0
Pilot sub-carriers
Symbol 1
Pilot sub-carriers
Pilot subcarrier index
Uses Permutation Type 3 for Tone
Distribution (Eq. 108 [20])
9k+3m+1,
for k=0,1,……, Npilots and
m=[symbol index] mod 3
Optional DL and UL Adjacent Subcarrier
Permutation: Advanced Modulation and Coding
(AMC)
This method uses adjacent subcarriers to form
subchannels. When used with fast feedback channels it
can rapidly assign a modulation and coding combination
per subchannel. The AMC subchannels enable the use of
“water-pouring” types of algorithms, and it can be used
effectively with an AAS option.
Symbol 2
Figure 4: UL PUSC tile structure
Table 8 summarizes the AMC subcarrier allocation
parameters. In AMC, pilots are mapped as specified
below.
Table 7 summarizes the parameters of UL OPUSC
subcarrier allocation. UL OPUSC uses a tile structure, as
illustrated in Figure 5, that spans over three OFDM
symbols (in time) of three subcarriers each with one pilot
subcarrier per tile.
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
206
Intel Technology Journal, Volume 8, Issue 3, 2004
Table 7: Optional UL distributed subcarrier
permutation (OPUSC)
Parameters
Values
Table 8: UL/DL adjacent subcarrier permutation
(optional AMC)
Parameters
Values
System bandwidth
1.25
2.5
5
10
20
System bandwidth
1.25
2.5
5
10
20
FFT size (NFFT)
128
N/A
512
1024
2048
FFT size (NFFT)
128
N/A
512
1024
2048
Number of guard
subcarriers
19
N/A
79
159
319
Number of guard subcarriers
19
N/A
79
159
319
Number of used
subcarriers
109
N/A
433
865
1729
Number of used subcarriers (Nused)
109
N/A
433
865
1729
Number of tiles
36
N/A
144
288
576
Number of pilots (Npilots)
12
N/A
48
96
192
Number of tiles per
subchannel
6
N/A
6
6
6
Number of data subcarriers
96
N/A
384
768
1536
Number of data
subcarriers per subchannel
48
Number of bands
3
N/A
12
24
48
Number of bins per band
4
N/A
4
4
4
Number of subchannels
6
Number of subcarriers per
bin (8 data +1 pilot)
9
N/A
9
9
9
Number of subchannels
2
N/A
8
16
32
Subcarrier permutation
N/A
N/A
48
24
48
48
48
96
Uses Permutation Type 4 for Tone
Distribution (Eq. 111 [20])
Sub-carrier permutation
Pilot subcarrier index
None
9k+3m+1,
for k=0,1,……, Npilots and
m=[symbol index] mod 3
Figure 5: UL OPUSC tile structure
Figure 6: Multiple zones in Uplink and Downlink subframes
Zone Switching
OFDMA PHY also supports multiple subcarrier
allocation zones within the same frame to enable the
possibility of support for and coexistence of different
types of SS’s.
Figure 6 illustrates zone switching within the DL and UL
subframes. The switching is performed using an
information element included in DL-MAP and UL-MAP.
DL and UL subframes both start in PUSC mode where
groups of subchannels are assigned to different segments
by the use of dedicated FCH messages. The PUSC
subcarrier allocation zone can be switched to a different
type of subcarrier allocation zone through a directive
from the PUSC DL-MAP. Figure 6 shows the zone
switching from the perspective of a PUSC segment. In the
figure, the PUSC FCH/DL-MAP for a segment with
IDCell X is followed with another possibly data PUSC
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
207
Intel Technology Journal, Volume 8, Issue 3, 2004
zone for IDCell X. A PUSC zone for another sector/cell
with IDCell Y (Y in general is different from X) is
allocated next. An FUSC zone for IDCell Z is shown next
in the figure. Note that IDCell Z may be the same as
IDCell X which means that a PUSC to FUSC switching is
scheduled within the segment for Frequency Reuse One
operations. A switching to IDCell 0 can be planned for all
network broadcast operations.
users. The figure illustrates a four-antenna configuration
where the AAS preamble and AAS DL MAPs structure
are repeated multiples of four times to support the
corresponding four groups of users.
Optional PUSC, FUSC, and AMC zones in DL subframes
and optional PUSC and AMC zones in UL subframes can
be similarly scheduled. Allocation of AMC zones enables
the simultaneous support of fixed, portable, and nomadic
mobility users along with high mobility users (supported
in PUSC/FUSC zones).
DIVERSITY OPTIONS
OFDMA PHY supports AAS and also a set of second-,
third-, and fourth-order transmit diversity options.
With the AAS option, the system uses a multiple-antenna
transmission to improve the coverage and capacity of the
system while minimizing the probability of outage
through transmit diversity, beam forming, and null
steering.
Transmit diversity options consist of a comprehensive set
of methods based on second- or fourth-order diversity in
DL and second-order diversity in UL that can be flexibly
chosen to tradeoff capacity and coverage. The set
includes both closed- and open-loop options and also
supports Spatial Multiplexing (SM) for maximum
spectral efficiency.
Advanced Antenna Systems
Two optional AAS modes are supported in OFDMA
PHY: Diversity-Map Scan and Direct Signaling Method.
Diversity-Map Scan supports both diversity (FUSC and
PUSC) and adjacent (AMC) subcarrier permutation
options. The Direct Signaling Method supports adjacent
subcarrier permutation with less overhead in control
signaling.
We now discuss the Diversity-Map Scan option when
applied to the AMC subcarrier allocation method.
Figure 7 shows the AAS Diversity Map Zone within a
frame. The DL subframe includes a non-AAS section and
an AAS section specified by information elements
provided in the DL MAP.
Within the AAS zone, subchannel numbers 4 and N-4 (N
is the index for the last logical subchannel) are allocated
to the AAS DL MAP where a pointer to a beamformed
broadcast DL MAP is specified. The broadcaset DL MAP
provides beamformed private DL and UL MAPs for AAS
Figure 7: AAS diversity MAP zone
Within the AAS zone, the AAS BS specifies allocations
to be used for SS Ranging. In TDD mode, the BS can
extract the channel information required for beam
forming from the Ranging Request messages received
from the SS’s. In FDD mode, beam forming is done
through the AAS Feedback Request and Response
messages where channel response information along with
mean Received Signal Strength Indicator (RSSI) and
Carrier to Interference plus Noise Ratio (CINR) are
reported back to the BS by the SS.
Transmit Diversity
OFDMA mode supports second-, third- and fourth-order
transmit diversity options in DL and second-order
transmit diversity in UL. All diversity options are
applicable to both diversity and adjacent subcarrier
permutations.
Space Time Coding (STC) based on Alamouti algorithm
[19] and Frequency Hopping Diversity Code (FHDC) are
two options for second-order diversity in DL. Although
not specified by the standard, the number of receive
antennas can be specified depending on the performance
required.
Second-Order STC
Second-order STC in DL supports coding rates of 1 and 2
using the following two transmission format matrices.
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
208
Intel Technology Journal, Volume 8, Issue 3, 2004
S
A= 
S
−S 

S 
*
i+1
i
Equation (4)
*
i +1
i
S 
B= 
S 
Equation (5)
i
i+1
Here S ’s are OFDM symbols in the frequency domain
right before IFFT operation.
k
The optional STC transmit diversity is also supported in
UL using the transmission format matrix A of Equation
(4). Matrix B of Equation (5) can be used by two SS’s in
a collaborative special multiplexing mode.
S

C = S
 S
S

S
A=
0

 0
−S
S

S
B=
S

 S
−S
i
*
i +1
*
i +1
S
0
0
S
0
S
i
i
S
i +1
*
i +1
*
i
i+ 2
i +3
S
1
2
3
4
*
i +3
*
i+ 2
0 

0 
−S 

S 
*
i +2
−S
S
S
C=
S
S

0
Equation (6)
i+ 3
*
i+ 3
i +2
−S 

−S 
S 

S 
*
S
i+ 4
S
i +5
i+ 6
*
i+ 7
*
S
i +6
S
i +7
Equation (7)
~
S
~
S
~
S
~
S
~
S
~
S
~
S
~
S
1
=S +S
2
=S +S
k
Third-Order STC
The third-order transmit diversity in DL supports rates 1,
2, or 3 using the following transmission format matrices
A, B, and C, respectively.
 S~
~
B = S
 S~

~
−S
~
S
~
S
2
1
2
7
*
2
*
1
*
2
*
1
8
0
~
S
~
S
0
~
−S
~
S
~
S
~
S
~
S
~
−S
~
S
~
S
3
4
5
6
3
1I
3Q
2I
4Q
4
=S +S
=S +S
4I
2Q
5
=S +S
7Q
=S +S
8Q
=S +S
= S +S
7I
5Q
8I
6Q
3
6
8
3I
1Q
5I
6I
Equation (12)
θ = (tan 2) / 2 ,
,
S = X ⋅e
for
k = 1,2,L,8 and X ' s are OFDM symbols in the frequency
domain right before the IFFT operation.
where
−1
S = S + j⋅S
k
kI
jθ
kQ
k
k
k
Precoding
A general KxL precoding matrix W is specified to be
applied to the output X of any second-, third- or fourthorder diversity option mentioned earlier. This way an L th
order output vector Z of the STC block is transformed
into a final K th order vector for transmission on
antennas.
Z =W ⋅ X
Equation (8)
~
−S
~
S
0
Equation (11)
*
i+ 5
Here, S ’s are OFDM symbols in the frequency domain
right before the IFFT operation.
1
3




i+ 4






 S~
~
A = S
0

2
In Equations (9) and (10), we have
7
Fourth-Order STC
The fourth-order transmit diversity in DL supports rates
1, 2, or 4 using the following transmission format
matrices A, B, and C, respectively.
1
4
*
3
*
6
*
5
4





Equation (9)





Equation (10)
*
Equation (13)
Precoding can be performed either in closed-loop or
open-loop form. In the case of open-loop, the BS weights
the transmission according to the channel measurement
performed on the UL signal, where a reciprocity
assumption can be made for a TDD mode, for example.
In the case of closed-loop, BS uses the Channel Quality
Indications feedback from the SS.
RANGING IN OFDMA
The OFDMA PHY specifies a ranging allocation that can
be used for ranging as well as bandwidth request. Initial
and periodic ranging processes are supported to
synchronize the SS’s with the BS at the initial network
entry and also periodically during the normal operation.
Bandwidth request mechanism is supported so that SS’s
can request UL allocations for transmission of data to the
BS. A set of 256 special pseudo-noise 144 bit-long
ranging codes are divided into three groups for Initial
Ranging, Periodic Ranging, and Bandwidth Requests,
such that the BS can determine the purpose of the
received code by the subset to which the code belongs.
One or more groups of six adjacent subchannels are
allocated to ranging where the ranging codes are BPSK
modulated to the allocation. The SS randomly selects one
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
209
Intel Technology Journal, Volume 8, Issue 3, 2004
code from the allocated set of codes and transmits back to
the BS through ranging allocation. Different SS’s can
collide on their ranging and/or bandwidth requests and
the BS is still able to receive simultaneous requests.
further increases the capability of the system to support
larger numbers of synchronization mismatches.
To process an Initial Ranging request, a ranging code is
repeated twice and transmitted in two consecutive OFDM
symbols with no phase discontinuity between the two
OFDM symbols (see Figure 8). This way, the BS can
properly receive the requests from un-ranged SS’s with a
larger value of synchronization mismatch when the first
attempt is made to enter the network. The SS can
optionally use two consecutive ranging codes transmitted
during a four-OFDM symbol period (see Figure 9). This
option decreases the probability of failure and increases
the ranging capacity to support larger numbers of
simultaneous ranging SS’s while at the same time it
Figure 8: Initial ranging transmission
Figure 9: Initial ranging using two ranging codes
brief summary of the supported mandatory and optional
modes are given here.
Figure 10: Periodic ranging and bandwidth
request transmission
For Periodic Ranging or Bandwidth Requests, the
options are either to use one or three consecutive
ranging codes transmitted during a one or three OFDM
symbol period (see Figure 10 and Figure 11). In the case
of three ranging codes, the probability of failure
decreases at the same time as the ranging capacity
increases, to support larger numbers of simultaneous
ranging SS’s.
CHANNEL CODING
A detailed discussion of channel coding options in
OFDMA PHY is beyond the scope of this paper; only a
Based on terminology used in WirelessMAN OFDMA
PHY, channel coding consists of Randomization,
Forward Error Correction (FEC), bit interleaving, and
modulation. Repetition code is used on various control
messages to further enhance the error correction
performance of the system. Repetition codes of 2, 4, or
6 are implemented by utilizing multiple subchannels.
Randomization is performed on both UL and DL data.
The data are randomized using a PN sequence generator
with a polynomial of degree 15 that is reinitialized at the
beginning of each FEC block with a seed, which is a
function of the OFDM symbol offset (from the start of
the frame) and the starting subchannel number
corresponding to the FEC block.
The OFDMA PHY supports mandatory tail-biting
Convolutional Coding and three optional coding
schemes:
Zero
Tailing
Convolutional
code,
Convolutional Turbo code along with H-ARQ, and
Block Turbo code.
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
210
Intel Technology Journal, Volume 8, Issue 3, 2004
The tail biting is implemented by initializing the
encoders memory with the last data bits of the FEC
block being encoded, and the zero tailing is
implemented by appending a zero tail byte to the end of
each burst.
H-ARQ mitigates the effect of impairments due to
channel and external interference by effectively
employing time diversity along with incremental
transmission of parity codes (subpackets in this case). In
the receiver, previously erroneously decoded subpackets
and retransmitted subpackets are combined to correctly
decode the message. The transmitter decides whether to
send additional subpackets, based on ACK/NAK
messages received from the receiver.
based on the number of coded bits per encoded block
size. The interleaving is performed using a two-step
permutation process. The first permutation ensures that
adjacent coded bits are mapped onto nonadjacent
subcarriers. The second permutation ensures that
adjacent coded bits are mapped alternately onto less or
more significant bits of the constellation, thus avoiding
long runs of lowly reliable bits.
CONCLUSION
The IEEE 802.16 WirelessMAN OFDMA supports a
comprehensive set of system parameters and advanced
optional features for mobile, portable, and fixed usage
models. Scalability enables the technology to operate
optimally in different usage scenarios.
Bit interleaving is performed on encoded data at the
output of FEC. The size of the interleaving block is
Figure 11: Periodic ranging and bandwidth request transmission using three codes
ACKNOWLEDGMENTS
The author thanks Dr. C.K. Bright for the support
provided during the writing of this paper and the
valuable help on the graphics. I also thank T.J. Cox, D.
Andelman, R.C. Schwartz, Y. Lomnitz, G. Begis and S.
Talwar for their valuable reviews and comments.
REFERENCES
[1]. IEEE P802.16-2004, standard for local and
metropolitan area networks Part 16: Air Interface
for Fixed Broadband Wireless Access Systems
Name (To be published).
[2]. ETS 300 744 rev 1.2.1, (1999-01), “digital
broadcasting systems for television, sound and data
services (DVB-T); framing structure, channel
coding and modulation for digital terrestrial.”
[3]. IEEE Std 802.11a-1999, Part 11, “Wireless LAN
Medium Access Control (MAC) and Physical
Layer (PHY) specifications; high-speed physical
layer in the 5 GHz band.”
[4]. IEEE 802.11g-2003, “IEEE Standard for
Information technology, telecommunications and
information exchange between systems, local and
metropolitan area networks, specific requirements,
Part 11: Wireless LAN Medium Access Control
(MAC) and Physical Layer (PHY) specifications,
Amendment 4: further higher-speed physical layer
extension in the 2.4 GHz band.”
[5]. IEEE P802.16e, “draft amendment to IEEE
standard for local and metropolitan area networks,
Part 16: air interface for fixed and mobile
broadband wireless access systems, amendment for
physical and medium access control layers for
combined fixed and mobile operation in licensed
bands.”
[6]. IEEE C802.16d-04_47, “applying scalability for the
OFDMA PHY layer.”
[7]. IEEE C802.16REVd-04/50r1, “OFDMA PHY
enhancements for better mobility performance.”
[8]. IEEE C802.16d-04/72, “additional optional symbol
structure.”
[9]. IEEE C802_16e-04/88-r3, “128 FFT sizes for
OFDMA PHY.”
[10].
C802.16REVd-04_50r3,
“OFDMA
PHY
enhancements for better mobility performance.”
[11].
IEEE C802.16d-04/90, “AAS enhancements
for OFDMA PHY.”
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
211
Intel Technology Journal, Volume 8, Issue 3, 2004
[12].
IEEE 802.16d-04/65, “Enhancing MIMO
features for OFDMA PHY layer.”
[13].
IEEE C802.16e-04_72r2, “STC Enhancements
for optional FUSC and AMC zones for OFDMA
PHY layer.”
[14].
IEEE C802.16e-04/208r2, “space-time codes
for 3 transmit antennas for the OFDMA PHY.”
[15].
Rappaport, T.S., Wireless Communications
Principles and Practice, Second Edition 2002,
Prentice Hall PTR, Upper Saddle River, NJ.
[16].
Li, Y., Cimini, L.J., “Bounds on the
Interchannel Interference of OFDM in TimeVarying Impairments,” IEEE Transactions ON
Communications, Vol. 49, No. 3, March 2001, pp.
401-404.
[17].
IEEE 802.16.3c-01/29r4, “channel models for
fixed wireless applications.”
[18].
Recommendation ITU-R M.1225, “Guidelines
for evaluation of Radio transmission technologies
for IMT-2000, 1997.”
[19].
Alamouti, S. A., “Simple Transmit Diversity
Technique for Wireless Communications,” IEEE
Journal on Select Areas in Communications, Vol.
16, No. 8, October 1998.
[20].
IEEE P802.16REVd/D5-2004, standard for
local and metropolitan area networks Part 16: Air
Interface for Fixed Broadband Wireless Access
Systems Name.
solutions. He is a member of the IEEE 802.16 and
802.20 working groups. He also serves as secretary of
the sub 11 GHz Technical Working Group for the
WiMAX forum, an industry group focused on
interoperability of systems that conform to the IEEE
802.16 standard. Prior to Intel, he worked on design and
modeling of wireless terrestrial and satellite receivers
for Stanford Telecom and on RF network design of
mobile wireless systems for LCC international. His email is hassan.yaghoobi at intel.com.
Copyright © Intel Corporation 2004. This publication
was downloaded from http://developer.intel.com/.
Legal notices at
http://www.intel.com/sites/corporate/tradmarx.htm.
AUTHOR’S BIOGRAPHY
Hassan Yaghoobi received a B.S. degree from Sharif
University of Technology, Tehran, Iran, in 1989 and
M.S. and Ph.D. degrees from the University of
Maryland, in 1993 and 2000, respectively, all in
Electrical Engineering. His academic research interests
include nonlinear control theory, communications
theory, and digital signal processing.
Hassan’s industrial experience includes communications
systems
engineering,
silicon
design/functional
definition, and standards development in the area of
broadband communications. Since 2000, he has been
working at Intel Corporation. As an engineer for Intel’s
Broadband Product Group, he worked on silicon
functional definition, algorithm design, system design
verification, and validation of various cable modem
products. He represented Intel at the DOCSIS2.0 Radio
Frequency Interface Specification (RFI) and Acceptance
Test Plan (ATP) standard committees at Cablelabs.
Hassan is currently working as a Strategic Technologist
for Intel’s Broadband Wireless Division working on
product definitions of Intel’s 802.16d/e silicon
Scalable OFDMA Physical Layer in IEEE 802.16 WirelessMAN
212
IEEE 802.16 Medium Access Control
and Service Provisioning
Govindan Nair, Intel Communications Group, Intel Corporation
Joey Chou, Intel Communications Group, Intel Corporation
Tomasz Madejski, Intel Communications Group, Intel Corporation
Krzysztof Perycz, Intel Communications Group, Intel Corporation
David Putzolu, Intel Communications Group, Intel Corporation
Jerry Sydir, Intel Communications Group, Intel Corporation
Index words: 802.16 MAC, OFDM, OFDMA, QoS, Service Provisioning, IXA, IA, MIB, WiMAX
ABSTRACT
In this paper we describe the IEEE 802.16 Orthogonal
Frequency Division Multiplexing (OFDM) and the 802.16
Orthogonal Frequency Division Multiple Access
(OFDMA), Medium Access Control (MAC) protocols,
both of which are key elements of the Worldwide
Interoperability for Microwave Access Forum (WiMAX)
deployments. We also discuss the types of provisioning
and Quality of Service (QoS) that can be achieved using
the features of this MAC protocol to facilitate the
WiMAX deployments. Finally, we review the challenges
inherent in implementing this MAC protocol on
architectures such as the Intel® IXP network processors
and embedded Intel architecture processors to support the
application of MAC functionality to the wide range of
potential QoS and provisioning approaches.
INTRODUCTION
The success of cellular networks in the last decade and
the integration of narrowband data solutions into these
networks are the first indications that wireless solutions
may be able to solve the last mile, a.k.a. the consumer
broadband problem. The emergence of Wi-Fi networks
has demonstrated that high-bandwidth radio networks are
feasible and desirable for both fixed and mobile clients.
Finally, recent advances in Radio Frequency (RF)
technology, coding algorithms, Medium Access Control
(MAC) protocols, and packet processing horsepower
have made it possible to achieve the high bandwidths of
®
Intel is a registered trademark of Intel Corporation or its
subsidiaries in the United States and other countries.
Wi-Fi networks over the extended coverage areas of
cellular networks. This fusion, which is realized in the
IEEE 802.16 architecture, not only addresses the
traditional last mile problem, but also supports nomadic
and mobile clients on the go. The architecture enables a
“hotzone” deployment model, where high-speed Internet
access is provided over large portions of urban areas and
along major freeways. In this model, laptops and PDAs
operate as Subscriber Stations (SS’s) allowing users to
connect to the network in parks, buildings, or wherever
they may be.
The broadband wireless architecture is being standardized
by the IEEE 802.16 Working Group (WG) and the
Worldwide Interoperability for Microwave Access
(WiMAX) forum. The 802.16 WG is developing
standards for the Physical (PHY) and MAC layers, as
well as for the security and higher-layer network model.
In this paper we concentrate on the MAC layer and the
Quality of Service (QoS) support that is provided by the
IEEE 802.16 standard. Throughout the paper, we use the
terms 802.16 and WiMAX interchangeably.
In the MAC section we describe the major functions of
the 802.16 MAC operating on the Orthogonal Frequency
Division Multiplexing (OFDM) and Orthogonal
Frequency Division Multiple Access (OFDMA) PHY
layers [8]. We then explore the differences in QoS
mechanisms in the 802.11 and 802.16 with a view
towards pointing out the challenges associated with largescale WiMAX deployment. We also describe service
provisioning and the WiMAX management model that
supports self-install and auto-configuration. Finally, in
the Implementation Challenges section, we describe two
IEEE 802.16 Medium Access Control and Service Provisioning
213
Intel Technology Journal, Volume 8, Issue 3, 2004
alternate implementations of the 802.16 MAC on an Intel
IXP network processor and on an Intel IA processor.
THE MEDIUM ACCESS CONTROL
(MAC) LAYER
The IEEE 802.16 MAC [8] layer performs the standard
Medium Access Control (MAC) layer function of
providing a medium-independent interface to the 802.16
Physical (PHY) layer. Because the 802.16 PHY is a
wireless PHY layer, the main focus of the MAC layer is
to manage the resources of the airlink in an efficient
manner. The 802.16 MAC protocol is designed to support
Point to Multipoint (PMP) and Mesh network models. In
this paper we focus on the PMP network model. The
802.16 MAC protocol is connection oriented. Upon
entering the network, each Subscriber Station (SS) creates
one or more connections over which their data are
transmitted to and from the Base Station (BS). The MAC
layer schedules the usage of the airlink resources and
provides Quality of Service (QoS) differentiation. It
performs link adaptation and Automatic Repeat Request
(ARQ) functions to maintain target Bit Error Rates (BER)
while maximizing the data throughput. The MAC layer
also handles network entry for SS’s that enter and leave
the network, and it performs standard Protocol Data Unit
(PDU) creation tasks. Finally, the MAC layer provides a
convergence sub layer that supports Asynchronous
Transfer Mode (ATM) cell- and packet-based network
layers.
In the remainder of this section we provide an overview
of the functions of the MAC layer. We start with a brief
description of the Orthogonal Frequency Division
Multiplexing (OFDM) and Orthogonal Frequency
Division Multiple Access (OFDMA) PHY layers, and
show how they motivate some of the functions that must
be performed by the MAC layer for these specific PHYs.
We then describe the major functions of the 802.16 MAC
protocol.
(CC) scheme, supporting code rates of 1/2, 2/3, 3/4, and
5/6. Variable-rate Block Turbo Code (BTC) and
Convolutional Turbo Code (CTC) are also optionally
supported. The standard supports multiple modulation
levels, including Binary Phase Shift Keying (BPSK),
Quadrature Phase Shift Keying (QPSK), 16-Quadrature
Amplitude Modulation (QAM) and 64-QAM. Finally, the
PHY supports (as options) transmit diversity in the
Downlink (DL) using Space Time Coding (STC) and
Adaptive Antenna Systems (AAS) with Spatial Division
Multiple Access (SDMA).
The transmit diversity scheme uses two antennas at the
BS to transmit an STC encoded signal, in order to provide
the gains that result from second-order diversity. Each of
two antennas transmits a different symbol (two different
symbols) in the first symbol time. The two antennas then
transmit the complex conjugate of the same two symbols
in the second symbol time. The resulting data rate is the
same as without transmit diversity. AAS is used in the
802.16 specification to describe beam forming
techniques, where an array of antennas is used at the BS
to increase gain to the intended SS, while nulling out
interference to and from other SS’s and interference
sources. AAS techniques can be used to enable SDMA,
where multiple SS’s that are separated in space can
receive and transmit on the same subchannel at the same
time. By using beam forming, the BS is able to direct the
desired signal to the different SS’s and can distinguish
between the signals of different SS’s even though they are
operating on the same subchannel(s).
The OFDM Physical Layer
The WirelessMAN-OFDM PHY layer is based on OFDM
modulation. It is intended mainly for fixed access
deployments, where SS’s are residential gateways
deployed within homes and businesses in much the same
way as DSL and cable modems are deployed to provide
broadband over wireline networks. The OFDM PHY
layer supports subchannelization in the Uplink (UL).
There are 16 subchannels in the UL. The OFDM PHY
layer supports Time Division Duplexing (TDD) and
Frequency Division Duplexing (FDD) operations, with
support for both FDD and Half-Duplex FDD (H-FDD)
SS’s. The specification defines as mandatory, a combined
variable-rate Read-Solomon (RS)/Convolutional Coding
Figure 1: Frame structure
Figure 1 illustrates the frame structure for a TDD system.
The frame is divided into DL and UL subframes. The DL
subframe is made up of a preamble, Frame Control
Header (FCH), and a number of data bursts. The FCH
specifies the burst profile and the length of one or more
DL bursts that immediately follow the FCH. The DLMAP, UL-MAP, DL Channel Descriptor (DCD), UL
IEEE 802.16 Medium Access Control and Service Provisioning
214
Intel Technology Journal, Volume 8, Issue 3, 2004
Channel Descriptor (UCD), and other broadcast messages
that describe the content of the frame are sent at the
beginning of these first bursts. The remainder of the DL
subframe is made up of data bursts to individual SS’s.
Each data burst consists of an integer number of OFDM
symbols and is assigned a burst profile that specifies the
code algorithm, code rate, and modulation level that are
used for those data transmitted within the burst. The UL
subframe contains a contention interval for initial ranging
and bandwidth allocation purposes and UL PHY PDUs
from different SS’s. The DL-MAP and UL-MAP
completely describe the contents of the DL and UL
subframes. They specify the SS’s that are receiving
and/or transmitting in each burst, the subchannels on
which each SS is transmitting (in the UL), and the coding
and modulation used in each burst and in each
subchannel.
If transmit diversity is used, a portion of the DL frame
(called a zone) can be designated to be a transmit
diversity zone. All data bursts within the transmit
diversity zone are transmitted using STC coding. Finally,
if AAS is used, a portion of the DL subframe can be
designated as the AAS zone. Within this part of the
subframe, AAS is used to communicate to AAS-capable
SS’s. AAS is also supported in the UL.
In FDD systems, the DL and UL frame structure is
similar, except that the UL and DL are transmitted on
separate channels. When H-FDD SS’s are present, the BS
must ensure that it does not schedule an H-FDD SS to
transmit and receive at the same time.
The OFDMA Physical Layer
data. Also, because a number of different
subchannelization schemes are defined, the frame is
divided into a number of zones, each using a different
subchannelization
scheme.
(Most
of
the
subchannelization schemes are optional, so it is not
expected that all schemes will be used in all
deployments). The MAC layer is responsible for dividing
the frame into zones and communicating this structure to
the SS’s in the DL and UL maps. As in the OFDM PHY,
there are optional transmit diversity and AAS zones, as
well as a MIMO zone.
MAC Header Types and Management
Messages
There are two types of MAC headers: a generic header
and a Bandwidth Request (BR) MAC header. The generic
header is used to transmit data or MAC messages. The
BR header is used by the SS to request more bandwidth
on the UL. The maximum length of the MAC PDU is
2048 bytes, including header, payload, and Cyclic
Redundancy Check (CRC). For Point to Multi Point
(PMP), the MAC defines ARQ Fast-Feedback,
Fragmentation, Packing, and Grant Management
subheaders. ARQ Fast-Feedback and Grant Management
subheaders are used to communicate ARQ and bandwidth
allocation states between the BS and SS. Fragmentation
and Packing subheaders are used to utilize the bandwidth
allocation efficiently. The standard defines a number of
MAC management messages that are used to pass control
information between the SS and BS. These messages are
divided into broadcast messages, initial ranging messages,
basic messages, and primary management messages.
The WirelessMAN-OFDMA PHY layer is also based on
OFDM modulation. It supports subchannelization in both
the UL and DL. The standard supports five different subchannelization schemes. The OFDMA PHY layer
supports both TDD and FDD operations. CC is the
required coding scheme and the same code rates are
supported as are supported by the OFDM PHY layer.
BTC and CTC coding schemes are optionally supported.
The same modulation levels are also supported. STC and
AAS with SDMA are supported, as well as Multiple
Input, Multiple Output (MIMO). MIMO encompasses a
number of techniques for utilizing multiple antennas at
the BS and SS in order to increase the capacity and range
of the channel. (A full discussion of the implications of
supporting MIMO are outside the scope of this paper.)
Network Entry
The frame structure in the OFDMA PHY layer is similar
to that of the OFDM PHY layer. The notable exceptions
are that subchannelization is defined in the DL as well as
in the UL, so broadcast messages are sometimes
transmitted at the same time (on different subchannels) as
When an SS wishes to enter the network, it scans for a
channel in the defined frequency list. Normally an SS is
configured to use a specific BS with a given set of
operational parameters, when operating in a licensed
band. If the SS finds a DL channel and is able to
synchronize at the PHY level (it detects the periodic
In order to communicate on the network an SS needs to
successfully complete the network entry process with the
desired BS. The network entry process is divided into DL
channel synchronization, initial ranging, capabilities
negotiation,
authentication
message
exchange,
registration, and IP connectivity stages. The network
entry state machine moves to reset if it fails to succeed
from a state. Upon completion of the network entry
process, the SS creates one or more service flows to send
data to the BS. Figure 2 depicts the network entry
process. The following subsections describe each of these
stages in more detail.
Downlink Channel Synchronization
IEEE 802.16 Medium Access Control and Service Provisioning
215
Intel Technology Journal, Volume 8, Issue 3, 2004
frame preamble), then the MAC layer looks for DCD and
UCD to get information on modulation and other DL and
UL parameters.
manufacturer and a description of the supported
cryptographic algorithms to its BS. The BS validates the
identity of the SS, determines the cipher algorithm and
protocol that should be used, and sends an authentication
response to the SS. The response contains the key
material to be used by the SS. The SS is required to
periodically perform the authentication and key exchange
procedures to refresh its key material.
Registration
After successful completion of authentication the SS
registers with the network. The SS sends a registration
request message to the BS, and the BS sends a
registration response to the SS. The registration exchange
includes IP version support, SS managed or non-managed
support, ARQ parameters support, classification option
support, CRC support, and flow control.
IP Connectivity
The SS then starts DHCP (IETF RFC 2131) to get the IP
address and other parameters to establish IP connectivity.
The BS and SS maintain the current date and time using
the time of the day protocol (IETF RFC868). The SS then
downloads operational parameters using TFTP (IETF
RFC 1350).
Figure 2: Network entry process
Initial Ranging
When an SS has synchronized with the DL channel and
received the DL and UL MAP for a frame, it begins the
initial ranging process by sending a ranging request MAC
message on the initial ranging interval using the minimum
transmission power. If it does not receive a response, the
SS sends the ranging request again in a subsequent frame,
using higher transmission power. Eventually the SS
receives a ranging response. The response either indicates
power and timing corrections that the SS must make or
indicates success. If the response indicates corrections,
the SS makes these corrections and sends another ranging
request. If the response indicates success, the SS is ready
to send data on the UL.
Capabilities Negotiation
After successful completion of initial ranging, the SS
sends a capability request message to the BS describing
its capabilities in terms of the supported modulation
levels, coding schemes and rates, and duplexing methods.
The BS accepts or denies the SS, based on its capabilities.
Authentication
After capability negotiation, the BS authenticates the SS
and provides key material to enable the ciphering of data.
The SS sends the X.509 certificate of the SS
Transport Connection Creation
After completion of registration and the transfer of
operational parameters, transport connections are created.
For preprovisioned service flows, the connection creation
process is initiated by the BS. The BS sends a dynamic
service flow addition request message to the SS and the
SS sends a response to confirm the creation of the
connection. Connection creation for non-preprovisioned
service flows is initiated by the SS by sending a dynamic
service flow addition request message to the BS. The BS
responds with a confirmation.
Convergence Sublayer
The 802.16 MAC layer provides a convergence sublayer
for the transport of ATM cells and IP packets. The MAC
layer classifies the packets and steers them into the
required 802.16 connection and packet header
suppression in order to avoid the transmission of
redundant information over the airlink.
Protocol Data Unit Creation and Automatic
Repeat Request
The 802.16 MAC performs the standard PDU creation
functions. It applies the MAC header and optionally
calculates the CRC. Because airlink resources are very
precious, the 802.16 MAC layer performs both
fragmentation of MAC SDUs and packing of MAC
SDUs. Small SDUs are packed to fill up airlink
IEEE 802.16 Medium Access Control and Service Provisioning
216
Intel Technology Journal, Volume 8, Issue 3, 2004
allocations and large SDUs are fragmented when they
don’t fit into an airlink allocation. MAC PDUs may be
concatenated into bursts having the same modulation and
coding.
Each SS to BS connection is assigned a service class as
part of the creation of the connection. When packets are
classified in the convergence sublayer, the connection
into which they are placed is chosen based on the type of
QoS guarantees that are required by the application.
Figure 4 depicts the 802.16 QoS mechanism in supporting
multimedia services, including TDM voice, VoIP, video
streaming, TFTP, HTTP, and e-mail.
Figure 3: PDU and SDU in protocol stack
ARQ processing is the process of retransmitting MAC
SDU blocks (“ARQ blocks”) that have been lost or
garbled. The 802.16 MAC uses a simple sliding windowbased approach, where the transmitter can transmit up to
a negotiated number of blocks without receiving an
acknowledgement. The receiver sends acknowledgement
or negative acknowledgement messages to indicate to the
transmitter which SDU blocks have successfully been
received and which have been lost. The transmitter
retransmits blocks that were lost and moves the sliding
window forward when SDU blocks are acknowledged to
have been received.
Service Classes
The 802.16 MAC provides QoS differentiation for
different types of applications that might operate over
802.16 networks. The 802.16 standard defines the
following types of services:
! Unsolicited Grant Services (UGS): UGS is designed
to support Constant Bit Rate (CBR) services, such as
T1/E1 emulation, and Voice Over IP (VoIP) without
silence suppression.
! Real-Time Polling Services (rtPS): rtPS is designed
to support real-time services that generate variable
size data packets on a periodic basis, such as MPEG
video or VoIP with silence suppression.
! Non-Real-Time Polling Services (nrtPS): nrtPS is
designed to support non-real-time services that
require variable size data grant burst types on a
regular basis.
! Best Effort (BE) Services: BE services are typically
provided by the Internet today for Web surfing.
Figure 4: QoS mechanism for multimedia services
There are two types of polling mechanisms:
Unicast: When an SS is polled individually, it is allocated
bandwidth to send bandwidth request messages.
Contention-based: Contention-based bandwidth request
is used when insufficient bandwidth is available to
individually poll many inactive SS’s. The allocation is
multicast or broadcast to a group of SS’s that have to
contend for the opportunity to send bandwidth requests.
Scheduling and Link Adaptation
The goal of scheduling and link adaptation is to provide
the desired QoS treatment to the traffic traversing the
airlink, while optimally utilizing the resources of the
airlink. Scheduling in the 802.16 MAC is divided into two
related scheduling tasks: scheduling the usage of the
IEEE 802.16 Medium Access Control and Service Provisioning
217
Intel Technology Journal, Volume 8, Issue 3, 2004
airlink among the SS’s and scheduling individual packets
at the BSs and SS’s.
architecture by using the MAC functionalities as
described above.
The airlink scheduler runs on the BS and is generally
considered to be part of the BS MAC layer. This
scheduler determines the contents of the DL and UL
portions of each frame. When optional modes such as
transmit diversity, AAS, and MIMO are used, the MAC
layer must divide the UL and DL subframes into normal,
transmit diversity, AAS, and MIMO zones, to
accommodate SS’s that are to be serviced using one of
these modes. Having divided the subframes into zones,
the scheduler allocates transmission opportunities to
individual SS’s within the zone in which they operate. In
the OFDM, DL transmission opportunities are time slots,
while in the OFDM UL and OFDMA UL and DL,
transmission opportunities are time slots within individual
subchannels. When AAS with SDMA is employed within
the BS, a given time slot on a given subchannel can be
allocated to multiple SS’s. This means that the twodimensional scheduling problem (with time slots along
one axis and subchannels along the other) becomes a
three-dimensional problem, with the third axis being the
spatial axis. The MAC must determine which SS’s have
orthogonal spatial signatures, making them good
candidates for sharing the same subchannel/time slot
combinations.
802.16 and 802.11 QoS Comparison
The airlink scheduler must also determine the appropriate
burst profile for communication with each SS. The BS
monitors the signal to noise ratio (SNR) and increases or
decreases the coding rate and modulation level
accordingly for traffic for an SS. This achieves the
highest possible throughput, while maintaining a given
BER level.
The airlink scheduler determines the bandwidth
requirements of the individual SS’s based on the service
classes of the connections and on the status of the traffic
queues at the BS and SS. The BS monitors its own queues
to determine the bandwidth requirements of the DL and
utilizes a number of different communication
mechanisms (such as polling and unsolicited bandwidth
requests) to keep informed of the bandwidth requirements
of the SS’s for the UL.
The key characteristic of a Wi-Fi network is its
simplicity. An SS can roam into any Access Point (AP) or
hotspot almost without any user intervention. However,
the simplicity also comes with limitations. Even with the
QoS enhancement in the 802.11e, it can still only support
limited QoS parameters (i.e., eight user priorities) and a
single connection. 802.11 is based on a distributed
architecture, where the operation of the MAC is
coordinated among APs and SS’s. On the other hand,
WiMAX is based on a centralized control architecture,
where the scheduler in the BS has complete control of the
wireless media access among all SS’s. WiMAX can
support multiple connections that are characterized with
the complete set of QoS parameters. Moreover, WiMAX
provides the packet classifier to map these connections
with various user applications and interfaces, ranging
from Ethernet, TDM, ATM, IP, VLAN, etc. However, the
rich feature set and flexibility in WiMAX also increase
the complexity in the service deployment and
provisioning for fixed and mobile broadband wireless
access networks. In the following subsections we describe
service provisioning, auto-configuration, and the WiMAX
Management Information Base (MIB).
Service Provisioning and AutoConfiguration
Figure 5 shows the management reference model of
Broadband Wireless Access (BWA) networks. The model
consists of a Network Management System (NMS),
managed nodes, and a Service Flow Database. BS and SS
managed nodes collect and store the managed objects in
an 802.16 MIB format. Managed objects are made
available to NMSs using the Simple Network
Management Protocol (SNMP). The Service Flow
Database contains the service flow and the associated
QoS information that directs the BS and SS in the
creation of transport connections when a service is
provisioned or an SS enters the network.
Finally, there is a packet scheduler in the BS and SS. This
scheduler schedules packets from the connection queues
into the transmission opportunities allocated to the SS
within each frame.
SERVICE PROVISIONING
We first explore the differences in QoS mechanisms in
the 802.11 and 802.16 with a view towards pointing out
the challenges associated with large-scale WiMAX
deployment. Then, we describe the service provisioning
IEEE 802.16 Medium Access Control and Service Provisioning
218
Intel Technology Journal, Volume 8, Issue 3, 2004
•
S S #1
M an age d
N od e
•
M IB
BS
PHY
M AC
..
.
M ana ge d
N ode
In terne t
M IB
S S #N
M an age d
N od e
M IB
PHY
PHY
M AC
N etw o rk
M ana ge m e nt
S yste m
S e rvic e
F lo w
D a taba s e
M AC
Figure 5: Network management reference model
Figure 6 shows the MIB structure of wmanIfMib [11] for
802.16. wmanIfMib is composed of three groups:
•
wmanIfBsObjects: This group contains managed
objects to be implemented in the BS.
•
wmanIfSsObjects: This group contains managed
objects to be implemented in the SS.
•
wmanIfCommonObjects: This group contains
common managed objects to be implemented in the
BS and SS.
wmanIfMib contains the following tables to support the
service flow provisioning.
wmanIfBsProvisionedSfTable: This table contains the
pre-provisioned service flow information to be used to
create connections when a user enters the network.
•
•
•
•
•
•
Maximum traffic burst: Specifies the maximum burst
size that can be transported.
Minmum reserved rate: The rate in bits per second
specifies the minimum amount of data to be
transported on the service flow when averaged over
time.
Tolerated jitter: Specifies the maximum delay
variation (jitter) for the service flow.
Maximum latency: Specifies the maximum latency
between the reception of a packet by the BS or SS on
its network interface and the forwarding of the
packet to its RF interface.
wmanBsClassifierRuleTable: This table contains rules
for the packet classifier to map DL and UL packets to the
service flow.
• In the DL direction, when a packet is received from
the network, the classifier in the BS may use the
MAC address or IP address to determine which SS
the packet shall be forwarded to, and may use Type
of Service (TOS) or Differentiated Service Code
Point (DSCP) parameters to select the service flow
with suitable QoS.
• In the UL direction, when a packet is received from
the customer premise, the classifier in the SS may
use the source/destination MAC address or IP
address and port number, TOS/DSCP, Virtual Local
Area Network (VLAN) ID to forward the packet to a
service flow with the appropriate QoS support.
SS MAC address: a unique SS identifier to associate
the service flow with an SS.
Direction: the direction of this service flow (e.g., UL
or DL).
Service class index: a pointer to the QoS parameter
set for such service flow.
Service flow state: there are three states (i.e.,
provisioned, admitted, and activated) indicating
whether the resource is provisioned, admitted, or
active.
wmanIfBsServiceClassTable: This table contains the
QoS parameters that are associated with service flows.
The key parameters include the following:
•
•
Traffic priority: The value (0 .. 7) specifies the
priority assigned to a service flow. When two service
flows have identical QoS parameters besides priority,
the higher priority service flow should be given lower
delay and higher buffering preference.
Maximum sustained rate: Specifies the peak
information rate of the service flow in bits per
second.
IEEE 802.16 Medium Access Control and Service Provisioning
219
Intel Technology Journal, Volume 8, Issue 3, 2004
wmanIfMib
(1.3.6.1.2.1.10.184)
wmanIfBsObjects
wmanIfBsSystem
wmanIfBsRegisteredSsTable
wmanIfBsPacketCs
wmanIfBsProvisionedSfTable
wmanIfBsServiceClassTable
wmanIfBsClassifierRuleTable
wmanIfBsCps
wmanIfBsConfigurationTable
wmanIfBsChMeasurementTable
wmanIfBsPkm
wmanIfBsPkmBaseTable
wmanIfBsPkmAuthTable
wmanIfBsPkmTekTable
wmanIfBsNotification
wmanIfSsObjects
wmanIfSsSystem
wmanIfSsConfigFileEncodingTable
wmanIfSsCps
wmanIfSsConfigurationTable
wmanIfSsPkm
wmanIfSsPkmAuthTable
wmanIfSsPkmTekTable
wmanIfSsDeviceCertTable
three levels of QoS: Gold, Silver, and Bronze. sfIndex
points to the entry in the wmanBsClassifierRuleTable,
indicating which rules shall be used to classify packets on
the given service flow.
When the SS with MAC address 0x123ab54 registers into
the BS, the BS creates an entry in the
wmanIfBaseRegisteredTable in Table 7D. Based on the
MAC address, the BS will be able to find the service flow
information that has been pre-provisioned in Table 7A,
7B, and 7C. The BS will use a Dynamic Service Addition
(DSA) message to create service flows for sfIndex
100001 and 100002, with the pre-provisioned service
flow information. It creates two entries in
wmanIfCmnCpsServiceFlowTable in Table 7E. The
service flows will then be available for the customer to
send data traffic.
wmanIfSsNotification
wmanIfCommonObjects
wmanIfCmnPacketCs
wmanIfCmnClassifierRuleTable
wmanIfCmnCps
wmanIfCmnCpsServiceFlowTable
wmanIfCmnBsSsConfigurationTable
wmanIfCmnSsChMeasurementTable
A.
wmanIfBsRegisteredSsTable
D.
SS MAC
ssIndex idIndex Addr
…
wmanIfCmnPrivacy
wmanIfCmnCryptoSuiteTable
wmanIfCmnOfdmPhy
wmanIfOfdmUplinkChannelTable
wmanIfOfdmDcdBurstProfileTable
Direction
1
D
100002 123ab54
2
U
45feda1
1
D
200
1
123ab54
100003
201
2
45feda1
203
2
245ad56
100004 45feda1
Note: D – Downlink
Use DSA messages
To create service flows
and entries in the table
wmanIfOfdmUcdBurstProfileTable
QoS
Index
123ab54
100001
B.
wmanIfOfdmDownlinkChannelTable
wmanIfBsProvisionedSfTable
MAC
sfIndex SS
Addr
Figure 6: wmanIfMib structure
U
2
U - Uplink
wmanIfBsServiceClassTable
QoS
Service Max
Max
Index
Class
Data Latency
Gold
2000000
2
Silver
1000000
100
3
Bronze
512000
150
1
…
…
50
wmanIfCmnCpsServiceFlowTable
Minimizing customer intervention and truck roll is very
important for WiMAX deployments. The following
describes the service provisioning features by configuring
the Provisioned Service Flow Table, Service Class Table,
and Classifier Rule Table as described above, in order to
support self-installation and auto-configuration.
When a customer subscribes to the service, he or she will
tell the service provider the service flow information
including the number of UL/DL connections with the data
rates and QoS parameters, along with what kind of
applications (e.g., Internet, voice, or video) he or she
intends to run. The service provider will pre-provision the
services by entering the service flow information into the
Service Flow database. When the SS enters the BS by
completing the network entry and authentication
procedure, the BS will download the service flow
information from the Service Flow Database. Figure 7
provides an example describing how the service flow
information is populated. Tables 7A, 7B, and 7C
indicates that two SS’s, identified by MAC address
0x123ab54 and 0x45fead1, have been pre-provisioned.
Each SS has two service flows, identified by sfIndex, with
the associated QoS parameters that are identified by
qosIndex 1 and 2, respectively. qosIndex points to a QoS
entry in the wmanIfBsServiceClassTable that contains
E.
sfIndex
100001
sfCid
101
QoS
Index
1
100002
102
2
100003
053
1
100004
054
2
…
C.
wmanIfBsClassifierRuleTable
sfIndex
Src IP
Addr
100001
Dest IP
Addr
TOS
…
1.0.1.48
100003
1.0.1.45
115455
6.12.6.4
100002
6.12.6.5
7
100004
6.12.6.6
4
Figure 7: Service flow provisioning
IMPLEMENTATION CHALLENGES OF
THE WIMAX MAC AND QOS MODELS
The tasks performed by the 802.16 MAC protocol can be
roughly partitioned into two different categories: periodic
(per-frame) “fast path” activities, and aperiodic “slow
path” activities. Fast path activities (such as scheduling,
packing, fragmentation, and ARQ) must be performed at
the granularity of single frames, and they are subject to
hard real-time deadlines. They must complete in time for
transmission of the frame they are associated with. In
contrast, slow path activities typically execute according
to timers that are not associated with a specific frame or
the frame period and as such do not have stringent
deadlines.
IEEE 802.16 Medium Access Control and Service Provisioning
220
Intel Technology Journal, Volume 8, Issue 3, 2004
The two categories of tasks described above interact in
that the slow path activities described above typically
dictate the mode of operation of the fast path activities.
For instance, SS registration and association with a BS,
which occurs through the exchange of several messages,
results in the creation of several connections and
associated state between the SS and BS. These
connections can include state to be tracked in the fast
path such as fragmentation status, ARQ retransmissions,
and packing.
In addition to supporting the QoS and MAC functionality
described above, a set of virtualization challenges are
faced by 802.16 MAC implementers as well. Specifically,
it is expected that at system setup time it will be possible
to configure single systems to treat multiple air channels
as separate MAC instances. Thus a single BS (and
associated MAC implementation) might for example
utilize two 10 MHz channels in parallel as two separate
MAC instances. This type of virtualization is necessary
because the usage and allocation of available air
bandwidth is highly dependent on carrier policies, system
loading, and radio environment.
Supporting virtualization of the MAC layer has subtle
implications for 802.16 MAC implementation. Gross
attributes of system design such as total air bandwidth,
and thus the above-MAC data rate (Mbps) and packet
rate (PPS), is unchanged. Similarly, very fine-grained
details, such as state machines for connection setup or for
packing CS SDUs into a MAC PDU, remain the same.
However, virtualization affects intermediate-level MAC
abstracts, in that MAC state machines that deal with
states such as the list of authenticated SS’s, or whether
admission control can allow another bandwidth request,
must now be virtualized so that a set of independent
instances of each of these state machines must be
executed and coordinated with each other. Furthermore,
PHY indications must be provided such that frames from
separate bands can be distinguished and delivered to the
correct set of state machine instantiations for processing.
Finally, the multiple instantiations, while independent
from the point of view of shared state, are all executing
on the same hardware, and as such care must be taken to
ensure that MAC timeliness deadlines are still met for all
state machine instances.
In addition to virtualization, another key architectural
feature that must be supported by MAC implementations
is extensibility. Extensibility, in terms of differentiating
features such as alternative QoS scheduling algorithms,
which may not be present in the base implementation of
the MAC, is a second key challenge for MAC
implementers. Extensibility is an important feature of the
MAC protocol in that it is expected that BS
manufacturers along with their customers will desire the
ability to easily customize the scheduler and other aspects
of the MAC to differentiate their offerings from others.
The 802.16 leaves a wide variety of options and
functionality up to the implementer to determine how
best to achieve a robust service offering.
The following two sections review the implementation
challenges discussed above in the context of two
processor architectures: the Intel IXP Network processor
architecture, which utilizes core-multiprocessing and
hardware threading support, and the Intel Architecture
Pentium® M general-purpose processor architecture.
IXP Implementation
Intel IXP network processors are especially suited for
implementing
high-density
networking-related
applications like access points, routers, and gateways. It
is also a natural choice for WiMAX BSs. (It may also be
used for SS’s playing the role of residential routing
gateways). While the BS feature set is user-specific, the
802.16 MAC software is one of the most important BS
components. The provided MAC software is designed to
cooperate seamlessly with other ready-to-use IXP library
routines, available with the IXA Software Development
Kit (SDK) tool chain. Therefore it is easy to combine the
MAC with chosen IXA SDK forwarding modules, be they
IPv4, IPv6, or Multiprotocol Label Switching (MPLS).
Moreover, a rich choice of network access interfaces is
supported, e.g., Ethernet (100M, 1G, 10G), ATM
(including TM4.1), and Packet Over SONET (POS).
Figure 8 shows a sample WiMAX BS software
partitioning. The fast path activities are often referred to
as Data Plane (DP) activities, and slow path activities are
known as Control Plane (CP) activities. The CP-related
code modules deal with policies, while the DP-related
modules are concerned with execution. The CP sets
control tables used by the DP.
An IXP network processor hosts both the DP modules
and CP modules. As shown in the figure, the DP modules
run partly on IXP microengines (and are frequently
referred to as “microblocks”) and partly on the IXP
XScale® integrated control processor (the code directly
cooperating with microblocks is called “core
components”). The microblocks utilize hardware
multithreading, while the XScale code uses an embedded
®
Pentium is a registered trademark of Intel Corporation
or its subsidiaries in the United States and other
countries.
®
XScale is a registered trademark of Intel Corporation
and its subsidiaries in the United States and other
countries.
IEEE 802.16 Medium Access Control and Service Provisioning
221
Intel Technology Journal, Volume 8, Issue 3, 2004
operating kernel (e.g., Linux* or VxWorks*) to work in
multiprogramming mode. More information on the IXP
hardware, software, and tools is available at the Intel web
site [1]; see also the Intel Technology Journal [2], [3],
[4], [5].
The IXP code is directly portable across the IXP 2xxx
network processor range.
(FAPI) [6]. The core components include MAC-related
code, and also the code cooperating with the forwarder
(so-called “slow path” implementation). On top of FAPI,
there is the remaining CP software, including the MAC
signaling stack, management and monitoring applications,
etc. It is worth mentioning that it is possible to remote the
FAPI to some external control processor, using the
ForCES framework [7], Remote Procedure Call (RPC), or
Common Object Request Broker Architecture (CORBA).
For the WiMAX BS, the XScale processing power is
adequate to run all the necessary CP software by itself,
however.
The CP also controls the PHY hardware, via driver
software that is accessed by using the FAPI.
IXP Data Flows
Figure 8: Sample WiMAX BS software partitioning
The DP part includes 802.16 MAC, including UL and DL
schedulers, and typically also some forwarder module
(e.g., IPv4 router with DiffServ support). From the RF
side, it interfaces to the 802.16 PHY (OFDM, OFDMA),
implementing baseband processing, using a so-called
PHY Service Access Point Application Programming
Interface (SAP API). From the network side, this may be,
for example, a Gigabit Ethernet or ATM network,
accessible via a CS API that is compliant with an IXA
SDK framework. The interface to the CP is done using
IXP shared memory.
Some of the tasks such as handling the MAC
management messages are serviced either by the DP or
CP, depending on their relative frequency. For example,
the 802.16 DP will service Bandwidth Requests (in),
ARQ (in, out), DL-MAP (out), UL-MAP (out),
DCD/UCD (out), while the other MAC messages that are
not time critical will be passed to the CP for processing.
We call this class “signaling messages”; they are handled
according to the state machines maintained by the CP.
The CP part contains the IXA SDK infrastructure code
(implementing generic communication mechanisms
between XScale and microengines), the core components,
and Network Processing Forum (NPF)-style control API
Figure 8 also shows the data flows within the IXP
network processor. The main data stream is transferred
between the RF side and the external network. The IXP
microblocks are responsible for handling this data stream.
A part of the data stream (containing non time-critical
MAC Management messages) terminates at the CP; it is
handled by the 802.16 MAC signaling stack. Lastly, the
CP management software sets or gets configuration and
monitoring data (shared with microengines) using the
FAPI.
CP-DP Cooperation
The CP cooperates with the DP across the FAPI. The CP
issues requests, which may convey configuration data,
queries, or they may contain MAC Management
messages (to be sent to a remote SS), and it receives
responses to those requests and also asynchronous events
(e.g., MAC Management messages coming from remote
SS’s).
MAC-PHY Cooperation
The MAC and PHY layers cooperate across the PHY
SAP API. This interface enables a fast and low-latency
exchange of traffic data between PHY and MAC, and
also supports in-band PHY configuration (setting TX/RX
Vector, a data structure equivalent to DL-MAP and ULMAP, which has to be provided for the PHY frame after
frame). The interface is asynchronous and supports
multiple MAC instances, which enables parallel servicing
of many transmission channels.
It is assumed that it is PHY that maintains precise time
synchronization needed to transmit or receive a frame.
MAC is loosely coupled with PHY over the PHY SAP
API.
*
All other brands and names are the property of their
respective owners.
IEEE 802.16 Medium Access Control and Service Provisioning
222
Intel Technology Journal, Volume 8, Issue 3, 2004
MAC-Forwarder Cooperation
The CS interface utilizes a “no packet copying” approach.
The MAC prepares a handle to a control structure
pointing at a data buffer (a portion of a buffer or even a
buffer chain) when passing an SDU to a forwarder. A
forwarder uses the same mechanism when passing an
SDU to the MAC for transmission.
The MAC and a forwarder are loosely coupled via an
elasticity buffer between the two.
IXP Microblocks
Figure 9 shows the microblocks implementing the fastpath processing on IXP microengines. The current code
supports the OFDM PHY and multiple MAC instances.
The chosen architecture guarantees that the
implementation constitutes a good starting point for
implementation of future 802.16 standard extensions as
well as for cooperation with other PHY types. Part of the
code may be reused for the SS MAC implementation.
The microblocks optimize usage of the radio link and
support all service flow types on the UL direction;
they provide efficient DL traffic handling in both the
TDD and FDD mode of operation, including
handling of half-duplex SS’s. The microcode blocks
cooperate using messages passed via ring structures as
depicted in Figure 8. Because the message formats are
well-defined, it is possible to customize or even replace
certain blocks to enable easy product differentiation. In
particular, it is possible to introduce customer-designed
schedulers. This way, extensibility of the design is
guaranteed.
The other important data structures include the
Connection Record and Frame Definition. The
Connection Record holds all connection data on a per
CID and MAC instance basis. Its contents are defined by
the CP and used by the DP. The Frame Definition
structure determines the DL-MAP and UL-MAP for the
current frame.
The microblocks are described below. They are grouped
into UL Path, DL Path, and Service Blocks.
Figure 9: Data plane MAC software modules on IXP
microengines
UL Path
PHY SDU RX reassembles messages received from PHY
into PHY SDUs, prepares MAC PDUs (with validated
HCS and CRC, and decrypted if needed). It also extracts
Grant Requests (from stand-alone headers).
MAC PDU RX prepares MAC SDUs from MAC PDUs
(with unpacking and defragmentation, in two versions:
with and without ARQ), extracts ARQ feedback IEs,
piggybacked Grant Requests, and MAC Management
messages destined for the CP. It detects missing blocks
and (for ARQ connections) signals this to the ARQ
Engine. Complete MAC SDUs are passed to the
forwarder.
The UL Scheduler receives Grant Requests and plans
when those requests may be fulfilled, based on the
service parameters associated with a given connection. It
prepares the UL portion of the Frame Definition
structure. It operates on an abstract allocation unit.
Because the UL Scheduler processes input in the form of
a grant request message, and produces output to a shared
memory, a Frame Definition structure, it is possible to
move it to an XScale core component.
DL Path
MAC SDU TX handles MAC SDUs arriving from the
forwarder, CP (i.e., MAC Management messages), and
from retransmit queues (ARQ connections only). This
block performs fragmentation, if necessary. It forms
incomplete MAC PDUs (which can be later packed). For
ARQ use, it saves a copy of the portion prepared for
transmission and starts the retransmission timer.
MAC PDU TX performs MAC PDU queuing per CID,
destination SS, and Burst Profile. The amount of queued
data depends on the free space remaining in the currently
prepared frame (the information is available in the Frame
definition structure). It also does dequeuing of MAC
IEEE 802.16 Medium Access Control and Service Provisioning
223
Intel Technology Journal, Volume 8, Issue 3, 2004
PDUs for final processing and transmission. At this stage
packing and concatenation take place.
Map Builder is a PHY-specific module, which processes
the Frame Definition structure contents and produces
specifically formatted RX/TX information both for the
local PHY (as TX/RX Vector) and for remote SS PHYs
(as DL-MAP and UL-MAP MAC Management
messages).
The table below shows the raw frame sizes and
corresponding speeds possible to attain with the selected
profile (not all possible combinations are shown).
Table 1: Raw frame size and speed calculations
PHY SDU TX finalizes processing of each MAC PDU,
by preparing HCS, encrypting its payload (if required)
and generating a CRC. MAC PDUs belonging to the same
burst are then sent as a multisegment PHY SDU to the
PHY for transmission. This microblock also passes the
TX/RX Vector to the PHY and processes confirmations
from PHY (forwarded by the PHY SDU RX microblock).
Timer is a universal block, receiving wake-up requests
from the remaining microblocks and processing them in
the expiration time sequence. The Timer also processes
timeout cancellation orders. When the active timer
expires, a message is sent to the requested microblock
with sufficient context information to handle the event
correctly.
IXP MAC Performance
The 802.16 MAC microcode has been modeled using the
Intel Architecture Development Tool for IXP 2850 and
IXP 2350 network processors. The performance
estimations done on the model indicate a large processing
headroom, guaranteeing scalability and making IXP
network processors a perfect choice for multichannel and
multisector WiMAX BS implementations. The analysis
shows that both types of IXP processors can easily handle
four RF channel/four sector configurations on a single
chip.
Number of
symbols
1/32
23 6/55
215
Gaps
[physical slots]
TTG
RTG
45
45
Raw frame size per modulation [bytes]
16QAM1/2
16QAM3/4
64QAM2/3
10320
15480
20640
64QAM3/4
23220
Raw speed per modulation [Mbps]
Service Blocks
The ARQ Engine processes ARQ feedback IEs arriving
from remote SS’s and also signals coming from the local
timer and from the MAC SDU TX. It runs state machines
to maintain RX window and TX window data structures,
used to control MAC SDU reassembly and
retransmission. This block also handles resynchronization
between SS’s and BS’s, if they get out of sync.
CP
Total
symbol
length [us]
16QAM1/2
16QAM3/4
64QAM2/3
64QAM3/4
16.5120
24.7680
33.0240
37.1520
The load and headroom estimates were done for the
following scenario:
•
•
Four 10 MHz channels are used in parallel.
Modulation/coding is 64-QAM3/4. From Table 1, the
aggregate raw throughput amounts to 4 * 37.152 =
148.608 Mbps.
• DES encryption/decryption on all connections.
• ARQ active on all connections.
• Symmetric UL/DL traffic.
• IPv4 forwarder code included together with
6-tuple classifier (from DiffServ).
• mix of UL traffic: UGS (30%), nrtPS (30%), BE
(40%).
The analysis was performed using Intel’s IXP
Architecture Development Tool (ADT) implementing a
model of the 802.16 MAC software being developed by
Intel. The results of this analysis are given below. They
are preliminary and subject to change.
For estimation purposes, the following assumptions were
made:
•
•
•
•
802.16 MAC works in point-to-multipoint mode.
the PHY layer is OFDM (as defined in clause
8.3 of [8]).
Frame length is set at 5 ms.
Used profile is ProfP3_10 (10 MHz – see [8]).
IEEE 802.16 Medium Access Control and Service Provisioning
224
Intel Technology Journal, Volume 8, Issue 3, 2004
Table 2: Summary of microengine utilization for
IXP2850 (at 1.4 GHz) and IXP2350 (at 900 MHz)
IXP2850
(1.4 GHz)
IXP2350 (900
MHz)
Internal Bus
Bandwidth
3%
7%
Memory Bus
Bandwidth
7%
19%
Microengine
(ME) Utilization
4%
9%
Crypto Unit
Utilization
1%
n/a1
MAC implementation. Scalability, both in the design of
the software MAC as well as in the BS design itself, was
another key requirement. Portability of the MAC
implementation was also a key design consideration,
which goes hand in hand with scalability. A portable
MAC implementation should be able to execute on any of
the wide range of Intel architecture and XScale
architecture general-purpose processors. This section
describes in detail the scalability and portability
requirements that drove the Intel Architecture MAC
design, while the following section describes the
architectural approach chosen to satisfy the requirements
given here and in the introduction to the Implementation
Challenges section.
Scalability
Table 3: A preliminary code space and local memory
occupancy estimations for MEv2
Micro
engine
Num. of
Threads
Num. of
Instr.
Local
Mem
Words
ME #1
6 (2 free)
5350
(34% free)
480
(53
free)
%
ME #2
8 (0 free)
6100
(25% free)
576
(43% free)
ME #3
8 (0 free)
5120
(37% free)
530
(48% free)
5100
(37% free)
96
(90% free)
ME #4
6 (2 free)
Intel Architecture MAC Implementation
Goals
The 802.16 specification defines a complex, powerful
MAC protocol for achieving high bandwidth and robust
service offerings. In addition to the MAC features and
functionality described in the first part of this paper, the
following design considerations were used in architecting
the Intel Architecture BS MAC implementation of 802.16
with the OFDMA PHY. Extensibility, as described above
was a primary requirement in the Intel Architecture BS
1
Scalability is a key feature of the MAC in that it is
envisioned that BSs will have a wide variety of physical
configurations, ranging from “pico” BS’s to “macro”
systems.
In this context, a pico BS might be deployed mounted on
a pole with a small, single sector and single
omnidirectional antenna, perhaps with limited bandwidth
and tight power and heat limitations, and subject to
outdoor environment-level temperatures. At the other
extreme, a heavy iron BS might be rack mounted, support
multiple sectors, have many antennae, and be in an
environmentally controlled cabinet or small building, with
a large antenna tower connected to it.
As such, it must be possible for the MAC software
implementation to be usable with the wide range of
processor performance levels available with generalpurpose processors such as Intel Architecture processors.
The system must be implemented such that performance
scales in a predictable fashion with processor
performance, allowing appropriate processors to be
chosen for executing the MAC software.
Portability
Portability is a key feature of the Intel Architecture MAC
implementation for similar reasons. The wide range of
performance and price points likely to be associated with
WiMAX BSs creates the need to easily choose different
processors based on power, price, heat, and performance
metrics. The Intel Architecture MAC design takes this
feature as a primary goal, providing a complete and
robust MAC offering while at the same time allowing it to
be ported across the range of Intel general-purpose
The current ADT version does not support IXP2350
crypto unit modeling. It is assumed that the crypto unit
will handle the expected load, since its bandwidth is 200
Mbps.
IEEE 802.16 Medium Access Control and Service Provisioning
225
Intel Technology Journal, Volume 8, Issue 3, 2004
processor architectures, including Pentium® M, Pentium
4, Xeon®, XScale, and Celeron®.
added to the low-priority queue to process the message
(by invoking the QoS admission control event handler).
Intel Architecture MAC Features and Design
I/O-driven events are events that are added to one of the
priority queues based on reception of I/O of some sort.
Thus, notification by the 802.16 PHY that a new UL
frame has been fully received results in an event being
added to the medium-priority event queue for parsing of
the received frame. Similarly, delivery of an Ethernet
packet to the packet convergence sublayer results in an
event being queued to the low-priority event queue for
classification of the Ethernet frame into a per-CID queue.
As explained previously, the tasks that make up 802.16
can be divided into two categories: time-critical, periodic
operations that must occur on every frame, and slower,
less demanding aperiodic operations that typically operate
over the duration of several frames. In order to support
this mixture of processing tasks, the Intel Architecture
implementation of 802.16 uses a multilevel hard real-time
priority-based scheduling system. The scheduling system
utilizes three priority levels of events: high, medium, and
low. High-priority events are those events that must
always be serviced in a timely fashion, and must not be
executed past their deadline or the basic functionality of
the MAC will be compromised. Medium-priority events
are events that have strict time requirements, but if their
deadlines are missed they may be skipped without
causing a catastrophic failure. Finally, low-priority events
are events that typically do not have strict processing
requirements associated with them; they are processed on
a best-effort basis whenever processing time is available.
All functionality involved in 802.16 MAC processing is
implemented as one or more events, all of which fall into
one of three categories: periodic events, protocol-driven
events, and I/O-driven events. Periodic events are events
that occur with a known and fixed regularity. For
example, delivery of a ready frame by the MAC software
to the PHY device driver for transmission is a highpriority event that occurs exactly once every frame period
(typically 2.5-5 ms). Generating the UL-MAP that is part
of the ready frame (by the UL scheduler) is another
periodic high-priority event that occurs exactly once
every frame period.
Protocol-driven events are events that are added to one of
the priority queues based on external stimuli associated
with the 802.16 MAC itself. For example, reception of a
DSA-REQ message from an SS results in an event being
®
Pentium is a registered trademark of Intel Corporation
or its subsidiaries in the United States and other
countries.
®
Xeon is a registered trademark of Intel Corporation or
its subsidiaries in the United States and other countries.
®
Celeron is a registered trademark of Intel Corporation
or its subsidiaries in the United States and other
countries.
All events have associated with them an earliest
acceptable start time and a deadline time. If the
associated event handler is invoked within this time
interval it runs to completion, with medium- and lowpriority event handlers always being implemented such
that they have relatively small run times (perhaps
resulting in scheduling of another follow-up event to
continue processing later). If an event handler is not
executed before its deadline, it instead will have a special
late invocation call made that allows it to triage the
missed event as best as possible.
The combination of event priority levels and controlled
execution times allows the entire system to scale in a
predictable, controlled fashion. Low-priority events, such
as handling of newly received Ethernet frames or
negotiating a request to set up a new connection, will
never cause the system as a whole to miss high-priority
event deadlines such as frame transmission times. This
ensures that the system will always function correctly no
matter what the maximum load, dropping low-priority
traffic rather than becoming unsynchronized with the
PHY, for instance. Conversely, as available processing
power is increased, the system can scale to handle more
and more medium- and low-priority events, thus being
scalable to higher bandwidth configurations through the
use of more powerful processors. The need for scalability
of the 802.16 MAC is one of the key challenges identified
in the implementation of this protocol, and the use of an
event-based, real-time-scheduled system is a powerful
and flexible method for achieving such scalability.
The use of an event-based system with the associated
event handlers allows for great flexibility in
implementation. Each event handler can be customized in
its implementation, and as long as the specified pre- and
post-conditions are met, along with the maximum
execution time, the system implementation will work
correctly. This predictable execution behavior, eventbased system, and flexibility of the system allows the
virtualization requirement to be easily met, because the
events and associated state machines for the MAC can be
IEEE 802.16 Medium Access Control and Service Provisioning
226
Intel Technology Journal, Volume 8, Issue 3, 2004
multi-instanced in order to utilize multiple virtual MACs
in support of multiple air channels.
The requirement of portability in this implementation is
achieved through the selection of programming language
for the entire implementation, which is ANSI C.
Furthermore, the implementation is implemented in an
endian-neutral fashion and only uses explicitly sized
types. This makes it very simple for Telecommunications
Equipment
Manufacturers
(TEMs)
or
carrier
programmers to understand the existing code and port it
across the range of Intel architecture general-purpose
processors to suit their heat, price, and performance
needs. Furthermore, the use of event handlers with known
pre- and post-conditions and maximum execution
periods, along with the use of ANSI C, allows for simple
extensibility and customization of the 802.16 MAC. Thus
the key challenge of extensibility is met and the goal of
portability is achieved in the Intel Architecture-based
implementation while still providing a complete and
robust MAC implementation.
CONCLUSION
The IEEE 802.16 is a very complicated standard,
featuring high adaptiveness to maximize airlink usage;
therefore, it requires sophisticated algorithms. At the
same time, its implementation should expose ease-of-use
for users and provide adequate QoS. Consequently, the
802.16 MAC poses significant challenges to the BS
software implementer. Hard real-time deadlines must be
met while still maintaining high throughput and
predictable behavior. The two MAC implementations
described above, which are available on Intel IXP
network processors and Intel Architecture Pentium M
processors, provide complete, robust implementations of
the 802.16 specification, while at the same time also meet
the additional stated goals of virtualization and
extensibility presented in the introduction to this paper.
The existence of two 802.16 BS MAC implementations
enables equipment manufacturers to select the MAC
software and associated processor architecture that best
meets their power, price, portability, and performance
needs.
[2] Adiletta, Matthew et al., “The Next Generation of
Intel IXP Network Processors,” in Intel Technology
Journal, Volume 6, Issue 3, 2002.
[3] Naik, Uday et al., IXA Portability Framework:
Preserving Software Investment in Network Processor
Applications in Intel Technology Journal, Volume 6,
Issue 3, 2002.
[4] Deval, Manasi et al., “Distributed Control Plane
Architecture for Network Elements,” in Intel
Technology Journal, Volume 7, Issue 4, 2003.
[5] Vinnakota, Bapi et al., “Scalable Intel IXA and its
Building Blocks for Networking Platforms,” in Intel
Technology Journal, Volume 7, Issue 4, 2003.
[6] FAPI Model & Usage Guidelines, June 10, 2004,
npf2002.340.32 http://www.npforum.org/*
[7] IETF ForCES Working Group:
http://www.ietf.org/html.charters/forcescharter.html.*
[8] IEEE™ P802.16-REVd/D5-2004: “Air Interface for
Fixed Broadband Wireless Access Systems.”
[9] Govindan Nair, MAC 802.11 Point Coordination
Function:
http://www.intel.com/cd/ids/developer/asmona/eng/52768.htm
[10] Carl Eklund, Reger B. Marks, Kenneth L. Stanwood,
and Stanley Wang, “IEEE™ Standard 802.16:
Technical Overview of the Wireless MAN Air
Interface for Broadband wireless Access,” IEEE™
Communications Magazine, June 2002.
[11] J. Chou, R. Reynold, V. Yanover, S. Eini, R. Selea,
B. Moldoveanu, “MAC and PHY MIB for
WirelessMAN and WirelessHUMAN BS and SS,”
http://grouper.ieee.org/groups/802/16/mgt/contrib/C80216m
gt-04_04.pdf*
[12] WiMAX PICS wiMAX Forum, “PICS for
WirelessMAN-OFDM and WirelessHUMAN
(-OFDM).”
ACKNOWLEDGMENTS
[13] IEEE™ 802.11 “Wireless LAN Medium Access
Control (MAC) and Physical (PHY) Layer
Specifications.”
The authors thank the reviewers: Al Dabagh Baraa,
Shlomo Ovadia, and Henry Mitchel.
AUTHORS’ BIOGRAPHIES
REFERENCES
[1]
For Intel IXA Network Processor resources: visit
http://developer.intel.com/design/network/products/np
family/index.htm.
Govindan Nair is a senior software engineer in the
Broadband Wireless Division where he is involved in
software architecture, design and implementation of the
802.16 MAC and device drivers. Govindan co-authored
PPP Static Interoperability Testing–Working Text (WT)
52 in DSL Forum, and he published the 802.11 MAC
IEEE 802.16 Medium Access Control and Service Provisioning
227
Intel Technology Journal, Volume 8, Issue 3, 2004
PCF implementation in the Intel Services Forum.
Govindan received his M.S. degree in Computer Science
from Manonmanium Sundaranar University, India. His email is govindan.nair at intel.com.
Joey Chou is a customer architect for the Wireless
Broadband Division. Joey is actively involved in the
IEEE 802.16 Working Group and WiMAX Forum by
leading the 802.16 MIB and Service Provisioning works,
respectively. Joey was a key contributor to the VoDSL
and VoIP standard works in the ATM Forum and DSL
Forum, and was editor of the VoDSL Implementation
Guideline and Interoperability Test Plan in the OpenVoB
Forum. Prior to joining Intel in 1999, he worked at GTE,
Siemens, AT&T, and Motorola on numerous telephony
and narrowband and broadband wireless projects. He
received a M.S. degree in Electrical Engineering from the
Georgia Institute of Technology in 1985. His e-mail
address is joey.chou at intel.com.
degree in Computer Science from the University of
Illinois at Urbana-Champaign. His e-mail is david.putzolu
at intel.com.
Jerry Sydir is a senior srchitect in the Broadband
Wireless Division where he is involved in architecture of
802.16 baseband processors. Jerry has worked in a
variety of hardware and software projects in the
telecommunications area. Jerry received an M.S. degree
in Systems Engineering and a B.S. degree in Computer
Engineering from the Case Western Reserve University.
Jerry’s
professional
interests
include
wireless
communications, smart antenna technologies, and
network protocols. His e-mail is jerry.sydir at intel.com.
Copyright © Intel Corporation 2004. This publication
was downloaded from http://developer.intel.com/.
Legal notices at
http://www.intel.com/sites/corporate/tradmarx.htm.
Tomasz Madejski received an M. Sc. degree from the
Technical University of Gdansk, Poland in 1995. He
joined Intel in 1999. He has 11 years of networking
industry experience, mainly in the area of ATM.
Currently, he is a senior architect in the Modular
Communication Platform Division responsible for the
design of an 802.16 MAC/scheduler implementation on
the IXP2xxx line of network processors. His main
interest areas are in quality of service in wireless
networks and architecture of traffic schedulers. His email is tomasz.madejski at intel.com.
Krzysztof Perycz is a senior staff architect on Intel’s
MCPD team. He is currently responsible for 802.16
MAC/Scheduler design for the IXP2xxx series network
processors. He holds an M.S. degree from the Technical
University of Gdansk, Poland and has 30 years of
industrial experience. He held various R&D positions at
ZETO DP Center, Telecommunication Institute Poland,
CrossComm Corp. and Olicom, before joining Intel in
1999. He authored nine papers, and holds one Polish
patent and has filed for four US patents. His main
professional interest focuses on telecommunication and
computer science. His e-mail is krzysztof.perycz at
intel.com.
David Putzolu is a senior staff architect in the
Broadband Wireless Division where he is involved in
software architecture and implementation of the 802.16
MAC. David’s areas of interest are wireless networks and
modular software architectures for network equipment.
David was principal editor and co-authored several
Implementation Agreements in the Software Working
Group of the Network Processing Forum, and he is cochair of the IETF Forwarding and Control Element
Separation Working Group. David received his M.S.
IEEE 802.16 Medium Access Control and Service Provisioning
228
Multiple-Antenna Technology in WiMAX Systems
Atul Salvekar, Intel Communications Group, Intel Corporation
Sumeet Sandhu, Corporate Technology Group, Intel Corporation
Qinghua Li, Corporate Technology Group, Intel Corporation
Minh-Anh Vuong, Intel Communications Group, Intel Corporation
Xiaoshu Qian, Intel Communications Group, Intel Corporation
Index words: Alamouti, MIMO, diversity, AAS, WiMAX, broadband wireless
ABSTRACT
WiMAX is a wireless technology that provides broadband
data at rates over 3 bits/second/Hz. In order to increase
the range and reliability of WiMAX systems, the IEEE
802.16-2004 standard supports optional multiple-antenna
techniques such as Alamouti Space-Time Coding (STC),
Adaptive Antenna Systems (AAS) and Multiple-Input
Multiple-Output (MIMO) systems. In this paper, we
focus on techniques that do not require channel
knowledge at the transmitter, which include both
Alamouti STC and MIMO, but not AAS.
In the first half of the paper, we present simple diversity
schemes that require only a single RF chain at the
receiver. The performance of STC is compared with nonSTC performance. Simulations show that STC buys 2-10
dB over a single antenna system, which can double the
cell range and quadruple the cell coverage. For STC,
multiple Radio Frequency (RF) chains are implemented at
the Base Station (BS) to shift cost away from Subscriber
Stations (SS), thus enabling market penetration for firstgeneration, high-performance IEEE 802.16-2004
networks. We then concentrate on other simple standardcompliant diversity schemes that require only a single
receive chain at the SS: delay diversity and selection
diversity.
The second half of the paper investigates standardcompliant MIMO techniques and proposes new nonstandard advanced algorithms for open-loop MIMO. A
novel space-frequency bit-interleaver that buys 2-4 dB
over a frequency-only interleaver is presented. A 2x2
MIMO can double the throughput at a reduced range. An
iterative receiver is introduced to recover range, which
buys up to 5 dB with additional baseband complexity.
Multiple-Antenna Technology in WiMAX Systems
The intent of this paper is to provide an idea of the
benefits of multiple antenna systems over single antenna
systems in WiMAX-type deployments.
INTRODUCTION
Wireless broadband promises to bring high-speed data to
multitudes of people in various geographical locations
where wired transmission is too costly, inconvenient, or
unavailable. WiMAX is a technology devoted to making
broadband wireless commercially available to the mass
market by employing IEEE 802.16 standards-based
technology. Other important wireless standards include
IEEE 802.11, which is devoted to high-speed Local Area
Networks (LANs) and IEEE 802.15, which is devoted to
short-range Personal Area Networks (PANs).
WiMAX technology is based on the IEEE 802.16
specification of which IEEE 802.16-2004 and 802.16e
amendment are Physical (PHY) layer specifications. The
IEEE 802.16-2004 standard is primarily intended for
stationary transmission while IEEE 802.16e amendment
is intended primarily for both stationary and mobile
deployments.
While there are multiple modulations defined in the IEEE
802.16 standards, in this paper, we examine Orthogonal
Frequency Division Multiplexing (OFDM) because of
OFDM’s robustness to multipath propagation and its ease
for utilizing multiple antenna techniques [1].
Furthermore, we focus on IEEE 802.16-2004 technology
as it has already been ratified.
IEEE 802.16-2004 currently supports several multipleantenna options including Space-Time Codes (STC),
Multiple-Input Multiple-Output (MIMO) antenna systems
and Adaptive Antenna Systems (AAS).
229
Intel Technology Journal, Volume 8, Issue 3, 2004
There are several advantages to using multiple-antenna
technology over single-antenna technology:
•
Array Gain: This is the gain achieved by using
multiple antennas so that the signal adds
coherently.
•
Diversity Gain: This is the gain achieved by
utilizing multiple paths so that the probability
that any one path is bad does not limit
performance. Effectively, diversity gain refers to
techniques at the transmitter or receiver to
achieve multiple “looks” at the fading channel.
These schemes improve performance by
increasing the stability of the received signal
strength in the presence of wireless signal
fading. Diversity may be exploited in the spatial
(antenna), temporal (time), or spectral
(frequency) dimensions.
•
Co-channel Interference Rejection (CCIR): This
is the rejection of signals by making use of the
different channel response of the interferers.
A common scheme that exhibits both array gain and
diversity gain is maximal ratio combining: this combines
multiple receive paths to maximize Signal to Noise Ratio
(SNR). Selection diversity, on the other hand, primarily
exhibits diversity gain; the signals are not combined;
rather, the signal from the best antenna is chosen.
For AAS, multiple overlapped signals can be transmitted
simultaneously using Space Division Multiple Access
(SDMA), which is a technique that exploits the spatial
dimension to transmit multiple beams that are spatially
separated [3]. SDMA makes use of CCIR, diversity gain,
and array gain. A good tutorial on AAS can be found in
[3].
For MIMO systems, spatial multiplexing is often
employed. Spatial multiplexing transmits coded data
streams across different spatial domains. Some
techniques, such as BLAST [6] do not require feedback,
while others, such as vector coding on the modes of the
channel [7], do. MIMO techniques can also make use of
CCIR, diversity gain, and array gain. A form of
transmission codes used in MIMO systems are STC. A
good review of techniques for STC and MIMO can be
found in [13 and 14].
The higher performance and lower interference
capabilities of MIMO and AAS make them attractive over
other high-rate techniques for WiMAX systems in costly,
licensed bands.
Base Station (BS). The Alamouti code provides maximal
transmit diversity gain for two antennas [2]. Another
transmit diversity scheme is cyclic delay diversity. A key
advantage of transmit diversity is that it can be
implemented at the BS, which can absorb higher costs of
multiple antennas and associated RF chains. This shifts
cost away from the SS, which enables faster market
penetration of 802.16 products.
One of the many advantages of OFDM technology is the
ease with which multiple-antenna techniques can be
utilized to increase range and throughput (a system
description is given below). Using this general system
model, we show the primary advantage of OFDM systems
over single-carrier systems in multipath propagation
environments to explain why OFDM is conceptually less
complex in AAS and MIMO systems. We then discuss a
fixed point implementation of the Alamouti receiver. The
fixed point simulations show several performance
enhancements. Several practical aspects of the technology
are also discussed. Next, we discuss several other simple
diversity options, cyclic delay diversity and selection
diversity, to improve system performance. We then
describe more advanced schemes that could be used to
achieve even higher throughput. We introduce open-loop
techniques for multiple-antenna systems, which include
standard compliant MIMO equalization, spatialfrequency interleaving, and iterative decoding. Simulation
models are discussed that show large performance
improvements.
SYSTEM DESCRIPTION
We describe the Physical (PHY) layer of the general
communication system. The performance of the PHY
layer is strongly correlated to the overall system
performance. However, higher-level entities such as
Automatic Request (ARQ) for retransmission can also
impact system performance.
A wireless environment suffers from multipath
propagation. Multipath propagation, also known simply
as multipath, is a condition where multiple reflections of
the transmitter waveform arrive at the receiver at
different times. This is shown in Figure 1, where a and b
are the gains of the paths and τ1 and τ2 are the delays. The
reflected path is actually the sum of multiple reflections
from the obstruction, which causes fading. Multipath
propagation induces Inter-Symbol Interference (ISI)
which is traditionally compensated for by equalizers in
single-carrier systems [4].
For WiMAX, the simplest MIMO system is actually a
Multiple-Input Single-Output (MISO) STC code called
the Alamouti code. This requires two antennas at the
Multiple-Antenna Technology in WiMAX Systems
230
Intel Technology Journal, Volume 8, Issue 3, 2004
X1[K]
Y1 [K]
H11[K]
H21[K]
X2[K]
Y2[K]
H12[K]
H22[K]
H23[K]
H31[K]
Y3[K]
Transmitter
Figure 1: Conventional wireless system
Equalizers are computationally intense compared to the
processing required in OFDM systems. Hence, OFDM is
preferable in multipath propagation scenarios. A block
diagram of OFDM is shown in Figure 2. As long as the
CP, or Cyclic Prefix, is longer than the difference in
multipath propagation arrival times, or multipath spread,
an equalizer is not needed. The CP prepends the output of
the Inverse Fast Fourier Transform (IFFT) with the last L
samples of the IFFT output, where L is the length of the
CP.
Receiver
Figure 4: MIMO channel
In Figure 4, Yi[k] is the kth subcarrier output for receive
antenna i, Hij[k] is the kth subcarrier gain from the jth
transmit antenna to the ith receive antenna, and Xj[k] is
the kth subcarrier input from antenna j.
In the single carrier case, each of the matrix elements
would be multipath propagation channel responses.
Conceptually, the signal processing is much more
complicated; however, such systems can be simplified.
So, without loss of generality, rewriting the above
equation, for an OFDM system would be
Y =H∗X +N
Figure 2: The OFDM system
For terminology, X[k] is the transmitted information
symbol on subcarrier k. For subcarrier k, H[k] is the
scalar subcarrier response and its value is related to the
FFT of the digitized channel response h(t), V[k] is the
noise, and Y[k] is the output. The complete set of inputs
{X[k]} is called the transmit OFDM symbol, and the set
of demodulated signals {Y[k]} is called the receive
OFDM symbol. On a subcarrier by subcarrier basis, there
is no need for an equalizer.
Consider a MIMO system without noise as shown in
Figure 4. In this figure, each ray corresponds to a
multipath propagation channel. From the point of view of
a subcarrier, each multipath propagation channel
collapses to a single scalar tap. For subcarrier k, this can
be expressed as shown in Figure 3 below.
Y1 [k ]   H 11[k ] H 12 [k ]
Y [k ] =  H [k ] H [k ] ∗  X 1 [k ] 
22
  X [k ]
 2   21
Y3 [k ]  H 31 [k ] H 32 [k ]  2 
Figure 3: MIMO channel model
Multiple-Antenna Technology in WiMAX Systems
(eq. 1)
where Y, H, and X are the appropriate generalizations of
the 2 transmit x 3 receive antenna system and N is the
noise and interference. For general systems, H is an Mr
by Mt matrix representing the number of transmit and
receive antennas, respectively.
For an Additive White Gaussian Noise (AWGN) channel,
the maximum achievable theoretical data rate of this
system is given by the Shannon capacity formula [11]
(
C = log det I +
P
BNo M t
)
HH *
where P is the transmit power, No is the noise power
spectral density, and B is the signal bandwidth. An Mt x
Mr MIMO system can provide up to M=min (Mt, Mr)
times the spectral efficiency of a 1x1 system. This linear
relationship also holds true for outage capacity, which is
equal to percentiles of the cumulative distribution
function of C.
STC AND OTHER STANDARDCOMPLIANT DIVERSITY SCHEMES
In order to increase the rate and range of the modem,
there are several considerations. Generally, the BS can
incur more cost and complexity than the SS, so multipleantenna chains are a good option at the BS, which can
then apply receiver diversity techniques. These
231
Intel Technology Journal, Volume 8, Issue 3, 2004
techniques include switched diversity and maximal ratio
combining. To balance the link, the SS needs to have
improved performance. Transmission diversity schemes
are utilized at the BS that require only one receive
antenna at the SS. Two transmit diversity schemes are
cyclic delay diversity and Alamouti transmission. We
focus on Alamouti transmission.
Alamouti Transmission
The Alamouti transmission scheme is an STC in that it
sends information on two transmit antennas and consists
of two consecutive transmissions in time. Hence it
transmits information in space and time.
In IEEE 802.16-2004 OFDM-256 the Alamouti code is
applied to a specific subcarrier index k. For instance,
suppose that in the uncoded system S1[k] and S2[k] are
sent in the first and second OFDM symbol transmissions.
The Alamouti encoded symbols send S1[k] and S2[k] off
the first and second antennas in the first transmission and
-S2*[k] and S1*[k] off the first and second antennas in the
next transmission.
The receiver demodulates the received waveform by a
few simple operations as follows. Let Y1[k] and Y2[k] be
the first and second receive OFDM symbols, respectively.
Let C1[k] and C2[k] be the channel response for the kth
subcarrier of the first and second transmit antennas.
C1 * [k ]Y1[k ] + C 2 [k ]Y2 * [k ] =
(|| C [k ] || 2 + || C [k ] || 2 ) Sˆ [k ] +
1
2
1
C1 * [k ]V1[k ] + C 2 [k ]V2 * [k ]
(eq. 2)
C 2 * [k ]Y1 [k ] − C1 [k ]Y2 * [k ] =
(|| C [k ] || 2 + || C [k ] || 2 ) Sˆ [k ] +
1
2
2
C 2 * [k ]V1 [k ] − C1 [k ]V2 * [k ]
If the noise V1[k] and V2[k] are uncorrelated, then the
overall SNR is the maximum achievable and equal to
(||C1[k]||2+||C2[k]||2)(Signal Energy/Noise Energy). Notice
that both C1[k] and C2[k] need to be in a fade for the
overall processed symbol to be in a deep fade. This
system has two-fold diversity. For k-fold diversity, the Bit
Error Rate (BER) is proportional to (1/SNR)k in a fading
environment.
subcarrier 2. This means that each set of data needs to be
appropriately smoothed, which is done in these
simulations. The second is that the pilots have certain
degenerate situations: for the first Alamouti transmitted
symbol, the pilots destructively add and for the second
Alamouti transmitted symbol, the pilots constructively
add. Hence, the pilots cannot always be useful. Properly
processing the pilot symbols is required. In the
simulations, such a technique is used.
We present block diagrams detailing the flow of an
Alamouti implementation. This implementation has two
parts. The first calculates the parameters that are
necessary for data demodulation such as channel
estimates. The second part is the actual data
demodulation and tracking.
Figure 5 describes the parameter estimation portion. In
this part, two channels are estimated, and those channel
estimates are used to calculate the Viterbi equalizer
coefficients. Ei, is the average energy of the ith transmit
path. This is a computationally intensive portion of the
Alamouti reception; however, it is a one-time
computation per burst, so is feasible.
Alamouti Performance Simulations
For the purposes of simulations, three scenarios are
simulated each of which are important to typical system
vendors. The first set uses an AWGN channel that is the
baseline for performance results. In AWGN, BER is the
most important metric. The second set of simulations
uses a frequency selective channel normalized so that its
average SNR is equal to the instantaneous SNR. These
simulations show the performance in frequency selective
channels. In fixed wireless scenarios, the receive SNR
does not change rapidly, so the average BER during
multiple instantiations of the channel is of interest.
Finally, in the third set of simulations, the channel is
fading. In non-mobile situations, the fading rate is slow,
so it is of interest to determine how often the system does
not provide good performance. The Packet Error Rate
(PER) is a good metric. A fixed-point model of the
Alamouti scheme is simulated under the following
conditions:
•
Full bandwidth IEEE 802.16-2004 OFDM-256
•
Stanford University Interim (SUI)-3 model
•
3.5 MHz bandwidth
Alamouti Implementation Details
•
Varying SNR
There are a number of features to IEEE 802.16-2004
OFDM-256 Alamouti transmission that are of interest.
The first is that the preamble for Alamouti transmission is
transmitted from both antennas with the even subcarriers
used for antenna 1 and the odd subcarriers used for
•
No timing/frequency offset or drift
Multiple-Antenna Technology in WiMAX Systems
All the blocks in Figure 5 are executed. The results are
shown in Figures 6 and 7.
232
Intel Technology Journal, Volume 8, Issue 3, 2004
The 3 dB theoretical gain, as indicated by Equation 2, is
not met at BER=10-3. We expect that at lower BERs, the
curves will be closer to expected theoretical gains.
Figure 5: Alamouti parameter estimation
A normalized channel has the average channel energy
normalized to a constant so that instantaneous SNR for
the realization is equal to the average SNR. We show
the performance results in Figures 8 and 9.
Figure 6: BER vs. SNR(dB) for BPSK rate ½
Figure 8: BER vs. SNR (dB) for BPSK rate ½ in
SUI-3 channel
In the normalized SUI-3 configurations the gain is more
than 3 dB. The main conclusion to draw is that the
frequency selectivity can cause deep notches, which the
error correction cannot correct; however; the sum
channel may not have as deep notches, thereby
improving performance beyond the simple 3 dB gain
found in AWGN channels.
We now reproduce the results in a fading environment.
The main difference between the next simulation and
the earlier ones is that SUI-3 fading channels are used.
Figure 7: BER vs. SNR (dB) for 64-QAM rate 3/4
To judge the scheme in the presence of frequency
selectivity, we simulate a SUI-3 normalized channel.
Multiple-Antenna Technology in WiMAX Systems
In fading channels, PER is a better performance metric,
since in slowly fading channels, the channel will be in a
233
Intel Technology Journal, Volume 8, Issue 3, 2004
fade for a long period of time. The results are shown in
Figures 10 and 11.
At a 1% PER rate, the gain is quite significant. The PER
increase is over 5 dB for the BPSK transmission and
over 10 dB for the 64-QAM transmission.
Figure 11: PER vs. SNR (dB) for 64-QAM rate 3/4
SUI-3 channel
OTHER DIVERSITY SCHEMES
Figure 9: BER vs. SNR (dB) for 64-QAM rate ¾
SUI-3 channel
In the rest of this section we compare various diversity
schemes using floating point models. We primarily
depict relative gains since some of the non-ideal modem
behavior will not be simulated. We focus on the
subscriber side. SS’s are typically cost sensitive, hence
we focus on single receive chain systems.
The primary forms of diversity we examine are
selection diversity and cyclic delay diversity. These are
two forms of diversity that do not necessarily have an
impact on standards-compliant modems.
Consider the following block diagram:
Figure 10: PER vs. SNR (dB) for BPSK rate ½ SUI3 channels
Multiple-Antenna Technology in WiMAX Systems
Figure 12: Example of selection diversity
In selection diversity, the receiver chooses the “best”
antenna to receive. The additional hardware requirement
is simply a switch and an antenna. Many performance
metrics can be optimized. For non-multipath
propagation channels, the strongest received signal is
typically the “best” antenna. For multipath propagation
channels, the optimization can be more complicated, for
example, the maximum geometric SNR [5]. In the
following simulations, the selected antenna was that
234
Intel Technology Journal, Volume 8, Issue 3, 2004
which had the highest signal power. Selection diversity
is a form of receive diversity.
Figure 13: Transmit diversity scheme using cyclic
delay diversity
Figure 13 depicts cyclic delay diversity. As shown
cyclic delay diversity is a transmission diversity scheme.
Details of cyclic delay diversity can be found in [2].
Basically, consider the transmit sequence before
appending the CP, x[n]. Then the “delayed version” that
is transmitted off the second antenna is x’[n]=x[((nm))NFFT], where m is the delay, ((.))a represents the
modulo operation, and NFFT is the FFT size. Z(t) and
Z’(t) are the outputs from the antennas following digital
and analog processing.
Simulation Results
In this section we compare these two simple diversity
techniques. The setup is the same as in the Alamouti
case, where the channel model is a correlated SUI-3
channel including fading as found in IEEE 802.16. We
simulate 64 byte packets, which represent the ACK
from Ethernet transmission/reception. Figure 14 shows
the simulation results.
In typical WiMAX environments, simple schemes such
as selection diversity and cyclic delay diversity can give
over 4 dB in performance gains. Such simple schemes
can increase coverage and throughput. For selection
diversity, a switch and another antenna are needed, and
for cyclic delay diversity, an additional transmit chain is
necessary. As this cost is at the BS, the extra transmit
chain is usually acceptable.
Figure 14: PER as a function of SNR
MULTIPLE-INPUT MULTIPLE-OUTPUT
FOR THROUGHPUT AND RANGE
MIMO multiplies the point-to-point spectral efficiency
by using multiple antennas and RF chains at both the BS
and the SS. MIMO achieves a multiplicative increase in
throughput compared to Single Input, Single Output
(SISO) architecture by carefully coding the transmitted
signal across antennas, OFDM symbols, and frequency
tones. This gain is achieved at no cost in bandwidth or
transmit power. These simulation results assume ideal
channel estimation, channel estimate smoothing, and
perfect synchronization.
We concentrate on open-loop systems in this paper.
These do not require feedback of channel information to
the transmitter. AAS and some MIMO techniques
require some amount of channel knowledge at the
transmitter. This information can be implicitly estimated
using reciprocity in Time Division Duplex (TDD)
systems or may be explicitly signaled back to the
transmitter in Frequency Division Duplex (FDD)
systems. In a slowly changing system such as IEEE
802.16-2004, channel knowledge may remain valid for a
long time. In a mobile system like that defined in the
IEEE 802.16e amendment, however, the channel may
change quickly and require frequent feedback updates.
The overhead of channel feedback may become
significant for mobile FDD systems. MIMO is an
attractive solution for such systems because some
methods do not require channel knowledge: it maintains
the link by exploiting spatial diversity.
Outage capacity is closely related to PER, which is
often used to evaluate performance. In the next subsection, we present the design and performance of a
space-frequency interleaver for mapping coded bits to
Multiple-Antenna Technology in WiMAX Systems
235
Intel Technology Journal, Volume 8, Issue 3, 2004
tones and antennas. With an optimal receiver, this
interleaver can provide M times the spectral efficiency
of a 1x1 system at a given range depending on the
channel conditions.
data
bits
802.16d
encoder +
puncturer
Spacefrequency
interleaver
Bits to QAM ,
Spatial
mapper
IFFT - CP
IFFT - CP
Transmitter
FFT - CP
FFT - CP
Spatial
demapper
(MMSE/MAP)
QAM to
soft bits
Deinterleaver
depuncturer
decoder
estimated
bits
Receiver
Figure 15: System block diagram for spacefrequency interleaving
The simplest MIMO receiver is the zero-forcing
receiver that inverts the channel, thus recovering M =
min(Mt , Mr) transmitted data streams. However, this
inversion can cause noise enhancement. A better
receiver is the Minimum Mean Squared Error (MMSE)
receiver that performs a weighted inverse so as not to
magnify noise in the poor channel modes. In general,
the optimal receiver that minimizes the probability of
error (and achieves capacity) is the Maximum
Likelihood (ML) receiver or the Maximum A Posteriori
Probability (MAP) receiver. The transmission source
may also have a code incorporated. For instance, the
OFDMA section of IEEE 802.16-2004 contains
transmission matrices for STC that can be used in
conjunction with these reception techniques. The
performance of some MIMO receivers is outlined in the
following sub-sections.
MIMO Transmitter: Space-Frequency
Interleaving
Space-frequency interleaving is a simple way to provide
diversity gain to a spatially multiplexed, coded data
stream. This method is not currently standardcompliant. The block diagram for the Space-Frequency
Interleaver (SFI) transmitter and receiver is illustrated in
Figure 15. Information bits are first encoded by a
Forward Error Correction (FEC) encoder, which is a
concatenation of Reed-Solomon and convolutional
encoders in OFDM-256 IEEE 802.16-2004. After
puncturing, the binary coded bits are sent to an SFI,
which maps bits to antennas and tones so as to exploit
full diversity in both space and frequency. The
interleaved bits are then mapped to Gray coded QAM
data symbols. The receiver uses the MMSE receiver,
and it sends the soft bits into the concatenated
convolutional and Reed Solomon decoders.
Multiple-Antenna Technology in WiMAX Systems
Details of the interleaver design are available in [12].
We provide a short description here. Let q be the
number of bits per QAM symbol, assume 192 data tones
(256 point FFT with 64 guard tones+pilots) and M
transmit antennas. The interleaver consists of three
steps: (1) serial-to-parallel multiplexing of incoming
q*192*M bits to M antennas, (2) IEEE 802.16-2004
interleaving on each antenna, and (3) forward circular
shift of the bits on each antenna by q*cts, where cts =
“cyclic tone shift” is a parameter that must be optimized
for each data mode and MIMO configuration.
For example, the IEEE 802.16-2004 interleaver output
for BPSK modulation is shown in Figure 16 below:
1
2
3 ..
1 3 5 ..
384 362 364 …
13 14 15
25 27 29
2
25 26 27
49 51 53
26 28 30
37 38 39
73 75 77
50 52 54
:
:
:
4
6
Figure 16: IEEE 802.16-2004 bit interleaver BPSK,
SF interleaver for 2x2 MIMO on antenna 1 and on
antenna 2
In Figure 16 bits are mapped to tones column-bycolumn. Therefore bits indexed by 1, 13, 25, 37, … are
mapped to tones 1, 2, 3, 4, ….etc. Our proposed
interleavers are shown in the second two boxes of
Figure 16.
Simulation results for this interleaver with BPSK, rate
½, 192 data tones are shown in Figure 17. SUI-3
channel models without spatial correlation are used
throughout this section.
Also shown for reference is a simpler interleaver labeled
SM, which does not interleave bits across antennas.
Instead, it takes contiguous blocks of q*192 bits and
maps them to antennas, followed by IEEE 802.16-2004
interleaving on each antenna.
236
Intel Technology Journal, Volume 8, Issue 3, 2004
10
Space-frequency interleaving for 3x3 MIMO (lowest mode)
0
10
0
Space-frequency interleaving for 3x3 MIMO (highest mode)
1x1
3x3 SM
3x3 cts=1
10
10
10
-1
PER (64 byte packets)
PER (64 byte packets)
10
1x1
3x3 SM
3x3 cts=1
3x3 cts=64
3x3 svd
-2
10
-1
-3
-4
2
3
4
5
6
7
SNR (dB)
8
9
10
10
11
-2
26
27
28
29
30
31
SNR (dB)
32
33
34
35
Figure 17: SFI for 3x3 MIMO (lowest mode)
Figure 18: SFI for 3x3 MIMO at highest mode
Our interleaver provides gains of 2 to 4 dB over this
simple interleaver because it has been designed to
extract maximal space-frequency diversity.
MIMO Advanced Receivers: Iterative Decoding
At the highest IEEE 802.16-2004 mode with 64-QAM,
rate ¾ coding, gains with the interleaver are not as high,
but still significant at 1 to 2 dB over SM, as shown in
Figure 18.
Figure 18 shows two values of cts: cts=1 and cts=64.
Performance of the MMSE receiver is sensitive to the
choice of cts, although cts=1 works well for most
modes, channel conditions, and MIMO architectures.
Performance of the ML receiver is not sensitive to the
choice of cts (not shown here).
This suggests that the MMSE receiver induces
correlation across space-frequency blocks. The MMSE
induces correlation across antennas because of crosstalk, and the channel induces correlation across tones
because of limited delay spread. A combination ends up
correlating adjacent tones on all antennas. The proposed
interleaver places bits on uncorrelated tones and
antennas, thereby improving performance with the
MMSE receiver.
Figure 18 also shows performance with an SVD
receiver, which requires channel feedback to the
transmitter in order to diagonalize the channel matrix.
Note in Figures 17 and 18 that the 3x3 architectures fall
short of 1x1 by 3 to 5 dB. Therefore these MMSEMIMO architectures do not maintain range at the higher
throughputs.
Advanced receivers are required to improve range at
high rates, and they are the subject of the next subsection.
Multiple-Antenna Technology in WiMAX Systems
A non-iterative receiver similar to that used in the
previous sub-section is shown in Figure 19.
d̂ (i )
FEC
Decoder
Lb (i )
Unpuncturer
Lb1 (i )
SpaceFrequency
De-interleaver
Lb2 (i )
n1 (i )
QAM
to Bit
QAM
to Bit
r1 (i )
sˆ1 (i )
sˆ2 (i )
Spatial
Demapper
x1 (i )
x2 (i )
r2 (i )
n2 (i )
Figure 19: Illustration of non-iterative receiver
The spatial demapper above decouples the data streams
mixed by the channel matrix over the air. The MAP
demapper has the best performance and the highest
complexity, while linear demappers such as MMSE and
ZF have low complexity but poor performance
compared to MAP. Recently, techniques such as sphere
decoding have been proposed to reduce the complexity
of MAP receivers.
After the spatial streams are separated, the “QAM to
bit” functional block converts the noisy QAM symbols
into Log Likelihood Ratios (LLR) for each punctured,
coded bit. For the non-iterative receiver, these LLRs are
eventually sent to the FEC decoder and bit decisions are
made.
For the iterative receiver, there are many more steps
before bit decisions are made. Figure 20 shows an
iterative receiver based on the turbo principle [8]. The
channel matrix H is treated as a rate one linear block
code, which is concatenated with the convolutional and
Reed-Solomon codes. Iterations are conducted between
the spatial demapper and the FEC decoder by passing
extrinsic information (i.e., LLRs) back and forth.
237
Intel Technology Journal, Volume 8, Issue 3, 2004
802.16d, 1x1, 2x2, 2x3, SUI 3
25
SpaceFrequency
Interleaver
Puncturer
lb (i )
-
lb1 (i ) -
+
Lb (i )
BCJR FEC
Decoder
r1 (i )
x1 (i )
20
and
and
and
and
1x2,
1x2,
1x2,
1x3,
MMSE
MAP
SVD
MMSE
+
SpaceFrequency
De-interleaver
Unpuncturer
lb2 (i )
MAP Spatial
Demapper
-
x2 (i )
r2 (i )
+
n2 (i )
Figure 20: Illustration of an iterative receiver
Performance of this iterative receiver is shown in Figure
21 for 2x2 and 2x3 MIMO architectures. All seven rate
modes specified in IEEE 802.16-2004 are used to
generate this throughput versus SNR curve. For each
architecture, all SIMO subsets such as 1x2 and 1x3 are
allowed in the set of possible modulations. For each
SNR and each channel realization, all possible
combinations of data rate and antenna subsets are run to
compute throughput and only the maximum throughput
is reported. The maximum throughputs of all channel
realizations for that SNR are then averaged and the
mean throughput is plotted. The number of iterations for
2x2 MAP curve is 4. The SVD curve includes 2x2 with
spatial mode puncturing and 1x2 Maximal Ratio
Combining (MRC).
We observe the following:
1. 2x2 MAP buys 3-5 dB gain over 2x2 MMSE at
higher throughputs.
2. 2x2 SVD buys 2-3 dB over 2x2 MMSE at higher
throughputs.
3. 2x3 MMSE buys 5-7 dB over 2x2 MMSE at higher
throughputs.
Therefore we buy the most gain by adding an extra
receive chain. This hardware cost can be transferred to
baseband complexity by using the MAP [9] and BCJR
[10] iterative algorithms instead of MMSE, by taking a
1 dB performance hit. The complexity of the MAP
(
K
)
, where M is the size of the
spatial demapper is O M
QAM symbol and K is the number of data streams. This
can be rather large for higher order QAMs. We are
looking at methods to reduce the complexity of
advanced receivers.
Throughput (Mbps)
d̂ (i )
n1 (i )
1x1
2x2
2x2
2x2
2x3
15
10
5
0
-5
0
5
10
15
SNR = E s /N0 (dB)
20
25
30
Figure 21: Advanced receivers for 2x2 and 2x3
CONCLUSION
We have shown that multiple-antenna techniques can
greatly enhance the performance of wireless
transmission systems. Systems are currently trending
towards using multiple antennas at the BS and future
systems may evolve to multiple antenna systems at the
SS. We have demonstrated that Alamouti reception,
circular delay diversity, and selection diversity are
simple schemes that can increase performance greatly.
More advanced MIMO techniques can increase
performance well beyond the current limits of data rate
and reach.
ACKNOWLEDGEMENTS
The authors thank Hakim Mesiwala, Wendy Wong,
Sigang Qiu, Tony Liu, and Bo Xia for simulation and
channel modeling support.
REFERENCES
[1] H. Heiskala and J. Terry, “OFDM Wireless LANs: A
Theoretical and Practical Guide,” SAMS, 2002.
[2] S. Alamouti, “A Simple Transmit Diversity
Technique for Wireless Communications,” IEEE
Journal on Select Areas in Communications, Vol.
16, No. 8, pp. 1451-1458, October 1998.
[3] R. Monzingo and T. Miller, Introduction to Adaptive
Arrays, Scitech Publishing, Raleigh, NC, 2004.
[4] M. Simon, S. Hinedi, and W. Lindsey, Digital
Communication Techniques Signal Design and
Detection, Prentice Hall, Englewood Cliffs, NJ,
1995.
[5] J. Cioffi, “Course Reader for EE379A,” Stanford
University, 2002.
Multiple-Antenna Technology in WiMAX Systems
238
Intel Technology Journal, Volume 8, Issue 3, 2004
[6] G. J. Foschini, et al., “On Limits of Wireless
Communications in a Fading Environment Using
Multiple Antennas,” Wireless Personal
Communications, vol. 6, no. 3, pp. 311-335, March
1998.
processing for cognitive networks. Prior to Intel, she has
held positions at Iospan Wireless, Hughes Research
Laboratories, and Bell Laboratories. She holds a Ph.D.
from Stanford University and a B.S and M.S from MIT.
Her e-mail is sumeet.sandhu at intel.com.
[7] G. G. Raleigh and J. M. Cioffi, “Spatio-temporal
Coding for Wireless Systems,” IEEE Trans.
Communication, Vol. 4, No. 3, pp. 357-366, 1996.
Qinghua Li is a researcher in Intel’s Corporate
Technology Group. He is currently developing high
throughput techniques for Intel’s WLAN products and
IEEE 802.11n standard. Before he joined Intel in 2001,
he worked for Ericsson and Nokia for short periods. His
research lies in the hot areas of wireless
communications including MIMO, SDMA, UWB,
MAC, indoor wireless channel modelling, CDMA, FEC
coding, multiuser detection, and interference mitigation.
He received B.E., M.E., and Ph.D. degrees from South
China University of Technology, Tsinghua University,
and Texas A&M University, respectively in 1992, 1995,
and 2001, all in Electrical Engineering. His e-mail is
qinghua.li at intel.com.
[8] J. Hagenauer, “The turbo principle–tutorial
introduction and state of the art,” in Proceedings
International Symposium on Turbo Codes & Related
Topics, Brest, France, pp. 1-11, Sept. 1997.
[9] J. G. Proakis, Digital Communications, McGraw
Hill, 4th Ed., Aug. 2000.
[10] L. R. Bahl, et al., “Optimal decoding of linear
codes for minimizing symbol error rate,” IEEE
Transactions on Information Theory, pp. 284–287,
March 1974.
[11] I. E. Telatar, “Capacity of multi-antenna Gaussian
channels,” AT&T Bell Labs Tech. Memo., 1995.
[12] “Space-frequency interleaving for MIMOOFDM,” in IEEE TG802.11n, S. Sandhu, December
2003.
[13] E. Larsson and Petre Stoica, Space Time Block
Coding for Wireless Communications, Cambridge
University Press, Cambridge, UK, 2003.
[14] A. Paulraj, R. Nabar, and D. Gore, Introduction to
Space-Time Wireless Communications, Cambridge
University Press, Cambridge, UK, May 2003.
AUTHORS’ BIOGRAPHIES
Atul Salvekar is a member of the technical staff for the
Broadband Products Group. His last assignment was
designing algorithms for the IEEE 802.16-2004 modem.
Atul’s primary interest is in signal processing and
communications. He is also an avid tennis player and
loves playing the piano. Atul received his B.S. degree in
Electrical Engineering from Caltech and his M.S. and
Ph.D. degrees in Electrical Engineering from Stanford
University in 1996, 1998, and 2002, respectively. He
also has an M.S. degree in statistics from Stanford
University. His e-mail is atul.a.salvekar at intel.com.
Minh-Anh Vuong is a senior engineer in the
Broadband Wireless Division. He is working on
multiple projects. He has worked on algorithms,
firmware, and system modelling. Minh-Anh received his
B.S. degree in Electrical Engineering from Grenoble,
France and his M.S. degree in Electrical Engineering
from Arizona State University. His e-mail is minhanh.q.vuong at intel.com.
Xiaoshu Qian currently manages the system group in
the Broadband Wireless Division at Intel Corporation to
help develop the next-generation broadband wireless
communication chips. In the past, he has worked
primarily in the areas of algorithm development, DSP
architecture, and logic design for multimedia and
communication chips. He received a Ph.D. degree in
Electrical Engineering, an M.S. degree in Computer
Science, and an M.S. degree in Mathematics, all from
the University of Rhode Island. He also holds a B.S.
degree in Physics from Zhejiang University in China.
His e-mail is xiaoshu.qian at intel.com.
Copyright © Intel Corporation 2004. This publication
was downloaded from http://developer.intel.com/.
Legal notices at
http://www.intel.com/sites/corporate/tradmarx.htm.
Sumeet Sandhu is a senior staff researcher in the
Corporate Technology Group in Santa Clara. As CTG
MIMO lead for the 802.11n standards effort, she
developed a number of algorithms and IP which are part
of the Intel 802.11n proposal. Her primary interests are
space-time coding, signal processing, and FEC for
point-to-point wireless systems and distributed
Multiple-Antenna Technology in WiMAX Systems
239
Intel Technology Journal, Volume 8, Issue 3, 2004
THIS PAGE INTENTIONALLY LEFT BLANK
Multiple-Antenna Technology in WiMAX Systems
240
Fully Integrated CMOS Radios from RF to
Millimeter Wave Frequencies
Luiz M. Franca-Neto, Intel Communications Group, Intel Corporation
Roger Eline, Intel Communications Group, Intel Corporation
Bisla Balvinder, Intel Communications Group, Intel Corporation
Index words: 802.16, RF CMOS, microwave, millimeter wave, deep nwell, mixed-signal, RFIC,
analog IC, flip-chip package, passives on the package, System-on-a-Chip (SoC), System-on-aPackage (SoP), backing-off method, optimum-pump method, 60 GHz, 100 GHz circuits, Wi-Fi,
WiMAX.
ABSTRACT
This paper reviews (a) recent CMOS demonstrations of
capabilities for Radio Frequency (RF), microwave, and
millimeter wave circuits from 1 GHz to 100 GHz, (b)
advances in on-die isolation structures for integrating
radio’s delicate circuits with very noisy general-purpose
processors on the same die, and (c) entirely novel design
methods for complex RF passive networks on the
package substrate by engineering the physical design of
the package substrate (no discrete passive components
added to the package) that diminish the silicon area
requirements for multiband multiprotocol CMOS radios
and frees silicon area to host complex digital processing
and communication engines. Circuit design techniques
are discussed to cope with intrinsic CMOS challenges and
technology scaling. Building upon these developments, a
vision for CMOS technology and platform direction is
proposed.
INTRODUCTION
From 1995 to 2004, CMOS technology has proven its
Radio-Frequency (RF), microwave, and millimeter wave
capabilities by demonstrations of fully integrated key
circuit blocks from 1 GHz to 100 GHz [1-7]. Low Noise
Amplifiers (LNAs) with noise figures as low as
previously reported for compound semiconductor
technology started to be reported for fully integrated
CMOS realizations. The intrinsic higher 1/f (flicker)
noise corner in CMOS technology compared to bipolar
technologies found compensation in novel circuit-level
methods.
CMOS scaling enabled the technology to reach for higher
GHz frequencies, and the higher speeds offer other
opportunities to compensate at the circuit level for
intrinsic technology drawbacks.
Only one intrinsic technology problem appeared to be
fundamentally unsuited for technology scaling: RF
transmission power levels. As CMOS scales, lower
voltages are tolerated at the transistor terminals. Circuitlevel solutions using power-combining techniques to add
the power of parallel Power Amplifiers (PAs) in CMOS
have met with success. Power combination of parallel
PAs have being used on die [8], and in this paper, we
discuss novel power-combining circuits on the package.
These power-combining circuits on the package become
en passant the supporting structure for MIMO or general
antenna-diversity/beam-forming-based radios. This last
step means the circuits on the package support what can
be recognized as power combining on air to cope with
CMOS RF power transmission limitations.
Nevertheless, CMOS technology’s full potential would
not be realized if only standalone radios are fabricated.
Integration of delicate radio and general-purpose
processors is the next goal. The co-habitation of delicate
RF circuits and a very noisy general-purpose processor
such as a 1 GHz 55 W Pentium® 4 processor on the same
die was shown to be possible by proper circuit
techniques, special deep nwell isolation structures, and
exploitation of the digital substrate noise spectrum
structure [9]. Novel entire designs of complex RF
passive networks realized by trace engineering (no
®
Pentium is a registered trademark of Intel Corporation
or its subsidiaries in the United States and other
countries.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
241
Intel Technology Journal, Volume 8, Issue 3, 2004
discrete components added) on the package substrate
diminished the silicon area required for multiband
multiprotocol radios and freed silicon area for hosting
more
digital
circuits,
processing units,
and
communications system new features [10].
Building on these developments, at the platform level,
considering the PC motherboard, we articulate a vision
for a new CMOS computing and communication
platform. Instead of trying to integrate multiband
multiprotocol radio circuits into already densely packed
chips like a Pentium processor and its companion
chipsets, it might be more promising to re-think the PC
motherboard as a multiprocessor platform, where the
processor and chipset will make a new ecosystem with
two new chips that provide for multiband multiprotocol
radios. In other words, a platform which is capable of
supporting the variety of current standards and is also
able to support always evolving standards is proposed.
This new platform can extend its reach to encompass in a
modular fashion wireless communications from 700
MHz, over the newly vacant TV bands, all the way to 60
GHz, where 7 GHz of bandwidth enables indoor high
data rate omnidirectional wireless links and outdoor lineof-sight (LOS) high data rate backbone links.
The next sections in this paper detail the CMOS
technology scaling effects on its RF, microwave, and
millimeter wave capabilities; the new developments in
package technology and novel CMOS-compatible
devices; and further elaborate on the opportunities in the
area of platforms for CMOS.
Fortunately, the opposite has happened so far and even
though the product gmRo decreases with nanometer
scaling, the device transconductance (gm), with typical RF
device loading, still provides higher gain with CMOS
scaling to compensate for the additional noise in the
channel high field transport. Moreover, carrier transport
in the channel of 90 nm CMOS and future nodes may
experience a qualitative change in properties that leads to
less carrier velocity dispersion due to a diminishment in
the likelihood of carrier scattering events in such
extremely short channels. If this becomes a new trend it
will progressively benefit CMOS technology and will
offer unprecedented lower noise figures with scaling at
frequencies above 10 GHz. Currently, minimum noise
figure (NFmin) numbers for CMOS transistors in 0.18 µm
and 90 nm are respectively 1 dB and 0.5 dB at 5.5 GHz.
These are at par with the best numbers offered by SiGe
and other compound semiconductor technologies at these
frequencies.
In reality, CMOS transistors are becoming virtually
“noiseless” for practical purposes below 10 GHz. That
completely shifted the design and optimization
procedures for LNAs to include noise contribution from
passives. A circuit-level method was developed by one of
the authors to globally optimize the noise figure of LNAs,
taking into account noise contributions from both passive
and active devices. It became an extension of S-parameter
methods used in traditional microwave methods and was
named the “backing-off” method [6].
CMOS (EXCESS) THERMAL NOISE.
In the radio receiver front-end, the Low Noise Amplifier
(LNA) is the first key component in which CMOS
technology needed to prove its adequacy. By the
aggressive scaling of CMOS technology, there was
always the concern that the high field transport in the
channel could produce too large a carrier velocity
dispersion, and therefore microwave noise, significantly
above thermal noise. That’s because apart from not
always having consistent definitions in the literature for
excess thermal noise, a conductor or semiconductor is
only guaranteed to develop thermal noise levels in
thermal equilibrium, and will tend to develop noise levels
above this equilibrium level whenever a dc current flows
through them and more so as the electric field applied to
the transport increases [11, 12]. The concern was that the
noise level could become progressively higher with
scaling in such a way that the gain of the device could not
compensate and, in this case, scaling would start to
produce higher noise figure transistors at some point.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
242
Intel Technology Journal, Volume 8, Issue 3, 2004
Zin*
Zload
Trans 1
L1
L2
RFin+
MC1
25pH
3.0
4.0
5.0
2.0
0.6
0.8
1.0
0.4
0.2
0
0
.
-2
-1.0
-0.8
-0
.6
10.0
-3
.
-4 0
-5..0
0
-10.0
2
10.0
0.2
L2
.4
-0
Zload
(10dB gain)
0
3.
0
4. 0
5.
L1
-0.
Z*in
(for 10dB gain)
10dB
15dB
5.2dB (NF)
4.2dB (NF)
2.
0
0.
4
Total length
of Impedance
Transformer
0.8
6
0.
Back-off
1.0
Zin
17GHz
Figure 1: Backing off from active device’s NFmin:
length of impedance transformer’s transmission line
is shortened to diminish the noise contributions from
lossy passives at the expense of a small increase in the
contribution from the active devices, but still
producing a lower noise figure for the final LNA
would have been as close to L1+L2 (note “L” stands for
length rather then inductance in this discussion) as an
acceptable input mismatch would allow. However, once
low-Q passives are used, making the length of Trans1 be
shortened to L1, despite an increase in the cascode
structure’s noise, leads to smaller noise figures for the
final LNA. Moreover, the pair Zload and Zin* for 10 dB
gain, identified in Figure 1, stresses that backing off can
lead not only to a lower noise figure but can also lead to
minimal mismatch at the input port of the amplifier. In
general, the amount of back off is a function of how low
the Q of the passives is and how slowly the active
devices’ noise figure changes as their driving source
move away from their optimum (i.e., how small the active
device’s noise parameter Rn is). The disposition of
constant gain circles, constant noise circles, and stability
circles in the Smith-chart will change with transistor size,
amount of source inductive degeneracy, and frequency of
operation of the LNA. In the designs presented in this
paper, for every step in the optimization process, backing
off is always checked around every design point iteration.
Lload2
Lload1
Trans2
Trans1
RFin +
bias 2
M C1
M2
M1
bias4
MC2
M3
Mstb
Le
bias 3
AGC
Metal 1 ground-plane
Traditional microwave design approaches assume highquality passives. Thus, low noise amplifier designs
primarily seek the transistor’s NFmin and implement its
optimum source (driving) impedance, so far as this does
not compromise unacceptably the gain of the amplifier or
its input match [13]. In contrast, when passives are
implemented on a CMOS die, because of geometric
constraints (more on geometric implications for on-die
realizations later), their low Q makes such an approach
sub-optimal. In effect, low noise amplifiers are optimally
designed if backing off from the active device’s NFmin is
used. This new approach minimizes the final noise figure
of the LNA by trading off a small increase in transistor
noise for a much lower noise contribution from the lossy
passives.
Figure 1 illustrates how the backing-off approach is
applied to the definition of transmission line length of the
impedance transformer (Trans1) in the Input Matching
Network (IMN) of the amplifier. Constant gain circles
(15 dB and 10 dB gain, Zload referred) and constant noise
figure circles (Zin* referred) for a cascode structure with
inductive source degeneracy is depicted. Note that if the
design of the input matching network was done with highquality passives the length of the impedance transformer
RFout +
M4
RFin +
RFout +
RFin -
Rfout -
MpAGC
Towards other half
of differential circuit
Virtual-ground axis
(differential operation)
metal6
h
metal1
Microstrip-on-die
Figure 2: Schematic and corresponding layout for a
17 GHz LNA in 0.18 µm CMOS
Even though the procedure was illustrated for 17 GHz
and 24 GHz LNAs, and on-die microstrip segments were
used for the passives, the method is readily applicable to
both lower and higher frequencies, with the only visual
effect of using lumped passive components (spiral
inductors) at lower GHz frequencies and distributed
passive components (microstrips) at higher frequencies.
Figure 2 shows the simplified schematics and the
corresponding layout for a 17 GHz LNA in 0.18 µm
CMOS.
Figure 3 shows die photos of both 17 GHz and 24 GHz
LNAs in 0.18 µm. These designs weren’t the most
compact designs possible since they also aimed at
proving abrupt curves make for only minimal affect in
microstrip performance on-die, and that on-die
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
243
Intel Technology Journal, Volume 8, Issue 3, 2004
transmission lines can be used for robust design of
inductance as small as 25 pH. These 17 GHz and 24 GHz
LNAs delivered final noise figures below 6 dB (world
record in 0.18 µm). LNA designs at 2.4 GHz and 5.2 GHz
using backing-off method delivered under 3 dB noise
figures even with input matching network realized on-die,
also a world record for 0.18 µm. These results qualify
CMOS for 802.16 applications with a healthy margin.
[14].
LNA 17GHz die photo
LNA 24GHz die photo
Figure 3: Die photos of 17 GHz and 24 GHz LNAs in
0.18 µm CMOS. Noise figures below 6 dB (world
record) at these frequencies include input matching
network (IMN) on die. At 2.4 GHz and 5.2 GHz noise
figures are below 3 dB, IMN included, for the same
CMOS technology.
Both 17 GHz and 24 GHz LNA designs were successful
at first trial. They were designed using S-parameter
measurements from laid out CMOS transistors on wafer.
No CMOS modeling was used, and device sizing was still
retained as a designer’s degree of freedom. Larger or
smaller devices could have their S-parameter calculated
on the computer straightforwardly since larger devices
are just smaller ones in parallel. All the microstrips on the
die were electromagnetic field solved, and their Sparameter behavior was determined as well. Again, no
modeling of these passives to their constitutive
components was necessary. All this indicates CMOS RF
and microwave designs will benefit from seamlessly
merging methods and techniques from both VLSI and
microwave domains, and this is discussed later in this
paper.
CMOS 1/F (FLICKER) NOISE
Due to the very nature of carrier transport in CMOS
transistors taking place at the interface between SiO2 and
Si, the 1/f corner frequency in CMOS transistors is much
higher than the corner frequency for bipolar transistors.
At the intrinsic device level, therefore, CMOS suffers
from a physically-based drawback. And, the introduction
of new high-κ dielectric material in the gate of future
CMOS technology nodes will tend to increase the 1/f
noise levels of CMOS transistors.
RF CMOS designers have worked successfully through
mitigation procedures. First, whenever fast enough,
PMOS transistors are used instead of NMOS transistors
as the device for oscillators and Voltage Controlled
Oscillators (VCOs). It was already noted that the CMOS
drawback of typically 10 dB in 1/f noise in comparison to
bipolar devices could be compensated for by (a not
always desirable) 4X increase in power dissipation in the
final oscillator and VCO designs in CMOS. That
stemmed from the experiment of paralleling four identical
coupled oscillators to produce a single oscillator signal.
Since the individual oscillators’ signals add in amplitude,
and the uncorrelated noise from the identical oscillators
add in power, paralleling oscillators yield lower phase
noise oscillations for the final assembly [15].
More importantly however, new understanding of the
manifestation of the upconversion of 1/f noise as close in
phase-noise in oscillators and VCOs opened the
perspective of more sophisticated circuit-level approaches
to low noise oscillators and VCO designs in CMOS [1622]. First of all, contrary to the assumption of many
designers, an assumption encouraged by Leeson’s
formula [15, 22], the 1/f corner frequency of the CMOS
transistors will not be the first corner frequency of the
oscillator phase noise spectrum [18-20]. Actually,
experimental results have frequently indicated the
incorrectness of this assumption [22]. In reality, circuitlevel considerations of topological and current drive
symmetry can push the oscillator phase noise first corner
to within kHz frequencies from the carrier’s frequency,
thus yielding very low noise oscillations, even in CMOS
technology where the 1/f corner can be in the hundreds of
MHz frequencies. This new appreciation of phase noise
readily led to a demonstration of unprecedented lowphase noise oscillators and VCO designs in CMOS,
without the need to increase unduly the power
consumption.
On top of that, VCOs are used in PLL- or DLL-based
synthesizers in radio designs. This allows for another
level of circuit design techniques to be used for further
diminishing phase noise. The idea is to make the loop
force the internal VCO to follow the much lower-phase
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
244
Intel Technology Journal, Volume 8, Issue 3, 2004
noise oscillations of an external reference source. The
larger the loop gain and the loop bandwidth, the lower the
phase noise for the synthesizer output signal. The loop
bandwidth of PLL and DLL designs is typically 10% of
the lowest frequency into the phase detector, which is
normally the reference source signal. In integer-N
synthesizer architecture, the source frequency (or a
suitable division of that reference frequency) is defined
by the finesse of the separation between the wireless
channels, which might conflict with the intention of
having a higher frequency for the source reference signal
(for larger loop bandwidths). A solution to this conflict is
found in the ever more popular Fractional-N synthesizer
architecture, where the source reference frequency is
allowed to be much higher and the finesse of channel
separation is attended by periodically or randomly
(sigma-delta synthesizers) alternating the division used in
the PLL or DLL loop [15,22].
Other non-linear techniques have been proposed for lownoise synthesizers in CMOS, with ever larger loop
bandwidth [16], and all these new techniques only help
the case for CMOS to prove its intrinsic higher 1/f noise
is no impediment to the use of this technology in radio
frequency designs and systems.
Another benefit comes straight from CMOS scaling,
however. Higher Q passive components can be achieved
at higher GHz frequencies, since for the same geometry
available for passives, Q increases with the square root of
the frequency [23, 24]. Once CMOS scaling enables
oscillator and VCO designs at higher frequencies, lowerphase noise operation is achieved by the use of these
higher Q components, since thermal noise from passives
also affects phase noise as well as the transistor’s 1/f
noise. Finally, dividing the output of these high-frequency
oscillators, VCOs, and PLLs to get the actually used final
lower frequency will also allow for another decrease in
phase noise. And, as a beneficial side effect, starting with
higher frequency oscillators and VCOs may result in
significant savings in foot print in the silicon die, since
passives at higher frequencies are smaller. Therefore,
having higher frequency capabilities enabled from CMOS
scaling does lead to improvements in designs, even for
radios operating at much lower GHz frequencies than the
frequency limits for a given CMOS technology.
CMOS RF STABILITY, MODELS AND
METHODOLOGY
90
nm
40
50
40
load
stability
circles
(lsc)
source
stability
circles
(ssc)
CMOS
0.
6
1.
0
0.
8
30
2.
0
30 0.4
20
3.0
4.0
20
5.0
0.2
10 10.0
10
0
0.
2
0.
4
0.
6
0.
8
1.
0
2.
0
10
.0
3. 4. 5.
0
0 0
-10.0
port 2
-0.2
-5.0
-4.0
port 1
-0.4
M1
-3.0
2.
0.
0.
1.
Figure 4: 90 nm CMOS stability circles:
unconditional stability only after 40 GHz
As can be seen in Figure 4, a typical 90 nm CMOS
transistor is only unconditionally stable above 40 GHz.
As CMOS scales, the unconditionally stability region will
only start at higher and higher frequencies. This doesn’t
necessarily preclude future RF designs at 2.4 GHz and
lower frequencies necessarily, but it does require RF
designers to pay close attention to the source and load
impedances they use in their circuit designs when they
move to use more advanced CMOS technology nodes.
Not being careful will lead to oscillatory behavior in
amplifiers and failure in other active circuits. As
multiband radios spanning from 700 MHz to 60 GHz may
be fabricated in the same CMOS process, it is very
unlikely that CMOS modeling, using detailed network
representations of transistors traditionally used in VLSI
design, will accurately represent the devices behavior
across such a large span of frequencies. Since accurate
RF/microwave behavior and noise performance
parameters are required, merging VLSI and microwave
methods holds more promise. CMOS models plus full
disclosure of S-parameter/Noise-parameter data and other
relevant experimental results, will mark the new
methodology to be followed by CMOS foundries and the
CMOS-based industry.
This change is more pressing still when, counting on the
expected CMOS scaling, some companies release
transistor models with “forward-looking” adjustments
that do not agree with current silicon behavior. These
companies borrow from traditional VLSI methods that
expect the performance of silicon transistors to always
improve with time. Thus, they release CMOS models they
think will be correct some time in the future when the
designers eventually tape out their designs. That is not a
methodology suited for RF and microwave design, which
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
245
Intel Technology Journal, Volume 8, Issue 3, 2004
depend on much more accurate representations of the
devices used. A simple compromising change in the
methods is to make these companies fully disclose the
current silicon data, S-parameter, and Noise-parameter
companion to any “forward-looking” CMOS model
release. This way even if present silicon data and
futuristic models do not agree, designers have full
knowledge of this “gap” and are able to assess how
significantly this gap affects their designs.
gain
(dB) Maximum unilateral power gain, Gu,max
45
40
35
30
25
fmax ~ 110GHz
20
Designers on their side should be equally proficient at
designing from CMOS models, experimental S-parameter
and Noise-parameter data. These are complementary sets
of information. Each one is useful for different aspects of
the design and frequency of operation.
RF passive design also experiments with similar
methodological changes. Electromagnetic field solvers
have become commonly present amongst the set of CAD
tools used in both silicon and package RF design to the
extent of almost dispensing with detailed lump element
models of silicon and package interconnects.
In the next section, designs at 64 GHz and 100 GHz are
not only based on S-parameter measurements (not CMOS
modeling), but they also extrapolate these measurements
to both larger device sizes and much higher frequencies
prior to design. That accurate representation of
transistors’ behavior led to success at first try, despite it
being a design of completely uncharted and
unprecedented high millimeter wave frequencies for
CMOS.
Millimeter Wave Capabilities: 64 GHz and 100
GHz VCOs in CMOS
In order to demonstrate CMOS technology capabilities
well above 10 GHz, and establish the technology
potential for the full 802.16 standard, voltage-controlled
oscillators were designed and demonstrated for
operations at 64 GHz and 100 GHz. These were
frequencies close to CMOS transistors’ fmax. It was
therefore not only a CMOS technology intrinsic
capabilities demonstration, but also a circuit-level design
advance in concept and methods that renders itself very
well in CMOS. The transistors used were thicker gate, nostrained CMOS that exhibited fmax ~ 110 GHz (Figure 5).
The unconditional stability above 40 GHz in 90 nm
CMOS technology (see Figure 4) is exploited in these
novel designs. Since the device is unconditionally stable
above that frequency, it allows the use of simultaneous
complex conjugate matching at input and output ports of
every transistor in the VCO. This matching pumps energy
from the active device to the passive network optimally,
optimum pumping, which is essential at frequencies close
to fmax, where transistors offer little gain.
15
10
5
0
-5
1
10
100
1000
freq (GHz)
Figure 5: Maximum unilateral power gain and fmax of
a thick gate non-strained 90 nm CMOS technology
used in 64 GHz and 100 GHz VCO designs
In a typical negative-Gm LC oscillator/VCO (Figure 6a),
it is required that the negative resistance, Rin, appearing at
terminals “a” and “b” (Figure 6b) be smaller than the
parallel resistance of the tank network [15]. No
consideration is given to an optimum value for Rin.
Nevertheless, optimum pumping is accomplished by
considering the generalization of the LC oscillator
network and its equivalent unraveled version shown in
Figure 6b. A signal entering transistor M1’s gate (node
“a”), appears at M1’s drain and travels through the
general passive network to reach the gate of transistor
M2. This signal enters the gate of M2, appears at its drain,
travels through the general passive network and reappears back at point “a.” After this whole cycle, this
signal will have experienced the same change in phase
and amplitude as if it had traveled along the equivalent
unraveled infinite network shown in Figure 6b from its
node “a” to its node “a*.”
Every single transistor in the unraveled infinite network
can now be thought of as part of a chain of amplifiers.
Since the transistors are unconditionally stable at
frequencies close to fmax, the required ZG* and ZL for
simultaneous conjugate matching is promptly calculated
from their reflection coefficients (Figure 6) [13,23,24].
Hence the general network (Figure 6c) transforms the
impedance at the gate of each transistor, ZG, into the
required load impedance, ZL, at the drain of the transistor
of the preceding stage. In a lossless passive network, this
impedance transformation preserves the coefficient of
mismatching, MS, along the unraveled chain (Figure 6c)
[13], which makes this oscillator topology a physically
realizable one.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
246
Intel Technology Journal, Volume 8, Issue 3, 2004
ZLOAD
Case1:
|ΓG| = |ΓL|
ΓL
L2
Rin = -2/gm
L1
a
L1
b
ΓG
delay at the cross of
real axis
Z
Z*
L2
Z GATE
Z LOAD
C1
Z* LOAD
C2
L2
DELAY
L1
OUT
buffer
Figure 7: Strategic delay element introduction: “L1”
and “L2” are lengths of transmission lines
(a)
general net
…
a
M2
general net
general net
…
a
M1
a*
node “ a* ”
node “ a ”
o
a to a* needs gain>1 and integer number of 360 phase shift
(b)
MS
MS
MS
MS
impedance
converter
…
ΓS
Γ L Γ*L
MS
impedance
converter
Γ*S Γ S
(c)
In this work, no commercial CMOS model was used. 90
nm logic CMOS transistors were laid out and
characterized by S-parameter measurements up to 50
GHz. The transistor S-parameters were extrapolated to 64
GHz and 100 GHz. The distributed passive networks
were realized using microstrip-on-die, with ground plane
in metal-1 and traces in metal-7 layers. These passive
networks were Electromagnetic Field solved using a
commercial program [25]. Ground plane in metal-1
isolated the passive networks from silicon substrate
losses.
…
ΓL
Figure 6: From negative-gm to optimum pumping
Depending on the transistor technology, the number of
stages required for a multiple-of-360o phase shift in the
signal may be awkwardly high. Figure 7 shows how delay
lines are added in one of the three possible general cases
for the optimum pair Γs and ΓL [7]. The impedance
transformation along these distributed networks crosses
the horizontal-axis (real impedance axis) of the Smithchart along one of its transmission lines. At this cross, a
lossless transmission line segment of characteristic
impedance defined by the point of cross can be added to
the VCO’s passive network without disturbing the
optimum-pumping impedance transformation. The length
(delay) added depends on the number of stages desired
for the final VCO. It is important to note the optimumpumping method exploits the unconditional stability of
the transistor whereas the standard microwave approach
exploits the device instability for oscillator design
[13,23,24].
Figure 8 shows the photo die with a description of the
components of both 64 GHz and 100 GHz VCOs. Signals
were taped from the VCO’s core at its lowest impedance
(lowest swing) with a high-impedance tap for minimum
disturbance of oscillations.
¼ λ t.lines
GND
Vctrl
VCC
Vctrl
VCO
Vctrl
GND
GND
OUT
64 GHz VCO
GND
delay
GND
¼ λ t.lines
VCC
Vctrl
OUT
bias
GND & tap
100 GHz VCO
Figure 8: 64 GHz and 100 GHz VCOs: singletransistor core die photo
The ¼-wavelength-transmission-line tap from the VCO
core to the transistor buffer further diminished the signal
to be measured. The pads are part of the output network
of the buffer; microstrip stubs were added to properly
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
247
Intel Technology Journal, Volume 8, Issue 3, 2004
tune the pad impedance to maximum buffer gain (Figure
8).
Quadracture outputs (I & Q)
delicate RF circuits typically sensitive to at least -76
dBm signals from the antenna, we started by measuring
how noisy the substrate of a Pentium 4 is.
VCC
core
transistors
BLOCK 1
BLOCK 2
delay
¼ λ t.lines
a
b
µProcessor
VCO’s
core
100 GHz VCO
Figure 9: 100 GHz VCO: 4-transistor-core die photo
The 64 GHz and 100 GHz VCO signals were measured to
be centered at 63.6 GHz and 103.9 GHz. This was
calculated by simulating that the measured signal for both
64 GHz and 100 GHz meant a 0.4 Vp-p swing at the
VCOs’ cores at their largest swing point. Both VCOs
used a 1.0 V power supply and drew 20 mA (64 GHz)
and 30 mA (100 GHz) of current. Both VCOs were
completely functional from -50oC to 110oC. Center
frequency changed approximately 5 GHz (100 GHz) and
3 GHz (64 GHz) in this temperature range, because of the
relatively small temperature dependence of the phase
shift contribution from the passive network in the VCO
core. Consistently, the gains for both VCOs were in the
range of 2 GHz/V, either through body bias or supply
voltage control. These were successful designs at first try
that firmly established CMOS capabilities well into
millimeter wave frequencies.
Figure 9 shows a 100 GHz 4-transistor core oscillator that
was designed by adjusting the delay elements. In this
topology quadracture output signals are produced.
RF AND DIGITAL PROCESSOR IN THE
SAME DIE
Once CMOS technology capabilities for RF applications
is established even to extreme 100 GHz frequencies, the
next step is to go beyond standalone radio design.
Communications and computing have synergies that can
be exploited in RF and digital processor integrations in
the same die.
In order to demonstrate that such an integration is
possible even in the extreme case of a very noisy digital
processor with clock frequencies in the GHz range and
Figure 10: Redundant logic delay chains: unused
transistors engineered into noise sensors
Intel processors are taped out with extra logic delay
chains for pre-product investigations (Figure 10). These
extra transistors are left in the commercial
microprocessors without doing any work. These
redundant transistors were engineered as substrate noise
sensors by Focused Ion Beam (FIB) work from the back
of the die. These transistors were unconnected from the
rest of the microprocessor circuits and their terminals
were brought externally onto the back of the wafer
(Figure 11). Substrate activity (noise), which modulates
the body bias of these transistors, is displayed by the
spectrum analyzer (Figure 12). Figure 11 shows the
layout as seen from the back of the die, and it shows the
vias and wires for connecting the source, drain, and gate
of a test transistor. 1.2 mVrms noise measured at the
drain of a 5 µm-wide noise sensor located at the center of
the die translates (by the noise sensor transfer function)
to 100 mVrms noise on the substrate. These
measurements corresponded to 15 W power dissipation
produced by the excitation of the 1 GHz clock grid as can
be seen by the noise spectrum developed. Assuming
substrate noise power is directly related to the
microprocessor (dynamic) power dissipation [26], the
same microprocessor dissipates 55 W and thus produces
190 mVrms substrate noise at full operation. Because of a
typical activity of 10% (for the logic gates), this
additional substrate noise on an actual application of this
microprocessor, is concentrated from dc to 150 MHz.
The fundamental insight guiding this research is that
high-performance microprocessors, with clocks at GHz
frequencies, develop substrate noise with a spectrum
structure that can be exploited to place RF narrow-band
signals in valleys of low-substrate noise levels in the
frequency spectrum. This can be achieved by placing and
retrieving feeble bandwidth-limited RF information
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
248
Intel Technology Journal, Volume 8, Issue 3, 2004
signals between the harmonics of the clock, and away
from the intense substrate noise’s components generated
by the random logic gate activity (Figure 13). A
commercial 1.5 V 55 watts 1 GHz 104-million-transistor
(Pentium 4) digital microprocessor and a 50 MHzbandwidth-76 dBm-sensitivity wireless receiver with a
carrier frequency at 2.4 GHz and 5.2 GHz FCC’s ISM
bands is analyzed for possible integration on the same
die. For simplicity, but without loss of generality, a
direct-conversion architecture is assumed for the RF
receiver. A band-selective LNA amplifies the feeble RF
signals from the antenna to bring their level sufficiently
above the (attenuated by isolation) substrate noise upon
downconversion to baseband (Figure 13). Isolation
requirements for Signal-to-substrate Noise-Ratio (SsNR)
higher than 20 dB (BER<10-9) are derived, and an
isolation scheme with only minimal technology addition
(deep nwell structures) is presented. The measured high
frequency performance of 140 nm logic CMOS (fmax and
ft at 100 GHz and 60 GHz, respectively) is not
significantly affected by placing them in the required
deep nwell structure.
Back of wafer FIB work
source
Test vector
gate
drain
Vd
bias-t
Vg
bias-t
substrate
Spectrum Analyzer
-50
-55
-60
-65
-70
-75
-80
-85
-90
-95
-1000
1
2
3
4
5
6
7
GHz
Figure 12: Pentium 4 substrate noise spectrum
The isolation for wireless receiver integration needs at the
very least to guarantee that the substrate noise (both inband and out-of-band frequency components) does not
disturb the bias voltages used in the wireless receiver
front-end. Gate overdrive in MOS transistors used in the
RF front-end, are typically around 200 mV, and a 10%
disturbance in these voltages means isolation should
provide 20 dB (|20log(20 mV/190 mV)|) across the entire
relevant spectrum. There are, however, much more strict
requirements for isolation, as will be seen next. The
typical 10% activity factor on the logic gates makes the
random logic activity develop only a 0.0025 fraction of its
substrate noise power over the 50 MHz bandwidth on the
2.4 GHz ISM band and only a 0.0014 fraction over a
similar 50 MHz bandwidth on the 5.2 GHz ISM band,
(Table 1) [9].
Figure 11: Focused ion beam (FIB) work: transistornoise-sensor is accessible from the back of the
packaged processor. The processor is excited by
standard test vectors.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
249
Intel Technology Journal, Volume 8, Issue 3, 2004
Power
random
logic
digital clock
Since the 50 µV (-76 dBm) RF signal from the antenna
needs to be amplified enough before being
downconverted to baseband (where it will face the lowfrequency components of the substrate noise produced by
the random activity of the logic gates), combinations of
LNA’s gain and isolation levels are presented with the
final SsNR achieved in Table 2 [9].
harmonics
f
Power
X
RF signal
Y
baseband
LNA
Local oscillator
f
RF signal
Figure 13: Exploiting substrate noise spectrum
structure: feeble RF signals are placed at valleys of
low noise levels
Table 1: Substrate noise power
Total
substrate
noise at die
center
Substrate
noise from
1GHz clock
grid
Substrate
noise from
digital logic
(10%
activity
relative to
clock
frequency)
2.4GHz inband
substrate
noise from
logic
(50MHz
bandwidth)
5.2GHz inband
substrate
noise from
logic
(50MHz
bandwidth)
190.0mVrms
109.7mVrms
155.1mVrms
7.76mVrms
5.80mVrms
As an enabling requirement, SsNR>20 dB aims for a
healthy margin for achieving system Bit Error Rate
(BER) better than 10-9. Hence, LNA’s gain of 20 dB and
its isolation level of 70 dB are the borderline enabling
values for integration. Note that the out-of-band
components of the substrate noise are not amplified by
the band-selective LNAs, and any mixing of out-of-band
substrate noise and the RF signal will be attenuated by the
conversion loss of the operating non-linearity [9]. Note
also that the 50 MHz bandwidth at RF frequencies
becomes 25 MHz bandwidth at the baseband, diminishing
the amount of substrate noise captured at the baseband.
As mentioned, the final 70 dB isolation requirement for
integration is much stricter than the requirements for
merely not disturbing the bias voltages in the RF frontend. As will be seen next, this total of 70 dB isolation will
be partitioned into on-die isolation and layout- and
circuit-level isolation. We achieve the 50 dB of on-die
isolation by use of a deep nwell, and therefore we
guarantee that the bias voltages of the RF front-end are
not disturbed by the substrate noise from the digital
circuits.
70 dB substrate noise isolation between integrated
subsystems is realized by adding isolations from on-die
implanted deep nwell structures (>50 dB) to isolations
from layout and fully differential circuit topology (20
dB). An on-die isolation higher than 50 dB is realized by
implanted double deep nwell structures using two circuitlevel methods: substrate noise trapping and floating deep
nwell, shown in Figure 14.
Table 2: Isolation and LNA gain tradeoff
Goal:
SsNR>20dB after down-conversion inside 25MHz of base-band bandwidth
(direct conversion RF receiver assumed for both 2.4GHz and 5.2GHz ISM
bands)
LNA
gain
isolati
on
2.4GHz
in-band
SNR
(from
logic
activity
at LNA
input
after
isolation)*
5.2GHz
in-band
SNR
(from
logic
activi-ty
at LNA
input
after
isolation)*
Baseband
substrate
noise
from
logic
(25MHz
bandwidth
after
isolation)†
2.4 GHz
transceiver
25MHz
Base-band
SsNR,
(after downconversion
to baseband)‡
5.2GHz
transceiver
25MHz
Base-band
SsNR (after
downconversion
to baseband)‡
10dB
80dB
36dB
39dB
2.5µVrms
30dB
31dB
20dB
70dB
26dB
29dB
7.8µVrms
24dB
26dB
30dB
60dB
16dB
19dB
25µVrms
15dB
18dB
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
250
Intel Technology Journal, Volume 8, Issue 3, 2004
Vcc (analog)
Vss (digital)
Vss (analog)
die. Moreover, the signaling between the two subsections
is differential. This differential signaling leaves the
analog and digital subsystems to “fluctuate” relative to
each other.
Vss (digital)
Vcc (digital)
RF circuit
digital circuit
section
section
S21 ( dB)
-60
p-
p-
n
p-
n
-80
p-
Floating deep
nwell
Double deep
Common
-120
trapping nwell
nwell isolation
-140
Figure 14: Deep nwell hook ups and biasing
The deep nwell covering the digital circuit section
attenuates the substrate noise passing through the deep
nwell’s walls towards the common substrate (substrate
noise trapping). Once into the common substrate, the
attenuated substrate noise will proceed towards the deep
nwell protecting the RF circuit section, making that whole
deep nwell change its electric potential uniformly
(floating deep nwell). These two mechanisms are more
effective the more conductive is the deep nwell implant in
relation to the substrate, and the smaller the area of the
floating deep nwell.
low impedance
Differential signal
same-die housing
differential loop interface
Analog
+
power choke
choke
Analog
Digital
Sub
Sub
+ Digital
choke power
R
50
-100
substrate
Substrate noise
choke
bare
choke
high impedance common-mode loop
Figure 15: Differential signaling between analog and
digital subsystems
In order to realize these two effects, the power supply for
the subcircuits sections and signaling between the two
sections follows the description in Figure 14 and 15. Note
both Vcc and Vss power supply connections for both
subcircuits are kept independent and never connected on-
-160
2.4
double dnwell
4.4
6.4
8.4
Frequency ( GHz)
10
Figure 16: Double deep nwell isolation (aggressor and
victim surrounded by nwell) vs. no deep nwell
isolation
The differential interface defines a preferred path for
signals from one subsystem to another. Substrate noise
from the digital subsystem will travel towards the analog
subsystem through the common mode loop. Since this
common mode loop has several choke inductors along its
path, it will present a high impedance for currents (Figure
15) and will strongly attenuate the substrate noise sensed
in the analog subsystem. This on-die isolation was
analyzed using a commercial electromagnetic field solver.
Isolation between subcircuits reaches 50 dB even for
highly conductive (lossy) substrate (1x103 S/m) with no
epi-layer, and separation between analog and digital
subcircuits as small as 200 µm. On-die isolation exceeds
50 dB if an epi-layer (5-125 S/m conductivity) is added in
the field solver simulations, thus benefiting the state-ofart technology of high-performance logic CMOS
technology.
Figure 16 shows results for 50 µm-thick deep nwell
lateral walls, which is appropriate for subsystem isolation
(not for individual transistor isolation). The 50 dB
isolation levels for thick-wall double-deep nwell isolation
(both aggressor and victim covered by deep nwells) and
frequency characteristics agree with experimental results
obtained by CMOS foundries later [27].
For the additional layout- and circuit-level isolation, RF
circuits are fully differential with layout of matched
devices based on a common centroid. Transistors sized
with W=250 µm-460 µm (common in 2.4 GHz/5.2 GHz
RF designs with 140 nm CMOS technology) Vt and Lmin
mismatches lead to gm mismatches smaller than 10%,
which imply >20 dB common-mode rejection. Once
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
251
Intel Technology Journal, Volume 8, Issue 3, 2004
substrate activity passes to the circuit signal lines with
some attenuation, layout- and circuit-level isolation
combined reaches above 20 dB as desired. An additional
“vertical grid” was simulated in the electromagnetic field
solver to minimize coupling between on-die metal traces,
which imposes also an isolation higher than 70 dB for this
path. This vertical grid was connected to the digital Vss
and finally completed the total isolation enabling RFdigital processor integration.
Table 3: Body bias and RF performance
Id
(mA)
|S21|
@ 5.2
GHz
Fmin
@ 5.2
GHz
(dB)
-0.5
30.02
3.253
1.05
0.7
-0.25
35.36
3.248
1.07
0.7
0.7
+0.25
49.82
3.122
1.14
0.7
0.7
+0.50
60.58
2.979
1.20
Vg (V)
Vd
(V)
0.7
0.7
0.7
Vb
(V)
Due to the introduction of deep nwells, body biasing
becomes an additional degree of freedom for both digital
and RF circuits. Reverse and forward body bias,
respectively, diminishes and augments the current driving
capability (hence gm) of the devices as can be seen by the
change in Ids with Vb in Table 3. However, as the current
driving capability increases, the drain junction
capacitance also increases as that junction becomes less
reverse biased, and the overall effect is a diminishment in
the RF performance with forward bias as represented by
the measured |S21| in Table 3. That is a clear departure
from the effect of body bias in digital circuitry where the
capacitances at the output of logical gates are dominated
by the gate capacitance of the following gate, and any
increase in current-driving capability implies a gain in
performance. Note also that, as expected, there is no
apparent effect of body bias on the carrier heating (drain
current excess thermal noise) by the high horizontal
electric field in the channel as Fmin merely tracks
variations in the device gain (|S21|) – higher |S21| (Table
3). Table 4 compares the behavior of identical transistors
inside and outside the deep nwell. A small but perceptible
increase in channel resistance diminishes the driving
current capability, diminishes the RF performance (|S21|),
augments Fmin, and augments Rn (the device in the deep
nwell departs from the optimum noise performance faster
with source impedance than the identical device outside
the well). Nevertheless, these are not compromising
effects. Similarly, from 25oC to 110oC the device
performance inside and outside the deep nwell showed a
less than 10% variation in RF and noise parameters.
Table 4: Transistor inside vs. outside deep nwell
Ids
(mA)
Ids
(dnwell)
(mA)
|S21|
@5.2
GHz
|S21|
@5.2
GHz
(dnwell)
Fmin
@5.2
GHz
(dB)
Fmin
@5.2
GHz
(dnwell)
(dB)
Rn/5
0
@5.2
GHz
(dB)
Rn/50
@5.2
GHz
(dnwell)
(dB)
48.0
46.3
3.338
3.036
1.15
1.71
0.22
0.35
Mixed-signal integration for high-performance Systemon-a-Chip (SoC) is thus enabled with minimal technology
modification, by adding deep nwell structures. By
exploring the spectrum structure of the substrate noise of
a high-performance microprocessor (1 GHz 55 W) with a
clock at GHz frequencies, and placing the feeble RF
signals (for a 50 MHz, –76 dBm-sensitivity receiver)
received from the antenna between harmonics of the
clock, we have shown that 70 dB of isolation is sufficient
to enable RF-high-performance digital processor
integration.
Note that this was an extreme case of RF and digital
processor integration in the same die. RF delicate circuits
are more likely to be integrated with sub 2 W digital
circuits or processors, instead of 55 W and above digital
processors. Therefore, less isolation between digital and
RF circuits is likely to be the typical case and deep nwell
structures might be only seldom used. Our investigation,
accordingly, supported the feasibility of RF and digital
processor integration in the same die with comfortable
margins.
CMOS SYSTEM ON A PACKAGE (SOP)
After digital substrate noise is successfully handled,
silicon area availability is the next and final road block to
be cleared in the path to RF and digital processor
integration in the same die. Multiband radios do require a
large number of passive components that take
considerable area on-die and are also non-scaling
components. This area road block will be adequately
cleared in this section as a side effect of a creating highperformance wireless SoP.
Higher performance RF and microwave transceivers
require high-performance active devices and high-quality
passives. On the silicon die, only the former is available.
Integrated passives have poor quality factors (Q),
typically around 5 or 7 at low GHz frequencies. A
significant improvement in such a scenario is found when
passives are implemented on the package substrate.
That’s because Q is a ratio of energy accumulated in the
component to its losses per cycle of the operating
frequency. Therefore, since energy is associated with the
volume occupied by the electric and magnetic fields of
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
252
Intel Technology Journal, Volume 8, Issue 3, 2004
the passive component and the losses are predominantly
associated with the surface of the conductors used in
those passives at low GHz frequencies, just by being able
to use more volume leads to higher quality passives. In
this sense, the height from the bottom metal layer to the
top metal layers in the silicon back-end is around 5 µm,
whereas the metal layer separation on the package
substrate is around 30 µm. Then, a 6X improvement in
quality for the passives could be expected by moving a
passive component from the silicon die to the low-cost
package substrate. Unfortunately, dielectric losses on the
package substrate (which uses organic materials) is
significantly higher than the SiO2-based interlayer
dielectric used on the die, and final quality factor
improvement, though still realizable, is somehow lower
than 6X [10].
Back of wafer/die ~ 100-400µm
and a high quality impedance converter needs to be
placed between the antenna and transistors’ drain for
high-power transmissions. High-quality impedance
converters are just not available on-die. Moreover, the
impedance converters solve the problem of high voltage
swing by trading it for high current handling capabilities.
This means the RF choke used in the PA of Figure 19
needs to carry currents on the 1A peak levels or more at
times. For the sake of reliability, such wide metal traces
have to be used on die to support these currents that the
RF choke becomes plagued with parasitics and then it is
useless even at low GHz frequencies. Moving the
passives to the package neatly solves the problems of the
high-quality impedance converter and the high-current
handling capabilities of the RF choke.
Metal 6 to metal 1 separation ~ 5µm
RF in+ Vn
~
R1
L1
M1
C1
Metal layer separations ~ 30µm
L2
IM
Package pins
NFIMN=f (L1, QL1, fop)
Figure 17: Die and package metal stacking and
dielectric separations
It is interesting to realize that despite being higher than
on-die, the still relatively modest Qs of passives on the
package warrant optimization of the final LNA design by
backing off, explained earlier. In order to better
appreciate this, Figure 18 shows that depending on the Q
being 5 or 10 for inductors in the IMN, a floor of 1 dB
noise figure for the LNA is already established if too-high
inductor values are placed in that IMN. This means that
transistors in the LNA need to be properly sized with the
minimization of inductance in the IMN included. Note
that according to Figure 18, for instance, for an LNA at
2.4 GHz it is necessary to use inductor values below 8.5
nH in the Q=10 curve, to have any chance of making a 1
dB noise figure LNA, even if the rest of the LNA is
completely noiseless.
Similar benefits are accrued by other RF key components.
Oscillators and VCOs also develop less phase noise if
higher quality passive components are used, since all the
noise contributors (not only transistor’s 1/f noise)
influence phase noise, as was discussed in the 1/f section
of this paper.
In RF Power Amplifiers (PAs), the moving of passives
from the die to the package is not just a benefit but
actually an enabling development. As CMOS transistors
scale, less voltage swing is tolerated at their terminals,
NF
5
4.5
( dB)
Q=5
4
3.5
5.2
Q=10
3
Q=5
2.4
Q=10
2.5
2
1.5
1 dB
1
0.5
0
0
2
4
6
8
10
12
14
16
L1
(
Figure 18: Input-matching-network-limited LNA’s
noise figure
2
V /R
V2/R
Vcc
RFchoke
Impdnc
Trfrmr
M1
RF_out
Figure 19: Conceptual description of power
amplifiers
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
253
Intel Technology Journal, Volume 8, Issue 3, 2004
Figure 20 shows the area of a flip-chip package where
passives can be realized by trace engineering (no discrete
components added), and a 3D blowup of an example of
moving impedance transformers and RF chokes to the
package substrate. Figure 21 depicts the 1-to-1
correspondence between schematics and trace layout
realization for the PA’s network of Figure 20.
2.4
GHz
transmission
serpentine
given the already densely occupied processors and
chipset dies, if we are to integrate radios into those dies,
the radio needs to be of minimum area. Moving the
passives to the package then becomes again an enabling
technology.
towards the dipole antenna
3.1cm
dense
line
flip-chip
Antenna
contacts
4 RF-chokes based on
4 ¼ wavelength transmission lines
¼ wavelength
transmission lines
for power combination
at antenna
~1cm
Multi-layer
3.1cm
Die
Impedance transformers
Area
available
for circuit on
k
PA1PA2 PA3PA4
Antenna
contacts
Silicon die
Figure 20: High-quality RF passives realized by trace
engineering on the flip-chip package substrate
More compact realizations of the PA’s network can be
developed, and Figure 22 shows how a power
combination of four identical PAs can be realized on-die,
occupying a small area. The PA’s transmission power in
the range of 0.5 W at 5.5 GHz can be readily achieved
with low voltage (1.2 V) transistors.
package
Figure 22: Power combining of 4 PAs on the package:
0.5 W of power transmission with 1.2 V transistors
2.4
GHz/5.2
GHz
Synthesizer
2.4
PA
GHz
5.2
PA
GHz
2.4 GHz LNA
passives
PA Mx Synth Mx LNA
Impedance
RF-choke1
5.2 GHz LNA
passives
Mem buffer
tx1
d1
Vcc
General purpose
processor
(MAC, windows)
Transformer 1
RF
chk1
die
Impdnc
M
All active devices
on-die make for a
thin area along the
top die side
RF_out +
Trfrmr
On-die
d1
tx1
Vcc
Pa_on
Ms
tx2
d2
M
Impdnc
Trfrmr
Dipole
antenna
RF-choke2
Impedance
RF_out -
RF
chk2
Vcc
d2
tx2
Figure 21: Schematic and trace layout
correspondence for a PA with its entire passive
network on the package substrate
Moving RF passives to the package substrate can be
carried on to the extreme of moving all the big passives
out of the silicon die and onto the package. Such a move
is supported by the large pin count capability of flip-chip
packages and ultimately will lead to leaving only the
transistors on-die. Having only the radio’s transistors ondie makes the radio occupy only a small area in that die.
Figure 23 shows a concept where all the radio’s passives
are moved to the package surface and the radio becomes
a slim area on the north side of the die. The rest of the die
is now available to host a digital processor. Note that
package
Figure 23: All-passives-on package radio concept.
Radio becomes a slim silicon area north of the die,
and a general-purpose digital processor is hosted on
the same die.
Another path for the SoP with all (or most of) the RF
passives moved to the package substrate is explored later
in this paper.
SIGMA-DELTA ADCS AND DACS:
TRADING VOLTAGE RESOLUTION FOR
TIME RESOLUTION
Analog to Digital Converters (ADCs) and Digital to
Analog Converters (DACs) are the specialty circuits at
the interface between the RF front-end and the digital
communication processing circuits. It is important to
point out these analog specialty circuits are in fact
influenced on the architecture and circuit level by CMOS
scaling and the new requirements of high performance
CMOS radios.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
254
Intel Technology Journal, Volume 8, Issue 3, 2004
As CMOS scales, lower voltages and integrated
components variations (mismatches) become higher and
higher barriers for voltage resolution. Fortunately, scaling
delivers higher and higher speeds in the circuits and
enables large oversampling ratios for both ADCs and
DACs operations. Large oversampling ratios allow for a
directly proportional diminishment in (kT/C)-limited
(thermal noise limited) capacitor sizes used in switching
circuits [28, 29]. Clever high oversampling circuit
techniques can make systematic offset produced by
mismatch in the components show up as high-frequency
noise, which can be readily filtered in order to achieve a
high number of bits resolution. Oversampling also
alleviates the anti-aliasing analog filtering order prior to
the ADC blocks, and quantization noise can be shaped so
that its frequency content is higher away from the
frequencies of interest for the information being
processed [30].
More importantly, radio signals are bandpass in nature.
Before downconversion to baseband, those signals are
made of a narrow band information signal on a higher
frequency carrier. These signals can appropriately be
tackled with sigma-delta ADCs and DACs, in particular
the bandpass version of these, where all the benefits
mentioned above for trading off voltage resolution for
time resolution are at the core of these converter
concepts. Oversampling ratios can reach values well
above 50 (bandpass signal width to sampling frequency)
[28, 29].
It is important to relate this change in gears for ADC and
DAC converters due to CMOS scaling to a disruptive
effect in radio transceiver architecture: these new ADCs
and DACs allow for simplified RF front-ends and
synthesizers in multiband multiprotocol radios. That is
because the RF front-end will not, for instance, chase the
narrow channels anymore during communications as
defined by 802.16. Chasing narrow channels is now
moved to the digital domain, since high-speed ADCs
support a much larger bandwidth to be processed in the
receiver, and high-speed DACs allow for offsetting
signals for proper channelization prior to sending them to
the RF transmitter.
Flexible Radios: a Practical Vision
The CMOS-based computer industry takes full advantage
of CMOS scaling to produce always changing ever more
powerful computing platforms. This was thought to be in
fundamentally stark contrast to the standards-driven
communications industry.
P4
XScale
cores
DSP+RF
chipset
open standard
Figure 24: Flexible radios: a PC platform for merging
computing and communications under alwaysevolving communication standards
Despite that, the reality of recent years and plans for the
foreseeable future appear to show a path for further
integration of computing and communications. The
apparent paradox disappears when one considers
communication standards are always evolving documents.
The standards themselves generate new standards and
addenda to standards are always being made. Standards
serve as guidelines for products, and these products at a
given time only attend to a limited subset of
recommendations in the standards to warrant a
compatibility stamp like “Wi-Fi” or “WiMAX.” Other
parts of the standards are left for future implementation.
In such a state of affairs, computing and communications
do share the always-evolving aspects that are the spirit of
the CMOS business model.
At the implementation level, multiband, multiprotocol
radio for always-evolving communication standards is too
complex a system to fit in a Pentium or chipset die. It
would be probably better to start thinking that the PC
motherboard will be a multiprocessor platform whose
ecosystem will be populated by new chips besides the
Pentium and chipset.
Figure 24 illustrates in a highly simplified abstraction the
addition of a flexible radio to a multiprocessor PC
platform. Note the suggestion that Intel® XScale®
technology should become the general-purpose processor
to handle all aspects of reprogrammability and hardware
switchability of the flexible radio. XScale would be also
in
charge
of
a
software-MAC
(for
easy
reprogrammability) and all network layers of operation
for the radio.
Figure 25 depicts a flexible radio concept and its
realization as one package and two silicon chips. This
concept exploits all the CMOS technology and package
®
Intel XScale is a registered trademark of Intel
Corporation or its subsidiaries in the United States and
other countries.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
255
Intel Technology Journal, Volume 8, Issue 3, 2004
advancements discussed in this paper. Note that the two
chips on the same package is only a suggestion, not the
only possible solution. Nevertheless, this particular
configuration allows for more surfaces on the package
onto which the RF passives can be realized.
Note the addition of an Ethernet interface in die 1 of this
flexible radio realization (Figure 25) allows for easy
access and reprogramability/reconfigurability in case this
flexible radio is taken out of the multi-processor PC
platform and placed as a stand alone core component of
radio base stations.
communications, and CMOS is very well posed to be the
technology to enable this merging.
ACKNOWLEDGMENTS
We acknowledge Tim Teckman (ICG/BWD) for support
and discussions on goals for silicon technology and
802.16.
REFERENCES
[1] Craninckx, J., Steyaert, M., “A CMOS 1.8 GHz lowphase-noise Voltage Controlled Oscillator with
Prescaler,” Int. Solid-State Circ. Conf. (ISSCC), San
Francisco, Feb. 1995.
[2] Shaeffer, D.K.; Lee, T.H., “A 1.5 V, 1.5 GHz CMOS
Low Noise Amplifier,” IEEE J. Solid-State Circ.
(JSSC), May 1997.
[3] King-Chun Tsai, Gray, P.R., “A 1.9 GHz, 1 W CMOS
class E power amplifier for wireless
communications,” IEEE J. Solid-State Circ. (JSSC),
July 1999.
[4] Lam, C., Razavi, B., “A 2.6 GHz/5.2 GHz frequency
synthesizer in 0.4 µm CMOS technology,” IEEE J.
Solid-State Circ. (JSSC), May 2000.
[5] Samavati, H., Rategh, H. R., Lee, T.H., “A 5 GHz
CMOS wireless LAN receiver front-end,” IEEE J.
Solid State Circ. (JSSC), May 2000.
Figure 25: A flexible radio realization: top and lateral
view of a realization with two dies and one package.
More package surface for the RF passive network.
CONCLUSION
CMOS radio capabilities were demonstrated from RF
circuits at 1 GHz in 1995 to millimeter wave circuits at
100 GHz in 2004. Intrinsic CMOS transistors’ physical
deficiencies have found adequate compensation in
innovative circuit-level solutions. These solutions exploit
and advance the understanding of fundamental
mechanisms behind excess thermal noise and 1/f noise
processes in semiconductor devices and how it affects
circuit performance. Exploitation of digital substrate
noise spectrums and advances in CMOS packaging
enabled superior performance for CMOS-based wireless
SoP solutions.
Building on these developments, a practical flexible radio
concept can be realized. This concept recognizes the
always-evolving nature of communications standards as
akin to the constantly evolving computer industry. This
concept supports the seamless merging of computing and
[6] L. M. Franca-Neto et al., “17 GHz and 24 GHz LNA
Designs based on Extended-S-parameter with
Microstrip-on-Die in 0.18 µm Logic CMOS
Technology,” European Solid-State Circ. Conf.
(ESSCIRC), Lisbon, September 2003.
[7] L. M. Franca-Neto et al., “64 GHz and 100 GHz
VCOs in 90 nm CMOS Using Optimum Pumping
Method,” Int. Solid-State Circ. Conf. (ISSCC), San
Francisco, Feb. 2004.
[8] Aoki, I. et al., “Distributed active transformer-a new
power-combining and impedance-transformation
technique,” IEEE Trans. Microwave Theory &
Techniques, Jan. 2002.
[9] L. M. Franca-Neto et al., “Enabling HighPerformance Mixed-Signal System-on-a-Chip (SoC)
in High Performance Logic CMOS Technology,”
IEEE VLSI Circ. Symp., Hawaii, June 2002.
[10] L. M. Franca-Neto (invited paper), “System-on-apackage (SoP) Solution for High Performance
RF/Microwave Systems,” Progress in
Electromagnetic Research Symp., Cambridge, July
2002.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
256
Intel Technology Journal, Volume 8, Issue 3, 2004
[11] S. Sze, Physics of Semiconductor Devices, Wiley
Inter Science, 2nd ed., 1981.
[12] L. M. Franca-Neto, “Noise in High Electric Field
Transport: the Ergodic Method,” Ph.D. Thesis,
Stanford University, 1999.
[13] R. E. Collin, Foundations of microwave
engineering, 2nd ed., McGraw Hill, 1992.
[14] IEEE 802.16 standard.
[15] B. Razavi, RF Microelectronics, Prentice Hall, 1998.
[16] Farjad-Rad, R. et al., “A low-power multiplying DLL
for low-jitter multigigahertz clock generation in highly
integrated digital chips,” IEEE J. Solid-State Circ.
(JSSCC), Dec. 2002.
[17] B. Razavi, “A Study of Phase Noise in CMOS
Oscillators,” IEEE J. Solid-State Circ. (JSSC), March
1996.
[18] A. Hajimiri, T.H. Lee, “A General Theory of Phase
Noise in Electrical Oscillators,” IEEE J. Solid-State
Circ. (JSSC), Feb. 1998.
[19] A. Hajimiri, T.H. Lee, “Phase Noise in CMOS
Differential LC Oscillators,” IEEE VLSI Symp on
Circ., 1996.
[20] A. Hajimiri et al., “Phase Noise in Multi-Gigahertz
CMOS Ring Oscillators,” IEEE Custom Integrated.
Circ. Conf. (CICC), 1998.
[21] Rael, J.J., Abidi, A. A., “Physical Processes of Phase
Noise in Differential LC Oscillators,” IEEE Custom
Integrated. Circ. Conf. (CICC), 2000.
[22] T. H. Lee, The Design of CMOS Radio-Frequency
Integrated Circuits, Cambridge University Press,
1998.
[23] G. D. Vendelin, A. M. Pavio, and U. L. Rohde,
Microwave Circuit Design using Linear and
Nonlinear Techniques, John Wiley, 1990.
[24] I. Bahl and P. Bhartia, Microwave Solid State
Circuit Design, John Wiley, 1988.
[25] Applied Wave Research, http://www.awr.com.*
[26] M. van Heijningen et al., “Substrate noise generation
in complex digital systems: efficient modeling and
simulation methodology and experimental
verification,” Int. Solid State Circ. Conf. (ISSCC),
San Francisco, Feb. 2001.
[27] TSMC documents.
[28] B. Leung, VLSI for wireless communications,
Prentice Hall, 2002.
[29] J. C. Candy and G. C. Temes, Oversampling DeltaSigma Data Converters: theory, design and
simulation, Wiley Inter-Science, 1992.
[30] R. van de Plassche, CMOS integrated Analog-toDigital and Digital-to-Analog converters, 2nd ed.,
Kluwer, 2003.
AUTHORS’ BIOGRAPHIES
Luiz M. Franca-Neto earned his Electronic Engineering
degree, with distinction, from ITA/CTA, SJC, Sao Paulo,
Brazil, in 1989, and he received the TASA award for
being first in class in communications. He received his
M.Sc. and Ph.D. degrees from Stanford University, all in
Electrical Engineering, in 1995 and 1999, respectively.
From 1990 to 1992, he was a design engineer with
ALCATEL/Elebra
Telecom
for
public
telecommunications and optical line terminal equipment.
In USA from 1993-1996, he has worked for HP-Labs,
Palo Alto, CA, and Texas Instruments, Dallas, TX. He
was with Intel R&D Labs from 1999-2004, where he led
research on CMOS for RF/Microwave/Millimeter wave
frequencies. He created new circuit design methods such
as “backing off” for LNAs and “optimum pump” for
VCOs with demonstrated circuits operating from 2.4 GHz
to 100 GHz (a world record for CMOS). He led the
investigations for substrate noise in Pentium 4 processors
and deep nwell isolation where he articulated how
substrate noise spectrum structure can be exploited for
full integration of digital processors and RF delicate
circuits in the same die. Also in the labs, Luiz led the
research to move all RF passives from the die to the
substrate package in order to realize higher performance
RF System-on-Package and free silicon area for hosting
more digital functions and general-purpose processors.
Since February 2004, Luiz has led the WiMAX RF &
Analog IC internal development within the ICG/BWD
group in Santa Clara. His homepage is http://wwwsnow.stanford.edu/~franca.*
Roger Eline received a B.S.E.E. degree from UC Davis
and an M.S.E.E from Santa Clara University in 1991.
Since then his work has focused on RF and microwave
communication system development. He currently works
for the BroadBand Wireless Division of Intel, where he
manages the Platform Engineering Group. He has been
with Intel for one and a half years developing low-cost
IEEE 802.16 baseband and radio reference platforms
based on Intel’s IEEE 802.16 baseband processor/modem
ASIC. His e-mail is Roger.j.eline at intel.com.
Balvinder Bisla received his B.Sc. degree at Sussex
University, England in 1984. He then worked at
Rutherford Appleton Labs in the UK before moving to
the USA to work on wireless metering and global
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
257
Intel Technology Journal, Volume 8, Issue 3, 2004
positioning systems. He was a principal RF engineer with
Iospan Wireless where they developed the world’s first
MIMO-OFDM system. Currently he is working at Intel
on RF and microwave communication issues for WiMAX
products. His e-mail is Balvinder.s.Bisla at intel.com.
Copyright © Intel Corporation 2004. This publication
was downloaded from http://developer.intel.com/.
Legal notices at
http://www.intel.com/sites/corporate/tradmarx.htm.
Fully Integrated CMOS Radios from RF to Millimeter Wave Frequencies
258
For further information visit:
developer.intel.com/technology/itj/index.htm
Copyright © 2004 Intel Corporation. All rights reserved.
Intel is a trademark or registered trademark of Intel Corporation or its subsidiaries in the United States and other countries.
For a complete listing of trademark information visit: www.intel.com/sites/corporate/tradmarx.htm
易迪拓培训
专注于微波、射频、天线设计人才的培养
网址:http://www.edatop.com
射 频 和 天 线 设 计 培 训 课 程 推 荐
易迪拓培训(www.edatop.com)由数名来自于研发第一线的资深工程师发起成立,致力并专注于微
波、射频、天线设计研发人才的培养;我们于 2006 年整合合并微波 EDA 网(www.mweda.com),现
已发展成为国内最大的微波射频和天线设计人才培养基地,成功推出多套微波射频以及天线设计经典
培训课程和 ADS、HFSS 等专业软件使用培训课程,广受客户好评;并先后与人民邮电出版社、电子
工业出版社合作出版了多本专业图书,帮助数万名工程师提升了专业技术能力。客户遍布中兴通讯、
研通高频、埃威航电、国人通信等多家国内知名公司,以及台湾工业技术研究院、永业科技、全一电
子等多家台湾地区企业。
易迪拓培训推荐课程列表: http://www.edatop.com/peixun/tuijian/
射频工程师养成培训课程套装
该套装精选了射频专业基础培训课程、射频仿真设计培训课程和射频电
路测量培训课程三个类别共 30 门视频培训课程和 3 本图书教材;旨在
引领学员全面学习一个射频工程师需要熟悉、理解和掌握的专业知识和
研发设计能力。通过套装的学习,能够让学员完全达到和胜任一个合格
的射频工程师的要求…
课程网址:http://www.edatop.com/peixun/rfe/110.html
手机天线设计培训视频课程
该套课程全面讲授了当前手机天线相关设计技术,内容涵盖了早期的
外置螺旋手机天线设计,最常用的几种手机内置天线类型——如
monopole 天线、PIFA 天线、Loop 天线和 FICA 天线的设计,以及当前
高端智能手机中较常用的金属边框和全金属外壳手机天线的设计;通
过该套课程的学习,可以帮助您快速、全面、系统地学习、了解和掌
握各种类型的手机天线设计,以及天线及其匹配电路的设计和调试...
课程网址: http://www.edatop.com/peixun/antenna/133.html
WiFi 和蓝牙天线设计培训课程
该套课程是李明洋老师应邀给惠普 (HP)公司工程师讲授的 3 天员工内
训课程录像,课程内容是李明洋老师十多年工作经验积累和总结,主要
讲解了 WiFi 天线设计、HFSS 天线设计软件的使用,匹配电路设计调
试、矢量网络分析仪的使用操作、WiFi 射频电路和 PCB Layout 知识,
以及 EMC 问题的分析解决思路等内容。对于正在从事射频设计和天线
设计领域工作的您,绝对值得拥有和学习!…
课程网址:http://www.edatop.com/peixun/antenna/134.html
`
易迪拓培训
专注于微波、射频、天线设计人才的培养
网址:http://www.edatop.com
CST 学习培训课程套装
该培训套装由易迪拓培训联合微波 EDA 网共同推出,是最全面、系统、
专业的 CST 微波工作室培训课程套装,所有课程都由经验丰富的专家授
课,视频教学,可以帮助您从零开始,全面系统地学习 CST 微波工作的
各项功能及其在微波射频、天线设计等领域的设计应用。且购买该套装,
还可超值赠送 3 个月免费学习答疑…
课程网址:http://www.edatop.com/peixun/cst/24.html
HFSS 学习培训课程套装
该套课程套装包含了本站全部 HFSS 培训课程,是迄今国内最全面、最
专业的 HFSS 培训教程套装,可以帮助您从零开始,全面深入学习 HFSS
的各项功能和在多个方面的工程应用。购买套装,更可超值赠送 3 个月
免费学习答疑,随时解答您学习过程中遇到的棘手问题,让您的 HFSS
学习更加轻松顺畅…
课程网址:http://www.edatop.com/peixun/hfss/11.html
ADS 学习培训课程套装
该套装是迄今国内最全面、最权威的 ADS 培训教程,共包含 10 门 ADS
学习培训课程。课程是由具有多年 ADS 使用经验的微波射频与通信系统
设计领域资深专家讲解,并多结合设计实例,由浅入深、详细而又全面
地讲解了 ADS 在微波射频电路设计、通信系统设计和电磁仿真设计方面
的内容。能让您在最短的时间内学会使用 ADS,迅速提升个人技术能力,
把 ADS 真正应用到实际研发工作中去,成为 ADS 设计专家...
课程网址: http://www.edatop.com/peixun/ads/13.html
我们的课程优势:
※ 成立于 2004 年,10 多年丰富的行业经验,
※ 一直致力并专注于微波射频和天线设计工程师的培养,更了解该行业对人才的要求
※ 经验丰富的一线资深工程师讲授,结合实际工程案例,直观、实用、易学
联系我们:
※ 易迪拓培训官网:http://www.edatop.com
※ 微波 EDA 网:http://www.mweda.com
※ 官方淘宝店:http://shop36920890.taobao.com
专注于微波、射频、天线设计人才的培养
易迪拓培训
官方网址:http://www.edatop.com
淘宝网店:http://shop36920890.taobao.com
Was this manual useful for you? yes no
Thank you for your participation!

* Your assessment is very important for improving the work of artificial intelligence, which forms the content of this project

Download PDF

advertising