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US 20140254827A1
(19) United States
(12) Patent Application Publication (10) Pub. No.: US 2014/0254827 A1
Bailey
(43) Pub. Date:
Sep. 11, 2014
(54)
(57)
METHOD AND CIRCUITRY FOR
PROCESSING AUDIO SIGNALS
ABSTRACT
(71) Apphcam: APHEX’ LLC’ Burbank’ CA (Us)
An audio signal processing method and circuitry that pro
cesses an input audio signal by ?ltering the input audio signal
(72)
With a high pass ?lter to produce a ?ltered audio signal, Which
Inventor,
James L Bailey Pomona CA (Us)
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is input to a compressor. A ?rst intermediate audio signal is
(73) Assignee: APHEX’ LLC Burbank CA (Us)
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produced based on the compressor output signal. The ?ltered
3
audio signal is also input to a harmonics generator that pro
(21) App1_ NO; 13/788,845
duces harmonics of the ?ltered audio signal. A second inter
mediate audio signal is produced based on such harmonics. A
(22) Filed;
third intermediate signal is produced based upon the input
audio signal. An output audio signal is produced by combin
ing the ?rst intermediate audio signal, the second intermedi
ate audio signal and the third intermediate audio signal. The
Mar, 7, 2013
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components of the ?ltered audio signal that contribute to the
?rst intermediate audio signal relative to the dynamic range of
(2006-01)
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the harmonics that contribute to the second intermediate
USPC .......................................................... .. 381/98
audio signal, thus enhancing the input audio signal.
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US 2014/0254827 A1
Sep. 11,2014
US 2014/0254827 A1
METHOD AND CIRCUITRY FOR
PROCESSING AUDIO SIGNALS
pressor in reducing the masking of the harmonics by the
frequencies that pass through the high pass ?lter.
BACKGROUND OF THE INVENTION
dynamic range compression on a ?ltered audio signal pro
duced by a high pass ?lter, which causes the effect of the
?ltering to be more consistently audible when the reduced
[0009] Moreover, the compressor preferably performs
[0001] 1. Field of the Invention
[0002] The present invention relates to the processing of
audio signals to enhance the quality and clarity and/or other
characteristics of the audio signals.
[0003] 2. State oftheArt
[0004] In general, the concept of processing an audio signal
to enhance the quality, clarity and/or other characteristics of
the audio signal is known. US. Pat. No. 4,150,253 to Knoppel
addresses this concept and describes a circuit for generating
low order and high order harmonics of an input audio signal.
[0005] Another relevant patent in the prior art is US. Pat.
No. 5,424,488 that describes a circuit for generating transient
discriminate harmonics of an input audio signal.
dynamic range components of the ?ltered input audio signal
are integrated with the input audio signal. In an illustrative
embodiment, the compressor reduces dynamic range of the
?ltered audio signal as compared to said input audio signal at
a ratio between 5 to 1 and 15 to 1.
[0010] In the preferred embodiment, the ?rst intermediate
audio signal is produced by attenuating the compressor out
put signal, the second intermediate audio signal is produced
by attenuating the harmonics of the ?ltered audio signal out
put by the harrnonics generator, and the third intermediate
audio signal is produced by the input audio signal. The attenu
ating of the compressor output signal as well as the attenuat
ing the harmonics of the ?ltered audio signal can be con
SUMMARY OF THE INVENTION
trolled dynamically by user input. The attenuating of the
[0006] The prior art references discussed above suffer from
limitations in that the harmonics can be masked by certain
attenuation. The attenuating of the harmonics produced by
?ltered audio signal preferably causes attenuation of such
?ltered audio signal in a range from Zero attenuation to full
higher frequency components of the audio signal (such as
frequency components higher than at least 5 KHZ and possi
bly additional higher frequency components in the range
between 500 HZ and 5 KHZ), thereby reducing the audibility
the harmonics generator preferably causes attenuation of
of such harmonics. There is a signi?cant need for an improved
method and circuit to address this problem.
[0007] The present application is an audio signal process
form or combination thereof.
ing method and circuitry that processes an input audio signal
by ?ltering the input audio signal with a high pass ?lter to
produce a ?ltered audio signal. For example, the high pass
?lter can have a low frequency cutoff in the range between
500 HZ and 5 KHZ. In this con?guration, the ?ltered audio
signal includes frequency components of the input audio
signal higher than at least 5 KHZ and possibly additional
higher frequency components in the range between 500 HZ
and 5 KHZ. The lower frequency components of the input
audio signal lower than the low frequency cutoff of the high
pass ?lter are ?ltered from the ?ltered audio signal. The
?ltered audio signal is input to a compressor that produces a
compressor output signal. A ?rst intermediate audio signal is
produced based on the compressor output signal. The ?ltered
audio signal is also input to a harmonics generator that pro
duces harmonics of the ?ltered audio signal. A second inter
mediate audio signal is produced based on the harmonics of
the ?ltered audio signal. A third intermediate signal is pro
duced based upon the input audio signal. An output audio
such harmonics in a range from 0 dB to 20 dB.
[0011] The audio signals processed by the method and cir
cuitry of the present application can be in digital form, analog
BRIEF DESCRIPTION OF THE DRAWINGS
[0012]
FIG. 1 is a block diagram of an audio signal pro
cessing system according to the present application.
[0013] FIG. 2 is a block diagram of an illustrative embodi
ment of the compressor circuitry of FIG. 1.
[0014] FIG. 3 is a schematic of an analog circuit implemen
tation that embodies the compressor circuitry of FIG. 2.
[0015] FIG. 4 is a block diagram of an illustrative embodi
ment of the harmonics generator circuitry of FIG. 1.
[0016] FIG. 5 is a schematic diagram of an analog circuit
implementation that embodies the harmonics generator cir
cuitry of FIG. 4.
[0017] FIG. 6 is a schematic diagram of another analog
circuit implementation of the harmonics generator circuitry
of FIG. 1.
[0018] FIG. 7 is a block diagram of a stereo audio signal
processing system according to the present application.
signal is produced by combining the ?rst intermediate audio
DETAILED DESCRIPTION OF THE PREFERRED
EMBODIMENTS
signal, the second intermediate audio signal and the third
intermediate audio signal.
cessing system according to the present application. An elec
[0019]
FIG. 1 is a block diagram of an audio signal pro
[0008] In the preferred embodiment. the compressor is con
?gured to reduce the dynamic range of components of said
?ltered audio signal that contribute to the ?rst intermediate
audio signal relative to the dynamic range of the harmonics
that contribute to the second intermediate audio signal. In this
con?guration, when the ?rst and second intermediate audio
signals are combined to produce the output audio signal (and
trical audio signal is provided by an audio signal source 101
such as a microphone, radio tuner, CD player, audio receiver,
thus the harmonics that contribute to the second intermediate
transmission or storage, or other suitable audio component.
audio signal are integrated with the reduced dynamic range
The audio signal can be passed through an optional ampli?er
components that contribute to the ?rst intermediate audio
signal), the harmonics become more apparent and audible in
the output audio signal. This is due to the effect of the com
to two discrete signal processing paths. The ?rst path 105
audio ampli?er, audio preampli?er, portable music player,
mobile phone, computer or other data processing system that
stores audio ?les in digital form and possibly plays the stored
audio ?les, automobile audio head unit, satellite or cable
set-top box, a television, an audio processor for audio signal
103 and then split (or copied in the digital domain) for supply
passes the audio signal without any enhancement. The second
Sep. 11,2014
US 2014/0254827 A1
path includes a series of signal processing stages comprising
and a second attenuator 115. The outputs of the three signal
[0023] The mixer 117 and ampli?er 119 also preferably
operate over the full audio bandwidth with unity gain output
of the ampli?er 119 as compared to the audio signal supplied
by the audio signal source 101. The ampli?er 119 can provide
for impedance matching for the output audio signal. It can
also provide for ampli?cation of the output audio signal to a
level that compensates for any gain changes caused by the
processing paths (i.e., the outputs of the attenuators 111, 115)
signal processing of the system.
are supplied to a mixer 117 for recombination. The relative
[0024] The highpass ?lter 107 can have a low frequency cut
off in the range from 500 HZ to 5,000 HZ. The low frequency
cut off of the high pass ?lter 107 can be dictated by user input
via HPF user input control 121. In this con?guration, the
a high-pass ?lter 107, a compressor 109 and an attenuator
111. The signal output from the high-pass ?lter 107 is split (or
copied in the digital domain) for supply to the compressor 109
and to a third signal processing path that includes a series of
signal processing stages comprising harmonics generator 113
gain of the signal output from the three signal processing
paths (i.e., the output of the attenuators 111, 115 and the input
signal) in the recombined signal can be adjusted by the mixer
117. The resultant recombined signal produced by the mixer
117 can be passed through an ampli?er 119 for output as an
output audio signal as shown. The output audio signal can be
?ltered audio signal includes frequency components of the
input audio signal higher than at least 5 KHZ and possibly
additional higher frequency components in the range between
fed to an output transducer (such as an audio speaker) or to
500 HZ and 5 KHZ. The lower frequency components of the
another suitable audio device.
[0020] The compressor 109 of FIG. 1 is an ampli?er that
outputs one decibel of volume increase per X number of
decibels of increase present at its input that exceeds a thresh
old Y. The parameters X and Y can be ?xed parameters,
dynamic parameters or user-controlled parameters. A 10:1
compressor would output 1 dB of volume increase for every
10 dB of volume increase present at its input that exceeds the
threshold. Below the threshold, 1 dB in of volume increase
equals 1 dB out of volume increase. The compressor 109
reduces the volume of loud sounds or ampli?es quiet sounds
input audio signal lower than the low frequency cutoff of the
high pass ?lter are ?ltered from the ?ltered audio signal that
is passed by the high pass ?lter 107.
[0025]
The compressor 109 can provide for a ?xed or
dynamic compression ratio or other ?xed or dynamic param
eters (such as attack time or release time parameters or com
pression density), which can be dictated by user input via
compressor user input control 123. The attenuator 111 can
provide for an adjustable attenuation factor between full
attenuation and zero attenuation, which can be dictated by
user input via user input control 125. The attenuator 127 for
by narrowing or “compressing” the dynamic range of the
audio signal. Compression is often used to make music sound
louder without increasing its peak amplitude. By compress
the third signal processing path can provide for an adjustable
attenuation factor preferably between 20 dB attenuation and
ing the peak (or loudest) signals, it becomes possible to
input controls 127.
[0026] The audio circuit of FIG. 1 provides for audio signal
processing that involves both dynamic level compression of
components of the input audio signal and the generation of
harmonics of components of the input audio signal together
with functionality to integrate the enhancements of both func
increase the overall gain (or volume) of a signal without
exceeding the dynamic limits of a reproduction device or
medium. The net effect, when compression is applied along
with a gain boost, is that relatively quiet sounds become
louder, while louder sounds remain unchanged or are reduced
in volume.
[0021] The harmonics generator 113 of FIG. 1 generates
harmonics of the components of the input audio signal that are
passed by the high pass ?lter. A harmonic is a signal whose
frequency is an integral (whole-number) multiple of the fre
quency of some reference signal, in this case the components
of the input audio signal that are passed by the high pass ?lter.
For example, if f represents the main (or fundamental) fre
quency of the components of the input audio signal that are
passed by the high pass ?lter, then the frequency f is the
frequency at which most of the energy is contained, or at
which the signal is de?ned to occur. For a signal whose
fundamental frequency is f, the second harmonic has a fre
quency 2f, the third harmonic has a frequency of 3f, and so on.
Signals occurring at frequencies of 2f, 4f, 6f, etc. are called
even harmonics; and the signals at frequencies of 3f, 5f, 7f,
etc. are called odd harmonics. The frequency f is typically
expressed in hertz. A signal can, in theory, have in?nitely
many harmonics. The introduction of harmonics can signi?
cantly enhance the sound quality of the audio signal.
[0022] The ampli?er 103, if present, can provide for imped
ance matching with respect to the audio signal source 101
and/or provide for common mode rejection. It can also pro
vide for ampli?cation of the input audio signal to a level that
is optimal for the signal processing functions of the system as
described herein. The ampli?er 103 preferably operates over
the full audio bandwidth, which is generally de?ned as
encompassing frequencies between 20 HZ and 20,000 HZ.
zero attenuation, which can be dictated by user input via user
tions as part of an output audio signal. In the preferred
embodiment, the compressor is con?gured to reduce the
dynamic range of ?ltered components of the input audio
signal relative to the dynamic range of the harmonics pro
duced by the harmonic generator. In this con?guration, when
the harmonics are integrated with the reduced dynamic range
?ltered components of the input audio signal as part of an
output audio signal, the harmonics become more apparent
and audible in the output audio signal. Moreover, the com
pressor preferably performs dynamic range compression on
?ltered audio signal produced by a high pass ?lter, which
causes the effect of the low-frequency ?ltering to be more
consistently audible when the reduced dynamic range com
ponents of the ?ltered input audio signal are integrated with
the input audio signal.
[0027] FIG. 2 is a high level block diagram of an illustrative
embodiment of the compressor 109 of FIG. 1. The compres
sor 109 includes a voltage-controlled ampli?er 201 electri
cally coupled between an input signal path 203 and an output
signal path 205, a recti?er 207 electrically coupled to the
output signal path 205 for rectifying the output signal and
producing a recti?ed output signal 209, and an adaptive ?lter
211 that processes the recti?ed output signal 209 to generate
a feedback signal 213 for controlling operation of the voltage
controlled ampli?er 201. The adaptive ?lter 211 operates with
a multiplicity of interactive layered time constants such that
feedback control signal 213 is instantaneously and continu
ously dependent upon both the average and the transient peak
Sep. 11,2014
US 2014/0254827 A1
value of the recti?ed output signal 209. Certain characteris
tics of the adaptive ?lter 211 can affect the operating param
eters of the compressor 109 (such as compression ratio, attack
time or release time parameters or compression density) and
can be dictated by user input via compressor user input con
trol 123 for dynamically updating the operating parameters of
the compressor 109.
[0028] FIG. 3 is a schematic diagram showing an exem
plary analog circuit implementation of the compressor of
FIG. 2.
[0029] The voltage-controlled ampli?er 201 of FIG. 2 is
implemented in FIG. 3 by the combination of U1 through U4,
and resistor R1 through resistor R12. These components form
a practical voltage-controlled ampli?er circuit with an audio
input, an audio output, and a control input. The components
U1, U2, and U4 can be implemented with any suitable type of
operational ampli?er such as the LF347 quad bifet op amp
sold commercially by Texas Instruments, Inc. of Dallas, Tex.
The component U3 is a voltage-controlled attenuator such as
the VCA1001 chip available from Aphex, LLC of Burbank,
Calif. The component U1 acts as a phase invertor for the input
signal. Pins 2 and 7 of the voltage-controlled attenuator U3
receives as inputs the audio input signal as well as the phase
inverted input signal (180 degrees out of phase) produced the
component U1. The voltage-controlled attenuator U3 attenu
ates the audio input signal as well as the phase-inverted input
signal input on pins 2 and 7 under control of feedback signal
supplied to pin 9 of the voltage-controlled attenuator U3. The
attenuated signals are output on pins 17 and 13 of the voltage
the series-coupled capacitors C8 and C9 is coupled to the
intermediate node of the series-coupled resistors R14 and
R15. The resistor R15 is a variable resistor (such as a poten
tiometer) whose resistance is controlled by compressor user
input control 123, which can be realized by a knob or other
suitable user input mechanism. By way of example only, the
capacitor C9 is 1 uF and the capacitor C8 is 4.7 uF, while the
resistor R14 is 20 k-ohms and the variable resistor R15 ranges
from 100 k-ohms to 1 M-ohm.
[0033]
It can be seen that there are a multiple of time con
stants within the RC network of FIG. 3. The recti?ed current
from resistor R13 can be called the “charging current” since
this current charges up the capacitors C9 and C8. The dis
charge path of capacitors C9 and C8 includes only resistors
R14 and R15. The charging and discharging of the capacitors
C9 and C8 charge dictate the function of the adaptive ?lter.
Assuming a suf?ciently large recti?ed audio signal is applied
to diode D1, a current will ?ow through resistor R14 and
capacitor C9. Several paths of current are possible. The resis
tors R14 and R15 will carry a current to ground directly, and
capacitor C9 will pass current through resistor R15 and
capacitor C8. Capacitor C8 receives a current through capaci
tor C9 and resistor R14. As the capacitors C9 and C8 build a
charge, the current divides in different proportions among
these paths. This is due to asymptotic charging curve of a
capacitor.
[0034]
More speci?cally, capacitor C8 acts as a relatively
slow charging ?lter, while capacitor C9 acts relatively fast.
?er comprised of U2, and resistor R7 through resistor R11,
Since they are stacked in a series, the net ?lter output voltage
is the sum of the voltages on capacitors C9 and C8. Capacitor
which rejects the common mode output of U3 and renders the
?nal VCA audio output signal 205. The output of U2 feeds the
C8 charges up from current brought down through the branch
resistors R13 and R14, and also through the branch of resistor
controlled attenuator U3 and supplied to a differential ampli
diode D1 which acts as a half wave recti?er. Negative half
R13 and capacitor C9. Capacitor C8 charges initially faster
waves conducted through diode D1 generate the charging
current of the adaptive ?lter, as described below in great
through the branch of resistor R13 and capacitor C9 because
capacitor C9 is accepting maximum charge and dumps a
detail. The negative polarity of the feedback control signal
relatively large current through capacitor C8. This rapidly
output from the adaptive ?lter as described below is buffered
by the op amp U4 (which is in a follower con?guration) and
supplied to pin 9 of the voltage-controlled attenuator U3 to
control the attenuation function of the voltage-controlled
adds a partial charge to capacitor C8, but the charging of
capacitor C8 by the current through capacitor C9 is short lived
attenuator U3. High amplitude transient output signals output
from pins 17 and 13 of the voltage-controlled attenuator U3
are ?ltered by the adaptive ?lter to cause greater attenuation
by the voltage-controlled attenuator U3. This produces an
adaptive amplitude regulating effect due to the nature of the
adaptive ?lter as described below.
[0030] Note that the resistors labeled Trims 1 and 2 can be
adjusted to obtain the lowest control feed through with an
average offset of zero volts DC at the output of U2 as
described in US. Pat. No. 5,483,600.
[0031] The recti?er 207 ofFlG. 2 is implemented in FIG. 3
by diode D1 and resistor R13, which serves as a half-wave
recti?er. By way of example only, resistor R13 can be 20
k-ohms. The diode D1 also functions as a reverse current
blocker for the adaptive ?lter such that reverse current cannot
?ow from the adaptive ?lter to the audio output node.
[0032] The adaptable ?lter 211 of FIG. 2 is implemented in
FIG. 3 by a resistor-capacitor (“RC”) network including
capacitors C9 and C8 that are connected in series between the
recti?ed output signal (produced at the output of the recti?er
output of resistor R13) and ground together with resistors
since capacitor C9 rapidly charges to nearly the input voltage
and its charging current then stops. If the input voltage is now
removed, the output voltage of the adaptable ?lter is dictated
by the voltage developed across capacitor C9. If the input
voltage remains longer, capacitor C8 will sustain further
charging through the branch of resistors R13 and R14. This
will be much slower than the initial charge of capacitor C8 by
the charging current of capacitor C9. As the voltage charge of
capacitor C8 rises, the total voltage across capacitors C8 and
C9 will nearly equal the input voltage. This does not bring a
halt to charging currents, because capacitor C9 will begin to
discharge through resistor R14 as the charge of capacitor C9
rises. There will be a transition period wherein the charge of
capacitor C9 relatively slowly rises and the charge of capaci
tor C8 falls. Equilibrium will be reached when the voltage
charges on capacitors C9 and C8 equal the voltage division of
the ladder of resistors R13, R14 and R15. Since resistor R15
may have a variable resistance, the charge ratio of capacitors
C9 and C8 may also be variable.
[0035] When the input voltage is removed, capacitors C9
and C8 begin to discharge. The discharge paths of capacitors
R14 and R15 that are also connected in series between the
C9 and C8 tend to circulate through their parallel resistances
of resistors R14 and R15, respectively, since the path up
through resistor R13 is blocked by the 10 reverse impedance
recti?ed output signal and ground. The intermediate node of
of diode D1. The time constant of R14-C9 is much faster than
Sep. 11,2014
US 2014/0254827 A1
that of R15-C8, so capacitor C9 can discharge relatively fast
while capacitor C8 discharges more slowly.
would be desirable to prevent the listener from perceiving that
the level of the average sounds surrounding the transients are
[0036]
In the preferred embodiment, the adaptive ?lter is
modulated by the fast peak ?owing gain reduction. If the
con?gured to react to different types of input signals as set
forth in US. Pat. No. 5,483,600.
sounds are mainly non transient, the ripple of the output
signal of the adaptive ?lter should be minimized to reduce
waveform distortion of the audio output. This requires gen
erally slower ?lter averaging. It has been shown that the
responses of the adaptive ?lter of FIG. 3 convolve to meet this
various conditions.
[0037] Speci?cally, if the input signal is a short transient,
capacitor C9 will charge and discharge relatively fast with
little charge going to capacitor C8. The output of the adapt
able ?lter will contain a fast rise and fall.
[0038] If the input is a repeating series of short transients,
capacitor C9 will ?rst charge up then gradually charge and
discharge at a proportionately lesser average voltage as
capacitor C8 progressively builds up its charge. The output of
the adaptive ?lter will contain a fast attack but the output
ripple will slowly diminish. Finally, the output will contain a
relatively slow fall time.
[0039] If the input contains a fairly steady signal with a fast
attack and decay, capacitor C9 at ?rst attains a high charge,
but subsequently its charge gives way to the charge which
builds up on capacitor C8. The output of the adaptive ?lter
contains a fast rise followed by a slight fall to a steady value
followed by a slow fall.
[0040] If the input is a slow rising and relatively steady
signal, then capacitor C9 does not attain much charge because
capacitor C8 can attain a charge fast enough through 40 the
branch of resistor R13 and resistor R14 to track the input rate
of rise. The output of the adaptable ?lter is basically that of the
voltage on capacitor C8 alone with relatively little contribu
tion from capacitor C9.
[0041] As noted above, the value of resistor R15 dictates
the relative weight of charge on capacitor C9 and capacitor C8
[0044]
Resistor R15 can have a variable resistance which
controls the “release time constant” of the adaptive ?lter of
FIG. 3 because it has the effect of slowing down or speeding
up the discharge of the stored voltage of the adaptive ?lter. A
faster release time constant, which results from a reduction in
the value of resistor R15, causes the compressor to release
faster, thus maintaining a higher average output level.
Another consequence of reducing the value of resistor R15 is
that of changing the relative charge equilibrium of capacitor
C9 and capacitor C8. Speci?cally, when resistor R15 is
smaller, the branch of resistor R13, resistor R14 and resistor
R15 divides differently and a smaller proportion of the input
voltage is developed across the capacitor C8. This makes
capacitor C9 contain a larger relative charge and capacitor C8
a smaller relative charge when a sustained non-transient input
signal. Thus, not only is the average compression release
made faster, but the transformation of peak following to aver
age following is less complete. This leaves peak following
more present in the VCA control, and the density of compres
sion consequently increases. The density of compression is
the extent to which the amplitudes of audio signal peaks are
made uniform at the expense of dynamic range. Such an
by changing of the point of charge equilibrium. Speci?cally,
increase in density of compression is a useful effect because
as resistor R15 becomes smaller, the point of charge equilib
rium weighs the charge of capacitor C9 heavier and the output
signal of the ?lter output contains a higher portion of the
the usual reason for speeding up the release time of a com
capacitor C9 charge for the conditions of more sustained and
less transient input signals.
[0042] It can be generalized that the faster time constants
related to capacitor C9 more closely follow the peaks of the
input signal and the slower time constants related to capacitor
C8 more closely follow the average of the input signal. There
fore, the output of the adaptable ?lter contains both average
and peak following components which are interactive.
[0043] Moreover, it can be generalized that the output of the
adaptive ?lter of FIG. 3 acts mainly like a peak following ?lter
for a single transient input. The adaptive ?lter transforms
from peak following to average following for repetitive tran
sient inputs. For a non-transient input, the adaptive ?lter
transforms from peak to average following if the input has a
fast attack, but remains mainly an average follower if the
non-transient input has a slower attack. In this manner, the
adaptive ?lter of FIG. 3 adapts to the transient nature of the
pressor is to gain greater compression density. Typically,
however, the increased density so derived comes at the cost of
a much more intense gain modulation effect causing transient
peaks to audibly modulate the level of smaller signals. The
compressor of FIG. 3 reduces this effect considerably by the
combination of time constants and the tendency to convolve
between peak and average following.
[0045] It is also contemplated that resistor R13 can have a
variable resistance which is dictated by user input controls to
control the “attack time constant” of the adaptive ?lter of FIG.
3 because the resistor R13 has the effect of speeding up or
slowing down the charging of the stored voltage of the adap
tive ?lter. Speci?cally, the attack time constant is determined
by the values of resistor R13 and the capacitors C8 and C9. A
faster attack time constant, which results from a reduction in
the value of resistor R13, causes the compressor to attack
faster toward peak following. A slower attack time constant,
which results from an increase in the value of resistor R13,
causes the compressor to attack slower toward average fol
input signal, and to the input signal’s average characteristics.
lowing.
The bene?t of this action is in allowing the adaptive ?lter of
[0046] The compression ratio of the adaptive ?lter of FIG.
3 is dependent on the gain of the feedback loop through
FIG. 3 to react in the manner which is optimum for any audio
signal. This statement is based upon the supposition that a
transient signal sounds better if compressed with fast time
constants, allowing the compressor to effectively reduce the
transient amplitude but not sustain the gain reduction long
enough for the listener to notice a level reduction of the
average sound level surrounding the transient peak. The tran
sient peaks which repeat at su?icient frequency should at ?rst
be compressed with a fast action but if their repeating pattern
is sustained long enough, a slower averaging gain reduction
op-amp U4 and R5 to pin 9 of the voltage-controlled attenu
ator U3. It is contemplated that such gain (and the resulting
compression ratio) can be varied according to user input
controls if desired.
[0047] It is noted that the analog circuit implementation of
FIG. 3 shows a speci?c polarity for diode D1. This polarity is
shown for illustration only. It should be obvious that the
polarity of the diode D1 can be reversed and the circuit will
function identically except the polarity of the input and output
Sep. 11,2014
US 2014/0254827 A1
voltages would be reversed. If capacitors C9 and C8 are
implemented as polarized capacitor types, their polarization
in the circuit can be properly arranged.
[0048] FIG. 4 is a high level block diagram ofan embodi
ment of the harmonics generator 113 of FIG. 1. The harmon
ics generator 113 of FIG. 4 includes a multiplier 401 electri
cally coupled between an input signal path 403 and an output
signal path 405 and an automatic gain control (AGC) function
407. The multiplier 401 can be a linear multiplier having a
transfer function of (XY)/K where X is the signal at an X
input, andY is the signal at aY input, and K is a constant. The
resistor R108. The AGC OTA U100A is con?gured by the
resistors VR100, VR103, R109, C100, R103, and R102 to
generate an X-input signal that modulates the multiplier cir
cuit 401 in accordance with the input signal 403 and the
output X-signal. Feedback control of the AGC OTA U1 00A is
provided by the signal path from pin 8 of the Darlington pair
transistors (which produces a signal corresponding to the
buffered output X-input signal on pin 9) to pin 2 of the AGC
OTA U100A via resistor-capacitor network R102, R103, and
C100. The parameters for the AGC circuit 407, such as the
limiter threshold, the attack and/ or release times as well as the
multiplier circuit 401. The parameters of the AGC function
compression ratio of the AGC can be determined for the best
effect in a particular application. One or more of the param
eters of the AGC circuit 407 can also be dictated by user input
controls for user control of the audio effect, if desired.
407, such as a speci?c compression ratio as well as attack and
release times, can be determined for best audio effect in a
FIG. 5 can be con?gured as a transient discriminator harmon
AGC function 407 generates the X-input signal that modu
lates the multiplier circuit 401 in accordance with the input
signal (Y input) and the output X signal supplied to the
particular application. Nonetheless, the attack time is
required to be of a ?nite value. One or more parameters of the
AGC function 407 can also be dictated by user input controls
for user control of the audio effect, if desired. The AGC
function 407 and the multiplier 401 cooperate to generate
harmonics of components of the input signal 403 at the output
405.
[0049]
FIG. 5 is a schematic diagram showing an exem
plary analog circuit implementation of the harmonics genera
tor 113 of FIG. 4. The harmonics generator 113 includes two
functional circuits (multiplier circuit 401 and an AGC circuit
407) based on an LM13700 dual operational transconduc
[0053]
In one embodiment, the harmonics generator 113 of
ics generator that generates a relatively high level of harmon
ics at the start of the amplitude envelope of the input signal
and then reduces the level of harmonics generated during an
intermediate portion of the amplitude envelope of the input
signal and then generates a relatively low level of harmonics
at the terminal part of the amplitude envelope of the input
signal. The relative timing of the leading edge of the interme
diate portion of the amplitude envelope where the level of
harmonics is reduced is dictated by the attack time parameter
of the AGC circuit 407 of FIG. 5. The attack time parameter
of the AGC circuit 407 represents how quickly the gain of the
AGC circuit 407 is adjusted when the input signal magnitude
tance ampli?er (OTA) chip sold commercially by Texas
exceeds a threshold limit. The attack time parameter of the
Instruments, Inc. of Dallas, Tex. The LM13700 includes two
OTAs (labeled U100A and U100B in FIG. 5) as well as two
AGC circuit 407 is dictated by the capacitor C101, the resistor
R108 and the variable resistance VR102. User input controls
Darlington transistor pairs. The multiplier circuit 401 is real
ized by the OTA U100B of the LM13700 integrated circuit.
The AGC function 407 is realized by the OTA U100A and the
two Darlington transistor pairs of the LM13700 integrated
circuit. Note that the pin numbers 1 through 16 of FIG. 5 are
the pin numbers of the LM13700 integrated circuit.
[0050] The multiplier circuit 401 also includes capacitor
C101, resistors R106, R107 and R108, and variable resistors
VR101 and VR1 02. By way of example only, capacitor C101
may be 22F, resistors R106 and R107 may be 10 k-ohms,
resistor R108 may be 5 k-ohms, variable resistor VR101 may
be 50 k-ohms, and variable resistor VR102 may be 1 k-ohm.
[0051] The AGC circuit 407 also includes capacitor C100,
resistors R100, R101, R102, R103, R104, R105 and R109,
can control the variable resistance VR102 in order to adjust
the attack time and the corresponding relative timing of the
leading edge of the intermediate portion of the amplitude
envelope where the level of harmonics is reduced. The dura
tion of the intermediate portion of such amplitude envelope is
?xed by the design of capacitor C101 and the resistors R108
and R106 of FIG. 5. It is also contemplated that one or both
values for resistors R1 08 and R1 06 (and the resulting duration
of the intermediate portion of such amplitude envelope) can
be varied according to user input controls if desired. The
compression ratio of the AGC circuit 407 is dependent on the
gain of the feedback loop through resistors R106 and R108 to
pin 13 of the multiplier OTA U100B. It is contemplated that
such gain (and the resulting compression ratio) can be varied
and variable resistors VR100 and VR103. The dotted block
501 are the two Darlington transistor pairs of the LM13700
according to user input controls if desired.
integrated circuit, which is used to buffer the output of the
OTA 100A. By way of example only, capacitor C100 may be
4.7 uF, resistor R100 may be 250 k-ohm, resistor R101 may
be 30 k-ohm, resistors R102, R103 and R104 may be 10
k-ohm, resistor R105 may be 100 k-ohm, resistor R109 may
be 10 M-ohm, variable resistorVR100 may be 25 k-ohm, and
of the two Darlington transistor pairs at pin 9 of the LM13700
integrated circuit, is supplied as a current input to pin 16 of the
multiplier OTA U100B by the input resistance of variable
[0054]
The X-input signal, which is generated at the output
resistorVR101. TheY input signal is supplied to the input pin
13 of the multiplier OTA U100B by coupling capacitor C101
The input signal 403 is split and supplied to the AGC
and resistor R108. The multiplier OTA U100B is con?gured
by variable resistor VR102 and R107. The multiplier OTA
U100B generates the harmonics output signal 405 at pin 12 of
the multiplier OTA U100B. It is noted that pin 12 is a high
impedance output and must not be signi?cantly loaded by any
external impedances for correct operation.
[0055] In alternate embodiments, the multiplier circuit 401
can be realized by any variety of voltage-controlled ampli?
ers (VCAs) with a signal input, signal output and gain control
circuit 407 via resistor R100 and to the multiplier OTA
U100B as the Y-input via DC coupling capacitor C101 and
plier if the signal input is equated to the “Y” channel input and
variable resistorVR103 may be 50 k-ohm. The variable resis
tors VR103 and R109 provide for nulling of the control feed
through of the OTA 100A. Capacitor C101 isolates the DC
input offset voltage of the multiplier circuit. In addition, posi
tive voltage +E and negative voltage —E are provided to the
circuit. By way of example only, the positive voltage +E may
be 15 V and the negative voltage —E may be —15 V.
[0052]
input. If a VCA is utilized, it can be de?ned as an XY multi
Sep. 11,2014
US 2014/0254827 Al
the gain control input is equated to the “X” channel input. The
only difference between a linear multiplier and a VCA used as
a multiplier is that the transfer function of a VCA “X” input is
case, stage 611 can be realized by a summing ampli?er stage
that generates an output signal that represents the difference
usually exponential, so that the output transfer function
between the negative voltage transient component signal gen
erated by the signal path 603 and the input signal supplied by
would be generally (Y)(exp X)/K. Nevertheless, the output of
signal path 602.
the linear multiplier or the VCA used as a multiplier will
[0058]
contain harmonics of the input signal, and the circuit shown in
processing system based upon the circuitry of FIG. 1 repli
cated for processing left and right audio signal channels,
FIG. 4 remains valid.
FIG. 7 is a block diagram of a stereo audio signal
FIG. 6 is a schematic diagram showing another
respectively. The system can include a user input control 121'
exemplary analog circuit implementation of the harmonics
that dictates operation of the respective high pass ?lter cir
cuits 107L, 107R that process the left and right audio signal
[0056]
generator 113 of FIG. 1. The harmonics generator 113 of FIG.
channels in a manner similar that described above for the user
6 splits the input signal for supply to two discrete signal
processing paths. The ?rst signal processes path 601 passes
the input signal without any enhancement. The second signal
input control 121. The system can also include a user input
processing path 603 includes a series of audio signal process
ing stages 605, 607, 609. Stage 605 includes an RC network
tors 111L, 111R in a manner similar that described above for
the user input control 125. The system can also include a user
control 125' that dictates operation of the respective attenua
(resistor R235, capacitor C220, resistor R236) and diode
input control 127' that dictates operation of the respective
D210 that cooperate to generate negative voltage transient
component signal based upon the input signal. The RC net
attenuators 115L, 115R in a manner similar that described
work generates a transient component waveform based upon
above for the user input control 127. The system can also
include compressor user input control 123' that dictates
operation of the respective compressors 109L, 109R in a
the input signal. The diode D210 is con?gured for half wave
recti?cation where negative voltages of the transient compo
nent waveform generated by the RC network pass through the
manner similar that described above for the user input control
diode D210. In this manner, diode D210 produces a negative
a stereo linking function that mixes or otherwise processes the
voltage transient component signal that is based upon the
input signal. Stage 607 is an inverting ampli?er stage that
produces a positive voltage transient signal of inverse polarity
left and right channel compressors 109L, 109R so that each
with respect to the negative voltage signal produced by the
diode D210. Stage 607 is realized by the operational ampli?er
U203-A and resistors R237 and R238. The gain of the invert
123. Note that the compressoruser input control 123' employs
feedback control signals generated by the adaptive ?lter of the
channel is compressed to the same extent.
[0059]
It is emphasized that the circuits described herein
are not intended as a limitation to the embodiments of the
present invention. There are many more circuits that can be
gain control stage that employs variable resistor VR205 to
adapted to follow the teaching of the present application.
Moreover, it is contemplated that the circuits (or parts therein)
can be implemented by a programmed data processing sys
provide an adjustable voltage gain to the positive voltage
transient component signal output from stage 607. The output
tem (such as a digital signal processor) that operates on audio
signals in the digital domain. In this case, an analog audio
of stage 609 is the output of the signal path 603. In this
manner, the signal path 603 outputs a positive voltage tran
sient signal that is derived from the input signal. The input
signal supplied by signal path 601 as well as the positive
input signal is converted into a digital audio signal by sample
and hold circuitry and suitable analog-to-digital conversion
ing ampli?er stage 607 is proportional to (—R238/R37),
which is set to (—10K/4K) or —2.5 as shown. Stage 609 is a
voltage transient component signal generated by the signal
circuitry well known in the electronic arts. In order to output
an analog audio signal, the digital audio signal is converted
into an analog signal by suitable digital-to-analog conversion
path 603 are supplied to a difference ampli?er stage 611 that
generates an output signal that represents the difference
circuitry well known in the electronic arts.
between the positive voltage transient component signal gen
erated by the signal path 603 and the input signal supplied by
signal path 602. Stage 611 is realized by an operational ampli
described herein can be integrated as part of an audio com
?er U203-B and resistors R245, R246, R247, and R248. The
input signal supplied by signal path 602 is coupled to the
inverting input of the operational ampli?er U203-B via the
resistor network R245, R246, and the positive voltage tran
sient component signal generated by the signal path 603 is
supplied to the non-inverting input of the of the operational
ampli?er U203-B via the resistor network R247, R248. The
output signal of the difference ampli?er stage 611 has asym
metry that represents harmonics of the input signal supplied
by signal path 602 and thus is labeled harmonics output in
FIG. 6. One or more parameters of the signal processing path
603 (such as the gain afforded by the variable resistor VR205
of the gain control stage 609) and/orparameters of the mixing
stage 611 can be dictated by user input controls for user
control of the audio effect, if desired.
[0057] A variety of modi?cations can be made to the circuit
of FIG. 6. For example, stage 607 can be realized by a non
inverting ampli?er stage that generates a negative voltage
transient component signal based on the input signal. In this
[0060]
Moreover, it is contemplated that the circuit
ponent such as a microphone, radio tuner, CD player, audio
receiver, audio ampli?er, portable music player, mobile
phone, computer or other data processing system that stores
audio ?les in digital form and possibly plays the stored audio
?les, automobile audio head unit, satellite or cable set-top
box, a television, an audio processor for audio signal trans
mission or storage, or other suitable audio components.
[0061]
Moreover, it is contemplated that the audio signal
processing functions can be embodied in software (such as an
application or app) that is loaded onto a data processing
system (such as a digital signal processor or computer or
mobile phone). During operation, the software is executed by
the data processing system to carry out the audio signal pro
cessing functions of the circuitry as described herein in the
digital domain in order to enhance an input audio signal. The
input audio signal can be stored in the memory of the data
processing system orpossibly streamed to the data processing
system via network communication.
[0062] There have been described and illustrated herein
several embodiments of a method and circuitry for processing
audio signals. While particular embodiments of the invention
Sep. 11,2014
US 2014/0254827 A1
have been described, it is not intended that the invention be
limited thereto, as it is intended that the invention be as broad
in scope as the art will allow and that the speci?cation be read
likewise. Thus, while particular circuit elements and compo
nent values have been disclosed, it will be appreciated that
other circuit elements and component values can be valid as
well. It will therefore be appreciated by those skilled in the art
that yet other modi?cations could be made to the provided
invention without deviating from its spirit and scope as
claimed.
What is claimed is:
1. An audio signal processing method that processes an
input audio signal, comprising:
a) ?ltering the input audio signal with a high pass ?lter to
produce a ?ltered audio signal;
b) inputting said ?ltered audio signal to a compressor that
produces a compressor output signal;
c) producing a ?rst intermediate audio signal based on the
compressor output signal;
d) inputting said ?ltered audio signal to a harmonics gen
erator that produces harmonics of said ?ltered audio
signal;
d) producing a second intermediate audio signal based on
the harmonics of the ?ltered audio signal;
e) producing a third intermediate signal based upon the
input audio signal; and
f) producing an output audio signal by combining the ?rst
intermediate audio signal, the second intermediate audio
signal and the third intermediate audio signal.
2. An audio signal processing method according to claim 1,
wherein:
the compressor is con?gured to reduce dynamic range of
components of said ?ltered audio signal that contribute
to the ?rst intermediate audio signal relative to dynamic
range of the harmonics that contribute to the second
intermediate audio signal.
3. An audio signal processing method according to claim 1,
wherein:
the compressor is con?gured to cause the harmonics that
contribute to the second intermediate audio signal to
become more apparent and audible in the output audio
diate audio signal involves amplifying at least one of the
?rst, second and third intermediate audio signals.
8. An audio signal processing method according to claim 1,
further comprising:
amplifying the output audio signal.
9. An audio signal processing method according to claim 1,
wherein:
the input audio signal is output from an ampli?er whose
input is electrically coupled to an audio source.
10. An audio signal processing method according to claim
1, wherein:
the compressor reduces dynamic range of said ?ltered
audio signal as compared to said input audio signal at a
ratio between 5 to 1 and 15 to 1.
11. An audio signal processing method according to claim
1, wherein:
the ?rst intermediate audio signal is produced by attenuat
ing the compressor output signal;
the second intermediate audio signal is produced by attenu
ating the harmonics of the ?ltered audio signal output by
the harmonics generator; and
the third intermediate audio signal is a copy of said input
audio signal.
12. An audio signal processing method according to claim
11, wherein:
the attenuating of the compressor output signal and the
attenuating of the input audio signal is controlled
dynamically in a reciprocal fashion in accordance with
user input; and
the attenuating the harmonics of the ?ltered audio signal is
controlled dynamically by user input.
13. An audio signal processing method according to claim
11, wherein:
the attenuating of said ?ltered audio signal causes attenu
ation of such ?ltered audio signal in a range from zero
attenuation to full attenuation.
14. An audio signal processing method according to claim
11, wherein:
the attenuating of the harmonics produced by the harmon
ics generator causes attenuation of such harmonics in a
range from 0 dB to 20 dB.
signal by reducing the dynamic range of the ?rst inter
mediate audio signal.
4. An audio signal processing method according to claim 1,
15. An audio signal processing method according to claim
1, wherein:
the audio signals processed by the method are in digital
wherein:
the compressor is con?gured to cause the effect of the
?ltering to be more consistently audible when the ?rst
intermediate signal is combined with the third interme
16. An audio signal processing method according to claim
1, wherein:
the audio signals processed by the method are in analog
diate audio signal.
5. An audio signal processing method according to claim 1,
17. Audio signal processing circuitry that processes an
wherein:
operational parameters of the compressor dynamically
adjusted by user input.
6. An audio signal processing method according to claim 1,
wherein:
the combining of the ?rst intermediate audio signal, the
second intermediate audio signal and the third interme
diate audio signal involves mixing the ?rst, second and
third intermediate audio signals.
7. An audio signal processing method according to claim 1,
wherein:
the combining of the ?rst intermediate audio signal, the
second intermediate audio signal and the third interme
form.
form.
input audio signal, comprising:
a) a high pass ?lter that ?lters the input audio signal to
produce a ?ltered audio signal;
b) a compressor that compresses said ?ltered audio signal
to produce a compressor output signal;
c) means for producing a ?rst intermediate audio signal
based upon the compressor output signal;
d) a harmonics generator with an input that receives said
?ltered audio signal, the harmonics generator producing
harmonics of said ?ltered audio signal;
e) means for producing a second intermediate audio signal
based on the harmonics produced by said harmonics
generator;
US 2014/0254827 A1
Sep. 11,2014
8
i) means for producing a third intermediate audio signal
based on said input audio signal; and
g) means for producing an output audio Signal by combim
ing the ?rst intermediate audio signal, the second inter-
mediate audio signal and the third intermediate audio
signal.
_
_
_
_
_
_
_
18. Audio signal processing c1rcu1try accord1ng to cla1m
17’ Whereln:
tho compressor is con?gured to reduce dynamic range of
components of said ?ltered audio signal that contribute
to the ?rst intermediate audio signal relative to dynamic
range of the harmonics that contribute to the second
intermediate audio signal.
19_ Audio signal processing circuitry according to claim
18, wherein;
the compressor is con?gured to cause the harmonics that
contribute to the second intermediate audio signal to
become more apparent and audible in the output audio
signal by reducing the dynamic range of the ?rst intermediate audio signal.
20. Audio signal processing circuitry according to claim
17, Wherein:
the compressor is con?gured to cause the effect of the
?ltering to be more consistently audible When the ?rst
intermediate 'signal is combined With the third interme
dlate audlo Slgnal'
_
_
_
_
_
21' Au‘llo Slgnal processmg Clrcultry accord1ng to Clalm
17’ Wherelp:
.
.
.
.
the ?rst 1ntermed1ate audio signal 1s produced by attenuat
mg the compressor output Signal;
the second intermediate audio signal is produced by attenu
ating the harmonics of the ?ltered audio signal output by
the harmonics generator; and
the third intermediate audio signal is produced by passing
said input audio signal Without any signal processing
being performed on said input audio signal.
22. Audio signal processing circuitry according IO claim
17, Wherein:
the? audiP Signals are? Processed by the alldio Signal Process
mg “@111er 111 dlgltal $011111
_
_
_
23. Audio s1gnal processing c1rcu1try accord1ng to cla1m
17> Whereln:
the_ audio_ signals
are processed by the audio signal process
_
_
1ng c1rcu1try 1n analog form.
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