HEWLETT-PACKARDJOURNAL JANUARY1971 © Copr. 1949-1998 Hewlett-Packard Co. A New High-Speed Multifunction DVM Plug-ins provide true rms ac capability as well as dc and ohms. Reading speed is 1000 per second of ohms and dc. By Craig Walter, H. Mac Juneau and Lee Thompson THERE is A NEED today for 'horizontal expansion' of the capability to measure dc and ac voltage, and resistance â€” more accuracy in general applications, with good repeat ability. In addition, there is a need to reduce the difficulty of eliminating errors under conditions such as making a floating dc measurement in the presence of both common mode and normal mode error signals, or avoiding large errors when measuring distorted sinusoids or waveforms without zero axis symmetry. For bench users, the need is not to make an already difficult measurement with greater precision, but there is a need to make measurements of adequate resolution with more ease and reliability. The measurement prob lems of the bench user have often been ignored in favor of 'greater' or 'more' rather than 'better'. Recently some instruments have been compromised in favor of the 'sys tem' aura â€” instrument optimization for system use at the expense of, rather than for, the bench user. The system user can generally find a unique solution to his unique problem. Having found it, he can operate his system properly. He has the time and generally the capital to find unique solutions to his individual prob lems. The bench user has a difficult problem each time he uses the instrument. A lash-up that provides optimum results for a measurement one day cannot be expected to yield the same results if the measurement problem changes. The bench user's measurement problems vary from day to day, and he seldom has time or money to invent solutions for each problem. Nevertheless, the problems associated with instrument use in both applications are not totally unique to either. A great degree of commonality exists. Providing solu tions for the common measurement errors that exist in bench applications can result in an instrument useful for systems use at little or small expense to the bench user. Indeed, the primary distinction between the two appli cation areas often is in the speed (and end use) of the measurement, not in the measurement itself. The instru ment, in bench applications, is interfaced to a human; in system applications, to a machine. If the same meas urement can be made with greater speed (and if, in addi tion, the data collected is easily transferred to a machine as well as a human), the instrument can also satisfy many system requirements. It should be possible, then, to pro vide an instrument that could properly be called a hybrid â€” optimized for bench and widely useful in systems. Cover: Both half-module and rack versions of the Hewlett-Packard Model 3480A/B are shown. Its reading speed is 1000 per second for ohms and dc to 1000 volts. True rms ac is an option. This article dis cusses dc processing and preconditioning as related to the Model 3480 A/ B. The ac conversion technique will be covered in detail in a future issue of the Hewlett-Packard Journal. In this issue: A New High-Speed Multifunction DVM, by Craig Walter, H. Mac Juneau and Lee Thompson PRINTED IN U.S.A. C HEM © Copr. 1949-1998 Hewlett-Packard Co. page 2 Fig. multi-function DVM new Hewlett-Packard Model 3480A/B is a high speed multi-function DVM capable of making 1000 readings per second up to WOO volts dc, and ohms down to 100 ohms full scale. It comes in both halt module and rack versions. Also shown here are the Models 3481 A Butter Amplifier (with one 10 V dc range), 3482A DC Range Unit and the 3484A Multifunction Unit. Bench Features Many features of the new HP Model 3480A/B, Fig. 1, which may seem mundane by themselves, combine to make the instrument easy to use. The display is easily readable, function and range information and the instru ment's functions are readily apparent. Autoranging is complete through all ranges and functions. The first read ing after an autorange cycle will be correct. The sample rate is fully controllable, from a sample initiated by a front panel pushbutton to >25 readings per second; higher sample rates â€” to 1000 per second â€” can be initi ated by external commands. Selectable filtering is pro vided for normal mode signals so that the user has a variety of choices between instrument settling time and interference rejection. The instrument is fully protected from damage from overvoltage, on any range, in any function. Overload recovery of the amplifiers is fast enough that a correct reading upon removal of the over load is guaranteed, even though the reading is started at the same time the overvoltage is removed. Zeroing requirements are held to a minimum. The instrument has accuracy commensurate with its resolu tion â€” at moderate or extreme speeds. The ac converter uses a thermopile to provide true rms conversion to elimi nate common ac measurement errors. And of great im portance, the instrument has minimum effect on the device or circuit under test â€” all injected currents have been eliminated or reduced to an insignificant level. Measurement Speed Terminology usually undergoes transformation when applications are changed; hence, for system use, the DVM is more properly called an A/D (analog to digital) converter. There can be significant differences. © Copr. 1949-1998 Hewlett-Packard Co. For bench applications, the sequence of readings per second should be quick compared with human reaction time. A good performance criterion is the time required for a single measurement, limited by the ability of the analog circuits to respond to sudden input changes and settle to the final value. Most systems are without similar restrictions â€” data can be absorbed at much faster rates. Thus, high reading speed is a primary concern, provided, of course, that response and settling times within the instrument are commensurate with its reading rate. Often speed is a major distinction. Because of the design compromises generally required to obtain high system speeds the system-designed analog to digital con verter (A/D) is ordinarily less versatile than its bench counterpart. It is task dedicated. A DVM, although slow er, has varied functional capability and more versatile signal preconditioning. Does the versatility necessary for bench use preclude the measurement speed deemed necessary for systems applications? In many system applications, the Model 3480A repre sents a solution to this apparent paradox. For systems use it can be considered a comparatively slow A/D (its maximum sampling speed is 1000/s) with 15 bit resolu tion and much greater signal-conditioning capability than the typical A/D. As such, it may be the best candidate for certain system situations. The digitizing technique used â€” successive approxima tion â€” provides moderate speed at low cost. It is inher ently simple and quite reliable. Reed relays in the A/D converter are replaced with semiconductor switches. The instrument accommodates plug-ins to provide signal pre conditioning (giving great measurement versatility), and the main frame contains the necessary power supplies and the A/D. The performance of either section is comple mented, not compromised, by the other. To avoid placing an undesirable burden on the bench user â€” added costs without benefit â€” the interfacing cir cuitry required to communicate with other instrumenta tion is not included within the basic instrument. These interfacing circuits are, instead, available as options. The emphasis during development was to capitalize on the digitizing speed made available by the successive approximation technique without compromising bench performance or increasing the instrument's basic cost; to solve those problems common to all traditional measure ments, not those related to instrument use in a single application â€” bench or system. Measurement Errors Errors associated with instrument use commonly fall within three groups. One group is caused by the measure ment circuit's interaction with its surroundings. The most common sources are normal mode and common mode generators which, directly or by magnetic coupling, in duce unwanted currents in the measuring loop; the sources can be magnetic fields from other instrumentation or voltages generated because of the flow of relatively large currents in the ground connections among instru ments. A second error group is caused by interactions between the measuring instrument and the circuit or de vice under test. Common mode or normal mode sources exist within virtually all instrumentation, and may force 'injected currents' into the circuits being measured; this may occur between 'high' and 'low' or between the instrument's chassis and its other input terminals. These currents can create errors by flowing through unbalanced impedances in the input circuit or, more importantly, may actually upset or change the characteristics of the circuit under test. Third among error sources is those caused directly by the instrument. The most obvious are errors in amplification, attenuation, or conversion that somehow modify the information being sought so that the data presented as absolute may actually be in error. Unfortunately, reducing the errors of one group does not guarantee reduction of the others. In fact, the opposite is often true. These sources of error, although classified separately, must be treated simultaneously to minimize the entire error matrix. ERRORS CAUSED BY THE MEASUREMENT CIRCUIT AND/OR INTERACTION WITH THE DVM CM and NM Sourcesâ€” Outside the DVM The most general measurement situation is shown in Fig. 2. A floating (above earth or chassis ground) meas urement is to be made across a resistance bridge. Al though the instrument ground and the source ground are on the same line, a voltage generator (the common mode source) will exist between them. The difference in the ground voltage is primarily due to induced currents and ground currents that flow between the two physically isolated grounding points. This current creates a voltage difference (the ground line or plane will always have some impedance) whose magnitude depends on the hook up and the environment into which it is placed. The common mode generators (ac and dc) will not, by themselves, cause measurement errors if the instru ment impedance from chassis to each of its input termi nals is infinite. Unfortunately, this is not generally the © Copr. 1949-1998 Hewlett-Packard Co. R, Â¡s Parallel Combination To High of R! and R3 Rb is Parallel Combination of R i and R4 DC Common Mode is the Voltage Across R* b = R2//R4 To Low Guard Should Go Here, but this Point Doesn't Exist in Actual Circuit Common Mode Generators Fig. 2. A general measurement situation where a floating measurement is made across a resistance bridge. Its ThÃ©venin equivalent is shown. Possible Common Mode Currents 3480A DVM normal mode errors â€” the common-mode signal has been converted to one in series with the primary measurement loop, that is, to a normal mode error signal. (The treat ment of common mode errors, once the conversion to normal mode occurs is then identical to that for normal mode errors.) These signals create undesirable errors â€” either dc offsets or a time varying voltage which may cause the DVM's display to 'rack.' Because the instrument's 'high' terminal is generally a point or line rather than a plane, its impedance to chassis is generally quite high and common mode errors in this input lead can generally be ignored. If, however, the im pedance from high to low (> 1010n, <50 pF for each of the 3480's plug-ins) is not extremely high and a guard is either not available or used, significant errors can result â€” a 60 Hz common mode voltage of 10 V, an imbalance R of 1 kn, and an input capacity of 1000 pF combine to generate an error of 3.7 mV. Normal mode filtering can, of course, reduce the ac errors but not without a corre sponding increase in measurement time (the response time of the filter must be added to the instrument's basic digitizing time). The instrument impedance from low to chassis is usually much lower because of the instrument's physical construction; 'low' will generally be a plane â€” capacities of several thousand picofarads are not unusual. Guarding must then be used to reduce these errors. Connecting the guard, Fig. 4, will effectively bootstrap that portion of the impedance between low and chassis that is terminated on guard. It can be of little help, obviously, for that imped ance not interrupted by guard. nection Shunts Common Mode Current Away From Source Resistances Source Ground Instrument Common Mode Source Ground (Chassis) Fig. 3. Possible common mode currents from external sources. Assume Z6 Â« Z, and Z,. case. Because impedance is finite between the instru ment's high and low terminals and the point to which the common-mode generator is referenced (the instrument's chassis), Fig. 3, current will flow through each of the imbalance resistors. The resulting voltage drop across these imbalance R's will enhance or oppose Eln, creating © Copr. 1949-1998 Hewlett-Packard Co. Common Instrument Ground (Chassis) Mode Source Added Â¡, AddÂ¡tjonal strumentation Without Isolation is Used. Fig. 4. Proper guard connections will shunt most com mon mode current away from source resistances. The degradation in Z6 that is so undesirable can easily occur when the instrument is interfaced to other equip ment. If, for example, the data generated by the DVM is to be used to provide hard copy, a printer or other record ing device will be electrically tied to the DVM's data output lines. If, too, the instrument is to be remotely programmed or externally commanded, the program source must be electrically connected to the DVM. As the DVM's data programming lines are referenced to the instrument's low terminal, a floating measurement will be impossible unless this added instrumentation can also be floated. Floating this instrumentation, unfortunately, will reduce ZCl and cause deterioration of the system's CMR. Injected currents from low to chassis will also be significantly increased â€” unless the output or controlling circuitry is completely isolated from its chassis. If the output and control lines can be so referenced, and if iso lation can be provided within the DVM, these system errors can be eliminated. The Model 3480A/B and its plug-ins have digital output and programming options for necessary electrical isolation without degrading other performance criteria (measurement speed, susceptibility to electrical interference). The isolated programming op tion also provides program storage. The physical architecture of the Model 3480A/B has eliminated the necessity for costly 'box-within-a-box-construction' (all internal circuitry surrounded by guard) â€” yet CMR for the Model 3480A/B and any of its plug-ins is >80 dB at 60 Hz for a 1 kn imbalance. Physical spac ing between the internal circuitry ('low') and chassis is as large as practicable. Where large spacings are impractical, individual shields are employed. 'Box-within-a-box' con struction is necessary only when much higher levels of CMR are required. While reducing errors caused by ex ternal CM voltages, this type of construction may accen tuate measurement errors caused by injected currents. Where guarding is required within the instrument (the transformer, power supply heat sink, and plug-in covers are the principal guard shields) the necessary care has been taken to eliminate time-varying voltages in their vicinity. The result is an extraordinarily favorable set of tradeoffs. The complete measurement problem has been taken into account. Errors from all sources, not just a few, have been reduced together in appropriate amounts. Normal Mode Filtering Normal mode filtering, again, can be used to reduce these errors, but measurement speed is reduced. The most obvious solution to this tradeoff is to use a digitizing technique that provides filtering â€” i.e., integration. Hence, the recent popularity of dual-slope DVM's. This com promise does not solve the entire measurement problem unless the injected currents are also minimized. Those currents may do nothing to the instrument because of its inherent rejection, but can and do create subtle and seem ingly mysterious changes in the circuit under test. Inte gration is also only effective for noise whose frequencies are related to (and multiples of) the converter's integration period. Although filtering by integration can, theoreti cally, give superior results at these discrete frequencies, Fig. 5, little help is afforded the user whose noise is not exactly synchronized to the DVM's integration period. Moreover, an ideal converter using integration is limited to a maximum sample rate of 60 Hz (and a correspond ing minimum aperture time of 16.6 ms) if the CM fre quency is exactly 60 Hz. l/10sGate(100ms) I Essentially 'infinite1 46 dB Typical Integrating DVM (100 ms Period) Fig. 5. Normal mode rejection at discrete frequencies characteristic of a typical integrating DVM. Normal mode rejection can also be achieved by passive or active filtering or by a combination of filtering and frequency conversion. A chopper stabilized amplifier is a good example. Filtering may occur before, within or after this input amplifier. All have relative advantages and dis advantages â€” none is an ideal solution to the general measurement problem. The degree of filtering required for different applications will, of course, be different as will the measurement times desired. There will always be a speed/rejection tradeoff, stated or implied. Rather than restrict the user to a fixed compromise â€” the amount of filtering and the delays that are a necessary consequence â€” filtering in the plug-ins for the Model 3480A/B is selectable (see Specifications). The user can choose the © Copr. 1949-1998 Hewlett-Packard Co. compromise that best suits his needs. Strong magnetic fields near the DVM can contribute both common mode and normal mode errors if care is not exercised in the design of the instrument. Here, filtering may be of no direct benefit, for the injected currents may be induced in the instrument's filter or in the circuitry following the filter or the integrator (if the DVM uses integration). A five gauss 60 Hz field (typically found near the primary power section of most instrumentation) can induce a peak-to-peak error as large as 30 Â¿uV if the circuitry within the field encloses an area of one square inch. Loops can be generated by the improper layout or design of either or both of the DVM's high and low leads. Shielding will be of little help if the field is large enough to cause saturation. Reducing these errors within the Model 3480A/B and its plug-ins is accomplished in several ways. All the input wiring â€” high and low â€” form tightly twisted pairs to re duce the area that may enclose the field. Where twisted pairs are impractical, compensating loops have been added. Wirewound resistors, notorious for their ability to sense magnetic fields, even when 'non-inductively' wound, have been replaced in sensitive areas by precision metal film resistors. CM and NM Sources Inside the DVM Not all common mode currents (and the normal mode errors they create) come from the circuit being measured. Some common mode error sources are generated within the measuring instrument. These are caused by currents induced into the ground or guard shields by voltages referenced to chassis, or into chassis from sources refer enced to low. These internal common mode sources are generally constant current sources and force 'injected currents' into the circuits being measured, Fig. 6. Direct measurement errors are the normal result, but by upset ting the circuit under test or by changing its characteristics during the measurement period, indirect errors often oc cur â€” these errors, because they occur only when the device under test is connected to the DVM, are often impossible to isolate and identify. The most common source of these injected currents is the instrument's power transformer. The transformer, to reduce its capacity from low (secondary) to chassis (pri mary) is guarded. Capacity from chassis to low is inter rupted by a guard shield between the two windings, Fig. 7. If the guarding is complete (C, quite small) the injected current flowing through low will be correspondingly small (if E, is 100 V at 60 Hz and d = 10 pF, the result cur rent â€” 400 nA â€” will develop 400 pV across an Rb of 1 k, Fig. 6). Injected Currents (From Transformer Primary) Fig. 6. Internally generated common mode sources ref erenced to chassis. The injected current flowing through the guard termi nal â€” caused by C2 â€” is of no consequence as it shunts the measurement circuit. If, however, the guard is connected to low, all the injected current will flow through the meas urement circuit. This error can be hundreds or thousands of times larger than the one previously calculated (C2 ^ 1000 pF). This error source can be reduced by placing an additional shield between the existing guard shield and the transformer primary winding. Other sources of similar magnitudes exist between the transformer's secondary winding and its guard shield, Figs. 7 and 8. These inject current from low to guard. This injected current will only go through Rb if the guard is properly connected; if tied to low the current is shunted around the measurement circuit. So connecting the guard to maximize CMR will also maximize the effect of this particular injected current. Connecting it to minimize the injected current will correspondingly reduce the CMR. The need for this tradeoff can be eliminated if still another shield â€” between guard and low â€” is added to the transformer, Fig. 9. The transformer construction in the Model 3480A/B incorporates all three shields to reduce all of these injected currents. Both the primary and secondary windings are completely enclosed in 'box' shields tied to their respec tive grounds. A guard shield between these two boxed windings is used as a guard to maximize CMR. In this manner injected currents are limited to a few nanoamps. Shielding requirements for the transformer must also extend to all primary and secondary wiring â€” in fact to all time varying voltages within the DVM. The primary wir- © Copr. 1949-1998 Hewlett-Packard Co. Fig. 7. A typical guarded transformer and capacitances resulting. Fig. 8. Internally generated common mode sources ref erenced to 'low.' ing is physically isolated and shielded from circuitry referenced to low. The secondary wiring and all the recti fication and regulation circuitry used for the DVM's in ternal power supplies are also shielded and isolated from chassis (the instrument's frame). Timing, gating, and external display circuitry must also be shielded (and guarded, if necessary) from the instrument chassis. Logic circuitry that generates internal timing and gating is located on a single printed-circuit card in the middle of the instrument â€” boards on either side shield it from chassis. The sample rate generator that initiates each sample is coupled to the plug-in by a steady state voltage, not one that is time varying. Time varying voltages on the 'mother board' are shielded and guarded from the chassis by another board below. This board, physically attached to the mother board, also provides the mechanical stiffening necessary to insert and extract the other plug-in cards from the mother board. Gas discharge display tubes are also isolated. Because of the differences in the glow area of the various segments (glow tube cathodes), each has a different sustaining voltage. Because of the differences in the spatial arrange ment of the segments, every unlit segment assumes a unique voltage â€” a voltage that will be dependent on the lighted segment. These voltage variations can be in excess of 40 V. When the display is changed, these voltages also change, and create large voltage transients that can gen erate large injected currents (the capacity between the Nixie segments and the instrument's chassis is relatively large because of the glow tube's large surface area). Isola tion is achieved by depositing a metallic coating (tied to low) on the inside of the plastic window that is the front of the mainframe. This conformal coating, although it does provide significant attenuation of the broadband noise generated by these display tubes, is not sufficient to reduce the injected currents to the nanoamp levels de sired. Therefore, buffering between the decoder drivers and the D/A logic has been added. The display is changed only once â€” at the completion of each reading. From a visual standpoint, buffering is not required because of the relatively small digitizing time of the A/D â€” the human eye could not detect the change in the voltage displayed during digitizing. Shielding has also been added to the board on which the D/A converter and the comparator are located. Shielding is required here because the physical spacing between the components on the board and top cover is not sufficient to reduce the injected currents from these sources to acceptable levels. These components are delib erately near the top of the instrument to provide easy accessibility to calibration potentiometers. The heat sink on the regulator card must of necessity, be tied to guard (its capacity to the instrument top cover cannot otherwise be tolerated). Here again, all time varying circuitry refer enced to low has been physically removed from the vicin ity of this shield to reduce injected currents. Time varying lines between the mainframe and plug-in are either shielded (and guarded, as necessary) or have voltage levels reduced commensurate with the injection desired. This method of guarding and shielding to minimize injected currents without the necessity of sacrificing CMR in actual use has also been used in the plug-ins, the iso- © Copr. 1949-1998 Hewlett-Packard Co. Fig. 9. Manner in which the transformer in the Model 3480A is shielded and guarded. lated digital output option (for the mainframe), and the isolated programming option (for the plug-ins) . The total injected current from all sources has been reduced to a few nanoamps â€” a level sufficient to eliminate any error when moderate unbalances are used, regardless of the guard connection. Current injection from the instrument high terminal to low will also cause an obvious error. Leakage or injected currents are dc rather than time varying (currents used to bias the input amplifier or currents from improperly shielded power supply voltages). The total input error involves not only the instrument's input resistance but also this dc leakage current. An instrument with 1012f2 input resistance may have a leakage current of 1 nanoamp. If a source resistance of 1 Mn is used, little loading error will result, but the offset caused by the leakage current will be 1 millivolt. The plug-ins for the Model 3480 A/B have an initial offset current of < 10 picoamps; its change with temperature (perhaps of more impor tance) is less than 1 picoamp/Â°C. Input impedance (and offset current) of the instrument is also constant with time or sample rate. 'Kickback' cur rents that might adversely affect measurements from rela tively high source impedances have been eliminated. ERRORS CAUSED DIRECTLY BY THE DVM DC Signal Preconditioning Conditioning the signal voltage to the nominal value required by the A/D (10 V) within the accuracies desired (Â±0.005%) can be achieved easily. It is more difficult though, if characteristics other than accurate amplifica tion or attenuation are also required. Some of the more obvious requirements are: moderate bandwidth (20 kHz @ A = 40 dB); wide dynamic range (0 V to Â±15 V at the instrument's input); extremely high input resistance ( > 1010n) ; very low offset voltage and current Â«1 Â¡J.V, 1 pA at the amplifier input); and low sensitivity to power supplies, source and load impedances, temperature and humidity. All but the requirement for bandwidth are associated with the design of any low level dc amplifier. The actual operating characteristics desired are fast recovery from overload ( >50 /is) and a slew rate and settling time fast enough to make useful the A/D digitizing time. These requirements imply a bandwidth in excess of 20 kHz. To satisfy the bandwidth requirement only is simple. But to satisfy both the requirement for dc preconditioning and for bandwidth requires greater sophistication. Chopper amplifiers (which up-convert the signal to some low carrier frequency, amplify, then down-convert) will not normally satisfy the bandwidth requirement. Such amplifiers are normally used only to amplify dc and lowfrequency voltages near dc. To amplify the higher fre quencies, an ac-coupled amplifier can be paralleled, but this is, of course, more complicated and more costly. Up-conversion to a much higher frequency carrier (mega hertz), as in a parametric amplifier, accommodates the frequency range requirement, but adequate amplifier ac curacy and dc stability (variation of the offset voltage and current at the amplifier input with time and tempera ture) are difficult to achieve. A direct-coupled amplifier of unusual design was the final choice for the instrument, satisfying performance requirements at reasonable cost. A matched pair of field-effect transistors is used for the differential input stage, Fig. 10. The FET offers both the high input resistance and the small leakage current re quired. Bipolar devices, though not necessarily limited by their lower input resistance (although lower than the FET, it is boosted by the amplifier's loop gain) have con siderably more leakage current. The FETs are operated in a balanced common drain configuration to achieve minimum sensitivities of offset voltage and gain to varia tions in power supply and device parameters. High CMR, >80 dB, is not readily obtainable in a common source configuration because of the inherent mismatch of the two discrete devices. To reduce parameter variations caused by temperature fluctuations, the FET environment is temperature con trolled at a temperature higher than the maximum ex- © Copr. 1949-1998 Hewlett-Packard Co. Bootstrap Zener (Voltage Shifter) Fig. 10. Input stage ot the Model 3482A and Model 3484A dc preconditioning amplifier. pected ambient. An integrated circuit is used as an 'oven' to maintain the FET at constant temperature. The mono lithic 1C has within it all the circuitry normally associated with an oven and its control circuitry â€” heaters, tem perature sensors, and amplification. The FET dice, mounted atop the 1C, assume the temperature of the larger chip. Although the temperature control does oper ate open loop (the sensing devices are within the 1C, not the FETs), the resultant thermal gain (AVGS without temperature control/AVGs with temperature control) can be quite high (AT > 100). A high thermal gain, however, does not guarantee a reduced offset voltage temperature coefficient. The thermal gains of the two devices must be matched because of their large initial TC (â€”600 /Â¿V/Â°C), i.e., if the two devices are ideally matched initially but AT = 100 for one side and 1000 for the other, the net TC will be 5.4 fiV/Â°C, not zero. Compensation must be used to reduce the effects of open loop control. The device is constructed with obvious symmetry about the two FET dice, Fig. 1 1 . Thermal gradients across the face of the 1C have been reduced by an anodized aluminum heat sink (0.001" X 0.030") be tween the 1C and the FET chips. The epoxy used for mounting the FETs and the 1C is thermally conductive although electrically resistive. The aluminum bonding wires (1.5 mil diameter) used for connecting the FETs are thermally bootstrapped. Such bootstrapping is re quired to minimize the effect of the heat conduction through the bonding wires â€” >60% of the heat lost. If heat is lost unevenly, the gradients that result are severe enough to seriously degrade both the absolute value of thermal gain achieved and the resulting match in thermal gain between devices. The FETs are operated at 80Â°C, a temperature high enough above ambient to allow regulation when the instrument is operated at elevated temperatures. Com pensation reduces the offset current initially to < 10 pA and attains a composite TC of <1 pA/Â°C. To keep the resulting offset current independent of input voltage level, the compensation circuitry is bootstrapped. Even though the temperature of the FET is controlled, the offset voltage temperature coefficient of the FET, combined with the rest of the amplifier, may still be greater than desired. To reduce this to <Â±1 /Â¿V/Â°C the entire amplifier is temperature compensated. The ampli fier's temperature is varied and its TC calculated. Resistive compensation is added, that is the amplifier's TC is changed and the amplifier is rerun until the desired TC is achieved. Compensation is achieved by varying the VBE match of the first bipolar gain stage. Its TC match will change by approximately 1 /xV/Â°C for each 300 /Â¿V mismatch in its base-emitter voltage. (1) Resistors are added in series with the emitters of these transistors to get the required mismatch. As the stage is driven by a current source, the voltage drops across the emitter resistors act as small batteries to mismatch AVBE. To take full advantage of the DVM's reading speed, field effect transistors are used for dc range switching. 10 © Copr. 1949-1998 Hewlett-Packard Co. FET Chips AL Strip Fig. compensation. Construction of the FETs on a chip with temperature compensation. Heaters are on the same chip. Input voltages from 10 to 1000 V are divided down to 10 V by reed relays. Ranging is then by FETs. The FET 'on' resistance, although several hundred ohms, is in series with the amplifier's input impedance ( > 10Â°n) and thus creates little error. The leakage current from the inverting side of the amplifier (and from the 'off switches) flows through the 'on' switch and, on the two lower ranges, 10 kn, Fig. 12. Even though this leakage current can be as large as 1 nA, it is relatively constant because of the controlled environment of the FET used as the amplifier input stage. The offset it creates can be removed initially by zeroing. Any changes with temperature are accounted for by the compensation technique previously described. Fig. 1 2. DC amplifier gain switching. 11 © Copr. 1949-1998 Hewlett-Packard Co. What is the HP Model 3480 A? The Model 3480A/B is a 4-digit digital voltmeter (with 50% overrange), an A/D converter with moderate speed which, when combined with one of its plug-ins, can provide multifunction measurement capability without many of the limitations created by traditional measurement errors. The 3480A/B mainframe (the 'A' is a half module; the 'B', a full rack module) uses plug-ins â€” Models 3481A, 3482A or 3484A. The Model 3484A, with all options, has five dc and true rms ac voltage ranges, and six ohms ranges. The Model 3482A has the same dc capability as the Model 3484A (it cannot, however, be expanded to provide ac and ohms). The Model 3481A has only a single dc voltage range. All plug-ins fit either mainframe configuration. Successive approximation is used for A/D conversion. Because of the design of the analog processing portions of the instrument (within the plug-in) and the means employed for data or programming transfer, reading and recording speeds up to 1000 per second are possible without per formance degradation. A true rms ac converter (an option within the 3484A) en ables accurate voltage measurements to be made of wave forms with frequency components from dc to 10 MHz. The converter eliminates significant errors (when other conver sion techniques are used) caused by small amounts of har monic distortion present in most sinusoidal signals. Accurate measurement of non-sinusoidal phenomena is also possible â€” the full scale crest factor is 7:1. Because the converter can be dc coupled, it also measures the rms value of a com bined ac and dc signal. A dual-thermopile makes the con version and is 30 times more sensitive than a single thermo couple. This sensitivity permits accurate measurements on the 100 millivolt range. DC input errors between the high and low input terminals are virtually eliminated by the combination of a constant input resistance of >10'Â° ohms and a leakage current of <10 pA. A three position input filter can be used to reduce or eliminate measurement errors caused by normal-mode noise. Errors caused by common-mode noise can be re duced by using the guard. Injected currents flowing from the low and guard terminals to chassis have been signifi cantly reduced to minimize the effect the instrument has on the device or circuit under test. System options include isolated or non-isolated BCD out puts and isolated programming inputs. Everything (except terminal selection) on the DVM is programmable. A two range, three terminal, dc ratio option is also available. The variety of possible configurations available and the innate operating features allow the user to easily adapt the instru ment to his specific needs while simultaneously reducing measurement errors. Fig. 13. FET range switch. Ql and its associated components drive Q2 'on' or 'off', Fig. 13. When the range line is open (high), Ql is off and the gate is biased to the negative supply through R4 and CR1. Grounding the range line reverses biases CR1 â€” Q2 turns on and is zero biased through R3. Acknowledgments Paul Baird and Ken lessen contributed significantly to the project's definition. George Latham was responsible for the 3480A/B's electrical design, Dave Luttropp for its mechanical design. Jim Arnold's suggestions greatly increased the instrument's serviceability. John Hettrick bore the primary responsibility for coordination of the production transfer, Gregg Boxleitner the responsibility for production and electrical design of the 3481 A. Karl Waltz and Barry Taylor were jointly responsible for the 3482A, Mike Aken for the 3484A. Jerry Blanz was responsible for the mechanical design of the plug-ins. Approbation must also be given to Larry Lopp and Jerry Harmon who contributed to the SFET's development and to Larry Linn who made many significant contributions. Too numerous to name but nonetheless indispensable are the many other people involved in layout and manufac turing. Â£ References 1 A. H. Hoffait and R. D. Thornton, "Limitations of Tran sistor DC Amplifiers," Proceedings of the IEEE, February 1964. 12 © Copr. 1949-1998 Hewlett-Packard Co. Electrical Isolation: Coupling from Low to Chassis Transformer coupling is used to transfer the information used for programming or digital output â€” from circuitry ref erenced to low, to or from circuitry referenced to chassis. This solution offers both speed and reliability â€” neither achievable with reed relays. It has one inherent disadvan tage, however. The successive approximation technique, unlike integration, presents the data in a parallel, rather than serial format. The information to be transferred is in its final form. Data transfer, then, requires a separate trans former for each bit â€” 32 for data, 15 for programming. Integrating or digitizing techniques that use voltage to frequency converters need provide transfer only for their clock pulses (a single line). Decoding is then done in the 'out guard,' or chassis section of the instrument. Program ming isolation is inherent in the reed relays typically used, so additional transfer is not necessary. The costs involved in implementing a multiple transformer scheme at first appear prohibitive. But to convert first from the parallel format to serial, transfer, and then reconvert is also costly. Serial transfer also requires clocking to main tain cogency. At the speeds deemed necessary for data transfer, the injected currents caused were considered ex cessive. The use of light isolators, although attractive, is still somewhat expensive. To reduce costs, the transformers used are simply pairs of molded RF chokes. DC isolation is achieved by mounting the chokes and their respective circuits on interfacing PC boards â€” one referenced to low and one to chassis. Power for the isolated or chassis side is provided by a separate winding on the transformer and by a separate regulator â€” both isolated from the instrument's ground (low). The basic coupling circuit is shown below. these common mode voltages may be switched at rapid rates (when used with a scanner switching large high, low, and guard voltages), the shielding and the guarding it im plies are necessary. A small capacity Â«1 pF) exists between each of the two coils used for transfer. In the example shown, a large and relatively fast common mode voltage (caused by switching) may inject enough current into the base of the transistor to cause the latch to change state. This injected current â€” caused by the external circuitry, not by the DVM â€” can be eliminated if a shield, tied to chassis, is used to interrupt the capacity between coils. If information is to be trans ferred in the opposite direction, the shield must be tied to low. Unless these shields are properly terminated, current injection will be enhanced and noise sensitivity reduced. These shields, although providing the reduced sensitivi ties desired, greatly decrease CMR. An additional shield, tied to guard, is added to obtain a net reduction in capacity. A compromise between CMR and noise sensitivity is re quired, however, as common mode voltages may also exist between guard and low or guard and chassis. As the largest voltage change allowed is between low and chassis, only shielding (no guarding) is incorporated on output lines; the resulting decrease in CMR is tolerable (the instrument's sys tem specification, regardless of option or plug-in, is 80 dB at 60 Hz with a 1 k imbalance in either input lead). Guarding of the input lines is tolerable because of the reduced voltage allowed between low and guard. Li and L; (100 /Â¿H and 220 /iH, respectively) provide a cur rent step-up of approximately 2:1 â€” low to chassis. A high current, low-voltage pulse, on transfer, is forced into the transistor's base causing saturation and a resultant change in state of the latch used to provide storage. Programming transfer is accomplished in an identical manner â€” the grounds are simply reversed. The coefficient of coupling, M, is 0.3 to 0.5. It is imperative that the programming and digital output circuitry remain insensitive to externally changing voltage; otherwise, false triggering or programming, or a change in the output data could invalidate the measurement being made. As the low and earth grounds can vary by as much as 700 V (the maximum voltage that can be tolerated from low to guard is 200 V; 500 V from guard to chassis), and as A multilayer flex circuit is used to implement the shielding and guarding between the coils. The shields are constructed in a manner that eliminates any eddy currents that could be induced in a solid sheet that would result in unacceptable coupling losses. Double shields at different potentials are offset to reduce capacitance. 13 © Copr. 1949-1998 Hewlett-Packard Co. PARTIAL SPECIFICATIONS HP Model 3480A/B (With 3481A Buffer Amplifier) HP Model 3480A/B (With 3484A Multifunction Unit) DC VOLTAGE RANGE FULL RANGE DISPLAY: Â±10.000 V. OVERRANGE: 50% PERFORMANCE ACCURACY: 90 days (25Â°C, <95% R.H.): Â±(0.01% of reading +0.01% of range) TEMPERATURE COEFFICIENT: 0Â°C to 55Â°C: Â±(0.001% of reading +0.0003% of range) per Â°C MEASURING SPEED: Reading Period: 950 /is. Reading Rate: Variable from 1 to 25 per s plus manual with front panel controls; 0 to 1000 per s with external trigger. RESPONSE TIME: 1 ms. Reads to 'within 1 count of final reading when triggered coincident with step input voltage. INPUT CHARACTERISTICS INPUT RESISTANCE: >10'Â° 0. COMMON MODE REJECTION: >80 dB, dc to 60 Hz with 1 kS2 in either lead. RANGES FULL RANGE DISPLAY: Â±100.00 mV. Â±1000.0 mV. Â±10.000 V. Â±100.0 V. Â±1000.0 V. OVERRANGE: 50% on all ranges. Â±1200 V max input. RANGE SELECTION: Manual, automatic or remote. AUTOMATIC RANGING: Upranges at 140% of range; downranges at 10% of range. PERFORMANCE MEASURING SPEED: Reading Period: 950 /Â¿s. Reading Rate (without range change): Variable from 1 to 25 per s plus manual with front panel controls; 0 to 1000 per s with ex ternal trigger. Autorange Time: Filter Out: 4 ms per range change. Filter A: 200 ms per range change. Filter B: 1 s per range change. Response Time: (without range change): Filter Out: 1 ms to within 1 count of final reading when triggered coincident with step input voltage. Filter A: 200 ms to within 1 count of final reading. Filter B: 1 s to within 1 count of final reading. INPUT CHARACTERISTICS INPUT RESISTANCE: 100 mV, 1000 mV, 10 V ranges: >1010 S2. 100 V, 1000 V ranges: 10MS2Â±0.1%. EFFECTIVE COMMON MODE REJECTION (ECMR): ECMR is the ratio of the peak common-mode voltage to the resultant error in reading with 1 kii unbalance in either lead. DC: >80 dB. AC (50-60 Hz): Filter Out: >80 dB. Filter A: >110 dB. Filter B: >160 dB. NORMAL MODE REJECTION (NMR): NMR is the ratio of the peak normal mode signal to the resultant error in reading. Filter Out: 0 dB. Filter A: >30 dB at 50 Hz and above. Filter B: >80 dB at 50 Hz and above. FILTER SELECTION: Manual or Remote. OHMS, Option 042 RANGES FULL RANGE DISPLAY: 100.00 Si 1 000.0 Q 10.000 kS2 100.00 kiJ 1000.0 kS2 10.000 MO OVERRANGE: 50% on all ranges. RANGE SELECTION: Manual, automatic or remote. AUTOMATIC RANGING: Upranges at 140% of range; downranges at 10% of range. PERFORMANCE ACCURACY: 90 days (25Â°C Â±5Â°C, <95% R.H.): 1000 Ã! thru 1000 kS! ranges: Â±(0.01% of reading +0.01% of range). 100 G range: Â±(0.02% of reading Â±0.05% of range). 10 Mfi range: Â±(0.1% of reading +0.01% of range). MEASURING SPEED: Reading Period: 950 /Â¿s. Reading Rate (without range change): Variable from 1 to 25 per s plus manual with front panel controls; 0 to 1000 per s with ex ternal trigger. HP Model 3480A/B (With 3482A DC Range Unit) RANGES FULL RANGE DISPLAY: Â±100.00 mV. Â±1000.0 mV. Â±10.000 V Â±100.0 V Â±1000.0 V OVERRANGE: 50% on all ranges. Â±1200 V max input. RANGE SELECTION: Manual, automatic or remote. AUTOMATIC RANGING: Upranges at 140% of range; downranges at 10% of range. PERFORMANCE MEASURING SPEED: Reading Period: 950 iis. Reading Rate (without range change): Variable from 1 to 25 per s plus manual with front panel controls; 0 to 1000 per s with ex ternal trigger. Autorange Time: Filter Out: 4 ms per range change. Filter A: 200 ms per range change. Filter B: 1 s per r&nge change. Response Time (without range change): Filter Out: 1 ms. Reads to within 1 count of final reading when trig gered coincident with step input voltage. Filter A: 200 ms to within 1 count of final reading. Filter B: 1 s to within 1 count of final reading. INPUT CHARACTERISTICS INPUT RESISTANCE: 100 mV, 1000 mV, 10 V ranges: >1010 fi. 100 V, 1000 V ranges: 10MÃ2 Â±0.1%. EFFECTIVE COMMON MODE REJECTION (ECMR): ECMR is the ratio of the peak common-mode voltage to the resultant error in reading with kl2 unbalance in either lead. DC: >80 dB. AC: (50-60 Hz): Filter Out: >80 dB. Filter A: >110 dB. Filter B: >160 dB. NORMAL MODE REJECTION (NMR): NMR is the ratio of the peak normal mode signal to the resultant error in reading. Filter Out: 0 dB. Filter A: <30 dB at 60 Hz and above. Filter B: <80 dB at 60 Hz and above. FILTER SELECTION: Manual or Remote. 14 © Copr. 1949-1998 Hewlett-Packard Co. Response Time (full scale step input): 100 !2 thru 100 kSi ranges (no filtering): 1 ms. Reads to within 1 count of final reading. 100 kS2 range (Filter A): 200 ms to within 1 count of final reading. 10 M'.J range (Filter A): 2 s to within 1 count of final reading. Note: may to noise generated in the unknown resistance, filtering may be required for quiet readings with inputs >100 kl.'. Response times with below are proportional less than those shown for inputs below full scale. INPUT CHARACTERISTICS VOLTAGE ACROSS UNKNOWN: 1 V at full scale on all ranges. FLAG (Print Command): Line remains 'High' during reading period. Line changes to 'Low' to Indicate completion of reading period and remains 'Low' until start of next reading period. PROGRAM FLAG (3482A Option 021, 3484A Option 041 only): Line remains 'Low' until Program is executed. Line then goes 'High' upon execution, then 'Low' after programming is completed (~1 ms). NON-ISOLATED REMOTE CONTROL Non-isolated Remote Control is standard on the 3481A, 3482A and 3484A. ISOLATED REMOTE CONTROL (3482A Option 021, 3483A Option 041) 3482A Option 021, 3484A Option 041 will operate only with 3480A/B mainframes equipped with Isolated Digital Output (Option 004). Iso lated Remote Contr.ol for the 3481A is provided in the mainframe Isolated Digital Output option. True rms ac Voltage Option 043 RANGES FULL RANGE DISPLAY: 100.00 mV. 1000.00 mV. 10.000 V. 100.00 V. 1000.0 V. DIGITAL OUTPUT OPTIONS The Digital Output Options provide measurement data outputs in digital form for printer and systems applications. In addition, input lines are included to remotely control triggering of the 3480A/B. NON-ISOLATED DIGITAL OUTPUT, 3480A/B Option 003 Non-isolated Digital Output is available both as a factory-installed option (3480A/B Option 003) and a field installable accessory (HP 11147A). GENERAL POWER: 115 V or 230 V Â±10%, 50 Hz to 400 Hz, 60 W max (including plug-in, options, normal environmental conditions). INPUT TERMINALS: High, Low and Guard terminals on both front and rear panels of 3481A, 3482A and 3484A. Front/Rear selector switch on front-panel of plug-in. High and Low Ratio Reference Input terminals on 3480A/B rear panel. Low Ratio and Low Input termi nals are electrically common. WEIGHT: 3480A Basic Instrument: 11 Ibs, 12 oz (5,25 kg) Including Options: 12 Ibs, 8 oz, (5,7 kg) Shipping: 17 Ibs, (7,75 kg) 3480B Basic Instrument: 12 Ibs, 12 oz (5,71 kg) Including Options: 13 Ibs, 8 oz, (6,15 kg) Shipping: 18 Ibs, (8,1 kg) 3481A Net Weight: 2 Ibs, 11 oz (1,2 kg) Shipping: 5 Ibs, (2,3 kg) 3482A Basic Instrument: 4 Ibs, (1,8 kg) Including Options: 4 Ibs, 4 oz, (1,9 kg) Shipping: 7 Ibs, (3,15 kg) 3484A Basic Instrument: 4 Ibs, 6 oz (1,97 kg) Including All Options: 6 Ibs, 2 oz, (2,76 kg) Shipping: 8 Ibs, (3,6 kg). ACCESSORIES AVAILABLE: HP 11148A Plug-In Extender Cable for servicing all plug-ins.. $ 45 HP 11149A Remote Control Cable for all plug-Ins $ 25 The following accessories add optional capabilities not included with not basic instrument. Optional capabilities which are not listed as accessories can be ordered only at the time of initial purchase. The Isolated Remote Accessory, HP 11 151 A, can be used only when the 3480A/B has the Isolated Digital Output Option 004, which is not available as an accessory. HP 11147A Non-isolated Digital Output for 3480A/B $200 HP 11151A Isolated Remote Control for 3482A 3 4 8 4 A ( r e q u i r e s 3 4 8 0 A / B O p t i o n 0 0 4 ) $ 2 0 0 H P 1 1 1 5 2 A O h m s C o n v e r t e r f o r 3 4 8 4 A $ 2 0 0 H P 1 1 1 5 3 A A C C o n v e r t e r f o r 3 4 8 4 A $ 8 0 0 PRICES: H P 3 4 8 0 A V 2 M o d u l e M a i n F r a m e $ 8 0 0 H P 3 4 8 0 B F u l l R a c k W i d t h M a i n F r a m e $ 9 0 0 Main Frame Options: O p t i o n 0 0 2 D C R a t i o $ 2 0 0 O p t i o n 0 0 3 D i g i t a l O u t p u t $ 2 0 0 O p t i o n 0 0 4 I s o l a t e d D i g i t a l O u t p u t $ 3 7 5 HP 3481A Buffer Amplifier (includes Single Range DC Voltage a n d N o n - i s o l a t e d R e m o t e C o n t r o l ) $ 3 5 0 HP3482A DC Range Unit (includes 5 Range DC Voltage and N o n - I s o l a t e d R e m o t e C o n t r o l ) $ 7 0 0 Option 021 Isolated Remote Control (requires Main Frame with Option 004, HP 11149A Remote Cable furnished) $200 HP 3484A Multifunction Unit (includes 5 Range DC Voltage and N o n - I s o l a t e d R e m o t e C o n t r o l ) $ 9 0 0 Option 041 Isolated Remote Control (requires Main Frame with Option 004. HP 11149A Remote Cable furnished) $200 O p t i o n 0 4 2 O h m s C o n v e r t e r $ 2 0 0 O p t i o n 0 4 3 T r u e R M S A C C o n v e r t e r $ 8 0 0 MANUFACTURING DIVISION: LOVELAND DIVISION Loveland, Colorado 80537 OVERRANGE: 50% on all ranges. 1500 V peak max input. RANGE SELECTION: Manual, automatic or remote. AUTOMATIC RANGING: Upranges at 140% of range; downranges at 10% of range. PERFORMANCE MEASURING SPEED: Reading Period: 950 jus. Reading Rate (without range change): Variable from 1 to 25 per s plus manual with front panel controls; 0 to 1000 per s with ex ternal trigger. Response Time (full scale step input, without range change): AC Coupled: 1 s to within 5 counts of final reading. DC Coupled: 15 s to within 5 counts of final reading. INPUT CHARACTERISTICS INPUT RESISTANCE: 2 Mi! Â±1%. CREST FACTOR: 7:1 at full scale. 70:1 at 10% of full scale. GENERAL SPECIFICATIONS Mainframes, Plug-ins and Options DC Ratio 3480A/B Option 002 DISPLAYED RATIO: Display in all functions is proportional to the ratio of the input voltage to the external 10 V dc reference voltage ap plied to rear-panel Ratio terminals. ACCURACY (with respect to external reference voltage): 10 V or 100 V Â±5% external reference: Same as basic instrument accuracy specifications. 10 V, 100 V +5% to +35% or 10 V, 100 V -5% to -13%: Add Â±0.02% of reading to basic instrument accuracy specifications. INPUT CHARACTERISTICS (ratio reference terminals): INPUT VOLTAGE: +10 V or +100 V (referenced to Low side of measurement). INPUT RESISTANCE: 10 V Ratio Range: 100 k!2 Â±1.5%. 100 V Ratio Range: 100 kG Â±0.5%. RATIO MEASUREMENT SELECTION: Manual or Remote. RATIO RANGE SELECTION: Manual. REMOTE CONTROL Remote controls are selected by application of a 'Low' state (logical '0') to the remote lines through a rear-panel connector. REMOTE CONTROL LINES ENCODE (external trigger): Initiates a measurement period. Actu ated by application of 'Low' state for >50 /is. Line must be in 'High' state >50 tis before applying 'Low' state. Minimum time between ENCODE commands: 1 ms. INHIBIT (Interface Hold): Disables front-panel Sample Rate control. RATIO SELECT (Non-isolated Remote Control only): Selects Ratio Measurement (if mainframe has the Ratio option). FILTER SELECT (3482A, 3483A only): Selects Filter A or Filter B; one line per filter. RANGE SELECT (3482A, 3484A only): Selects measurement range; one line per range. FUNCTION SELECT (3484A only): Selects measurement function; one line per function. PROGRAM (3482A Option 021, 3484A Option 041 only): Accepts program commands when 'Low' state is applied for >50 us. Prevents changes in previously selected program when 'High' state is applied for >50 /is. Does not affect operation of ENCODE line. A minimum of 1 ms must be allowed between PROGRAM and ENCODE commands. 15 © Copr. 1949-1998 Hewlett-Packard Co. H. Mac Juneau â€¢ Mac received his BSEE from â€¢ Swarthmore College in 1961 I and his MSEE and Ph.D. from I the University of Minnesota in 1 1965 and 1967 respectively. After graduation Mac came to Hewlett-Packard and worked on ac converters for DVM's, a job â€” which has kept him occupied for three years. Outside working hours, Mac spends his time woodcarving and welding. Lee Thompson I Lee Thompson received his BSEE degree from the University ^M^^_ aaÃ¯. oÃ- Texas at Austin in 1966. f He joined HP's Loveland Division P ^ J that same year as a product designer. Lee did the product design and some circuit design Jy ^3^,^ on the rms converter in the HP 3450A before becoming involved with the true rms J c o n v e r t e r i n t h e H P 3 4 8 4 A , w h e r e he has worked primarily on the amplifier design. * Lee received his MSEE degree from Colorado State University in 1968 as a participant in the HP Honors Cooperative Program. He is a member of Tau Beta Pi. Craig Walter L Craig earned a BSME from Stanford University in 1961, then continued at graduate school while working part time for Hewlett-Packard. After receiving his MSEE in 1963, Craig joined the HP Loveland Division. In 1968 he assumed responsibility for the four-digit DVM program. Craig's interests include sailing, bridge and wood working. He is also an ardent sports enthusiast. HEWLETT-PACKARDJOURNAL<â€¢"'JANUARY1971volume22â€¢Number5 TECHNICAL U.S.A. FROM THE LABORATORIES OF HEWLETT-PACKARD COMPANY 1501 PAGE MILL ROAD. PALO ALTO, CALIFORNIA 94304 U.S.A. Hewlett-Packard S. A. 1 21 7 Meynn - Geneva, Switzerland â€¢ Yokagawa-Hewlett-Packard Ltd., Shibuya-Ku, Tokyo 1 51 Japan Editor: R. Assistant: Jordan Editorial Board: R. P. Dolan. H. L. Roberts, L. D. Shergalis Art Director: Arvid A. Danielson Assistant: Maridel Jordan © Copr. 1949-1998 Hewlett-Packard Co.
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