A GPS-Based Frequency Standard
By Brooks Shera, W5OJM
A GPS-Based Frequency
Standard
This modern and highly accurate
frequency standard is something
you can readily have!
requency accuracy has been a
topic of special interest to many
amateurs and experimenters
since the early days of radio.
Until recently, the best frequency standard
available to most hams was a crystal oscillator carefully adjusted to zero-beat with a
station of known frequency, such as WWV.
Unpredictable variations in ionospheric
propagation make achieving high accuracy
by this method an art as well as a science.
In fact, until 1981, the ARRL sponsored
frequency measuring tests. 1 Only the best
entries were closer than 0.1 ppm. This is
plenty good enough to keep your transmitter within the HF band edges, but for applications such as EME work and VHF weaksignal detection, it leaves something to be
desired.2
Today, the potential for accurately measuring frequency is vastly improved. We
now have more than two dozen atomic
clocks constantly circling the Earth and
typically six or more of these are in line-ofsight view all the time! I refer, of course, to
the GPS satellites kindly provided by the
Department of Defense (DoD). The spreadspectrum signals from these satellites are
very weak and quite complex. However, the
hard work necessary to receive and process
these highly accurate frequency signals has
already been done by the folks that design
and build the little GPS navigation receivers that are now available for less than the
price of a hand-held transceiver.
These GPS receivers process the signals
from four or more satellites, and once every
second they compute latitude, longitude and
altitude. To make this computation, the receiver must also compute the current time
with very high accuracy. As a side benefit of
the position computation, many receivers
output a timing pulse once each second for
the use of people who like to know exactly
what time it is.
Perhaps we can use this once-per-second pulse, which is derived from the orbiting atomic clocks and is typically accurate
to a few tens of nanoseconds (ns), to con-
F
1
Notes appear on page 43.
trol (discipline) the frequency of our Earthbased frequency standard in much the same
way that previous generations of hams have
manually adjusted crystal oscillators to
WWV. If so, we might expect to achieve an
accuracy of perhaps a few parts in 1011,
about 10,000 times better than the timehonored WWV zero beat method!
The challenge of this project is to use
the GPS timing pulse—which occurs only
once per second—to control the frequency
of a crystal oscillator that vibrates perhaps
five million times a second. The direct approach I chose was to use a phase-locked
loop (PLL). Because of the long time constants involved, I built the loop using digital—rather than the traditional analog—
components.
Circuit Description
As shown in the block diagram of Figure 1, the device consists of five sections:
a commercial GPS receiver, a voltage-controlled crystal oscillator (VCXO), a phasemeasuring circuit, a microprocessor (CPU)
with a few interface chips, and a digital-toanalog converter (DAC) to control the
VCXO. Figure 2 shows the controller
board, which includes almost everything
except the GPS receiver and the VCXO.
Figure 3 is the schematic of the controller.
VCXO
A stable, temperature-controlled oscillator is desirable because we will rely on it
to keep the output on frequency between
GPS pulses. Good oscillator stability also
helps us overcome the slight jitter in the
time of the GPS pulses that is purposely
introduced by the DoD to reduce the navigational accuracy of GPS for nonmilitary
users. High-quality VCXOs are widely
used in cellular-telephone transmitter sites.
These VCXOs typically oscillate in the 5 to
10 MHz range and are often available on
the surplus market. Excellent temperaturecontrolled crystal oscillators have also been
used for many years as components in better-quality frequency counters. One example is the HP 10811 crystal oscillator
subassembly, which is often available at a
reasonable price.3 The controller circuit can
accommodate any stable oscillator in the
range of 0.4 to 10 MHz, but be certain you
July 1998
37
Figure 1—Block diagram of the GPS-based frequency standard.
get a unit that has a voltage-control input
that can vary the frequency over a narrow
range. In the following discussion, I will
assume we are using a 5 MHz VCXO.
Phase Detector
To control the VCXO, we need to measure the phase difference between its output
and the 1 pps GPS pulses. This can be done
with less ambiguity if the measurement is
done at a frequency lower than 5 MHz, say
300 kHz. Therefore, after amplification and
buffering by U1 (see Figure 1), the 5 MHz
signal from the VCXO is divided by 16 in a
4-bit counter, U2A, to produce an output
near 312 kHz. The exact frequency is unimportant. It is only necessary that it be phaserelated to the VCXO. The phase difference
between the GPS and the VCXO is measured
by counting the number of pulses (U2B/U4)
from a separate, garden-variety 24 MHz
crystal oscillator (U7) that occur during the
time between the GPS pulse and the next
VCXO pulse from U2A. Each count of the
24 MHz signal indicates a phase difference
of 42 ns. If this count is constant, we know
the phase difference is constant, and thus,
that our LO is synchronized to the GPS
atomic clocks.
Interestingly, it is desirable to have the
frequency of U7 drift slightly rather than
being synchronized with the VCXO. A
slight random drift averages out the ±1
count ambiguity that is inherent in any
pulse-counting device. My measurements
indicate that the simple phase-measuring
circuit I use is consistently accurate to 2 or
3 ns (for a 30-second measurement), while
without drift, the resolution would be limited to 42 ns. The $5 crystal oscillator mod38
July 1998
ule drifts adequately and also provides a
divided 6 MHz output to clock the CPU.
DAC
It is the microcontroller’s (U8) job to
read the count from U2B/U4 and adjust the
control voltage applied to the VCXO to keep
this count constant. U8 controls the VCXO
by sending a message to the DAC (U9). The
DAC I chose is a relatively low-cost 18-bit
unit designed for digital audio applications
such as high-quality CD players. It can control the VCXO over a range of ±3 V in steps
as small as 23 µV. The DAC output voltage
is attenuated by the resistor network R6 and
R5 before it is applied to the VCXO control
input. The resistor network serves to further
decrease the voltage step size and provides
a convenient way to adapt the controller
circuit to VCXOs that have different control-input sensitivities.
CPU
The CPU is the brains of the controller
and its interfacing and software dominated
the design effort that I devoted to this
project. The inexpensive PIC16C73 micro I
chose is quite versatile but, like most small
micros, it has a limited number of input/output pins. Therefore, I have used serial communications between the PIC and the other
controller components. The ’16C73 is well
suited for such communications because it
includes two built-in serial ports. One of
these ports is used to send ASCII messages
to an external PC so the performance of the
VCXO and the PLL can be monitored. (This
ASCII port also made debugging the hardware and software much easier.)
Serial communication is also used to
read the phase count value from U2A/U4.
Whenever the CPU needs to get the current
phase, it sends a load pulse to the parallelin/serial-out shift registers, U5 and U6.
This pulse loads the count data from U2A/
U4 into the shift registers, and a reset pulse
is then sent to clear the counters so they
commence to count from zero when the
next GPS pulse arrives. The CPU uses its
synchronous serial port to upload the phase
data from the shift registers.
The PCM-61A DAC (U9) is also a serial-input device. The CPU’s software manipulates the three output pins that are connected to the DAC to download the data
one bit at a time. After a full 18-bit data
value is loaded, the DAC automatically
latches this value into an internal register
and sets its output voltage accordingly. The
voltage is held indefinitely, or until a new
data value is loaded. The voltage-hold feature makes it possible to arrange the DAC
and the VCXO as a separate detachable unit
that is connected to the controller only
when the VCXO needs to be set on frequency. After setting, the VCXO-DAC unit
can be moved to wherever a precise frequency reference is needed.
Two more bits of hardware should be
mentioned since their appearance in the circuit may be confusing. The controller uses
two ’4046 PLL ICs, but neither of them is
used for their designed purpose. U1 is used
only as a sensitive, high-input-impedance
amplifier section to buffer the VCXO, while
U3 buffers the GPS input line and supplies
a fast RS flip-flop that gates the phase
counter. The 4046 is a readily available,
inexpensive IC that has several uses and is
worth exploring. 4
Software
As mentioned earlier, the primary task
of the CPU is to monitor the phase difference between GPS and the VCXO and respond appropriately. If this phase difference begins to drift, the CPU must make a
correction to the VCXO frequency. The
tricky part is to make a correction of the
right size. If the correction is too large, the
VCXO frequency will consistently overshoot the mark and jitter around the correct
value. Conversely, too small a correction
will cause a sluggish, overdamped response. The CPU should also smooth over
the small second-to-second and minute-tominute GPS timing fluctuations, while giving GPS full control of the VCXO frequency in the longer term. This is basically
a filtering function, so we need to design
software that makes the CPU act like a lowpass digital filter.
Fortunately, design help is available.
The theory of analog filters in feedback
control systems is described in several reference books, 5 and methods have been developed to implement these designs as digital-computer algorithms. 6 Our feedback
control and filtering problem is another aspect of digital-signal processing (DSP). You
might reasonably wonder why I chose a little
8-bit microcontroller instead of a DSP processor if we need to do DSP. There are several reasons. First of all, high CPU speed is
not required because phase values come
only once per second and the program averages 30 seconds of phase data together before computing a new DAC setting. Half a
minute gives plenty of time to compute almost anything that could be needed. The 8bit word size of the microcontroller also
proved to be no limitation. The software
does highly precise arithmetic simply by
stacking five consecutive 8-bit words to
form a 40-bit integer. Software routines do
all the usual arithmetic operations on these
40-bit words. Lastly, perhaps the most important factor in my CPU choice is that the
PIC micro is a lot of fun to program!
The feedback filter I have programmed
is the digital equivalent of what control
systems specialists call a PI filter (proportional integral filter—its response is proportional to the input signal, plus its time
integral). You can make an analog version
of a PI filter with a high-gain op amp, two
resistors and a capacitor, but you might
have trouble getting the long integration
time we need in this application. The PI
filter has several useful and interesting
properties. First, because it integrates the
input signal, it provides low-pass filtering.
Moreover, after a period of operation, the
filter learns the average drift rate of its input signal and automatically adjusts its
output to correct for it! In our case, such a
linear drift might result from a steady
change in ambient temperature, or the aging of the VCXO crystal. The PI filter is
programmed as a second-order infinite
impulse response (IIR) filter, which has the
Figure 2—The double-sided controller board measures 21/ 2×51 / 2 inches.
advantage that it requires only a few arithmetic operations and needs to store only
three data values: the current input and the
previous input and output values.
The software provides six different filter time constants, ranging from no filtering at all, to a time constant of many hours.
The time constant—which is chosen by
setting S1 through S3 of U11—can be
changed on-the-fly while the controller is
running without causing a transient that
might temporarily perturb the VCXO frequency.7 The usual procedure is to start
with no filtering, which permits the controller to quickly lock the VCXO to GPS
and then, over a period of time, select progressively longer constants. The ultimate
selection is a balance between reduction of
GPS jitter (which needs a long time constant) and elimination of medium and longterm VCXO drift (which favors a shorter
time constant). The longest time constants
are suitable only for very stable oscillators,
such as high-quality oven-controlled crystals and Rubidium (Rb) atomic oscillators.
Although Rb oscillators are very stable,
even they can be improved by locking them
to the GPS Cesium clocks.
In addition to implementing a versatile
digital filter, the software provides other
useful functions. Each new phase value is
checked for consistency against the immediately preceding values by a deglitching
algorithm. Momentary phase jumps are discarded before they can affect the VCXO
frequency. The CPU also continually monitors the status of the PLL and reports potential error conditions via the front-panel
LEDs and the ASCII port, as described in
the following sections.
Construction
PC boards are available from A&A Engineering (see Figure 3 caption). In many
applications, the controller will run essentially unattended for days, weeks, or
months, so it is important to use reliable
and safe construction practices and ad-
equate power-supply fusing. A metal enclosure and shielded I/O leads protect the
unit from nearby RF fields and reduce EMI
from the digital signals generated within.
Use IC sockets. I installed a digital voltmeter module on the front panel of the prototype to monitor the DAC output voltage.
The meter provides a convenient operational check and indicates when VCXO
aging may require a manual adjustment of
the frequency pot to keep the DAC within
its ±3 V operating range. (A test point for
use of an external voltmeter can be substituted.)
Setup
VCXO
The circuit can accommodate most any
stable VCXO that you want to use. The price
of this flexibility is that a few setup steps
are required. First, set S4 (U11, pin 4) to
correspond to the polarity of the VCXO
control voltage. If the VCXO frequency increases when the control voltage increases,
open S4. If the frequency decreases with
increasing voltage, close S4. Install a
jumper to connect U3 pin 3 to one of the
four output pins of U2A. U2A divides the
input VCXO frequency in binary steps from
2 to 16. Select a pin that provides a VCXO
output frequency in the 150 to 700 kHz
range. Solder pads are provided at the edge
of the PC board to make jumper installation
easy. Third, the values of R6 and R5 must
be selected 8 to match the sensitivity of the
control voltage input of your VCXO. The
goal is to obtain a relative frequency
change, ∆F/F = 7.5 × 10–9, when the voltage
applied by the DAC to R6 changes by 1 V.
Lastly, you should check the values of the
input resistors R1 and R2. The values given
in Figure 2 should work fine with most
VCXOs because they were selected to provide a fairly high input impedance to avoid
loading the VCXO output. If you change
the value of these resistors, keep in mind
that the peak-to-peak voltage at U1 pin 14
July 1998
39
40
July 1998
should be in the range 50 mV to 2 V. If your
VCXO provides a low-impedance TTL output, bypass the input network (including
C1) and connect directly to U1 pin 14.
GPS Receiver
Several GPS receiver models are available that provide a 1-pps timing pulse. This
controller was developed using a Motorola
PVT-6, which has been recently superseded
by the Motorola Oncore VP. The Garmin
GPS-25 and the Trimble SK-8 also provide
1 pps pulses.9 The best timing results are
usually obtained when the receiver is used
in “position hold” mode. Basically, the idea
is to let the receiver determine its location,
then manually lock that location into the
software so that timing is the only variable
the receiver needs to consider. Consult your
receiver manual on how to set up this mode.
The receivers typically provide a standard
TTL signal that can be connected directly
to the input at U3 pin 14, however the discussion earlier regarding the VCXO input
network also applies here since both inputs
feed 4046s. U3 triggers on the positive rising edge of the input and the circuitry assumes that this edge provides the GPS time
reference. The GPS receiver’s CPU clock
introduces a granularity in the timing pulse
output, but the effect of this is greatly reduced—along with jitter from other
sources—by the 30-second averaging and
by the low-pass filter.
Controller
Figure 3— Schematic of the controller. Unless otherwise specified, resistors are
1
/ 4 W, 5% tolerance carbon-composition or film units. Equivalent parts can be
substituted. The controller is constructed on a double-sided PC board measuring
approximately 2.75×5.25 inches. A controller PC board is available from A&A
Engineering, 2521 W La Palma Ave Unit K, Anaheim, CA 92801; tel 714-952-2114,
fax 714-952-3280; stock no. 217-PCB, $19.95 plus $1.50 shipping and handling per
order; foreign orders, shipping and handling is 15% of the total order price for
surface mail. Programmed PICs are available from the author for $22 each plus $3
shipping in the US and Canada, $7 elsewhere. The source code for the controller
software is available from the author, see Note 14. All the controller parts are
available from Digi-Key Corp, 701 Brooks Ave S, Thief River Falls, MN 56701-0677;
tel 800-344-4539, 218-681-6674, fax 218-681-3380 http://www.digikey.com and
Radio Shack. For part numbers in parentheses, DK = Digi-Key; RS = Radio Shack.
C1-C12, C150.1 µF, 50 V polystyrene
(DK P4593)
C13, C14—47 µF, 16 V electrolytic
(DK P1210)
DS1-DS3—LED (RS 276-069)
U1, U3—74HCT4046 (RS 276-2913)
U2—74HCT4520 (DK CD74HCT4520E)
U4—74HC4040 (DK MM74HC4040)
U5, U6—74HCT166 (DK 74HCT166E)
U7—24 MHz oscillator module ECS300S-24 (DK XC-313)
U8— Programmed PIC 16C73A; unprogrammed devices are available from Digi-Key
(DK PIC16C73A-20/SP)
U9—PCM61P (DK PCM61P)
U10—MAX233 (DK MAX233CPP)
U11—8-pole DIP switch (DK A5208)
U12—LM2940 (DK LM2940T-5.0)
U13—LM2990 (DK LM2990T-5.2)
U14—47 kΩ, 9-resistor SIP (DK Q9473)
Misc: PC board (see Note 14), enclosure (Elma type 33 [DK 260-1001] used here),
optional voltmeter (DK JDPM601; less expensive units will suffice), IC sockets,
connectors and hardware.
In addition to the six time constants
mentioned earlier, the controller provides a
setup mode to help you make a coarse adjustment of VCXO frequency that places
the desired lock frequency near the center
of the voltage control range. Depending on
the VCXO used, the manual adjustment
could be a trimmer capacitor or a controlvoltage trim pot. The controller’s LED status lights HIGH and LOW indicate when the
frequency is too high or too low. Select
setup mode (U11 S1, S2, S3 all closed) and
carefully adjust the VCXO frequency until
the two high/low lights are extinguished.
Small adjustments and a little patience are
needed because the status lights are updated
only at the 30-second intervals when the
phase-measuring circuit produces a new
reading. When the lights remain off for
several minutes, switch to run mode and
proceed to progressively lengthen the time
constant as described later. A third status
light (HEARTBEAT) blinks once per second
to indicate that the GPS pulses are being
received and that the controller CPU is running. It may be helpful, although not essential, to monitor the controller’s performance by connecting a PC to the controller’s ASCII port as described next.
Performance
The controller is programmed to send
the GPS-VCXO phase-difference data (averaged over 30-second intervals) to its
July 1998
41
Figure 4—Locking an LO to the GPS clocks. Initially, the phase between GPS and the
oscillator changes with time, indicating that the oscillator is slightly off frequency. After
several hours, the controller feedback loop was closed and the oscillator becomes
locked to the GPS atomic clocks!
Figure 5—Dynamic behavior of the controller. Rapid response and absence of ringing are
displayed when the controller is forced to correct a manual frequency offset (see text).
42
July 1998
ASCII output where it can be captured by a
PC for analysis. 10 Data from the ASCII port
is quite revealing, especially when accumulated over a period of a few hours, and
with the PLL feedback loop both open and
closed. 11 Even when the PLL is locked, the
phase data shows random noisy fluctuations from one 30-second averaging period
to the next that have a standard deviation of
about 35 ns. I call this “GPS jitter,” and its
primary cause is presumably the intentional
DoD dithering of the GPS signals. 12 It is not
evident when I monitor the phase difference between the VCXO and a second
stable crystal oscillator that has been divided to generate 1 pps pulses. If this jitter
is passed on directly to the VCXO by the
PLL, it would induce short-term frequency
shifts of about one part in 10 9 . Although
such performance isn’t too bad, it can be
improved by low-pass filtering. How much
filtering improvement can be achieved depends on the VCXO stability, which itself
can be estimated from open-loop phase
observations. As programmed, the loop filter has a slope of 6 dB/octave, so each doubling of the time constant cuts the GPS jitter in half. The maximum filter attenuation
is about 40 dB, but as mentioned earlier,
this can only be used effectively with a
highly stable oscillator. A good oven-controlled VCXO should be able to operate
with a time constant that will attenuate the
jitter by at least 30 dB, and should yield a
frequency accuracy of a few parts in 10 11.
The drift rate of a high-stability oscillator is conventionally specified by computing the Allan variance (AV). The AV is a
measure of the fractional difference in frequency we would expect to observe if two
measurements of the oscillator frequency
were made separated by a time, T. If the
time between the two measurements is long,
the AV of our disciplined oscillator will be
essentially the same as the long-term stability of the GPS atomic clocks. A typical value
for T = 24 hours would be AV = 2×10–13 . If
the time between the measurements is
shorter—in the range of a few minutes to a
few hours—the AV is determined by the
stability of our VCXO and the CPU filter
time constant. For a good-quality VCXO,
we can expect to obtain an AV of perhaps
5 × 10 –12 for T = 10 min. 13
The best way to judge your disciplined
VCXO is to compare it with a hydrogen
maser, but lacking that, a lot can be inferred
from the ASCII data provided by the controller. To select the time constant that
gives the optimum performance from a
particular VCXO, attach a PC to the
controller’s ASCII port and record the data
for awhile. Locking an oscillator to a shortterm-noisy, long-term-accurate reference
like GPS is a relatively unexplored field. I
encourage you to experiment. My assembly-language PIC software is available as a
starting point if you want to try different
filtering techniques. 14 The code is highly
commented and the free software program-
ming tools you need are available on the
Internet. 15
Figure 4 is an illustration of phase-locking to the GPS signal. Initially, the VCXO
was allowed to run open-loop, ie, not under
GPS control. Then, after several hours, the
loop is closed. The plot shows the phase
difference (expressed in time units) between the VCXO and the GPS pulses. The
phase difference slowly decreases, indicating that the VCXO frequency is a little too
high. From the slope of the plot, we can
estimate that it is too high by about two
parts in 10 9 , or 0.01 Hz for our 5 MHz
VCXO. Note that the plot is quite linear,
indicating that although the frequency is
off a bit, it is quite constant—the trademark
of a stable VCXO. Note also the noise on
the plot caused by GPS jitter. After about
5.5 hours, I flipped U11 S5 to allow the
DAC to control the VCXO thereby closing
the loop. After a short transient, the plot
abruptly becomes horizontal indicating that
the phase difference is now constant and
that the VCXO is locked to the GPS atomic
clocks. It was an exciting moment when I
first watched this happen!
The closed-loop transient response of
the controller is shown in Figure 5. The
upper plot shows phase difference data like
that of Figure 4, and the lower plot is the
DAC output voltage fed to the VCXO. The
two curves represent the input and output
of the digital filter. About three hours into
the run, I manually offset the frequency of
the VCXO by 0.035 Hz, forcing the controller to make a large correction. The phase
difference begins to increase and the controller responds by decreasing the DAC
output. After about 45 minutes, the transient is complete and the VCXO has been
disciplined back onto frequency as shown
(upper plot) by the phase difference returning to the setpoint. The lower plot shows
the DAC voltage output that was needed to
correct the manual frequency offset. A
shorter filter time constant would make the
response faster, but at the cost of greater
GPS jitter. After about 14 hours, I removed
the manual offset and the transient is repeated, but with the opposite polarity. The
clean response of the controller and the
small nearly optimal overshoot is evident
for both transients.
Figure 6 shows the DAC output voltage
during normal operation of the controller.
The data were accumulated over a period of
several days by recording the output from
the ASCII port. Several interesting things
can be observed. There is a slow decrease
in voltage, which corrects the long-term
aging of the VCXO crystal. Converting the
VCXO input voltage to relative frequency
indicates that the aging process is about 7 ×
10 –12 per day; which is typical of a good
quality VCXO that was manufactured a few
years ago and has had a while to stabilize.
Superimposed on the slow decrease is a
nearly sinusoidal 24-hour variation caused
by a day-night temperature shift of about
Figure 6—Normal operation of the controller. A day-night temperature effect and the
long-term aging of the VCXO crystal oscillator are evident in this plot of the correction
voltage applied to the VCXO by the controller during normal operation. The controller
adjusts the VCXO input voltage as needed to eliminate the frequency shifts that would
otherwise occur if there were no GPS stabilization. The data point scattering is primarily
caused by residual GPS jitter that is present even after low-pass filtering. The solid line
is a curve fitted to the data to obtain the aging and temperature parameters.
4 ° C in my workshop. The data indicate a
temperature coefficient of about 4 × 10 –12
per oC, a respectable value for an ovenized
crystal. The GPS phase lock has prevented
these normal aging and temperature effects
from significantly affecting the VCXO frequency. GPS jitter is also quite evident as
noise on the plot. The RMS jitter amplitude
has been reduced from about 35 ns at the
filter input to about 0.5 ns equivalent at the
output. Still, the noise dominates the plot
and suggests that an even-longer time constant could be used with this VCXO to further improve the short-term stability of the
output frequency.
Figure 6 also suggests that two major
causes of frequency instability—temperature shift and aging—could be predicted and
largely eliminated by tracking the performance of the VCXO for a while to estimate
the aging parameters and by measuring the
ambient temperature. The predicted corrections could be applied to the VCXO independently of the PLL, which might allow
much longer loop filtering time constants to
be used, further reducing GPS jitter. Although this scheme would be ultimately limited by sources of crystal frequency instability that are random and inherently unpredictable, it might be interesting to explore.
Only minor additions to the hardware would
be needed since the PIC microprocessor has
an unused ADC input that is available for
temperature monitoring.
A Demonstration
As a demonstration, I tested the controller by disciplining the master oscillator (a
10811) in an HP 5328A frequency counter
(see the title photo). The 1-MHz frequency
output on the rear panel of the ’5328A was
connected to the controller VCXO input
and a pair of wires was added to carry the
DAC feedback voltage to the master-oscillator board inside the HP counter. After
only a few moments of operation, and a
little adjustment of its master-oscillator
frequency pot, the HP5328A was locked
onto the GPS. The result was a counter with
an accuracy and stability far surpassing
anything that could have been imagined by
Hewlett-Packard for the 5328A when they
built this counter nearly 20 years ago.
Notes
1
The end of the almost 50-year span of ARRLsponsored frequency measuring tests was
announced in “Operating News,” QST , Dec
1981, p 99—Ed.
2
For an interesting discussion of ultra-weak signal detection and the importance of high frequency accuracy, see the article by Darrel
Emerson, AA7FV “The Mars Global Survey
Relay Test,” AMSAT Journal, Volume 20, No 1,
Jan/Feb 1997, or visit Darrel’s Web site at
http://www.tuc.nrao.edu/~demerson/
marsspec/marsspec.htm. SETI information
can be found at http://seti1.setileague.org
and interesting EME work has been reported
by Michael Cook, AF9Y (http://www.
webcom.com/af9y).
3
A possible source of HP10811s is Gary
Glassmeyer, N6ZD, at Certified Used Test
Equipment, Lockwood CA; tel 408-385-0301;
also Douglas Dwyer e-mail: ddwyer@ddwyer.
demon.co.uk and Joakim Langlet, e-mail
joakim@seaview.se, have VCXOs available.
For used Rb oscillators, try Wade Lehman,
http://www.lehman.scientific.com.
4
For more information about the 4046, see Neil
Heckt, “A PIC-Based Digital Frequency Display,” QST, May 1997, pp 36-38.
5
A good introduction to PLLs and digital filters
appears in Chapters 14 and 18 of The ARRL
Handbook, 75th ed, 1997. The excellent and
classic work in the field is F. M. Gardner,
Phaselock Techniques, (New York: John
Wiley & Sons, second ed, 1979).
July 1998
43
6 Digital
PLLs are discussed by C. L. Phillips
and H. T. Nagle, Digital Control System
Analysis and Design , (Englewood Cliffs:
Prentice-Hall, Inc third ed, 1995), and by R.
E. Best, Phase-Locked Loops, (New York:
McGraw-Hill, 3rd ed, 1997); watch out for typos in the equations). The MatLab software
package (MathWorks, Inc, Natick, MA; tel
508-653-1415, has a digital filter package that
is useful for checking designs. A student edition of the software is available.
7 Switches S1 through S3 of U11 can be considered as controlling a three-bit number, N, in
the range of 0 to 7, where closed is 0 and
open is 1. Then, N = 0 is setup mode, N = 1
implements a first-order PLL with no filtering
beyond the 30-second integration of the
phase measuring circuit, and N = 2 through 7
implement second-order PLLs with time constants, τ, starting at 1500 seconds and increasing by a factor of two at each step to
approximately 13 hours. Here, τ = 2 π ÷ ω n
where ω n is the natural loop frequency. It is
approximately the time required for the PLL
to fully recover from a transient (see Notes 5
and 6).
8 A procedure for selecting R6 and R5 is as follows. (1) Construct a VCXO control input network that is suitable for your VCXO. Use highstability resistors and/or wire-wound multiturn
pots. Typical values are in the 10 to 20 kΩ
range for the COARSE adjustment and 1 kΩ for
the FINE adjustment pot. (2) Ground the bottom end of the FINE adjustment pot, set it to the
center of its range and adjust the coarse pot to
bring the VCXO to approximately the correct
frequency. (3) Now, measure the resistance
values you have for RA and RB . (4) Determine
the relative frequency change (∆ F/F) that occurs when you adjust the pots to change the
voltage by 1 V at the control input. This value
is the control sensitivity, S. A triggerable ’scope
and a second stable oscillator may be useful in
determining S. Trigger the ’scope from the second oscillator and observe the VCXO signal.
The VCXO signal will probably drift slowly
across the screen. The change in the rate of
drift when the VCXO control voltage is changed
by 1 V is S. (5) Select a value for R5 around
100 Ω (the value should be much less than RA
and RB ). (6) Finally, compute the value of R6
you need from the equation
R6 = R5[(RA / (RA+RB) × (S / 7.5×10 –9)) – 1]
(Eq 1)
9 The Oncore VP and the GPS-25 receiver
boards are presently available via a bulk purchase from TAPR, Tucson, AZ; tel 940-3830000. The Oncore is more expensive, but
offers a more-stable timing output. The recently introduced Oncore UT+ model is a
less-expensive version of the VP that is said
to retain its high quality timing. It is available
from Synergy Systems, Inc; tel 619-5660666; http://www.synergy-gps.com.
10 At each 30-second DAC update, the controller prints three five-digit (16-bit) numbers,
each separated by CR and LF characters. The
first of the three numbers is the total count
from the phase detector counter (U2A/U4) for
the previous 30 seconds. When multiplied by
the constant, 41.7 ns per count divided by 30
counting intervals = 1.39, this number is the
phase difference in nanoseconds between
GPS and the VCXO. The controller attempts
to keep this count constant at the value 1024.
By using a phase difference offset from zero
the controller can easily track both positive
and negative phase changes. The second of
the three ASCII numbers is the value the digital filter is currently sending to the DAC to
control the VCXO frequency. This number can
be either positive or negative (depending on
whether a positive or negative VCXO disciplining voltage is needed) and is in two’s
complement binary notation (values larger
than 32768 are interpreted as negative and
equal to the value minus 65536). Only the
most-significant 16 bits of the 18-bit DAC input are printed. The third ASCII value indicates the status of the controller. This number
is a combination of three values arranged so
44
July 1998
that it is easy to see the status at a glance.
The current filter switch setting determines
the lowest decimal place (0-7). If the phase
difference is far from the set point, suggesting
that the PLL is not locked, the value 100 is
added to the number. If the phase has just
changed abruptly, which invokes a “deglitching” algorithm in the software, the value 10 is
added. For example, the value 105 indicates
that filter time constant 5 is in use and that the
phase lock may be questionable.
11 Opening S5 will hold the DAC voltage at its
current setting, thereby preventing the controller from further changing the VCXO frequency. However, the ASCII output continues to provide data as before so that the
“open loop” GPS-VCXO phase drift can be
monitored. In normal operation, S5 should be
closed, but opening it for short periods is an
effective method of eliminating “GPS jitter”
that can be useful when ultra-stable shortterm performance is needed.
12 Variations in the electron density of the ionosphere can also cause the arrival time of the
GPS signals to vary; however this is partially
corrected in the GPS receiver using data
broadcast by the satellites. The residual uncorrected variation is only a few nanoseconds.
13 These AV performance numbers are based
on my estimates and the unpublished GPS
stability measurements of W3IWI in connection with the Totally Accurate Clock project
(see www.tapr.org).
14 Contact me at RR 2 Box 423 Santa Fe, NM
87505, or by e-mail at shera@rt66.com. The
binary software is free for noncommercial
uses. The software is in the file GPSCNTRL
.ZIP and can be found on the Internet (ftp to
oak.oakland.edu/pub/hamradio/arrl/qstbinaries and on the ARRL BBS 860-5940306.
15 Start with the Microchip Web site at http://
www.microchip.com.
Brooks Shera, W5OJM, was first licensed in
1961 as K8WGA. He received his MS in Physics
from the University of Chicago and a PhD in
Physics from Case Western Reserve University.
Brooks has been employed for most of his professional career at Los Alamos National Laboratory in New Mexico, where he has conducted
basic research on the structure and properties
of atomic nuclei. More recently, Brooks has developed laser-based techniques for detecting
single atoms in liquid solutions and is applying
these methods to biological and medical research. He is presently an independent consultant focusing primarily on developing instrumentation for research. Brooks holds several patents, is the author of more than 100 research
papers, and is a Fellow of the American Physical Society. He can be reached at RR 2 Box 423
Santa Fe, NM 87505; e-mail shera@rt66.com.
New Products
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contact Rocky Mountain Antennas, 1409
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HAMCALC VERSION 31: 200 FREE
ELECTRONIC DESIGN PROGRAMS
◊ George Murphy’s venerable HAMCALC,
now in version 31, is a handy collection of
electronic design and math reference programs that has been a favorite or hams, engineers and instructors for more than five
years. The easy-to-use program requires a
DOS-compatible PC with a 3.5-inch disk
drive. Run HAMCALC from a floppy or install it to your hard drive. Windows 95/98
users can run HAMCALC in DOS mode.
To stay on top of the often-upgraded release, author George Murphy, VE3ERP,
suggests that users get the latest, most upto-date version directly from him, as the
copies found on-line and in many CD-ROM
software collections aren’t current and
some don’t run correctly.
To get your copy send $5 (US check or
money order only, no stamps or IRCs) to
cover materials and airmail postage to
George Murphy, VE3ERP, 77 McKenzie
St, Orillia ON L3V 6A6, CANADA.
ELECTROINSTRUMENT KEY-8
PADDLE/KEYER FROM MILESTONE
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◊ Now available from Milestone Technologies is the Key-8 paddle/keyer made by
ElectroInstrument, a former Russian military contractor specializing in aerospace
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body is polished chrome, and the base plate,
machined from solid brass, brings the unit’s
weight to almost 3.5 pounds. This is one
paddle that won’t slide around on your
desktop!
The internal keyer (non Iambic) features
an adjustable sidetone oscillator, 5-50
WPM keying speeds and an electrically isolated reed-relay output. All connections are
made via a 5-pin DIN socket (matching plug
included). Other features include precision
tension and spacing adjustments, and silver
contacts.
Price: $129.95. For more information,
contact Milestone Technologies, 3140 S
Peoria St, Unit K-156, Aurora, CO 800143155; tel 303-752-3382; http://www.
mtechnologies.com.
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