EVALUATION KIT AVAILABLE MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current General Description Features The MAX16974 is a 2A, current-mode, step-down converter with an integrated high-side switch. It is designed to operate with 3.5V to 28V input voltages, while using only 35FA quiescent current at no load. The switching frequency is adjustable from 220kHz to 2.2MHz by an external resistor and can be synchronized to an external clock. The output voltage is pin selectable to be 5V fixed or 1V to 10V adjustable. The wide input voltage range makes the device ideal for automotive and industrial applications. SWide 3.5V to 28V Input Voltage Range S42V Input Transient Tolerance S5V Fixed or 1V to 10V Adjustable Output Voltage SIntegrated 2A Internal High-Side Switch SAdjustable Switching Frequency (220kHz to 2.2MHz) SOperates Through Cold Crank with High Duty Cycle SFrequency Synchronization Input SInternal Boost Diode S35µA Skip-Mode Operating Current S5µA Typical Shutdown Current SAdjustable Power-Good Output Level and Timing S3.3V Logic Level to 42V Compatible Enable Input SCurrent-Limit, Thermal-Shutdown, and Overvoltage Protections SAutomotive Temperature Range: -40°C to +125°C SAEC-Q100 Qualified The device operates in skip mode for reduced current consumption in light-load applications. An adjustable reset threshold helps keep microcontrollers alive down to their lowest specified input voltage. Protection features include cycle-by-cycle current limit, overvoltage, and thermal shutdown with automatic recovery. The device also features a power-good monitor to ease powersupply sequencing. The device operates over the -40°C to +125°C automotive temperature range and is available in a 16-pin TSSOP-EP package. Ordering Information PART Applications MAX16974AUE/V+ Automotive TEMP RANGE PIN-PACKAGE -40NC to +125NC 16 TSSOP-EP* /V denotes an automotive qualified part. +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad. Industrial Typical Application Circuit VBAT CIN1 47µF CIN2 4.7µF CIN3 0.1µF EN SUP SUPSW RCOMP 20kI FB BIAS CCRES 100pF RESETI VBIAS FOSC CBIAS 1µF L1 4.7µH BST D1 OUT MAX16974 RFOSC 12.1kI CIN5 4.7µF VOUT COMP CCOMP2 OPEN CIN4 0.1µF VOUT = 5V AT 2A LX FSYNC CCOMP1 5600pF CBST 0.1µF COUT 22µF VBIAS RRES 10kI RES CRES GND PLACE CIN2 (4.7µF) AND CIN3 (0.1µF) NEXT TO SUP. PLACE CIN4 (0.1µF) AND CIN5 (4.7µF) NEXT TO SUPSW. For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com. 19-5630; Rev 3; 12/16 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current ABSOLUTE MAXIMUM RATINGS SUP, SUPSW, EN, LX to GND................................-0.3V to +45V SUP to SUPSW......................................................-0.3V to +0.3V BST to GND ...........................................................-0.3V to +47V BST to LX ................................................................-0.3V to +6V OUT to GND...........................................................-0.3V to +12V RESETI, FOSC, COMP, BIAS, FSYNC, CRES, RES, FB to GND...................................................-0.3V to +6V Output Short-Circuit Duration.....................................Continuous Continuous Power Dissipation (TA = +70NC) TSSOP (derate 26.1mW/NC above +70NC)........... 2088.8mW* Operating Temperature Range......................... -40NC to +125NC Junction Temperature......................................................+150NC Storage Temperature Range............................. -65NC to +150NC Lead Temperature (soldering, 10s).................................+300NC Soldering Temperature (reflow).......................................+260NC *As per JEDEC 51 standard (multilayer board). Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. PACKAGE THERMAL CHARACTERISTICS (Note 1) TSSOP Junction-to-Ambient Thermal Resistance (qJA)........38.3°C/W Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial. ELECTRICAL CHARACTERISTICS (VSUP = VSUPSW = 14V, L1 = 4.7FH, VEN = 14V, CIN = 10FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 12.1kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.) PARAMETER Supply Voltage Supply Current SYMBOL VSUP ISUP Shutdown Supply Current BIAS Regulator Voltage CONDITIONS Normal operation MIN BIAS Undervoltage Lockout VUVBIAS BIAS Undervoltage Lockout Hysteresis VUVBIAS_ MAX UNITS 28 V Normal operation, ILOAD = 1.5A 2 3 mA Skip mode, no load, VOUT = 5V 35 50 FA VEN = 0V VBIAS TYP 3.5 5 10 FA VSUP = VSUPSW = 6V to 42V, VOUT > 6V 4.9 5.1 5.5 V VBIAS rising 2.85 3.05 3.25 V HYS Thermal-Shutdown Threshold 350 mV 175 NC OUTPUT VOLTAGE (OUT) Output Voltage Skip-Mode Output Voltage VOUT Normal operation, VFB = VBIAS, ILOAD = 2A, TA = +25°C 4.95 5 5.05 Normal operation, VFB = VBIAS, ILOAD = 2A, -40°C < TA <+125°C 4.9 5 5.1 4.95 5.05 5.2 VOUT_SKIP No load, VFB = VBIAS (Note 2) Load Regulation VOUT = 5V, VFB = VBIAS; 400mA < ILOAD < 2A Line Regulation 6V < VSUP < 28V BST Input Current IBST LX Current Limit ILX Skip-Mode Current Threshold 2 ISKIP_TH V 100% duty cycle, VBST - VLX = 5V 2.5 V 1 % 0.02 %/V 1.5 3 3 3.5 240 mA A mA Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current ELECTRICAL CHARACTERISTICS (continued) (VSUP = VSUPSW = 14V, L1 = 4.7FH, VEN = 14V, CIN = 10FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 12.1kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.) PARAMETER SYMBOL Power-Switch On-Resistance LX Leakage Current RON ILX,LEAK CONDITIONS MIN RON measured between SUPSW and LX, ILX = 500mA TYP MAX UNITS 185 400 mI VSUP = 28V, VLX = 0V, TA = +25°C 1 VSUP = 28V, VLX = 0V, TA = +125°C 0.04 FA TRANSCONDUCTANCE AMPLIFIER (COMP) FB Input Current IFB FB Regulation Voltage VFB 20 nA FB connected to an external resistive divider, TA = +25°C 0.99 1.0 1.01 FB connected to an external resistive divider, -40°C < TA < +125°C 0.985 1.0 1.015 V FB Line Regulation DVLINE 6V < VSUP < 28V 0.02 %/V Transconductance (from FB to COMP) gm,EA VFB = 1V, VBIAS = 5V 1000 FS tMIN,ON 120 ns DCCC 92 % Minimum On-Time Cold-Crank Event Duty Cycle OSCILLATOR FREQUENCY Oscillator Frequency RFOSC = 120kI 190 260 310 kHz RFOSC = 12.1kI 2.00 2.20 2.48 MHz EXTERNAL CLOCK INPUT (FSYNC) External Input Clock Acquisition Time External Input Clock High Threshold tFSYNC 4 VFSYNC_HI VFSYNC rising 1.5 V External Input Clock Low Threshold VFSYNC_LO VFSYNC falling FSYNC Pulldown Resistance Soft-Start Time 0.5 IFSYNC tSS Cycles 510 fSW = 220kHz 9.3 fSW = 2.2MHz 0.93 V kI ms ENABLE INPUT (EN) Enable-On Threshold Voltage Low VEN_LO Enable-On Threshold Voltage High VEN_HI Enable Threshold Voltage Hysteresis Enable Input Current 0.7 2.2 V V VEN,HYS 0.35 V IEN 0.5 FA RESET Reset Internal Switching Level RESETI Threshold Voltage CRES Threshold Voltage CRES Threshold Hysteresis RESETI Input Current Maxim Integrated VTH_RISING VFB rising, VRESETI = 0V 0.88 0.90 0.92 VTH_FALLING VFB falling, VRESETI = 0V 0.83 0.85 0.87 VRESETI_LO VRESETI falling 1.13 1.2 1.27 V 1.1 1.25 1.45 V VCRES_HI VCRES rising VCRES_HYS IRESET VRESETI = 0V V 0.04 V 0.02 FA 3 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current ELECTRICAL CHARACTERISTICS (continued) (VSUP = VSUPSW = 14V, L1 = 4.7FH, VEN = 14V, CIN = 10FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 12.1kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.) PARAMETER SYMBOL CRES Source Current ICRES CRES Pulldown Current CONDITIONS VOUT in regulation ICRES_PD MIN TYP MAX 9.5 10 10.5 UNITS FA VOUT out of regulation 1 mA RES Sink Current VRES pulls low, VRES > 0.4V 1 mA RES Leakage Current (OpenDrain Output) VOUT in regulation, TA = +25NC Reset Debounce Time tRES_DEB 1 VRESETI falling 25 FA Fs Note 2: Guaranteed by design; not production tested. Typical Operating Characteristics (VSUP = VSUPSW = 14V, VOUT = 5V, FSYNC = GND, fOSC = 400kHz, TA = +25NC, unless otherwise noted. See Figure 1.) 5V/400kHz 70 5V/2.2MHz 60 50 40 VSUP = 14V 80 EFFICIENCY (%) EFFICIENCY (%) 80 90 5V/400kHz 70 60 90 80 SUPPLY CURRENT (µA) VSUP = 14V 100 MAX16974 toc02 90 SUPPLY CURRENT vs. SUPPLY VOLTAGE EFFICIENCY vs. ILOAD 100 MAX16974 toc01 100 5V/2.2MHz 50 40 70 60 50 30 30 20 20 20 10 10 0 0.00001 0.0001 0.001 0.01 0.1 0.01 ILOAD (C) 1 8 10 12 14 16 18 20 22 24 26 28 SUPPLY VOLTAGE (V) STARTUP INTO HEAVY LOAD MAX16974 toc05 MAX16974 toc04 SHUTDOWN CURRENT (µA) 6 10 ILOAD (A) SHUTDOWN CURRENT vs. INPUT VOLTAGE 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 5V/400kHz 0 0 0.1 5V/2.2MHz 40 30 10 MAX16974 toc03 EFFICIENCY vs. ILOAD (SKIP MODE) 5V/div VSUP 0V TA = +125°C 5V/div TA = +25°C 0V VOUT 2A/div TA = -40°C 4 6 8 10 12 14 16 18 20 22 24 26 28 ILOAD SF = 400kHz 0A 10ms/div INPUT VOLTAGE (V) 4 Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Typical Operating Characteristics (continued) (VSUP = VSUPSW = 14V, VOUT = 5V, FSYNC = GND, fOSC = 400kHz, TA = +25NC, unless otherwise noted. See Figure 1.) LOAD TRANSIENT UNDERVOLTAGE PULSE MAX16974 toc06 MAX16974 toc07 200mV/div VOUT 5V/div VSUP 0V 0V SF = 400kHz VOUT ACCOUPLED ILOAD 5V/div VOUT 0V 1A/div 0A 5V/div 0V SF = 2.2MHz VRES 20ms/div SWITCHING FREQUENCY vs. TEMPERATURE RESET TIMEOUT PERIOD vs. CRES 80 70 60 50 40 30 20 500 MAX16974 toc09 RESET TIMEOUT PERIOD (ms) 90 VOUT = 5V 490 SWITCHING FREQUENCY (kHz) MAX16974 toc08 100 480 470 460 450 440 430 420 410 10 400 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) CRES (µF) LOAD DUMP TEST (5V/2.2MHz) SHORT-CIRCUIT RESPONSE MAX16974 toc10 MAX16974 toc11 2.2MHz VSUP 20V/div 5A/div ILX 0 VOUT 5V/div 0A 5V/div VOUT 0 VRES 5V/div 0 100ms/div Maxim Integrated 0V 5V/div 0V VRES 2ms/div 5 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Typical Operating Characteristics (continued) (VSUP = VSUPSW = 14V, VOUT = 5V, FSYNC = GND, fOSC = 400kHz, TA = +25NC, unless otherwise noted. See Figure 1.) 2.4 2.0 1.6 1.2 0.8 420 ILOAD = 1A 418 SWITCHING FREQUENCY (kHz) MAX16974 toc12 0.4 416 414 412 410 408 406 404 402 400 0 6 12 22 32 42 52 62 72 82 92 102 112 122 8 10 12 14 16 18 20 22 24 26 28 SUPPLY VOLTAGE (V) RFOSC (kI) FSYNC TRANSITION FROM INTERNAL TO EXTERNAL FREQUENCY (5V/2.2MHz CONFIGURATION) VOUT vs. ILOAD MAX16974 toc14 VOUT 5.10 5.06 0 0 5.04 VOUT (V) 2V/div EXTERNAL CLOCK AT SYNC VSUP = 14V 5.08 5V/div MAX16974 toc15 SWITCHING FREQUENCY (MHz) 2.8 MAX16974 toc13 SWITCHING FREQUENCY vs. SUPPLY VOLTAGE SWITCHING FREQUENCY vs. RFOSC 5.02 5V/2.2MHz 5.00 4.98 4.96 VLX 10V/div 4.94 0 4.92 5V/400kHz 4.90 1µs/div 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 ILOAD (A) STARTUP INTO HEAVY LOAD (5V/2.2MHz) SWITCHING FREQUENCY vs. ILOAD VSUP = 14V SWITCHING FREQUENCY (kHz) 418 MAX16974 toc17 MAX16974 toc16 420 416 VSUP 5V/div 414 412 0 VOUT 2V/div 0V 410 408 406 VRES 5V/div 0 ILOAD 2A/div 0V 404 402 400 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 400µs/div ILOAD (A) 6 Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Typical Operating Characteristics (continued) (VSUP = VSUPSW = 14V, VOUT = 5V, FSYNC = GND, fOSC = 400kHz, TA = +25NC, unless otherwise noted. See Figure 1.) OUTPUT RESPONSE TO SLOW INPUT RAMP UP (5V/400kHz) OUTPUT RESPONSE TO SLOW INPUT RAMP DOWN (5V/2.2MHz) MAX16974 toc18 ILOAD = 2A MAX16974 toc19 10V/div VSUP VSUP 0 VOUT 5V/div 0 VLX 10V/div 2V/div 0 VOUT 0 2V/div 0 VLX 2V/div 5V/div 0 VRES 0 10s/div 10s/div DIPS AND DROP TEST (5V/2.2MHz) MAX16974 toc20 10V/div VSUP 0 VOUT 5V/div 0 VLX 10V/div 0 2A/div 0 ILOAD 10ms/div Maxim Integrated 7 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current VBAT CIN1 47µF CIN2 4.7µF CIN3 0.1µF SUP EN SUPSW CBST 0.1µF RCOMP 20kI L1 VOUT = 15µH 3.3V AT 2A AT 300kHz LX VOUT COMP MAX16974 RESETI CRES COUT1 47µF VBIAS RFB2 56kI BIAS CCRES 1nF COUT 47µF RFB1 124kI FB FOSC CBIAS 1µF D1 OUT RFOSC 82kI CCOMP2 OPEN CIN5 4.7µF BST FSYNC CCOMP1 5600pF CIN4 0.1µF RRES 10kI RES GND PLACE CIN2 (4.7µF) AND CIN3 (0.1µF) NEXT TO SUP. PLACE CIN4 (0.1µF) AND CIN5 (4.7µF) NEXT TO SUPSW. Figure 1. 3.3V Fixed Output Voltage Configuration Pin Configuration 16 15 14 13 12 11 10 BIAS BST LX SUP SUPSW EN RES RESETI TOP VIEW 9 MAX16974 EP 6 7 8 GND FSYNC 5 OUT CRES 4 FB 3 COMP 2 I.C. 1 FOSC + TSSOP 8 Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Pin Description PIN NAME 1 CRES Analog Reset Timer. CRES sources 10FA (typ) of current into an external capacitor to set the reset timeout period. Reset timeout period is defined as the time between the start of output regulation and RES going high impedance. Leave unconnected for minimum delay time. 2 FOSC Resistor-Programmable Switching-Frequency Setting Control Input. Connect a resistor from FOSC to GND to set the switching frequency. 3 FSYNC Synchronization Input. The device synchronizes to an external signal applied to FSYNC. The external signal period must be 10% shorter than the internal clock period for proper operation. 4 I.C. 5 COMP Error Amplifier Output. Connect an RC network from COMP to GND for stable operation. See the Compensation Network section for more details. 6 FB Feedback Input. Connect an external resistive divider from OUT to FB and GND to set the output voltage. Connect to BIAS to set the output voltage to 5V. 7 OUT Supply Input. OUT provides power to the internal circuitry when the output voltage of the converter is set between 3V and 5V. 8 GND Ground 9 BIAS Linear Regulator Output. BIAS powers up the internal circuitry. Bypass with a 1FF capacitor to ground. 10 BST High-Side Driver Supply. Connect a 0.1FF capacitor between LX and BST for proper operation. 11 SUP Voltage Supply Input. SUP powers up the internal linear regulator. Connect a minimum of 1FF capacitor from SUP to GND close to the IC. Connect SUP to SUPSW. 12 LX 13 SUPSW Internal High-Side Switch Supply Input. SUPSW provides power to the internal switch. For most applications, connect 4.7FF and 0.1FF capacitors between SUPSW and GND close to the IC. See the Input Capacitor section for more details. 14 EN Battery-Compatible Enable Input. Drive EN low to disable the device. Drive EN high to enable the device. 15 RES Open-Drain Active-Low Reset Output. RES asserts when VOUT is below the reset threshold set by RESETI. 16 RESETI — EP Maxim Integrated FUNCTION Internally Connected. Connect to GND. Inductor Switching Node. Connect a Schottky diode between LX and GND. Reset Threshold Level Input. Connect to a resistive divider to set the reset threshold for RES. Connect to GND to enable the internal reset threshold. Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power dissipation. Do not use as the only IC ground connection. EP must be connected to GND. 9 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Detailed Description The MAX16974 is a constant-frequency, current-mode, automotive buck converter with an integrated high-side switch. The device operates with 3.5V to 28V input voltages and tolerates input transients up to 42V. During undervoltage events, such as cold-crank conditions, the internal pass device maintains up to 92% duty cycle. An open-drain, active-low reset output helps monitor the output voltage. The device offers an adjustable reset threshold that helps keep microcontrollers alive down to their lowest specified input voltage. A capacitor programmable reset timeout ensures proper startup. SUP BIAS The switching frequency is resistor programmable from 220kHz to 2.2MHz to allow optimization for efficiency, noise, and board space. A clock input, FSYNC, allows the device to synchronize to an external clock. During light-load conditions, the device enters skip mode that reduces the quiescent current down to 35FA. The 5V fixed output voltage option eliminates the need for external resistors and reduces the supply current by up to 50FA. See Figure 2 for the internal block diagram. Supply Voltage Range (SUP) The device’s supply voltage range (VSUP) is compatible with the typical 3.5V to 28V automotive battery voltage range and can tolerate transients up to 42V. FOSC BST SUPSW DRV ISENSE LEVEL SHIFT FSYNC LX OUT EN STANDBY SUPPLY OSC SUM ILIM REF EA PWM COMP LOGIC LDO MUX FB COMP UVLO LOGIC FOR DROPOUT VBIAS SOFTSTART RES 10µA CRES COMP RESETI B.G. REF COMP MAX16974 GND Figure 2. Internal Block Diagram 10 Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Linear Regulator Output (BIAS) The device includes a 5V linear regulator, VBIAS, that provides power to the internal circuitry. Connect a 1FF ceramic capacitor from BIAS to GND. If the output voltage is set between 3.0V and 5.6V, the internal linear regulator only provides power until the output is in regulation. The internal linear regulator turns off once the output is in regulation and load current is below 50mA, allowing the output to provide power to the device. External Clock Input (FSYNC) The device synchronizes to an external clock signal applied at FSYNC. The signal at FSYNC must have a 10% shorter period than the internal clock period for proper synchronization. The internal clock signal takes over if the externally applied signal has a frequency lower than the internal clock frequency. Adjustable Reset Level The device features a programmable reset threshold using a resistive divider between OUT, RESETI, and GND. Connect RESETI to GND for the internal threshold. RES asserts low when the output voltage falls to 85% of its programmed level. RES deasserts when the output voltage goes above 90% of its set voltage. Some microprocessors have a wide input voltage range (5V to 3.3V) and can operate during device dropout. Use a resistive divider at RESETI to adjust the reset activation level (RES goes low) to lower levels. The reference voltage at RESETI is 1.2V (typ). The device also offers a capacitor-programmable reset timeout period. Connect a capacitor from CRES to GND to adjust the reset timeout period. When the output voltage goes out of regulation, RES asserts low, and the reset timing capacitor discharges with a 1mA pulldown current. Once the output is back in regulation, the reset timing capacitor recharges with 10FA (typ) current. RES stays low until the voltage at CRES reaches 1.25V (typ). Dropout Operation The device has an effective maximum duty cycle to help refresh the BST capacitor when continuously operated in dropout. When the high-side switch is on for three consecutive clock cycles, the device forces the highside switch off during the final 35% of the fourth clock cycle. When the high-side switch is off, the LX node is pulled low by current flowing through the external Maxim Integrated Schottky diode. This increases the voltage across the BST capacitor. To ensure that the inductor has enough current to pull LX to ground, an internal load sinks current from VOUT when the device is close to dropout and when the external load is small. Once the input voltage is increased above the dropout region, the device continues to regulate without restarting. If the device has neither external clock nor external load, the effective maximum duty cycle is 92% when operating deep into dropout. This effective maximum duty cycle is influenced by the external load and by the external synchronized clock, if any. System Enable (EN) An enable-control input (EN) activates the device from its low-power shutdown mode. EN is compatible with inputs from automotive battery level down to 3.3V. The highvoltage compatibility allows EN to be connected to SUP, KEY/KL30, or the INH pin of a CAN transceiver. EN turns on the internal regulator. Once VBIAS is above the internal lockout level, VUVL = 3.05V (typ), the controller activates and the output voltage ramps up within 2048 cycles of the switching frequency. A logic-low at EN shuts down the device. During shutdown, the internal linear regulator and gate drivers turn off. Shutdown is the lowest power state and reduces the quiescent current to 5FA (typ). Drive EN high to bring the device out of shutdown. Overvoltage Protection The device includes an overvoltage protection circuit that protects the device when there is an overvoltage condition at the output. If the output voltage increases by more than 10% of its set voltage, the device stops switching. The device resumes regulation once the overvoltage condition is removed. Overload Protection The overload protection circuitry is activated when the device is in current limit and VOUT is below the reset threshold. Under these conditions the device enters a soft-start mode. If the overcurrent condition is removed before the soft-start mode is over, the device regulates the output voltage to its set value. Otherwise, the softstart cycle repeats until the overcurrent condition is removed. 11 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Skip Mode During light-load operation, IINDUCTOR P 240mA, the device enters skip-mode operation. Skip mode turns off the internal switch and allows the output to drop below regulation voltage before the switch is turned on again. The lower the load current, the longer it takes for the regulator to initiate a new cycle. Because the converter skips unnecessary cycles, the converter efficiency increases. During skip mode the quiescent current drops to 35FA. VOUT RFB1 RESETI MAX16974 RFB2 FB Overtemperature Protection Thermal-overload protection limits the total power dissipation in the device. When the junction temperature exceeds +175°C (typ), an internal thermal sensor shuts down the internal bias regulator and the step-down controller, allowing the IC to cool. The thermal sensor turns on the device again after the junction temperature cools by +15°C. Applications Information Output Voltage/Reset Threshold Resistive Divider Network Although the device’s output voltage and reset threshold can be set individually, Figure 3 shows a combined resistive divider network to set the desired output voltage and the reset threshold using three resistors. Use the following formula to determine the RFB3 of the resistive divider network: R FB3 R TOTAL × VREF OUT where VREF = 1V, RTOTAL = selected total resistance of RFB1, RFB2, and RFB3 in ohms, and VOUT is the desired output voltage in volts. R FB2 = R TOTAL × VREF_RES - R FB3 V RES where VREF_RES is 1.2V (see the Electrical Characteristics table), and VRES is the desired reset threshold in volts. The precision of the reset threshold function is dependent on the tolerance of the resistors used for the divider. Care must be taken to choose the values of the resistors. Too small a resistor value adds to the device’s quiescent current, whereas if the resistors are too large, there is some noise susceptibility to the FB pin. 12 RFB3 Figure 3. Output Voltage/Reset Threshold Resistive Divider Network Boost Capacitor for Dropout Operation The device has an internal boost capacitor refresh algorithm for dropout operation. This is required to ensure proper boost capacitor voltage, which delivers power to the gate drive circuitry. If the high-side MOSFET is on consecutively for 3.65 clock cycles, the internal counter detects this and turns off the high-side MOSFET for 0.35 clock cycles. This is of particular concern when VIN is falling and approaching VOUT and a minimum switching frequency of 220kHz is used. The worst-case condition for boost capacitor refresh time is with no load on the output. For the boost capacitor to recharge completely, the LX node must be pulled to ground. If there is no current in the inductor, the LX node does not go to ground. To solve this issue, an internal load of approximately 100mA is turned on at the 6th clock cycle, which is determined by a separate counter. In the worst-case condition with no load, the LX node does not go below ground during the first detect of the 3.65 clock cycles. It must wait for the next 3.65 clock cycles to finish. This means the soonest the LX node can go below ground is 4 + 3.65 = 7.65 clock cycles. This time does not factor in the size of the inductor and the time it takes for the inductor current to build up to 100mA (internal load). So no-load minimum time before refresh is: dt (no load) = 7.65 clock cycles = 7.65 x 5µs (at 220kHz) = 34.77µs Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current where: SWITCHING FREQUENCY vs. RFOSC MAX16974 toc12 2.8 SWITCHING FREQUENCY (MHz) IBST(DROPOUT) = 3mA (worst case) 2.4 dt (no load) = 16 clock cycles dV (BST capacitor) = VOUT - 2.7V. Reset Timeout Period 2.0 The device offers a capacitor-adjustable reset timeout period. Connect up to 0.1FF capacitor from CRES to GND to set the timeout period. CRES can source 10FA of current. Use the following formula to set the timeout period: 1.6 1.2 0.8 0.4 RESET_TIMEOUT = 0 12 22 32 42 52 62 72 82 92 102 112 122 RFOSC (kI) Figure 4. Switching Frequency vs. RFOSC Assume a full 100mA is needed to refresh the BST capacitor. Depending on the size of the inductor, the time it takes to build up a full 100mA in the inductor is given by: dt (inductor) = L x di/dV (current buildup starts from the 6th clock cycle) L = inductor value chosen in the design guide di is the required current = 100mA dV = voltage across the inductor (assume this to be 0.5V), which means VIN is greater than VOUT by 0.5V If dt (inductor) < 7.65 - 6 (clock cycles), the BST capacitor should be sized as follows: BST_CAP ≥ IBST(DROPOUT) x dt (no load)/dV (BST capacitor) dt (no load) = 7.65 clock cycles = 34.77µs dV (BST capacitor) for (3.3V to 5V) output = VOUT - 2.7V (2.7V is the minimum voltage allowed on the bst capacitor) If dt (inductor) > 7.65 - 6 clock cycles, then wait for the next count of 3.65 clock cycles making dt (no load) = 11.65 clock cycles. Considering the typical inductor values used for 220kHz operation, the safe way to design the BST capacitor is to assume: dt (no load) as 16 clock cycles So the final BST_CAPACITOR equation is: BST_CAP = IBST(DROPOUT) x dt (no load)/dV (BST capacitor) Maxim Integrated 1.25V × C 10 × 10 -6 A (s), where C is the capacitor from CRES to GND in Farads. Internal Oscillator The switching frequency (fSW) is set by a resistor (RFOSC) connected from FOSC to GND. See Figure 4 to select the correct RFOSC value for the desired switching frequency. For example, a 2.2MHz switching frequency is set with RFOSC = 12.1kI. Higher frequencies allow designs with lower inductor values and less output capacitance. Consequently, peak currents and I2R losses are lower at higher switching frequencies, but core losses, gatecharge currents, and switching losses increase. Inductor Selection Three key inductor parameters must be specified for operation with the device: inductance value (L), inductor saturation current (ISAT), and DC resistance (RDCR). To select inductance value, the ratio of inductor peak-topeak AC current to DC average current (LIR) must be selected first. A good compromise between size and loss is a 30% peak-to-peak ripple current to average current ratio (LIR = 0.3). The switching frequency, input voltage, output voltage, and selected LIR determine the inductor value as follows: V (V − VOUT ) L = OUT SUP VSUP fSWI OUTLIR where VSUP, VOUT, and IOUT are typical values so that efficiency is optimum for typical conditions. The switching frequency is set by RFOSC (see the Internal Oscillator section). The exact inductor value is not critical and can be adjusted to make trade-offs among size, cost, efficiency, and transient response requirements. 13 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Table 1. Inductor Size Comparison INDUCTOR SIZE SMALLER LARGER Lower price Smaller ripple Smaller form factor Higher efficiency Faster load response Larger fixed-frequency range in skip mode Table 1 shows a comparison between small and large inductor sizes. The inductor value must be chosen so the maximum inductor current does not reach the minimum current limit of the device. The optimum operating point is usually found between 10% and 30% ripple current. When pulse skipping (light loads), the inductor value also determines the loadcurrent value at which PFM/PWM switchover occurs. IRMS has a maximum value when the input voltage equals twice the output voltage (VSUP = 2VOUT), so IRMS(MAX) = ILOAD(MAX)/2. Choose an input capacitor that exhibits less than +10NC self-heating temperature rise at the RMS input current for optimal long-term reliability. The input-voltage ripple is comprised of DVQ (caused by the capacitor discharge) and DVESR (caused by the equivalent series resistance (ESR) of the capacitor). Use low-ESR ceramic capacitors with high ripple-current capability at the input. Assume the contribution from the ESR and capacitor discharge equal to 50%. Calculate the input capacitance and ESR required for a specified input-voltage ripple using the following equations: ∆VESR ESRIN = ∆I I OUT + L 2 Find a low-loss inductor having the lowest possible where: DC resistance that fits in the allotted dimensions. Most inductor manufacturers provide inductors in standard (V − VOUT ) × VOUT ∆IL = SUP values, such as 1.0FH, 1.5FH, 2.2FH, 3.3FH, etc. Also VSUP × fSW × L look for nonstandard values, which can provide a better compromise in LIR across the input voltage and: range. If using a swinging inductor (where the no-load I × D(1 − D) VOUT inductance decreases linearly with increasing current), = CIN OUT = and D evaluate the LIR with properly scaled inductance values. ∆VQ × fSW VSUPSW For the selected inductance value, the actual peak-topeak inductor ripple current (DIINDUCTOR) is defined by: where IOUT is the maximum output current, and D is the duty cycle. VOUT (VSUP − VOUT ) ∆IINDUCTOR = VSUP × fSW × L Output Capacitor The output filter capacitor must have low enough ESR to where DIINDUCTOR is in A, L is in H, and fSW is in Hz. meet output ripple and load-transient requirements, yet have high enough ESR to satisfy stability requirements. Ferrite cores are often the best choices, although The output capacitance must be high enough to absorb powdered iron is inexpensive and can work well at the inductor energy while transitioning from full-load 220kHz. The core must be large enough not to saturate to no-load conditions without tripping the overvoltage at the peak inductor current (IPEAK): fault protection. When using high-capacitance, low-ESR ∆IINDUCTOR = IPEAK ILOAD(MAX) + capacitors, the filter capacitor’s ESR dominates the 2 output voltage ripple. So the size of the output capacitor depends on the maximum ESR required to meet the outInput Capacitor put voltage ripple (VRIPPLE(P-P)) specifications: The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage VRIPPLE(P −P) = ESR × ILOAD(MAX) × LIR ripple on the input caused by the circuit’s switching. The input capacitor RMS current requirement (IRMS) is defined by the following equation: IRMS = ILOAD(MAX) 14 VOUT (VSUP − VOUT ) VSUP The actual capacitance value required relates to the physical size needed to achieve low ESR, as well as to the chemistry of the capacitor technology. Thus, the capacitor is usually selected by ESR and voltage rating rather than by capacitance value. Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Transient Response When using low-capacity filter capacitors, such as The inductor ripple current also impacts transient ceramic capacitors, size is usually determined by the response performance, especially at low VSUP - VOUT capacity needed to prevent VSAG and VSOAR from differentials. Low inductor values allow the inductor causing problems during load transients. Generally, current to slew faster, replenishing charge removed from once enough capacitance is added to meet the overthe output filter capacitors by a sudden load step. The shoot requirement, undershoot at the rising-load edge total output-voltage sag is the sum of the voltage sag is no longer a problem (see the VSAG and VSOAR while the inductor is ramping up and the voltage sag equations in the Transient Response section). However, before the next pulse can occur: low-capacity filter capacitors typically have high-ESR zeros that can affect the overall stability. Other important 2 L(∆ILOAD(MAX) ) ∆ILOAD(MAX) (t − ∆t) criteria in the choice of the total output capacitance = VSAG + are the device’s soft-start time and maximum current C OUT 2C OUT ((VSUP × D MAX ) − VOUT ) capability (see the Soft-Start Time and Maximum Allowed Output Capacitance section). where DMAX is the maximum duty factor (see the Electrical Characteristics table), L is the inductor value Soft-Start Time and Maximum in FH, COUT is the output capacitor value in FF, t is the Allowed Output Capacitance switching period (1/fSW) in Fs, and δt equals (VOUT/ The device’s soft-start time depends on the selected VSUP x t when in fixed-frequency PWM mode, or L x 0.2 switching frequency. The soft-start time is fixed to 2048 x IMAX/(VSUP - VOUT) when in skip mode. The amount cycles, regardless of the switching frequency. This of overshoot (VSOAR) during a full-load to a no-load means at 2.2MHz the soft-start time is ~0.93ms, and at transient due to stored inductor energy can be 220kHz the soft-start time is ~9.3ms. calculated as: The device is a 2A-capable switching regulator and the 2 amount of load present at startup determines the total VSOAR ≈ ∆ILOAD(MAX) × L/ (2 x C OUT × VOUT ) output capacitance allowed for a particular application. ( C OUT(MAX) ≈ 2048/fSW × 1/∆VOUT × ILX(MIN) - ILOAD(MAX) Keeping the above equation in mind, see the following table to ensure that COUT is less than maximum allowed values. FREQUENCY = 400kHz VOUT (V) ILOAD (STARTUP) (A) COUT (MAX ALLOWED) 3.3 2 775FF 5 2 512FF 3.3 0 3.9mF 5 0 2.6mF FREQUENCY = 2.2MHz VOUT (V) ILOAD (STARTUP) (A) COUT (MAX ALLOWED) 3.3 2 140FF 5 2 93FF 3.3 0 705FF 5 0 465FF Maxim Integrated ) Rectifier Selection The device requires an external Schottky diode rectifier as a freewheeling diode. Connect this rectifier close to the device using short leads and short PCB traces. Choose a rectifier with a continuous current rating greater than the highest output current-limit threshold (3.5A), and with a voltage rating greater than the maximum expected input voltage, VSUPSW. Use a low forward-voltage-drop Schottky rectifier to limit the negative voltage at LX. Avoid higher than necessary reverse-voltage Schottky rectifiers that have higher forward-voltage drops. Compensation Network The device uses an internal transconductance error amplifier with its inverting input and output available to the user for external frequency compensation. The output capacitor and compensation network determine the loop stability. The inductor and the output capacitor are chosen based on performance, size, and cost. Additionally, the compensation network optimizes the control-loop stability. The controller uses a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor, so the device uses 15 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current the voltage drop across the high-side MOSFET. Currentmode control eliminates the double pole in the feedback loop caused by the inductor and output capacitor resulting in a smaller phase shift and requiring less elaborate error-amplifier compensation than voltage-mode control. A simple single-series resistor (RC) and capacitor (CC) are all that is required to have a stable, high-bandwidth loop in applications where ceramic capacitors are used for output filtering (Figure 5). For other types of capacitors, due to the higher capacitance and ESR, the frequency of the zero created by the capacitance and ESR is lower than the desired closed-loop crossover frequency. To stabilize a nonceramic output capacitor loop, add another compensation capacitor (CF) from COMP to GND to cancel this ESR zero. The basic regulator loop is modeled as a power modulator, output feedback divider, and an error amplifier. The power modulator has a DC gain set by gmc O RLOAD, with a pole and zero pair set by RLOAD, the output capacitor (COUT), and its ESR. The following equations approximate the value for the gain of the power modulator (GAINMOD(DC)), neglecting the effect of the ramp stabilization. Ramp stabilization is necessary when the duty cycle is above 50% and is internally done for the device. GAINMOD(DC) = g mc × R LOAD where RLOAD = VOUT/ILOUT(MAX) in I and gmc = 3S. In a current-mode step-down converter, the output capacitor, its ESR, and the load resistance introduce a pole at the following frequency: fpMOD = 1 2π × C OUT × R LOAD VOUT R1 RC VREF When COUT is composed of “n” identical capacitors in parallel, the resulting COUT = n O COUT(EACH) and ESR = ESR(EACH)/n. Note that the capacitor zero for a parallel combination of alike capacitors is the same as for an individual capacitor. The feedback voltage-divider has a gain of GAINFB = VFB/VOUT, where VFB is 1V (typ). 16 CF CC Figure 5. Compensation Network The transconductance error amplifier has a DC gain of GAINEA(DC) = gm,EA O ROUT,EA, where gm,EA is the error amplifier transconductance, which is 1000FS (typ), and ROUT,EA is the output resistance of the error amplifier 50MI. A dominant pole (fdpEA) is set by the compensation capacitor (CC) and the amplifier output resistance (ROUT,EA). A zero (fZEA) is set by the compensation resistor (RC) and the compensation capacitor (CC). There is an optional pole (fpEA) set by CF and RC to cancel the output capacitor ESR zero if it occurs near the crossover frequency (fC), where the loop gain equals 1 (0dB)). Thus: fpdEA = 1 2π × C C × (R OUT,EA + R C ) fzEA = 1 2π × C C × R C fpEA = 1 2π × C F × R C The output capacitor and its ESR also introduce a zero at: 1 fzMOD = 2π × ESR × C OUT COMP gm R2 The loop-gain crossover frequency (fC) should be set below 1/5th of the switching frequency and much higher than the power-modulator pole (fpMOD): f fpMOD << fC ≤ SW 5 Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current The total loop gain as the product of the modulator gain, the feedback voltage-divider gain, and the error amplifier gain at fC should be equal to 1. So: GAINMOD(fC) × VFB × GAINEA(fC) = 1 VOUT For the case where fzMOD is greater than fC: GAINEA(fC) = g m,EA × R C GAIN = MOD(fC) GAINMOD(DC) × fpMOD fC V GAINMOD(fC) × FB × g m,EA × R C = 1 VOUT Solving for RC: VOUT g m,EA × VFB × GAINMOD(fC) Set the error-amplifier compensation zero formed by RC and CC (fzEA) at the fpMOD. Calculate the value of CC as follows: CC = 2π × fpMOD × R C 1 2π × fzMOD × R C As the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly and the crossover frequency remains the same. For the case where fzMOD is less than fC: The power-modulator gain at fC is: GAIN = MOD(fC) GAINMOD(DC) × Maxim Integrated GAINMOD(fC) × f VFB 1 × g m,EA × R C × zMOD = VOUT fC RC = VOUT × fC g m,EA × VFB × GAINMOD(fC) × fzMOD Set the error-amplifier compensation zero formed by RC and CC at the fpMOD (fzEA = fpMOD): CC = 1 2π × fpMOD × R C If fzMOD is less than 5 O fC, add a second capacitor, CF, from COMP to GND. Set fpEA = fzMOD and calculate CF as follows: CF = 1 2π × fzMOD × R C PCB Layout Guidelines 1 If fzMOD is less than 5 x fC, add a second capacitor, CF, from COMP to GND and set the compensation pole formed by RC and CF (fpEA) at the fzMOD. Calculate the value of CF as follows: CF = Therefore: f GAINEA(fC)= g m,EA × R C × zMOD fC Solving for RC: Therefore: RC = The error-amplifier gain at fC is: fpMOD fzMOD Careful PCB layout is critical to achieve low switching losses and clean, stable operation. Use a multilayer board whenever possible for better noise immunity and power dissipation. Follow these guidelines for good PCB layout: 1)Use a large contiguous copper plane under the IC package. Ensure that all heat-dissipating components have adequate cooling. The bottom pad of the device must be soldered down to this copper plane for effective heat dissipation and getting the full power out of the IC. Use multiple vias or a single large via in this plane for heat dissipation. 2) Isolate the power components and high-current path from the sensitive analog circuitry. This is essential to prevent any noise coupling into the analog signals. 3) Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. The high-current path composed of input capacitor, high-side FET, inductor, and the output capacitor should be as short as possible. 17 MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current 4)Keep the power traces and load connections short. This practice is essential for high efficiency. Use thick copper PCBs (2oz vs. 1oz) to enhance full-load efficiency. 5) The analog signal lines should be routed away from the high-frequency planes. This ensures integrity of sensitive signals feeding back into the IC. 6)The ground connection for the analog and power section should be close to the IC. This keeps the ground current loops to a minimum. In cases where only one ground is used, enough isolation between analog return signals and high power signals must be maintained. Chip Information PROCESS: BiCMOS Package Information For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”, “#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE PACKAGE CODE OUTLINE NO. LAND PATTERN NO. 16 TSSOP-EP U16E+3 21-0108 90-0120 7) Ensure a high-frequency decoupling capacitor of 0.1µF is placed next to the SUP pin of the IC. This capacitor prevents high-frequency noise from entering the SUP pin. Adding a resistor between the SUPSW and SUP pins along with the decoupling capacitor at the SUP pin is recommended to reduce noise sensitivity. 18 Maxim Integrated MAX16974 High-Voltage, 2.2MHz, 2A Automotive StepDown Converter with Low Operating Current Revision History REVISION NUMBER REVISION DATE 0 11/10 Initial release — 1 7/11 Corrected the GAINMOD(DC) and fpMOD equations in the Compensation Network section 16 2 3/15 Updated first two lines in Absolute Maximum Ratings section 2 3 12/16 Changed BIAS Regulator Voltage max in Electrical Characteristics from 5.4 to 5.5 2 DESCRIPTION PAGES CHANGED Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance. Maxim Integrated 160 Rio Robles, San Jose, CA 95134 USA 1-408-601-1000 © 2016 Maxim Integrated Products, Inc. 19 Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
* Your assessment is very important for improving the work of artificial intelligence, which forms the content of this project