power density optimization of emi filters for power

power density optimization of emi filters for power
Dottorato di ricerca in Energia e Tecnologie dell’Informazione
Dipartimento di Energia, Ingegneria dell'Informazione e Modelli Matematici.
Settore Scientifico Disciplinare ING-IND/31 - Elettrotecnica
POWER DENSITY OPTIMIZATION OF EMI FILTERS
FOR POWER ELECTRONIC CONVERTERS
IL DOTTORE
IL COORDINATORE
Ing. Graziella Giglia
Prof.ssa Maria Stella Mongiovì
IL TUTOR
IL CO-TUTOR
Prof. Ing. Guido Ala
Prof. Ing. G. Costantino Giaconia
IL CO-TUTOR ESTERNO
(CNR-ISSIA)
Dr. Ing. Maria Carmela Di Piazza
CICLO XXIX
ANNO CONSEGUIMENTO TITOLO 2017
To the person I love the most,
My son Giuseppe
2
ACKNOWLEDGEMENTS
I would like first of all to express appreciation and gratitude to my advisors, Prof. Ing. Guido Ala,
Dr. Ing. Maria Carmela Di Piazza and Prof. Ing. G. Costantino Giaconia, for their availability and
competence, for their guidance and support throughout my PhD studies.
I would like also to express my gratitude to Dr. Ing. Gianpaolo Vitale and Dr. Ing. Massimiliano
Luna for their kind cooperation during the PhD course.
I also thank the technicians Giuseppe Scordato and Antonio Sauro for their contribution in the
realization of the prototypes and for their support during the tests performed in the EMC laboratory.
I would also like to acknowledge the Prof. Pericle Zanchetta who gave me the opportunity to spend
a period of study and research at the Power Electronics, Machines and Control Research Group University of Nottingham.
Finally and the most importantly, I would like to thank my husband Eugenio and my son Giuseppe.
I appreciate their love, support, and sacrifice. My greatest wish is to balance work and family life
better and be more present in their lives.I would like also my parents and the rest of the family for
their continuous support over the years.
Thank you everyone,
Graziella
3
CONTENTS
LIST OF FIGURES ......................................................................................................................... 6
LIST OF TABLES ......................................................................................................................... 10
LIST OF ACRONYMS ................................................................................................................. 11
INTRODUCTION .......................................................................................................................... 13
CHAPTER I – ELECTROMAGNETIC COMPATIBILITY AND POWER DENSITY
ISSUES IN POWER ELECTRONIC CONVERTERS .............................................................. 15
1.1
EMC: GENERAL CONCEPTS AND DEFINITIONS ................................................................. 15
1.2
EMI ISSUES IN POWER ELECTRONIC CONVERTERS ........................................................... 18
1.2.1
1.3
EMI Mitigation Strategies……………………………………………….………….23
POWER DENSITY ISSUES IN POWER ELECTRONIC CONVERTERS ....................................... 30
1.3.1
Scopes of action for the power density improvement ............................................... 31
CHAPTER II – EMI ANALYSIS ................................................................................................. 35
2.1
INTRODUCTION .................................................................................................................. 35
2.2
CONDUCTED EMI AND NOISE PROPAGATION PATHS ....................................................... 35
2.3
CM AND DM EMI SEPARATION TECHNIQUES .................................................................. 37
2.4
2.3.1
Separation technique using RF current probes .......................................................... 40
2.3.2
Hardware-based separation technique ....................................................................... 42
2.3.3
Software-based separation technique ........................................................................ 43
EXPERIMENTAL VALIDATION OF THE SOFTWARE BASED SEPARATION TECHNIQUE ......... 44
CHAPTER III – EMI FILTER DESIGN .................................................................................... 48
3.1
INTRODUCTION .................................................................................................................. 48
3.2
CRITERIA FOR THE CHOICE OF EMI FILTER TOPOLOGY ..................................................... 48
3.3
REAL BEHAVIOR OF PASSIVE COMPONENTS ...................................................................... 52
3.4
3.3.1
Capacitors behavior including parasitic effects......................................................... 53
3.3.2
Inductors behavior including parasitic effects ........................................................... 54
EMI FILTER GENERAL DESIGN PROCEDURE ..................................................................... 56
3.4.1
Design of CM choke and DM extra inductor ............................................................ 59
3.4.2
Considerations on magnetic cores saturation ............................................................ 62
3.4.3
Considerations on materials of the EMI filter components ....................................... 63
4
Contents
CHAPTER IV – OPTIMIZED DESIGN OF HIGH POWER DENSITY EMI FILTER ....... 72
4.1
INTRODUCTION .................................................................................................................. 72
4.2
OPTIMIZED DESIGN PROCEDURE ....................................................................................... 73
4.3
ODEF APPLICATION .......................................................................................................... 77
4.4
SUMMARY.......................................................................................................................... 82
CHAPTER V – EXPERIMENTAL VALIDATION OF THE OPTIMIZED EMI FILTER
DESIGN PROCEDURE ................................................................................................................ 84
5.1
INTRODUCTION .................................................................................................................. 84
5.2
EXPERIMENTAL SETUPS ..................................................................................................... 84
5.3
CASE STUDY #1: INVERTER-FED INDUCTION MOTOR DRIVE ............................................. 90
5.4
CASE STUDY #2: INVERTER-FED SYMMETRIC LOW POWER RESISTIVE LOAD .................... 98
5.5
CASE STUDY #3: DC MOTOR DRIVE SUPPLIED BY A DC/DC BOOST CONVERTER ........... 102
5.6
CASE STUDY #4: DC MOTOR DRIVE SUPPLIED BY A DC/DC BUCK CONVERTER ............. 108
CONCLUSIONS AND FUTURE DEVELOPMENTS ............................................................. 113
REFERENCES ............................................................................................................................. 115
PUBLICATIONS ......................................................................................................................... 124
5
LIST OF FIGURES
Figure I.1
- Main elements in the EMC.
Figure I.2
- Scheme of EMC Problems.
Figure I.3
- Electromagnetic disturbances related to the frequency bands.
Figure I.4
- Typical current or voltage waveform generated by an electronic power system.
Figure I.5
- Discrete spectrum of a train of trapezoidal pulses with T=2τ.
Figure I.6
- Spectral envelope of trapezoidal pulse train in Bode diagram.
Figure I.7
- Feedback type active filters. (a) Current detecting and voltage compensating. (b)
Current detecting and current compensating. (c) Voltage detecting and current
compensating. (d) Voltage detecting and voltage compensating.
Figure I.8
- Feedforward type active filters. (a) Current detecting and current compensating. (b)
Voltage detecting and voltage compensating.
Figure I.9
- Generic scheme of a common air cooled power electronic system.
Figure II.1
- CM and DM noise paths.
Figure II.2
- CM/DM voltage and current generated by a single phase power electronic equipment.
Figure II.3
- Conducted emissions measurement circuit.
Figure II.4
- Circuit scheme of the high voltage (HV) AMN (dual LISN).
Figure II.5
- Impedance ideal curve and measured impedance curve of the LISN 1.
Figure II.6
- Separation of CM and DM current by using a current probe.
Figure II.7
- Comparison between the ideal and measured CM (upper) and DM (lower) LISN
impedance.
Figure II.8
- Separation of CM and DM noise via hardware.
Figure II.9
- Block diagram of the time domain EMI measurement method.
Figure II.10 - Test bench used to conducted EMI measurements.
Figure II.11 - Comparison between CM EMI obtained by the software-based separation technique
and by RF measurement-based technique.
Figure II.12 - Comparison between DM EMI obtained by the software-based separation technique
and by RF measurement-based technique.
Figure III.1 - EMI filter circuit configurations.
Figure III.2 - Schematic representation of noise source and victim without (a) and with (b) filter.
Figure III.3 - Equivalent circuit of capacitors.
Figure III.4 - Bode plot of impedance Zc(f).
6
List of Figures
Figure III.5 - Equivalent circuit of a inductor.
Figure III.6 - Bode plot of impedance ZL(f).
Figure III.7 - Generic EMI filter configuration (a), CM equivalent circuit (b) and DM equivalent
circuit (c).
Figure III.8 - Steps of EMI filter design.
Figure III.9 - CM choke.
Figure III.10 - Winding angle example.
Figure III.11 - Electrical representation of a CM inductor.
Figure III.12 - Magnetic properties for ferrites, iron powder and metal alloys: permeability vs.
frequency.
Figure III.13 - Magnetic permeability curves versus frequency (a), Magnetization curves (b) and the
saturation induction versus temperature (c) of the VITROPERM 500F and a typical
Mn-Zn ferrite.
Figure III.14 - Hysteresis loop of VITROPERM 500F and a typical ferrite.
Figure III.15 - Comparison of magnetization losses of typical materials for CM choke and DM
inductance.
Figure III.16 - CM choke set up by using an N30 ferrite core (left) and a VITROPERM core (right).
Figure III.17 - 100 nF capacitor impedance (a) 250Vdc ceramic capacitor (measured data) and (b)
300Vac,1000Vdc polypropylene capacitor (datasheet).
Figure III.18 - Measured impedance module (upper) and phase (lower) of a 47µF electrolytic
capacitor with nominal voltage equal to 160V and 400V.
Figure III.19 - Measured impedance module (upper) and phase (lower) of a 47µF electrolytic
capacitor with nominal voltage equal to 160V of different manufacturers and for
different application fields.
Figure IV.1 - Concept of the optimized EMI filter design procedure.
Figure IV.2 - Flowchart of the optimized EMI filter design procedure.
Figure IV.3 - Screenshot of the web page for ODEF application download.
Figure IV.4 - Screenshot of ODEF application: Noise Profile tab.
Figure IV.5 - Screenshot of ODEF application: Computation tab.
Figure IV.6 - Screenshot of ODEF application: Extra tab.
Figure V.1
- Scheme of the experimental rigs: (a) case study #1; (b) case study #2; (c) case study
#3; (c) case study #4.
Figure V.2
- Cyclone III FPGA Starter Board equipped of the Nial Stewart GPIB expansion board
used in the experimental setups.
Figure V.3
- Board with the display SSD used in the experimental setups.
7
List of Figures
Figure V.4
- MIL-STD-461F: CE102 limit (EUT power leads, AC and DC) for all applications.
Figure V.5
- CISPR 25: Limits for broadband conducted disturbances (peak detector).
Figure V.6
- View of the PWM induction motor drive experimental setup.
Figure V.7
- CM and DM EMI generated by inverter-fed induction motor drive.
Figure V.8
- Photo of conventionally designed EMI filter (on the left) and optimized EMI filter (on
the right), in case study #1.
Figure V.9
- Comparison of optimized and conventionally designed EMI filter performance (case
study #1).
Figure V.10 - Distribution of all feasible configurations (case study #1).
Figure V.11 - Distribution of the best 15 configurations (case study #1).
Figure V.12 - Scatter plot of the best 15 configurations (case study #1).
Figure V.13 - Volume of the best configuration for each number of stages (case study #1).
Figure V.14 - Distribution of the best 100 configurations for different n. of stages (case study #1).
Figure V.15 - Volume variation of the best design for increasing CM attenuation (case study #1).
Figure V.16 - Number of stages of the best design for increasing CM attenuation (solid line). CM
core index of the best design for increasing CM attenuation (dashed line). - case study
#1.
Figure V.17 - CM and DM EMI generated by inverter-fed symmetric low power resistive load.
Figure V.18 - Photo of conventionally designed EMI filter (on the left) and optimized EMI filter (on
the right), in case study #2.
Figure V.19 - Comparison of optimized and conventionally designed EMI filter performance (case
study #2).
Figure V.20 - CM and DM EMI generated by a DC motor drive supplied by a DC/DC boost
converter.
Figure V.21 - Comparison of components used to EMI filters setup (case study #3).
Figure V.22 - Comparison of optimized and conventionally designed EMI filter performance (case
study #3).
Figure V.23 - Distribution of feasible configurations without extra LDM (case study #3).
Figure V.24 - Distribution of the best 30 configurations without extra LDM (case study #3).
Figure V.25 - Scatter plot of the best 30 no extra LDM configurations (case study #3).
Figure V.26 - Volume of the best configuration for each number of stages (case study #3).
Figure V.27 - Distribution of the best 100 configurations for different n. of stages (case study #3).
Figure V.28 - CM and DM EMI generated by DC motor drive supplied by the DC/DC buck
converter 1.
8
List of Figures
Figure V.29 - CM and DM EMI generated by DC motor drive supplied by the DC/DC buck
converter 2.
Figure V.30 - Comparison between CM EMI generated by the DC motor drive supplied by the buck
converter 1 (solid line) or by the buck converter 2 (dashed line).
Figure V.31 - Comparison between DM EMI generated by the DC motor drive supplied by the buck
converter 1 (solid line) or by the buck converter 2 (dashed line).
Figure V.32 - Measured EMI with and without EMI filter (case study #4).
9
LIST OF TABLES
Table II.1
Performance indices.
Table III.1
Filter topology selection based on impedance mismatching criterion.
Table III.2
Tables of AWG wire sizes (solid wire).
Table III.3
Comparison of different magnetic cores characteristics to set up a LCM=0.8 mH.
Table V.1
Input data for ODEF application – Case study #1.
Table V.2
Comparison between optimized and conventionally-designed EMI filters (Case study
#1).
Table V.3
Input data for ODEF application – Case study #2.
Table V.4
Comparison between optimized and conventionally-designed EMI filters (Case study
#2).
Table V.5
Input data for ODEF application – Case study #3.
Table V.6
Comparison between optimized and conventionally-designed EMI filters (Case study
#3)
Table V.7
Input data for ODEF application – Case study #4 with buck converter 1.
Table V.8
Features of the optimized EMI filter (case study #4).
10
LIST OF ACRONYMS
AEF
active EMI filter
AMN
artificial mains network
AWG
american wire gauge
CISPR
Comité International Spécial des Perturbations Radioélectriques (International Special
Committee on Radio Interference)
CM
common mode
DAEF
digital active EMI filter
DFFT
discrete fast Fourier transform
DM
differential mode
DPDT
double pole - double throw
DSO
digital storage oscilloscope
EEPROM
electrically erasable programmable read only memory
EM
electromagnetic
EMC
electromagnetic compatibility
EME
electromagnetic emission
EMI
electromagnetic interference
EPC
equivalent parallel capacitance
EPR
equivalent parallel resistance
ESL
equivalent series inductance
ESR
equivalent series resistance
EUT
equipment under test
HF
high frequency
HSMC
high speed mezzanine card
HV
high voltage
IEC
International Electrotechnical Commission
IL
insertion loss
LISN
line impedance stabilization network
PLLs
phase-locked loop
PSD
power spectral density
PWM
pulse width modulation
QP
quasi peak
11
List of Acronyms
RCFMFD
random carrier-frequency modulation with fixed duty cycle
RF
radiofrequency
RPWM
random pulse width modulation
SMPS
switched mode power supply
SRF
self resonant frequency
SSD
seven segments display
SSRAM
synchronous static random access memory
VNA
vector network analyzer
12
INTRODUCTION
The switching power converters are used in a broad variety of applications, from the consumer
electronics to the DC distribution systems, from the vehicle applications (road vehicles, marine
vehicles, aircraft) to the industrial automation. In each of these application fields, the conversion
systems which present more compact size and reduced weight, at the same power, are strongly
required in relation to stringent design constraints. In this context, the optimization of the power
density of the converter becomes an essential requirement. The increase of the switching frequency of
the static devices allows an improvement of the power density, thanks to the possibility of reducing
the sizes of the energy storage passive components (inductors and capacitors). On the other hand, the
increase of the switching frequency determines, with high probability, the generation of more relevant
conducted electromagnetic interferences (EMI) in the frequency range 150 kHz – 30 MHz. In
particular, the high switching frequency is responsible for several serious problems affecting both the
reliability and the electromagnetic compatibility of the systems of which the converter is part. For this
reason, noise mitigationg is, more than ever, one of the major issues in power electronic system
design, particularly when dealing with stringent standard regard the maximum emission limits, which
however are mandatory for the marketing of these systems.
EMI filters are the most efficient among the different possible solutions to mitigate the conducted
electromagnetic interferences. On the other hand, EMI filters are part of the power electronic
converters and they have significant impact on the overall converter volume and weight. In order to
take on this issue, besides satisfying EMI limits, a further optimization in terms of filter size and
weight during the design stage is advantageous to maximize the overall converter’s power density.
The identification of the configuration leading to the best power density, in terms of minimum
volume/weight, is a nontrivial task. The conventional design of EMI filters disregards the power
density issue. The trial and error approach requires a significant effort in terms of time spent and it
does not guarantee the optimal choice of filter configuration in order to obtain the maximum power
density. For this reason, an automatic optimized design procedure of discrete EMI filters has been
developed. Once the power electronic converter characteristics are known and based on databases,
suitably set up, of commercially available devices for the realization of EMI filters, the optimized
procedure enables EMI engineers or scientists to obtain the best EMI filter configuration in term of
power density.
On the basis of the developed automatic design procedure, an interactive software, ODEF
(Optimized Design of EMI Filters), has been developed to make the new design procedure more
13
Introduction
accessible to EMI designer. Moreover, the developed application is provided of a graphical interface
which allows to analyze and compare simultaneously different EMI filter designs. The optimization
algorithm can be used as a EMI filter design tool but also as a tool for the analysis of the EMI filters
performance.
The thesis is organized as follows.
The first chapter gives an overview of the conducted electromagnetic interference issues and the
power density issues in power electronic converters. A literature review and a summary of the main
mitigation strategies adopted to suppress the conducted EMI are provided and the scopes of actions for
a power density improvement are explained.
Chapter II discusses the characterization of the EMI noise, such as the difference between the
common-mode (CM) and differential-mode (DM) noise. The CM and DM noise paths are evaluated
and three CM and DM separation techniques are described. In particular, a software-based CM/DM
separation technique, developed within the PhD work, has been validated by comparing the measured
EMI spectra with those obtained by measurements coming from a high bandwidth radio-frequency
current probe and a spectrum analyzer. Furthermore, the deviation of the results obtained by the two
techniques has been computed in terms of normalized root mean square error and normalized average
error.
Chapter III is dedicated to the conventional EMI filter design. In the first step, the criteria for the
correct choice of EMI filter topology and the real high frequency behavior of filter components, that
can heavily influence the filter performance, are discussed. Then the EMI filter general design steps
are presented. Finally, the chapter ends with some considerations on the material of filter components
and on their impact on filter performance and size.
Chapter IV presents the new optimized EMI filter design technique for the optimal and fast
selection of discrete EMI filter components and configuration, aimed at obtaining the minimum
volume/weigth. A general description of the ODEF implementation and functionality is given as well.
The results of EMI filters designs according to the optimized and conventional procedure in four
case studies are discussed in chapter V. A comparison of the obtained optimized filters with the
conventionally designed ones, is carried out in terms of volume, weight and performance. Futhermore,
an analysis of the fleasible configurations returned by the algorithm is performed, for some of the case
studies, by a series of comparative plots generated by ODEF application.
The end of the thesys contains the conclusions and the possible future developments.
14
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
CHAPTER I – Electromagnetic Compatibility and Power Density issues in
Power Electronic Converters
This chapter starts with an overview of the general concepts and main definitions of the
Electromagnetic Compatibility. It follows with the background of conducted electromagnetic
interference issues in power electronic converters. Then, a literature review and a summary of main
mitigation strategies adopted to suppress the conducted EMI are provided. Finally, also the power
density issues in power electronic converters are treated and the scopes of action for a power density
improvement are explored.
1.1
EMC: General Concepts and Definitions
Electromagnetic Compatibility (EMC) deals with electromagnetic problems existing between the
“devices” and the environment in which they are located.
The legislative decree 05/18/2016 no. 80 implements the Directive 2014/30/UE, drafted in date
26/02/2014, which provides the definition of electromagnetic compatibility as follows:
“Electromagnetic Compatibility is the ability of a device, equipment or system to function
satisfactorily in its electromagnetic environment without introducing intolerable electromagnetic
disturbance to anything in that environment”. According to the Directive 2014/30/UE, the term
“devices” indicate all electrical and electronic devices together with equipments and systems
containing electrical and/or electronic components.
The term EMC, covers both electromagnetic emission and electromagnetic susceptibility [1], [2].
The electromagnetic emission is referred to the disturbance level emitted by a device which can
degrade the performance of other devices operating in the same environment; the electromagnetic
susceptibility (or immunity) is the ability of a device to maintain the correct functional performance in
presence of external EM interference. Then an electromagnetically compatible system must satisfy the
following requirements:
• it does not cause interference with other systems;
• it must not be susceptible to electromagnetic radiation generated by other systems;
• it does not cause interference to himself.
The EMC study is focused on the generation, the transmission and the reception of electromagnetic
energy intended as a disturbance in relation to the correct functioning of the "devices". Therefore a
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
problem related to electromagnetic compatibility requires the identification of a source or emitter, of a
propagation path and the coupling channel and a receiver or victim, as shown in Figure I.1.
Figure I.1 - Main elements in the EMC.
It is possible to introduce a further distinction between natural (i.e. lightning, electrostatic
discharge) and artificial sources; the latter can be classified as intentional and unintentional sources.
An intentional radiation source is specifically designed to generate radiation to perform a specific
function (e.g. a mobile phone or a radio transmitter), while for an unintentional one, the emissions are
an undesirable consequence (e.g., the radiation emitted by a computer or a monitor).
Concerning the effects that electromagnetic radiation causes on receivers, a similar distinction
applies on them: if the received radiation generates a desired behavior it is called "useful signal"
(intentional receiver); instead, if the received radiation generates a malfunction, it is called
“interference signal” (unintentional receiver) or Electromagnetic Interference (EMI).
With regard to the electromagnetic interference effects, it is possible to observe that the EMI can
determine a simple reduction of the devices/equipment/systems performance, or a malfunction or fault
conditions of the same apparatus and, in certain critical applications, it can compromise irreparably
things and/or people safety.
It must be remarked that the intentional sources and receivers can generate or receive
electromagnetic radiation in frequency bands different from those typical of normal operation; even
then they must comply the electromagnetic compatibility requirements.
On the basis of EMI propagation mode, the EMI are distinguished in conducted and radiated
disturbances. The scheme in Figure I.2 summarizes electromagnetic compatibility problems [3]. It is
common to define the EMI study into four different groups: conducted emissions, radiated emissions,
conducted susceptibility and radiated susceptibility. The radiated emissions are the electromagnetic
waves which propagate into the surrounding environment due to irradiation of the currents circulating
in the conductor elements (e.g. cables or screens). It comes to radiated susceptibility if a component
acts as an antenna that intercepts the emissions generated by other systems. The conducted emissions
are undesired voltage or current signals which propagate from a system to another through the
connection cables (power cables, signal and communication cables); the sensitivity of a component to
this type of interference defines the conducted susceptibility. In fact, a variable signal that flows in a
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
conductor cable generates an EM field in the surrounding space and at the same time an EM field
induces an electrical signal on a conductor. Then, conducted and radiated phenomena are related.
The EMI sources can be located inside the system (internal problem or intrasystem problem), or the
interference can be generated by external sources (external problem or intersystem problem). A very
common interference source, internal or external in the system, is due to a signal which, although
specifically generated for a given circuit, also reaches one or more circuits in the system to which the
signal itself is not dedicated.
Figure I.2 - Scheme of EMC Problems.
According to the definitions listed above, the electromagnetic compatibility is related to the
generation, the transmission and the reception of the electromagnetic energy between the source and
the receiver by means a coupling path in which the interference is an unwanted phenomenon.
To prevent interference it is possible implement three strategies:
• to suppress the generated EMI;
• to make the coupling path less efficient as possible;
• to make the receiver less susceptible to interference.
It is therefore important to manage the generation of the electromagnetic radiation as well as the
susceptibility to it, during the design phase of the device. If the noise sources and the possible
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
susceptibilities are not taken into account in the initial design, it can result underperforming, expensive
and time-consuming procedures during the production and deployment process [4].
The standardization work on electromagnetic compatibility is spread mainly according to the
following definitions:
1. a typical electromagnetic environment (public networks, residential or industrial environment,
control areas plants, outdoor areas, etc.);
2. a compatibility level for each type of interference, given a specific interference level that has a
defined probability of being exceeded in a given environment;
3. a susceptibility level, for different categories of devices, given by the maximum interference
level that a device must be able to support maintaining its performance;
4. an emission limit level as the maximum interference level that a device can generate.
1.2
EMI issues in Power Electronic Converters
The power electronic converters can be used wherever it is necessary to modify the characteristics
of the waveforms related to the electrical energy conversion, for example, varying the voltage and
current levels, the waveform or the frequency [5], [6].
A power electronic converter is defined as the system consisting of one or more electronic
switching devices and, if necessary, transformers, filters and other auxiliary devices necessary for the
power electronic conversion. Electronic switching device is a component including one or more
conductive paths in a single direction, not actuated or controlled in bistable mode [6]. Often the
switching electronic devices used in power electronic converters are named as elementary conversion
unit.
The main power electronic converters can be classified on the basis of their fundamental functions,
on the basis of the converter switching mode and on the basis of the voltage (V) - current (I) plane
quadrants in which they can work.
According to the first classification criterion, the power electronic converters can be identified as
follows:
•
rectifier (AC/DC converter);
•
inverter (DC/AC converters);
•
AC/AC converter;
•
DC/DC converter.
The first two types of converter realize the conversion from AC to DC current and vice versa,
respectively. Using the AC/AC and the DC/DC converters, it is possible to realize the voltage
amplitude/polarity variation; furthermore the AC/AC converter makes also possible the variation of
the frequency and of the number of phases.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
A further distinction can be made among the AC/AC converters:
•
AC power regulators to modify the current characteristics keeping constant the frequency.
•
Direct frequency converters to modify the voltage, the frequency or the number of phases
without an intermediate energy storage circuit.
•
Indirect frequency converters that have an intermediate DC voltage connections. They modify
the voltage, the frequency or the number of phases including an energy storage device in the
intermediate circuit. In such converters, the output frequency is independent from the input
one.
DC-DC converters are also referred as direct regulators of continuous current (also called chopper)
and realize the DC voltage variation without employing any intermediate circuit. Except for some
types of rectifiers, which can also operate in an uncontrolled way, all converters require a controllable
elementary conversion unit.
With regard to the switching mode, it is possible to recognize the following types of converters:
•
Natural switching converters in which the switching event is imposed by an external circuit
and it occurs with a frequency equal to the network supply frequency.
•
Forced switching converters in which the switching event is imposed by a control operation on
the driving devices conditions of the converter. This event occurs with a frequency higher than
the network supply frequency.
•
Resonant or quasi-resonant converters in which the switching event occurs when the condition
of zero voltage and/or zero current on the component is verified [2].
Finally, taking into account the operating quadrants of the converters on the plane V-I, it is possible
to classify the following converters type.
•
Converters which allow the power flux in a single direction, therefore their operation is only in
the first quadrant.
•
Reversible converters (also called current converters) whose operation can be represented in
the first and second quadrant.
•
Bidirectional converters are composed of two reversible converters with the electronic devices
oriented in opposite direction, so as to obtain the possibility of reversal of both the current of
the voltage; their operation may therefore be represented in all four quadrants of the V-I plane.
Power electronic converters are particularly interesting systems in electromagnetic compatibility.
Due to non-linear effects of the static conversion devices and to the switching operation, power
electronic converters generate a wide range of electromagnetic disturbances. The generated noise
propagate towards the power supply network and to the load; then EMC problems occur [7].
In recent years, the power electronics development has contributed to progress in the power
converters technology and in the market deployment evolution. In particular, the progress achieved in
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
the power electronic components control, during their power on/off phase, has allowed to obtain a
drastic reduction of the turn-on and of the turn-off times of the switched voltage and current
waveforms. Then a relative increase of the switching frequencies has been obtained. Indeed, some
semiconductor power electronic devices controllable in turn-on and turn-off phase such as the GTO
(Gate Turn-Off Thyristors) exhibit turn-on and turn-off time of tens of microseconds, and switching
frequencies of some kHz. The BJT (Bipolar Junction Transistors) and the IGBT (Insulate Gate Bipolar
Transistors) exhibit turn-on and turn-off time lower than microseconds and allowing switching
frequency about to 100 kHz. Recently, power electronics market has been boosted by new high-speed
MOSFET (Metal-Oxide-Semiconductor Field Effect Transistors), as the wide-band gap devices based
on Silicon Carbide (SiC) or gallium nitride (GaN) [8], [9], allowing faster switching operation. These
devices exhibit turn-on and turn-off time of tens of nanoseconds and switching frequency of the order
of MHz. These devices are characterized by low switching losses and allow to obtain a beneficial
effect on the reliability.
The increase of the switching frequency of static devices allows to reduce the dimensions of the
energy storage passive elements (inductors and capacitors). Then, for the same power, more compact
conversion circuits are obtained with an increasing of the system power density. However, the increase
of the switching frequency leads to a significant extension in the harmonics frequency spectrum
produced by power electronic converters. For this reason power electronic converters are considered
unintentional sources of high frequency electromagnetic interference and they determine several
problems affecting both the reliability and the electromagnetic compatibility of the systems of which
the converter is a part.
Finally, it should be noted that the digital electronic devices of the control apparatus (in particular
of the processor with an internal clock of the tens of MHz) can also determine the radio frequency
interference emission. Even the driving circuits (drivers) of switching devices contribute to EMI
emission because they amplify the high frequency signals and their connection, being traversed by
high frequency current signals, radiate electromagnetic field.
Electromagnetic disturbances, in relation to their frequency content, can be related to well defined
frequency bands (Figure I.3). Following a distinction into subharmonic frequencies (below 50 Hz),
disturbances in the harmonic frequencies range (50 Hz to about 2 kHz), disturbances in the frequency
band between the acoustic frequencies and the radio frequency, radio frequency conducted
disturbances (in the frequency band 150 kHz - 30 MHz) and radiated interference (above 30 MHz).
20
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
Figure I.3 - Electromagnetic disturbances related to the frequency bands.
In order to characterize the high frequency spectral content of the generated noise by a generic
switching power electronic device, an analysis in the frequency domain of a series of trapezoidal
pulses can be usefully performed [1].
The trapezoidal pulse, shown in Figure I.4, represents a typical current or voltage waveform
generated by the power electronic circuits, where the variation speed of the signal is related to the
switching speed of semiconductor devices.
Each trapezoidal pulse is described by the amplitude A, the rise time τr, the fall time τf and a pulse
width τ of the 50% amplitude (Figure I.4). The time period of the pulse repetition is indicated as T.
Figure I.4 – Typical current or voltage waveform generated by an electronic power system.
As it is well known, a periodic function x(t) is expressible in Fourier series. The complex
exponential form of the Fourier series of a periodic function x (t) is defined as follows:

x(t )  c0   cn cos(n0t   cn )
(1.1)
n 1
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
where ω0 is the angular frequency of the signal and
c0  A

T
cn  2 cn
cn 
with positive integer n
1 t1 T
x(t )e jn 0 t dt

t
T 1
For a trapezoidal pulses series, the expansion coefficients have the following expressions:
A  jn o
cn   j
e
2n
(  r )
2
1
1


 sin( 2 n0 r ) jn 0 2 sin( 2 n0 f )  jn 0 2 
e

e
 1

1


n0 r
n0 f
2
 2

(1.2)
Equation (1.2) is not of immediate interpretation. By imposing that τr=τf in (1.2), the expression of
complex expansion coefficients cn becomes:
1
1
(  )
 sin( 2 n0 ) sin( 2 n0 r )  jn0 2 r
cn  A
e
1
T 1 n 
n0 r
0
2
2
Since 0 = 2/T, it results:


sin( n ) sin( n r )

T
T
cn  2 cn  2 A
r
T n 
n
T
T
(1.3)
Equation (1.3) allows to determine the discrete spectrum of the signal harmonics amplitudes when
τr=τf. It is clear that this discrete spectrum contains spaced rows of intervals equal to 1/T and that the
first zero occurs at n/T=1/. Figure I.5 shows the trend of the trapezoidal waveform spectrum in case
where /T ratio (i.e. the duty cycle) is equal to ½.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
Figure I.5 – Discrete spectrum of a train of trapezoidal pulses with T=2.
To obtain other information, the envelope of the spectrum previously identified can be analyzed by
means of a Bode diagram. By replacing f=n/T, the conversion of the discrete spectrum in its
continuous envelope versus frequency f is obtained with the following relation:
envelope  2 A
 sin(f ) sin( r f )
T f
 r f
(1.4)
The overall Bode diagram is the sum of the three diagrams:

diagram1  20 log10 (2 A )
T
sin(f )
diagram2  20 log10 (
)
f
diagram3  20 log10 (
sin( r f )
)
 r f
In the Bode plot the diagram 1 has a slope of 0 dB/decade and a level of 2A/T. The diagram 2
instead has two asymptotes, one with a slope of 0 dB/decade and a level equal to unity, the other with
a slope of -20 dB/decade, the cutoff frequency equal to 1/(). The diagram 3 likewise presents other
two asymptotes, respectively with a slope of 0 dB/decade (level equal to unity) and of -20 dB/decade,
with a cutoff frequency equal to 1/(r). Therefore, the overall asymptote is composed of three
segments (Figure I.6): the first with a slope of 0 dB/decade, the second with a slope of -20 dB/decade
and finally the third with a slope of -40 dB/decade.
Since r < , the first spectral envelope cutoff frequency will be equal to 1/() therefore related to
the trapezoidal pulse width . Instead the second cutoff frequency will be related to the rise time r.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
Figure I.6 - Spectral envelope of trapezoidal pulse train in Bode diagram.
By the Bode diagram shown in Figure I.6, it is deduced that:
 the spectral envelope overall level of a trapezoidal pulses train depends on both the amplitude A
and the duty cycle /T of the pulses sequence;
 the behavior of the spectral envelope at low frequencies depends on the pulse width ;
 the behavior at high frequencies is related to the rise time r and fall time f of the pulses.
By consideng for example the CMF20120D device which is a Silicon Carbide Power MOSFET
with high speed switching. This device exhibits the r = 38 ns and the f = 24 ns (datasheet data).
According to the analysis perfomed above, it results that this device generates high frequency
harmonics up to tens of MHz due to its rise time value. So, more relevant conducted EMI are
generated at high frequencies. From this perspective, the implementation of a proper mitigation
technique is a very crucial requirement.
1.2.1.
EMI Mitigation Techniques
Due to the switching operation, the power electronic converters are unintentional sources of high
frequency electromagnetic interference for equipment placed nearby. Therefore, EMI attenuation
systems are necessary to ensure both the reliability and the electromagnetic compatibility of the
system of which the power electronic converter is part. In particular, EMI containment techniques
should be developed to ensure the compliance with the emission limits imposed by the technical
standards which are binding for the marketing of those systems.
Technical literature provides a large number of contributions concerning the reduction of
electromagnetic noise [10]. One way to reduce the level of conducted noise is by ensuring that less
noise is generated by the noise source itself. On the other hand, the noise can be mitigated along the
noise propagation path by filtering and other means.
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Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
The noise generation can be reduced by a proper design of circuit layout and selection of circuit
components, a better switching-control scheme, and soft-switching transition technique.
The selection/design of appropriate components and/or better physical layout of the circuit to
minimize the EMI generation, require a particular attention during the initial design stage.
About the switching-control scheme, it is possible to select a low switching frequency resulting in
lower noise spectrum contribution at the high frequencies or to apply the more appropriate modulation
technique. A common compromise is to set the switching frequency value lower than the half of the
considered standard lower frequency limit [11], [12]. Of course, the proper operation of the switching
devices imposes the limitations on the increase of the transition times. In [13] the impact of the used
modulation technique on the generated EMI level in dc–dc power converters is analysed. A
comparative investigation is performed into the use of different random modulation schemes as
against the classic pulse width modulation (PWM). The effectiveness of randomization on spreading
those dominating frequencies, that normally exist in constant frequency PWM schemes, is evaluated
by power spectral density (PSD) estimations in the low-frequency range. Limited speed PWM driving
of the power switches with appropriate snubber circuits guarantees reduced conductive EMI.
However, this investigation shows that, among all the random schemes under consideration, the
random pulse width modulation (RPWM) and the random carrier-frequency modulation with fixed
duty cycle (RCFMFD) produce a minimum low-frequency harmonic spectrum and are, therefore,
considered the best choice for dc–dc converter applications.
Moreover it is possible to improve the current and voltage waveforms associated with the switching
of the power devices to reduce the EMI generated by the noise source. Snubbers, gate-drive
modifications and soft-switching techniques all fall under this category. Snubber circuits coul be
considered as low-pass filters; they allow to soften the switching transitions and also to aid in damping
the high frequency waveform oscillations during the switching. However the snubbers produce a slight
increase in overall power loss. In [14] and [15] the reduction of the generated EMI is obtained by the
implementation of the active voltage control (AVC); it is applied and improved successfully to define
IGBT switching dynamics with a smoothed Gaussian waveform. The general idea of AVC is to use a
high-speed feedback to force the collector–emitter voltage to follow a well predefined reference. In
this way a constant control of the collector–emitter voltage and voltage clamping may be achieved.
The high-performance proportional–derivative and multiple-loop AVC controller provides a practical
solution to force the IGBT voltage to follow a smoothed Gaussian reference, so reducing the highfrequency EMI generation. In fact, the Gaussian reference implies a very high dv/dt in the middle
slope, but the duration is short, and it is part of the S shape. The switching speed is not limited by the
Gaussian reference but mainly by the drive capability of the driver. The successful shaping of IGBT
switching is dependent on two aspects: the controller and the quality of the reference. The Gaussian
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
shaping requirement for the controller is very stringent. The Gaussian waveform sampling is limited
by the FPGA and the digital-to-analog conversion rate. A high-quality reference generation might
further improve the shaping performance.
Another method of reducing di/dt and dv/dt in order to reduce EMI generation is to use softswitching techniques. The soft-switched converters have generally reduced conducted EMI. However,
the soft-switched converters may require auxiliary resonant circuits and extra devices with additional
control complexity that can increase the converter cost, can decrease its reliability, and can create extra
losses that can adversely affect the efficiency [16].
A different solution to limit the EMI would be to make the receiver less susceptible to the
interference. In other words, rather than to limit the signals amplitude that interfere with the noise
receiver, it could make sure that they have the minimum possible effect on the correct operation of the
receiver.
Another EMI mitigation strategy would be to make the coupling path as less efficient as possible.
This could be achieved by placing the receiver in a metal box (shield) or by using the shielded cables
to realize all connections between the devices. However, this solution is very expensive and the
obtained performance are often below the expectations. The use of filters allows to modify more
efficiently the characteristics of the noise propagation path so as to reduce the noise at the receiver
end. This filter can be a separate unit kept on the front end or it can be integrated into the power
converter itself. This leads to a further subdivision into external EMI filters and internal filters. In the
internal filters, the noise currents are internally circulated within the converter itself by layout or
topology modifications. The external EMI filters are adopted more frequently and they can be further
classified into passive and active filter types [17] - [26]. Typically, an external EMI filter acts as a lowpass filter. It has a negligible effect at the power frequency, while it offers large attenuation to the
noise currents in the conducted EMI frequency range.
The discrete passive EMI filters are generally realized with capacitors and inductors connected
according to different topologies in single or multi-stage configuration. Each filter configuration can
result useful for some applications while it can not ensure the required performance for others
applications. Therefore, it is important to choose the filter configuration depending on the system in
which it will be adopted. The main advantage of passive filters is the relative “simplicity” of the
design and their pratical implementation while their main limitation is related to their high frequency
performance degradation due to the parasitic phenomena [27] - [29]. Moreover, the passive EMI filter
performance are closely related to the EMI source and the EMI receiver impedances. The passive EMI
filter design is considered a “black art” because little is known about the EMI source; the interaction
between the EMI source and the EMI filter impedances can cause poor noise attenuation. A proper
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Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
EMI filter design must suitably takes into account the criterion of maximum impedance mismatching
between the source and the receiver [19], [69] - [71].
The active EMI filters (AEFs) use active electronic circuits to cancel or suppress the conducted
noise and they are possible alternatives to bulky passive EMI filters. There can be numerous ways of
implementing active filters in different applications, but the same theory of operation applies. The
function of an active filter is the detection and the compensation of the noise signal (current or
voltage) from the noise source or receiver. Different active filters topologies have been proposed in
technical literature, depending on the method of compensation. There are two groups of active filters:
the first one, referred as the feedback-type active filter, detects noise at the receiver while the other
one, referred as the feedforward-type active filter, detects the noise at the noise. Active filters can vary
in type according to different detection and compensation signals. Figure I.7 and Figure I.8 show,
respectively, possible configurations of feedback and feedforward type active filters according to
different measures of detection and compensation signals. In the figures, zs is the impedance of a noise
receiver, which measures noise power caused at the noise source is; zn is an internal impedance of the
noise source in, ic and vc are the compensation signals. The impedance relationship between source and
receiver must be taken into account when selecting and locating various active filters.
In the systems employing the power electronic converters, the active EMI filters have been
employed for both the input and output EMI mitigation compensation [22].
Figure I.7 - Feedback type active filters. (a) Current detecting and voltage compensating. (b) Current detecting and
current compensating. (c) Voltage detecting and current compensating. (d) Voltage detecting and voltage
compensating. [20]
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
Figure I.8 - Feedforward type active filters. (a) Current detecting and current compensating. (b) Voltage detecting
and voltage compensating. [20]
The main advantage of the active filters is their very compact dimensions due to their small size
integrated components; however the high frequency performance is limited by the bandwidth of the
operational amplifier (op-amp) that makes the filtering action less efficient. Some studies have in fact
shown that the active filters have good performance only up some MHz. In [24] the operational
amplifier is used at unity gain to mantain its maximum bandwidth and the desired gain of the active
filter can be achieved by the injection transformer. However, depending on the type of the operational
amplifier selected for the application, there is always a minimum phase error and a distortion of the
input signal at very high frequencies due to the parasitic elements inherent in the operational amplifier
itself. Since the AEF requires op-amps with good high frequency characteristics and wide bandwidth,
the traditional active EMl filters are expensive. A recent paper [24] proposes an improved topology
structure for an active filtering with ordinary op-amps to suppress the CM interference. It includes two
same closed-loop feedback circuits. Besides, the cost is less, the filtering effect and stability of the
two-stage active filter are better.
Instead in [26] a method to enhance the op-amp gain bandwidth product is presented to improve the
active EMI filter performance.
Another disadvantage is related to the possible instability of the entire system that can limit the
filter dynamic performance [30], [31], [38], [39]. It should be noted that, generally, the presence of
active components reduces the filter reliability. Furthermore, it is necessary a carefully design of the
entire filtering system so avoiding possible interaction between the sensing/injector circuits and the
input/output of the active filter [25].
Even if the AEFs are more compact, their high speed active components require an additional
power supply, which in turn, will increase the size and weight of the active filter and their integration
is difficult. Therefore, active EMI filters are still not widely accepted by the industry. To solve the
problem related to the power supply of the active components, a novel AEF topology used for DC-DC
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Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
power converters is proposed in [37]. The most distinguished feature of this filter is that it shares the
same power supply with the DC-DC converter; no power is supplied to the active part of the proposed
filter, only the passive components are used. The size of the filtering circuit as well as its power supply
cell is very small and compact. It is easy and feasible to be integrated in the future commercial large
scale manufacture.
The performance versus cost reduction trends of the digital circuits has made possible their
application for power converters digital controller techniques. They are usually based on FPGA
technology that exploits their mathematical oriented resources. Some authors propose FPGA-based
EMI suppression techniques, referred to as digital active EMI filter (DAEF) [32] - [34]. The DAEF
presents stronger competitive application in medium to high current converters. The size and the
losses of the passive EMI filter are proportional to the rated current and voltages of the power
converter. Hence, the DAEF provide a feasible solution to overcome these drawback with good
attenuation performance. The conducted noise signal is the noise voltage that is sensed through an RC
high-pass circuit with the cutoff frequency tuned to the lower spectrum frequency of the conducted
emission standard. The sensed noise voltage is sampled by using high-speed analog-to-digital
converter (ADC) in order to be processed through a phase reversal algorithm. The discrete-time noise
signal is then reconstructed by using a digital-to-analog converter (DAC). The output signal of the
DAC is then electrically injected at input leads of the power converter, for the EMI noise suppression;
by an injection circuit which is a simple low-pass filter tuned to the higher frequency spectrum of the
conducted emission standard (30 MHz). ADC and DAC devices with high bit resolution are necessary
to achieve an adequate signal sampling. If this requirement is not satisfied, a phase error between the
sensed signal (sampled) and the injected signal (reconstructed) occurs, and consequently, a significant
degradation in the DAEF performance will result. However, the cost of the DAEF remains an
important disadvantage as compared to the passive EMI filter counter-part.
Finally, it is possible to combine more than one approach to come up with a “hybrid” mitigation
approach. For example, the active and passive filtering techniques may be combined to develop a
“hybrid” filter [35] - [39]. In the hybrid filters, the active filtering part mitigate the conducted EMI at
low frequencies (it provide good noise attenuation for the first several harmonics of the switching
frequency) and the passive filtering part to mitigate the conducted EMI at high frequencies. This
approach allows to exceed the limits related to the passive and active filters with satisfactory results
about the filter performance and the realization of a compact layout. Thus, the active filter can
significantly reduce the size of the passive filter whose cut-off frequency can be set at much higher
frequency. Hovewer the hybrid filters imply an inevitable increase in the complexity of the filtering
system.
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Taking into consideration the EMI mitigation techniques described above, it is possible to conclude
that an external front-end passive filter is the most-established method to meet the EMI standards and
it is extensively used at present. The main drawback of a passive filter is its large volume.
1.3
Power Density issues in Power Electronic Converters
The more increasing development of the power electronic converters in a wide range of
applications requires, besides the electromagnetic compatibility compliance, some improvements in
terms of higher efficiency, lower losses, lower volume, lower weight and lower production costs. A
high effiecient power converter ensures a good utilization of the energy resources and a low operating
cost. Low losses are basic requirement to enable a compact realization, which also allows a flexible
deployment of the converter system.
The integration of the power electronic converters in the final application is very common in many
applications such as the variable speed drives, that are used in a range of the industrial systems, in the
hybrid vehicles and in the More Electric aircraft. This trend allows to reduce the installation cost but
the converter volume is strongly limited by the main dimensions of the load system. In addition to the
low volume requirement also the low weight is very important in mobile systems applications to
facilitate the installation, handling and maintenance operations of the power converter [84]. High
power density power electronic converters become increasingly essential for future markets.
The converter design is a complex engineering due to the interaction of many aspects: a designer
must choice the more appropriate design among various possible designs and technologies, finding the
optimal allocation of each component in terms of its mass and its volume and for each component
must meets the electrical and thermal specifications. Therefore, an efficient design considers
simultaneously the system-wide electrical, mechanical and thermal problems; it is not enough to
design each component individually based on its electrical specification alone. A generic power
electronic converter is composed of :
– the power semiconductor devices,
– the modulation and control circuits,
– the power passive components (filters and transformers),
– the cooling system,
– the interconnections and the packaging.
It is necessary to introduce the power density concept. Power density of power electronics is a
Figure of Merite (FoM) to compare the technological status and performance of power electronic
converters. This parameter serves to characterize the degree of compactness of a converter or the
volume required for realization at a given rated power. The power density, ρ, is defined by the ratio of
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Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
the power output, Po, by the total volume, Vol, where the total volume is typically a factor of two more
than the sum ofthe partial volumes ΣVoli.
P
P
o
ρ= Volo = ∑ 𝑉𝑜𝑙
(2.1)
𝑖
A side view of a common air-cooled power electronic converter is shown in Figure I.9. It is evident
that the passive components (capacitors and inductors) and the cooling system, if air volume is not
taken into account, have major impact on the bulky power electronic system [40]-[42]. They are
regarded as the main barriers to the improvement power density. However a proper evaluation of the
power density of a power converter should take into account also the volume requirements of the EMC
filter, the power semiconductors with driver circuitry and auxiliary power supply, as well as the
control electronics and the housing in the construction volume Vol [44], [45]. Unfortunately, in the
literature the heatink is often not taken into account or the EMI filter is omitted.
Figure I.9 – Generic scheme of a common air cooled power electronic system.
1.3.1
Scopes of action for the power density improvement
Since 1970, power density of power electronic converters has been approximately doubled every
decade. This evolution was mainly driven by the increase of the switching frequency by a factor of 10
every ten years [84]. At present, an ever more radical increase of power density continue to be
required.
One of the possible action regards the power inductors design in the switched-mode power supply
(SMPS). The losses and the consequent temperature rise are the main intrinsic issues of power
inductors operation. Commonly, the inductors are designed so that they can work in the weak
saturation region but, in recent years, inductor saturation has been the subject of several scientific
investigations. Some authors have experimentally verified that smaller volume inductors working in
partial saturation, could help in achieving more compact SMPSs with an acceptable amount of power
losses; in this context, it is necessary to assess the inductor saturation effects. No useful information on
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the real magnitude of the current ripple for inductors working in partial saturation and how to
efficiently analyze the sustainability of such operating conditions are given in the scientific literature
and in the inductors datasheets. Recently, some authors have proposed [46] a method for ripple
analysis of saturated inductors, in order to allow the investigation of effective SMPS design solutions
with minimum size inductors, as well as the identification of optimum design tradeoff of the
parameters of the inductors. In [47], the potential impact on the reduction of the volume and the
weight of power inductors allowed by the adoption of partial saturation operation in Aerospace Power
Supply Units, is described. Although a traditional bigger core non-saturated inductor presents lower
power losses, current ripple and temperature rise, a smaller core inductor working in a partial
saturation may operate with acceptable power losses, current ripple and temperature rise.
Another scope of action regards the use of power devices with higher switching speed [40]. The
switching frequency increase allows to decrease the volume and weight of the passive energy storage
elements (inductors and capacitors) and thus to decrease the converter's global volume.
However, the switching frequency increase determines a power losses increase of the power
semiconductors and of the magnetic materials; this choice can finally leads to a thermal limit since a
minimum volume is reached and no more energy can be dissipated from the surface area [43]. The
power losses lead to worst performance, to a cost increase and to a minor power density of the
converters. The power devices normally take large share of system losses and they are directly coupled
to the cooling system; then they have major influence on the cooling system size. Advanced magnetic
materials, dielectric materials, wide bandgap devices with better electrical and thermal properties are
investigated to allow a wider operating frequency range.
Besides the power losses increase, higher switching frequency can determine more relevant high
frequency conducted EMI, since the higher magnitude harmonics (1st, 2nd, etc) of the generated noise
to be attenuated are located close to or within the frequency range limited by the reference EMC
standards.
In [45] the implication of the switching frequency and the power rating on the converter weight is
shown. In particular, the relationship between the per-unit weight of EMI filter, the switching
frequency and the power level is analysed: for the same switching frequency, as the power rating
increases, the EMI filter weight increases; for the same power level, the weight is not linearly
dependent on the switching frequency. Then the optimal switching frequency could be chosen taking
into consideration the EMI filter weight: for a power level higher than 10 kW, the tradeoff between
cooling and passive components is more important for converter weight.
Instead in [48], based on the low frequency attenuation requirement, a smaller EMI filter size can
be obtained by pushing the switching frequency higher. However, in the hardware implementation the
real limitation comes from the spectrum at high frequencies. In fact the EMI filter performace are less
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Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
effective at 20 MHz - 30 MHz range due to the variation of the permeability of the magnetic material
with the frequency, due to the parasitics phenomena of the EMI filter and to the inter-component
coupling.
The current ripple influences the harmonic content, increasing the noise amplitude. In [49] an
analysis towards the influences of the current ripple and the switching frequency on the overall losses,
the minimization possibility and the EMI filter design is presented.
The EMI filters can contribute substantially to the volume and the weight of the power converter;
then the optimization of the EMI filter size is an important requirement in the design stage. There are
some scopes of action to improve the EMI filter power density and conseguently of the power
converter. They are listed in the following.
-
Choice of the converter topology based on EMI filter volume and weight. Each converter
topology generates different EMI, harmonics and it is characterized by different loss
performance and therefore it has different impact on the convert power density [50].
-
Optimal switching frequency. Depending on the application field and then on the EMC
standard frequency range, the switching frequency value can have a major impact on the EMI
spectrum amplitude, and then on the EMI filter size, if the fundamental and the other
harmonics are inside the standard frequency range. The choice of the optimal switching
frequency allows to reduce the EMI filter size and weight [12].
-
Optimal number of filter stages. The optimum number of filter stages depends on the required
attenuation value, on the design frequency and on the rated power [90]. A multistage EMI
filter configuration can occupy a smaller volume than a single stage one. Then the evaluation
of the optimal number of the EMI filter stages allows to increase its power density.
-
High performance magnetic materials for the inductance implementation. The use of high
performance magnetic materials allows to achieve high inductance value with a smaller
number of turns and consequently a reduction of the number of stages, the size and the weight
of the filter [86].
-
Integrated EMI filter. These integrated structures use the printed circuit board technology to
realize the filter components, implementing techniques to cancel or to compensate the
parasitic phenomena, and they use appropriate packaging technologies to obtain better
performance at high frequencies and more compact layout [51]-[59]. The distinctive feature of
the integrated filters is a planar structure that integrates inductors and capacitors. This
structure consists of alternating layers of conductors, dielectrics, insulators and magnetic
materials with characteristics similar to the discrete components ones. The integrated EMI
filter presents some limits due to the materials, the electromagnetic aspects, the structure and
the limits related to the material processing technologies. These problems have to be
33
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter I - Electromagnetic Compatibility and Power
Density issues in Power Electronic Converters
considered during the design stage of integrated EMI filters. However, the integrated EMI
filter could not ensure good performance in the overall frequency range.
On basis of the considerations given in this section, it is evident that the power converter design
oriented to the power density is very complex. In fact, an approach can improve the power density of a
converter component but it can also deteriorate the performance or cause the size increase of other
converter components.
It is of considerable importance the correlation between the electromagnetic compatibility and the
power density issues in the power electronic converters. The implementation of a technique to solve
these two issues at the same time has been the subject of the research activity conducted during the
PhD course. It will be described in the following chapters.
34
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
CHAPTER II – EMI analysis
2.1
Introduction
Electromagnetic interference (EMI) emission is an important matter for any electric and electronic
equipment. When the noise emission of an equipment fails to satisfy the Standard limits, it is
necessary to adopt solutions to reduce the noise emission level. Measured emissions are a mixture of
common mode (CM) and differential mode (DM) noise. Furthermore, the design procedure for EMI
filters is usually divided in CM and DM filters design. Therefore, it is very important to measure
separately the two modes in order to design an efficient EMI filter.
In this chapter the CM and DM noise paths are evaluated and CM and DM separation techniques
are described. They can be roughly classified into three main groups: separation technique using RF
current probes, hardware-based separation technique and software-based separation technique. In
particular, the validation of the software-based CM/DM separation technique has been done by
comparing the results with those obtained by measurements coming from a high bandwidth RF current
probe and a spectrum analyzer. Furthermore, the deviation of the results obtained by the two
techniques has been computed in terms of normalized root mean square error and normalized average
error.
2.2
Conducted EMI and Noise Propagation Paths
The conducted EMI noise is usually decoupled and characterized by two noise components:
- Common Mode, defined as the noise flowing between the power circuit and the ground.
- Differential Mode, defined as the current flowing the same path as the power delivery.
The CM and DM noise propagation paths are shown in Figure II.1.
Figure II.1 – CM and DM noise paths.
35
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
Conducted EMI can propagate through a coupling channel given by metallic planes used for the
equipment ground connection or through a coupling channel generated from the equipment power
supply.
Regarding the emissions that propagate through the supply network, it is necessary to distinguish
between internal network noise sources and external noise sources. For example, among the first, there
are both impulsive and non-impulsive overvoltages, whereas lightning phenomenon and switching
transients, due to electronic converters used for the network voltage regulation, are external noise
sources.
To better understand the relationship between the CM and DM currents, a single-phase power
application where the mains cable of the EUT (Figure II.2), can be considered. It consists of three
parallel conductors: phase, neutral, and ground. The EUT is supplied by the power source by an
artificial mains network (AMN), that consists of two line impedance stabilization networks (LISNs,
described in section 2.3). Therefore, from the power flow point of view, the EUT is the load but from
the conducted EMI point of view, the EUT is the source because it produces the noise.
Figure II.2 – CM/DM voltage and current generated by a single phase power electronic equipment.
Sometimes the power cable might consist of only two conductors, phase and neutral, in which case
the EUT is floating. The two wires P and N are characterized respectively by two voltage levels VP
and VN and the currents flowing through the phase and neutral conductors are denoted with IP and IN,
respectively. These currents can be decomposed into two components, which are referred as the CM
current ICM and the DM current IDM. Then:
36
Power Density Optimization of EMI Filters for Power Electronic Converters
I P  I CM  I DM ,
I N  I CM  I DM ,
Chapter II – EMI analysis
I G  2I CM
I CM 
IP  IN
,
2
I DM 
IP  IN
2
(2.1)
VCM 
VP  V N
,
2
VDM  VP  VN
(2.2)
Therefore the CM and DM emissions can be regarded as the two components of an electromagnetic
disturbance. Indeed, given a conducted electromagnetic noise generated by a power electronic
equipment, it is possible to identify:
-
DM noise component (DM current) that flows from the P conductor closing on the N
conductor, through the parasitic capacitances between the conductors;
-
CM noise component (CM current) flowing on the P and N conductors and closes again on the
ground line, through the parasitic capacitances between the aforementioned conductors and the
equipment parts connected to the ground line.
On the basis of the these considerations, it can be assessed that CM currents are equal in magnitude
and have the same direction in both conductors while DM currents are equal in magnitude but opposite
in direction in the two conductors.
2.3
CM and DM EMI Separation Techniques
In conducted noise compliance tests the CM and DM noise components are irrelevant. However,
they are of basic importance in the design and in the analysis of passive EMI filters which are one of
the most common possible solutions to mitigate conducted EMI. The components of an EMI filter
attenuate CM and DM disturbances differently. For this reason, it is impossible to design or select an
appropriate EMI filter without knowing the levels of the CM and DM noise.
In this section, the CM and DM EMI measurement methods are described.
Firstly it is necessary to focus attention on an essential circuit for the conducted EMI measurement:
the artificial mains network placed between the power source and the EUT power supply cable (Figure
II.3).
37
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
Figure II.3 - Conducted emissions measurement circuit.
The measurement of conducted emissions consists in evaluating the noise currents which are
generated by an equipment and which propagate through its power supply cable. The measurement
made with only a current probe is not sufficient to evaluate conducted EMI since it is necessary to
obtain comparable measurements made at different environments. In fact, the power source impedance
is variable from plant to plant and it depends on frequency value; the variability of the load connected
to the EUT is a complication since it affects the intensity of the conducted interference. Consequently,
it is necessary to stabilize the impedance seen from the EUT to make consistent and comparable
measurements made at different environments and at a different time.
The LISN provides repeatability of the conducted EMI measurements by fulfilling several
important functions.
Firstly, it allows to obtain a constant impedance on the power supply of the EUT; this impedance
should also be constant for all frequencies of the conducted EMI measurement range. Generally, the
LISN impedance value must be equal to 50 Ω: i.e. it is the impedance value that the LISN must ensure
between the phase/neutral conductor and the ground plane.
The amount of noise existing on the external network of the energy distribution varies from place
to place; this external noise reaches the EUT via the power supply cable and, if it is not isolated in
some way (such as through a filtering or via the intrinsic operation of the apparatus itself), it is added
inevitably to the measured conducted emissions. The second objective of the LISN is to block
conducted emissions coming from the power supply, which are not due to the EUT in order to measure
only conducted emissions generated by the equipment under test.
Finally, the third objective of LISN is to facilitate the power flow from the mains to the EUT.
Figure II.4 shows the electrical circuit of AMN [60]. It is a dual circuit formed of two identical
LISNs placed in the same metal enclosure, one for each DC power line (positive and negative wire,
referred to ground). According to CISPR 25 standard requirements [61], each LISN has a 5 μH
38
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
inductance, L, a 1 μF capacitor, C1, on the mains side, and a 50 Ω resistance, R, as output to the
measuring instrument, with a series connected coupling capacitor C of 0.1 μF, on the EUT side. A 47
pF capacitor, parallel connected to the 50 Ω resistance, allows the LISN impedance to follow the ideal
impedance curve according to CISPR 25 in the whole frequency range of interest, as shown in Figure
II.5. In the same plot, the LISN ideal impedance curve, with the upper and lower dotted line curves,
defining the range of 10% admitted tolerance, is reported.
Figure II.4 - Circuit scheme of the high voltage (HV) AMN (dual LISN).
Figure II.5 - Impedance ideal curve and measured impedance curve of the LISN 1 [60].
As previously said, in order to properly design the EMI filter components, it is necessaryl to
evaluate separately CM and DM noise component in the frequency range of interest.
As already stated, the total current in the phase/neutral conductors is given, respectively, by the
sum and the difference of the CM and DM components:
39
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
I P / N  I CM  I DM
If a noise component is predominant on the other, it is necessary to operate on those passive EMI
filter components that act on the dominant component. This can take place on the whole frequencies
range, on certain ranges or on specific frequencies.
To determine which of the current component is dominant at a given frequency, the CM and DM
components should be known at every frequency in the range of interest. In this way, it can effectively
operate on the passive components to act directly on the dominant component in the frequency range
in which this occurs.
There are several CM and DM separation techniques; they can be roughly classified into three main
groups:
1. Separation technique using RF current probes;
2. Hardware-based separation technique;
3. Software-based separation technique.
The three separation techniques of CM and DM components will be described in detail in the
following sections.
2.3.1
Separation technique using RF current probes
The RF separation method is based on the use of a high bandwidth current probe to measure, via
spectrum analyzer, the CM and DM noise between the LISN and the disturbance source. This
separation method is a very reliable technique. It should be considered that, during the tests, the RF
current probe is exposed to the typically high supply current. Therefore, saturation or sensitivity
problems should be taken into account in the choice of a suitable current probe, especially in high
power-low voltage applications. This involves the use of expensive RF current probes. [62]
The measuring principle is shown in Figure II.6.
Figure II.6 - Separation of CM and DM current by using a current probe.
40
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
The spectrum analyzer (or EMI test receiver), connected to the current probe, measures twice the
CM current and twice the DM current depending on the wire placement respect to the probe. The EMI
instrument measures the voltage over a 50Ω resistor and scales it according to (2.3):
EMI dBV   20 log10 V  120
(2.3)
In order to obtain the current level in dB(µA) it is necessary to add the correction factor k of the
current probe to the voltage level in dB(µV):
  V 
I dBA  V dBV   k dB 
  A 
(2.4)
As well established, the LISN ideally provides a 50Ω load (per power wire) for the conducted EMI
in the frequency range between 150 kHz and 30 MHz.
Considering the propagation paths of the CM/DM currents, the CM current flows on an impedance
equal to 25Ω while the DM current flows on an impedance of 100Ω (corresponding to parallel or
series connection of the two impedances from 50Ω). As a result, the CM and DM EMI components are
defined by the following equations:
25 

EMI CM  20  log10  2 I CM    20  log10 2  I CM   20  log10 12,5
2 

(2.5)
100 

EMI DM  20  log10  2 I DM 
  20  log10 2  I DM   20  log10 50
2 

(2.6)
Although the LISN impedance should be equal to 50Ω in the frequencies range of interest, the CM
and DM LISN impedance values may be subjected to some variations. Figure II.7 shows the CM and
DM LISN impedances module values of Figure II.4, measured by a precision RLC meter Agilent
4285ALCR. It can be noted that the actual impedance of the LISN is not strictly constant within the
frequency range of the conducted EMI.
Therefore, a suitable frequency-dependent conversion factor, corresponding to the CM and DM
LISN impedance, respectively ZCM_LISN(f) and ZDM_LISN(f), must be used in order to obtain an accurate
measurement of the evaluation of the CM/DM noise. Then,
41
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
 Z CM _ LISN ( f ) 

EMI CM  20  log10 2  I CM   20  log10 
2


(2.7)
 Z DM _ LISN ( f ) 

EMI DM  20  log10 2  I DM   20  log10 
2


(2.8)
Figure II.7 - Comparison between the ideal and measured CM (upper) and DM (lower) LISN impedance.
2.3.2
Hardware-based separation technique
The second category of CM and DM noise separation techniques requires the use of a "noise
separator" between the dual LISN and the spectrum analyzer (Figure II.8) [1], [63].
The device carries out the sum or the difference of the phase and neutral conductors voltages, by
providing (obviously doubled value) only the CM or DM component. For this operation, the device
uses two broadband transformers: on the primary windings of transformers are applied phase and
neutral voltages on dual LISN outputs. The secondary windings of the transformers are connected in
series and a DPDT (Double Pole Double Throw) reverses the neutral voltage polarity: with the switch
in the normal position, the sum of the phase and neutral voltages is carried out, while after the
switching, the difference of the same signals is carried out. The spectrum analyzer, spanning the whole
42
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
frequency range of interest, provides a view about the conducted measures, allowing to do all
evaluations of the case study.
Figure II.8 - Separation of CM and DM noise via hardware.
The use of hardware separators (e.g., Paul Hardin separators, power combiners) may give phase
coherency problems and results degradation without a good characterization of the unavoidable modal
conversion of the complete set-up (i.e. the measured level at the DM output in case of pure CM
excitation and vice-versa). Moreover the EMI separators are suitable for low-medium power levels,
due to the saturation of their wideband transformer [63].
2.3.3
Software-based separation technique
Software-based separation techniques using time domain measurements are generally considered as
easy and cost-effective methods for designing the EMI filters, provided that a multi-port digital
oscilloscope with suitable sampling capability is available. Moreover, these techniques can reduce the
phase coherency problems without any dedicated hardware.
The signals on the phase and neutral conductors of the dual LISN, are simultaneously measured by
a multi-channel Digital Storage Oscilloscope (DSO) with a high sampling frequency. Figure II.9
shows a block diagram, synthesizing the software based CM/DM separation technique concept.
Considering that the signals retrieved from CH1 (VP) and CH2 (VN) are tied to CM and DM
voltages, VCM and VDM, referring to Figure II.9, by relationships in (2.9) and (2.10), respectively,
2VCM  VP  VN
(2.9)
43
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
VDM  VP  VN
(2.10)
their separation can be achieved in time domain by simply summing and subtracting the measured
signals. This operation is done by using the DSO processing features.
The concurrent signals measurement guarantees phase coherency; the high sampling
frequency/processing speed of the instrument and the suitability of time domain signals to be
processed by a DFFT (discrete fast Fourier transform) algorithm enables to obtain an accurate
frequency spectrum of the CM and DM EMI in the range 150 kHz - 30 MHz. In particular, according
to the Nyquist-Shannon sampling theorem [64], the DSO sampling rate should be at least two times
that of the highest signal frequency. The DFFT algorithm can be simply executed on a PC or
implemented on an embedded system as a part of an automated tool for EMI filter design.
Figure II.9 - Block diagram of the time domain EMI measurement method.
2.4
Experimental Validation of the Software based separation technique
As described in the previous section, in software-based CM and DM noise separation techniques
for conducted EMI, CM and DM signals are acquired in time domain and their frequency spectra are
computed using a software-based post-processing. In particular Welch and Bartlett periodograms are
proposed in [65] to represent the frequency spectra of CM and DM noise. In [66] a Labview-based
measurement system is used. Both contributions state the advantage of the technique and show a
qualitative comparison with RF measurement-based separation techniques in the case of low power
44
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
switched mode power supplies (SMPSs); on the other hand, the difference between CM and DM mode
obtained by the different techniques is not quantified.
In this section the software-based separation technique is validated against a consolidated full
experimental-based technique using a current probe, a spectrum analyzer and a correction factor of the
DC LISN versus frequency that has been measured experimentally, instead of using the theoretical
value of such a correction factor, which is constant. In this way the validation of the separation
technique is more accurate.
Differently from other contributions in technical literature, the difference between the results
obtained by the two techniques, has been evaluated in terms of normalized root mean square error
(NRMSE) and of normalized average error (NAVE).
The test bench shown in Figure II.10 has been used and it is composed as follows. A PWM IGBT
Voltage Source Inverter (VSI), equipped with a STGIPS10K60A module; an Altera Cyclone III FPGA
board equipped with a Nial Stewart GPIB expansion board, implementing the PWM modulator; a 48V
induction motor with a rated power of 1.1 kW. The VSI switching frequency is equal to 20 kHz. The
use of an intelligent module for the VSI allows a very compact layout of the power electronic stage. A
dual DC LISN with a voltage capability up to 600V has been set-up and used to measure the
conducted EMI.
In particular, a Tektronix TDS7254B 2.5 GHz – 20 GS/s - 4 channels is used for the time domain
measurements needed for the software-based CM/DM separation technique. A RF current probe R&S
EZ-17 that allows measurements in the frequency range 20 Hz – 100 MHz with a maximum DC
current of 300 A, and an Agilent E4402 (9 kHz – 3 GHZ) spectrum analyzer have been employed for
the RF measurement-based separation technique. A 10 kHz resolution has been chosen for both the
measurement instruments to obtain directly comparable frequency spectra.
An accurate time triggering of the signals is done with the aim of capturing an integer number of
the PWM switching periods. Moreover the DSO sampling rate and the length of measurement should
be chosen so that the resulting resolution of the DFFT is the same of the RF based measurements. In
this way the DFFT results can be directly compared with the RF measurements without the need of
windowing techniques, used to reduce spectral leakage issues. In this way a simplification is obtained
in the subsequent phase of EMI filter design since the use of appropriate margins to compensate the
reduction of signal spectra amplitude due to windowing can be avoided.
45
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
Figure II.10 – Test bench used to conducted EMI measurements.
Figure II.11 and Figure II.12 show the comparison between CM and DM EMI obtained with the
separation techniques described in sections 2.3.1 and 2.3.3. It is possible to observe that the obtained
results are very close to each other.
To quantify the difference of the software-based method results respect to the RF measurementbased ones, the NRMSE and the NAVE have been considered over the considered frequency range
[67]. Such indices have been normalized with respect to the maximum amplitude of the RF measured
emission, ymax, and they give a quantitative evaluation of the software based CM/DM separation
technique. Table II.1 summarizes the obtained performance indices by Eq. (2.11) and (2.12):
 n ( yˆi  yi ) 2 

NRMSE   i 1


n


ymax
(2.11)
  n ( yˆ i  yi ) 
 y
NAVE   i 1

 max
n


(2.12)
where ŷi and yi are respectively the RF emission amplitudes, measured to the i-th frequency, with
the RF current probe and with the software based separation technique and n indicates the number of
frequencies measured in the range of interest.
The low values of the performance indices demonstrate the goodness of the software based
separation technique and allows to consider it as a valid tool for the noise mode separation in cases
where high operating currents make critical the use of other methods.
46
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter II – EMI analysis
140
Magnitude (dBuV)
120
100
80
60
40
20
0
CM EMI obtained with the use of the current probe
CM EMI obtained with the software-based technique
6
10
7
10
Frequency (Hz)
Figure II.11 - Comparison between CM EMI obtained by the software-based separation technique and by RF
measurement-based technique.
140
Magnitude (dBuV)
120
100
80
60
40
20
0
DM EMI obtained with the use of the RF current probe
DM EMI obtained with the software-based technique
6
10
7
10
Frequency (Hz)
Figure II.12 - Comparison between DM EMI obtained by the software-based separation technique and by RF
measurement-based technique.
Table II.1
Noise Mode
CM
DM
PERFORMANCE INDICES.
NRMSE (%)
13.3
2.3
47
NAVE (%)
0.28
0.24
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
CHAPTER III – EMI Filter Design
3.1 Introduction
Starting from the considerations of the previous chapters, it is evident that the use of EMI filters to
reduce the conducted emissions has become necessary in various application fields. The EMI filters
used for the attenuation of conducted emissions are generally low-pass type configuration; there are
different circuit configurations with different characteristics obtained by the combination of passive
components (capacitors and inductors).
Often the filter components are defined by a trial and error approach. This approach leads to
oversizing of the components to obtain the desired performance of the EMI filter and therefore to an
increase of the EMI filter volume and cost of realization. Appropriate considerations should be made
during the design phase to avoid those advantages and they will be treated in this chapter.
In the previous chapter, the CM and DM noise paths have been evaluated and the noise separation
techniques have been described.
The beginning of this chapter presents the criteria for the correct choice of EMI filter topology and
the real HF behavior of filter components that can heavily influence the filter performance. Then the
EMI filter general design steps, the design of CM choke and DM extra inductor with specific attention
to the absence of magnetic core saturation are presented. Finally, the chapter ends with some
considerations on the material of filter components and on their filter performance impact. In
particular, the first part of this subsection deals with the available material technology that could help
in reducing the volume of the filter, such as the nanocrystalline materials. Later on, the key parameters
of the CM choke, such as the permeability, the saturation flux density and the magnetic losses of the
magnetic material core, are analyzed and pushed to their limits. Also a study on capacitors is carried
out in terms of the material performance, the nominal voltage and the application field. Lastly, the
effect on the EMI filter performance due to the filter components tolerances is evaluated.
3.2 Criteria for the choice of EMI filter topology
EMI filters are generally low-pass passive filter realized with inductors and capacitors. The main
adopted topology are the Γ, T and π configurations (Figure III.1). Each filter configuration can result
useful for some applications whereas it could not ensure the required performance for others
applications [68]. It is therefore important to choose the type of the filter depending on the application
in which it will be used. As it can be seen in Figure III.1, each reactive element is arranged so as to
48
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
provide a theoretical attenuation of 20 dB/decade, in this way the π and T filters introduce a 60
dB/decade attenuation, on the other hand the Γ-filter theoretically provides a 40 dB/decade
attenuation.
The term “theoretical attenuation” means the filter attenuation in the absence of parasitic effects.
Really, parasitic phenomena occur at high frequencies resulting in a degradation of the filter
performance.
With regard to the double stage configuration, it is noted that the intermediate capacitor in the π
filter has a double value due to the parallel connection of the first stage output capacitor and the
second stage input capacitor. Analogous consideration applies on the intermediate inductance of a T
double stage filter.
Figure III.1 - EMI filter circuit configurations.
The EMI filter design based only on the theoretical attenuation of the filter topology, without
taking into account the source and the receiver impedances, generally does not allow complying with
the limits imposed by the EMC standards.
The choice of the passive filter topology is closely related to the noise source and to the receiver
impedances module, respectively ZS and ZL. Therefore, in addition to the theoretical attenuation value
of the chosen filter configuration, a proper EMI filter design must suitably takes into account the
criterion of maximum impedance mismatching between the source and the receiver [69] - [71]. In fact,
49
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
the transfer of disturbance from the noise source to the victim (receiver or load) is dependent on the
source/load impedance module ratio. In particular, if the system is schematised according to Figure
III.2 (a), this transfer is described by (3.1), where VL is the load voltage.
VL  VS
ZL
ZS  ZL
(3.1)
It is then evident that the condition
ZS  ZL
(3.2)
is required to have a load voltage attenuation. On the other hand, if this condition is not satisfied or
the attenuation is not enough to comply with the standard limits, a filter should be interposed between
the source and the load. In this case the corresponding scheme is that shown in Figure III.2 (b).
Figure III.2 - Schematic representation of noise source and victim without (a) and with (b) filter.
When the filter is inserted the source will “see” the impedance Zi instead of ZL and the victim will
see the impedance Zo instead of ZS.
Then it is necessary to define the insertion loss of a filter (IL). This parameter is a figure of merit
for a filter, it is defined as a ratio of the signal level in a test configuration without the filter (VL) to the
signal level with the filter (VL'). This ratio is given in dB unit by the following equation:
V 
ILdB  20 log10  L 
 VL ' 
(3.3)
50
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
For most filters VL' will be smaller than VL. Then, the insertion loss is positive and measures how
much smaller the signal is after adding the filter. A high IL value means a good filter attenuation.
In the case shown in Figure III.2 (b), by means of impedance parameters, the IL equation becomes
[72]:
IL 
Z11  Z S   Z L  Z 22   Z12  Z 21
Z S  Z L   Z 21
(3.4)
where Z11, Z12, Z21 e Z22 are the elements of the filter impedance matrix. From (3.4) it is evident that
the IL is dependent on noise source impedance and load impedance, as well as on the intrinsic filter
parameters. Eq. (3.4) can be rearranged as follows:

Z  Z  Z Z
1  S 1  22   12 21
Z11 
Z L  Z11Z L
IL  
 Z S  Z 21
1 

 Z L  Z11
(3.5)
With reference to (3.5) it is possible to observe that the maximization of IL can be obtained if the
two terms in round brackets appearing in the numerator are increased. This is obtained if the following
conditions are imposed in the filter design:
Z S  Z11

Z L  Z 22
(3.6)
Table III.1 gives the filter topology to be used according to the noise source and load impedances.
The following rules can be assessed:

Filter series inductance should be connected to low impedance source or low impedance
load.

Filter parallel capacitors should be connected to high impedance source or high impedance
load.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Table III.1
3.3
Chapter III – EMI Filter Design
FILTER TOPOLOGY SELECTION BASED ON IMPEDANCE MISMATCHING CRITERION.
Real behavior of passive components
Another important consideration on the EMI filter design and implementation regards the parasitic
phenomena due both to non-ideal behavior of filter components and to the circuit layout. These
aspects can heavily influence the filter performance.
Passive EMI filters components can have specific current and/or rating requirements, but they all
must have good HF characteristics that are a purely capacitive behavior and purely inductive behavior,
for capacitors and inductors respectively, up to high frequencies. Due to present parasitic phenomena,
any suppression component resonates at a frequency, namely the self-resonant frequency fr, given by:
fr 
1
(3.7)
2 LC
where L and C are component’s inductance and capacitance. For capacitors and inductors one of
these parameters is the intrinsic characteristic while the other is a parasitic parameter. The fr value of
most suppression capacitors and inductors falls in the frequencies range 150 kHz – 30 MHz so it is
necessary to choose passive components with a fr as high as possible to obtain good HF characteristics.
In the following subsections, an analysis of the real behavior of capacitors and inductors will be
provided.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
3.3.1 Capacitors behavior including parasitic effects
For low-frequency and DC signals, such as the supply currents in AC or DC power lines, a
capacitor provides a high impedance path, but for HF signals, such as the noise currents in the 150
kHz – 30 MHz range, it provides a low impedance path. For this reason capacitors are connected in
parallel with the noise source then they attenuate the conducted EMI by shunting them.
A real capacitor has not a purely capacitive behavior (even at low frequencies), since the leakage
resistance of the isolation (REPR) and the equivalent series resistance (RESR) cannot be neglected in
every case. At higher frequencies, the effect of stray inductance (LESL) is also to be considered. The
characteristics of a real capacitor can be properly discussed in a relatively wide frequency range by
means of the equivalent circuit shown in Figure III.3 [72].
Figure III.3 – Equivalent circuit of capacitors.
The impedance of a capacitor, ZC, is defined in (3.8):
Z C  RESR  jLESL 
REPR

1  jREPRC
RESR
  2 LESLC
REPR
L

 jLESLC  j  ESL  RESRC 
 REPR

1
1
REPR
(3.8)
At DC and low frequency, the impedance of a capacitor is affected only by the resistive effects,
RESR+REPR. With increasing frequency, the capacitive behavior predominates over the resistive one and
Eq. (3.8) is semplified to the form below:
53
Power Density Optimization of EMI Filters for Power Electronic Converters
Z C  RESR  jLESL 
Chapter III – EMI Filter Design
1
jC
(3.9)
Over fr the inductive component predominates on capacitive one; then the real capacitor behave
like an inductor at high frequencies. The impedance of a real capacitor is just RESR at fr.
The Bode plot of the impedance as a function of frequency, defined by Eq. (3.8), with the
following parameters: C=10 µF, RESR=0.3 Ω, REPR=10 kΩ and LESL=96 nH, is shown in Figure III.4 as
an example. As seen from the curve, the capacitor can be regarded as purely capacitance only in the
angular frequency range 10÷106 rad/s.
Figure III.4 - Bode plot of impedance Zc(f).
3.3.2 Inductors behavior including parasitic effects
Unlike capacitors, inductors provide a low impedance path for mains frequency signals, and high
impedance for HF signals. There are two types of suppression inductors: DM inductors and CM
chokes.
Suppression inductor is affected by disadvantages:

it is bulky;

the real inductance value can differ to the nominal value given by the manufacturer, due to
high tolerance value;

its HF characteristic depends on operating temperature or current (in fact the nominal
inductance value decreases as a result of the core saturation).
In EMI filter design the Y-capacitor values are limited for safety reasons and the required CM
attenuation must be achieved mainly by the inductive components of the filter. Despite the
54
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
disadvantages listed above, least one inductor is used in EMI filters implementation. In comparison
with capacitors, inductors have non-linear characteristics and they can present more resonant
frequencies.
To understand the terms that define the performance and limitation of inductors in the frequency
range of conducted noise suppression, it is useful to consider the equivalent circuit in Figure III.5. The
resistance RESR in the equivalent circuit represents the coils losses. Parasitic effects at higher
frequencies, due to the stray capacitances between turns, cannot be neglected. Although the turncapacitance is distributed, a parallel connected concentrated capacitor, CEPC, provides a suitable
approximation. [72]
Figure III.5 – Equivalent circuit of a inductor.
The impedance of the inductor according to the equivalent circuit is:
ZL 
RESR  jL
1   LC EPC  jRESRC EPC
(3.10)
2
The DC impedance is equal to RESR. At low frequencies, impedance ZL is dominated by inductive
component and its value increases proportionally with the frequency. At the resonance frequency, the
inductor, L, resonates with the parallel capacitor, CEPC, and the impedance reaches its maximum value.
Over the resonance frequency the inductor impedance decreases because the capacitive contribution
dominates and the inductor behaves like a capacitor.
The Bode plot of the impedance as a function of frequency, defined by Eq. (3.10), with the
following parameters: L=1 mH, CEPC=10 pF, and RESR=10 Ω, is shown in Figure III.6 as an example.
The inductor can be regarded as purely inductance only in the angular frequency range 104÷107 rad/s.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
Figure III.6 - Bode plot of impedance ZL(f).
Finally, the inductor impedance at low frequencies (more precisely, in the frequency range below
the resonance frequency) can be approximated as follows:
Z L  RESR  jL
(3.11)
At resonance frequency it is:
ZL  fr  
L
RESRC EPC
(3.12)
Indeed, for frequencies higher than the resonant frequency, ZL is approximated as follows:
ZL 
jL
1

 LC EPC jC EPC
(3.13)
2
3.4 EMI Filter General Design Procedure
As well established, the conducted electromagnetic emission can be decomposed in DM and CM
noise. The generation and coupling mechanisms as well as the CM and DM EMI paths are different
and therefore separated filter sections are needed in order to obtain suitable attenuation for EMC
compliance. Both CM and DM filter components are usually embedded into a single filter
configuration, as shown in Figure III.7 (a). In particular, a Γ type filter is used to attenuate CM noise,
56
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
as shown in Figure III.7 (b) (with capacitors placed on source side, assuming a high source impedance
and a low load impedance), and a π type filter is used to DM noise mitigation, as shown in Figure III.7
(c). The filter is composed of passive components acting on CM or DM noise separately, and other
elements simultaneously affecting both types of noise. Therefore, it is necessary to separately evaluate
the two noise modes as a basic step to design an effective EMI filter [73].
Figure III.7 - Generic EMI filter configuration (a), CM equivalent circuit (b) and DM equivalent circuit (c).
The capacitors Cy attenuate both CM and DM noise and they are generally in the order of
magnitude of nF; their value is very small compared to that of Cx1 and Cx2, which are in the µF range,
so their effect on the DM noise is almost negligible. On the other hand, the capacitor Cx between the
electrical lines only attenuates the DM noise. The bulkiest component of the filter is the common
mode choke, LCM, that ideally suppresses only the CM noise; however its leakage inductance (Lleakage)
is usually sufficient to attenuate the DM noise as well [68]. Sometimes an additional inductance in
series with the LCM (LDM_extra) might be useful to increase the total value of the DM inductance.
The procedure followed in designing the CM and DM filter components is shown schematically in
the block diagram in Figure III.8 and it is described below.
The first step is the identification of the crucial point on the experimental curve of the EMI
emission, i.e., the most relevant emission peaks or the emission peaks at the lowest frequency.
Therefore it is necessary to measure the CM and DM components separately and to obtain their
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
frequency spectra. The required attenuation for the CM e DM noise, Attreq_CM e Attreq_DM, are calculated
as follows:
Att req _ CM  Ah _ CM  L  SM
(3.14)
Att req _ DM  Ah _ DM  L  SM
(3.15)
where Ah_CM, Ah_DM are the amplitudes of the harmonic to be attenuated, L is the maximum
amplitude allowed by the reference standard at the frequency of interest and SM denotes an additional
safety margin usually set to 6 dBµV.
Figure III.8 – Steps of EMI filter design.
The cut-off frequency, fo, of the CM or DM filter is given by (3.16):
𝑙𝑜𝑔10 𝑓
𝑓ℎ_𝑎𝑡𝑡
0_𝐶𝑀/𝐷𝑀
𝐴𝑡𝑡𝑟𝑒𝑞
= 𝐴𝑡𝑡
𝑓𝑖𝑙𝑡𝑒𝑟
,
f h _ att
f o _ CM / DM 
Attreq
10
(3.16)
Att filter
where fh_att is the harmonic frequency to be attenuated and Attfilter is the filter theoretical attenuation,
relating to its circuit configuration [68]. In order to improve the filter performance and to avoid any
amplification of the switching frequency harmonics, the designer could choose to fix the fo value lower
than the converter switching frequency value (fPWM).
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
When the both corner frequencies are known, the inductance and capacitance values of the
CM/DM sections of the EMI filter can be determined according to (3.17) and (3.18).
LCM 

1
CCM 2 f o _ CM
CCM  n phase C y

2
(3.17)
where nphase is the number of AC phases/DC lines of the power electronic system.
CDM  C x1  C x 2 
LDM
1
2 f o _ DM


(3.18)
2
From Eq. (3.17) and (3.18), it is necessary to define a value of filter components to obtain all the
others. Usually firstly the CM parameters are defined. The CM capacitors, Cy, are connected from the
line to the ground/chassis and the value of these specific capacitors is therefore regulated by standards
for safety. Some of the possible choices for the designer are cited below:
-
Cy =100 nF (SAE AS 1831 - Society of Automobile Engineers [74]);
-
Cy =100 nF for 60 Hz equipment or Cy =20 nF for 400 Hz equipment (MIL-STD-461F [75]).
With reference to DM parameters some degrees of freedom exist; in particular, by increasing the
value of CDM, the size of LDM will be reduced and vice versa.
The DM inductance is usually obtained by the leakage inductance of the CM choke; in this case its
value is approximately 0.1-2% of the CM inductance value, depending on the core material. Then, the
required DM capacitance is defined on the basis of the cutoff frequency. On the other hand, if a higher
LDM value is preferred to reduce the value or the size of DM capacitors, then dedicated DM inductors
can be considered in the filter design [76].
Once the values of the EMI filter components are defined, the chokes/inductors and capacitors
should be accurately chosen to obtain an effective attenuation. In particular, with regard to magnetic
cores, both geometrical dimensions and magnetic properties of the material must be appropriately
selected to prevent magnetic saturation, as it will be described in subsections 3.4.1 and 3.4.2.
In subsection 3.4.3 some considerations on the impact of magnetic cores’ and capacitors’ material
on filter performance will be presented.
3.4.1 Design of CM choke and DM extra inductor
Once the values of the filter inductive components are defined, it is necessary to perform the
following analysis.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
The first parameter that needs to be set is the windings wire section A. This parameter is defined by
the value of the operating current I as: A  I / J where J is the current density set equal to 4 A/mm².
The value of the wire section defines the wire AWG (American Wire Gauge), as in Table III.2. The
AWG is a standardized wire gauge system used for the diameters of round, solid, nonferrous,
electrically conducting wire.
The AWG of a stranded wire is determined by the cross-sectional area of the equivalent solid
conductor. Because there are also small gaps between the strands, a stranded wire will always have a
slightly larger overall diameter than a solid wire with the same AWG. The cross-sectional area of each
gauge is an important factor to determine its current-carrying capacity.
Table III.2
TABLES OF AWG WIRE SIZES (SOLID WIRE).
AWG gauge Conductor Diameter (mm) Conductor Section (mm2)
14
1.628
2.08
15
1.450
1.65
16
1.291
1.31
17
1.150
1.04
18
1.024
0.823
19
0.912
0.653
20
0.812
0.518
The next step is to choose a core size and material depending on the number of turns needed to
implement LCM.
To avoid saturation phenomena in the core material, due to high LCM value (generally of the order
of millihenries) and the current that runs through it, CM inductance is implemented with a CM choke.
Common mode inductors are wound with two or three windings with equal numbers of turns. The
number of windings is the same as the number of phases. As depicted in Figure III.9, the windings are
placed on the core so that the line currents in each winding create fluxes that are equal in magnitude
but opposite in phase in the case of DM currents, and identical in the case of CM currents. The fluxes
of DM currents are thus ideally cancelling out each other and the related current is not influenced by
the inductors. It will be shown in section 3.4.2 that the cancellation is in practice not complete, and
this is called leakage inductance. The fluxes due to CM currents, on the contrary, are accumulated
inside the core. Therefore, the common mode choke coil works as an inductor against common mode
current.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
Figure III.9 – CM choke [98].
The inner circumference of the toroidal core, I.C., the maximum number of achievable turns on
toroidal core, Nmax, and the required number of turns for LCM/DM implementation, Nrequired, are
determined by the following equations in which σwinding represents the maximum angle that the
winding subtends on half of the core. It is acceptable to assume σwinding equal to 145º, as shown in
Figure III.10.
Figure III.10 - Winding angle example.
I .C.   IDcore  Dwire 
N max 
2 winding I .C.
360 Dwire
 L1  2  LCM

CM 
L1
 N required  2 A
L

 L1  LDM / 2

DM 
L1
 N required  A
L

(3.19)
where IDcore is the inner diameter of the toroidal core, Dwire is the wire diameter, L1 is the
inductance value of a winding and AL is the inductance value obtained with a turn realized on a
specific core. Of course the condition N required  N max must be verified.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
3.4.2 Considerations on magnetic cores saturation
After having chosen a suitable toroidal core in terms of geometric dimensions to implement LCM or
LDM, it’s necessary verify the suitability of the core in terms of absence of magnetic saturation.
The design criterion for the magnetic core selection to implement LCM and LDM is based on the
following analysis.
Let us consider the CM choke in Figure III.11. The maximum flux density (Bmax) in the CM choke
magnetic core depends on the flowing peak current in its inductors. If L1 and L2 are the inductance of
the two windings of the CM choke and M12=M21=M the mutual one, the currents flowing on such
inductors are respectively (IDM+ICM) and (-IDM+ICM).
Figure III.11 - Electrical representation of a CM inductor.
The magnetic flux due to each of the two windings of the CM choke, ϕ1 and ϕ2, is given by:
1  L1  I DM  I CM   M   I DM  I CM  
(3.20)
 L1  M   I DM  L1  M   I CM
2  L2   I DM  I CM   M  I DM  I CM  
  L2  M   I DM  L2  M   I CM
(3.21)
Therefore, as an upper limit, the overall flux ϕtot_CMchoke on the core is given by:
tot _ CMchoke  1  2  L1  L2   I DM  L1  L2  2M   I CM
(3.22)
The term proportional to the DM current is tied to the leakage flux. Moreover, having a coupling
factor k 
M
close to unity, the following approximation can be done: M=L1=L2.
L1L2
Then Eq. (3.22) can be rewritten as follows:
62
Power Density Optimization of EMI Filters for Power Electronic Converters
tot _ CMchoke  Lleakage  I DM  4L1  I CM
Chapter III – EMI Filter Design
(3.23)
In the case of an extra DM inductor the magnetic flux is generated by a single winding and the
peak current flowing on the same is composed of both the DC current that the noise current. So the
magnetic flux due to DM inductor, ϕtot_DMinductor, is given by:
tot _ DMinductor 
LDM
I DC  I DM  I CM 
2
(3.24)
Moreover the maximum magnetic flux can be expressed as:
tot _ max B max S  N
(3.25)
where N is the total number of turns of the windings, S is the cross section of the magnetic core, Bmax is
the maximum value of the magnetic flux density. Therefore,
Bmax 
tot _ max
(3.26)
(h  (rout  rint ))  N
where h, rout and rint are the height, the outer radius and the inner radius of the toroidal core,
respectively.
Once the peak current is known, it is possible to suitably choose the toroidal core for the CM choke
or DM inductor implementation, according to the condition B max < Bsat (Bsat is the saturation value of
the magnetic flux density at the operating temperature).
3.4.3 Considerations on materials of the EMI filter components
For the implementation of the CM choke and the DM inductance, generally toroidal cores of
magnetic materials are used.
The typical requirements of a common mode choke are: high permeability of the magnetic core,
high saturation level and low magnetic losses.
Three basic materials used in the design of the traditional CM choke are ferrite cores, powder
materials (iron) and metal alloys (nanocrystalline and amorphous structure). Other materials with high
performance at high frequencies and high permeability are commercially available. Those materials
allow to achieve high inductance value with a smaller number of turns and thus reducing the number
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
of stages, the size and the weight of the filter. The choice of materials has therefore a significant
impact on the filter performance.
Figure III.12 presents an overview of the permeability as a function of the frequency of
nanocrystalline, ferrites, iron and amorphous materials [77]. The permeability µ, as the ratio of the
magnetic flux density B and the magnetic field H, is the most important parameter of magnetic
materials. The highest permeabilities are found in nanocrystalline materials. Amorphous alloys have
somewhat smaller values whereas iron powder cores have relatively low permeabilities. Ferrite cores
can be used over a wide frequency range but exhibit a significantly lower level of saturation. Higher
flux densities can be found in nanocrystalline and amorphous materials as well as in iron powder cores
but are limited in the used frequency range. Nanocrystalline materials are a good alternative to the
traditional chokes made of ferrite or iron powder. So, the main advantage is the high level of
saturation and the resulting smaller size.
Figure III.12 - Magnetic properties for ferrites, iron powder and metal alloys: permeability vs. frequency.
A comparison of the magnetic characteristics of a nanocrystalline material and a common ferrite is
given in Figure III.13. It shows the magnetic permeability curves versus frequency (a), the
magnetization curves (b) and the saturation flux density versus temperature (c) of the VITROPERM
500F and of a typical Mn-Zn ferrite.
In Figure III.13 (a), it can be observed that the permeability of VITROPERM 500F is significantly
higher than ferrite in the low frequency range. At higher frequencies the μ of the nanocrystalline
material remains above that of ferrite. A high choke impedance is preferred for higher attenuation.
This can be achieved more effectively by using high permeability core materials than by increasing the
64
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
number of turns, as a lower number of turns results in lower winding capacitance and hence improved
HF properties. In Figure III.13 (b), the magnetization curve of VITROPERM 500F in comparison to
typical Mn-Zn ferrite shows noticeable differences in permeability (slope of the curve) and saturation
flux density (maximum value of the curve). In particular, as shown in Figure III.13 (c), the saturation
flux density of the VITROPERM suffers of a small variation in the operating temperature range up to
150 °C, while the ferrite exibits a decrease of about 40% at 100 °C, compared to the environment
temperature value. The high Curie temperature of VITROPERM alloys (above 600 °C), allows short
term maximum operating temperatures as high as 180 – 200 °C; then the VITROPERM is more
suitable for applications with high operating temperatures.
Figure III.13 - Magnetic permeability curves versus frequency (a), Magnetization curves (b) and the saturation
induction versus temperature (c) of the VITROPERM 500F and a typical Mn-Zn ferrite.
Another feature to take into consideration is related to the power losses in the magnetic materials.
These losses occur when the magnetic material is subjected to a time variable magnetic flux density.
There are several specific losses that contribute to the total power loss due to the magnetic hysteresis
phenomenon and to the eddy currents.
The magnetic hysteresis phenomenon occurs when a ferromagnetic material core is subjected to an
alternative cyclic magnetization. This occurs, for example, when a ferromagnetic material core is
subjected to a time variable magnetic field, as produced by a time variable current. The energy
provided to the core during the magnetization phase is not entirely returned during the
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
demagnetization phase; at each cycle, a quantity of energy proportional to the area enclosed by the
hysteresis loop (Figure III.14) is stored in the magnetic core. Therefore a dissipation of heat energy
occurs in these cores; it is equivalent to the hysteresis power loss and it depends on the frequency f of
the magnetizing current. The Steinmetz’s equation allows calculating the specific hysteresis loss pi per
volume [W/m3] or per mass [W/kg], depending on the expression of the constant kist, as follows:
𝛼
𝑝𝑖 = 𝑘𝑖𝑠𝑡 ∙ 𝑓 ∙ 𝐵𝑚𝑎𝑥
(3.27)
where Bmax is the maximum value of the magnetic flux density, α can range from 1.6 to 2, and kist is a
coefficient depending on the magnetic material.
Figure III.14 – Hysteresis loop of VITROPERM 500F and a typical ferrite [78].
In addition to the hysteresis loss, the eddy currents in the cores mass induced by the variations of
the magnetic flux cause a further dissipation of energy. This loss is determined by the electrical
resistance ρ and the thickness δ of the core. The eddy currents loss pc is calculated using the equation:
𝑝𝑐 =
2
𝑘𝑐′ ∙𝛿 2 ∙𝑓2 ∙𝐵𝑚𝑎𝑥
𝜌
2
= 𝑘𝑐 ∙ 𝑓 2 ∙ 𝐵𝑚𝑎𝑥
(3.28)
where kc’ is a coefficient depending on the magnetic material.
For the nanocrystalline material VITROPERM 500F the coercitivity is very small and therefore the
hysteresis loss is very low. The loss is much lower than in ferrite materials. VITROPERM 500F is a
metallic material and it shows therefore eddy current losses. Because this loss depends not only on the
conductivity but also on the ribbons thickness (Eq. 3.28), that is only 20 μm [80], minimal eddy
currents loss is ensured.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
The sum of hysteresis loss and eddy currents loss constitutes the total loss in the material.
Manufacturers of ferromagnetic materials provide a very important technical data, known as specific
loss figure Cp. Known the value of the specific loss figure, the evaluation of the total loss pt in the
ferromagnetic material, absorbed by a core of mass M subjected to a variable magnetic flux with a
maximum flux density Bmax and frequency f, is performed by means of the following semi-empirical
relation:
𝑓 1.2
50
2
𝑝𝑡 = 𝐶𝑝 ∙ 𝐵𝑚𝑎𝑥
∙( )
∙𝑀
(3.29)
Figure III.15 shows the comparison of magnetization losses in different materials. Just in the range
of the operating frequencies of the switching power devices of 10 – 100 kHz, the nanocrystalline
VITROPERM-alloys are superior to other magnetic materials by exhibiting lower losses. For this
reason the VITROPERM is suitable to power applications from a few kW, e.g. medical or industrial
applications, up to powers of MW used in the field of modern railway traction techniques [78].
Figure III.15 – Comparison of magnetization losses of typical materials for CM choke and DM inductance [79].
On the basis of the considerations aforesaid, VITROPERM nanocrystalline alloys are optimized to
combine highest permeability and lowest coercive field strength. The combination of very thin tapes
and the relatively high electrical resistance (1.1 – 1.2 µΩm) ensures minimal eddy current losses and
an outstanding frequency vs. permeability behavior. Along with saturation flux density of 1.2 T and
wide operational temperature range, these features combine to make VITROPERM a universal
solution for most EMC problems and it presents better performance in many aspects to commonly
used ferrite and amorphous materials [80].
The choice of magnetic materials with high permeability allows achieving high inductance value
with a smaller number of turns: a reduction of the number of stages, the size and the weight of the
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Power Density Optimization of EMI Filters for Power Electronic Converters
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filter is obtained. Two CM chokes with a CM inductance equal to 0.8 mH have been realized by using
a N30 ferrite core and a VITROPERM 500F core. A comparison between the main geometrical and
magnetic characteristics of the two cores is reported in Table III.3. The nanocrystalline core allows
reducing the CM choke size of about 87.5% in volume respect to the ferrite core. Similarly, a
reduction of the CM choke weight of 82.5% is obtained. The CM choke set up with ferrite and
VITROPERM cores is shown in Figure III.16. The size reduction with the use of nanocrystalline
material is evident.
Table III.3
COMPARISON OF DIFFERENT MAGNETIC CORES CHARACTERISTICS TO SET UP A LCM=0.8 mH.
Parameter
Symbol
Toroidal core size
Ferrite N30
50x30x20 mm
VITROPERM
500F
25x16x10 mm
Volume
V
50 cm3
AL value
AL
8700 nkz
Saturation flux density
Bs
0.45 T
1.2 T
114 g
20 g
12
5
Weight
N° of turns per winding
N
6.25 cm3
65.5kz
Figure III.16 - CM choke set up by using an N30 ferrite core (left) and a VITROPERM core (right).
[68] Suitable considerations must be complied about the capacitors. They are very critical
components; an uncorrect choice of the type to use can cause damage and malfunctions of the system.
In particular, they are sensitive to overvoltages that may cause permanent damage to the insulation.
The main specification for the capacitors is MIL-STD-15573. The capacitors must meet various
voltage ratings and for AC capacitors, the level must be 4.2 times the RMS voltage of the system. For
example, in a 220V RMS system, the capacitor must be designed to handle 924V, usually rounded up
to 1 kV. For the DC capacitor, the multiplier is 2.5 times the system voltage. For example, in a 50V
DC system, the DC capacitor must be designed to handle 125V DC. Moreover, the RMS peak voltage
and the maximum applied DC voltage are used to determine this, not the nominal or average voltage.
If this is a 120V AC system, then we can safely assume ±10%; the peak value of 132V is multiplied
by 4.2=554V, which is the final test voltage for the capacitor [68].
Noise suppression capacitors are typically made of metalized film. This choice has the advantage to
increase the stability of the component versus time and temperature. They are also self healing: the
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
components repairs itself after a voltage spike: each layer acts as a single capacitor, if any of them is
damaged the total capacitance can decrease slightly without affecting the performance of the part. In a
worst case scenario the failure mode is an open circuit. This specific structure also allows high values
of capacitance (several μF). These capacitors are more expensive than general purpose capacitors.
In most filters, if the capacitor value is reasonably higher, the filter will work much better and give
more insertion loss (but that will not suffice if the value is limited). From that point of view, ceramic
or polypropylene capacitors are generally preferred. In the following, the impedances module and
phase of different capacitors measured by a precision RLC meter Agilent 4285ALCR are shown to
perform a comparison on their high frequencies behavior. In Figure III.17 the impedances module of
ceramic and polypropylene capacitors are shown: a capacitive behavior can be noted up to a few
megahertz. The ceramic capacitors are characterized by a low parasitic inductance due to their small
size. The polypropylene capacitors have low losses and can support overvoltages.
Figure III.17 – 100 nF capacitor impedance (a) 250Vdc ceramic capacitor (measured data) and (b) 300Vac,1000Vdc
polypropylene capacitor (datasheet).
The DM capacitance is generally of the order of tens of microfarads. In these cases electrolytic
capacitors are used even though their high frequency performance are not good compared to
ceramic/polypropylene capacitors. As illustrated in Figure III.18, the capacitor behavior is generally
capacitive until the tens of kilohertz range, and then become resistive and finally inductive after 1
MHz. In particular, in Figure III.18 and Figure III.19 it can be observed that, for a given capacitance
value, the performance of an electrolytic capacitor varies both as a function of the nominal voltage and
the application field. This confirms the importance of the choice of suitable capacitors for EMI
mitigation.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter III – EMI Filter Design
Figure III.18 – Measured impedance module (upper) and phase (lower) of a 47µF electrolytic capacitor with
nominal voltage equal to 160V and 400V.
Figure III.19 – Measured impedance module (upper) and phase (lower) of a 47µF electrolytic capacitor with
nominal voltage equal to 160V of different manufacturers and for different application fields.
Another feature that can influence the EMI filter performance is the tolerance of its components.
The tolerance value is the extent to which the actual component is allowed to vary from its nominal
value listed in the datasheet.
The cut-off frequency fo of the EMI filter is described by the Eq. (3.27):
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fo 
Chapter III – EMI Filter Design
1
(3.27)
2 LC
The deviation on the required fo due to the inductor and capacitor tolerance is obtained as follows:
𝜕𝑓
𝜕𝑓
∆𝑓𝑜 = | 𝜕𝐶𝑜 | ∙ ∆𝐶 + | 𝜕𝐿𝑜 | ∙ ∆𝐿
(3.28)
According to Eq. (4.27), the Eq. (4.28) can be rewritten as follows:
∆𝑓𝑜 = 4𝜋
1
∙
√𝐿𝐶
∆𝐶
𝐶
+ 4𝜋
1
√𝐿𝐶
∙
∆𝐿
𝐿
(3.29)
where ΔL/L and ΔC/C are the tolerance values of the inductor and capacitor respectively.
From the Eq. (3.30) that defines the deviation on fo value in percent, it is evident that the inductor
and capacitor tolerances determine a cut-off frequency value unlike the required one.
∆𝑓𝑜
%
𝑓𝑜
1
2
∆𝐶
%
𝐶
= ∙(
+
∆𝐿
%)
𝐿
(3.30)
Usually the tolerance rating is expressed as a percentage (±%). The tolerance value of the
capacitors can range anywhere from -20% up to +80% in some cases. Thus a 100µF capacitor with a
±20% tolerance could legitimately vary from 80µF to 120µF and still remain within tolerance. The
tolerance value of the inductors depends to the AL tolarance of the core magnetic material used to
realize the desidered inductance. For example, the AL tolerance of a VITROPERM core can range
from -25% to 45% while for a typical ferrite core it can range from -30% to 30%.
The negative tolerance percentage determines a negative impact on filter performance because it
involves a real value of the component lower than the nominal value and consequently a cut-off
frequency higher than the required one; hence the filter will begin to attenuate at different frequency
than the desired one.
Depending the filter topology, the tolerances effect can not be neglected and it can be analysed by a
Monte Carlo analysis or worst case method.
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Chapter IV – Optimized Design of High Power
Density EMI Filter
CHAPTER IV – Optimized Design of High Power Density EMI Filter
4.1 Introduction
Now that the baseline design, topology and guidelines for an accurate choice the EMI filter
components have been revised, it is essential to look at the optimization of the EMI filter and
especially at its size reduction to achieve a high power density.
Recently power electronics market has been boosted by new high-speed devices allowing faster
switching operation as the wide-band gap devices based on Silicon Carbide (SiC) or Gallium Nitride
(GaN) [8], [9]. On the other hand, their operation in power electronic converters leads to an increase of
electromagnetic interference. For this reason noise filtering is, more than ever, one of the major issues
in power electronic system design, particularly when dealing with stringent standard limits [81]-[83].
Besides satisfying EMI limits, a further optimization in terms of filter size and weight can be
performed; in fact, the EMI filter can contribute up to 30% of the total size and weight of power
electronic converters. Therefore, a filter design matching the maximum power density is strongly
desired, especially for the applications (e.g. airplanes, electric vehicles, etc.) in which compactness
and low weight are the primary constraints [84].
Scientific literature proposes several techniques dealing with high-power-density design of discrete
EMI filters for power electronic converters. Some techniques are based on setting up a compact layout
by using suitable winding structures and/or high performance magnetic materials for the inductor
cores [85], [86]. Other approaches, starting from an accurate high frequency model of the system
under investigation, propose the use of optimization algorithms to minimize either the volume of the
whole EMI filter, i.e. related to common mode (CM) and differential mode (DM) sections, or the
volume of some parts of it. It should be noted that a relevant computational effort is anyhow needed
[87]. The use of heuristic procedures, mostly genetic algorithms (GA), to perform an EMI filter design
oriented to power density maximization, is proposed in [88]. In those cases the high number of
iterations, usually needed to obtain optimal or sub-optimal solutions, results in a time consuming
procedure. A PC-based automatic EMI filter design method without any volume minimization
implications is presented in [89]. Finally, a minimization of the DM EMI filter volume, utilizing some
interpolated volumetric parameters, has been done in [90], where it is demonstrated that the selection
of an optimal number of filter stages leads to the minimum occupied volume. However, this approach
cannot be applied to minimize CM EMI filter volume.
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It is worth noting that, once the filter topology has been chosen and the values of its components
(Common Mode/Differential Mode inductors/capacitors) have been defined, there is a huge amount of
possibilities for practical configurations. Moreover, the identification of the configuration leading to
the best power density in terms of minimum volume/weight is a nontrivial task. According to a trial
and error approach, the conventional design of EMI filters requires a significant effort in terms of time
spent and it does not guarantee the optimal choice of filter components in order to obtain the
maximum power density.
For this reason, an optimized design procedure of discrete EMI filters oriented to obtain high
performing filters and high power density is presented in this chapter. This is ultimately the main
focus of this PhD thesis.
The optimized design of EMI filters is based on an automatic rule-based computer aided procedure
and presents easy implementation features and low computational demand. Both CM and DM sections
of the EMI filter are considered within the procedure.
Moreover, to make the new design procedure more accessible to EMI engineers or scientists
involved in investigation of filter performance/configurations/power density, a software tool based on
the optimized design procedure has been developed, namely ODEF (Optimized Design of EMI
Filters).
The optimized design procedure and the developed tool are described in the following sections.
4.2 Optimized Design Procedure
The optimized design procedure starts from the basic principle of the conventional EMI filter
design illustrated in Section 3.4, introducing the additional objective of pursuing the best power
density for the EMI filter. It is a rule-based algorithm that takes into account the main characteristics
of the filter application: the power electronic circuits under study, the filter design constraints and
databases with parameters extracted by datasheets of commercial components for the realization of
EMI filters.
The overall concept of the optimized technique is summarized in Figure IV.1.
The following Input Data are needed to run the rule-based algorithm:

EMI filter topology;

nphase: number of AC phases/DC lines of the power electronic system;

UN: nominal voltage of the power converter;

Imax_phase: maximum operating current;

ICM_max, IDM_max: maximum CM and DM currents;

the filter design can be performed either on the basis of the measured CM/DM spectra (EMICM,
EMIDM), given as input and compared with the limits of a chosen standard, or explicitly giving
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the required CM/DM attenuations (Attreq_CM, Attreq_DM) and the CM/DM harmonic frequencies
to be attenuated (fCM_att_h , fDM_att_h);

Standard for Cy_value selection: the CM capacitance can be chosen either on the basis of SAE
AES 1831 standard requirements or by explicitly setting the maximum ground leakage current;

kvol, kweight: two coefficients provided by the designer allow to optimize the design assuming any
linear combination of volume and weight as the objective function.
In order to properly select the most suitable components for the EMI filter, two databases of
commercially available devices have been set up.
Figure IV.1 – Concept of the optimized EMI filter design procedure.
The first one is a database of magnetic cores, including 110 toroidal cores, with both nanocrystalline (Vitroperm 500F) and ferrite (N30) materials.
The database of commercial cores contains the following information:

core material and model;

geometric dimensions and weight;

inductance factor AL (µH/1 turn) at 10 kHz and saturation flux density value.
The cores’ dimensions have been chosen so as to design EMI filters for applications able to manage
both low powers and powers up to some kWs.
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In addition, a suitable database of capacitors, including Y-type (for the CM noise mitigation) and
X-type (for the DM noise mitigation), has been built as well. As for Y-type capacitors, ceramic
devices with different rated voltage levels have been included in the database. As far as the X-type
capacitors are concerned, 160V, 250V and 400V aluminum electrolytic capacitors for applications
with high ripple currents at high frequencies and polypropylene capacitors for EMI suppression, have
been included.
The database of commercial capacitors contains the following information:

brand, material, series, model and package;

rated capacitance and voltage;

geometric dimensions and weight.
In addition, a third database, including conducting wires, is provided. So, the volume/weight
contribution given by the inductor wires (non-negligible when dealing with rated power of hundreds of
watt and beyond) is included in the EMI filter calculations.
Once all the input data have been entered, the rule-based algorithm repeats the steps of the
conventional design procedure for different configurations (e.g., varying core material and model,
number of stages, etc.) and chooses the configuration exhibiting the best power density. Since multistage filter can occupy a smaller volume than single stage one, depending on the used components, the
optimized design procedure considers the possibility to span a number of filter stages (n) ranging from
1 to 5. The evaluation of a maximum number of five stages is a reasonable choice, since it is very
unlikely that a greater number of filter stages can allow to obtain a more compact filter than a single
stage one.
In particular, the algorithm performs the filter design according firstly to CM requirements; then,
DM requirements are fulfilled according to the steps described hereinafter. The different steps of the
optimized procedure are summarized in Figure IV.2, whose symbols are defined as follows. AWG:
conductor diameter expressed in American Wire Gauge unit; n: number of filter stages in range 1÷5;
Nmax: maximum number of turns for each core; Cy, Cx: capacitance of phase-to-ground/phase-to-phase
capacitors; Vrated, Crated: capacitors’ rated voltage/capacitance; UN: nominal voltage of the power
converter; Bmax, Bsat: maximum/saturation magnetic induction.
The first step is the selection of wire AWG on the basis of the maximum operating current value
given by the designer as input data. Follows a computation of the maximum number of turns (Nmax) for
each core of the database.
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A. CM section design
As for the CM section design, the procedure computes, for n=1,…,5, the following quantities: the
CM capacitance (Cy), the cutoff frequency, the CM inductance and the number of turns (NCM) needed
to set up the required inductance.Then, the Cy capacitor with the minimum volume is selected from the
database according to the design constraint 𝑉𝐶𝑦 = 𝑘 ∙ 𝑈𝑁 where k is a multiplier factor (equal to 2.5 for
DC systems and 4.2 for AC systems). Also the cores allowing the practical realization of the CM
choke according to the required value of CM inductance (i.e., NCM <Nmax) are selected from the
database, taking into account the further constraint related to the absence of saturation.
B. DM section design
Two different procedures allow the designer to compare the results obtained considering either the
leakage inductance of the CM choke (No extra LDM) or the use of separate DM inductors (Extra LDM).
The first step, i.e., the evaluation of the cutoff frequency versus the number of stages, is common to
both procedures. Then:

The “No extra LDM” procedure computes, for n=1,…,5, the leakage inductance of the feasible
CM chokes and the corresponding required value of DM capacitance. After the computation of
the CDM capacitance, the corresponding capacitor with the minimum volume is selected from the
database according to the design constraint 𝑉𝐶𝑥 = 𝑘 ∙ 𝑈𝑁 .

With the “Extra LDM” procedure, for n=1,…,5, the DM inductance candidate values are obtained
on the basis of the X-capacitors values in the database (ranging between 10nF and 330µF).
Then, the number of turns for each DM core is calculated and the cores allowing a practical
realization are selected from the database, according to the condition NDM <Nmax and to the
absence of saturation. Finally, the best pairs LDM-Cx that allow to set up the DM section
according to the design constraints 𝑉𝐶𝑥 = 𝑘 ∙ 𝑈𝑁 , are selected.
As already underlined, the constraint on the absence of saturation of the magnetic core has been
imposed in both the CM and DM section design procedures. In particular, the fulfillment of the
condition Bmax <Bsat has been verified for the magnetic materials (nano-crystalline or ferrite) taken
into account in the procedure. Thus the designed EMI filters are free from saturation issues with no
degradation of the desired performance. Finally, the algorithm calculates the EMI filter volume and
weight of all possible configurations and select the one with the best power density.
The algorithm allows a more extensive evaluation of EMI filter components and configuration
impact on power density in terms of both volume and weight and, therefore, more effective results.
Furthermore, it does not discard suboptimal designs, allowing to compare them with the best solution.
The automatic rule-based procedure can be easily implemented by using a common programming
language, within either an open source or a commercial environment, and it does not require a long
lasting execution time but it provides the real-time output data. Therefore, it can be advantageously
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used both by EMI engineers for obtaining an optimized filter design and by scientists/experts for the
evaluation of filter performance versus configuration and power density features. Moreover, to make
the new design procedure more accessible to EMI designer, a software tool based on the optimized
design procedure has been developed, namely ODEF (Optimized Design of EMI Filters) and described
in the next section. All the design options, steps, outputs and design-related supplementary analyses
are managed by ODEF tool in a user-friendly mode.
Figure IV.2 - Flowchart of the optimized EMI filter design procedure.
4.3 ODEF Application
ODEF is an interactive application running in Matlab® environment that enables a simple and fast
selection of EMI filter components, circuit configuration and number of stages for achieving optimal
power density. Furthermore, ODEF allows to compare the optimal EMI filter design to the suboptimal
results, so as to leave the final choice to the designer.
ODEF application is distributed as freeware for noncommercial use. A first version of ODEF can
be downloaded from www.issia.cnr.it/wp/?page_id=8070 (Figure IV.3) and an updated version v.2.0.
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will be provided. The application has been designed using Matlab GUIDE to layout the user interface
and by manually writing the code for the callback functions associated with each graphical object. It
has a simple and intuitive user interface, which is organized in three tabs for increased usability,
namely Noise Profile, Computation, Extra (Figure IV.4 - Figure IV.6). In particular, the main
algorithm is executed interacting with the graphical objects of Computation tab, whereas the other tabs
present additional features that complement the main algorithm.
Figure IV.3 – Screenshot of the web page for ODEF application download.
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A general description of the implementation of ODEF and the main functionality of the application
will be given hereinafter.
A. Starting the application and defining inputs
After installation, the application is simply started typing ODEF at Matlab® prompt. A window
will open, as shown in Figure IV.4. If the user already knows the required CM/DM attenuations and
cutoff frequencies, he can directly switch to Computation tab and enter these values in the related
fields of the third panel, together with filter topology (e.g., Γ, Π, T). The circuit schematic of the
chosen filter topology will be shown in the upper right figure of the tab.
Then, the user can provide the other input quantities, filling the fields of the other panels. In
particular, the system parameters to be entered in the first panel are the following: system type
(frequency and number of phases), nominal voltage, the maximum load current and maximum values
of the CM/DM noise currents. In the second panel, the user can choose how to determine the value of
the CM capacitors: either according to SAE AS 1831 standard (20 nF for 400 Hz systems; 100 nF in
the other cases) or imposing a maximum leakage current. Furthermore, in the Extra_Ldm panel the
user can express his preference about the realization of the DM filter. Choosing Always or Never the
algorithm will include in the search space only the DM filters realized using an extra DM inductor or
exploiting the leakage inductance of the CM choke, respectively. For example, aiming to achieve an
increased reliability, the user might want to force the use of extra DM inductors to be sure that
electrolytic capacitors are not used for the realization of the DM filter. On the other hand, if Auto is
selected, the search space will include DM filters realized according to both techniques. Finally, the
volume and weight coefficients of the objective function, expressed in percentage, can be entered in
the last panel. If the user does not know the required CM/DM attenuations and frequencies, he can
open Noise Profile tab and load previously acquired CM/DM noise spectra in Excel ® file format or
import them from Matlab® workspace. The data will be plotted in the two graphs of the tab. Then,
selecting an item from the related drop-down menu, the user can choose a reference standard, whose
limit curve will be superimposed on the data as a red line. The following reference standards are
available: MIL-STD-461F, EN55011 class A, EN55011 class B, DO160F cat. B, DO160F cat. L.
Finally, as soon as the user selects filter topology and safety margin, the application automatically
computes the CM/DM harmonic frequencies to be attenuated and the corresponding required
attenuations, fills the related text boxes and highlights the related points on the graphs with red circles.
At this point, the user can confirm the data and switch to the Computation tab to provide the other
input data, as previously described.
B. Executing the algorithm
When the user clicks on the Compute button of the Computation tab the algorithm described in
Section 4.2 is executed, processing the input values and the database content. Then, the results are
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shown in the Results panel of the same tab. In particular, the feasible filter configurations are arranged
according to increasing values of the objective function and the Configuration drop-down menu is
populated. When the user selects an item from this menu, the details about the chosen configuration,
including the objective function’s value, are shown in the Results panel. Besides the numerical values
of the physical quantities (e.g., LCM, Cx, Cy, etc.), other data are extracted from the databases, e.g., wire
type and code, core material and type, capacitor brand and model, etc. The obtained filter
configurations depend on the specific components of the chosen database, which can easily be
modified or expanded. In the latter case the execution time of the algorithm will increase, as expected.
Finally, a series of buttons allow to load/save either the complete input dataset, including the CM/DM
spectra, or the sorted set of feasible configurations. In this way, saved information can be recalled at
later time.
C. Plotting data
Sometimes, it is useful to compare suboptimal designs to the best solution returned by the
algorithm. For example, it could happen that the best design is a two-stage filter, but the second best
design is a one-stage filter, whose objective function’s value is slightly higher than the global
minimum. In such cases, the designer could choose the second best design. For this purpose, besides
exploring the Results panel of the Computation tab, it is possible to exploit the features of the Extra
tab. In particular, after selecting the number of configurations to consider among the entire feasible
set, it is possible to generate a series of comparative plots, as those shown in Chapter V for the chosen
case study.
Furthermore, it could be interesting to check whether the best design varies or not when the CM
attenuation, imposed by the user, is higher than the minimum required value. Sometimes the best
design remains the same, due to the discrete nature of the problem (discrete set of values for L and C,
integer number of turns for the inductor windings, etc.). To this aim, a series of buttons of the Extra
tab allow to load feasible configuration sets, previously saved after different runs of the algorithm, and
to generate some plots that compare the best designs, as those shown in Chapter V to discuss the
results for the chosen case study.
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Figure IV.4 - Screenshot of ODEF application: Noise Profile tab.
Figure IV.5 - Screenshot of ODEF application: Computation tab.
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Figure IV.6 - Screenshot of ODEF application: Extra tab.
4.4 Summary
In this chapter a new optimized EMI filter design technique for the optimal and fast selection of
discrete EMI filter components and configuration, aimed at obtaining the minimum volume/weigth,
has been presented. This tecnnique has been implemented as a feedback to the demand of a wide range
of applications in which the power density of power converter systems is a stringent design constraint.
Therefore a filter design that implements the maximum power density is strongly desired.
The optimized procedure relies on a suitably devised rule-based algorithm and on databases of
commercially available magnetic cores, capacitors and conducting wires. It takes as inputs some
parameters that are computed from noise measurements and others that define the power electronic
circuits under study system. Once all the input data have been entered, the rule-based algorithm
repeats the steps of the conventional design procedure for different configurations (e.g., varying core
material and model, number of stages, etc.) and chooses the configuration exhibiting the best power
density. Easy implementation features and low computational demand characterize the rule-based
algorithm.
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On the basis of this procedure, an interactive software tool for the optimized design of discrete EMI
filters in terms of power density, namely ODEF (Optimized Design of EMI Filters), has been
developed . ODEF is an application running in Matlab® that also allows to compare the optimal EMI
filter design to the suboptimal results, so as to leave the final choice to the designer.
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Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
CHAPTER V – Experimental Validation of the Optimized EMI filter
Design Procedure
5.1 Introduction
In chapter 3, the basic steps of the general EMI filter design procedure and some considerations on
the EMI filter performance and size due to the topology and to the components type have been
presented. In the chapter 4 an optimized EMI filter design procedure which allows an optimal and fast
selection of discrete EMI filter components and configuration, aiming to obtain the miminum volume
or weigth and high performance, has been presented.
In this chapter, an experimental assessment of the optimized technique is performed by using
different suitably devised experimental setup. A comparison of the optimized filter obtained with the
conventionally designed one, is carried out in terms of volume, weight and performance: the optimized
design procedure allows to obtain the compliance of the power electronic system under study with the
standard, using EMI filters with higher compactness and power density, with a low computational
effort.
Futhermore, an analysis of the fleasible configurations returned by the algorithm is performed, for
some of the case studies, by a series of comparative plots generated by ODEF application; interesting
results and a very considerable number of configurations are evaluated. This evaluation is practically
cumbersome without the developed software tool.
5.2 Experimental setups
In order to validate the optimized EMI filter design procedure, an experimental investigation has
been carried out on four suitable experimental setups:
 case study #1: inverter-fed induction motor drive (240W);
 case study #2: inverter-fed symmetric low power (7.2W) resistive load;
 case study #3: DC motor drive (30W) supplied by a DC/DC boost converter;
 case study #4: DC motor drive (190W) supplied by a DC/DC buck converter.
Figure V.1 shows a scheme of the experimental arrangement for the case studies.
It should be observed that PWM inverter-fed loads/induction motor drives supplied by a DC power
grid, such as those considered in the first and second case study, are very common, for example either
in vehicle applications (road vehicles, marine vehicles, aircrafts) either in DC distribution systems,
such as those used in some residential/commercial smart buildings for energy saving [91], [92].
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Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
The third and fourth case study are typical applications for automotive environment in which the
presence of low-power loads supplied with different voltage levels requires the use of DC/DC
converters [93].
Figure V.1- Scheme of the experimental rigs: (a) case study #1; (b) case study #2; (c) case study #3; (c) case study #4.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
In all case studies, the PWM modulation has been implemented on an Altera Cyclone III FPGA
board [94], shown in Figure V.2 on the left. It includes an Altera FPGA EP3C25F324 controlled by a
50 MHz oscillator and it presents the following main features:

25K logic elements;

66 M9K memory blocks (0.6 Mbits);

four PLLs (Phase-Locked Loop);

214 I/Os.
The board can be easily programmed in VHDL language and it represents a low-cost and highperformance platform that allows to implement a broad range of designs of different complexity.
The main features of the Cyclone III starter board are the low-power consumption, the availability
of SSRAM and EEPROM memories and and the expandability via the HSMC connector (High Speed
Mezzanine Card), which allows the connection of expansion boards with different capabilities.
In the concerned cases, it was necessary to connect the Nial Stewart GPIB expansion board [95],
equipped with different I/O connectors with its level-shifter to interconnect the 2.5V CMOS logic to
3.3V CMOS logic and TTL logic, and with 10-bit A/D and D/A converters with 8 channels each. This
expansion board is shown on the right of the Figure V.2.
A further advantage of the Altera board consists of the the Cyclone III device can be configured via
the on-board USB-Blaster™ or through the JTAG interface using an external programming cable (sold
separately).
For the writing and the compiling code, the simulation, the debugging and the programming phases
of the device, the Quartus II Web Edition software, which is a free valuable graphical development
environment provided by Altera [96], has been used.
Figure V.2 - Cyclone III FPGA Starter Board equipped of the Nial Stewart GPIB expansion board used in the
experimental setups.
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Optimized EMI filter Design Procedure
The programming of the Altera Cyclone III FPGA board has allowed to obtain a system with the
following characteristics:
• the PWM carrier frequency can be modified according to the output frequency (synchronous or
asynchronous modulation);
• the startup and shutdown of the system is settled by means of acceleration and deceleration ramps
with configurable time, allowing to gradually change the frequency even after unexpected changes
of reference;
• the presence of START and STOP buttons with LED display;
• the possibility to use the RESET button to quickly disconnect the connected motor;
• the possibility to view on a 7-segment display (SSD) the following information: the frequency, the
actual output frequency set point, the output voltage, the modulation index, the carrier frequency,
the indication of a block due to the overmodulation, the overflow indication.
In the SSD board there are a rotary switch with four possible positions, a 7-segment display with
four digits (SSD display) and all the circuitry necessary for the multiplexed driving, as shown in
Figure V.3. The SSD board allows to display information given by the Altera board, selecting them by
the rotary switch, and it connects to the DIP24 socket of the Nial Stewart board using a specific 26-pin
flat cable. When an overflow condition occurs, the display shows "0.0.0.0.". The LED is used as the
overmodulation block indicator.On overflow, the display shows "0.0.0.0.". The sign LED is used as
the indicator of block condition due to overmodulation.
The system also includes the following components:
• a DB15-RCA cable to connect the Nial Stewart board to the driving inputs of the switching devices;
• a 1 kΩ linear multi-turn potentiometer to set the frequency reference; it needs to be connected to
ADC0 input of the A/D converter in the Nial Stewart board.
Figure V.3 - Board with the display SSD used in the experimental setups.
A dual LISN with a voltage capability up to 600V, a RF current probe R&S EZ-17 that allows
measurements in the frequency range 20 Hz – 100 MHz with a maximum DC current of 300 A, and an
Agilent E4402 (9 kHz – 3 GHz) spectrum analyzer have been employed to measure the conducted
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Optimized EMI filter Design Procedure
EMI. A Tektronix TDS7254B 2.5GHz - 20GS/s - 4 channels has been used for the time domain
measurements.
In all case studies has been verified that the systems are characterized by high CM noise source
impedance and low CM noise receiver impedance. Then according to the criterion of maximum
impedance mismatching between the source and the receiver, a Γ network topology has been chosen
for the CM filter. The noise source characterization for DM noise is more complex to define due to the
DM input impedance of the motor drive. For this reason, a Π network topology has been chosen for
the DM filter: if the real DM noise source impedance is high, a theoretical attenuation of 60 dB/dec is
expected. Otherwise the theoretical attenuation from one of the capacitors Cx is insignificant and a
lower attenuation could be expected (40 dB/dec) and consequently the value of the DM capacitors will
be adjusted.
Moreover, the EMI filters performance has been verified against both military and civilian
technical standards. Many standards exist to accommodate the wide variety of applications where EMI
is an issue. Most of the standards differ either by their frequency range of application or the amplitude
of the noise limits and whether the type of measured noise is voltage or current. They also have their
own experimental and noise measurement setup as well as their own LISN circuit. However this Ph.D.
thesis is based on the military standard 461F described in [75] and on the civilian standard CISPR 25
described in [97].
The MIL-STD-461F, entitled “Requirements for the control of electromagnetic interference
characteristics of subsystems and equipment”, establishes interface and associated verification
requirements for the control of the electromagnetic interference emission and susceptibility
characteristics of electronic, electrical, and electromechanical equipment and subsystems designed or
procured for use by activities and agencies of the Department of Defense (DoD) USA. Such
equipment and subsystems may be used independently or as an integral part of other subsystems or
systems. In particular the CE102 applicability of this standard has been taken into account; this
requirement is applicable from 10 kHz to 10 MHz for all power leads, including returns, which obtain
power from other sources not part of the EUT. Figure V.4 defines the maximum noise limit for the
conducted EMI noise. It is important to mention that the basic curve is given for a voltage of 28 V, and
as the voltage increases some relaxation of this limit is permitted. For this Ph.D. thesis, the basic curve
is anyway considered in the comparison with the measured EMI in order to consider the most
restrictive limits.
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Optimized EMI filter Design Procedure
Figure V.4 - MIL-STD-461F: CE102 limit (EUT power leads, AC and DC) for all applications.
The CISPR 25 [97], entitled “Radio disturbance characteristics for the protection of receivers used
on board vehicles, boats, and on devices – Limits and methods of measurement”, describes the limits
and the procedures for the measurement of radio disturbances in the frequency range of 150 kHz to
1000 MHz. The standard applies to any electronic/electrical component intended for use in vehicles
and devices. The limits are intended to provide protection for receivers installed in a vehicle from
disturbances produced by components/modules in the same vehicle. The limits in this standard are
recommended and subject to modification as agreed between the vehicle manufacturer and the
component supplier. This standard is also intended to be applied by manufacturers and suppliers of
components and equipment, to devices under test, which are to be added and connected to the vehicle
harness or to an on-board power connector after delivery of the vehicle. Moreover the electromagnetic
disturbance sources are divided into two main types:
• narrowband source whose emission has a bandwidth less than that of a particular measuring
apparatus or receiver (e.g. vehicle electronic components which include clocks, oscillators, digital
logic from microprocessors and displays);
• broadband source whose emission has a bandwidth greater than that of a particular measuring
apparatus or receiver (e.g. electrical motors and ignition system).
The noise emission limits are referred to five different classes, in rising order of required reduction
of the maximum electromagnetic disturbance level that the devices can produce on board. The class
refers to a performance level agreed upon by the purchaser and the supplier and documented in the test
plan. In the case studies the EMI source is a broadband source, therefore the noise limits (peak
detector) for broadband conducted disturbances, shown in Figure V.5, have been considered.
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Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
The possible variation of the emission limits established by the standard, in later editions, does not
affect in any case the optimized design procedure itself.
140
Class 1 limit
Class 2 limit
Class 3 limit
Class 4 limit
Class 5 limit
Amplitude (dBuV)
120
100
80
60
40
20
6
7
10
10
Frequency (Hz)
Figure V.5 - CISPR 25 [97]: Limits for broadband conducted disturbances (peak detector).
5.3 Case Study #1: inverter-fed induction motor drive
In the first case study, the conventional and optimized design of an EMI filter for a low voltage
high current induction motor drives supplied by DC power grids are presented. A comparison of the
optimized filter with the conventionally designed one has been carried out in terms of EMI filter
configuration characteristics, size and performance. Futhermore, useful considerations on filter design
result by an analysis of comparative plots of the best solution and suboptimal designs returned by the
algorithm.
The test bench is composed of:
-
a PWM IGBT voltage source inverter (VSI), realized by an intelligent module
STGIPS10K60A;
-
an Altera Cyclone III FPGA board equipped with a Nial Stewart GPIB expansion board,
implementing the PWM modulator;
-
a 48V induction motor with a rated power of 240W.
The switching frequency fPWM is equal to 20 kHz.
The use of an intelligent module for the VSI allows a very compact layout of the power electronic
stage. In Figure V.6 a view the experimental arrangement of the drive under study is shown.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
Figure V.6 - View of the PWM induction motor drive experimental setup.
Design, set up and comparison of optimized and conventionally designed EMI filter
Measured conducted disturbances have been compared with the limit reported in the MIL 461 F
standard. As shown in Figure V.7, both the CM and DM emission profile exceeds the limit of the
chosen reference standard. This calls for a suitable input EMI filtering.
160
Standard Limit (Mil. 461F)
CM EMI
DM EMI
Amplitude (dBuV)
140
120
100
80
60
40
20
0
10
6
10
7
Frequency (Hz)
Figure V.7 – CM and DM EMI generated by inverter-fed induction motor drive.
On the basis of the conventional and optimized design procedures described in sections III.4 and
IV.2, EMI filters have been set up. In particular, as for the filter realized according to the conventional
procedure, the following data have been used:
-
CM emission peak at the lowest frequency: 96dBµ[email protected];
-
DM emission peak at the lowest frequency: 124dBµ[email protected];
-
cut-off frequencies fo_CM=12.5 kHz and fo_DM= 18 kHz (according to the constraint fo < fPWM);
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Power Density Optimization of EMI Filters for Power Electronic Converters
-
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
CCM=200 nF (maximum value allowed according to SAE AS 1831 Standard).
As regards the filter realized according to the optimized procedure, the measured CM/DM spectra,
shown in Figure V.7, have been loaded into ODEF application and the automatic processing returned
the following filter parameters:
-
Attreq_CM = 30 dBμ[email protected] kHz;
-
Attreq_DM = 60 dBμ[email protected] kHz.
Then, the input data shown in Table V.1 have been entered using the Computation tab of Figure
IV.5 and the computation has been started. It is worth noting that the limit curve of the Military
Standard 461F has been used as EMI limit stardard and the SAE AS 1831 standard has been used for
Cy selection in this case study.
Table V.1
INPUT DATA FOR ODEF APPLICATION – CASE STUDY #1.
Γ-Π
Military Standard 461F
DC system
48 V
5A
32 mA
150 mA
SAE AS 1831
100%
0%
Auto
Filter topology
Reference standard
System type
UN
Imax_phase
Icm_max
Idm_max
Cy standard
Volume coefficient
Weight coefficient
Extra_Ldm_mode
The conventional design procedure leds to a single stage configuration, whereas the optimized
procedure selected a double stage configuration without separate DM inductors. A comparison of the
optimized filter with the conventionally designed one has been carried out verifying their size and
performance. Table V.2 summarizes the results obtained with the two procedures.
Evaluating the volume and weight of the two filters, it is possible to observe that the optimized
design leads to a reduction in volume and weight. In particular, a reduction of 52% in volume and of
56% in weight is obtained. Figure V.8 shows the photo of the optimized EMI filter compared to the
conventionally designed one: the higher compactness is evident.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Table V.2
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
COMPARISON BETWEEN OPTIMIZED AND CONVENTIONALLY-DESIGNED EMI FILTERS (CASE STUDY #1).
Conventional Design
Optimized Design
Number of stages
1
2
[email protected]
0.8 mH
126 µH (each stage)
27.9x13.6x12.5
12x8.0x4.5 (each stage)
65.5 µH
28 µH
CM inductor core dimensions
(mm x mm x mm)
CM core [email protected]
Number of turns per CM
winding
CCM
Cy
[email protected]
CDM
Wire size
(Vitroperm 500F, model T60006-
(Vitroperm 500F, model T60006-L2012-
L2025-W380)
W902) (each stage)
5
3 (each stage)
200 nF
100 nF (each stage)
100 nF, ceramic, 250 V, (Murata
47 nF, ceramic, 250 V, (Murata
RDER72E104K2)
RDER72E473K2) (each stage)
1.6 µH (Lleakage=0.2% LCM)
252 nH (Lleakage=0.2% LCM) (each stage)
47 µF, electrolytic, 400 V,
33 µF, electrolytic, 160 V,
(Panasonic EEUEE2G470)
(Kemet ESG336M160AH4) (each stage)
15 AWG
15 AWG
3
Volume
25.87 cm
13.88 cm3 (all stages)
Weight
44 g
19.12 g (all stages)
Figure V.8 - Photo of conventionally designed EMI filter (on the left) and optimized EMI filter (on the right), in
case study #1.
Comparison of Optimized and Conventionally Designed EMI filter Performance
Finally, in order to evaluate the EMI filters mitigation performance, EMI measurements have been
carried out without any filter, with the conventionally designed filter and with the optimized filter
(Figure V.9). Limit curve relating to the EMI limit imposed by the Military Standard 461F is shown as
well. It is possible to observe in Figure V.9 that both filters show a satisfactory behavior since the EMI
filtered meet the limits imposed by the reference standard.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
Therefore, despite the higher compactness and power density achieved, the optimized EMI filter
still allows to obtain the compliance of the power electronic system under study with the reference
standard.
140
Standard Limit (Mil. 461F)
EMI emission w/o filter
1 stage (conventional design)
2 stage (optimized design)
Magnitude (dBuV)
120
100
80
60
40
20
0
10
6
10
7
Frequency (Hz)
Figure V.9 - Comparison of optimized and conventionally designed EMI filter performance (case study #1).
Considerations on feasible configurations
As described in section 4.3, ODEF allows to compare the optimal EMI filter design to the
suboptimal results, so as to leave the final choice to the designer. Moreover, after selecting the number
of configurations to be considered among the entire feasible set in the Extra tab of ODEF application,
it is possible to generate a series of comparative plots for the chosen case study.
In this case study the application selects the best design (2 stages, no extra LDM, CM core index 27,
total volume=13.88 cm3) among a total of 910 feasible configurations.
Finally, the features of the Extra tab have been used to analyze and compare the feasible
configurations, which can be classified as shown in Figure V.10, according to the number of filter
stages and to the presence/absence of the extra DM inductor. As shown in the figure, the search space
is quite large and the filter volume reaches 6591 cm3 in the worst configuration. The fleasible
configurations selected by ODEF include both configuration with and without extra DM inductor. In
addition it is very important to consider that the number of fleasible configurations is related to the
number of components in the database; so increasing the number of magnetic cores and capacitors in
the database, the number of fleasible configurations will increase. The evaluation by the designer of
such considerable number of configurations is practically cumbersome without the software tool.
94
3
spanned volume (cm )
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
8000
6000
no extra Ldm
extra Ldm
4000
2000
0
1
2
3
n. of stages
4
5
3
n. of stages
4
5
n. of configurations
150
100
no extra Ldm
extra Ldm
50
0
1
2
Figure V.10 - Distribution of all feasible configurations (case study #1).
In order to provide an insight into the configurations that belong to a small neighborhood around
the optimal solution, the distribution of the 15 best designs is plotted in Figure V.11 and Figure V.12.
The 15 best designs include 1-stage, 2-stage and 3-stage filter configurations with a volume range
from about 14 cm3 to 21 cm3. In particular the plot shown in Figure V.12 helps the designer to
3
spanned volume (cm )
compare the volume of different configurations.
50
40
no extra Ldm
extra Ldm
30
20
10
0
1
2
3
n. of stages
4
5
3
n. of stages
4
5
n. of configurations
15
10
no extra Ldm
extra Ldm
5
0
1
2
Figure V.11 - Distribution of the best 15 configurations (case study #1).
95
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
22
21
20
3
volume (cm )
19
18
17
16
15
1-stage
2-stage
3-stage
14
13
0
20
40
60
core index
80
100
120
Figure V.12 - Scatter plot of the best 15 configurations (case study #1).
Furthermore, to evaluate the proximity of the returned solution to other configurations, Figure V.13
shows the volume of the best configuration for each number of stages and Figure V.14 presents the
distribution of the best 100 configurations grouped for number of stages (50 for n=1, 29 for n=2, 11
for n=3, 8 for n=4, 2 for n=5). With reference to Figure V.14, the intersection of the curves related to
1-stage and 2-stage configurations demonstrates that the optimized design for achieving the best
power density is a non-trivial problem. Therefore, it cannot effectively be managed by a trial-and-error
approach.
35
30
3
volume (cm )
25
20
15
10
5
0
1
2
3
n. of stages
4
5
Figure V.13 - Volume of the best configuration for each number of stages (case study #1).
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Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
40
35
3
volume (cm )
30
25
20
1-stage
2-stage
3-stage
4-stage
5-stage
15
10
0
10
20
30
configuration n.
40
50
Figure V.14 - Distribution of the best 100 configurations for different n. of stages (case study #1).
As a further analysis, the design procedure has been repeated several times for increasing values of
the desired CM attenuation, starting from the minimum required value and keeping the other input
parameters constant. The following range has been swept: [30, 32, 34, 36, 38, 40, 42, 44, 46, 48, 50,
52, 54, 56] dBμV.
Figure V.15 and Figure V.16 show the obtained results in terms of volume, number of stages and
CM core index. In the considered case, the best configuration for each value of CM attenuation does
not require the use of an extra DM inductor.
It is worth noting that, as Figure V.15 shows, the CM attenuation of the filter can be increased up to
40 dBμV without increasing the design volume. This corresponds to an extra 10 dBμV safety margin
that allows obtaining a better filter performance, balancing further possible non-idealities in the filter
realization.
Instead in Figure V.16, it is possible observe that the double stage configuration is the best
configuration design for the evaluated CM attenuation range, except for a narrow range (42÷44 dBµV)
where the single stage configuration prevails.
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Optimized EMI filter Design Procedure
18.5
18
17.5
3
volume (cm )
17
16.5
16
15.5
15
14.5
14
13.5
30
35
40
45
50
CM attenuation (dBV)
55
60
Figure V.15 - Volume variation of the best design for increasing CM attenuation (case study #1).
2
40
20
CM core index
n. of stages
30
10
1
30
35
40
45
50
CM attenuation (dBV)
55
1
60
Figure V.16 - Number of stages of the best design for increasing CM attenuation (solid line). CM core index of the
best design for increasing CM attenuation (dashed line). - case study #1.
5.4 Case study #2: inverter-fed symmetric low power resistive load
In the second case study, the conventional and optimized designs of an EMI filter for a symmetric
low power resistive load supplied by DC power grid are reported. A comparison of the optimized filter
with the conventionally designed one has been carried out in terms of EMI filter configuration
characteristics, size and performance. In this case, the optimized filter configuration chosen is one of
the fleasible configurations proposed by ODEF. Even if it does not represent the best configuration
which allows to obtain the EMI filter with the minimum volume; a configuration with the extra LDM
has been chosen to validate the design procedure with extra LDM.
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Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
The test bench consists of a PWM IGBT Voltage Source Inverter that supplies a three phase
resistive load with the following characteristics: rated voltage UN = 48V, rated power PN = 7.2W,
maximum current Imax = 150mA. The VSI is based on a STGIPS10K60A power module and the
switching frequency is equal to 20 kHz.
Design, setup and comparison of optimized and conventionally designed EMI filter
Figure V.17 shows the measured CM and DM EMI; both emission profiles exceed the limits of the
standards that have been chosen as a reference. So a suitable input EMI filtering is necessary. It should
be noted that the load has a significant impact on the noise emission profile. In fact, the test bench of
case study #1 and #2 is the same and it differs only for the three phase load. The CM and DM EMI
profiles shown in Figure V.7 and Figure V.17 are different; in particular, the high frequency noise
contribution of the induction motor can be recognized in the spectrum.
160
Standard Limit (Mil. 461F)
CM EMI
DM EMI
Amplitude (dBuV)
140
120
100
80
60
40
20
0
10
6
10
7
Frequency (Hz)
Figure V.17 - CM and DM EMI generated by inverter-fed symmetric low power resistive load.
A single stage EMI filter designed according to the conventional procedure has been realized and
considered as a benchmark. It should be noted that the conventional design procedure, according to the
constraint on the cutoff frequency lower than the power converter’s switching frequency, leads to the
same filter of the previous case study.
As regards the filter designed according to the optimized procedure, the measured CM/DM spectra,
shown in Figure V.17, have been loaded into ODEF application and the automatic processing returned
the following filter parameters:
-
Attreq_CM = 25 dBμ[email protected] kHz;
-
Attreq_DM = 60 dBμ[email protected] kHz.
Then, the input data shown in Table V.3 have been entered and the computation has been started.
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Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
Table V.3 INPUT DATA FOR ODEF APPLICATION – CASE STUDY #2.
Γ-Π
Military Standard 461F
DC system
48 V
150 mA
45 mA
60 mA
SAE AS 1831
100%
0%
Auto
Filter topology
Reference standard
System type
UN
Imax_phase
Icm_max
Idm_max
Cy standard
Volume coefficient
Weight coefficient
Extra_Ldm_mode
In this case the optimized procedure selected a double stage configuration with separate DM
inductors. A comparison of the optimized filter with the conventionally designed one has been carried
out and Table V.4 summarize the results obtained with the two procedures.
Figure V.18 shows the photo of the optimized EMI filter compared to the conventionally designed
one: the optimized filter is evidently more compact. Evaluating the volume and the weight of the two
filters, it is possible to observe that the optimized design leads to a reduction in volume and weight. In
particular, a reduction of 65% in volume and of 67% in weight is obtained. Despite the realized
optimized filter is not the best configuration provided by ODEF, a considerable improvement in EMI
filter power density is obtained.
Figure V.18 - Photo of conventionally designed EMI filter (on the left) and optimized EMI filter (on the right), in
case study #2.
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Power Density Optimization of EMI Filters for Power Electronic Converters
Table V.4
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
COMPARISON BETWEEN OPTIMIZED AND CONVENTIONALLY-DESIGNED EMI FILTERS (CASE STUDY #2).
Conventional Design
Optimized Design
Number of stages
1
2
[email protected]
0.8 mH
56 µH (each stage)
27.9x13.6x12.5
14.1x6.6x6.3 (each stage)
65.5 µH
28 µH
CM inductor core dimensions
(mm x mm x mm)
CM core [email protected]
Number of turns per CM
winding
CCM
Cy
[email protected]
DM inductor core dimensions
(mm x mm x mm)
(Vitroperm 500F, model T60006-
(Vitroperm 500F, model T60006-L2012-
L2025-W380)
W902) (each stage)
5
2 (each stage)
200 nF
100 nF (each stage)
100 nF, ceramic, 250 V, (Murata
47 nF, ceramic, 250 V, (Murata
RDER72E104K2)
RDER72E473K2) (each stage)
1.6 µH (Lleakage=0.2% LCM)
459 µH (Extra LDM) (each stage)
n.a.
11.2x5.1x5.8 (two for each stage)
25.5 µH
DM core [email protected]
n.a.
(Vitroperm 500F, model T60006-L2009W914) (2 for each stage)
Number of turns per DM
winding
CDM
Wire size
n.a.
3 (each stage)
47 µF, electrolytic, 400 V,
33 nF, polypropylene, 560 V,
(Panasonic EEUEE2G470)
(Kemet 46KF23301P02) (each stage)
15 AWG
21 AWG
3
Volume
25.87 cm
9.88 cm3 (all stages)
Weight
44 g
14.42 g (all stages)
Comparison of Optimized and Conventionally Designed EMI filter Performance
EMI measurements have been carried out in order to evaluate the EMI filters mitigation
performance. As shown in Figure V.19, both filters allow to obtain a fully compliant behavior
concerning the standard limit in the whole frequency range. Then the optimized design procedure with
extra LDM has been validated.
101
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
140
Standard Limit (Mil. 461F)
EMI emission w/o filter
1 stage (conventional design)
2 stage (optimized design)
Amplitude (dBuV)
120
100
80
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.19 - Comparison of optimized and conventionally designed EMI filter performance (case study #2).
5.5 Case study #3: DC motor drive supplied by a DC/DC boost converter
In the third case study, the conventional and optimized designs of an EMI filter for a DC motor
drive supplied by a DC/DC boost converter are reported. A comparison of the optimized filter with the
conventionally designed one has been carried out in terms of EMI filter configuration characteristics,
size and performance. In this case, the best configuration given by the optimized design procedure
without extra LDM has been choosen as optimized filter configuration.
The DUT is composed of a voltage regulator based on a DC/DC boost converter and a DC motor
drive with rated voltage UN = 12 V, rated power PN = 30W and maximum current Imax = 2.5A. The
boost converter is based on the following devices: MURB820: Ultrafast Rectifier, IRFP150N: Power
MOSFET, Inductor 320µH, output capacitance 220µF. The switching frequency is equal to 20 kHz.
Design, set up and comparison of optimized and conventionally designed EMI filter
The measured CM and DM EMI emission profiles (Figure V.20) have been compared with the
limits imposed by the CISPR 25 because this standard is appropriate for the given DUT. Two limit
curves are reported in Figure V.20: the Class 5 limit, which is the most stringent, and the Class 4 limit.
Both CM and DM EMI exceed the limits of the standard that has been chosen as a reference.
EMI filters have been set up according to both design procedure.
In this case study, unlike the previous ones, the conventionally designed EMI filter does not take
into account the constraint on cutoff frequency (fo<fPWM) to verify the filter efficiency without this
rule.
102
Power Density Optimization of EMI Filters for Power Electronic Converters
120
Standard Limit (CISPR 25, Class 5)
Standard Limit (CISPR 25, Class 4)
CM EMI
DM EMI
100
Amplitude (dBuV)
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
80
boost + alzacristalli
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.20 - CM and DM EMI generated by a DC motor drive supplied by a DC/DC boost converter.
The following data have been used:
-
CM emission peak at the lowest frequency: 82.8dBµ[email protected];
-
DM emission peak at the lowest frequency: 111.9dBµ[email protected];
-
cut-off frequencies: fo_CM=56 kHz and fo_DM= 25 kHz;
-
CCM=200 nF (maximum value allowed according to SAE AS 1831 Standard).
The following attenuations have been used for the optimized design procedure:
Attreq_CM = 16.8dBμ[email protected] kHz;
Attreq_DM = 45.9dBμ[email protected] kHz.
Futhermore, the input data shown in Table V.5 have been used to run the optimized design
algorithm. It is worth noting the maximum ground leakage current has been considered for Cy
selection. Moreover, choosing Never in the Extra_Ldm panel, the algorithm included in the search
space only the configurations with the DM filter realized using the leakage inductance of the CM
choke.
Table V.5
Filter topology
Reference standard
System type
UN
Imax
Icm_max
Idm_max
Cy
Volume coefficient
Weight coefficient
Extra_Ldm_mode
INPUT DATA FOR ODEF APPLICATION – CASE STUDY #3.
Γ-Π
CISPR25 Class 5
DC system
12 V
2.5 A
30 mA
56.5 mA
102 nF (maximum ground leakage current = 0.85 mA)
100%
0%
Never
103
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
Also in this case study, the conventional design procedure leds to a single stage configuration,
whereas the optimized procedure selected a double stage configuration. Table V.6 summarizes the
design results. Evaluating the volume and weight of the two filters, it is possible to observe that the
optimized design leads to a reduction of 38% in volume and of 41% in weight.
Table V.6
COMPARISON BETWEEN OPTIMIZED AND CONVENTIONALLY-DESIGNED EMI FILTERS (CASE STUDY #3)
Conventional Design
Number of stages
[email protected]
CM inductor core dimensions
(mm x mm x mm)
CM core [email protected]
Number of turns per CM
winding
CCM
1
60µH
Optimized Design
2
51µH (each stage)
19x11x8.0
11.2x5.1x5.8 (each stage)
30µH
(Vitroperm 500F, model T60006L2017-W515)
25.5µH
(Vitroperm 500F, model T60006-L2009W914) (each stage)
2
2 (each stage)
200 nF
100 nF (each stage)
47 nF, ceramic, 100V, (Murata
RDER72A473K1)
(each stage)
102 nH (Lleakage=0.2% LCM) (each stage)
68 µF, electrolytic,
160 V, (Panasonic EEUEE2C680) (each
stage)
18 AWG
15.54 cm3 (all stages)
20.52 g (all stages)
Cy
100 nF, ceramic, 100V, (Murata
RDER72A104K1)
[email protected]
120nH (Lleakage=0.2% LCM)
CDM
330µF, electrolytic,
160 V, (Panasonic EEUEE2C331)
Wire size
Volume
Weight
18 AWG
25.06 cm3
34.83 g
The layout of the two filters is equal to those shown in Figure V.8. A comparison of the EMI filter
components is shown in Figure V.21: the volume difference is more evident.
Figure V.21 – Comparison of components used to EMI filters setup (case study #3).
104
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
Comparison of Optimized and Conventionally Designed EMI filter Performance
In order to evaluate the EMI filters mitigation performance, EMI measurements have been carried
out without any filter, with the conventionally designed filter and with the optimized filter (Figure
V.22). Both filters show a satisfactory behavior since the mitigated EMI not exceed the applicable
values imposed by the reference standard. In particular, the filters allow to obtain a fully compliant
behavior with the Class 4 limits and an acceptable behavior for the Class 5 limits.
Figure V.22 - Comparison of optimized and conventionally designed EMI filter performance (case study #3).
Considerations on feasible configurations
In this case study the ODEF application selected the best design (2 stages, no extra LDM, CM core
index 26, total volume=14.54 cm3) among a total of 519 feasible configurations without extra LDM
(1038 feasible configurations taking into account also configurations with extra LDM).
The features of the Extra tab have been used to analyze and compare the feasible configurations.
The fleasible configurations without extra LDM can be classified as shown in Figure V.23, according to
the number of filter stages. As the figure shows, the search space is quite large and the filter volume
reaches 6300 cm3 in the worst configuration.
In order to provide an insight into the configurations that belong to a small neighborhood around
the optimal solution, Figure V.24 and Figure V.25 show the distribution of the 30 best designs. The 30
best designs include 1-stage, 2-stage and 3-stage filter configurations with a volume range from about
15 cm3 to 22 cm3. In particular the plot shown in Figure V.25 helps the designer to compare the
volume of the different configurations.
105
spanned volume (cm3)
Power Density Optimization of EMI Filters for Power Electronic Converters
8000
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
no extra Ldm
extra Ldm
6000
4000
2000
0
1
n. of configurations
150
2
3
n. of stages
4
5
3
n. of stages
4
5
no extra Ldm
extra Ldm
100
50
0
1
2
spanned volume (cm3)
Figure V.23 - Distribution of feasible configurations without extra LDM (case study #3).
30
20
10
0
no extra Ldm
extra Ldm
1
2
3
n. of stages
4
5
n. of configurations
20
15
10
5
0
no extra Ldm
extra Ldm
1
2
3
n. of stages
4
5
Figure V.24 - - Distribution of the best 30 configurations without extra LDM (case study #3).
106
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
22
21
volume (cm3)
20
19
18
17
1-stage
2-stage
3-stage
16
15
0
20
40
60
core index
80
100
120
Figure V.25 - Scatter plot of the best 30 no extra LDM configurations (case study #3).
Furthermore, to evaluate the proximity of the returned solution to other configurations, Figure V.26
shows the volume of the best configuration for each number of stages; it can be noted that the 2-stage
and 3-stage configurations occupy a lower volume respect to the 1-stage configuration.
Figure V.27 presents the distribution of the best 100 configurations grouped for number of stages
(32 for n=1, 26 for n=2, 20 for n=3, 14 for n=4, 8 for n=5): there are different intersection points of the
curves related to 1-stage, 2-stage and 3-stage configurations. With reference to Figure V.27, it should
be noted that 14 2-stage configurations and 9 3-stage configurations occupy a lower volume respect to
the best 1-stage configuration. This occurs because the case study requires low attenuation for the CM
noise (16.8dBμ[email protected] kHz) and high attenuation for the DM noise (45.9dBμ[email protected] kHz): then DM
filter components have greater impact on the volume occupied by the EMI filter. Having chosen Never
(Table V.5) the algorithm included in the search space only the DM filters realized using the leakage
inductance of the CM choke. Since the DM required attenuation is high and LDM=Lleakage has generally
a very low value, a high DM capacitance is required and the algorithm chooses the electrolytic
capacitors. A multi stage configuration is preferable to reduce the value and conseguently the size of
electrolytic capacitors. Even if more components are required in the 2-stage and 3-stage configurations
than in a single stage configuration, more compact filter configurations are obtained.
107
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
30
25
volume (cm3)
20
15
10
5
0
1
2
3
n. of stages
4
5
Figure V.26 - Volume of the best configuration for each number of stages (case study #3).
34
32
30
volume (cm3)
28
26
24
22
20
1-stage
2-stage
3-stage
4-stage
5-stage
18
16
14
0
5
10
15
20
configuration n.
25
30
35
Figure V.27 - Distribution of the best 100 configurations for different n. of stages (case study #3).
5.6 Case study #4: DC motor drive supplied by a DC/DC buck converter
In this case study, the DUT is composed of a voltage regulator based on a DC/DC buck converter
and a DC motor drive with rated voltage UN = 12V, rated power PN = 190W and maximum current Imax
= 3.8A. As far as the buck converter is concerned, two different converters with the following devices
are considered:
-
buck converter 1: SKM50GB123D: Power IGBT, Diode with forward voltage VF = 2.2V and
maximum forward current IFmax = 40A, Inductor 500µH, output capacitance 1000µF;
108
Power Density Optimization of EMI Filters for Power Electronic Converters
-
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
buck converter 2: STTH12R06: Ultrafast Rectifier, IRFP150N: Power MOSFET, Inductor
500µH, output capacitance 1000µF.
The switching frequency of both buck converter devices is equal to 10 kHz. The swiching
frequency value is limited by the maximum frequency at which the power IGBT can operate. The
power MOSFET could operate at higher switching frequencies but, in order to maintain the same
operating conditions, the same swiching frequency has been maintained.
The effectiveness and usefulness of the optimized design procedure have been demonstrated in
previous case studies by a comparison of the optimized filters with the conventionally designed one in
terms of size and performance.
Then, in this case study, the EMI filter is designed according to the optimized procedure.
Optimized EMI filter design
At first the CM and DM EMI emission generated by the DC motor drive supplied by a DC/DC
buck converter 1 have been evaluated. In a second step, CM and DM EMI generated by DC motor
drive supplied by a DC/DC buck converter 2 have been measured and compared with those previously
measured to evaluate as the converters, with the same topology but different switching devices, can
influence the EMI generation.
The EMI profiles have been compared with the limits of the CISPR 25 that can be applied for the
given DUT. Both the CM and DM EMI, shown in Figure V.28 exceed the limits of the reference
standard.
120
Standard Limit (CISPR 25, Class 5)
Standard Limit (CISPR 25, Class 4)
CM EMI
DM EMI
Amplitude (dBuV)
100
buck IGBT + ventola
80
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.28 - CM and DM EMI generated by DC motor drive supplied by the DC/DC buck converter 1.
The attenuation values used for the optimized design procedure are: Attreq_CM = 14.4dBμ[email protected]
kHz, Attreq_DM = 37.3dBμ[email protected] kHz. Then, the input data shown in Table V.7 have been entered and
109
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
the computation has been started. Only the configurations with the DM filter realized using the
leakage inductance of the CM choke have been evaluated.
The application selected the best design (2 stages, no extra LDM, CM core index 26, total
volume=11.91 cm3, total weigth=16.61 g) among a total of 486 feasible configurations (except
configurations with extra Ldm). All the details of the optimized EMI filter are given in Table V.8.
INPUT DATA FOR ODEF APPLICATION – CASE STUDY #4 WITH BUCK CONVERTER 1.
Table V.7
Filter topology
Reference standard
System type
UN
Imax
Icm_max
Idm_max
Cy
Volume coefficient
Weight coefficient
Extra_Ldm_mode
Table V.8
Number of stages
[email protected]
CM inductor core dimensions
(mm x mm x mm)
CM core [email protected]
Number of turns per CM winding
CCM
Cy
[email protected]
CDM
Wire size
Volume
Weight
Γ-Π
CISPR25 Class 5
DC system
12 V
3.8 A
39 mA
46 mA
SAE AS 1831
100%
0%
Never
FEATURES OF THE OPTIMIZED EMI FILTER (CASE STUDY #4).
2
51 µH (each stage)
11.2x5.1x5.8 (each stage)
25.5 µH
(Vitroperm 500F, model T60006-L2009-W914) (each stage)
2 (each stage)
100 nF (each stage)
47 nF, ceramic, 100V, (Murata RDER72A473K1) (each stage)
102 nH (Lleakage=0.2% LCM) (each stage)
47 µF, electrolytic, 160 V, (Panasonic EEUEE2C470) (each stage)
15 AWG
11.91 cm3 (all stages)
16.61 g (all stages)
Afterwards CM and DM EMI generated by DC motor drive supplied by a DC/DC buck converter 2
have been measured. As shown in Figure V.29, CM and DM EMI exceed the limit curves. In
particular, Figure V.30 and Figure V.31 show a comparison between CM/DM EMI generated by the
DC motor drive supplied by the buck converter 1 or by the buck converter 2. It should be noted that
the EMI profiles are very similar until 10 MHz and 17 MHz, for CM and DM EMI respectively.
Beyond these frequencies the EMI generated with the buck converter 2 are more relevant. However
the emission peaks at the lowest frequency have the greatest impact on EMI filter design.
Consequently, the EMI filter design according to CM and DM spectrum shown in Figure V.29 leads to
the EMI filter with the same features given in Table V.8.
110
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
buck MOSFET + ventola
120
Standard Limit (CISPR 25, Class 5)
Standard Limit (CISPR 25, Class 4)
CM EMI
DM EMI
Amplitude (dBuV)
100
80
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.29 - CM and DM EMI generated by DC motor drive supplied by the DC/DC buck converter 2.
buck MOSFET + ventola
120
Standard Limit (CISPR 25, Class 5)
Standard Limit (CISPR 25, Class 4)
CM EMI (buck converter 1)
CM EMI (buck converter 2)
Amplitude (dBuV)
100
80
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.30 – Comparison between CM EMI generated by the DC motor drive supplied by the buck converter 1
(solid line) or by the buck converter 2 (dashed line).
111
Power Density Optimization of EMI Filters for Power Electronic Converters
Chapter V – Experimental Validation of the
Optimized EMI filter Design Procedure
120
Standard Limit (CISPR 25, Class 5)
Standard Limit (CISPR 25, Class 4)
DM EMI (buck converter 1)
DM EMI (buck converter 2)
Amplitude (dBuV)
100
buck + ventola
80
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.31 - Comparison between DM EMI generated by the DC motor drive supplied by the buck converter 1
(solid line) or by the buck converter 2 (dashed line).
Evaluation of optimized EMI filter Performance
Finally, in order to evaluate the EMI filter mitigation performance with both DC/DC buck
converter, EMI measurements have been carried out without filter and with the optimized filter
(Figure V.32). The EMI filter shows a satisfactory behavior since the filtered EMI comply with the
limits imposed by the reference standard in both cases.
120
Standard Limit (CISPR 25, Class 5)
Standard Limit (CISPR 25, Class 4)
EMI (buck converter 1)
EMI (buck converter 2)
EMI with optimized filter (buck converter 1)
EMI with optimized filter (buck converter 2)
Amplitude (dBuV)
100
80
60
40
20
0
6
7
10
10
Frequency (Hz)
Figure V.32 – Measured EMI with and without EMI filter (case study #4).
112
buck + ventola
Power Density Optimization of EMI Filters for Power Electronic Converters
Conclusions and Future Developments
CONCLUSIONS and FUTURE DEVELOPMENTS
The research conducted during the PhD course regards the power density issue in EMI filters used
for mitigating EMI in power electronic systems. For this purpose, a broad research on commercially
available materials, with which the discrete EMI filter components are realized (inductors and
capacitors), has been conducted; moreover, their characteristics and performance at high frequencies
(in particular in the range of frequencies of the conducted electromagnetic interference) have been
evaluated.
With regard to the design issues, once the filter topology and component values have been defined,
there are many practical configurations to realize the filter. The identification of the configuration
leading to the best power density in terms of minimum volume/weight is a nontrivial task. The
conventional EMI filter design requires a considerable computational effort and it could not guarantee
the optimal choice of filter components in order to obtain the maximum power density. Then, an
optimized design procedure of discrete EMI filters oriented at obtaining high performing filters with
the minimum volume/weigth, has been developed. The optimized procedure relies on a suitably
devised rule-based algorithm and on databases of commercially available magnetic cores, capacitors
and conducting wires. The optimized algorithm can be implemented by using a common programming
language, within either an open source or a commercial environment, and it exhibits low
computational demand. Therefore, it can be advantageously used both by EMI engineers for obtaining
an optimized filter design and by scientists/experts for the evaluation of filter performance versus
configuration and power density features. Moreover, to make the new design procedure more
accessible to EMI designer, a software tool based on the optimized design procedure has been
developed, namely ODEF (Optimized Design of EMI Filters). All the design options, steps, outputs
and design-related supplementary analyses are managed by ODEF tool in a very intuitive and userfriendly fashion.
The optimized EMI filter design procedure has been validated experimentally, both in terms of
performance and increased power density, in several case studies related to different power electronic
converters configurations and different application fields. In addition to the compliance of the power
electronic systems under study with the reference standards, the optimized procedure has allowed to
achieve EMI filters with considerable higher compactness and power density compared to the
conventional design. In particular, the EMI filters size comparison has been carried out for the
following case studies.
113
Power Density Optimization of EMI Filters for Power Electronic Converters
Conclusions and Future Developments
- In the first case study, in which the conventional and optimized design of an EMI filter for a low
voltage high current induction motor drives supplied by DC power grids have been presented, a
reduction of 52% in volume and of 56% in weight has been obtained.
- In the second case study, the conventional and optimized designs of an EMI filter for a
symmetric low power resistive load supplied by DC power grid have been reported. A reduction
of 65% in volume and of 67% in weight has been obtained.
- The third case study has regarded a DC motor drive supplied by a DC/DC boost converter. The
optimized design has led to a reduction of 38% in volume and of 41% in weight.
This thesis provides a considerable contribution to the power density improvement in power
electronic converters with regard to the EMC compliance of these systems.
Future developments include the investigation of the use of inductors working in partial saturation
and the approach with the digital active EMI filtering technique. The literature provide a few
contributions on these approachs, so it would be interesting to conduct an investigation on these topics
in order to analyze the possible developments and limits.
114
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