HSPICE User Guide, RF Analysis

HSPICE User Guide, RF Analysis
HSPICE User Guide: RF
Analysis
Version B-2008.09, September 2008
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__________________________________________ and its employees. This is copy number __________.”
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trademarks of Synopsys, Inc.
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SystemC is a trademark of the Open SystemC Initiative and is used under license.
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Saber is a registered trademark of SabreMark Limited Partnership and is used under license.
All other product or company names may be trademarks of their respective owners.
ii
HSPICE User Guide: RF Analysis
B-2008.09
Contents
1.
2.
3.
Inside This Manual. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
xiii
The HSPICE Documentation Set. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
xv
Searching Across the HSPICE Documentation Set. . . . . . . . . . . . . . . . . . . . .
xvi
Conventions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
xvii
Customer Support . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
xvii
HSPICE RF Features and Functionality . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
HSPICE RF Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
HSPICE RF Features. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
2
HSPICE and HSPICE RF Differences . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
6
Getting Started. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
9
Running HSPICE RF Simulations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
9
Netlist Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
10
Parametric Analysis Extensions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
10
Generating Output Files. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
10
HSPICE RF Output File Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
11
Using the CosmosScope Waveform Display . . . . . . . . . . . . . . . . . . . . . . . . . .
12
HSPICE RF Tutorial . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
15
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier . . . . . . . . .
15
Example 2: Using HB Analysis for a Power Amplifier . . . . . . . . . . . . . . . . . . .
20
Example 3: Using HB Analysis for an Amplifier . . . . . . . . . . . . . . . . . . . . . . . .
24
Device Model Cards . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
28
Example 4: Using HBOSC Analysis for a Colpitts Oscillator . . . . . . . . . . . . . .
29
Example 5: Using HBOSC Analysis for a CMOS GPS VCO . . . . . . . . . . . . . .
33
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer . . . . . . . . .
41
Two-tone HB Approach . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
42
iii
Contents
HBAC Approach . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
43
Comparing Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
44
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and
a Ring Oscillator. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
46
4.
5.
iv
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
46
Shooting Newton Analysis Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
46
Driven Phase Frequency Detector Example . . . . . . . . . . . . . . . . . . . . . .
47
Ring Oscillator Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
53
Other Shooting Newton Analyses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
57
Demonstration Input Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
58
Input Netlist and Data Entry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
59
Input Netlist File Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
59
Input Line Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
60
Delimiters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
66
Instance Names . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
66
Hierarchy Paths . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
67
Numbers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
68
Parameters and Expressions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
69
Input Netlist File Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
70
Schematic Netlists . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
70
Using Subcircuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
71
Hierarchical Parameters. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
72
DDL Library Access . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
75
Vendor Libraries. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
76
Subcircuit Library Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
77
Parameters and Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
79
Using Parameters in Simulation (.PARAM) . . . . . . . . . . . . . . . . . . . . . . . . . . .
79
Defining Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
79
Assigning Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
81
User-Defined Function Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
82
Predefined Analysis Function. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
83
Measurement Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
83
.PRINT and .PROBE Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
83
Using Algebraic Expressions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
83
Contents
6.
Built-In Functions and Variables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
84
Parameter Scoping and Passing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
88
Library Integrity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
89
Reusing Cells . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
90
Creating Parameters in a Library . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
90
Parameter Defaults and Inheritance. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
92
Parameter Passing Solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
95
Testbench Elements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
97
Passive Elements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
98
Resistors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
98
Capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
104
Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
109
Multi-Terminal Linear Elements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
123
W-element (Distributed Transmission Lines) . . . . . . . . . . . . . . . . . . . . . .
123
Scattering Parameter Data Element . . . . . . . . . . . . . . . . . . . . . . . . . . . .
128
Port Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
133
Port Element Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
133
Using the Port Element for Mixed-Mode Measurement . . . . . . . . . . . . . .
137
Active Elements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
137
Diode Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
137
Bipolar Junction Transistor (BJT) Element . . . . . . . . . . . . . . . . . . . . . . . .
140
JFETs and MESFETs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
142
MOSFETs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
144
Steady-State Voltage and Current Sources . . . . . . . . . . . . . . . . . . . . . . . . . . .
147
I and V Element Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
147
Steady-State HB Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
151
Phase Differences Between HB and SIN Sources . . . . . . . . . . . . . . . . . . . . .
152
Behavioral Noise Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
153
Using Noise Analysis Results as Input Noise Sources . . . . . . . . . . . . . .
155
Power Supply Current and Voltage Noise Sources . . . . . . . . . . . . . . . . .
155
Function Approximations for Distributed Devices . . . . . . . . . . . . . . . . . . . . . .
156
Foster Pole-Residue Form for Transconductance or Gain . . . . . . . . . . . .
157
Advantages of Foster Form Modeling . . . . . . . . . . . . . . . . . . . . . . . . . . .
157
G and E-element Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
157
v
Contents
Complex Signal Sources and Stimuli . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
7.
8.
vi
159
Vector-Modulated RF Source. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
159
Voltage and Current Source Elements. . . . . . . . . . . . . . . . . . . . . . . . . . .
161
SWEEPBLOCK in Sweep Analyses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
168
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
169
Using SWEEPBLOCK in a DC Parameter Sweep . . . . . . . . . . . . . . . . . .
169
Using in Parameter Sweeps in TRAN, AC, and HB Analyses . . . . . . . . .
170
Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
170
Clock Source with Random Jitter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
170
Syntax of SIN, COS, and Pulse Sources . . . . . . . . . . . . . . . . . . . . . . . . .
171
References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
174
Steady-State Harmonic Balance Analysis . . . . . . . . . . . . . . . . . . . . . . . . . .
175
Harmonic Balance Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
176
Harmonic Balance Equations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
177
Features Supported . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
178
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
178
HB Analysis Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
179
HB Analysis Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
181
Harmonic Balance Output Measurements . . . . . . . . . . . . . . . . . . . . . . . .
183
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
184
Calculating Power Measurements After HB Analyses . . . . . . . . . . . . . . .
187
Calculations for Time-Domain Output . . . . . . . . . . . . . . . . . . . . . . . . . . .
190
Output Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
192
Using .MEASURE with .HB Analyses . . . . . . . . . . . . . . . . . . . . . . . . . . .
192
HB Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
194
Errors and Warnings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
195
References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
198
Steady-State Shooting Newton Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . .
199
SN Steady-State Time Domain Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
199
SN Analysis Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
200
SN Analysis Output . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
202
Shooting Newton with Fourier Transform (.SNFT) . . . . . . . . . . . . . . . . . . . . . .
205
.SNFT Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
205
Contents
9.
Oscillator and Phase Noise Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
209
Harmonic Balance or Shooting Newton for Oscillator Analysis . . . . . . . . . . . .
209
Oscillator Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
209
Harmonic Balance Oscillator Analysis (.HBOSC) . . . . . . . . . . . . . . . . . . . . . .
210
Input Syntax for Harmonic Balance Oscillator Analysis . . . . . . . . . . . . . .
211
HB Simulation of Ring Oscillators . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
216
HBOSC Analysis Using Transient Initialization . . . . . . . . . . . . . . . . . . . . . . . .
217
Additional .HBOSC Analysis Options. . . . . . . . . . . . . . . . . . . . . . . . . . . .
218
.HBOSC Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
219
Troubleshooting Convergence Problems . . . . . . . . . . . . . . . . . . . . . . . . .
219
Oscillator Analysis Using Shooting Newton (.SNOSC) . . . . . . . . . . . . . . . . . .
224
.SNOSC Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
226
Phase Noise Analysis (.PHASENOISE) . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
227
Phase Noise Analysis Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
227
Identifying Phase Noise Spurious Signals . . . . . . . . . . . . . . . . . . . . . . . .
229
PHASENOISE Input Syntax. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
230
Phase Noise Algorithms. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
232
PHASENOISE Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
234
Phase Noise Analysis Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
237
Measuring Phase Noise with .MEASURE PHASENOISE . . . . . . . . . . . .
238
Amplitude Modulation/Phase Modulation Separation . . . . . . . . . . . . . . .
239
Accumulated Jitter Measurement for Closed Loop PLL Analysis . . . . . . . . . .
243
Jitter Measurements from Phase Noise . . . . . . . . . . . . . . . . . . . . . . . . . .
244
Small-Signal Phase-Domain Noise Analysis (.ACPHASENOISE). . . . . . . . . .
250
AC Phase Noise Analysis Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
251
ACPHASENOISE Analysis .PRINT/.PROBE Syntax . . . . . . . . . . . . . . . .
251
Errors and Warnings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
252
References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
253
10. Large Signal Periodic AC, Transfer Function, and Noise Analyses . . . . .
255
Multitone Harmonic Balance AC Analysis (.HBAC) . . . . . . . . . . . . . . . . . . . . .
255
Prerequisites and Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
256
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
256
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
257
HBAC Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
259
vii
Contents
Errors and Warnings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
viii
260
Shooting Newton AC Analysis (.SNAC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
261
Prerequisites and Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
261
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
262
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
262
SNAC Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
264
Errors and Warnings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
265
SNAC Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
265
Multitone Harmonic Balance Noise (.HBNOISE) . . . . . . . . . . . . . . . . . . . . . . .
266
Supported Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
267
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
267
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
269
Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
270
Measuring HBNOISE Analyses with .MEASURE . . . . . . . . . . . . . . . . . .
271
Errors and Warnings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
273
HBNOISE Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
273
Shooting Newton Noise Analysis (.SNNOISE) . . . . . . . . . . . . . . . . . . . . . . . .
274
Supported Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
274
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
275
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
278
Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
279
Measuring SNNOISE Analyses with .MEASURE . . . . . . . . . . . . . . . . . .
279
SNNOISE Analysis Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
281
Periodic Time-Dependent Noise Analysis (.PTDNOISE) . . . . . . . . . . . . . . . . .
282
PTDNOISE Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
283
PTDNOISE Output Syntax and File Format. . . . . . . . . . . . . . . . . . . . . . .
286
Error Handling and Warnings. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
289
Usage Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
289
Multitone Harmonic Balance Transfer Function Analysis (.HBXF). . . . . . . . . .
291
Supported Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
291
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
292
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
292
Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
293
Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
293
HBXF Test Listing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
294
Shooting Newton Transfer Function Analysis (.SNXF). . . . . . . . . . . . . . . . . . .
294
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
294
Contents
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
295
Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
296
Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
296
SNXF Test Listing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
297
References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
298
11. S-parameter Analyses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
299
Frequency Translation S-Parameter (HBLIN) Extraction . . . . . . . . . . . . . . . . .
300
HB Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
302
Port-element. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
302
HBLIN Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
303
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
306
Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
307
Large-Signal S-parameter (HBLSP) Analysis . . . . . . . . . . . . . . . . . . . . . . . . .
307
Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
308
Input Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
309
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
311
Output Data Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
311
12. Envelope Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
313
Envelope Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
313
Envelope Analysis Commands. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
314
Output Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
317
Envelope Output Data File Format. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
318
13. Post-Layout Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
321
Post-Layout Back-Annotation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
321
Standard Post-Layout Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
323
Selective Post-Layout Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
326
Additional Post-Layout Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
328
Selective Extraction Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
330
Overview of DSPF Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
331
Overview of SPEF Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
337
Linear Acceleration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
348
PACT Algorithm . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
349
PI Algorithm . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
350
ix
Contents
x
Linear Acceleration Control Options Summary . . . . . . . . . . . . . . . . . . . .
350
14. Statistical and Monte Carlo Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
353
Application of Statistical Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
353
Analytical Model Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
354
Simulating Circuit and Model Temperatures . . . . . . . . . . . . . . . . . . . . . . . . . .
355
Temperature Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
356
.TEMP Statement. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
358
Worst Case Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
358
Model Skew Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
358
Getting Started with Traditional Monte Carlo Simulations . . . . . . . . . . . . . . . .
363
Monte Carlo Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
365
Monte Carlo Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
366
Monte Carlo Output . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
368
.PARAM Distribution Function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
369
Monte Carlo Parameter Distribution . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
371
Monte Carlo Examples. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
372
Worst Case and Monte Carlo Sweep Example . . . . . . . . . . . . . . . . . . . . . . . .
378
Transient Sigma Sweep Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
380
Monte Carlo Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
381
Simulating the Effects of Global and Local Variations with Monte Carlo . . . . .
388
Variations Specified on Geometrical Instance Parameters . . . . . . . . . . .
388
Variations Specified in the Context of Subcircuits . . . . . . . . . . . . . . . . . .
389
Variations on a Model Parameter Using a Local Model in Subcircuit. . . .
391
Indirect Variations on a Model Parameter . . . . . . . . . . . . . . . . . . . . . . . .
391
Variations Specified on Model Parameters . . . . . . . . . . . . . . . . . . . . . . .
392
Variations Specified Using DEV and LOT . . . . . . . . . . . . . . . . . . . . . . . .
393
Combinations of Variation Specifications . . . . . . . . . . . . . . . . . . . . . . . . .
393
15. Using HSPICE with HSPICE RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
395
RF Numerical Integration Algorithm Control . . . . . . . . . . . . . . . . . . . . . . . . . .
395
RF Transient Analysis Accuracy Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
396
.OPTION SIM_ACCURACY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
396
Algorithm Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
396
RF Transient Analysis Output File Formats . . . . . . . . . . . . . . . . . . . . . . . . . . .
398
Contents
Tabulated Data Output . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
399
WDB Output Format. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
399
TR Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
400
NW Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
400
VCD Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
400
turboWave Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
401
Undertow Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
401
CSDF Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
401
Compressing Analog Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
401
Eliminating Voltage Datapoints . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
402
Eliminating Current Datapoints . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
402
16. Advanced Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
403
Creating a Configuration File . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
403
Inserting Comments in a .hspice File. . . . . . . . . . . . . . . . . . . . . . . . . . . .
406
Using Wildcards in HSPICE RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
406
Limiting Output Data Size . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
407
SIM_POSTTOP Option . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
407
SIM_POSTSKIP Option . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
408
SIM_POSTAT Option . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
408
SIM_POSTDOWN Option . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
408
SIM_POSTSCOPE Option. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
409
Probing Subcircuit Currents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
409
Generating Measurement Output Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
411
Optimization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
411
Optimizing AC, DC. and TRAN Analyses . . . . . . . . . . . . . . . . . . . . . . . . .
413
Optimizing HB Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
413
Optimizing HBOSC Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
414
Using CHECK Statements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
415
Setting Global Hi/Lo Levels . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
416
Slew, Rise, and Fall Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
416
Edge Timing Verification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
417
Setup and Hold Verification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
418
IR Drop Detection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
419
POWER DC Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
420
Power DC Analysis Output Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
420
xi
Contents
xii
POWER Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
421
Setting Default Start and Stop Times. . . . . . . . . . . . . . . . . . . . . . . . . . . .
422
Controlling Power Analysis Waveform Dumps . . . . . . . . . . . . . . . . . . . . .
422
Detecting and Reporting Surge Currents. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
423
Index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
425
About This Guide
This manual contains detailed reference information, application examples, and
design flow descriptions that show how HSPICE RF features can be used for
RF circuit characterization. The manual supplements the HSPICE user
documentation by describing the additional features, built on top of the
standard HSPICE feature set, that support the design of RF and high-speed
circuits. Where necessary, the manual describes differences that might exist
between HSPICE RF and HSPICE.
Note:
This manual discusses only HSPICE RF features. For information on other
HSPICE applications, see the other HSPICE manuals, listed in The HSPICE
Documentation Set on page xv.
Inside This Manual
This manual contains the chapters described below. For access to the other
manuals in the HSPICE documentation set, see the next section, Searching
Across the HSPICE Documentation Set on page xvi.
Chapter
Description
Chapter 1, HSPICE RF
Features and Functionality
Introduces HSPICE RF features and functionality.
Chapter 2, Getting Started
Describes how to set up your environment, invoke
HSPICE RF, customize your simulation, and
redirect input and output.
Chapter 3, HSPICE RF
Tutorial
Provides a quick-start tutorial for users new to
HSPICE RF.
Chapter 4, Input Netlist and
Data Entry
Describes the input netlist file and methods of
entering data in HSPICE or HSPICE RF.
Chapter 5, Parameters and
Functions
Describes how to use parameters within HSPICE
RF netlists.
HSPICE User Guide: RF Analysis
B-2008.09
xiii
Inside This Manual
xiv
Chapter
Description
Chapter 6, Testbench
Elements
Describes the specialized elements supported by
HSPICE RF for high-frequency analysis and
characterization and the syntax for the basic
elements of a circuit netlist in HSPICE or HSPICE
RF.
Chapter 7, Steady-State
Harmonic Balance Analysis
Describes how to use harmonic balance analysis for
frequency-driven, steady-state analysis.
Chapter 8, Steady-State
Shooting Newton Analysis
Describes HSPICE RF steady-state time domain
analysis based on Shooting-Newton.
Chapter 9, Oscillator and
Phase Noise Analysis
Describes how to use HSPICE RF to perform
oscillator and phase noise analysis on autonomous
(oscillator) circuits.
Chapter 10, Large Signal
Periodic AC, Transfer
Function, and Noise
Analyses
Describes how to use harmonic balance-based and
Shooting Newton AC analysis as well as nonlinear,
steady-state noise analysis and XF analysis.
Chapter 11, S-parameter
Analyses
Describes how to use periodically driven nonlinear
circuit analyses as well as noise parameter
calculation.
Chapter 12, Envelope
Analysis
Describes how to use envelope simulation.
Chapter 13, Post-Layout
Analysis
Describes the post-layout flow, including post-layout
back-annotation, DSPF and SPEF files, linear
acceleration, check statements, and power
analysis.
Chapter 14, Statistical and
Monte Carlo Analysis
Describes the features available in HSPICE RF for
statistical analysis.
Chapter 15, Using HSPICE
with HSPICE RF
Describes how various analysis features differ in
HSPICE RF as compared to standard HSPICE.
Chapter 16, Advanced
Features
Describes how to invoke HSPICE RF and how to
perform advanced tasks, including redirecting input
and output.
HSPICE User Guide: RF Analysis
B-2008.09
The HSPICE Documentation Set
The HSPICE Documentation Set
This manual is a part of the HSPICE documentation set, which includes the
following manuals:
Manual
Description
HSPICE User Guide:
Simulation and Analysis
Describes how to use HSPICE to simulate and
analyze your circuit designs, and includes simulation
applications. This is the main HSPICE user guide.
HSPICE User Guide:
Signal Integrity
Describes how to use HSPICE to maintain signal
integrity in your chip design.
HSPICE User Guide: RF
Analysis
Describes how to use special set of analysis and
design capabilities added to HSPICE to support RF
and high-speed circuit design.
HSPICE Reference
Manual: Commands and
Control Options
Provides reference information for HSPICE and
HSPICE RF commands and options.
HSPICE Reference
Manual: Elements and
Device Models
Describes standard models you can use when
simulating your circuit designs in HSPICE, including
passive devices, diodes, JFET and MESFET devices,
and BJT devices.
HSPICE Reference
Manual: MOSFET
Models
Describes available MOSFET models you can use
when simulating your circuit designs in HSPICE.
HSPICE Integration to
CadenceTM Virtuoso®
Analog Design
Environment User Guide
Describes use of the HSPICE simulator integration to
the Cadence tool.
AMS Discovery
Simulation Interface
Guide for HSPICE
Describes use of the Simulation Interface with other
EDA tools for HSPICE.
AvanWaves User Guide
Describes the AvanWaves tool, which you can use to
display waveforms generated during HSPICE circuit
design simulation.
HSPICE User Guide: RF Analysis
B-2008.09
xv
Searching Across the HSPICE Documentation Set
Searching Across the HSPICE Documentation Set
You can access the PDF format documentation from your install directory for
the current release by entering -docs on the terminal command line when the
HSPICE tool is open.
Synopsys includes an index with your HSPICE documentation that lets you
search the entire HSPICE documentation set for a particular topic or keyword.
In a single operation, you can instantly generate a list of hits that are hyperlinked to the occurrences of your search term. For information on how to
perform searches across multiple PDF documents, see the HSPICE release
notes.
Note:
To use this feature, the HSPICE documentation files, the Index directory,
and the index.pdx file must reside in the same directory. (This is the default
installation for Synopsys documentation.) Also, Adobe Acrobat must be
invoked as a standalone application rather than as a plug-in to your web
browser.
You can also invoke HSPICE and RF documentation in a browser-based help
system by entering-help on your terminal command line when the HSPICE
tool is open. This provides access to all the HSPICE manuals with the
exception of the AvanWaves User Guide which is available in PDF format only.
Known Limitations and Resolved STARs
You can find information about known problems and limitations and resolved
Synopsys Technical Action Requests (STARs) in the HSPICE Release Notes
shipped with this release. For updates, go to SolvNet.
To access the HSPICE Release Notes:
1. Go to https://solvnet.synopsys.com/ReleaseNotes. (If prompted, enter your
user name and password. If you do not have a Synopsys user name and
password, follow the instructions to register with SolvNet.)
2. Select Download Center> HSPICE> version number> Release Notes.
xvi
HSPICE User Guide: RF Analysis
B-2008.09
Conventions
Conventions
The following yypographical conventions are used in Synopsys HSPICE
documentation.
Convention
Description
Courier
Indicates command syntax.
Italic
Indicates a user-defined value, such as object_name.
Bold
Indicates user input—text you type verbatim—in syntax and
examples.
[ ]
Denotes optional parameters, such as:
write_file [-f filename]
...
Indicates that parameters can be repeated as many times as
necessary:
pin1 pin2 ... pinN
|
Indicates a choice among alternatives, such as
low | medium | high
+
Indicates a continuation of a command line.
/
Indicates levels of directory structure.
Edit > Copy
Indicates a path to a menu command, such as opening the
Edit menu and choosing Copy.
Control-c
Indicates a keyboard combination, such as holding down the
Control key and pressing c.
Customer Support
Customer support is available through SolvNet online customer support and
through contacting the Synopsys Technical Support Center.
HSPICE User Guide: RF Analysis
B-2008.09
xvii
Customer Support
Accessing SolvNet
SolvNet includes an electronic knowledge base of technical articles and
answers to frequently asked questions about Synopsys tools. SolvNet also
gives you access to a wide range of Synopsys online services, which include
downloading software, viewing Documentation on the Web, and entering a call
to the Support Center.
To access SolvNet:
1. Go to the SolvNet Web page at http://solvnet.synopsys.com.
2. If prompted, enter your user name and password. (If you do not have a
Synopsys user name and password, follow the instructions to register with
SolvNet.)
If you need help using SolvNet, click Help on the SolvNet menu bar.
Contacting the Synopsys Technical Support Center
If you have problems, questions, or suggestions, you can contact the Synopsys
Technical Support Center in the following ways:
■
Open a call to your local support center from the Web by going to
http://solvnet.synopsys.com/EnterACall (Synopsys user name and
password required).
■
Send an e-mail message to your local support center.
■
xviii
•
E-mail [email protected] from within North America.
•
Find other local support center e-mail addresses at
http://www.synopsys.com/support/support_ctr.
Telephone your local support center.
•
Call (800) 245-8005 from within the continental United States.
•
Call (650) 584-4200 from Canada.
•
Find other local support center telephone numbers at
http://www.synopsys.com/support/support_ctr.
HSPICE User Guide: RF Analysis
B-2008.09
1
HSPICE RF Features and Functionality
1
Introduces HSPICE RF features and functionality.
HSPICE RF is a special set of analysis and design capabilities that support the
design of RF and high-speed circuits. This functionality, built on top of the
standard HSPICE feature set, is also useful for analog and signal integrity
applications. Although the HSPICE and HSPICE RF simulators share a
common set of device models and simulation capabilities, HSPICE RF includes
several modeling, simulation, and measurement additions that augment the
ultimate-accuracy analog circuit simulation capabilities of HSPICE.
Note:
This manual describes the additional features and capabilities of HSPICE
RF. Where necessary, the manual describes differences between HSPICE
RF and HSPICE. For information about standard HSPICE device models,
syntax, and simulation control, you can refer to one of the other HSPICE
manuals in the HSPICE documentation set, listed in The HSPICE
Documentation Set on page xv.
These topics are covered in the following sections:
■
HSPICE RF Overview
■
HSPICE and HSPICE RF Differences
HSPICE RF Overview
HSPICE RF consists of:
■
The hspicerf simulation engine
■
The CosmosScope (cscope) waveform display tool
HSPICE User Guide: RF Analysis
B-2008.09
1
Chapter 1: HSPICE RF Features and Functionality
HSPICE RF Overview
The hspicerf simulation engine contains extensions to HSPICE for RF design.
These extensions are in the form of new analysis commands and new
elements. The hspicerf simulation engine processes command and element
syntax for new RF simulation features but also accepts standard HSPICE
netlist files as input.
The CosmosScope waveform display tool has been enhanced with special
features for reading and analyzing data created by the HSPICE RF simulation
engine. For a basic overview on how to use CosmosScope to view HSPICE RF
output, see Using the CosmosScope Waveform Display on page 12.
HSPICE RF Features
This section briefly introduces the features of both the simulation engine and
the waveform display tool.
HSPICE RF supports most HSPICE capabilities, and also includes:
■
Steady-state frequency-domain analyses for linear and nonlinear circuits.
■
High-performance transient analysis for faster simulation of high-speed
digital and analog circuits.
■
Port-wise automated .AC analyses for S (scattering) parameters. The.LIN
command invokes extraction of noise and linear transfer parameters of a
multi-port linear network. Extracts the S parameter and generates the Nport model.
This command is used in conjunction with the .AC command to measure
multiport S, Y, and Z parameters, noise parameters, stability and gain
factors, and matching coefficients. Additionally, it is used with the Port
element, which identifies the network ports and their impedances. You can
also use mixed mode with .LIN.
2
■
The Port (P) element identifies ports used in LIN analysis (multiport S, Y, or
Z parameter and noise parameter extraction). A port element behaves as a
noiseless impedance or a voltage source in series with an impedance,
depending on the simulation being performed. Different impedances can be
specified for DC, transient, AC, HB, and HBAC analyses.
■
The S element describes a linear network using multi-port S, Y, or Z
parameters in the form of a frequency table. These parameters can come
from a .LIN simulation or from physical measurement. The standard
Touchstone and CITIfile formats are supported in addition to a proprietary
HSPICE format.
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 1: HSPICE RF Features and Functionality
HSPICE RF Overview
■
The syntax of voltage and current sources as well as Port elements supports
the syntax for specifying power sources. In this case, the source value is
interpreted as a power value in Watts or dBm units, and the Port element is
implemented as a voltage source with a series impedance. The.HBLSP
command invokes periodically driven nonlinear circuit analyses for powerdependent S parameters.
■
Harmonic Balance (.HB) analysis using Direct and Krylov solvers. The.HB
command invokes the single and multitone Harmonic Balance algorithm for
periodic steady state analysis.
■
TRANFORHB element parameter to recognize V/I sources that include SIN
and PULSE transient descriptions as well as PWL and VMRF sources.
■
Harmonic balance-based periodic AC analysis. The .HBAC command
invokes periodic AC analysis for analyzing small-signal perturbations on
circuits operating in a large-signal periodic steady state.
■
Harmonic Balance-based Periodic Noise analysis (.HBNOISE) for noise
analysis of periodically modulated circuits, includes stationary,
cyclostationary, and frequency-dependent noise effects.
■
Autonomous Harmonic Balance analysis. The.HBOSC command invokes
the multitone, oscillator-capable Harmonic Balance algorithm for periodic
steady state analysis.
■
Perturbation analysis for Oscillator Phase Noise. The .HBAC command
invokes phase periodic AC noise for oscillators circuits operating in a largesignal steady-state.
■
Oscillator phase noise analysis, including both a nonlinear perturbation
method and a PAC method, and includes stationary, cyclostationary,
frequency-dependent, and correlated noise effects.
■
Frequency translation S-parameter and noise figure extraction with the
.HBLIN command.
■
Envelope analysis. The.ENV command: invokes standard envelope
simulation. The .ENVOSC command invokes envelope startup simulation.
The.ENVFFT command invokes envelope Fast Fourier Transform
simulation.
■
.OPTION HBTRANINIT, HBTRANPTS, and HBTRANSTEP for transient
analysis of ring oscillators.
■
Convolution for transient analysis of S-parameter data models (S-element).
HSPICE User Guide: RF Analysis
B-2008.09
3
Chapter 1: HSPICE RF Features and Functionality
HSPICE RF Overview
■
Calculation of the transfer function from an arbitrary source and harmonic in
the circuit to a designated output with the .HBXF command.
■
Reading encrypted netlists.
■
.OPTION SIM_ACCURACY provides simplified accuracy control for all
simulations while .OPTION SIM_ORDER and SIM_TRAP improve transient
analysis simulation controls.
■
DSPF Flow for fast analysis using parasitic data from layout.
■
.OPTION SIM_LA provides linear acceleration for RC network reduction for
faster simulation.
■
Saving .PRINT simulation output to a separate file.
■
HERTZ variable for frequency-dependent equations.
■
IC=OFF in element statements, IC parameter (initial conditions).
■
Shooting Newton steady-state time domain analysis; the Shooting Newton
algorithm provides functionality to support the following commands: .SN,
.SNAC .SNFT, .SNNOISE, .SNOSC, and .SNXF.
■
Periodic Time-Dependent Noise Analysis (.PTDNOISE) calculates the
noise spectrum and the total noise at a point in time. Jitter in a digital
threshold circuit can then be determined from the total noise and the digital
signal slew rate.
■
PSF output using the PSF and ARTIST options to include HSPICE RF
analyses such as Harmonic Balance, Shooting Newton, and their
associated small-signal analyses within the CadenceTM design
environment.
■
With the 2008.09 release, the SX-WaveView waveform viewing tool is
capable of displaying the following HSPICE RF output file formats: .tr# .sc#
.hb#, .hr#, .ev#, .fe#, .jt#, .pn#, .ls#, .p2d#, .sw#, .ac#, .msn#, .sn#, and
Touchstone formatted files.
HSPICE RF also adds the following measurement capabilities to HSPICE:
4
■
Small-signal scattering parameters.
■
Small-signal two-port noise parameters.
■
1 dB compression point.
■
Intercept points (for example, IP2, IP3).
■
Mixer conversion gain and noise figure.
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 1: HSPICE RF Features and Functionality
HSPICE RF Overview
■
VCO output spectrum.
■
Oscillator phase noise.
Options simplify specifying levels of accuracy. As a result, HSPICE RF provides
effective simulation solutions for RF, high-speed, and PCB signal integrity
circuit challenges.
Verilog-A is supported for all HSPICE RF analyses, including:
■
HB
■
HBOSC
■
HBAC
■
HBNOISE
■
HBXF
■
PHASENOISE
■
SN
■
SNOSC
■
ENV
There are standard restrictions for Verilog-A in periodic steady-state analysis
and are the same as other RF simulators that use Verilog-A. For example:
■
Verilog-A modules that are time dependent cannot be used for HB or SN
unless the time dependence is periodic with a period that matches the HB
or SN setup.
■
Verilog-A modules with "internal states" are not guaranteed to work correctly
in HB or SN because the internal state cannot be tracked by the engine, so
HB or SN may think it is converged to a periodic steady-state even though
the internal state may not be in periodic steady state.
■
Some event-driven constructs in Verilog-A may not be compatible with HB.
HSPICE User Guide: RF Analysis
B-2008.09
5
Chapter 1: HSPICE RF Features and Functionality
HSPICE and HSPICE RF Differences
HSPICE and HSPICE RF Differences
The following tables give an overview of which features (Table 1) and device
models (Table 2 on page 8) in HSPICE are not supported in HSPICE RF.
Table 1
HSPICE Features Not in HSPICE RF
Feature
See
Read hspice.ini file.
HSPICE User Guide: Simulation and Analysis
Short names for internal sub-circuits, such as
10:M1.
HSPICE User Guide: Simulation and Analysis
.MODEL types: AMP and PLOT for graphs
HSPICE Reference Manual: Commands and
Control Options
Parameter definition (.PARAM) for Monte Carlo
statistical functions
HSPICE Reference Manual: Commands and
Control Options
.PLOT simulation output (obsolete for HSPICE
too which uses .PRINT)
HSPICE Reference Manual: Commands and
Control Options
.GRAPH simulation output (uses PLOT model
type)
HSPICE User Guide: Simulation and Analysis
HSPICE Reference Manual: Commands and
Control Options
.WIDTH, and many .OPTION COMMANDS
HSPICE User Guide: Simulation and Analysis
HSPICE Reference Manual: Commands and
Control Options
.OPTION ACCT
HSPICE User Guide: Simulation and Analysis
HSPICE Reference Manual: Commands and
Control Options
Element template output
HSPICE User Guide: Simulation and Analysis
Condition-controls: IF-ELSEIF-ELSE-ENDIF
structure is not currently supported in HSPICE
RF; however, you may be able to use the "a ?
b : c" construct in expressions, which performs
if/then/else for expression evaluation
HSPICE User Guide: Simulation and Analysis
HSPICE Reference Manual: Commands and
Control Options
6
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 1: HSPICE RF Features and Functionality
HSPICE and HSPICE RF Differences
Table 1
HSPICE Features Not in HSPICE RF (Continued)
Feature
See
Group time delay parameters in AC analysis
output
HSPICE User Guide: Simulation and Analysis
.DISTO distortion analysis and associated
output commands
HSPICE User Guide: Simulation and Analysis
.SAVE and .LOAD
HSPICE Reference Manual: Commands and
Control Options
Options that activate unsupported features in
HSPICE RF:
FAST GSHDC GSHUNT
LIMPTS OFF RESMIN
TIMERES
All version options
HSPICE Reference Manual: Commands and
Control Options
Options ignored by HSPICE RF, because they
are not needed since they are replaced by
automated algorithms:
ABSH
ABSV ABSVAR
BELV
BKPSIZ CHGTOL
CONVERGE CSHDC CVTOL
DCFOR DCHOLD DCON
DCSTEP DI
DV
DVDT
FAST FS
FT
GMAX GRAMP
GSHDC GSHUNT ICSWEEP
IMAX
IMIN ITL3
ITL5
ITLPZ LIMPTS
LVLTIM MAXAMP MBYPASS
NEWTOL RELH RELI
RELQ
RELV RELVAR
TRTOL
All matrix options
All error options
Some Transient/AC input/output
(I/O) options. HSPICE RF does support POST
and PROBE options.
HSPICE User Guide: RF Analysis
B-2008.09
HSPICE Reference Manual: Commands and
Control Options
7
Chapter 1: HSPICE RF Features and Functionality
HSPICE and HSPICE RF Differences
Table 1
HSPICE Features Not in HSPICE RF (Continued)
Feature
See
Sub-circuit cross-listing in a .pa file
HSPICE User Guide: Simulation and Analysis,
Chapter 3
-r command-line argument for a remote host
HSPICE User Guide: Simulation and Analysis
.OP supports node voltage for any time, but
supports element values only for t=0.
HSPICE Reference Manual: Commands and
Control Options
Sensitivity analysis (.SENS)
HSPICE User Guide: Simulation and Analysis
HSPICE Reference Manual: Commands and
Control Options
DC mismatch analysis (.DCMATCH)
HSPICE User Guide: Simulation and Analysis
HSPICE Reference Manual: Commands and
Control Options
Table 2
Device Models Not in HSPICE RF
Model
See
B-element: IBIS buffer—
HSPICE User Guide: Signal Integrity
Bname n1 n2 [...] parameters
data-driven I element (current source)
HSPICE Reference Manual: Elements and Device
Models
data-driven V element (voltage source)
HSPICE Reference Manual: Elements and Device
Models
BJT LEVEL=10 (MODELLA)
HSPICE Reference Manual: Elements and Device
Models, Chapter 5
MOSFET Levels 4-8.
HSPICE Reference Manual: MOSFET Models
8
HSPICE User Guide: RF Analysis
B-2008.09
2
Getting Started
2
Describes how to set up your environment, invoke HSPICE RF, customize your
simulation, redirect input and output, and use the CosmosScope waveform
display tool.
Before you run HSPICE RF, you need to set up several environment variables.
You can also create a configuration file to customize your simulation run.
HSPICE RF accepts a netlist file from standard input and delivers the ASCII
text simulation results to HTML or to standard output. Error and warning
messages are forwarded to standard error output.
These topics are covered in the following sections:
■
Running HSPICE RF Simulations
■
Netlist Overview
■
Parametric Analysis Extensions
■
Generating Output Files
■
Using the CosmosScope Waveform Display
Running HSPICE RF Simulations
Use the following syntax to invoke HSPICE RF:
hspicerf [-a] inputfile [outputfile] [-n] [-h] [-v]
For a description of the hspicerf command syntax and arguments, see
section HSPICE RF Command Syntax in the HSPICE Reference Manual:
Commands and Control Options.
HSPICE User Guide: RF Analysis
B-2008.09
9
Chapter 2: Getting Started
Netlist Overview
Netlist Overview
The circuit description syntax for HSPICE RF is compatible with the SPICE and
HSPICE input netlist format. For a description of an input netlist file and
methods of entering data, see Chapter 4, Input Netlist and Data Entry in this
manual.
Parametric Analysis Extensions
All major HSPICE RF analyses (.TRAN, .AC, .DC, and .HB) support the
following parameter sweeps with the same syntax as standard HSPICE:
■
LIN
■
DEC
■
OCT
■
DATA
■
POI
You can also use the MONTE keyword for a Monte Carlo analysis or the
OPTIMIZE keyword for optimization.
Generating Output Files
HSPICE RF generates a table of simulation outputs.
■
If the output is text (the default), the text is put into a .lis file.
■
If you specify .OPTION POST, then HSPICE RF generates simulation
output in a format suitable for a waveform display tool.
■
The default output format for transient analysis in HSPICE RF is the same
as in HSPICE: the .tr0 file format. For additional information, see Standard
Output Files in the HSPICE User Guide: Simulation and Analysis.
The Synopsys interactive waveform display tool, CosmosScope, can display
both the text simulation results and binary output within the X-window
environment.
All output functions (.PRINT, .PROBE, .MEASURE, and so on) can use power
output variables in the form p(devicename), just as in HSPICE. You can also
use the “power” keyword.
10
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 2: Getting Started
Generating Output Files
Larger output files from multi-million transistor simulations might not be
readable by some waveform viewers. Options are available that enable you to
limit the output file size. See Limiting Output Data Size on page 407 for more
information.
HSPICE RF Output File Types
Table 3 shows the output file extensions that HSPICE RF analyses produce.
The base file name of each output file is the same as the input netlist file’s base
name. The # at the end of each file extension represents the .ALTER run from
which the file came.
In general, text output from .PRINT commands is intended to be read by
humans, while binary output from .PROBE or .OPTION POST is intended to be
read by the CosmosScope waveform display tool.
Table 3
HSPICE RF Output File Types
Command
Text Output
Output for CosmosScope
AC analysis (.AC)
.printac#
.ac#
AC noise analysis (.NOISE) .printac#
.ac#
DC sweep (.DC)
.printsw#
.sw#
Envelope analysis (.ENV)
.printev#
.ev#
Envelope FFT (.ENVFFT)
(none)
.fe#
Harmonic Balance (.HB)
.printhb#
.hb#
Harmonic Balance AC
(.HBAC)
.printhb#
.hb#
.HBLIN analysis
.PRINT output: .printhl#
S-param output: .SnP
.PROBE output: .hl#
S-paramr output: .SnP
.HBLSP large-signal
.PRINT output: .printls#
S-param output: .p2d#
.PROBE output: .ls#
S-param output: .p2d#
.HBLSP small-signal
.PRINT output: .printss#
S/noise output: .S2P#
.PROBE output: .ss#
S/noise output: .S2P#
HSPICE User Guide: RF Analysis
B-2008.09
11
Chapter 2: Getting Started
Using the CosmosScope Waveform Display
Table 3
HSPICE RF Output File Types
Command
Text Output
Output for CosmosScope
HBAC noise (.HBNOISE)
.printsnpn#
.pn#
Harmonic Balance OSC
(.HBOSC)
.printhb#
.hb#
Harmonic Balance TRAN
(.HBTRAN)
.printhr#
.hr#
Transfer Functions (.HBXF)
.printxf#
.xf#
Oscillator startup
(.ENVOSC)
.printev#
.ev#
.LIN analysis
.PRINT output: .printac#;
S/noise output: .sc#, .SnP,
.citi#
.PROBE output: .ac#;
S/noise output: .sc#,
.SnP, .citi#
Phase Noise
(.PHASENOISE)
.printsnpn#
.pn#
.SN analysis
.printsn#
.sn#
Transient analysis (.TRAN)
.printtr#
.tr#
Using the CosmosScope Waveform Display
CosmosScope has been enhanced to support viewing and processing of
HSPICE RF output files. This section presents a basic overview of how to use
CosmosScope to view HSPICE RF output.
12
■
Type cscope on the UNIX command line to start the CosmosScope tool.
■
Choose File > Open > Plotfiles (or just press CTRL-O) to open the Open
Plotfiles dialog. Use the Files of Type filter to find the HSPICE RF output file
that you want to open. Table 3 on page 11 lists the HSPICE RF file types.
When you open a file, its contents appear in the Signal Manager window.
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 2: Getting Started
Using the CosmosScope Waveform Display
■
The Signal Manager lists all open plot files. If you double-click a plot file
name, a new window appears, showing the contents of that plot file. To plot
one of the signals listed here in the active chart, double click on the signal
label.
■
To create a new chart, use the File > New menu. Select either XY Graph,
Smith Chart, or Polar Chart. You can also use the first three icons in the
toolbar to create new chart windows.
■
To display the Signal Menu, right-click a signal label in a chart. Using this
menu, you can change how signals look, delete signals, or move signals
from one chart panel to another.
■
•
Use the Attributes menu item to control how the signal looks.
•
Use the Stack Region menu to move signals. You can move a signal to
a new panel or an existing panel. The existing panels are named
“Analog 0”, “Analog 1”, and so on; “Analog 0” is the bottom panel on a
chart.
•
Use the To Time Domain command to convert a histogram plot (for
example, from a .hb0 file) to a time domain signal.
Right-click a horizontal or vertical axis to control an axis. Using the Axis
Attributes dialog, you can use the Axis Menu to configure the axis precisely.
•
Use the Range submenu to zoom in or out.
•
Use the Scale submenu to switch between linear and logarithmic
scales.
•
Lock Out New Signals creates an independent axis when you create a
new panel.
•
Display Range Slider displays a region next to the axis. Click in that
region to pan the display right, left, up, or down.
■
To zoom in and out, use the Axis Attributes dialog, the zoom buttons on the
tool bar, or the mouse directly on the chart window.
■
To attach a marker to a signal, click on a signal label, then click the Vertical
Marker or Horizontal Marker icons in the tool bar. You can use the mouse to
drag the marker along the signal to see the signal’s precise value at different
points.
■
Choose Tools > Calculator to open the Waveform Calculator tool. This tool
can be used to generate new waveforms from existing ones. It is described
in detail in the CosmosScope User Guide. The waveform calculator has no
RF-specific features.
HSPICE User Guide: RF Analysis
B-2008.09
13
Chapter 2: Getting Started
Using the CosmosScope Waveform Display
■
■
14
Tools > Measurement opens the Measurement Tool. Three RF
measurements have been added, under the RF submenu of the
measurement selection menu:
•
1db compression point (1DB CP).
•
IIP3/OIP3.
•
Spurious free dynamic range (SFDR).
Tools > RF Tool opens the RF Tool, which generates contour plots on Smith
or Polar charts. In HSPICE RF, the plotfile must be a file with a .sc#
extension that a .LIN command generates. HSPICE RF automatically finds
the S parameter and noise parameter data in the .sc# file, and uses it to
generate noise, gain, and stability circles.
HSPICE User Guide: RF Analysis
B-2008.09
3
HSPICE RF Tutorial
3
Provides a quick-start tutorial for users new to HSPICE RF.
This tutorial assumes you are familiar with HSPICE and general HSPICE
syntax, but new to RF analysis features. The most basic RF analysis features
are presented here, using simple examples. The end of this chapter contains a
listing of HSPICE RF demonstration files available for your use when you have
access to the HSPICE RF installation directory.
This tutorial covers the following examples:
■
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier
■
Example 2: Using HB Analysis for a Power Amplifier
■
Example 3: Using HB Analysis for an Amplifier
■
Example 4: Using HBOSC Analysis for a Colpitts Oscillator
■
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
■
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
■
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency
Circuit and a Ring Oscillator
In addition, there is a section listing all RF examples, including those not
covered in these tutorial examples:
■
Demonstration Input Files
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier
The .LIN command simplifies the calculation of linear multi-port transfer
parameters and noise parameters. In the LIN analysis, Port (P) elements are
used to specify port numbers and their characteristic impedances. The analysis
HSPICE User Guide: RF Analysis
B-2008.09
15
Chapter 3: HSPICE RF Tutorial
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier
automatically computes the frequency-dependent complex transfer coefficients
between ports. The result is a convenient means to get scattering parameters,
noise parameters, stability parameters, and gain coefficients. The .LIN
command renders obsolete the .NET command. The output from the .LIN
command is saved in the *.sc0 file format that can, in turn, be referenced as a
model file for the new S-parameter element.
To set up a linear transfer parameter analysis, the HSPICE input netlist must
contain the following:
■
Use the .AC command to activate small-signal AC analysis, and to specify
a frequency sweep. Also, use the .AC command to specify any other
parameter sweeps of interest.
■
Use the .LIN command with the .AC command to activate small-signal
linear transfer analysis. The .AC command specifies the base frequency
sweep for the LIN analysis. The LIN analysis automatically performs multiple
AC and NOISE analyses, as needed to compute all complex signal transfer
parameters.
■
The necessary number of port (P) elements, numbered sequentially
beginning with one to define the terminals of the multi-port network. For
example, a two-port circuit must contain two port elements with one listed
as port=1 and the other as port=2. The port elements define the ordering for
the output quantities from the .LIN command (for example, the terminals
for port=1 are used for S11, Y11, and Z11 measurements).
Much of the LIN analysis is automated so the HSPICE input netlist often does
not require the following:
■
AC signal sources. The .LIN command computes transfer parameters
between the ports with no additional AC sources needed.
■
DC sources. You can analyze a purely passive circuit without adding
sources of any kind.
The following tutorial example shows how to set up a LIN analysis for an NMOS
low noise amplifier circuit. This netlist is shipped with the HSPICE RF
distribution as gsmlna.sp and is available in the directory:
$installdir/demo/hspicerf/examples.
16
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier
Figure 1
Schematic Showing Instantiation for Low Noise Amplifier
HSPICE User Guide: RF Analysis
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17
Chapter 3: HSPICE RF Tutorial
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier
** NMOS 0.25um Cascode LNA for GSM applications
** setup for s-parameter and noise parameter measurements
* Revision: 2.0, change to HB analysis and add measurements
* 2 tone HB analysis, 1 tone as input, 1 tone as interfere
Port element as power source, sweep input power
**
.temp 27
.options post=2
.param
Vdd=2.3
.global gnd
**
** Cascode LNA tuned for operation near 1 GHz
**
M1 _n4 _n3 _n5 _n5 CMOSN l=0.25u w=7.5u
+
as=15p ad=15p ps=19u pd=19u m=80
M2 _n6 _n1 _n4 _n4 CMOSN l=0.25u w=7.5u
+
as=15p ad=15p ps=19u pd=19u m=80
M3 rfo _n6 gnd gnd CMOSN l=0.25u w=7.5u
+
as=15p ad=15p ps=19u pd=19u m=40
r1 _vdd _n6 400
l1 _n5 gnd l=0.9nH
l2 rfin _n3 l=13nH
vvb _n1 gnd
dc=1.19 $ bias for common base device
vvdd _vdd gnd dc=Vdd
rfb rfo _n6 120 $ feedback
**
** 50 Ohm input port (incl. bias), 255 Ohm output port.
**
P1 rfin gnd port=1 z0=50 dc = 0.595 $ input port includes DC bias
P2 rfo _vdd port=2 z0=255 $ port doubles as pull-up resistor
**
** Measure s-parameters and noise parameters
**
.AC DEC 50 100MEG 5G
.LIN noisecalc=1 sparcalc=1
.PRINT S11(DB) S21(DB) S12(DB) S22(DB) NFMIN
**
** Approximate parameters for TSMC 0.25 Process (MOSIS run T17B)
**
.MODEL CMOSN NMOS (
LEVEL
= 49
3.1
TNOM
= 27
TOX
= 5.8E-9
+XJ
= 1E-7
NCH
= 2.3549E17
VTH0
= 0.3819327
+K1
= 0.477867
K2
= 2.422759E-3
K3
= 1E-3
+K3B
= 2.1606637
W0
= 1E-7
NLX
= 1.57986E-7
+DVT0W
= 0
DVT1W
= 0
DVT2W
= 0
18
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 1: Using .LIN Analysis for a NMOS Low Noise Amplifier
+DVT0
+U0
+UC
+AGS
+KETA
+RDSW
+WR
+XL
+DWB
+CIT
+CDSCB
+DSUB
+PDIBLC2
+PSCBE1
+DELTA
+PRT
+KT1L
+UB1
+WL
+WWN
+LLN
+LWL
+CGDO
+CJ
+CJSW
+CJSWG
+CF
+PK2
*
.END
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
0.5334651
289.1720829
2.841618E-11
0.2874763
-2.395348E-3
178.7751373
1
3E-8
4.613042E-9
0
0
0.0463218
4.422611E-3
7.982649E10
0.01
0
0
-7.61E-18
0
1
1
0
5.62E-10
1.641005E-3
4.179682E-10
3.29E-10
0
2.650965E-3
DVT1
UA
VSAT
B0
A1
PRWG
WINT
XW
VOFF
CDSC
ETA0
PCLM
PDIBLCB
PSCBE2
RSH
UTE
KT2
UC1
WLN
WWL
LW
CAPMOD
CGSO
PB
PBSW
PBSWG
PVTH0
WKETA
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
0.7186877
-1.300598E-9
1.482651E5
-1.833193E-8
0
0.3774172
0
-4E-8
-0.0981658
2.4E-4
5.128492E-3
1.91946
-0.1
5.200359E-10
3.7
-1.5
0.022
-5.6E-11
1
0
0
2
5.62E-10
0.99
0.99
0.99
-8.385037E-3
7.293869E-3
DVT2
UB
A0
B1
A2
PRWB
LINT
DWG
NFACTOR
CDSCD
ETAB
PDIBLC1
DROUT
PVAG
MOBMOD
KT1
UA1
AT
WW
LL
LWN
XPART
CGBO
MJ
MJSW
MJSWG
PRDSW
LKETA
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
=
-0.5
2.3082E-18
1.6856991
-1E-7
0.4177975
-0.2
1.88839E-8
-1.2139E-8
1.2032376
0
6.18609E-4
1
0.9817908
9.31443E-3
1
-0.11
4.31E-9
3.3E4
0
0
1
0.5
1E-12
0.4453094
0.3413857
0.3413857
-10
-6.070E-3)
A LIN analysis also includes the following:
■
.LIN command:
.LIN noisecalc=1 sparcalc=1
This invokes a LIN analysis and activates noise calculations and S
parameter output files.
■
Two port elements:
P1 rfin gnd port=1 z0=50 dc=0.595
Specifies that an input port is assumed between terminals rfin and
ground, that it is has a 50 ohm termination, and it has a built-in DC bias of
0.595 V. The output (second) port is:
HSPICE User Guide: RF Analysis
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19
Chapter 3: HSPICE RF Tutorial
Example 2: Using HB Analysis for a Power Amplifier
P2 rfo _vdd port=2 z0=255
This syntax specifies that the output port is between terminals rfo and vdd,
and is being used as a pull up resistor with impedance of 255 ohms.
■
A .PRINT command for plotting the output S parameters in dB and the
noise figure minimum.
To run this netlist, type the following command:
hspicerf gsmlna.sp
This produces two output files, named gsmlna.sc0 and gsmlna.printac0,
containing the S-parameter and noise parameter results, and the requested
PRINT data.
To view the output:
1. Type cscope to invoke CosmosScope.
2. Open gsmlna.sc0 in the File > Open > Plotfiles dialog. (Be sure to change
the “Files of Type…” filter to find the sc0 file.)
3. To open a blank Smith chart, click the Smith chart icon, on the left side of
the upper toolbar.
4. Using the signal manager, select the S(1,1) and S(2,2) signals under the
S-Par heading from the gsmlna.sc0 file. You should see them plotted on the
Smith chart.
5. To open a blank Polar chart, click the Polar chart icon on the left side of the
upper toolbar. Now use the signal manager to select the S(2,1) signal under
the S-Par heading to plot the complex gain of the LNA.
6. Open a blank X-Y plot. Use the signal manager to plot K (the Rollett stability
factor) and Gas (the associated gain) under the Gain-Par heading, and
NFMIN (the noise figure minimum) under the Noise-Par heading.
Example 2: Using HB Analysis for a Power Amplifier
The .HB command computes periodic steady-state solutions of circuits. This
analysis uses the Harmonic Balance (HB) technique for computing such
solutions in the frequency domain. The circuit can be driven by a voltage,
power, or current source, or it may be an autonomous oscillator. The HB
algorithm represents the circuit’s voltage and current waveforms as a Fourier
series, that is, a series of sinusoidal waveforms.
20
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 2: Using HB Analysis for a Power Amplifier
To set up a periodic steady-state analysis, the HSPICE input netlist must
contain:
■
A .HB command to activate the analysis. The .HB command specifies the
base frequency (or frequencies, also called tones) for the analysis, and the
number of harmonics to use for each tone. The .HB command can specify
base tones so that the circuit solution is represented as a multi-dimensional
Fourier series. The number of terms in the series are determined by the
number of harmonics; more harmonics result in higher accuracy, but also
longer simulation times and higher memory usage.
■
One or more signal sources for driving the circuit in HB analysis, if the circuit
is driven. In the case of autonomous oscillator analysis, no signal source is
required. Signal sources are specified using the HB keyword on the voltage
or current source syntax. Power sources are specified by setting the power
switch on voltage/current sources to 1; in this case, the source value is
treated as a power value in Watts instead of a voltage or current.
Optionally, the netlist can also contain a set of control option for optimizing HB
analysis performance.
The following example shows how to set up a Harmonic Balance analysis on an
NMOS Class C Power Amplifier. The example compares transient analysis
results to Harmonic Balance results.
The following netlist performs both a transient and a Harmonic Balance
analysis of the amplifier driven by a sinusoidal input waveform. The accurate
option is set to ensure sufficient number of time points for comparison with HB.
This example is included with the HSPICE RF distribution as pa.sp and is
available in directory $<installdir>/demo/hspicerf/examples.
HSPICE User Guide: RF Analysis
B-2008.09
21
Chapter 3: HSPICE RF Tutorial
Example 2: Using HB Analysis for a Power Amplifier
Figure 2
Power Amplifier
.options POST accurate
.param f0=950e6 PI=3.1415926 Ld=2e-9 Rload=5 Vin=3.0
.param Lin=0.1n Vdd=2 Cd='1.0/(4*PI*PI*f0*f0*Ld)'
M1 drain gt 0 0 CMOSN L=0.35u W=50u AS=100p AD=100p
PS=104u PD=104u M=80
Ls in gt
Lin $ gate tuning
Ld drain vdd Ld $ drain tuning
Cd drain 0
Cd
Cb drain out
INFINITY $ DC block
Rload out
0
Rload
Vdd vdd 0
DC
Vdd
Vrf1 in
0
DC 'Vin/2.0'
+ SIN ('Vin/2' 'Vin/2' 'f0' 0 0 90)
+ HB 'Vin/2' 0.0 1 1
.hb tones=f0 nharms=10
.tran 10p 10n
.probe hb p(Rload)
.probe tran p(Rload)
.include cmos49_model.inc
.end
22
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 2: Using HB Analysis for a Power Amplifier
An HB analysis uses the following:
■
An .HB command:
.hb tones=f0 nharms=10
For a single tone analysis with base frequency 950 mHz and 10 harmonics.
■
The HB source in Vrf1:
HB ‘Vin/2’ 0.0 1 1.
This creates a sinusoidal waveform matching the transient analysis one.
The amplitude is Vin/2=1.5 V, and it applies to the first harmonic of the first
tone, 950 MHz.
■
A .PROBE command for plotting the output power:
.probe hb p(Rload)
To run this netlist, type the following command:
hspicerf pa.sp
This produces two output files named pa.tr0 and pa.hb0, containing the
transient and HB output, respectively. To view and compare the output:
1. Type cscope to invoke CosmosScope.
2. To open both files, use the File > Open > Plotfiles dialog. (Be sure to change
the “Files of Type…” filter to find the hb0 file.)
3. Using the signal manager, view the v(out) signals from the pa.tr0 file.
A time domain waveform opens.
4. View the v(out) signal from the pa.hb0 file.
The histogram shows lines at 950MHz, and multiples thereof, up to 9.5GHz.
5. Right-click on the waveform label for v(out) from the pa.hb0 file, and choose
To Time-Domain.
6. Change the X-End(sec) value to 10n.
7. Click OK to accept the default interval value.
You should now see a new waveform called timedomain(v(out)).
8. Left-click on the timedomain(v(out)) label, hold, and drag the signal to the
plot containing v(out).
HSPICE User Guide: RF Analysis
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23
Chapter 3: HSPICE RF Tutorial
Example 3: Using HB Analysis for an Amplifier
This should overlay the v(out) and timedomain(v(out)) signals on the same
panel. Zoom into the transitions to see the slight differences between the
waveforms.
Example 3: Using HB Analysis for an Amplifier
This example takes the LNA circuit of Example 1 and performs a simulation
using two closely spaced steady-state tones to study the compression and third
order distortion properties of the amplifier. The example file gsmlnaIP3.sp is
located at: /<install_dir>/demo/hspicerf/examples/
See Figure 1 on page 17 for the schematic view.
** NMOS 0.25um Cascode LNA for GSM applications
** Test bench setup for two-tone power sweep in dBm
** to extract IP3.
**
.temp 27
.options post=2
.param
Vdd=2.3
.global gnd
.param Pin:dBm=-30.0
.param Pin=Pin:dBm
.param Pin:W='1.0e-3*pwr(10.0,Pin/10.0)' $ Change to Watts for
sources
**
** Cascode LNA tuned for operation near 1 GHz
**
M1 _n4 _n3 _n5 _n5 CMOSN l=0.25u w=7.5u as=15p ad=15p ps=19u
pd=19u m=80
M2 _n6 _n1 _n4 _n4 CMOSN l=0.25u w=7.5u as=15p ad=15p ps=19u
pd=19u m=80
M3 rfo _n6 gnd gnd CMOSN l=0.25u w=7.5u as=15p ad=15p ps=19u
pd=19u m=40
r1 _vdd _n6 400
l1 _n5 gnd l=0.9nH
l2 rfin _n3 l=13nH $ 0.65n
vvb _n1 gnd
dc=1.19 $ bias for common base device
vinb rfinb gnd dc=0.595
lchk rfin rfinb INFINITY $ Choke
cblk rfin rfind INFINITY $ DC block
vvdd _vdd gnd dc=Vdd
rfb rfo _n6 120
$ feedback
**
** Two-tone input source (DC blocked at this point)
**
Vin rfind gnd dc=0 power=1 z0=50 $ 50 Ohm src
24
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 3: Using HB Analysis for an Amplifier
+ HB Pin:W 0 1 1 $ tone 1
+ HB Pin:W 0 1 2 $ tone 2
Rload rfo _vdd R=255
**
** HB test bench to measure IP3 and IP2
**
.HB tones=900MEG,910MEG nharms=11 11 intmodmax=7
+ SWEEP Pin:dBm -50.0 0.0 2.0
.print HB P(Rload) P(Rload)[1,0] P(Rload)[2,0] P(Rload)[2,-1]
.probe HB P(Rload) P(Rload)[1,0] P(Rload)[2,0] P(Rload)[2,-1]
**
** Approximate parameters for MOSIS 0.25um process (run T17B)
**
.MODEL CMOSN NMOS(
LEVEL
= 49
+VERSION
= 3.1
TNOM
= 27
TOX
= 5.8E-9
+XJ
= 1E-7
NCH
= 2.3549E17
VTH0
= 0.3819327
+K1
= 0.477867
K2
= 2.422759E-3
K3
= 1E-3
+K3B
= 2.1606637
W0
= 1E-7
NLX
= 1.579864E-7
+DVT0W
= 0
DVT1W
= 0
DVT2W
= 0
+DVT0
= 0.5334651
DVT1
= 0.7186877
DVT2
= -0.5
+U0
= 289.1720829
UA
= -1.300598E-9
UB
= 2.308197E-18
+UC
= 2.841618E-11
VSAT
= 1.482651E5
A0
= 1.6856991
+AGS
= 0.2874763
B0
= -1.833193E-8
B1
= -1E-7
+KETA
= -2.395348E-3
A1
= 0
A2
= 0.4177975
+RDSW
= 178.7751373
PRWG
= 0.3774172
PRWB
= -0.2
+WR
= 1
WINT
= 0
LINT
= 1.888394E-8
+XL
= 3E-8
XW
= -4E-8
DWG
= -1.213938E-8
+DWB
= 4.613042E-9
VOFF
= -0.0981658 NFACTOR = 1.2032376
+CIT
= 0
CDSC
= 2.4E-4
CDSCD
= 0
+CDSCB
= 0
ETA0
= 5.128492E-3
ETAB
= 6.18609E-4
+DSUB
= 0.0463218
PCLM
= 1.91946
PDIBLC1
= 1
+PDIBLC2 = 4.422611E-3
PDIBLCB
= -0.1
DROUT
= 0.9817908
+PSCBE1 = 7.982649E10 PSCBE2 = 5.200359E-10 PVAG = 9.314435E-3
+DELTA
= 0.01
RSH
= 3.7
MOBMOD
= 1
+PRT
= 0
UTE
= -1.5
KT1
= -0.11
+KT1L
= 0
KT2
= 0.022
UA1
= 4.31E-9
+UB1
= -7.61E-18
UC1
= -5.6E-11
AT
= 3.3E4
+WL
= 0
WLN
= 1
WW
= 0
+WWN
= 1
WWL
= 0
LL
= 0
+LLN
= 1
LW
= 0
LWN
= 1
+LWL
= 0
CAPMOD
= 2
XPART
= 0.5
+CGDO
= 5.62E-10
CGSO
= 5.62E-10
CGBO
= 1E-12
+CJ
= 1.641005E-3
PB
= 0.99
MJ
= 0.4453094
+CJSW
= 4.179682E-10
PBSW
= 0.99
MJSW
= 0.3413857
+CJSWG
= 3.29E-10
PBSWG
= 0.99
MJSWG
= 0.3413857
+CF
= 0
PVTH0
= -8.385037E-3
PRDSW
= -10
+PK2 = 2.650965E-3 WKETA= 7.293869E-3
LKETA = -6.070221E-3 )
*
.END
HSPICE User Guide: RF Analysis
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25
Chapter 3: HSPICE RF Tutorial
Example 3: Using HB Analysis for an Amplifier
First, notice that we have defined variables that allow power to be swept in dBm
units.
.param Pin:dBm=-30.0
.param Pin=Pin:dBm
.param Pin:W='1.0e-3*pwr(10.0,Pin/10.0)'
References to sources must use SI units in conjunction with the previous
equation to convert from dBm to Watts. The colon (:) is used as a labeling
convenience.
Second, a voltage source element is used as a two-tone power source by
setting the power flag and a source impedance of 50 ohms is specified. The HB
keyword is used to identify the amplitude (interpreted in Watts with the power
flag set), phase, harmonic index, and tone index for each tone.
Vin rfind gnd dc=0 power=1 z0=50 $ 50 Ohm src
+ HB Pin:W 0 1 1
$ tone 1
+ HB Pin:W 0 1 2
$ tone 2
Third, the .HB command designates the frequencies of the two tones and
establishes the power sweep using the dBm power variable. The intmodmax
parameter has been set to 7 to include intermodulation harmonic content up to
7th order effects.
.HB tones=900MEG,910MEG nharms=11 intmodmax=7
+ SWEEP Pin:dBm -50.0 0.0 2.0
Last, the HSPICE RF ability to specify specific harmonic terms is used in
the .PRINT and .PROBE statements to pull out the signals of particular
interest. Notice the three different formats:
.PRINT HB P(Rload)
This reference dumps a complete spectrum in RMS Watts for the power across
resistor Rload.
.PRINT HB P(Rload)[1,0]
This reference selectively dumps the power in resistor Rload at the first
harmonic of the 1st tone.
.PRINT HB P(Rload)[2,-1]
This reference selectively dumps the power in resistor Rload at the 3rd
intermodulation product frequency (890 MHz).
26
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Chapter 3: HSPICE RF Tutorial
Example 3: Using HB Analysis for an Amplifier
To run this simulation, type the following at the command line:
hspicerf gsmlnaIP3.sp
Viewing Results using CosmosScope
For this analysis, the .PRINT statement will generate a
<design_name>.printhb0 file. Assume you want to find out the output power
through the load resistor at the first tone when the input power is 0.1mW
To view the file:
1. Click the 4. Analysis button and then click on the Print tab.
2. Click the 3. Simulation button.
3. Invoke CosmosScope by clicking on the Waveform button.
4. Choose File > Open > Plotfiles to open the <design_name>.hb0 file. (Be
sure to select HSPICERF (*.hb*, *.pn*, *.hr*, *.jt*) from the Files of type
pulldown to find the <design_name>.hb0 file.)
5. Plot the signals Pr(rload) [1,0], Pr(rload) [2,0] and Pr(rload) [2 -1] on top of
each other. The X-axis will be the input power and the Y-axis will be the
output power.
Result: CosmosScope will display the input and output power in dBm. But,
there will be a (W) or (Watt) after the dBm label, this is incorrect.
6. To measure the 1dB compression point of the amplifier, open the
measurement tool by clicking on the caliper icon at the bottom tool bar. Use
the down arrow at the end of the Measurement field and select RF and
P1dB. The PowerOut field should contain the Pr(rload):(1,0) trace.
7. Select a PowerIn value from the list.(The power value should be as large as
possible, but still well within the linear range of the amplifier.) Try -25dbm.
8. Click the Apply button.
Result: CosmosScope will show the linear gain of the amplifier and the
1dBcompression point.
9. The 3rd order intercept point is also measured by using the measurement
tool. Use the down arrow at the end of the Measurement field and select RF
and IP3/SFDR. The PowerOut1 field should contain the Pr(rload):(1,0) trace
and the PowerOut3 field should contain the Pr(rload):(2, -1) trace.
10. Select a PowerIn value from the list. (The power value should be a value that
is as large as possible but, still well within the linear range of the amplifier.)
Try -25dbm.
HSPICE User Guide: RF Analysis
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27
Chapter 3: HSPICE RF Tutorial
Example 3: Using HB Analysis for an Amplifier
11. Click Apply.
Result: CosmosScope will show the 3rd order intercept point of the
amplifier.
Device Model Cards
The following is an NMOS model in cmos49_model.inc file used in the power
amplifier example. It is available in directory $<installdir>/demo/hspicerf/
examples.
.MODEL CMOSN NMOS (LEVEL
= 49
+VERSION = 3.1
TNOM
= 27
TOX
= 7.9E-9
+XJ
= 1.5E-7
NCH
= 1.7E17
VTH0
= 0.5047781
+K1
= 0.5719698
K2
= 0.0197928
K3
= 33.4446099
+K3B
= -3.1667861
W0
= 1E-5
NLX
= 2.455237E-7
+DVT0W = 0
DVT1W
= 0
DVT2W
= 0
+DVT0
= 2.8937881
DVT1
= 0.6610934
DVT2
= -0.0446083
+U0
= 421.8714618
UA
= -1.18967E-10
UB = 1.621684E-18
+UC
= 3.422111E-11
VSAT
= 1.145012E5
A0
= 1.119634
+AGS
= 0.1918651
B0
= 1.800933E-6
B1
= 5E-6
+KETA
= 3.313177E-3
A1
= 0
A2
= 1
+RDSW
= 984.149934
PRWG = -1.133763E-3 PRWB = -7.19717E-3
+WR
= 1
WINT
= 9.590106E-8
LINT
= 1.719803E-8
+XL
= -5E-8
XW
= 0
DWG
= -2.019736E-9
+DWB
= 6.217095E-9
VOFF
= -0.1076921
NFACTOR
= 0
+CIT
= 0
CDSC
= 2.4E-4
CDSCD
= 0
+CDSCB = 0
ETA0
= 0.0147171
ETAB
= -7.256296E-3
+DSUB
= 0.3377074
PCLM = 1.1535622
PDIBLC1 = 2.946624E-4
+PDIBLC2= 4.171891E-3 PDIBLCB = 0.0497942
DROUT = 0.0799917
+PSCBE1 = 3.380501E9
PSCBE2
= 1.69587E-9
PVAG = 0.4105571
+DELTA = 0.01
MOBMOD
= 1
PRT
= 0
+UTE
= -1.5
KT1
= -0.11
KT1L
= 0
+KT2
= 0.022
UA1
= 4.31E-9
UB1
= -7.61E-18
+UC1
= -5.6E-11
AT
= 3.3E4
WL
= 0
+WLN
= 1
WW
= -1.22182E-15
WWN
= 1.1657
+WWL
= 0
LL
= 0
LLN
= 1
+LW
= 0
LWN
= 1
LWL
= 0
+CAPMOD = 2
XPART
= 0.4
CGDO
= 3.73E-10
+CGSO
= 3.73E-10
CGBO
= 1E-11
CJ
= 8.988141E-4
+PB
= 0.8616985
MJ
= 0.3906381
CJSW
= 2.463277E-10
+PBSW
= 0.5072799
MJSW
= 0.1331717
PVTH0
= -0.0143809
+PRDSW = -81.683425
WRDSW = -107.8071189
PK2 = 1.210197E-3
+WKETA = -1.00008E-3
LKETA
= -6.1699E-3
PAGS
= 0.24968)
28
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 4: Using HBOSC Analysis for a Colpitts Oscillator
The following is the BJT model file, bjt.inc used in oscillator example. It is
available in directory $<installdir>/demo/hspicerf/examples.
* RF Wideband NPN Transistor die SPICE MODEL
.MODEL RF_WB_NPN
NPN
+ IS
= 1.32873E-015
BF
= 1.02000E+002
+ NF
= 1.00025E+000
VAF
= 5.19033E+001
+ EG
= 1.11000E+000
XTI
= 3.00000E+000
+ CJE
= 2.03216E-012
VJE
= 6.00000E-001
+ MJE
= 2.90076E-001
TF
= 6.55790E-012
+ XTF
= 3.89752E+001
VTF
= 1.09308E+001
+ ITF
= 5.21078E-001
CJC
= 1.00353E-012
+ VJC
= 3.40808E-001
MJC
= 1.94223E-001
Example 4: Using HBOSC Analysis for a Colpitts Oscillator
This section demonstrates HSPICE RF oscillator analysis using a single
transistor oscillator circuit. Oscillator analysis is an extension of Harmonic
Balance in which the base frequency itself is an unknown to be solved for. In
oscillator analysis, the user supplies a guess at the base frequency, and no
voltage or current source stimulus is needed.
To activate oscillator analysis, include a .HBOSC command with:
■
The TONE parameter set to a guess of the oscillation frequency.
■
The PROBENODE parameter set to identify an oscillating node or pair of
nodes. Always specify a pair of nodes; if only one node oscillates, specify
ground as the second node. To speed up the simulation, also supply a guess
at the magnitude of the oscillating voltage across these nodes.
■
The FSPTS parameter set to a frequency range and number of search
points. When you set FSPTS, HSPICE RF precedes the HBOSC analysis
with a frequency search in the specified range to obtain an optimal initial
guess for the oscillation frequency. This can accelerate the HB oscillator
convergence.
In conjunction with oscillator analysis, HSPICE RF can perform phase noise
analysis. Phase noise analysis measures the effect of transistor noise on the
oscillator frequency. Phase noise analysis is activated using the .PHASENOISE
command; this command sets a set of frequency points for phase noise
analysis. The .PRINT and .PROBE commands can be used to output phase
noise values.
HSPICE User Guide: RF Analysis
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29
Chapter 3: HSPICE RF Tutorial
Example 4: Using HBOSC Analysis for a Colpitts Oscillator
The following netlist, osc.sp, simulates an oscillator, and performs phase noise
analysis. This example is included with the HSPICE RF distribution as pa.sp
and is available in directory $<installdir>/demo/hspicerf/examples.
Figure 3
Colpitts Oscillator
Use the .HBOSC command with the PROBENODE and FSPTS parameters set.
PROBENODE=emitter,0,4.27
Identifies the emitter node as an oscillating node, and provides a guess value
of 4.27 volts for the oscillation amplitude at the emitter node.
FSPTS=40,9e6,1.1e7
Causes an initial frequency search using 40 equally-spaced points between 9
and 11 MHz.
In the .PHASENOISE, .PRINT, and .PROBE commands:
.PHASENOISE V(emitter) dec 10 10k 1meg
30
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 3: HSPICE RF Tutorial
Example 4: Using HBOSC Analysis for a Colpitts Oscillator
Runs phase noise analysis at the specified offset frequencies, measured from
the oscillation carrier frequency. The frequency points specified here are on a
logarithmic scale, 10 points per decade, 10 kHz to 1 MHz.
HSPICE User Guide: RF Analysis
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31
Chapter 3: HSPICE RF Tutorial
Example 4: Using HBOSC Analysis for a Colpitts Oscillator
■
.PROBE PHASENOISE PHNOISE and the similar .PRINT command
instruct HSPICE RF to output phase noise results to the osc.pn0 and
osc.printpn0 files.
**
** Uses emitter resistor limiting to keep output sinusoidal.
** Output can be taken at the emitter (eml node).
**
*--------------------------------------------------------* Options for Oscillator Harmonic Balance Analysis...
*
.OPTIONS post sim_accuracy=100 hbsolver=0
*--------------------------------------------------------* Bias NPN transistor for 5V Vce, 10mA Ic
* Emitter follower Colpitts design
Vcc collector 0
9V
Q1 collector base emitter emitter RF_WB_NPN
Re1
emitter
eml
100
RLoad eml
0
300
Rb1
collector base 4300
Rb2
base
0
5600
*
*--------------------------------------------------------* Capacitive feedback network
Ce
0
eml
100pF
Cfb base eml
100pF
Cbb base bb
470pF
Lb bb
0
6uH
*--------------------------------------------------------* Simulation control for automated oscillator analysis
*
.HBOSC tones=1.0e7 nharms=15
+PROBENODE=emitter,0,4.27
+FSPTS=40,9.e6,1.1e7
*
.PHASENOISE V(emitter) DEC 10 10K 1MEG
+METHOD=0 CARRIERINDEX=1
*
.print hbosc vm(eml) vp(eml) vr(emitter) vi(emitter)
.print hbosc vm(emitter) vp(emitter) P(Rload)
.print phasenoise phnoise
.probe phasenoise phnoise
.probe hbosc v(emitter) v(eml)
.include bjt.inc
.END
32
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
After you run this netlist, examine the osc.printhb0 file.
■
At the top is the oscillator frequency (about 10.14 MHz) and the .PRINT
HBOSC output.
■
The first 2 lines show that the eml node oscillates around 3V with an
amplitude of about 2.85V.
■
The emitter node oscillates around 4V with an amplitude of about 4.27V.
Also examine the osc.printpn0 file, which contains the phase noise results in
text form.
You can view the osc.hb0 and osc.pn0 files in CosmosScope.
1. To start CosmosScope, type cscope.
2. Use the File > Open > Plotfiles dialog to open osc.hb0.
Remember to set the file type filter to HSPICE RF HB (*.hb*).
3. From the signal manager, double click on v(emitter) to see that node’s
spectrum.
4. Right-click on the v(emitter) label in the chart, and choose “To Time Domain”
to create a time domain waveform.
5. To accept the defaults for range and interval, click OK.
You should see an oscillating time domain waveform.
To run a transient simulation for comparison:
1. Use the .TRAN 1n 10u command.
2. Add ic=10n to the Lb inductor.
The resulting waveforms should be the same as those from HB oscillator
analysis.
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
This second oscillator analysis example involves two negative resistance
oscillators coupled at 90 degrees. MOS capacitors are used as varactors. This
VCO topology is common for GPS applications and produces quadrature LO
outputs near 1550 MHz. The purpose of this example is to generate the VCO
tuning curve (output level and frequency as a function of tuning voltage) as well
as its phase noise characteristics as a function of tuning voltage.
HSPICE User Guide: RF Analysis
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33
Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
As in previous examples, the oscillator analysis is activated using the .HBOSC
command:
■
The TONE parameter sets an approximate oscillation frequency (near 1550
MHz).
■
The NHARMS parameter sets the harmonic content to 11th order.
■
The PROBENODE parameters identify the drain pins across the first oscillator
section as the pair of oscillating nodes. This is a differential oscillator, and
the approximate value for this differential amplitude is 6.1 V.
■
The FSPTS parameters set the search frequency range between 1500 and
1600 MHz.
■
The SWEEP parameters set a tuning voltage sweep from 2.0 to 3.2 V.
The following example is based on demonstration netlist gpsvco.sp, which is
available in directory $<installdir>/demo/hspicerf/examples. This netlist
simulates the oscillator schematic Figure 4 and performs phase noise analysis.
Figure 4
34
VCO Schematic
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
**
** NMOS IC Quadrature VCO circuit for GPS local oscillator
**
** Twin differential negative resistance VCOs using NMOS
** transistors for varactors, coupled to produce quadrature
** resonances.
** Design based on 0.35um CMOS process.
**
** References:
** >P. Vancorenland and M.S.J. Steyaert, "A 1.57-GHz fully
**
integrated very low-phase-noise quadrature VCO,"
**
IEEE Trans. Solid-State Circuits, May 2002, pp.653-656.
** >J. van der Tang, P. van de Ven, D. Kasperkovitz, and A.
Roermund,
** "Analysis and design of an optimally coupled 5-GHz quadrature
**
LC oscillator," IEEE Trans. Solid-State Circuits, May 2002,
**
pp.657-661.
** >F. Behbahani, H. Firouzkouhi, R. Chokkalingam, S. Delshadpour,
**
A. Kheirkhani, M. Nariman, M. Conta, and S. Bhatia,
** "A fully integrated low-IF CMOS GPS radio with on-chip analog
** image rejection," IEEE Trans. Solid-State Circuits, Dec. 2002,
**
pp. 1721-1727.
**
** Setup for Harmonic Balance Analysis
**
** Oscillation Frequency: ~ 1575 MHz (GPS L1 frequency)
** Amplitude: ~5 Volts peak-to-peak (zero to 5V)
** Vdd: 2.5 V
**
** HSPICE Simulation Options:
*.option delmax=1n ACCURATE LIST NODE
**
** HSPICE RF Simulation Options :
.option sim_accuracy=10
**
*.option savehb=’a.hbs’ loadhb=’a.hbs’
.option POST
.param Vtune=2.0 $ Failures: vtune=1
.param Cval=0.2p
*--------------------------------Vtune vc gnd DC Vtune
Vdd vdd gnd 2.5
*--------------------------------* First oscillator section
** Low-Q resonator with Vdd at center tap of inductors
R1a IP ri 100k $ These R’s set the Q
R1b ri IN 100k
L1 IP vdd 16.5nH
HSPICE User Guide: RF Analysis
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35
Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
L2 vdd IN 16.5nH
Cc1 IP gnd Cval $ I to Q
Cc2 IN gnd Cval $ -I to Q
** Differential fets
M1 IP IN cs gnd NMOS l=0.35u w=15u
M2 IN IP cs gnd NMOS l=0.35u w=15u
** Bias fet - bias at Vdd -- too high?
Mb cs vdd gnd gnd NMOS l=0.35u w=15u
** fets used as varactors
Mt1 vc IP vc gnd NMOS l=0.35u w=2u M=50
Mt2 vc IN vc gnd NMOS l=0.35u w=2u M=50
*--------------------------------** Second oscillator section
** Low-Q resonator with Vdd at center tap of inductors
R1a_b QP ri_b 100k $ These R’s set the Q
R1b_b ri_b QN 100k
L1_b QP vdd 16.5nH
L2_b vdd QN 16.5nH
Cc1_b QP gnd Cval $ -Q to -I
Cc2_b QN gnd Cval $ -Q to I
** Differential fets
M1_b QP QN cs_b gnd NMOS l=0.35u w=15u
M2_b QN QP cs_b gnd NMOS l=0.35u w=15u
** Bias fet - bias at Vdd -- too high? 2nd in parallel
Mb_b cs_b vdd gnd gnd NMOS l=0.35u w=15u
** fets used as varactors
Mt1_b vc QP vc gnd NMOS l=0.35u w=2u M=50
Mt2_b vc QN vc gnd NMOS l=0.35u w=2u M=50
*
*------------------------------* Differentiators Coupling transistors for quadrature
*
.param Cdiff=0.14p difMsize=50u
vidiff dbias gnd 1.25
viqdiff vdcdif gnd 1.75
Midiff1 dQP dbias gnd gnd NMOS l=0.35u w=difMsize
Midiff2 dQN dbias gnd gnd NMOS l=0.35u w=difMsize
Midiff3 dIN dbias gnd gnd NMOS l=0.35u w=difMsize
Midiff4 dIP dbias gnd gnd NMOS l=0.35u w=difMsize
Cdiff1 dQP QP Cdiff
Cdiff2 dQN QN Cdiff
Cdiff3 dIN IN
Cdiff
Cdiff4 dIP IP
Cdiff
Mc_QP1 IP vdcdif dQP gnd NMOS l=0.35u w=difMsize
Mc_QN2 IN vdcdif dQN gnd NMOS l=0.35u w=difMsize
Mc_QN3 QP vdcdif dIN gnd NMOS l=0.35u w=difMsize
Mc_QP4 QN vdcdif dIP gnd NMOS l=0.35u w=difMsize
*-------------------------------
36
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
* Transient Analysis Test Bench
*
* stimulate oscillation with 2mA pulse
*iosc IP IN PULSE ( 0 2m .01n .01n .01n 10n 1u )
*.probe tran v(IP) v(IN)
*.print tran v(IP) v(IN)
*.TRAN .01n 10n
*-----------------------------* Harmonic Balance Test Bench
*
.sweepblock vtune_sweep
+ 0 5 0.2
+ 2 3 0.1
.HBOSC tones=1550e6 nharms=12
+ PROBENODE=IP,QN,4
+ sweep Vtune sweepblock=vtune_sweep
**
.phasenoise dec 10 100 1e7
.print phasenoise phnz
.probe phasenoise phnz
.print hb v(IP,IN) v(IP,IN)[1] v(QP,QN) v(QP,QN)[1]
.probe hb v(IP,IN) v(IP,IN)[1] v(QP,QN) v(QP,QN)[1]
.probe hb hertz[1][1]
*
* NMOS Device from MOSIS 0.35um Process
*
* BSIM3 VERSION 3.1 PARAMETERS
*
* DATE: Mar 8/00
* LOT: n9co
WAF: 07
* Temperature_parameters=Default
*
.MODEL NMOS NMOS (
LEVEL
= 49
+VERSION = 3.1
TNOM
= 27
TOX
= 7.9E-9
+XJ
= 1.5E-7
NCH
= 1.7E17
VTH0
= 0.5047781
+K1
= 0.5719698
K2
= 0.0197928
K3
= 33.4446099
+K3B
= -3.1667861
W0
= 1E-5
NLX
= 2.455237E-7
+DVT0W
= 0
DVT1W
= 0
DVT2W
= 0
+DVT0
= 2.8937881
DVT1
= 0.6610934
DVT2
= -0.0446083
+U0
= 421.8714618
UA
= -1.18967E-10 UB
= 1.621684E-18
+UC
= 3.422111E-11 VSAT
= 1.145012E5
A0
= 1.119634
+AGS
= 0.1918651
B0
= 1.800933E-6
B1
= 5E-6
+KETA
= 3.313177E-3
A1
= 0
A2
= 1
+RDSW = 984.149934
PRWG = -1.133763E-3 PRWB = -7.19717E-3
+WR
= 1
WINT
= 9.590106E-8
LINT
= 1.719803E-8
+XL
= -5E-8
XW
= 0
DWG
= -2.019736E-9
+DWB
= 6.217095E-9
VOFF
= -0.1076921
NFACTOR = 0
+CIT
= 0
CDSC
= 2.4E-4
CDSCD
= 0
HSPICE User Guide: RF Analysis
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37
Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
+CDSCB = 0
ETA0
= 0.0147171
ETAB
= -7.256296E-3
+DSUB
= 0.3377074
PCLM
= 1.1535622
PDIBLC1 = 2.946624E-4
+PDIBLC2 = 4.171891E-3
PDIBLCB = 0.0497942
DROUT = 0.0799917
+PSCBE1 = 3.380501E9
PSCBE2 = 1.69587E-9
PVAG
= 0.4105571
+DELTA
= 0.01
MOBMOD = 1
PRT
= 0
+UTE
= -1.5
KT1
= -0.11
KT1L
= 0
+KT2
= 0.022
UA1
= 4.31E-9
UB1
= -7.61E-18
+UC1
= -5.6E-11
AT
= 3.3E4
WL
= 0
+WLN
= 1
WW
= -1.22182E-15
WWN
= 1.1657
+WWL
= 0
LL
= 0
LLN
= 1
+LW
= 0
LWN
= 1
LWL
= 0
+CAPMOD = 2
XPART
= 0.4
CGDO
= 3.73E-10
+CGSO
= 3.73E-10
CGBO
= 1E-11
CJ
= 8.988141E-4
+PB
= 0.8616985
MJ
= 0.3906381
CJSW
= 2.463277E-10
+PBSW
= 0.5072799
MJSW
= 0.1331717
PVTH0 = -0.0143809
+PRDSW = -81.683425
WRDSW = -107.8071189 PK2
= 1.210197E-3
+WKETA = -1.00008E-3
LKETA = -6.1699E-3
PAGS
= 0.24968
+AF
= 1.0
KF
= 1.0E-30
)
*
.END
The results of the analysis are displayed in Figure 5 on page 39, Figure 6 on
page 40, and Figure 7 on page 41 using CosmosScope for VCO waveforms,
tuning curves, and phase noise response.
38
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
Figure 5
VCO Waveforms Output in CosmosScope
HSPICE User Guide: RF Analysis
B-2008.09
39
Chapter 3: HSPICE RF Tutorial
Example 5: Using HBOSC Analysis for a CMOS GPS VCO
Figure 6
40
VCO Tuning Curves Output in CosmosScope
HSPICE User Guide: RF Analysis
B-2008.09
Chapter 3: HSPICE RF Tutorial
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
Figure 7
VCO Phase Noise Response in CosmosScope
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
The example in this section shows how to use HSPICE RF to analyze a circuit
driven by multiple input stimuli with different frequencies. Mixer circuits provide
a typical example of this scenario: in this case, there might be two input signals
(LO and RF), which are mixed to produce an IF output signal. In this case,
HSPICE RF offers two options:
■
Multi-tone HB analysis: specify the LO and RF base frequencies as two
separate tones on the .HB command.
■
Periodic AC analysis (HBAC): if one of the inputs is a small-signal, you can
use a faster linear analysis to analyze its effect. For example, if a mixer’s LO
is a large signal, but RF is a small signal, a single-tone HB analysis using
the LO frequency can be combined with HBAC in place of a 2-tone HB
analysis.
HSPICE User Guide: RF Analysis
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41
Chapter 3: HSPICE RF Tutorial
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
To demonstrate both techniques, this example analyzes an ideal mixer built
using behavioral elements. It is based on demonstration netlist mix_tran.sp,
which is available in directory $<installdir>/demo/hspicerf/examples.
* Ideal mixer example: transient analysis
.OPTIONS POST
vlo lo 0 1.0 sin (1.0 0.5 1.0g 0 0 90)
rrf1 rf1 rf 1.0
g1 0 if cur='1.0*v(lo)*v(rf)' $ mixer element
c1 0 if q='1.0e-9*v(lo)*v(rf)' $ mixer element
rout if ifg 1.0
vctrl ifg 0 0.0
h1 out 0 vctrl 1.0 $ convert I to V
rh1 out 0 1.0
vrf rf1 0 sin (0 0.001 0.8GHz 0 0 114)
.tran 10p 10n
.opt sim_accuracy=100
.end
This example uses behavioral controlled current and charge sources to
simulate a mixer. The LO signal is driven by a 0.5 Volt sinusoid at 1 GHz, and
RF is driven by a 10mV signal at 800 MHz. The mixer output is the voltage at
node out, v(out).
Two-tone HB Approach
To analyze this circuit using 2-tone HB, add:
■
HB source for LO: add HB 0.5 0 1 1 to the LO voltage source; this sets
the amplitude to 0.5, no phase shift for the first harmonic of the first tone,
which is 1 GHz.
■
HB source for RF: add HB 0.001 24 1 2 to the RF voltage source; this
sets the amplitude to 0.001, 24 degrees phase shift for the first harmonic of
the second tone (0.8 GHz).
■
An .HB command specifying both tones: .hb tones=1g 0.8g nharms=6 3;
only a small number of harmonics is required to resolve the signals.
The complete mix_hb.sp netlist for 2-tone HB analysis is then:
42
HSPICE User Guide: RF Analysis
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Chapter 3: HSPICE RF Tutorial
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
* Ideal mixer example: 2-tone HB analysis
.OPTIONS POST
vlo lo 0 1.0 sin (1.0 0.5 1.0g 0 0 90) HB 0.5 0 1 1
rrf1 rf1 rf 1.0
g1 0 if cur='1.0*v(lo)*v(rf)' $ mixer element
c1 0 if q='1.0e-9*v(lo)*v(rf)' $ mixer element
rout if ifg 1.0
vctrl ifg 0 0.0
h1 out 0 vctrl 1.0 $ convert I to V
rh1 out 0 1.0
vrf rf1 0 sin (0 0.001 0.8GHz 0 0 114) HB 0.001 24 1 2
.opt sim_accuracy=100
.hb tones=1g 0.8g nharms=6 3
.end
This example is available in directory $<installdir>/demo/hspicerf/examples.
HBAC Approach
To analyze this circuit using HBAC, start with the 2-tone HB analysis setup, and
modify it as follows:
■
Replace the RF HB signal with an HBAC signal: change HB 0.001 24 1
2 to HBAC 0.001 24; this deactivates the source for HB and activates it for
HBAC with the same magnitude and phase.
■
Specify the frequency in the .HBAC command.
■
Change the .HB command to single tone:
.HB tones=1g nharms=6
HBAC takes care of the second tone.
■
Add a .HBAC command
.HBAC lin 1 0.8g 0.8g
This command runs an analysis at a single frequency point, 0.8 GHz. In
general, HBAC analysis can sweep the RF frequency over a range of values.
The following is the complete mix_hbac.sp netlist for HBAC analysis of this
simple mixer. This netlist also contains commands for performing periodic noise
analysis. It is available in directory $<installdir>/demo/hspicerf/examples.
HSPICE User Guide: RF Analysis
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43
Chapter 3: HSPICE RF Tutorial
Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
* Ideal mixer example: HBAC analysis
.OPTIONS POST
vlo lo 0 1.0 sin (1.0 0.5 1.0g 0 0 90)
+ HB 0.5 0 1 1
rrf1 rf1 rf 1.0
g1 0 if cur='1.0*v(lo)*v(rf)' $ mixer element
c1 0 if q='1.0e-9*v(lo)*v(rf)' $ mixer element
rout if ifg 1.0
vctrl ifg 0 0.0
h1 out 0 vctrl 1.0 $ convert I to V
rh1 out 0 1.0
vrf rf1 0 sin (0 0.001 0.8GHz 0 0 114)
+ HBAC 0.001 24
.opt sim_accuracy=100
.hb tones=1g nharms=6
.hbac lin 1 0.8g 0.8g
* Noise analysis
.hbnoise v(out) rrf1 lin 40 0.1g 4g
.print hbnoise onoise nf
.probe hbnoise onoise nf
.end
Comparing Results
After running all three netlists above, you will have generated 3 output files:
■
mix_tran.tr0
■
mix_hb.hb0
■
mix_hbac.hb0
You can compare the results of the 3 analyses in CosmosScope.
1. To run the netlists and start CosmosScope, type:
hspicerf mix_tran.sp
hspicerf mix_hb.sp
hspicerf mix_hbac.sp
cscope &
2. Open the mix_tran.tr0 file: choose File > Open > Plotfiles and select
mix_tran.tr0.
3. To plot v(out), double-click v(out) in the signal manager.
4. Open the mix_hb.hb0 file: choose File > Open > Plotfiles and select
mix_hb.hb0.
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Example 6: Using Multi-Tone HB and HBAC Analyses for a Mixer
You might need to change the “Files of Type…” filter to “HSPICERF HB
(*.hb*)”.
5. Plot v(out) by double clicking v(out) in the signal manager.
A histogram displays.
6. Open the mix_hbac.hb0 file: choose File > Open > Plotfiles and select
mix_hbac.hb0.
You might need to change the “Files of Type…” filter to “HSPICERF HBAC
(*.hb*)”.
7. Plot v(out) by double clicking v(out) in the signal manager.
You should see a histogram similar to the one from mix_hb.hb0.
8. Convert the HB and HBAC histograms to time domain. For each of the two
v(out) histogram signals, right-click on the v(out) label and choose To Time
Domain. Accept the default range and interval settings.
Two new time domain waveforms should appear.
9. Overlay the three time domain plots. Right click on each
“timedomain(v(out))” label, and choose Stack Region/Analog 0.
The bottom panel should now display all three time domain signals. All three
are almost indistinguishable from each other.
You can also use HBAC to perform noise analysis on RF circuits by using
the .HBNOISE command, which is included in the mix_hbac.sp netlist.
■
The .HBNOISE command invokes noise analysis, identifying an output
node where the noise is measured, an input noise source (in this case, rrf1)
which serves as a reference for noise figure computation, and a frequency
sweep for the noise analysis.
■
The .PRINT and .PROBE hbnoise commands instruct HSPICE RF to
save the output noise and noise figure at each frequency in the
mix_hbac.printpn0 and mix_hbac.pn0 output files.
This ideal mixer is noiseless, except for the resistors at the input and output.
The mix_hbac.lis file contains detailed data on the individual noise source
contributions of the resistors. You can view mix_hbac.printpn0 to see the output
noise and noise figure at each frequency. In CosmosScope, you can view
mix_hbac.pn0 to plot the output noise and noise figure data as a function of
frequency.
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Chapter 3: HSPICE RF Tutorial
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Example 7: Using Shooting Newton Analysis on a Driven Phase
Frequency Circuit and a Ring Oscillator
Introduction
While the Harmonic Balance (HB) algorithm represents the circuit's voltage and
current waveforms as a Fourier series (a series of sinusoidal waveforms), the
Shooting Newton (SN) algorithm provides analysis capability for digital logic
circuits and RF components that require steady-state analysis, but operate with
waveforms that tend to be square instead of sinusoidal.
Simple examples of using the Shooting Newton analysis functions are
presented in this section:
■
Driven Phase Frequency Detector Example
■
Ring Oscillator Example
Shooting Newton Analysis Setup
To set up a time-domain, steady-state analysis, the HSPICE input netlist must
contain:
46
■
A .SN command to activate the analysis. The .SN command specifies:
■
The expected period of the steady-state waveforms, which must match the
period of any input waveforms. The period is specified in time domain units
(seconds). Alternatively, you may specify a frequency in Hz.
■
A time resolution, which is analogous to the transient analysis (.TRAN)
command’s TSTEP parameter and will affect the time step size selection. It
also affects the number of frequency terms used in small-signal analyses,
such as periodic AC or noise analysis. The time resolution is typically
specified in seconds but, alternatively, may be specified in the frequency
domain as a number of harmonics.
■
A transient initialization time that is used by HSPICE RF to run a basic
transient simulation of this length before attempting Newton-Raphson
iterations to converge on a steady-state solution. This parameter is optional.
If it is not specified, the specified period is used as the initialization time. The
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Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
initial transient analysis is used for circuit stabilization before the steady
state solution is found. Larger initialization values typically result in
convergence that is more robust.
■
For oscillator circuits, a .SNOSC command is used to activate the analysis.
The .SNOSC command specifies:
■
The approximate frequency of oscillation specified either as a frequency (in
Hertz) or as the time domain period.
■
The number of high frequency harmonics. Alternatively, a time resolution in
seconds can be specified.
■
A transient initialization time that is used by HSPICE RF to run a basic
transient simulation of this length before attempting Newton-Raphson
iterations to converge on a steady state solution. This parameter is optional.
If it is not specified, the period of the specified frequency of oscillation is
used as the initialization time. For oscillators we recommend specifying a
transient initialization time since the default initialization time is usually too
short to effectively stabilize the circuit.
■
A node at which to probe for oscillation conditions.
■
If the tuning curve of a VCO is to be analyzed, the optional parameter
MAXTRINITCYCLES can be specified.
■
One or more signal sources for driving the circuit in SN analysis, if the circuit
is driven. In the case of autonomous oscillator analysis, no signal source is
required. The sources are required to be time domain sources and must
match the period specified in the .SN command.
Driven Phase Frequency Detector Example
This example demonstrates the Shooting Newton-based analysis of a driven
phase-frequency detector. Extracted portions of the input file are presented
below. The complete phasefreqdet.sp input file for this example is located in the
following directory:
$installdir/demo/hspicerf/examples/
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Chapter 3: HSPICE RF Tutorial
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Figure 8
48
Driiven Phase Frequency Detector
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Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
* Phase Frequency Detector Example
*
.global vdd gnd
.options wl post
* DC sources
vsup vdd 0 DC 1.0
* Reference signal (sine wave)
vref xin gnd DC 0 sin( 0.5 0.5 0.5g)
* Input buffers (square up Ref sine wave)
xfin1 xin fin1 inv
xfin2 fin1 FIN inv3
* Compare signal (sine wave)
vcRef cin gnd DC 0 sin( 0.5 0.5 0.5g 0.0 0.0 phase)
$ phase shift
* Input buffers (square up compare sine wave)
xcfin1 cin cfin1 inv
xcfin2 cfin1 cFIN inv3
*
** Phase/frequency detector
xPFD cFIN FIN pdn pu phasedet
** Chargepump
xCP LFIN Ibias pdn pu chargepump
** Bias voltage
vIbias Ibias gnd 0.15
$ Sets charge pump bias!
Vload LFIN 0 0
* Harmonic Balance Test Bench
*
.param phase=0.0
$ phase shift in degrees
.opt snaccuracy=30
.SN tres=10p period=2n SWEEP phase POI 5 0.0 22.5 45.0 67.5 90.0
.SNNOISE V(pu,pdn) Vref
+ DEC 21 100 10MEG
$ offset frequency sweep
+ [0,1]
$ Take low frequency noise
*
.probe sn v(fin) v(cfin) v(pu) v(pdn) v(lfin) i(vibias)
.print snfd i(vload) i(vload) [0]
.probe snfd i(vload) i(vload) [0]
.probe SNNOISE ONOISE
.print SNNOISE ONOISE
.end
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Chapter 3: HSPICE RF Tutorial
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
During the analysis, the phase of the input signal is swept between 0 degrees
and 90 degrees using five equally spaced steps. This enables you to measure
the phase detector gain at the output load. The .SN command syntax specifies
specifies the expected period of the steady-state waveforms (2nS) and the time
resolution (10pS) in the time domain.
A periodic, time-varying AC noise analysis based on the Shooting Newton
algorithm is performed using the .SNNOISE command. The .SNNOISE
analysis requires an output node (v(pu, pdn)) where the noise is to be
measured, an input noise source (Vref) which serves as the reference for the
noise computation and, a frequency sweep for the noise analysis. Optionally,
an index term can be defined. The index term specifies the output frequency
band at which the noise is evaluated. For this case, you will evaluate the low
frequency noise of the phase frequency detector.
The time-domain signals v(cfin), v(fin), v(pu) and v(pdn), and v(lfin) are probed.
The gain of the phase frequency detector can be found by probing the
frequency domain value of v(lfin) at DC (frequency indices 0).
During the simulation, the simulation status is displayed on the screen. In
addition to the screen display, more detailed status, cpu time, and memory
usage information is also written to the phasefreqdet.lis file.
Viewing Results in CosmosScope
You can view the time-domain, phasefreqdet.sn0 file, the frequency domain,
phasefreqdet.snf0 file, and the noise results, phasefreqdet.snpn0 file in
CosmosScope:
1. Enter cscope at the prompt to start CosmosScope.
2. The time domain results are used to show the input and output waveforms
of the phase frequency detector. Use the File > Open > Plotfiles dialog to
open the phasefreqdet.sn0 file. Remember to set the file type filter to
HSPICE RF.
3. From the signal manager, double-click on the input signals, v(cfin) and v(fin),
and the output signals, v(pu) and v(pdn). Figure 9 shows the waveforms for
the selected input and output signals.
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Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Figure 9
Phase Frequency Detector Signals
4. The frequency domain results are used to show the gain of the phase
frequency detector. Use the File > Open > Plotfiles dialog to open the
phasefreqdet.snf0 file.
5. Open a new XY graph by clicking the waveform icon on the left side of the
icon bar.
6. From the signal manager, double-click on the signal i(vload):(0). The signal
is the DC component of the i(vload) signal spectrum. Both the magnitude
and phase of the load current are plotted. To measure the gain of the phase
frequency detector verses phase, only the magnitude is required.
7. The Y-axis should be a real value, not a dB value. To change the Y-axis,
right-click on the Y-axis and select Attributes from the menu. In the Signal
Attributes window, change the view from db(y) to real(y). Figure 10 shows
the gain of the phase frequency detector.
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Chapter 3: HSPICE RF Tutorial
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Figure 10
Phase Frequency Detector Gain
8. Next, plot the output noise of the phase frequency detector. Use the
File > Open > Plotfiles dialog to open the phasefreqdet.snpn0 file.
9. Open a new XY graph by clicking the waveform icon on the left side of the
icon bar.
10. From the signal manager, double-click on the signal inoise(onoise())). The
noise results are shown in Figure 11 on page 53. This displays the noise at
the output, v(pu, vpd) at each phase value swept in the .SN command.
11. Change the X-axis scale to log by right-clicking on the X-axis and selecting
scale -> log. You can change the color of each trace and control the labeling
by right-clicking on the signal name and selecting Member Attributes from
the menu. To assign each trace a different color, click on the rainbowcolored button in the Member Attributes menu. The labels are enabled by
clicking Show All Labels.
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Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Figure 11
Phase Detector Output Noise
Ring Oscillator Example
The Shooting Newton algorithm provides fast and effective analysis for ring
oscillators. The ringoscSN.sp input file for this example is located in the
following directory:
$installdir/demo/hspicerf/examples/
Figure 12
Ring Oscillator
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Chapter 3: HSPICE RF Tutorial
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
.title ringosc
.subckt inv in out vdd
mn1 out in 0 0 nmos l=0.25u w=2u
mp1 out in vdd vdd pmos l=0.25u w=6u
.ends
vdd vdd 0 3
x1 1 2 vdd inv
x2 2 3 vdd inv
x3 3 4 vdd inv
x4 4 5 vdd inv
x5 5 6 vdd inv
x6 6 7 vdd inv
x7 7 1 vdd inv
c1 1 0 0.022p
.ic v(1)=3
.options post
.options snaccuracy=50
.snosc tones=335meg nharms=10 oscnode=1 trinit=10n
.phasenoise v(7) dec 10 100 10meg
.probe
.probe
.print
.probe
sn v(7)
snfd v(7)
phasenoise phnoise v(7)
phasenoise phnoise v(7)
.end
This analysis finds the oscillation frequency of the ring oscillator. Since the
circuit is an oscillator, no input source is required. The oscillator is started by
setting an initial condition at the input of the ring (node 1). In the .SNOSC
command, the node that the analysis will probe for oscillation conditions is
specified, as well as the approximate frequency of oscillation. The number of
harmonics to include in the analysis is specified, as well.
The phase noise characteristics of the oscillator are analyzed by using the
.PHASENOISE command. The .PHASENOISE command requires that an
output node, pair of nodes, or a two-port element and a frequency sweep be
specified. The frequency sweep is used to calculate the phase noise analysis at
the specified offset frequencies, measured from the oscillation carrier
frequency. For this example phase noise analysis, the default Nonlinear
Perturbation (NLP) method is used.
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Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
The signals v(7) will be probed in both the frequency and time domain. The
measure statement is used to measure the fundamental frequency of the
oscillator.
Simulation Status Output
During the simulation, the simulation status is displayed on the screen. In
addition to the screen display, more detailed status, cpu time and memory
usage information is also written to the ringoscSN.lis file.
Viewing Results in CosmosScope
You can view the time-domain, ringoscSN.sn0 file, the frequency domain,
ringoscSN.snf0 file, and the phase noise, ringoscSN.snpn0 file in
CosmosScope.
1. Enter cscope at the prompt to start CosmosScope.
2. Use the File > Open > Plotfiles dialog to open the ringoscSN.sn0 file.
Remember to set the file type filter to HSPICE RF.
3. From the signal manager, double-click on the signal v(7). This is the time
domain trace shown at the top of Figure 13.
4. Use the File > Open > Plotfiles dialog to open the ringoscSN.snf0 file.
5. From the signal manager, double-click on the signals v(7). This is the
frequency domain spectrum shown at the bottom of Figure 13.
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Chapter 3: HSPICE RF Tutorial
Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Figure 13
Ring Oscillator Output
6. To view the phase noise of the ring oscillator, use the File > Open > Plotfiles
dialog to open the ringoscSN.snpn0 file.
7. Open a new XY graph by clicking the waveform icon on the left side of the
icon bar.
8. From the signal manager, double-click on the signal nlp_l(f). The noise
results are shown in Figure 13. Figure 14 on page 57 shows the resulting
phase noise results for the oscillator.
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Example 7: Using Shooting Newton Analysis on a Driven Phase Frequency Circuit and a Ring Oscillator
Figure 14
Ring Oscillator Phase Noise
Other Shooting Newton Analyses
The following Shooting Newton Analyses are also supported by HSPICE RF
but not used in this tutorial.
■
.SNFT is equivalent to the .FFT command in transient (.TRAN) analysis.
.SNFT uses Fourier transform to represent a time domain signal in the
frequency domain. For more information, see Shooting Newton with Fourier
Transform (.SNFT).
■
.SNAC is used to perform a linear analysis of a driven (or nonautonomous)
circuit, where the linear coefficients are modulated by a periodic, steadystate signal. The functionality is similar to the .HBAC command. For more
information, see Shooting Newton AC Analysis (.SNAC).
■
.SNXF is used to calculate transfer functions from an arbitrary number of
small signal sources to a designated output in a circuit under periodic steady
state conditions. For more information, see Shooting Newton Transfer
Function Analysis (.SNXF).
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Chapter 3: HSPICE RF Tutorial
Demonstration Input Files
Demonstration Input Files
The following is a listing of shipped demonstration files for illustrating HSPICE
RF functionality. All of these example files are available at:
$<installdir>/demo/hspicerf/examples
58
File Name
Description
acpr.sp
Envelope simulation example
bjt.inc
Transistor model library used by osc.sp
cmos49_model.inc
Transistor model library used by example circuits
cmos90nmWflicker.lib
Transistor model library used by phasefreqdet.sp
gpsvco.sp
Oscillator and Phase Noise analysis example
gsmlna.sp
LNA Linear analysis example
gsmlnaIP3_A.sp
3rd order intercept point example
mix_hb.sp
Mixer HB analysis example
mix_hbac.sp
MIxer HBAC analysis example
mix_snac.sp
Mixer Shooting Newton AC example
mix_tran.sp
Mixer transient analysis example
osc.sp
Oscillator tuning curve and phase noise analysis example
pa.sp
Power amplifier HB analysis example
pfdcpGain.sp
Shooting Newton analysis example
phasefreqdet.sp
Shooting Newton and noise analysis example
ringoscSN.sp
Shooting Newton and Phase Noise analysis example
tsmc018.m
Transistor model library used by ringoscSN.sp
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4
Input Netlist and Data Entry
Describes the input netlist file and methods of entering data.
For descriptions of individual HSPICE/HSPICE RF commands referenced in
this chapter, see Chapter 2, HSPICE and HSPICE RF Netlist Commands, in
the HSPICE Reference Manual: Commands and Control Options.
These topics are covered in the following sections:
■
Input Netlist File Guidelines
■
Using Subcircuits
■
DDL Library Access
■
Vendor Libraries
■
Subcircuit Library Structure
Input Netlist File Guidelines
HSPICE RF operates on an input netlist file, and store results in either an
output listing file or a graph data file. An input file, with the name <design>.sp,
contains the following:
■
Design netlist (subcircuits, macros, power supplies, and so on).
■
Statement naming the library to use (optional).
■
Specifies the type of analysis to run (optional).
■
Specifies the type of output desired (optional).
An input filename can be up to 1024 characters long. The input netlist file
cannot be in a packed or compressed format.
To generate input netlist and library input files, HSPICE or HSPICE RF uses
either a schematic netlister or a text editor.
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Chapter 4: Input Netlist and Data Entry
Input Netlist File Guidelines
Statements in the input netlist file can be in any order, except that the first line is
a title line, and the last .ALTER submodule must appear at the end of the file
and before the .END statement.
Note:
If you do not place an .END statement at the end of the input netlist file,
HSPICE RF issues an error message.
Netlist input processing is case insensitive, except for file names and their
paths. HSPICE RF does not limit the identifier length, line length, or file size.
Input Line Format
■
The input reader can accept an input token, such as:
•
a statement name.
•
a node name.
•
a parameter name or value.
Any valid string of characters between two token delimiters is a token.
You can not use a character string as a parameter value in HSPICE RF.
See Delimiters on page 66.
■
An input statement, or equation can be up to 1024 characters long.
■
HSPICE RF ignores differences between upper and lower case in input
lines, except in quoted filenames.
■
To continue a statement on the next line, enter a plus (+) sign as the first
non-numeric, non-blank character in the next line.
■
To indicate “to the power of” in your netlist, use two asterisks (**). For
example, 2**5 represents two to the fifth power (25)
■
To continue all HSPICE or HSPICE RF statements, including quoted strings
(such as paths and algebraics), use a single whitespace followed by a
backslash ( \) or a double backslash ( \\) at the end of the line that you
want to continue.
•
■
60
A single backslash preserves white space.
Parameter names must begin with an alphabetic character, but thereafter
can contain numbers and some special characters. See Special Characters,
below.
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Chapter 4: Input Netlist and Data Entry
Input Netlist File Guidelines
•
When you use an asterisk (*) or a question mark (?) with
a .PRINT, .PROBE, .LPRINT (HSPICE RF), or .CHECK (HSPICE RF)
statement, HSPICE or HSPICE RF uses the character as a wildcard.
•
When you use curly braces ( { } ), HSPICE converts them to square
brackets ( [ ] ) automatically.
•
Names are input tokens. Token delimiters must precede and follow
names. See Delimiters below.
•
Names can be up to 1024 characters long and are not case-sensitive.
•
Do not use any of the time keywords as a parameter name or node
name in your netlist.
•
The following symbols are reserved operator keywords:
() = " ‘
Do not use these symbols as part of any parameter or node name that
you define. Using any of these reserved operator keywords as names
causes a syntax error, and HSPICE or HSPICE RF stops immediately.
Special Characters
The following table lists the special characters that can be used as part of node
names, element parameter names, and element instance names. For detailed
discussion, see the appropriate sections in this chapter.
Note:
To avoid unexpected results or error messages, do not use the following
mathematical characters in a parameter name in HSPICE: * - + ^ and /.
Table 4
HSPICE/ HSPICE RF Netlist Special Characters
Special Character
Node Name
Instance Name
(cannot be the
first character;
element key
letter only)
Parameter Name
Delimiters
(cannot be the first
character, element
key letter only)
HSPICE 3 ,
Included only
for HSPICE RF
Included only
Included only
n/a
Included only
Included only
n/a
Note: 3 = character is
legal anywhere in the
string, first or included
~
tilde
!
exclamation
point
3
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Chapter 4: Input Netlist and Data Entry
Input Netlist File Guidelines
Table 4
HSPICE/ HSPICE RF Netlist Special Characters
Special Character
Node Name
Note: 3 = character is
legal anywhere in the
string, first or included
Instance Name
(cannot be the
first character;
element key
letter only)
Parameter Name
Delimiters
(cannot be the first
character, element
key letter only)
@
at sign
3
included only
Included only
n/a
#
pound sign
3
Included only
Included only
n/a
$
dollar sign
Included only
Included only
(avoid if after a
number in node
name)
Included only
In-line comment
character
%
percent
HSPICE 3 ,
Included only
for HSPICE RF
Included only
HSPICE: included n/a
only,
Illegal in HSPICE
RF
^
caret
HSPICE 3,
included only
for HSPICE RF
Included only
HSPICE: included
only (avoid usage),
Illegal in HSPICE
RF
“To the power of”,
i.e., 2^5, two
raised to the fifth
power
&
ampersand
HSPICE 3,
Included only
for HSPICE RF
Included only
Included only
n/a
*
asterisk
HSPICE:
Included only
included only
(avoid using * in
node names),
Illegal for
HSPICE RF
HSPICE: included
only (avoid using
in parameter
names),
Illegal in HSPICE
RF
Comment in both
HSPICE/HSPICE
RF. Wildcard
character. Double
asterisk (**) is “To
the power of”.
()
parentheses
Illegal
Illegal
Illegal
Token delimiter
-
minus
HSPICE:
included only
Included only
Included only
(avoid usage)
n/a
HSPICE RF 3
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Input Netlist File Guidelines
Table 4
HSPICE/ HSPICE RF Netlist Special Characters
Special Character
Node Name
Note: 3 = character is
legal anywhere in the
string, first or included
_
underscore
+
plus sign
3
HSPICE:
included only
Instance Name
(cannot be the
first character;
element key
letter only)
Parameter Name
Delimiters
(cannot be the first
character, element
key letter only)
Included only
Included only
n/a
Included only
HSPICE: included
only (avoid usage);
Illegal in HSPICE
RF
Continues
previous line,
except for quoted
strings
(expressions,
paths, algebraics)
Illegal
optional in
Token delimiter
HSPICE RF 3
=
equals
Illegal
.PARAM
statements
< >
less/more than
HSPICE 3,
included only
for HSPICE RF
Included only
Included only
n/a
?
question mark
HSPICE 3,
Illegal for
HSPICE RF
Included only
Included only
Wildcard in
character in both
HSPICE and
HSPICE RF
Included only
Illegal
n/a
3
/
forward slash
{}
curly braces
HSPICE:
Included only
included only,
converts { } to [ ]
No conversion
for HSPICE RF
Included only
Auto-converts to
square brackets
([])
Single ( { ) or ( } )
can be used in
Variation Blocks
[]
square
brackets
Included only
Included only
n/a
HSPICE User Guide: RF Analysis
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63
Chapter 4: Input Netlist and Data Entry
Input Netlist File Guidelines
Table 4
HSPICE/ HSPICE RF Netlist Special Characters
Special Character
Node Name
Instance Name
(cannot be the
first character;
element key
letter only)
Parameter Name
Delimiters
(cannot be the first
character, element
key letter only)
backslash
(requires a
whitespace
before to use
as a
continuation)
HSPICE:
included only,
Included only
Illegal in HSPICE, Continuation
Included only in
character for
HSPICE RF
quoted strings
(preserves
whitespace)
double
backslash
(requires a
whitespace
before to use
as a
continuation)
HSPICE:
included only,
Illegal
Illegal
Continuation
character for
quoted strings
(preserves
whitespace)
|
pipe
HSPICE 3,
Included only,
HSPICE RF
Included only
Included only
n/a
,
comma
Illegal
Illegal
Illegal
Token delimiter
.
period
Illegal
Included only
Included only
Netlist keyword,
(i.e., .TRAN, .DC,
etc.). Hierarchy
delimiter when
used in node
names
:
colon
Included only
Included only
Included only
Delimiter for
element attributes
semi-colon
Included only
Included only
Included only
n/a
Illegal
Illegal
Expression and
filename delimiter
Note: 3 = character is
legal anywhere in the
string, first or included
\
\\
;
""
64
HSPICE RF 3
HSPICE RF 3
double-quotes Illegal
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Chapter 4: Input Netlist and Data Entry
Input Netlist File Guidelines
Table 4
HSPICE/ HSPICE RF Netlist Special Characters
Special Character
Node Name
Instance Name
(cannot be the
first character;
element key
letter only)
Parameter Name
Delimiters
(cannot be the first
character, element
key letter only)
single quotes
Illegal
Illegal
Illegal
Blank
(whitespace)
Use before \ or
\\ line
continuations
Note: 3 = character is
legal anywhere in the
string, first or included
‘’
Tab
Expression and
filename delimiter
Token delimiter
Tab
Token delimiter
First Character
The first character in every line specifies how HSPICE RF interprets the
remaining line. Table 5 lists and describes the valid characters.
Table 5
First Character Descriptions
Line
If the First Character is...
Indicates
First line of a netlist
Any character
Title or comment line. The first
line of an included file is a
normal line and not a
comment.
Subsequent lines of
netlist, and all lines of
included files
. (period)
Netlist keyword. For example,
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.TRAN 0.5ns 20ns
c, C, d, D, e, E, f, F, g, G, h,
H, i, I, j, J, k, K, l, L, m, M,
q, Q, r, R, s, S, v, V,w,W
Element instantiation
* (asterisk)
Comment line
+ (plus)
Continues previous line
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Input Netlist File Guidelines
Delimiters
■
An input token is any item in the input file that HSPICE RF recognizes. Input
token delimiters are: tab, blank, comma (,), equal sign (=), and parentheses
( ).
■
Single (‘) or double quotes (“) delimit expressions and filenames.
■
Colons (:) delimit element attributes (for example, M1:VGS).
■
Periods (.) indicate hierarchy. For example, X1.X2.n1 is the n1 node on the
X2 subcircuit of the X1 circuit.
Instance Names
The names of element instances begin with the element key letter (see
Table 6), except in subcircuits where instance names begin with X. (Subcircuits
are sometimes called macros or modules.) Instance names can be up to 1024
characters long.
Table 6
66
Element Identifiers
Letter
(First
Char)
Element
Example Line
C
Capacitor
Cbypass 1 0 10pf
D
Diode
D7 3 9 D1
E
Voltage-controlled voltage source
Ea 1 2 3 4 K
F
Current-controlled current source
Fsub n1 n2 vin 2.0
G
Voltage-controlled current source
G12 4 0 3 0 10
H
Current-controlled voltage source
H3 4 5 Vout 2.0
I
Current source
IA 2 6 1e-6
J
JFET or MESFET
J1 7 2 3 GAASFET
K
Linear mutual inductor (general form) K1 L1 L2 1
L
Linear inductor
LX a b 1e-9
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Input Netlist File Guidelines
Table 6
Element Identifiers (Continued)
Letter
(First
Char)
Element
Example Line
M
MOS transistor
M834 1 2 3 4 N1
P
Port
P1 in gnd port=1 z0=50
Q
Bipolar transistor
Q5 3 6 7 8 pnp1
R
Resistor
R10 21 10 1000
S
S-parameter element
S1 nd1 nd2 s_model2
V
Voltage source
V1 8 0 5
W
Transmission Line
W1 in1 0 out1 0 N=1 L=1
T
““
U
““
X
Subcircuit call
X1 2 4 17 31 MULTI WN=100
LN=5
Hierarchy Paths
■
A period (.) indicates path hierarchy.
■
Paths can be up to 1024 characters long.
■
Path numbers compress the hierarchy for post-processing and listing files.
■
The .OPTION PATHNUM controls whether the list files show full path names
or path numbers.
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Input Netlist File Guidelines
Numbers
You can enter numbers as integer, floating point, floating point with an integer
exponent, or integer or floating point with one of the scale factors listed in
Table 7.
Table 7
Scale Factors
Scale Factor
Prefix
Symbol
Multiplying Factor
T
tera
T
1e+12
G
giga
G
1e+9
MEG or X
mega
M
1e+6
K
kilo
k
1e+3
MIL
n/a
none
25.4e-6
M
milli
m
1e-3
U
micro
μ
1e-6
N
nano
n
1e-9
P
pico
p
1e-12
F
femto
f
1e-15
A
atto
a
1e-18
Note:
Scale factor A is not a scale factor in a character string that contains amps.
For example, HSPICE interprets the 20amps string as 20e-18mps
(20-18amps), but it correctly interprets 20amps as 20 amperes of current,
not as 20e-18mps (20-18amps).
68
■
Numbers can use exponential format or engineering key letter format, but
not both (1e-12 or 1p, but not 1e-6u).
■
To designate exponents, use D or E.
■
The .OPTION EXPMAX limits the exponent size.
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Chapter 4: Input Netlist and Data Entry
Input Netlist File Guidelines
■
Trailing alphabetic characters are interpreted as units comments.
■
Units comments are not checked.
■
The .OPTION INGOLD controls the format of numbers in printouts.
■
The .OPTION NUMDGT=x controls the listing printout accuracy.
■
The .OPTION MEASDGT=x controls the measure file printout accuracy.
■
The .OPTION VFLOOR=x specifies the smallest voltage for which HSPICE
or HSPICE RF prints the value. Smaller voltages print as 0.
Parameters and Expressions
■
Parameter names in HSPICE RF use HSPICE name syntax rules, except
that names must begin with an alphabetic character. The other characters
must be either a number, or one of these characters:
! # $ % [ ] _
■
To define parameter hierarchy overrides and defaults, use the .OPTION
PARHIER=global | local statement.
■
If you create multiple definitions for the same parameter or option, HSPICE
RF uses the last parameter definition or .OPTION statement, even if that
definition occurs later in the input than a reference to the parameter or
option. HSPICE RF does not warn you when you redefine a parameter.
■
You must define a parameter before you use that parameter to define
another parameter.
■
When you select design parameter names, be careful to avoid conflicts with
parameterized libraries.
■
To delimit expressions, use single or double quotes.
■
Expressions cannot exceed 1024 characters.
■
For improved readability, use a double slash (\\) at end of a line, to continue
the line.
■
You can nest functions up to three levels.
■
Any function that you define can contain up to two arguments.
■
Use the PAR (expression or parameter) function to evaluate expressions in
output statements.
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Input Netlist File Guidelines
Input Netlist File Structure
An input netlist file should consist of one main program and can contain one or
more optional submodules. HSPICE RF uses a submodule (preceded by
an .ALTER statement) to automatically change an input netlist file; then rerun
the simulation with different options, netlist, analysis statements, and test
vectors.
You can use several high-level call statements (.INCLUDE,and .LIB) to
structure the input netlist file modules. These statements can call netlists,
model parameters, test vectors, analysis, and option macros into a file, from
library files or other files. The input netlist file also can call an external data file,
which contains parameterized data for element sources and models. You must
enclose the names of included or internally-specified files in single or double
quotation when they begin with a number (0-9).
Schematic Netlists
HSPICE RF typically use netlisters to generate circuits from schematics, and
accept either hierarchical or flat netlists.
The process of creating a schematic involves:
■
Symbol creation with a symbol editor.
■
Circuit encapsulation.
■
Property creation.
■
Symbol placement.
■
Symbol property definition.
■
Wire routing and definition
Table 8
Input Netlist File Sections
Sections
Examples
Definition
Title
.TITLE
The first line in the netlist is the title of the
input netlist file (optional in HSPICE RF).
Set-up
.OPTION .IC or .NODESET,
.PARAM, .GLOBAL
Sets conditions for simulation.
Initial values in circuit and subcircuit.
Set parameter values in the netlist.
Set node name globally in netlist.
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Using Subcircuits
Table 8
Input Netlist File Sections (Continued)
Sections
Examples
Definition
Sources
Sources and digital inputs
Sets input stimuli (I or V element).
Netlist
Circuit elements
.SUBKCT, .ENDS, or
.MACRO, .EOM
Circuit for simulation.
Subcircuit definitions.
Analysis
.DC, .TRAN, .AC, and so on.
.DATA,
.TEMP
Statements to perform analyses.
Save and load operating point information.
Create table for data-driven analysis.
Set temperature analysis.
Output
.PRINT, .PROBE,
.MEASURE
Statements to output variables.
Statement to evaluate and report userdefined functions of a circuit.
Library,
Model and
File
Inclusion
.INCLUDE
General include files.
.MODEL
Element model descriptions.
.LIB
Library.
.END
Required statement; end of netlist.
End of
netlist
Using Subcircuits
Reusable cells are the key to saving labor in any CAD system. This also applies
to circuit simulation, in HSPICE or HSPICE RF.
■
To create and simulate a reusable circuit, construct it as a subcircuit.
■
Use parameters to expand the utility of a subcircuit.
Traditional SPICE includes the basic subcircuit, but does not provide a way to
consistently name nodes. However, HSPICE or HSPICE RF provides a simple
method for naming subcircuit nodes and elements: use the subcircuit call name
as a prefix to the node or element name.
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Using Subcircuits
In HSPICE RF, you cannot replicate output commands within subcircuit
(subckt) definitions.
MP
MN
INV
Figure 15
Subcircuit Representation
The following input creates an instance named X1 of the INV cell macro, which
consists of two MOSFETs, named MN and MP:
X1 IN OUT VD_LOCAL VS_LOCAL inv W=20
.MACRO INV IN OUT VDD VSS W=10 L=1 DJUNC=0
MP OUT IN VDD VDD PCH W=W L=L DTEMP=DJUNC
MN OUT IN VSS VSS NCH W=’W/2’ L=L DTEMP=DJUNC
.EOM
Note:
To access the name of the MOSFET, inside of the INV subcircuit that X1
calls, the names are X1.MP and X1.MN. So to print the current that flows
through the MOSFETs, use .PRINT I (X1.MP).
Hierarchical Parameters
You can use two hierarchical parameters, the M (multiply) parameter and the S
(scale) parameter.
M (Multiply) Parameter
The most basic HSPICE RF subcircuit parameter is the M (multiply) parameter.
This keyword is common to all elements, including subcircuits, except for
voltage sources. M multiplies the internal component values, which, in effect,
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Chapter 4: Input Netlist and Data Entry
Using Subcircuits
creates parallel copies of the element. To simulate 32 output buffers switching
simultaneously, you need to place only one subcircuit; for example,
X1 in out buffer M=32
X1 in out inv M=2
M=8
mp out in vdd pch W=10 L=1 M=4
M=6
mn out in vss nch W=5 L=1 M=3
UNEXPANDED
Figure 16
EXPANDED
How Hierarchical Multiply Works
Multiply works hierarchically. For a subcircuit within a subcircuit, HSPICE RF
multiplies the product of both levels. Values of M must be positive. Do not
assign a negative value or zero as the M value.
S (Scale) Parameter
To scale a subcircuit, use the S (local scale) parameter. This parameter
behaves in much the same way as the M parameter in the preceding section.
.OPTION hier_scale=value
.OPTION scale=value
X1 node1 node2 subname S=valueM parameter
The OPTION HIER_SCALE statement defines how HSPICE RF interprets the
S parameter, where value is either:
■
0 (the default), indicating a user-defined parameter, or
■
1, indicating a scale parameter.
The .OPTION SCALE statement defines the original (default) scale of the
subcircuit. The specified S scale is relative to this default scale of the subcircuit.
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Using Subcircuits
The scale in the subname subcircuit is value*scale. Subcircuits can originate
from multiple sources, so scaling is multiplicative (cumulative) throughout your
design hierarchy.
x1 a y inv S=1u
subckt inv in out
x2 a b kk S=1m
.ends
In this example:
■
HSPICE RF scales the X1 subcircuit by the first S scaling value, 1u*scale.
■
Because scaling is cumulative, X2 (a subcircuit of X1) is then scaled, in
effect, by the S scaling values of both X1 and X2: 1m*1u*scale.
Using Hierarchical Parameters to Simplify Simulation
You can use the hierarchical parameter to simplify simulations. An example is
shown in the following listing and Figure 17 on page 74.
X1 D Q Qbar CL CLBAR dlatch flip=0
.macro dlatch
+ D Q Qbar CL CLBAR flip=vcc
.nodeset v(din)=flip
xinv1 din qbar inv
xinv2 Qbar Q inv
m1 q CLBAR din nch w=5 l=1
m2 D CL din nch w=5 l=1
.eom
Q
clbar
cl
Q
D
din
.Nodeset
Figure 17
D Latch with Nodeset
HSPICE does not limit the size or complexity of subcircuits; they can contain
subcircuit references, and any model or element statement. However, in
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DDL Library Access
HSPICE RF, you cannot replicate output commands within subcircuit
definitions. To specify subcircuit nodes in .PRINT statements, specify the full
subcircuit path and node name.
DDL Library Access
To include a DDL library component in a data file, use the X subcircuit call
statement with the DDL element call. The DDL element statement includes the
model name, which the actual DDL library file uses.
For example, the following element statement creates an instance of the
1N4004 diode model:
X1 2 1 D1N4004
Where D1N4004 is the model name.
See Chapter 6, Testbench Elements and the HSPICE Elements and Device
Models Manual for descriptions of element statements.
Optional parameter fields in the element statement can override the internal
specification of the model. For example, for op-amp devices, you can override
the offset voltage, and the gain and offset current. Because the DDL library
devices are based on HSPICE circuit-level models, simulation automatically
compensates for the effects of supply voltage, loading, and temperature.
HSPICE or HSPICE RF accesses DDL models in several ways:
■
The installation script creates an hspice.ini initialization file.
■
HSPICE or HSPICE RF writes the search path for the DDL and vendor
libraries into a .OPTION SEARCH=‘<lib_path>’ statement.
This provides immediate access to all libraries for all users. It also
automatically includes the models in the input netlist. If the input netlist
references a model or subcircuit, HSPICE or HSPICE RF searches the
directory to which the DDLPATH environment variable points for a file with
the same name as the reference name. This file is an include file so its
filename suffix is .inc. HSPICE installation sets the DDLPATH variable in the
meta.cfg configuration file.
■
Set .OPTION SEARCH=‘<lib_path>’ in the input netlist.
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Chapter 4: Input Netlist and Data Entry
Vendor Libraries
Use this method to list the personal libraries to search. HSPICE first
searches the default libraries referenced in the hspice.ini file, then searches
libraries in the order listed in the input file.
■
Directly include a specific model, using the .INCLUDE statement. For
example, to use a model named T2N2211, store the model in a file named
T2N2211.inc, and put the following statement in the input file:
.INCLUDE <path>/T2N2211.inc
This method requires you to store each model in its own .inc file, so it is not
generally useful. However, you can use it to debug new models, when you
test only a small number of models.
Vendor Libraries
The vendor library is the interface between commercial parts and circuit or
system simulation.
■
ASIC vendors provide comprehensive cells, corresponding to inverters,
gates, latches, and output buffers.
■
Memory and microprocessor vendors supply input and output buffers.
■
Interface vendors supply complete cells for simple functions and output
buffers, to use in generic family output.
■
Analog vendors supply behavioral models.
To avoid name and parameter conflicts, models in vendor cell libraries should
be within the subcircuit definitions.
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Subcircuit Library Structure
x1 in out vdd vss buffer_f
.OPTION search=‘/usr/lib/vendor’
/usr/lib/vendor/buffer_f.inc
/usr/lib/vendor/skew.dat
.macro buffer_f in out vdd vss
.lib ‘/usr/lib/vendor/skew.dat’ ff
.lib ff $ fast model
.param vendor_xl=-.1u
.inc ‘/usr/lib/vendor/model.dat’
.endl ff
.inc ‘/usr/lib/vendor/buffer.inc’
.eom
/usr/lib/vendor/buffer.inc
/usr/lib/vendor/model.dat
.model nch nmos level=28
+ xl=vendor_xl ...
Figure 18
.macro buffer in out vdd vss
m1 out in vdd vdd nch w=10 l=1
...
Vendor Library Usage
Subcircuit Library Structure
Your library structure must adhere to the .INCLUDE statement specification in
the implicit subcircuit. You can use this statement to specify the directory that
contains the <subname>.inc subcircuit file, and then reference the <subname>
in each subcircuit call.
The component naming conventions for each subcircuit is:
<subname>.inc
Store the subcircuit in a directory that is accessible by a.OPTION
SEARCH=‘<lib_path>’ statement.
Create subcircuit libraries in a hierarchy. Typically, the top-level subcircuit fully
describes the input/output buffer; any hierarchy is buried inside. The buried
hierarchy can include model statements, lower-level components, and
parameter assignments. Your library cannot use .LIB or .INCLUDE
statements anywhere in the hierarchy.
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Subcircuit Library Structure
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5
5
Parameters and Functions
Describes how to use parameters within HSPICE RF netlists.
Parameters are similar to the variables used in most programming languages.
Parameters hold a value that you assign when you create your circuit design or
that the simulation calculates based on circuit solution values. Parameters can
store static values for a variety of quantities (resistance, source voltage, rise
time, and so on). You can also use them in sweep or statistical analysis.
For descriptions of RF commands referenced in this chapter, see Chapter 2,
HSPICE and HSPICE RF Netlist Commands, in the HSPICE Reference
Manual: Commands and Control Options.
These topics are covered in the following sections:
■
Using Parameters in Simulation (.PARAM)
■
Using Algebraic Expressions
■
Built-In Functions and Variables
■
Parameter Scoping and Passing
Using Parameters in Simulation (.PARAM)
Defining Parameters
Parameters in HSPICE are names that you associate with numeric values.
(See Assigning Parameters on page 81.) You can use any of the methods
described in Table 9 to define parameters.
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Chapter 5: Parameters and Functions
Using Parameters in Simulation (.PARAM)
Table 9
.PARAM Statement Syntax
Parameter
Description
Simple
assignment
.PARAM <SimpleParam>=1e-12
Algebraic
definition
.PARAM <AlgebraicParam>=‘SimpleParam*8.2’
SimpleParam excludes the output variable.
You can also use algebraic parameters in .PRINT and .PROBE statements.
For example:
.PRINT AlgebraicParam=par(’algebraic expression’)
You can use the same syntax for .PROBE, statements. See Using Algebraic
Expressions on page 83.
User-defined
function
.PARAM <MyFunc( x, y )>=‘Sqrt((x*x)+(y*y))’
Character string
definition
.PARAM <paramname>=str(‘string’)
Subcircuit default
.SUBCKT <SubName> <ParamDefName>=<Value> str(‘string’)
.MACRO <SubName> <ParamDefName>=<Value> str(‘string’)
Predefined
analysis function
.PARAM <mcVar>=Agauss(1.0,0.1)
.MEASURE
statement
.MEASURE <DC | AC | TRAN> result TRIG ...
+ TARG ... <GOAL=val> <MINVAL=val>
+ <WEIGHT=val> <MeasType> <MeasParam>
.PRINT | .PROBE .PRINT | .PROBE
+ outParam=Par_Expression
A parameter definition in HSPICE always uses the last value found in the input
netlist (subject to local versus global parameter rules). The definitions below
assign a value of 3 to the DupParam parameter.
.PARAM DupParam=1
...
.PARAM DupParam=3
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Using Parameters in Simulation (.PARAM)
HSPICE assigns 3 as the value for all instances of DupParam, including
instances that are earlier in the input than the .PARAM DupParam=3
statement.
All parameter values in HSPICE are IEEE double floating point numbers. The
parameter resolution order is:
1. Resolve all literal assignments.
2. Resolve all expressions.
3. Resolve all function calls.
Table 10 shows the parameter passing order.
Table 10
Parameter Passing Order
.OPTION PARHIER=GLOBAL
.OPTION PARHIER=LOCAL
Analysis sweep parameters
Analysis sweep parameters
.PARAM statement (library)
.SUBCKT call (instance)
.SUBCKT call (instance)
.SUBCKT definition (symbol)
.SUBCKT definition (symbol)
.PARAM statement (library)
Assigning Parameters
You can assign the following types of values to parameters:
■
Constant real number
■
Algebraic expression of real values
■
Predefined function
■
Function that you define
■
Circuit value
■
Model value
To invoke the algebraic processor, enclose a complex expression in single
quotes. A simple expression consists of one parameter name.
The parameter keeps the assigned value, unless:
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Chapter 5: Parameters and Functions
Using Parameters in Simulation (.PARAM)
■
A later definition changes its value, or
■
An algebraic expression assigns a new value during simulation.
HSPICE does not warn you, if it reassigns a parameter.
Inline Parameter Assignments
To define circuit values, using a direct algebraic evaluation:
r1 n1 0 R=’1k/sqrt(HERTZ)’ $ Resistance for frequency
Parameters in Output
To use an algebraic expression as an output variable in a .PRINT, .PROBE
or .MEASURE statement, use the PAR keyword.
Example
.PRINT DC v(3) gain=PAR(‘v(3)/v(2)’) PAR(‘v(4)/v(2)’)
User-Defined Function Parameters
You can define a function that is similar to the parameter assignment, but you
cannot nest the functions more than two deep.
■
An expression can contain parameters that you did not define.
■
A function must have at least one argument, and can have up to 20 (and in
many cases, more than 20) arguments.
■
You can redefine functions.
The format of a function is:
funcname1(arg1[,arg2...])=expression1
+ [funcname2(arg1[,arg2...])=expression2] off
82
Parameter
Description
funcname
Specifies the function name. This parameter must be distinct from
array names and built-in functions. In subsequently defined functions,
all embedded functions must be previously defined.
arg1, arg2
Specifies variables used in the expression.
off
Voids all user-defined functions.
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Using Algebraic Expressions
Example
.PARAM f(a,b)=POW(a,2)+a*b g(d)=SQRT(d)
+ h(e)=e*f(1,2)-g(3)
Predefined Analysis Function
HSPICE includes specialized analysis types, such as Optimization and Monte
Carlo, that require a way to control the analysis.
Measurement Parameters
.MEASURE statements produce a measurement parameter. The rules for
measurement parameters are the same as for standard parameters, except
that measurement parameters are defined in a .MEASURE statement, not in
a .PARAM statement..
.PRINT and .PROBE Parameters
.PRINT,and.PROBE statements in HSPICE produce a print parameter. The
rules for print parameters are the same as the rules for standard parameters,
except that you define the parameter directly in a.PRINT or.PROBE
statement, not in a .PARAM statement
Using Algebraic Expressions
Note:
Synopsys HSPICE uses double-precision numbers (15 digits) for
expressions, user-defined parameters, and sweep variables. For better
precision, use parameters (instead of constants) in algebraic expressions,
because constants are only single-precision numbers (7 digits).
In HSPICE, an algebraic expression, with quoted strings, can replace any
parameter in the netlist.
In HSPICE, you can then use these expressions as output variables
in .PRINT, statements. Algebraic expressions can expand your options in an
input netlist file.
Some uses of algebraic expressions are:
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Chapter 5: Parameters and Functions
Built-In Functions and Variables
■
Parameters:
.PARAM x=’y+3’
■
Functions:
.PARAM rho(leff,weff)=’2+*leff*weff-2u’
■
Algebra in elements:
R1 1 0 r=’ABS(v(1)/i(m1))+10’
■
Algebra in .MEASURE statements:
.MEAS vmax MAX V(1)
.MEAS imax MAX I(q2)
.MEAS ivmax PARAM=’vmax*imax’
■
Algebra in output statements:
.PRINT conductance=PAR(‘i(m1)/v(22)’)
The basic syntax for using algebraic expressions for output is:
PAR(‘algebraic expression’)
In addition to using quotations, you must define the expression inside the
PAR( ) statement for output.The continuation character for quoted parameter
strings, in HSPICE, is a double backslash (\\). (Outside of quoted strings, the
single backslash (\) is the continuation character.)
Built-In Functions and Variables
In addition to simple arithmetic operations (+, -, *, /), you can use the built-in
functions listed in Table 11 and the variables listed in Table 10 on page 81 in
HSPICE expressions.
Table 11
84
Synopsys HSPICE Built-in Functions
HSPICE Form
Function
Class
Description
sin(x)
sine
trig
Returns the sine of x (radians)
cos(x)
cosine
trig
Returns the cosine of x (radians)
tan(x)
tangent
trig
Returns the tangent of x (radians)
asin(x)
arc sine
trig
Returns the inverse sine of x (radians)
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Built-In Functions and Variables
Table 11
Synopsys HSPICE Built-in Functions (Continued)
HSPICE Form
Function
Class
Description
acos(x)
arc cosine
trig
Returns the inverse cosine of x (radians)
atan(x)
arc tangent
trig
Returns the inverse tangent of x (radians)
sinh(x)
hyperbolic
sine
trig
Returns the hyperbolic sine of x (radians)
cosh(x)
hyperbolic
cosine
trig
Returns the hyperbolic cosine of x (radians)
tanh(x)
hyperbolic
tangent
trig
Returns the hyperbolic tangent of x (radians)
abs(x)
absolute
value
math
Returns the absolute value of x: |x|
sqrt(x)
square root
math
Returns the square root of the absolute value
of x: sqrt(-x)=-sqrt(|x|)
pow(x,y)
absolute
power
math
Returns the value of x raised to the integer
part of y: x(integer part of y)
pwr(x,y)
signed
power
math
Returns the absolute value of x, raised to the
y power, with the sign of x: (sign of x)|x|y
x**y
power
If x<0, returns the value of x raised to the
integer part of y.
If x=0, returns 0.
If x>0, returns the value of x raised to the y
power.
log(x)
natural
logarithm
math
Returns the natural logarithm of the absolute
value of x, with the sign of x: (sign of x)log(|x|)
log10(x)
base 10
logarithm
math
Returns the base 10 logarithm of the absolute
value of x, with the sign of x: (sign of
x)log10(|x|)
exp(x)
exponential
math
Returns e, raised to the power x: ex
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Built-In Functions and Variables
Table 11
Synopsys HSPICE Built-in Functions (Continued)
HSPICE Form
Function
Class
Description
db(x)
decibels
math
Returns the base 10 logarithm of the absolute
value of x, multiplied by 20, with the sign of x:
(sign of x)20log10(|x|)
int(x)
integer
math
Returns the integer portion of x. The fractional
portion of the number is lost.
nint(x)
integer
math
Rounds x up or down, to the nearest integer.
sgn(x)
return sign
math
Returns -1 if x is less than 0.
Returns 0 if x is equal to 0.
Returns 1 if x is greater than 0
86
sign(x,y)
transfer sign math
Returns the absolute value of x, with the sign
of y: (sign of y)|x|
def(x)
parameter
defined
control
Returns 1 if parameter x is defined.
min(x,y)
smaller of
two args
control
Returns the numeric minimum of x and y
max(x,y)
larger of two control
args
Returns the numeric maximum of x and y
val(element)
get value
various
Returns a parameter value for a specified
element. For example, val(r1) returns the
resistance value of the r1 resistor.
val(element.
parameter)
get value
various
Returns a value for a specified parameter of a
specified element. For example,
val(rload.temp) returns the value of the temp
(temperature) parameter for the rload
element.
val(model_type:
model_name.
model_param)
get value
various
Returns a value for a specified parameter of a
specified model of a specific type. For
example, val(nmos:mos1.rs) returns the value
of the rs parameter for the mos1 model, which
is an nmos model type.
Returns 0 if parameter x is not defined.
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Built-In Functions and Variables
Table 11
Synopsys HSPICE Built-in Functions (Continued)
HSPICE Form
Function
Class
Description
lv(<Element>)
or
lx(<Element>)
element
templates
various
Returns various element values during
simulation.
v(<Node>),
i(<Element>)...
circuit
output
variables
various
Returns various circuit values during
simulation.
[cond] ?x : y
ternary
operator
Returns x if cond is not zero. Otherwise,
returns y.
.param z=’condition ? x:y’
<
<=
>
>=
==
relational
operator
(less than)
Returns 1 if the left operand is less than the
right operand. Otherwise, returns 0.
relational
operator
(less than or
equal)
Returns 1 if the left operand is less than or
equal to the right operand. Otherwise, returns
0.
relational
operator
(greater
than)
Returns 1 if the left operand is greater than the
right operand. Otherwise, returns 0.
relational
operator
(greater
than or
equal)
Returns 1 if the left operand is greater than or
equal to the right operand. Otherwise, returns
0.
equality
Returns 1 if the operands are equal.
Otherwise, returns 0.
.para x=y<z (y less than z)
.para x=y<=z (y less than or equal to z)
.para x=y>z (y greater than z)
.para x=y>=z (y greater than or equal to z)
.para x=y==z (y equal to z)
!=
inequality
Returns 1 if the operands are not equal.
Otherwise, returns 0.
.para x=y!=z (y not equal to z)
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Table 11
Synopsys HSPICE Built-in Functions (Continued)
HSPICE Form
Function
Class
Description
&&
Logical
AND
Returns 1 if neither operand is zero.
Otherwise, returns 0. .para x=y&&z (y AND z)
||
Logical OR
Returns 1 if either or both operands are not
zero. Returns 0 only if both operands are zero.
.para x=y||z (y OR z)
Example
.parameters p1=4 p2=5 p3=6
r1 1 0 value='p1 ? p2+1 : p3'
HSPICE reserves the variable names listed in Table 12 on page 88 for use in
elements, such as E, G, R, C, and L. You can use them in expressions, but you
cannot redefine them; for example, this statement would be illegal:
.param temper=100
Table 12
Synopsys HSPICE Special Variables
HSPICE Form
Function
Class
Description
time
current simulation
time
control
Uses parameters to define the current
simulation time, during transient analysis.
temper
current circuit
temperature
control
Uses parameters to define the current
simulation temperature, during transient/
temperature analysis.
hertz
current simulation
frequency
control
Uses parameters to define the frequency,
during AC analysis.
Parameter Scoping and Passing
If you use parameters to define values in sub-circuits, you need to create fewer
similar cells, to provide enough functionality in your library. You can pass circuit
parameters into hierarchical designs, and assign different values to the same
parameter within individual cells, when you run simulation.
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For example, if you use parameters to set the initial state of a latch in its
subcircuit definition, then you can override this initial default in the instance call.
You need to create only one cell, to handle both initial state versions of the
latch.
You can also use parameters to define the cell layout. For example, you can
use parameters in a MOS inverter, to simulate a range of inverter sizes, with
only one cell definition. Local instances of the cell can assign different values to
the size parameter for the inverter.
In HSPICE, you can also perform Monte Carlo analysis or optimization on a cell
that uses parameters.
How you handle hierarchical parameters depends on how you construct and
analyze your cells. You can construct a design in which information flows from
the top of the design, down into the lowest hierarchical levels.
■
To centralize the control at the top of the design hierarchy, set global
parameters.
■
To construct a library of small cells that are individually controlled from
within, set local parameters and build up to the block level.
This section describes the scope of parameter names, and how HSPICE
resolves naming conflicts between levels of hierarchy.
Library Integrity
Integrity is a fundamental requirement for any symbol library. Library integrity
can be as simple as a consistent, intuitive name scheme, or as complex as
libraries with built-in range checking.
Library integrity might be poor if you use libraries from different vendors in a
circuit design. Because names of circuit parameters are not standardized
between vendors, two components can include the same parameter name for
different functions. For example, one vendor might build a library that uses the
name Tau as a parameter to control one or more subcircuits in their library.
Another vendor might use Tau to control a different aspect of their library. If you
set a global parameter named Tau to control one library, you also modify the
behavior of the second library, which might not be the intent.
If the scope of a higher-level parameter is global to all subcircuits at lower levels
of the design hierarchy, higher-level definitions override lower-level parameter
values with the same names. The scope of a lower-level parameter is local to
the subcircuit where you define the parameter (but global to all subcircuits that
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Parameter Scoping and Passing
are even lower in the design hierarchy). Local scoping rules in HSPICE prevent
higher-level parameters from overriding lower-level parameters of the same
name, when that is not desired.
Reusing Cells
Parameter name problems also occur if different groups collaborate on a
design. Global parameters prevail over local parameters, so all circuit
designers must know the names of all parameters, even those used in sections
of the design for which they are not responsible. This can lead to a large
investment in standard libraries. To avoid this situation, use local parameter
scoping, to encapsulate all information about a section of a design, within that
section.
Creating Parameters in a Library
To ensure that the input netlist includes critical, user-supplied parameters when
you run simulation, you can use “illegal defaults”—that is, defaults that cause
the simulator to abort if you do not supply overrides for the defaults.
If a library cell includes illegal defaults, you must provide a value for each
instance of those cells. If you do not, the simulation aborts.
For example, you might define a default MOSFET width of 0.0. HSPICE aborts,
because MOSFET models require this parameter.
Example 1
* Subcircuit default definition
.SUBCKT Inv A Y Wid=0 $ Inherit illegal values by default
mp1 <NodeList> <Model> L=1u W=’Wid*2’
mn1 <NodeList> <Model> L=1u W=Wid
.ENDS
* Invoke symbols in a design
x1 A Y1 Inv
$ Bad! No widths specified
x2 A Y2 Inv Wid=1u $ Overrides illegal value for Width
This simulation aborts on the x1 subcircuit instance, because you never set the
required Wid parameter on the subcircuit instance line. The x2 subcircuit
simulates correctly. Additionally, the instances of the Inv cell are subject to
accidental interference, because the Wid global parameter is exposed outside
the domain of the library. Anyone can specify an alternative value for the
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parameter, in another section of the library or the circuit design. This might
prevent the simulation from catching the condition on x1.
Example 2
In this example, the name of a global parameter conflicts with the internal
library parameter named Wid. Another user might specify such a global
parameter, in a different library. In this example, the user of the library has
specified a different meaning for the Wid parameter, to define an independent
source.
.Param Wid=5u
$ Default Pulse Width for source
v1 Pulsed 0 Pulse ( 0v 5v 0u 0.1u 0.1u Wid 10u )
...
* Subcircuit default definition
.SUBCKT Inv A Y Wid=0
$ Inherit illegals by default
mp1 <NodeList> <Model> L=1u W=’Wid*2’
mn1 <NodeList> <Model> L=1u W=Wid
.Ends
* Invoke symbols in a design
x1 A Y1 Inv
$ Incorrect width!
x2 A Y2 Inv Wid=1u
$ Incorrect! Both x1 and x2
$ simulate with mp1=10u and
$ mn1=5u instead of 2u and 1u.
Under global parameter scoping rules, simulation succeeds, but incorrectly.
HSPICE does not warn you that the x1 inverter has no assigned width,
because the global parameter definition for Wid overrides the subcircuit default.
Note:
Similarly, sweeping with different values of Wid dynamically changes both
the Wid library internal parameter value, and the pulse width value to the
Wid value of the current sweep.
In global scoping, the highest-level name prevails, when resolving name
conflicts. Local scoping uses the lowest-level name.
When you use the parameter inheritance method, you can specify to use local
scoping rules.
When you use local scoping rules, the Example 2 netlist correctly aborts in x1
for W=0 (default Wid=0, in the .SUBCKT definition, has higher precedence,
than the .PARAM statement). This results in the correct device sizes for x2. This
change can affect your simulation results, if you intentionally or accidentally
create a circuit such as the second one shown above.
As an alternative to width testing in the Example 2 netlist, you can
use .OPTION DEFW to achieve a limited version of library integrity. This option
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sets the default width for all MOS devices during a simulation. Part of the
definition is still in the top-level circuit, so this method can still make unwanted
changes to library values, without notification from the HSPICE simulator.
Table 13 compares the three primary methods for configuring libraries, to
achieve required parameter checking for default MOS transistor widths.
Table 13
Method
Methods for Configuring Libraries
Parameter
Location
Pros
Cons
Local
On a .SUBCKT
definition line
Protects library from global
circuit parameter definitions,
unless you override it. Single
location for default values.
Global
At the global
level and
on .SUBCKT
definition lines
Works with all HSPICE
versions.
An indiscreet user, another
vendor assignment, or the
intervening hierarchy can
change the library. Cannot
override a global value at a
lower level.
Special
.OPTION DEFW
statement
Simple to do.
Third-party libraries, or other
sections of the design, might
depend on .OPTION DEFW.
Parameter Defaults and Inheritance
Use the .OPTION PARHIER parameter to specify scoping rules.
Syntax:
.OPTION PARHIER=< GLOBAL | LOCAL >
The default setting is GLOBAL.
Example
This example explicitly shows the difference between local and global scoping
for using parameters in subcircuits.
The input netlist includes the following:
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.OPTION parhier=<global | local>
.PARAM DefPwid=1u
.SUBCKT Inv a y DefPwid=2u DefNwid=1u
Mp1 <MosPinList> pMosMod L=1.2u W=DefPwid
Mn1 <MosPinList> nMosMod L=1.2u W=DefNwid
.ENDS
Set the .OPTION PARHIER=parameter scoping option to GLOBAL. The
netlist also includes the following input statements:
xInv0 a y0 Inv
$ override DefPwid default,
$ xInv0.Mp1 width=1u
xInv1 a y1 Inv DefPwid=5u $ override DefPwid=5u,
$ xInv1.Mp1 width=1u
.measure tran Wid0 param=’lv2(xInv0.Mp1)’ $ lv2 is the
$ template for
.measure tran Wid1 param=’lv2(xInv1.Mp1)’ $ the channel
$ width
$ ‘lv2(xInv1.Mp1)’
.ENDS
Simulating this netlist produces the following results in the listing file:
wid0=1.0000E-06
wid1=1.0000E-06
If you change the .OPTION PARHIER=parameter scoping option
to LOCAL:
xInv0 a y0 Inv
$ not override .param
$ DefPwid=2u,
$ xInv0.Mp1 width=2u
xInv1 a y1 Inv DefPwid=5u
$ override .param
$ DefPwid=2u,
$ xInv1.Mp1 width=5u:
.measure tran Wid0 param=’lv2(xInv0.Mp1)’$ override the
.measure tran Wid1 param=’lv2(xInv1.Mp1)’$ global .PARAM
...
Simulation produces the following results in the listing file:
wid0=2.0000E-06
wid1=5.0000E-06
Parameter Passing
Figure 19 on page 94 shows a flat representation of a hierarchical circuit, which
contains three resistors.
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Parameter Scoping and Passing
Each of the three resistors obtains its simulation time resistance from the Val
parameter. The netlist defines the Val parameter in four places, with three
different values.
+
Sub1
Sub2
Sub3
r1
r2
r3
1V
-
Figure 19
TEST OF PARHIER
.OPTION list node post=2
+ ingold=2
+ parhier=<Local|Global>
.PARAM Val=1
x1 n0 0 Sub1
.SubCkt Sub1 n1 n2 Val=1
r1 n1 n2 Val
x2 n1 n2 Sub2
.Ends Sub1
.SubCkt Sub2 n1 n2 Val=2
r2 n1 n2 Val
x3 n1 n2 Sub3
.Ends Sub2
.SubCkt Sub3 n1 n2 Val=3
r3 n1 n2 Val
.Ends Sub3
.OP
.END
Hierarchical Parameter Passing Problem
The total resistance of the chain has two possible solutions: 0.3333Ω and
0.5455Ω.
You can use .OPTION PARHIER to specify which parameter value prevails,
when you define parameters with the same name at different levels of the
design hierarchy.
Under global scoping rules, if names conflict, the top-level assignment .PARAM
Val=1 overrides the subcircuit defaults, and the total is 0.3333Ω. Under local
scoping rules, the lower level assignments prevail, and the total is 0.5455Ω
(one, two, and three ohms in parallel).
The example in Figure 19 produces the results in Table 14, based on how you
set .OPTION PARHIER to local/global:
Table 14
94
PARHIER=LOCAL vs. PARHIER=GLOBAL Results
Element
PARHIER=Local
PARHIER=Global
r1
1.0
1.0
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Table 14
PARHIER=LOCAL vs. PARHIER=GLOBAL Results (Continued)
Element
PARHIER=Local
PARHIER=Global
r2
2.0
1.0
r3
3.0
1.0
Parameter Passing Solutions
The checklist below determines whether you will see simulation differences
when you use the default scoping rules. These checks are especially important
if your netlists contain devices from multiple vendor libraries.
■
Check your sub-circuits for parameter defaults, on the .SUBCKT or .MACRO
line.
■
Check your sub-circuits for a .PARAM statement, within a .SUBCKT
definition.
■
To check your circuits for global parameter definitions, use the .PARAM
statement.
■
If any of the names from the first three checks are identical, set up two
HSPICE simulation jobs: one with .OPTION PARHIER=GLOBAL, and one
with .OPTION PARHIER=LOCAL. Then look for differences in the output.
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Testbench Elements
6
Describes the syntax for the basic and specialized elements supported by
HSPICE RF for high-frequency analysis and characterization.
Elements are local and sometimes customized instances of a device model
specified in your design netlist.
For descriptions of the standard device models on which elements (instances)
are based, see the HSPICE Reference Manual: Elements and Device Models
and the HSPICE Reference Manual: MOSFET Models. For signal integrity
applications see the HSPICE User Manual: Signal Integrity.
HSPICE RF also supports several specialized elements for high-frequency
analysis and characterization.
These topics are covered in the following sections:
■
Passive Elements
■
Multi-Terminal Linear Elements
■
Port Element
■
Active Elements
■
Steady-State Voltage and Current Sources
■
Steady-State HB Sources
■
Phase Differences Between HB and SIN Sources
■
Behavioral Noise Sources
■
Function Approximations for Distributed Devices
■
Complex Signal Sources and Stimuli
■
SWEEPBLOCK in Sweep Analyses
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Chapter 6: Testbench Elements
Passive Elements
Passive Elements
This section describes the passive elements: resistors, capacitors, and
inductors. See Multi-Terminal Linear Elements for discussion of the W-, T-, and
S-elements.
This section includes:
■
Resistors
■
Linear Resistors
■
Behavioral Resistors
■
Skin Effect Resistors
■
Frequency-Dependent Resistors
■
Capacitors
■
Charge-Based Capacitors
■
Linear Capacitors
■
Frequency-Dependent Capacitors
■
Inductors
■
Mutual Inductors
■
Linear Inductors
■
Frequency-Dependent Inductors
■
Ideal Transformers
■
Coupled Inductor Element
■
Reluctance Format
■
Ideal Transformer Format in HSPICE RF
■
DC Block and Choke Elements
Resistors
Rxxx n1 n2 <mname> Rval <TC1 <TC2><TC>> <SCALE=val> <M=val>
+ <AC=val> <DTEMP=val> <L=val> <W=val> <C=val>
+ <NOISE=val>
Rxxx n1 n2 <mname> <R=>resistance <<TC1=>val>
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+ <<TC2=>val> <<TC=>val> <SCALE=val> <M=val>
+ <AC=val> <DTEMP=val> <L=val> <W=val>
+ <C=val> <NOISE=val>
Rxxx n1 n2 R=‘equation’ ...
Parameter
Description
Rxxx
Resistor element name. Must begin with R, followed by up to 1023
alphanumeric characters.
n1
Positive terminal node name.
n2
Negative terminal node name.
mname
Resistor model name. Use this name in elements, to reference a
resistor model.
TC
TC1 alias. The current definition overrides the previous definition.
TC1
First-order temperature coefficient for the resistor. See the Passive
Device Models chapter in the HSPICE Elements and Device Models
Manual for temperature-dependent relations.
TC2
Second-order temperature coefficient for the resistor.
SCALE
Element scale factor; scales resistance and capacitance by its value.
Default=1.0.
R=
resistance
Resistance value at room temperature. This can be:
■
■
■
■
■
a numeric value in ohms
a parameter in ohms
a function of any node voltages
a function of branch currents
any independent variables such as time, hertz, and temper
M
Multiplier to simulate parallel resistors. For example, for two parallel
instances of a resistor, set M=2, to multiply the number of resistors by
2. Default=1.0.
AC
Resistance for AC analysis. Default=Reff.
DTEMP
Temperature difference between the element and the circuit, in degrees
Celsius. Default=0.0.
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Parameter
Description
L
Resistor length in meters. Default=0.0, if you did not specify L in a
resistor model.
W
Resistor width. Default=0.0, if you did not specify W in the model.
C
Capacitance connected from node n2 to bulk. Default=0.0, if you did
not specify C in a resistor model.
user-defined Can be a function of any node voltages, element currents, temperature,
equation
frequency, or time
NOISE
■
■
NOISE=0, do not evaluate resistor noise.
NOISE=1, evaluate resistor noise (default).
Resistance can be a value (in units of ohms) or an equation. Required
parameters are the two nodes, and the resistance or model name. If you
specify other parameters, the node and model name must precede those
parameters. Other parameters can follow in any order. If you specify a resistor
model (see the Passive Device Models chapter in the HSPICE Elements and
Device Models Manual), the resistance value is optional.
The following are some basic examples for HSPICE RF.
Example 1
R1 is a resistor whose resistance follows the voltage at node c.
R1 1 0 ‘v(c)’
Example 2
R2 is a resistor whose resistance is the sum of the absolute values of nodes c
and d.
R2 1 0 ‘abs(v(c)) + abs(v(d))’
Example 3
R3 is a resistor whose resistance is the sum of the rconst parameter, and 100
times tx1 for a total of 1100 ohms.
.PARAM rconst=100 tx1=10
R3 4 5 ‘rconst + tx1 * 100’
R3 takes its value from the RX parameter, and uses the TC1 and TC2
temperature coefficients, which become 0.001 and 0, respectively.
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Example 4
You can use the HERTZ keyword to form frequency-dependent resistors.
HSPICE RF accurately analyzes these in all time-domain and frequencydomain simulations. In this example, R4 has resistance with both DC and skineffect contributions:
R4 in out R='100.0 + sqrt(HERTZ)/1000.0'
Linear Resistors
Rxxx node1 node2 < modelname > < R = > value < TC1=val >
+ < TC2=val > < W=val > < L=val > < M=val >
+ < C=val > < DTEMP=val > < SCALE=val >
Parameter
Description
Rxxx
Name of a resistor.
node1 and node2
Names or numbers of the connecting nodes.
modelname
Name of the resistor model.
value
Nominal resistance value, in ohms.
R
Resistance, in ohms, at room temperature.
TC1, TC2
Temperature coefficient.
W
Resistor width.
L
Resistor length.
M
Parallel multiplier.
C
Parasitic capacitance between node2 and the substrate.
DTEMP
Temperature difference between element and circuit.
SCALE
Scaling factor.
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Example
R1 1 2 10.0
Rload 1 GND RVAL
.param rx=100
R3 2 3 RX TC1=0.001 TC2=0
RP X1.A X2.X5.B .5
.MODEL RVAL R
In the example above, R1 is a simple 10Ω linear resistor and Rload calls a
resistor model named RVAL, which is defined later in the netlist.
Note:
If a resistor calls a model, then you do not need to specify a constant
resistance, as you do with R1.
■
R3 takes its value from the RX parameter, and uses the TC1 and TC2
temperature coefficients, which become 0.001 and 0, respectively.
■
RP spans across different circuit hierarchies, and is 0.5Ω.
Behavioral Resistors
HSPICE RF accepts equation-based resistors and capacitors. You can specify
the value of a resistor or capacitor as an arbitrary equation that involves node
voltages or variable parameters. Unlike HSPICE, you cannot use parameters to
indirectly reference node voltages in HSPICE RF.
Rxxx n1 n2 . . . <R=> ‘equation’ . . .
Example
R1 A B R=‘V(A) + I(VDD)’
Skin Effect Resistors
Rxxx n1 n2 R=value Rs=value
The Rs indicates the skin effect coefficient of the resistor.
The complex impedance of the resistor can be expressed as the following
equation:
R(f)=Ro + (1+j)*Rs*sqrt(f)
The Ro, j, and f are DC resistance, imaginably unit (j^2=-1) and frequency,
respectively.
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Frequency-Dependent Resistors
You can specify frequency-dependent resistors using the R=expression with
the HERTZ keyword. The HERTZ keyword represents the operating frequency.
In time domain analyses, an expression with the HERTZ keyword behaves
differently according to the value assigned to the CONVOLUTION keyword.
Syntax
Rxxx n+ n- R=expression(with HERTZ) <CONVOLUTION=0|1|2>
+ <FBASE=value> <FMAX=value>>
Parameter
CONVOLUTION
Description
Indicates which method is used.
■
■
■
FBASE
0: Acts the same as the conventional method. This is the
default.
1: Applies recursive convolution, and if the rational function
is not accurate enough, it switches to linear convolution.
2: Applies linear convolution.
Specifies the lower bound of the transient analysis frequency.
For CONVOLUTION=1 mode, HSPICE starts sampling at this
frequency. For CONVOLUTION=2 mode, HSPICE uses this
value as the base frequency point for Inverse Fourier
Transformation.
For recursive convolution, the default value is 0Hz, and for
linear convolution, HSPICE uses the reciprocal of the transient
period.
FMAX
Specifies the possible maximum frequency of interest. The
default value is the frequency point where the function reaches
close enough to infinity value, assuming that the monotonous
function is approaching the infinity value and that it is taken at
10THz.
The equation should be a function of HERTZ. If
CONVOLUTION is turned on when a HERTZ keyword is not
used in the equation, it is automatically be turned off to let the
resistor behave as conventional.The equation can be a function
of temperature, but it cannot be node voltage or branch current
and time.
The equation can only be a function of time-independent variables such as
hertz, and temperature.
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Example:
R1 1 2 r='1.0 + 1e-5*sqrt(HERTZ)' CONVOLUTION=1
Capacitors
The following general input syntax is for a capacitor.
Cxxx node1 node2 < modelname > < C = > capacitance
+ <TC1 = val> <TC2 = val> <W = val> <L = val>
+ <DTEMP = val> M = val> <SCALE = val> <IC = val>
Cxxx n1 n2 . . . C=‘equation’ CTYPE=[0|1|2]
Polynomial form:
Cxxx n1 n2 POLY c0 c1... <IC=val> <M=val>
Parameter
Description
Cxxx
Capacitor element name. Must begin with C, followed by up to 1023
alphanumeric characters.
POLY
Keyword, to specify capacitance as a non-linear polynomial.
c0 c1...
Coefficients of a polynomial, described as a function of the voltage
across the capacitor. c0 represents the magnitude of the 0th order
term, c1 represents the magnitude of the 1st order term, and so on.
You cannot use parameters as coefficient values.
node1 and node2 Names or numbers of connecting nodes.
104
capacitance
Nominal capacitance value in Farads.
modelname
Capacitance model name.
C
Capacitance at room temperature in Farads.
TC1, TC2
First-order and second-order temperature coefficient.
W
Capacitor width in meters.
L
Capacitor length in meters.
M
Multiplier to simulate multiple parallel capacitors.
DTEMP
Temperature difference between element and circuit.
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Parameter
Description
SCALE
Scaling factor.
IC
Initial capacitor voltage.
equation
Capacitance can be a function of any node voltage, and any branch
current, but not a function of time, frequency, or temperature.
CTYPE
Determines the calculation mode for elements that use capacitance
equations. Set this parameter carefully to ensure correct simulation
results. HSPICE RF extends the definition and values of CTYPE
relative to HSPICE:
■
■
■
0, if C depends only on its own terminal voltages—that is, a
function of V(n1<, n2>). This is consistent with HSPICE.
1, if C depends only on outside voltages or currents. This is
consistent with HSPICE.
2, if C depends on both its own terminal and outside voltages
(default for HSPICE RF). HSPICE does not use CTYPE=2.
You can specify capacitance as a numeric value, in units of farads, as an
equation, or as a polynomial of the voltage. The only required fields are the two
nodes, and the capacitance or model name.
■
If you use the parameter labels, the nodes and model name must precede
the labels. Other arguments can follow in any order.
■
If you specify a capacitor model (see the Passive Device Models chapter in
the HSPICE Elements and Device Models Manual), the capacitance value
is optional.
If you use an equation to specify capacitance, the CTYPE parameter
determines how HSPICE calculates the capacitance charge. The calculation is
different, depending on whether the equation uses a self-referential voltage
(that is, the voltage across the capacitor, whose capacitance is determined by
the equation).
To avoid syntax conflicts, if a capacitor model has the same name as a
capacitance parameter, HSPICE or HSPICE RF uses the model name.
Example 1
In the following example, C1 assumes its capacitance value from the model,
not the parameter.
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.PARAMETER CAPXX=1
C1 1 2 CAPXX
.MODEL CAPXX C CAP=1
Example 2
In the following example, the C1 capacitors connect from node 1 to node 2, with
a capacitance of 20 picofarads:
C1 1 2 20p
In this next example, Cshunt refers to three capacitors in parallel, connected
from the node output to ground, each with a capacitance of 100 femtofarads.
Cshunt output gnd C=100f M=3
The Cload capacitor connects from the driver node to the output node. The
capacitance is determined by the voltage on the capcontrol node, times 1E-6.
The initial voltage across the capacitor is 0 volts.
Cload driver output C=’1u*v(capcontrol)’ CTYPE=1 IC=0v
The C99 capacitor connects from the in node to the out node. The capacitance
is determined by the polynomial C=c0 + c1*v + c2*v*v, where v is the voltage
across the capacitor.
C99 in out POLY 2.0 0.5 0.01
Example 1
Cbypass 1 0 10PF
C1 2 3 CBX
.MODEL CBX C
CB B 0 10P IC = 4V
CP X1.XA.1 0 0.1P
In this example:
■
Cbypass is a straightforward, 10 pF capacitor.
■
C1 calls the CBX model, and its capacitance is not constant.
■
CB is a 10 pF capacitor with an initial voltage of 4V across it.
■
CP is a 0.1 pF capacitor.
Example 2
V1 1 0 pwl(0n 0v 100n 10v)
V2 2 0 pwl(0n 0v 100n 10v)
C1 1 0 C='(V(1) + V(2))*1e-12' CTYPE=2
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Example 3 (HSPICE RF Only)
C2 1 0 C='1 + TIME' $ Time-varying capacitor
Charge-Based Capacitors
You can also specify capacitors using behavioral equations for charge.
Syntax
Cxxx n1 n2 Q='equation'
dQ
C = -------, V = V (n1,n2) is equivalent to:
dV
Cxxx a b Q=’f(V(a,b))’
df ( x )
In the preceding equations, d ( x ) = ------------ .
dx
Example 1
C1 a b Q = ’sin(V(a,b)) + V(c,d)*V(a,b)’
This example is equivalent to:
C1 a b C = ’cos (V(a,b)) + V(c,d)’
Example 2
C3 3 0 Q = ‘TIME+TIME’
$ supported in HPICE RF only
Linear Capacitors
Cxxx node1 node2 < modelname > < C=> value < TC1=val >
+ < TC2=val > <W=val > < L=val > < DTEMP=val >
+ < M=val > < SCALE=val > < IC=val >
Parameter
Description
Cxxx
Name of a capacitor. Must begin with C, followed by up to 1023
alphanumeric characters.
node1 and node2
Names or numbers of connecting nodes.
value
Nominal capacitance value, in Farads.
modelname
Name of the capacitor model.
C
Capacitance, in Farads, at room temperature.
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Parameter
Description
TC1, TC2
Specifies the temperature coefficient.
W
Capacitor width.
L
Capacitor length.
M
Multiplier to simulate multiple parallel capacitors.
DTEMP
Temperature difference between element and circuit.
SCALE
Scaling factor.
IC
Initial capacitor voltage.
Frequency-Dependent Capacitors
You can specify frequency-dependent capacitors using the C=’equation’
with the HERTZ keyword. The HERTZ keyword represents the operating
frequency. In time domain analyses, an expression with the HERTZ keyword
behaves differently according to the value assigned to the CONVOLUTION
keyword.
Syntax
Cxxx n1 n2 C=’equation’ <CONVOLUTION=[0|1|2]
+ <FBASE=val> <FMAX=val>>
Parameter
Description
n1 n2
Names or numbers of connecting nodes.
equation
Expressed as a function of HERTZ. If CONVOLUTION=1 or 2
and HERTZ is not used in the equation, CONVOLUTION is
turned off and the capacitor behaves conventionally.
The equation can be a function of temperature, but it does not
support variables of node voltage, branch current, or time. If
these variables exist in the expression and CONVOLUTION=1 or
2, then only their values at the operating point are considered in
calculation.
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Parameter
Description
CONVOLUTION
Specifies the method used.
■
■
0 (default): HERTZ=0 in time domain analysis.
1 or 2: performs Inverse Fast Fourier Transformation (IFFT)
linear convolution.
FBASE
Base frequency to use for transient analysis. This value becomes
the base frequency point for Inverse Fast Fourier Transformation
(IFFT) when CONVOLUTION=1 or 2. If you do not set this value,
the base frequency is a reciprocal value of the transient period.
FMAX
Maximum frequency to use for transient analysis. Used as the
maximum frequency point for Inverse Fourier Transformation. If
you do not set this value, the reciprocal value of RISETIME is
taken.
Example
C1 1 2 C='1e-6 - HERTZ/1e16' CONVOLUTION=1 fbase=10 fmax=30meg
Inductors
General form:
Lxxx n1 n2 <L=>inductance <<TC1=>val>
+ <<TC2=>val> <SCALE=val> <IC=val> <M=val>
+ <DTEMP=val> <R=val>
Lxxx n1 n2 L=‘equation’ <LTYPE=val> <above_options...>
Polynomial form:
Lxxx n1 n2 POLY c0 c1... <above_options...>
Magnetic winding form:
Lxxx n1 n2 NT=turns <above_options...>
Parameter
Description
Lxxx
Inductor element name. Must begin with L, followed by up to
1023 alphanumeric characters.
n1
Positive terminal node name.
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Parameter
Description
n2
Negative terminal node name.
TC1
First-order temperature coefficient for the inductor. See the
Passive Device Models chapter in the HSPICE Elements and
Device Models Manual for temperature-dependent relations.
TC2
Second-order temperature coefficient for the inductor.
SCALE
Element scale parameter; scales inductance by its value.
Default=1.0.
IC
Initial current through the inductor, in amperes. HSPICE or
HSPICE RF uses this value as the DC operating point current.
L=inductance
Inductance value. This can be:
■
■
■
■
■
M
Multiplier, used to simulate parallel inductors. Default=1.0.
DTEMP
Temperature difference between the element and the circuit, in
degrees Celsius. Default=0.0.
R
Resistance of the inductor, in ohms. Default=0.0.
L=‘equation’
Inductance at room temperature, specified as:
■
■
■
LTYPE
110
a numeric value, in henries
a parameter in henries
a function of any node voltages
a function of branch currents
any independent variables such as time, hertz, and
temper
a function of any node voltages
a function of branch currents
any independent variables such as time, hertz, and
temper
Calculates inductance flux for elements, using inductance
equations. If the L inductance is a function of I(Lxxx), then set
LTYPE=0. Otherwise, set LTYPE=1. Use this setting correctly, to
ensure proper inductance calculations, and correct simulation
results. Default=0.
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Parameter
Description
POLY
Keyword that specifies the inductance, calculated by a
polynomial.
c0 c1...
Coefficients of a polynomial in the current, describing the
inductor value. c0 is the magnitude of the 0th order term, c1 is
the magnitude of the 1st order term, and so on.
NT=turns
Number of turns of an inductive magnetic winding.
In this syntax, the inductance can be either a value (in units of henries), an
equation, a polynomial of the current, or a magnetic winding. Required fields
are the two nodes, and the inductance or model name.
■
If you specify parameters, the nodes and model name must be first. Other
parameters can be in any order.
■
If you specify an inductor model (see the Passive Device Models chapter in
the HSPICE Elements and Device Models Manual), the inductance value is
optional.
Example 1
In the following example, the L1 inductor connects from the coilin node to the
coilout node, with an inductance of 100 nanohenries.
L1 coilin coilout 100n
Example 2
The Lloop inductor connects from node 12 to node 17. Its inductance is 1
microhenry, and its temperature coefficients are 0.001 and 0.
Lloop 12 17 L=1u TC1=0.001 TC2=0
Example 3
The Lcoil inductor connects from the input node to ground. Its inductance is
determined by the product of the current through the inductor, and 1E-6.
Lcoil input gnd L=’1u*i(input)’ LTYPE=0
Example 4
The L99 inductor connects from the in node to the out node. Its inductance is
determined by the polynomial L=c0 + c1*i + c2*i*i, where i is the current
through the inductor. The inductor has a specified DC resistance of 10 ohms.
L99 in out POLY 4.0 0.35 0.01 R=10
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Example 5
The L inductor connects from node 1 to node, as a magnetic winding element,
with 10 turns of wire.
L 1 2 NT=10
Mutual Inductors
General form:
Kxxx Lyyy Lzzz <K=coupling | coupling>
Mutual core form:
Kaaa Lbbb <Lccc ... <Lddd>> mname <MAG=magnetization>
112
Parameter
Description
Kxxx
Mutual inductor element name. Must begin with K, followed by up
to 1023 alphanumeric characters.
Lyyy
Name of the first of two coupled inductors.
Lzzz
Name of the second of two coupled inductors.
K=coupling
Coefficient of mutual coupling. K is a unitless number, with
magnitude > 0. If K is negative, the direction of coupling reverses.
This is equivalent to reversing the polarity of either of the coupled
inductors. Use the K=coupling syntax when using a parameter
value or an equation, and the keyword “k=” can be omitted.
Kaaa
Saturable core element name. Must begin with K, followed by up
to 1023 alphanumeric characters.
Lbbb, Lccc, Lddd
Names of the windings about the Kaaa core. One winding
element is required, and each winding element must use the
magnetic winding syntax. All winding elements with the same
magnetic core model should be written in one mutual inductor
statement in the netlist.
mname
Saturable core model name. (See the Passive Device Models
chapter in the HSPICE Elements and Device Models Manual for
more information.)
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Parameter
Description
MAG=
Initial magnetization of the saturable core. You can set this to +1,
0, or -1, where +/- 1 refer to positive and negative values of the
BS model parameter. (See the Passive Device Models chapter in
the HSPICE Elements and Device Models Manual for more
information.)
magnetization
In this syntax, coupling is a unitless value from zero upward, representing the
coupling strength. If you use parameter labels, the nodes and model name
must be first. Other arguments can be in any order. If you specify an inductor
model (see the Passive Device Models chapter in the HSPICE Elements and
Device Models Manual), the inductance value is optional.
You can determine the coupling coefficient, based on geometric and spatial
information. To determine the final coupling inductance, HSPICE or HSPICE
RF divides the coupling coefficient by the square-root of the product of the selfinductances.
When using the mutual inductor element to calculate the coupling between
more than two inductors, HSPICE or HSPICE RF can automatically calculate
an approximate second-order coupling. See the third example below for a
specific situation.
Note:
The automatic inductance calculation is an estimation, and is accurate for a
subset of geometries. The second-order coupling coefficient is the product
of the two first-order coefficients, which is not correct for many geometries.
Example 1
The Lin and Lout inductors are coupled, with a coefficient of 0.9.
K1 Lin Lout 0.9
Example 2
The Lhigh and Llow inductors are coupled, with a coefficient equal to the value
of the COUPLE parameter.
Kxfmr Lhigh Llow K=COUPLE
■
The K1 mutual inductor couples L1 and L2.
■
The K2 mutual inductor couples L2 and L3.
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Example 3
The coupling coefficients are 0.98 and 0.87. HSPICE or HSPICE RF
automatically calculates the mutual inductance between L1 and L3, with a
coefficient of 0.98*0.87=0.853.
K1 L1 L2 0.98
K2 L2 L3 0.87
Linear Inductors
Lxxx node1 node2 <L => inductance <TC1=val> <TC2=val>
+ <M=val> <DTEMP=val> <IC=val>
Parameter
Description
Lxxx
Name of an inductor.
node1 and node2
Names or numbers of the connecting nodes.
inductance
Nominal inductance value, in Henries.
L
Inductance, in Henries, at room temperature.
TC1, TC2
Temperature coefficient.
M
Multiplier for parallel inductors.
DTEMP
Temperature difference between the element and the circuit.
IC
Initial inductor current.
Example:
LX A B 1E-9
LR 1 0 1u IC=10mA
■
LX is a 1 nH inductor.
■
LR is a 1 uH inductor, with an initial current of 10 mA.
Frequency-Dependent Inductors
You can specify frequency-dependent inductors using the L=’equation’ with
the HERTZ keyword. The HERTZ keyword represents the operating frequency.
In time domain analyses, an expression with the HERTZ keyword behaves
differently according to the value assigned to the CONVOLUTION keyword.
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Syntax
Lxxx n1 n2 L=equation <CONVOLUTION=[0|1|2] <FBASE=valule>
+ <FMAX=value>>
Parameter
Description
Lxxx
Inductor element name. Must begin with L, followed by up to
1023 alphanumeric characters
n1 n2
Positive and negative terminal node names.
equation
The equation should be a function of HERTZ. If
CONVOLUTION is turned on when a HERTZ keyword is not
used in the equation, CONVOLUTION is automatically be
turned off and the inductor behaves conventionally.The
equation can be a function of temperature, but it does not
support variables of node voltage, branch current, or time. If
these variables exist in the equation with CONVOLUTION
turned on, only their values at the operating point are
considered in the calculation.
CONVOLUTION
Indicates which method is used.
■
■
■
FBASE
Specifies the lower bound of the transient analysis frequency.
■
■
■
■
FMAX
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0 (default): Acts the same as the conventional method.
1 : Applies recursive convolution, and if the rational function
is not accurate enough, it switches to linear convolution.
2 : Applies linear convolution.
For CONVOLUTION=1 mode, HSPICE starts sampling at
this frequency.
For CONVOLUTION=2 mode, HSPICE uses this value as
the base frequency point for Inverse Fourier Transformation.
For recursive convolution, the default value is 0Hz.
For linear convolution, HSPICE uses the reciprocal of the
transient period.
Specifies the possible maximum frequency of interest. The
default value is the frequency point where the function reaches
close enough to infinity value, assuming that the monotonous
function is approaching the infinity value and that it is taken at
10THz.
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Example
L1 1 2 L='0.5n + 0.5n/(1 + HERTZ/1e8)' CONVOLUTION=1 fbase=10
+ fmax=30meg
Ideal Transformers
You can use the IDEAL keyword with the K element to designate ideal
transformer coupling.
Syntax
Kxxx Ij Lj <k=IDEAL | IDEAL>
The IDEAL keyword replaces the coupling factor value. This keyword activates
the following equation set for non-DC values, which is presented here with
multiple coupled inductors. Ij is the current into the first terminal of Lj.
Equation 1
v1
v2
v3
v4
---------- = ---------- = ---------- = ---------- = ...
L1
L2
L3
L4
Equation 2
0 = ( il ⋅
L1 ) + ( i2 ⋅
L2 ) + ( i3 ⋅
L3 ) + ( i4 ⋅
L4 ) + ...
HSPICE RF can solve any i or v in terms of L ratios.
For two inductors (non-DC values):
Equation 3
v1
v2
---------- = ---------L1
L2
Equation 4
0 = ( il ⋅
Equation 5
v2 = v1 ⋅
Equation 6
i2 = i1 ⋅
L1 ) + ( i2 ⋅
L2 )
L2
-----L1
L1
-----L2
DC is treated as usual—inductors are treated as short circuits. DC ignores
mutual coupling.
You can couple inductors that use the INFINITY keyword to IDEAL K
elements. All inductors involved must have the INFINITY value, and for
K=IDEAL, the ratios of all L values is unity. Then, for two L values:
v2 = v1
i2 = -i1
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Example 1
This example is a standard 5-pin ideal balun transformer subcircuit. Two pins
are grounded for standard operation. With all K values being IDEAL, the
absolute L values are not crucial—only their ratios are important.
**
**
all K's ideal
**
**
o----in**
Lin=1
** 0 o------**
.subckt BALUN1 in
Lin
in
gnd
Lo1
out1 gnd
Lo2
gnd
out2
K12
Lin Lo1
K13
Lin Lo2
K23
Lo1 Lo2
.ends
-----o out1
Lo1=.25
-----o 0
Lo2=.25
-----o out2
out1 out2
L=1
L=0.25
L=0.25
IDEAL
IDEAL
IDEAL
Example 2
This example is a 2-pin ideal 4:1 step-up balun transformer subcircuit with
shared DC path (no DC isolation). Input and output have a common pin, and
both inductors have the same value. Note that Rload = 4*Rin.
**
**
all K's ideal
**in o-------------------o out=in
**
L1=1
**
-----o 0
**
L2=1
**
-----o out2
**
** With all K's ideal, the actual L's values are
** not important -- only their ratio to each other.
.subckt BALUN2 in out2
L1
in
gnd
L=1
L2
gnd out2 L=1
K12
L1
L2
IDEAL
.ends
Example 3
This example is a 3-pin ideal balun transformer with shared DC path (no DC
isolation). All inductors have the same value (here set to unity).
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**
**
all K's ideal -----o out1
**
Lo2=1
**
-----o 0
**
Lo1=1
**
-----o out2
**
in
Lin=1
**
o-------------------o in
**
.subckt BALUN3 in out1 out2
Lo2
gnd out1 L=1
Lo1
out2 gnd
L=1
Lin
in
out2 L=1
K12
Lin Lo1 IDEAL
K13
Lin Lo2 IDEAL
K23
Lo1 Lo2 IDEAL
.ends
Coupled Inductor Element
This section describes the multiport syntax for coupled inductor elements. This
syntax extends the existing linear (Lxxx) and mutual (Kxxx) inductor elements.
Two syntax configurations are available:
■
a reluctance format that is used by Star-RCXT for inductance extraction
■
an ideal transformer format that can be used to create balanced converter
(that is, balun) models in HSPICE RF.
Reluctance Format
Syntax
Reluctance Inline Form
Lxxx n1p n1n ... nNp nNn
+ RELUCTANCE=(r1, c1, val1, r2, c2, val2, ... , rm, cm, valm)
+ [SHORTALL=yes|no] [IGNORE_COUPLING=yes|no]
Reluctance External File Form
Lxxx n1p n1n ... nNp nNn RELUCTANCE
+ FILE=“filename1” [FILE="filename2" [...]]
+ [SHORTALL=yes|no] [IGNORE_COUPLING=yes no]
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Parameter
Description
Lxxx
Name of a reluctor. Must begin with L, followed by up to 1023
alphanumeric characters
n1p n1n ... nNp nNn
Names of the connecting terminal nodes. The number of
terminals must be even. Each pair of ports represents the
location of an inductor.
RELUCTANCE
Keyword to specify reluctance (inverse inductance).
r1, c1, val1,
r2, c2, val2, ...
rm, cm, valm
Reluctance matrix data. In general, K will be sparse and only
non-zero values in the matrix need be given. Each matrix entry
is represented by a triplet (r,c,val). The value r and c are
integers referring to a pair of inductors from the list of terminal
nodes. If there are 2*N terminal nodes, there will be N
inductors, and the r and c values must be in the range [1,N].
The val value is a reluctance value for the (r,c) matrix location,
and the unit for reluctance is the inverse Henry (H-1).
Only terms along and above the diagonal are specified for the
reluctance_matrix.
The simulator fills in the lower triangle to ensure symmetry. If
you specify lower diagonal terms, the simulator converts that
entry to the appropriate upper diagonal term.
If multiple entries are supplied for the same (r,c) location, then
only the first one is used, and a warning will be issued
indicating that some entries are ignored.
All diagonal entries of the reluctance matrix must be assigned
a positive value.
The reluctance matrix should be positive definite.
FILE=”<filename1>”
For the external file format, the data files should contain three
columns of data. Each row should contain an (r,c,val) triplet
separated by white space. The r, c, and val values may be
expressions surrounded by single quotes. Multiple files may be
specified to allow the reluctance data to be spread over several
files if necessary.
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Parameter
Description
SHORTALL
■
■
IGNORE_COUPLING
■
■
SHORTALL=yes, all inductors in this model are converted
to short circuits, and all reluctance matrix values are
ignored.
SHORTALL=no (default), inductors are not converted to
short circuits, and reluctance matrix values are not ignored.
IGNORE_COUPLING=yes, all off-diagonal terms are
ignored (that is, set to zero).
IGNORE_COUPLING=no (default), off-diagonal terms are
not ignored.
Example
This example has 9 segments (or ports) with 12 nodes, and can potentially
generate a 9x9 reluctance matrix with 81 elements.
L_ThreeNets a 1 1 2 2 a_1 b 4 4 5 5 b_1 c 7 7 8 8 c_1
+ RELUCTANCE=(
+ 1 1
103e9
+ 1 4
-34.7e9
+ 1 7
-9.95e9
+ 4 4
114e9
+ 4 7
-34.7e9
+ 7 7
103e9
+ 2 2
103e9
+ 2 5
-34.7e9
+ 2 8
-9.95e9
+ 5 5
114e9
+ 5 8
-34.7e9
+ 8 8
103e9
+ 3 3
103e9
+ 3 6
-34.7e9
+ 3 9
-9.95e9
+ 6 6
114e9
+ 6 9
-34.7e9
+ 9 9
103e9 )
+ SHORTALL = no IGNORE_COUPLING = no
Alternatively, the same element could be specified by using:
L_ThreeNets a 1 1 2 2 a_1 b 4 4 5 5 b_1 c 7 7 8 8 c_1 RELUCTANCE
+ FILE="reluctance.dat" SHORTALL = no IGNORE_COUPLING = no
Where reluctance.dat contains:
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+
+
+
+
+
+
+
+
+
+
+
+
+
+
+
+
+
+
1
1
1
4
4
7
2
2
2
5
5
8
3
3
3
6
6
9
1
4
7
4
7
7
2
5
8
5
8
8
3
6
9
6
9
9
103e9
-34.7e9
-9.95e9
114e9
-34.7e9
103e9
103e9
-34.7e9
-9.95e9
114e9
-34.7e9
103e9
103e9
-34.7e9
-9.95e9
114e9
-34.7e9
103e9
The following shows the mapping between the port numbers and node pairs:
------------------------------------------------------------------------------------|Ports
|
1
|
2
|
3
|
4
|
5
|
6
|
7
|
8
|
9
|
|Node pairs | (a,1) | (1,2) |(2,a_1)| (b,4) | (4,5) |(5,b_1)| (c,7) | (7,8) |(8,c_1)|
-------------------------------------------------------------------------------------
Ideal Transformer Format in HSPICE RF
The ideal transformer format simplifies modeling of baluns. Previously, baluns
were modeled using mutual inductors (K elements) with the IDEAL keyword.
Multiple L and K elements were needed for a given balun model. The ideal
transformer model allows modeling of a balun using a single L element.
In the ideal transformer format, no absolute inductance or reluctance values
are specified. Instead, the transformer’s coupling characteristics are specified
using inductor number-of-turns values. The behavior of the ideal transformer
depends on ratios of the inductors’ number of turns.
Syntax
Lxxx n1p n1n ... nNp nNn TRANSFORMER_NT=(nt1, ... , ntN)
Parameter
Description
Lxxx
Inductor element name. Must begin with L, followed by up to
1023 alphanumeric characters.
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Parameter
Description
n1p n1n ... nNp nNn
Positive and negative terminal node names. The number of
terminals must be even. Each pair of reports represents the
location of an inductor.
TRANSFORMER_NT Number of turns values. These parameters must match the
number of inductors.
The ideal transformer element obeys the standard ideal transformer equations:
ν
ν
νN
------1- = ------2- = … = ------nt 1
nt 2
nt N
i 1 nt 1 + i 2 nt 2 + …+ i N nt N = 0
Example
L1 1 0 0 2 3 0 transformer_nt=(1,2,2)
DC Block and Choke Elements
In HSPICE RF, you can specify an INFINITY value for capacitors and inductors
to model ideal DC block and choke elements. The following input syntax is for
the DC block (ideal infinite capacitor):
Syntax
Cxxx node1 node2 <C=> INFINITY <IC=val>
HSPICE RF does not support any other capacitor parameters for DC block
elements, because HSPICE RF assumes that the infinite capacitor value is
independent of temperature and scaling factors. The DC block acts as an open
circuit for all DC analyses. HSPICE RF calculates the DC voltage across the
circuit’s nodes. In all other (non-DC) analyses, a DC voltage source of this
value represents the DC block (that is, HSPICE RF does not then allow dv/dt
variations).
The following input syntax is for the Choke (ideal infinite inductor):
Syntax
Lxxx node1 node2 <L=> INFINITY <IC=val>
HSPICE RF does not support any other inductor parameters, because HSPICE
RF assumes that the infinite inductance value is independent of temperature
and scaling factors. The choke acts as a short circuit for all DC analyses.
HSPICE RF calculates the DC current through the inductor. In all other (non-
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DC) analyses, a DC current source of this value represents the choke (that is,
HSPICE RF does not then allow di/dt variations).
Multi-Terminal Linear Elements
A multi-terminal linear element such as a transmission line is a passive element
that connects any two conductors at any distance apart. One conductor sends
the input signal through the transmission line, and the other conductor receives
the output signal from the transmission line. The signal is voltage between the
conductors that is transmitted from one end of the pair to the other end.
See also, T-element (Ideal Transmission Lines) in the HSPICE User Guide:
Signal Integrity.
Examples of transmission lines include:
■
Power transmission lines
■
Telephone lines
■
Waveguides
■
Traces on printed circuit boards and multi-chip modules (MCMs)
■
Bonding wires in semiconductor IC packages
■
On-chip interconnections
The following sections discuss:
■
W-element (Distributed Transmission Lines)
■
Scattering Parameter Data Element
W-element (Distributed Transmission Lines)
The W-element supports five different formats to specify the transmission line
properties:
■
■
Model 1: RLGC-Model specification
•
Internally specified in a .model statement
•
Externally specified in a different file
Model 2: U-Model specification
•
RLGC input for up to five coupled conductors
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Multi-Terminal Linear Elements
■
•
Geometric input (planer, coax, twin-lead)
•
Measured-parameter input
•
Skin effect.
Model 3: Built-in field solver model
•
Standard format (using geometric data with the W-element)
•
Tabular format
■
Model 4: Frequency-dependent tabular model
■
Model 5: S-parameter Model
W-element Statement
The general syntax for a lossy (W-element) transmission line element is:
RLGC file form:
Wxxx in1 <in2 <...inx>> refin out1 <out2 <...outx>>
+ refout <RLGCfile=filename> N=val L=val
U Model form:
Wxxx in1 <in2 <...inx>> refin out1 <out2 <...outx>>
+ refout <Umodel=modelname> N=val L=val
Field solver form:
Wxxx in1 <in2 <...inx>> refin out1 <out2 <...outx>>
+ refout <FSmodel=modelname> N=val L=val
Table Model form:
Wxxx in1 in2 <...inx>> refin out1 <out2 <...outx>>
+ refout N=val L=val TABLEMODEL=name
S Model form:
Wxxx in1 <in2 <...inx>> refin out1 <out2 <...outx>>
+ refout <Smodel=modelname> <NODEMAP=XiYj...> N=val L=val
124
Parameter
Description
Wxxx
Lossy (W-element) transmission line element name. Must start
with W, followed by up to 1023 alphanumeric characters.
inx
Signal input node for xth transmission line (in1 is required).
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Multi-Terminal Linear Elements
Parameter
Description
refin
Ground reference for input signal
outx
Signal output node for the xth transmission line (each input port
must have a corresponding output port).
refout
Ground reference for output signal.
N
Number of conductors (excluding the reference conductor).
L
Physical length of the transmission line, in units of meters.
RLGCfile=filename
File name reference for the file containing the RLGC
information for the transmission lines (for syntax, see Using the
W-element in the HSPICE Signal Integrity Guide).
Umodel=modelname U-model lossy transmission-line model reference name. A
lossy transmission line model, used to represent the
characteristics of the W-element transmission line.
FSmodel=
modelname
Internal field solver model name. References the PETL internal
field solver as the source of the transmission-line
characteristics (for syntax, see Using the Field Solver Model
section in the HSPICE Signal Integrity Guide).
NODEMAP
String that assigns each index of the S parameter matrix to one
of the W-element terminals. This string must be an array of
pairs that consists of a letter and a number, (for example, Xn),
where
■
X= I, i, N, or n to indicate near end (input side) terminal of
the W-element
■
X= O, i, F, or f to indicate far end (output side) terminal of the
W-element.
The default value for NODEMAP is “I1I2I3...InO1O2O3...On”
Smodel
S Model name reference, which contains the S-parameters of
the transmission lines (for the S Model syntax, see the HSPICE
Signal Integrity Guide).
TABLEMODEL
Name of the frequency-dependent tabular model.
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The number of ports on a single transmission line is not limited. You must
provide one input and output port, the ground references, a model or file
reference, a number of conductors, and a length.
Example 1
The W1 lossy transmission line connects the in node to the out node:
W1 in gnd out gnd RLGCfile=cable.rlgc N=1 L=5
Where,
■
Both signal references are grounded
■
The RLGC file is named cable.rlgc
■
The transmission line is 5 meters long.
Example 2
The Wcable element is a two-conductor lossy transmission line:
Wcable in1 in2 gnd out1 out2 gnd Umodel=umod_1 N=2
+ L=10
Where,
■
in1 and in2 input nodes connect to the out1 and out2 output node
■
Both signal references are grounded.
■
umod_1 references the U-model.
■
The transmission line is 10 meters long.
Example 3
The Wnet1 element is a five-conductor lossy transmission line:
Wnet1 i1 i2 i3 i4 i5 gnd o1 gnd o3 gnd o5 gnd
+ FSmodel=board1 N=5 L=1m
Where,
126
■
The i1, i2, i3, i4 and i5 input nodes connect to the o1, o3, and o5 output
nodes.
■
The i5 input and three outputs (o1, o3, and o5) are all grounded.
■
board1 references the Field Solver model.
■
The transmission line is 1 millimeter long.
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Example 4: S Model Example
Wnet1 i1 i2 gnd o1 o2 gnd
+ Smodel=smod_1 nodemap=i1i2o1o2
+ N=2 L=10m
Where,
■
in1 and in2 input nodes connect to the out1 and out2 output node.
■
Both signal references are grounded.
■
smod_1 references the S Model.
■
The transmission line is 10 meters long.
You can specify parameters in the W-element card in any order. You can
specify the number of signal conductors, N, after the node list. You can also mix
nodes and parameters in the W-element card.
You can specify only one of the RLGCfile, FSmodel, Umodel, or Smodel
models, in a single W-element card.
Figure 20 shows node numbering for the element syntax.
N+1 conductor line
[i1]1
1.1 [i ]
12
1.2
[i1]N
1.N
1’
[v1]1
R(f), L(f), G(f), C(f)
[v2]1
[v1]2
Signal Conductors
[v2]2
.
.
.
Reference conductor
0
Figure 20
[i2]2
.
.
.
.
.
.
[v1]N
+
_
[i2]1
2.1
2.2
[i2]N
2.N
[v2]N
+
_
2’
x
Terminal Node Numbering for the W-element
For additional information about the W-element, see the W-element Modeling
of Coupled Transmission Lines chapter in the HSPICE User Manual: Signal
Integrity.
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Scattering Parameter Data Element
All HSPICE and HSPICE RF analyses can use the S-element. For more
information about S-parameters, see S-parameter Modeling Using the Selement in the HSPICE Signal Integrity Guide.
Frequency-Dependent Multi-Terminal (S-element)
When used with the generic frequency-domain model (.MODEL SP), an Selement is a convenient way to describe the behavior of a multi-terminal
network.
The S-element describes a linear time-invariant system, and provides a series
of data that describe the frequency response of the system. The S-element is
particularly useful for high-frequency characterization of distributed passive
structures. A common use of the S-element is in microwave circuits such as
spiral inductors, because electronic devices in this frequency domain no longer
act as they do in low frequencies. In this case, distributed system parameters
must be considered. See Example 9 below for an application of the state space
stamping to generate a frequency invariant modified nodal analysis (NMA)
matrix from frequency-dependent characteristics using the Shooting Newton
(.SN) algorithm.
The S-element uses the following parameters to define a frequency-dependent,
multi-terminal network:
■
S (scattering) parameter
■
Y (admittance) parameter
The S-parameter is the reflection coefficient of the system, which is measured
through ratios of incident and reflected sinusoidal waves. For passive systems,
the magnitude of an S parameter varies between zero and one. Because the
reflection coefficient is easy to measure in real microwave circuits, the Sparameter can be a very useful tool for microwave engineers.
You can use the S-element with a .MODEL SP, or with data files that describe
the frequency response of a network and provide discrete frequency
dependent data (Touchstone and CITIfile). You can measure this data directly
using network analyzers such as Hewlett-Packard's MDS (Microwave Design
System) or HFSS (High Frequency Structure Simulator). HSPICE can also
extract the S element from a real circuit system.
For a description of the S-parameter and SP analyses, see S-parameter
Modeling Using the S-element in the HSPICE User Manual: Signal Integrity.
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The nodes of the S-element must come first. If MNAME is not declared, you must
specify the FQMODEL. You can specify all the optional parameters in both the Selement and S model statements, except for MNAME argument.
You can enter the optional arguments in any order, and the parameters
specified in the element statement have a higher priority.
If the number of nodes in the element card is smaller than the number specified
in the model card (or external file) by 1, then the reference node is the default.
The default reference node is 0 (gnd).
For scattering parameter element and model syntax, see S-element Syntax
and S Model Syntax in the HSPICE User Manual: Signal Integrity.
.
.
.
.
.
.
[vinc]1
[i]1
[vref]1
.
.
.
N+1 terminal system
[vinc]N
[i]N
[vref]N
ndN
(+) [v]N
nd1
(+) [v]1
(-) ndR
(reference node)
Figure 21
Terminal Node Notation
Examples The FQMODEL, TSTONEFILE, and CITIFILE parameters
describe the frequency-varying behavior of a network. Only specify one of the
parameters in an S model card. If more than one method is declared, only the
first one is used and HSPICE issues a warning message.
FQMODEL can be set in S-element and S model statements, but both
statements must refer to the same model name.
The S-element is capable of reading in two-port noise parameter data from
Touchstone data files and then transform the raw data into a form used for
noise (and .LIN 2PNOISE) analysis.
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For example, you can represent a two-port system with an S element and then
perform a noise analysis (or any other analysis). The S-element noise model
supports normal, two-port and N-Port noise analysis (.NOISE =1) and .LIN
NOISECALC=1 for two-port, .LIN NOISECALC=2 for N-port).
Example 1
s1 n1 n2 n3 n_ref mname=smodel
.model smodel s n=3 fqmodel=sfqmodel zo=50 fbase=25e6
+ fmax=1e9
Example 2
s1 n1 n2 n3 n_ref fqmodel=sfqmodel zo=50 fbase=25e6 fmax=1e9
Examples 1 and 2 return the same result.
Example 3
s1 n1 n2 n3 n_ref mname=smodel zo=100
.model smodel s n=3 fqmodel=sfqmodel zo=50 fbase=25e6
+ fmax=1e9
In this example, the characteristic impedance of each port is 100 ohms, instead
of 50 ohms as defined in smodel, because parameters defined in the S
element statement have higher priority than those defined in the S model
statement.
Example 4
s1 n1 n2 n3 n_ref mname=smodel
.model smodel s n=3 fqmodel=sfqmodel zo=50 50 100
In this example, the characteristic impedance of port1 and port2 are 50 ohms,
and the characteristic impedance of port3 is 100 ohms.
Example 5
s1 n1 n2 n3 n_ref mname=smodel
.model smodel s tstonefile=expl.s3p
In this example, the name of the tstone file, expl.s3p, reveals that the network
has three ports.
Example 6
s1 n1 n2 n3 n_ref mname=smodel
.model smodel s fqmodel=sfqmodel tstonefile=expl.s3p
+ citifile=expl.citi0
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In this example, fqmodel, tstonefile, and citifile are all declared.
HSPICE uses only the fqmodel, ignores tstonefile and citifile, and
reports warning messages.
Example 7
s1 n1 n2 n3 n_ref mname=smodel fqmodel=sfqmodel_1
.model smodel s n=3 fqmodel=sfqmodel_2
In this example, fqmodel is declared in both the S element statement and the
S model statement, and they have different fqmodel names. This is not
allowed in HSPICE.
Example 8
s1 n1 n2 n3 n_ref mname=smodel fqmodel=sfqmodel
.model smodel s tstonefile=expl.s3p
In this example, fqmodel is already declared in the s1 statement, and
tstonefile is declared in the related smodel card. This is a conflict when
describing the frequency-varying behavior of the network, which is not allowed
in HSPICE.
Example 9
The following netlist and data file (test.rfm) show how the S-element “S1” uses
the “STAMP=YSTS” configuration which invokes the state space stamping to
generate a frequency invariant modified nodal analysis (NMA) matrix from
frequency-dependent characteristics. This stamping method allows the
Shooting-Newton algorithm (.SN) to obtain the steady state. Note that unless
RFM file input is given, the S-element first applies the rational function
approximation (equivalent behavior to RATIONAL_FUNCTION=1) to the
original S-parameters in order to generate the state space stamping.
======= main netlist =======
*** .SN with s-element example
P1 n1 gnd port=1 dc=1v ac=1v pulse(1 0 1n 1n 1n 10n 20n)
P2 n2 gnd port=2 dc=1v ac=1v pulse(1 0 1n 1n 1n 10n 20n)
S1 n1 n2 0 mname=s_model
.model s_model S n=2
+ rfmfile='test.rfm'
+ STAMP=YSTS
.SN tone=0.05Ghz nharms=32
.option post accurate
.end
The following is from the .lis file for this netlist.
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Chapter 6: Testbench Elements
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======= rational function matrix file (test.rfm) ======
VERSION 200600 NPORT 2 MATRIX_TYPE Y SYMMETRIC PRECFAC 0.75 Z0 50 50
BEGIN 1 1
BEGIN_REAL 9
DC 2.10290261e-02
2.80562648113e+07
1.36806220992e+08
1.16867967247e+09
1.23552099406e+09
1.92568095149e+09
4.15005808751e+09
1.00149288271e+10
2.27536895845e+10
3.54118199282e+10
1.791888661818e+00
-5.313505935943e+01
2.840375731037e+06
-4.257158329976e+06
3.038955064913e+06
-8.058749095413e+06
3.846931398394e+06
1.702938150800e+05
-1.243885701867e+07
BEGIN_COMPLEX 5
5.53251427579e+05 1.28282249537e+06 -3.17377193705e-03 1.20935639131e-03
2.39642428296e+09 1.39710928734e+08 -1.99538130185e+07 6.93072640638e+07
2.41275272760e+09 4.88535891322e+09 2.92904966609e+04
4.08311621367e+04
9.49575839142e+08 -2.82753080087e+10 -1.69178467311e+05 1.42790736653e+04
3.74702282735e+10 2.26461714292e+1
6.18960971035e+06 2.73309486084e+05 END BEGIN 2 2 DC 2.10290261e02
BEGIN REAL 9
2.80562648113e+07
1.36806220992e+08
1.16867967247e+09
1.23552099406e+09
1.92568095149e+09
4.15005808751e+09
1.00149288271e+10
2.27536895845e+10
3.54118199282e+10
1.79188866181e+00
-5.31350593594e+01
2.84037573103e+06
-4.25715832997e+06
3.03895506491e+06
-8.05874909541e+06
3.84693139839e+06
1.70293815080e+05
-1.24388570186e+07
BEGIN_COMPLEX 5
5.53251427579e+05 1.28282249537e+06 -3.17377193705e-03 1.20935639131e-03
2.39642428296e+09 1.39710928734e+08 -1.99538130185e+07 6.93072640638e+07
2.41275272760e+09 4.88535891322e+09 2.92904966609e+04
4.08311621367e+04
9.49575839142e+08 -2.82753080087e+10 -1.69178467311e+05 1.42790736653e+04
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Port Element
3.74702282735e+10 2.26461714292e+10
6.18960971035e+06 2.73309486084e+05 END
Port Element
The port element identifies the ports used in LIN analysis. Each port element
requires a unique port number. If your design uses N port elements, your netlist
must contain the sequential set of port numbers, 1 through N. For example, in a
design containing 512 ports, you must number each port sequentially, 1 to 512.
Each port has an associated system impedance, zo. If you do not explicitly
specify the system impedance, the default is 50 ohms.
The port element behaves as either a noiseless impedance or a voltage source
in series with the port impedance for all other analyses (DC, AC, or TRAN).
■
You can use this element as a pure terminating resistance or as a voltage or
power source.
■
You can use the RDC, RAC, RHB, RHBAC, and RTRAN values to override the
port impedance value for a particular analysis.
The port element accepts transient waveforms AM, EXP, PULSE, PWL, SFFM,
SIN, and, for signal integrity usage, the PAT source.
Port Element Syntax
Pxxx p n port=portnumber
+ $ **** Voltage or Power Information ********
+ <DC mag> <AC <mag <phase>>> <HBAC <mag <phase>>>
+ <HB <mag <phase <harm <tone <modharm <modtone>>>>>>>
+ <transient_waveform> <TRANFORHB=[0|1]>
+ <DCOPEN=[0|1]>
+ $ **** Source Impedance Information ********
+ <Z0=val> <RDC=val> <RAC=val>
+ <RHBAC=val> <RHB=val> <RTRAN=val>
+ $ **** Power Switch ********
+ <power=[0|1|2|W|dbm]>
Parameter
Description
port=portnumber
The port number. Numbered sequentially beginning
with 1 with no shared port numbers.
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Chapter 6: Testbench Elements
Port Element
Parameter
Description
<DC mag>
DC voltage or power source value.
<AC <mag <phase>>>
AC voltage or power source value.
<HBAC <mag <phase>>>
(HSPICE RF) HBAC voltage or power source value.
<HB <mag <phase <harm
<tone <modharm
<modtone>>>>>>>
(HSPICE RF) HB voltage, current, or power source
value. Multiple HB specifications with different harm,
tone, modharm, and modtone values are allowed.
■
■
■
<transient_waveform>
134
phase is in degrees
harm and tone are indices corresponding to the
tones specified in the .HB statement. Indexing starts
at 1 (corresponding to the first harmonic of a tone).
modtone and modharm specify sources for multitone simulation. A source specifies a tone and a
harmonic, and up to 1 offset tone and harmonic
(modtone for tones and modharm for harmonics).
The signal is then described as:
V(or I) = mag*cos(2*pi*
(harm*tone+modharm*modtone)*t + phase)
(Transient analysis) Voltage or power source waveform.
Any one of waveforms: AM, EXP, PULSE, PWL, SFFM,
or SIN. Multiple transient descriptions are not allowed.
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Port Element
Parameter
Description
<TRANFORHB=[0|1]>
■
0 (default): The transient description is ignored if an
HB value is given or a DC value is given. If no DC or
HB value is given and TRANFORHB=0, then HB
analysis treats the source as a DC source, and the
DC source value is the time=0 value.
■
1: HB analysis uses the transient description if its
value is VMRF, SIN, PULSE, PWL, or LFSR. If the
type is a non-repeating PWL source, then the
time=infinity value is used as a DC analysis source
value. For example, the following statement is treated
as a DC source with value=1 for HB analysis:
v1 1 0 PWL (0 0 1n 1 1u 1)
+ TRANFORHB=1
In contrast, the following statement is a 0V DC
source:
v1 1 0 PWL (0 0 1n 1 1u 1)
+ TRANFORHB=0
The following statement is treated as a periodic
source with a 1us period that uses PWL values:
v1 1 0 PWL (0 0 1n 1 0.999u 1 1u 0) R
+ TRANFORHB=1
To override the global TRANFORHB option, explicitly
set TRANFORHB for a voltage or current source.
DCOPEN
Switch for open DC connection when DC mag is not set.
■
■
<z0=val>
0 (default): P element behaves as an impedance
termination.
1: P element is considered an open circuit in DC
operating point analysis. DCOPEN=1 is mainly used in
.LIN analysis so the P element will not affect the selfbiasing device under test by opening the termination
at the operating point.
(LIN analysis) System impedance used when
converting to a power source, inserted in series with the
voltage source. Currently, this only supports real
impedance.
■
When power=0, z0 defaults to 0.
When power=1, z0 defaults to 50 ohms.
You can also enter zo=val.
■
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Parameter
Description
<RDC=val>
(DC analysis) Series resistance (overrides z0).
<RAC=val>
(AC analysis) Series resistance (overrides z0).
<RHBAC=val>
(HSPICE RF HBAC analysis) Series resistance
(overrides z0).
<RHB=val>
(HSPICE RF HB analysis) Series resistance (overrides
z0).
<RTRAN=val>
(Transient analysis) Series resistance (overrides z0).
<power=[0 | 1 | 2 | W | dbm]> (HSPICE RF) power switch
■
When 0 (default), element treated as a voltage or
current source.
■
When 1 or W, element treated as a power source,
realized as a voltage source with a series
impedance. In this case, the source value is
interpreted as RMS available power in units of Watts.
■
When 2 or dbm, element treated as a power source
in series with the port impedance. Values are in
dbms.
You can use this parameter for Transient analysis if the
power source is either DC or SIN.
Example
For example, the following port element specifications identify a 2-port network
with 50-ohm reference impedances between the “in” and “out” nodes.
P1 in gnd port=1 z0=50
P2 out gnd port=2 z0=50
Computing scattering parameters requires z0 reference impedance values.
The order of the port parameters (in the P-element) determines the order of
the S, Y, and Z parameters. Unlike the .NET command, the .LIN command
does not require you to insert additional sources into the circuit. To calculate
the requested transfer parameters, HSPICE automatically inserts these
sources as needed at the port terminals. You can define an unlimited number of
ports.
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Using the Port Element for Mixed-Mode Measurement
You can use a port element with three terminals as the port element for
measuring the mixed mode S-parameters. Except for the number of external
terminals, the syntax of the port element remains the same. The LIN analysis
function internally sets the necessary drive mode (common/differential) of
these mixed mode port elements. For analyses other than the LIN analysis
(such as DC, AC, TRAN, and so on), the mixed-mode P-element acts as a
differential driver that drives positive nodes with half of their specified voltage
and the negative nodes with a negated half of the specified voltage. Figure 22
shows the block diagram of the mixed mode port element.
P1 (Port element)
n1+
Z0
V+
Z0
Vn2-
n1_ref
Pl nl+ nl- nl_ref Zo=50
Figure 22
Mixed Mode Port Element
Active Elements
This section describes the active elements: diodes and transistors.
Diode Element
Geometric (LEVEL=1) or Non-Geometric (LEVEL=3) form:
Dxxx nplus nminus mname <<AREA=>area> <<PJ=>val>
+ <WP=val> <LP=val> <WM=val> <LM=val> <OFF>
+ <IC=vd> <M=val> <DTEMP=val>
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Dxxx nplus nminus mname <W=width> <L=length> <WP=val>
+ <LP=val> <WM=val> <LM=val> <OFF> <IC=vd> <M=val>
+ <DTEMP=val>
Fowler-Nordheim (LEVEL=2) form:
Dxxx nplus nminus mname <W=val <L=val>> <WP=val>
+ <OFF> <IC=vd> <M=val>
138
Parameter
Description
Dxxx
Diode element name. Must begin with D, followed by up to 1023
alphanumeric characters.
nplus
Positive terminal (anode) node name. The series resistor for the
equivalent circuit is attached to this terminal.
nminus
Negative terminal (cathode) node name.
mname
Diode model name reference.
AREA
Area of the diode (unitless for LEVEL=1 diode, and square meters for
LEVEL=3 diode). This affects saturation currents, capacitances, and
resistances (diode model parameters are IK, IKR, JS, CJO, and RS).
The SCALE option does not affect the area factor for the LEVEL=1
diode. Default=1.0. Overrides AREA from the diode model. If you do not
specify the AREA, HSPICE or HSPICE RF calculates it from the width
and length.
PJ
Periphery of junction (unitless for LEVEL=1 diode, and meters for
LEVEL=3 diode). Overrides PJ from the diode model. If you do not
specify PJ, HSPICE or HSPICE RF calculates it from the width and
length specifications.
WP
Width of polysilicon capacitor, in meters (for LEVEL=3 diode only).
Overrides WP in the diode model. Default=0.0.
LP
Length of polysilicon capacitor, in meters (for LEVEL=3 diode only).
Overrides LP in the diode model. Default=0.0.
WM
Width of metal capacitor, in meters (for LEVEL=3 diode only). Overrides
WM in the diode model. Default=0.0.
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Parameter
Description
LM
Length of metal capacitor, in meters (for LEVEL=3 diode only).
Overrides LM in the diode model. Default=0.0.
OFF
Sets the initial condition for this element to OFF, in DC analysis.
Default=ON.
IC=vd
Initial voltage, across the diode element. Use this value when you
specify the UIC option in the .TRAN statement. The .IC statement
overrides this value.
M
Multiplier, to simulate multiple diodes in parallel. The M setting affects
all currents, capacitances, and resistances. Default=1.
DTEMP
The difference between the element temperature and the circuit
temperature, in degrees Celsius. Default=0.0.
W
Width of the diode, in meters (LEVEL=3 diode model only)
L
Length of the diode, in meters (LEVEL=3 diode model only)
You must specify two nodes and a model name. If you specify other
parameters, the nodes and model name must be first and the other parameters
can appear in any order.
Example 1
The D1 diode, with anode and cathode, connects to nodes 1 and 2. Diode1
specifies the diode model.
D1 1 2 diode1
Example 2
The Dprot diode, with anode and cathode, connects to both the output node
and ground, references the firstd diode model, and specifies an area of 10
(unitless for LEVEL=1 model). The initial condition has the diode OFF.
Dprot output gnd firstd 10 OFF
Example 3
The Ddrive diode, with anode and cathode, connects to the driver and output
nodes. The width and length are 500 microns. This diode references the
model_d diode model.
Ddrive driver output model_d W=5e-4 L=5e-4 IC=0.2
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Bipolar Junction Transistor (BJT) Element
Qxxx nc nb ne <ns> mname <area> <OFF>
+ <IC=vbeval,vceval> <M=val> <DTEMP=val>
Qxxx nc nb ne <ns> mname <AREA=area> <AREAB=val>
+ <AREAC=val> <OFF> <VBE=vbeval> <VCE=vceval>
+ <M=val> <DTEMP=val>
Parameter
Description
Qxxx
BJT element name. Must begin with Q, then up to 1023 alphanumeric
characters.
nc
Collector terminal node name.
nb
Base terminal node name.
ne
Emitter terminal node name.
ns
Substrate terminal node name, which is optional. You can also use the
BULK parameter to set this name in the BJT model.
mname
BJT model name reference.
area,
AREA=area
Emitter area multiplying factor, which affects currents, resistances, and
capacitances. Default=1.0.
OFF
Sets initial condition for this element to OFF, in DC analysis.
Default=ON.
IC=vbeval,
Initial internal base-emitter voltage (vbeval) and collector-emitter
vceval, VBE, voltage (vceval). HSPICE or HSPICE RF uses this value when
VCE
the .TRAN statement includes UIC. The .IC statement overrides it.
140
M
Multiplier, to simulate multiple BJTs in parallel. The M setting affects all
currents, capacitances, and resistances. Default=1.
DTEMP
The difference between the element temperature and the circuit
temperature, in degrees Celsius. Default=0.0.
AREAB
Base area multiplying factor, which affects currents, resistances, and
capacitances. Default=AREA.
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Parameter
Description
AREAC
Collector area multiplying factor, which affects currents, resistances,
and capacitances. Default=AREA.
The only required fields are the collector, base, and emitter nodes, and the
model name. The nodes and model name must precede other fields in the
netlist.
Example 1
In the Q1 BJT element below:
Q1 1 2 3 model_1
■
The collector connects to node 1.
■
The base connects to node 2.
■
The emitter connects to node 3.
■
model_1 references the BJT model.
Example 2
In the following Qopamp1 BJT element:
Qopamp1 c1 b3 e2 s 1stagepnp AREA=1.5 AREAB=2.5
AREAC=3.0
■
The collector connects to the c1 node.
■
The base connects to the b3 node.
■
The emitter connects to the e2 node.
■
The substrate connects to the s node.
■
1stagepnp references the BJT model.
■
The AREA area factor is 1.5.
■
The AREAB area factor is 2.5.
■
The AREAC area factor is 3.0.
Example 3
In the Qdrive BJT element below:
Qdrive driver in output model_npn 0.1
■
The collector connects to the driver node.
■
The base connects to the in node.
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■
The emitter connects to the output node.
■
model_npn references the BJT model.
■
The area factor is 0.1.
JFETs and MESFETs
Jxxx nd ng ns <nb> mname <<<AREA>=area | <W=val>
+ <L=val>> <OFF> <IC=vdsval,vgsval> <M=val>
+ <DTEMP=val>
Jxxx nd ng ns <nb> mname <<<AREA>=area> | <W=val>
+ <L=val>> <OFF> <VDS=vdsval> <VGS=vgsval>
+ <M=val> <DTEMP=val>
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Parameter
Description
Jxxx
JFET or MESFET element name. Must begin with J, followed by up
to 1023 alphanumeric characters.
nd
Drain terminal node name
ng
Gate terminal node name
ns
Source terminal node name
nb
Bulk terminal node name, which is optional.
mname
JFET or MESFET model name reference
area,
AREA=area
Area multiplying factor that affects the BETA, RD, RS, IS, CGS, and
CGD model parameters. Default=1.0, in units of square meters.
W
FET gate width in meters
L
FET gate length in meters
OFF
Sets initial condition to OFF for this element, in DC analysis.
Default=ON.
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Parameter
Description
IC=vdsval,
vgsval, VDS,
VGS
Initial internal drain-source voltage (vdsval) and gate-source voltage
(vgsval). Use this argument when the .TRAN statement contains
UIC. The .IC statement overrides it.
M
Multiplier to simulate multiple JFETs or MESFETs in parallel. The M
setting affects all currents, capacitances, and resistances.
Default=1.
DTEMP
The difference between the element temperature and the circuit
temperature, in degrees Celsius. Default=0.0.
Only drain, gate, and source nodes, and model name fields are required. Node
and model names must precede other fields.
Example 1
In the J1 JFET element below:
J1 1 2 3 model_1
■
The drain connects to node 1.
■
The source connects to node 2.
■
The gate connects to node 3.
■
model_1 references the JFET model.
Example 2
In the following Jopamp1 JFET element:
Jopamp1 d1 g3 s2 b 1stage AREA=100u
■
The drain connects to the d1 node.
■
The source connects to the g3 node.
■
The gate connects to the s2 node.
■
1stage references the JFET model.
■
The area is 100 microns.
Example 3
In the Jdrive JFET element below:
Jdrive driver in output model_jfet W=10u L=10u
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■
The drain connects to the driver node.
■
The source connects to the in node.
■
The gate connects to the output node.
■
model_jfet references the JFET model.
■
The width is 10 microns.
■
The length is 10 microns.
MOSFETs
Mxxx nd ng ns <nb> mname <<L=>length> <<W=>width>
+ <AD=val> AS=val> <PD=val> <PS=val>
+ <NRD=val> <NRS=val> <RDC=val> <RSC=val> <OFF>
+ <IC=vds,vgs,vbs> <M=val> <DTEMP=val>
+ <GEO=val> <DELVTO=val>
.OPTION WL
Mxxx nd ng ns <nb> mname <width> <length> <other_options...>
144
Parameter
Description
Mxxx
MOSFET element name. Must begin with M, followed by up to 1023
alphanumeric characters.
nd
Drain terminal node name.
ng
Gate terminal node name.
ns
Source terminal node name.
nb
Bulk terminal node name, which is optional. To set this argument in the
MOSFET model, use the BULK parameter.
mname
MOSFET model name reference
L
MOSFET channel length, in meters. This parameter overrides
.OPTION DEFL, with a maximum value of 0.1m. Default=DEFL.
W
MOSFET channel width, in meters. This parameter overrides .OPTION
DEFW. Default=DEFW.
AD
Drain diffusion area. Overrides .OPTION DEFAD. Default=DEFAD, if
you set the ACM=0 model parameter.
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Parameter
Description
AS
Source diffusion area. Overrides .OPTION DEFAS. Default=DEFAS, if
you set the ACM=0 model parameter.
PD
Perimeter of drain junction, including channel edge. Overrides
.OPTION DEFPD. Default=DEFAD, if you set the ACM=0 or 1 model
parameter. Default=0.0, if you set ACM=2 or 3.
PS
Perimeter of source junction, including channel edge. Overrides
.OPTION DEFPS. Default=DEFAS, if you set the ACM=0 or 1 model
parameter. Default=0.0, if you set ACM=2 or 3.
NRD
Number of squares of drain diffusion for resistance calculations.
Overrides .OPTION DEFNRD. Default=DEFNRD, if you set ACM=0 or
1 model parameter. Default=0.0, if you set ACM=2 or 3.
NRS
Number of squares of source diffusion for resistance calculations.
Overrides .OPTION DEFNRS. Default=DEFNRS when you set the
MOSFET model parameter ACM=0 or 1. Default=0.0, when you set
ACM=2 or 3.
RDC
Additional drain resistance due to contact resistance, in units of ohms.
This value overrides the RDC setting in the MOSFET model
specification. Default=0.0.
RSC
Additional source resistance due to contact resistance, in units of
ohms. This value overrides the RSC setting in the MOSFET model
specification. Default=0.0.
OFF
Sets initial condition for this element to OFF, in DC analysis.
Default=ON. This command does not work for depletion devices.
IC=vds, vgs,
vbs
Initial voltage across external drain and source (vds), gate and source
(vgs), and bulk and source terminals (vbs). Use these arguments
with .TRAN UIC. .IC statements override these values.
M
Multiplier, to simulate multiple MOSFETs in parallel. Affects all channel
widths, diode leakages, capacitances, and resistances. Default=1.
DTEMP
The difference between the element temperature and the circuit
temperature, in degrees Celsius. Default=0.0.
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Parameter
Description
GEO
Source/drain sharing selector for a MOSFET model parameter value of
ACM=3. Default=0.0.
DELVTO
Zero-bias threshold voltage shift. Default=0.0.
The only required fields are the drain, gate and source nodes, and the model
name. The nodes and model name must precede other fields in the netlist. If
you did not specify a label, use the second syntax with the .OPTION WL
statement, to exchange the width and length options.
Example
In the following M1 MOSFET element:
M1 1 2 3 model_1
■
The drain connects to node 1.
■
The gate connects to node 2.
■
The source connects to node 3.
■
model_1 references the MOSFET model.
In the following Mopamp1 MOSFET element:
Mopamp1 d1 g3 s2 b 1stage L=2u W=10u
■
The drain connects to the d1 node.
■
The gate connects to the g3 node.
■
The source connects to the s2 node.
■
1stage references the MOSFET model.
■
The length of the gate is 2 microns.
■
The width of the gate is 10 microns.
In the following Mdrive MOSFET element:
Mdrive driver in output bsim3v3 W=3u L=0.25u DTEMP=4.0
146
■
The drain connects to the driver node.
■
The gate connects to the in node.
■
The source connects to the output node.
■
bsim3v3 references the MOSFET model.
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Steady-State Voltage and Current Sources
■
The length of the gate is 3 microns.
■
The width of the gate is 0.25 microns.
■
The device temperature is 4 degrees Celsius higher than the circuit
temperature.
Steady-State Voltage and Current Sources
The I (current source) and V (voltage source) elements include extensions that
allow you to use them as sources of steady-state sinusoidal signals for HB and
HBAC analyses. When you use a power parameter to specify the available
power, you can also use these elements as power sources.
For a general description of the I and V elements, see Power Sources in the
HSPICE User Guide: Simulation and Analysis.
I and V Element Syntax
Vxxx p n
+ $ **** Voltage or Power Information ********
+ <<dc> mag> <ac <mag <phase>>> <HBAC <mag <phase>>>
+<SNAC <mag <phase>>>
+ <hb <mag <phase <harm <tone <modharm <modtone>>>>>>>
+ <transient waveform> <TRANFORHB=[1|0]>
+ $ **** Power Switch ********
+ <power=[0 | 1 | W | dbm]> <z0=val> <rdc=val> <rac=val>
+ <RHBAC=val> <rhb=val> <rtran=val>
Ixxx p n
+ $ **** Current or Power Information ********
+ <<dc> mag> <ac <mag <phase>>> <HBAC <mag <phase>>>
+<SNAC <mag <phase>>>
+ <hb <mag <phase <harm <tone <modharm <modtone>>>>>>>
+ <transient waveform> <TRANFORHB=[1|0]>
+ $ **** Power Switch ********
+ <power=[0 | 1 | W | dbm]> <z0=val> <rdc=val> <rac=val>
+ <RHBAC=val> <rhb=val> <rtran=val>
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Parameter
Description
<<dc> mag>
DC voltage or power source value. You don’t need to
specify DC explicitly (default=0).
<ac <mag <phase>>>
AC voltage or power source value.
<HBAC <mag <phase>>> (HSPICE RF) HBAC voltage or power source value.
<SNAC <mag <phase>>> (HSPICE RF) SNAC voltage or power source value.
<hb <mag <phase <harm (HSPICE RF) HB voltage, current, or power source value.
<tone <modharm
Multiple HB specifications with different harm, tone,
<modtone>>>>>>>
modharm, and modtone values are allowed.
■
■
■
<transient waveform>
phase is in degrees
harm and tone are indices corresponding to the tones
specified in the .HB statement. Indexing starts at 1
(corresponding to the first harmonic of a tone).
modtone and modharm specify sources for multi-tone
simulation. A source specifies a tone and a harmonic,
and up to 1 offset tone and harmonic (modtone for
tones and modharm for harmonics). The signal is then
described as:
V(or I) = mag*cos(2*pi*
(harm*tone+modharm*modtone)*t + phase)
(Transient analysis) Any one of waveforms: AM, EXP,
PULSE, PWL, SFFM, or SIN. Multiple transient
descriptions are not allowed.
<power=[0 | 1 | W | dbm]> (HSPICE RF) Power Switch
■
When 0 (default), element treated as a voltage or
current source.
■
When 1 or W, element treated as a power source,
realized as a voltage source with a series impedance.
In this case, the source value is interpreted as RMS
available power in units of Watts.
■
When dbm, element treated as a power source in
series with the port impedance. Values are in dbms.
You can use this parameter for Transient analysis if the
power source is either DC or SIN.
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Parameter
Description
<z0=val>
(LIN analysis) System impedance used when converting
to a power source, inserted in series with the voltage
source. Currently, this only supports real impedance.
■
When power=0, z0 defaults to 0.
When power=1, z0 defaults to 50 ohms.
You can also enter zo=val.
■
<rdc=val>
(DC analysis) Series resistance (overrides z0).
<rac=val>
(AC analysis) Series resistance (overrides z0).
<RHBAC=val>
(HSPICE RF HBAC analysis) Series resistance (overrides
z0).
<rhb=val>
(HSPICE RF HB analysis) Series resistance (overrides
z0).
<rtran=val>
(Transient analysis) Series resistance (overrides z0).
<TRANFORHB=[0|1]>
■
0 (default): The transient description is ignored if an HB
value is given or a DC value is given. If no DC or HB
value is given and TRANFORHB=0, then HB treats the
source as a DC source, and the DC source value is the
time=0 value.
■
1: HB analysis uses the transient description if its value
is VMRF, SIN, PULSE, PWL, or LFSR. If the type is a
non-repeating PWL source, then the time=infinity value
is used as a DC source value. For example, the
following statement is treated as a DC source with
value=1 for HB:
v1 1 0 PWL (0 0 1n 1 1u 1) TRANFORHB=1
In contrast, the following statement is a 0V DC source:
v1 1 0 PWL (0 0 1n 1 1u 1) TRANFORHB=0
The following statement is treated as a periodic source
with a 1us period that uses PWL values:
v1 1 0 PWL (0 0 1n 1 0.999u 1 1u 0) R
TRANFORHB=1
To override the global TRANFORHB option, explicitly set
TRANFORHB for a V/I source.
Example 1
This example shows an HB source for a single tone analysis:
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.hb tones=100MHz harms=7
I1 1 2 dc=1mA hb 3mA 0. 1 1
I1 is a current source with a the following time-domain description:
I1=1mA + 3mA*cos(2*pi*1.e8*t)
Example 2
This example shows HB sources used for a two-tone analysis:
.hb tones=1.e9 1.1e9 intmodmax=5
Vin lo 0 dc=0. hb 1.5 90 1 1
Vrf rf 0 dc=0. hb 0.2 0 1 2
These sources have the following time-domain descriptions:
Vin=1.5*cos(2*pi*1.e9*t - 90*pi/180) V
Vrf = 0.2*cos(2*pi*1.1e9*t) V
Example 3
The following HB source uses a modtone and modharms:
.hb tones=2.e9 1.9e9 harms=5 5
Vm input gnd dc=0.5 hb 0.2 0. 1 1 -1 2
Vm has the following time-domain description:
Vm = 0.5 + cos(2*pi*1.e8*t)
Example 4
This example uses an HB source specified with a SIN source and
HBTRANINIT.
.hb tone=1.e8 harms=7
Vt 1 2 SIN(0.1 1.0 2.e8 0. 0. 90) tranforhb=1
Vt is converted to the following HB source:
Vt 1 2 dc=0.1 hb 1.0 0.0 2 1
Example 5
This example shows a power source (the units are Watts).
.hb tones=1.1e9 harms=9
Pt Input Gnd power=1 Z0=50. 1m 0. 1 1
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Pt delivers 1 mW of power through a 50 ohm impedance.
Steady-State HB Sources
The fundamental frequencies used with harmonic balance analysis are
specified with the .HB TONES command. These frequencies can then be
referenced by their integer indices when specifying steady-state signal sources.
For example, the .HB specification given by the following line:
.HB TONES=1900MEG,1910MEG INTMODMAX=5
This specifies two fundamental frequencies: f [ tone = 1 ] = 1.9GHz and
f [ tone = 2 ] = 1.91GHz . Their mixing product at 10 MHz can then be referenced
using indices as f [ 2 ] – f [ 1 ] , while their 3rd order intermodulation product at 1.89
GHz can be referenced as 2f [ 1 ] – f [ 2 ] .
Steady-state voltage and current sources are identified with the HB keyword
according to
<HB <mag <phase <harm <tone <modharm <modtone>>>>>>>
The source is mathematically equivalent to a cosine signal source that follows
the equation
A cos ( ωt + φ)
where
A = mag
ω = 2π harm ⋅ f [ tone ] + modharm ⋅ f [ mo dtone ]
π
φ = --------- ⋅ phase
180
Values for tone and modtone (an optional modulating tone) must be nonnegative integers that specify index values for the frequencies specified with
the .HB TONES command. Values for harm (harmonic) and modharm
(modulating tone harmonic) must be integers (negative values are OK) that
specify harmonic indices.
Example 1
The following example is a 1.0 Volt (peak) steady-state cosine voltage source,
which is at the fundamental HB frequency with zero phase and with a zero volt
DC value:
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Vsrc
in
gnd
DC
0
HB
1.0
0
1
1
Example 2
The following example is a steady-state cosine power source with 1.0mW
available power, which is implemented with a Norton equivalent circuit and a 50
ohm input impedance:
Isrc
in
gnd
HB
1.0e-3
0
1
1
power=1 z0=50
Example 3
Five series voltage sources sum to produce a stimulus of five equally spaced
frequencies at and above 2.44 GHz using modharm and modtone parameters.
These are commensurate tones (an integer relation exists); therefore, you only
need to specify two tones when invoking the HB analysis.
.param Vin=1.0
.param f0=2440MEG
.param deltaf=312.5K
.param fcenter='f0 + 2.0*deltaf'
Vrfa
in
ina
HB
'Vin'
0
1
GHz
Vrfb
ina
inb
HB
'Vin'
0
1
2.4403125
GHz
Vrfc
inb
inc
HB
'Vin'
0
1
2.440
GHz
Vrfd
inc
ind
HB
'Vin'
0
1
2.4409375
GHz
Vrfe
ind
gnd
HB
'Vin'
0
1
GHz
.HB tones=fcenter,deltaf intmodmax=5
1
$ 2.440625
1
-1
2
$
1
-2
2
$
1
+1
2
$
1
+2
2
$ 2.44125
Phase Differences Between HB and SIN Sources
The HB steady-state cosine source has a phase variation compared to the
TRAN time-domain SIN source. The SIN source (with no offset, delay or
damping) follows the equation:
Equation 7
A sin ( ωt + φ)
while the HB sources follow
Equation 8
A cos ( ωt + φ)
In order for the two sources to yield identical results it is necessary to align
them by setting their phase values accordingly using:
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Equation 9
A cos ( ωt + φ) = A sin ( ωt + φ + 90° )
Equation 10
A sin ( ωt + φ) = A cos ( ωt + φ – 90° )
To specify sources with matching phase for HB and TRAN analysis, use a
convention similar to:
** Example #1 with equivalent HB and SIN sources
** SIN source is given +90 phase shift
.param freq1=2400MEG Vin=1.0
Vsrc in gnd DC 0 HB 'Vin' 0 1 1 SIN(0 'Vin' 'freq1' 0 0 90)
.HB tones=freq1 intmodmax=7
** Example #2 with equivalent HB and SIN sources
** HB source is given -90 phase shift to align with SIN
.param freq1=2400MEG Vin=1.0
Vsrc in gnd DC 0 HB 'Vin' -90 1 1 SIN(0 'Vin' 'freq1' 0)
.HB tones=freq1 intmodmax=7
** Example #3 with equivalent .HB and .TRAN sources
** SIN source is activated for HB using "TRANFORHB"
.param freq1=2400MEG Vin=1.0
Vsrc in gnd DC 0 SIN(0 'Vin' 'freq1' 0) TRANFORHB=1
.HB tones=freq1 intmodmax=7
Behavioral Noise Sources
In HSPICE RF, you can use the G-element to specify noise sources. Frequency
domain noise analyses (.NOISE, .HBNOISE, and .PHASENOISE) take these
noise sources into account.
You can attach noise sources to behavioral models. For example, you can use
a G-element with the VCCAP parameter to model a varactor, which includes a
noise model. You can also simulate effects such as substrate noise, including
its effect on oscillator phase noise. You can also use this G element syntax to
simulate behavioral descriptions of substrate noise during any frequency
domain noise analysis, which includes phase noise analysis. For example,
gname node1 node2 noise=’noise_equation’
gname node1 node2 node3 node4 noise=’noise_equation’
The first line creates a simple two-terminal current noise source, whose value
is described in A2/(Hz). The output noise generated from this noise source is:
noise_equation*H
Where H is the transfer function from the terminal pair (node1,node2) to the
circuit output, where HSPICE RF measures the output noise.
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The second line produces a noise source correlation between the
(node1,node2) and (node3,node4) terminal pairs. The resulting output noise is
calculated as noise_equation*sqrt(H1*H2*); where,
■
H1 is the transfer function from (node1,node2) to the output
■
H2 is the transfer function from (node3,node4) to the output.
The noise_equation expression can involve node voltages and currents through
voltage sources.
For the PAC phasenoise simulation to evaluate the frequency-dependent noise,
the frequency-dependent noise factor in the phasenoise must be expressed in
between the parentheses. For example:
gname node1 node2 noise = '(frequency_dependent_noise)*
bias_dependent_noise'
This is only true when the total noise can be expressed in this form and when
the frequency-dependent noise can be evaluated in the PAC phasenoise
simulation. You can also input the behavioral noise source as a noise table with
the help of predefined Table() function. The Table() function takes two formats:
■
Noise table can be input directly through the Table() function. For example:
gname node1 node2 noise = 'Table(arg1,f1,v1,f2,v2,......)'
■
The f1,v1,f2,v2,..... parameters describe the noise table. When arg1 == f1,
the function returns v1. The arg1 can be an expression of either HERTZ,
bias, or both. For example, arg1 = 'HERTZ * 1.0E+3'.
■
The noise table can be input through a .DATA structure:
.DATA d1
+ x y
+ f1 v1
+ f2 v2
.ENDDATA
gname node1 node2 noise = 'TABLE(arg1,d1)'
The x, y parameters in the DATA structure are two placeholder strings that can
be set to whatever you prefer even if they are in conflict with other parameters
in the netlist. The arg1 parameter can be an expression of HERTZ and bias.
When arg1 == f2, the function will return v2.
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Using Noise Analysis Results as Input Noise Sources
SN phase noise and phase noise analyses can output simulation results as
ASCII data in *.printsnpn0 files for SNOSC and SNNOISE. By extending
the E and G voltage-controlled source syntax, the phase noise data in ASCII
phase noise files can used as input for specifying behavioral noise sources
Usage Model
The syntax for the voltage controlled voltage (E) or current (G) source is as
follows:
Exxx node1 node2 noisefile='filename' [mname='measname']
Gxxx node1 node2 noisefile='filename' [mname='measname']
Where,
noisefile='filename' is the name of the ASCII phase noise data file. The
file name is typically designated as 'design.printsnpn0', for a .SNOSC phase
noise analysis or .SNNOISE analysis. But it also supports .PHASENOISE,
.HBNOISE, .NOISE, and .ACPHASENOISE outputs.
mname='measname' is used to select the appropriate noise measurement
name to be taken from the *.printpn0 file.
measname can be one of the following:
■
NLP_L(f) - selects the nlp_L(f) phase noise data in units of dBc/Hz
■
PAC_L(f) - selects the pac_l(f) phase noise data in units of dBc/Hz
■
BPN_L(f) - selects the bpn_l(f) phase noise data in units of dBc/Hz
■
ONOISE - selects the onoise data based on .SNNOISE analysis
Power Supply Current and Voltage Noise Sources
You can implement the power supply noise source with G and E elements. The
G-element for the current noise source and the E-element for the voltage noise
source. As noise elements, they are two-terminal elements that represent a
noise source connected between two specified nodes.
Syntax
Expression form
Gxxx node1 node2 noise=‘expression’
Exxx node1 node2 noise=‘expression’
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The G noise element represents a noise current source and the E noise
element represents a noise voltage source. The xxx parameter can be set with
a value up to 1024 characters. The node1 and node2 are the positive and
negative nodes that connect to the noise source. The noise expression can
contain the bias, frequency, or other parameters.
Data form
Gxxx node1 node2 noise data=dataname
Exxx node1 node2 noise data=dataname
.data dataname
+ pname1 pname2
+ freq1 noise1
+ freq2 noise2
+ ...
.enddata
The data form defines a basic frequency-noise table. The .DATA statement
contains two parameters: frequency and noise to specify the noise value at
each frequency point. The unit for frequency is hertz, and the unit for noise is
A2/Hz (for G current noise source) or V2/Hz (for E voltage noise source).
Example
The following netlist shows a 1000 ohm resistor (g1) using a G element. The
g1noise element, placed in parallel with the g1 resistor, delivers the thermal
noise expected from a resistor. The r1 resistor is included for comparison: The
noise due to r1 should be the same as the noise due to g1noise.
* Resistor implemented using g-element
v1 1 0 1
r1 1 2 1k
g1 1 2 cur='v(1,2)*0.001'
g1noise 1 2
+ noise='4*1.3806266e-23*(TEMPER+273.15)*0.001'
rout 2 0 1meg
.ac lin 1 100 100
.noise v(2) v1 1
.end
Function Approximations for Distributed Devices
High-order rational function approximations constructed for distributed devices
used at RF frequencies are obtained in the pole-residue form (also known as
Foster canonical form). The popular method of recursive convolution also uses
this form.
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HSPICE supports the pole-residue form for its frequency-dependent controlled
sources (G and E elements). You can enter the pole-residue form directly
without first converting to another form.
Foster Pole-Residue Form for Transconductance or Gain
The Foster pole-residue form for transconductance G(s) or gain E(s) has the
form:
N
Equation 11
G ( s ) = k0 + k1 s +
⎛ A
A∗ ⎞
i
i
+ ---------------⎟
∑ ⎜⎝ -----------s – pi s – p ∗⎠
i
i=1
Where,
■
k0, k1 are real constants
■
residues Ai and poles pi are complex numbers (or real as a special case of
complex
■
asterisk (*) denotes the expression's complex conjugate
Advantages of Foster Form Modeling
The advantages of Foster canonical form modeling are:
■
models high-order systems. It can theoretically model systems having
infinite poles without numerical problems.
■
equivalent to Laplace and Pole-zero models
■
popular method of recursive convolution uses this form.
G and E-element Syntax
Transconductance G(s) form
Gxxx n+ n+ (Re{A1},
+ (Re{A2},
+ (Re{A3},
+ ...
FOSTER in+ in- k0
Im{A1})/ (Re{p1},
Im{A2})/ (Re{p2},
Im{A3})/ (Re{p3},
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Im{p1})
Im{p2})
Im{p3})
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Gain E(s) form
Exxx n+ n+ (Re{A1},
+ (Re{A2},
+ (Re{A3},
+ ...
FOSTER in+ in- k0
Im{A1})/ (Re{p1},
Im{A2})/ (Re{p2},
Im{A3})/ (Re{p3},
k1
Im{p1})
Im{p2})
Im{p3})
In the above syntax, parenthesis, commas, and slashes are separators—they
have the same meaning as a space. A pole-residue pair is represented by four
numbers (real and imaginary part of the residue, then real and imaginary part
of the pole).
You must make sure that Re[pi]<0; otherwise, the simulations will certainly
diverge. Also, it is a good idea to assure passivity of the model (for an N-port
admittance matrix Y, Re{Y} should be positive-definite), or the simulation is
likely to diverge).
Example
To represent a G(s) in the form,
Equation 12
0.0008 - + -------------------------------------------------------------------( 0.001 – j0.006 )
s + ----------------------------+
10
8
10
s + 1 × 10
s – ( – 1 × 10 + j1.8 × 10 )
(
0.001
+
j0.006
)
-------------------------------------------------------------------8
10
s – ( – 1 × 10 – j1.8 × 10 )
G ( s ) = 0.001 + 1 × 10
– 12
You would input:
G1 1 0 FOSTER 2 0 0.001 1e-12
+(0.0004, 0)/(-1e10, 0) (0.001, -0.006)/(-1e8, 1.8e10)
Note:
In the case of a real poles, half the residue value is entered, because it's
essentially applied twice. In the above example, the first pole-residue pair is
real, but we still write it as “A1/(s-p1)+A1/(s-p1)”; therefore, 0.0004 is
entered rather than 0.0008.
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Complex Signal Sources and Stimuli
To predict radio-frequency integrated circuit (RFIC) performance, some
analyses require simulations that use representative RF signal sources. Among
the representative sources available in HSPICE RF is the complex modulated
RF source. Also known as the Vector Modulated source, it allows digital
modulation of an RF carrier using in-phase and quadrature components
created from a binary data stream.
Vector-Modulated RF Source
Digital RF waveforms are typically constructed by modulating an RF carrier with
in-phase (I) and quadrature (Q) components. In HSPICE RF, this is
accomplished using the Vector Modulated RF (VMRF) signal source.
The VMRF signal source function is supported both for independent voltage
and current sources (V and I elements), and with controlled sources (E, F, G,
and H elements).
■
When used with independent sources, a baseband data stream can be input
in binary or hexadecimal format, and the scheme used to divide the data into
I and Q signals can be specified.
■
With controlled VMRF sources, the modulating I and Q signals can be
separately specified with other signal sources (such as a PWL source) and
then used as control inputs into the VMRF source.
Implementation
The VMRF source is a mathematical implementation of the following block
diagram:
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l(t)
cos(wt)
Data in
Serial to
Parallel
S(t)
sin(wt)
Q(t)
The following equation calculates the time and frequency domain stimuli from
the quadrature modulated signal sources:
Equation 13
s ( t ) = I ( t ) cos ( 2πf c t + φ0 ) – Q ( t ) sin ( 2πfct + φ0 )
The discrete ideal I (in-phase) and Q (quadrature) signal components are
digital. Discrete values allow uniform scaling of the overall signal. HSPICE RF
generates data streams for the I and Q signals based on interpreting the data
string, breaking the data string into a binary representation, and then using the
bit pairs to assign values for the I and Q data streams.
For BPSK (binary phase shift keying) modulation, the discrete signals are
scaled so that
160
2
2
I + Q = 1:
Data In
I Data
Q Data
0
–1
------2
–1
------2
1
1
------2
1
------2
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For QPSK (quadrature phase shift keying) modulation, the data stream is
broken into bit pairs to form the correct I and Q values. This function is
represented as the serial to parallel converter:
Data In
I Data
Q Data
00
–1
------2
–1
------2
01
–1
------2
1
------2
10
1
------2
–1
------2
11
1
------2
1
------2
To generate a continuous-time waveform, the VMRF source takes the resulting
digital I and Q data streams and passes them through ideal filters. Rectangular
and Nyquist (raised-cosine) filter options are available. The output waveforms
are therefore band-limited according to the specified data rate.
Voltage and Current Source Elements
The V and I elements can include VMRF signal sources that you can use to
generate BPSK and QPSK waveforms.
V and I Element Syntax
Vxxx n+ n- VMRF <(> AMP=sa FREQ=fc PHASE=ph MOD=MOD
+ FILTER=FIL FILCOEF=filpar RATE=Rb BITSTREAM=data
+ <TRANFORHB=0/1> <)>
Ixxx n+ n- VMRF <(> AMP=sa FREQ=fc PHASE=ph MOD=MOD
+ FILTER=FIL FILCOEF=filpar RATE=Rb BITSTREAM=data
+ <TRANFORHB=0/1> <)>
Parameter
Description
Vxxx
Independent voltage source.
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Parameter
Description
Ixxx
Independent current source.
n+ n-
Positive and negative controlled source connecting nodes.
VMRF
Keyword that identifies and activates the Vector Modulated RF signal
source.
AMP
Signal amplitude (in volts or amps).
FREQ
Carrier frequency in hertz. Set fc=0.0 to generate baseband I/Q
signals. For harmonic balance analysis, the frequency spacing must
coincide with the .HB TONES settings.
PHASE
Carrier phase (in degrees). If fc=0.0,
■
■
■
MOD
One of the following keywords identifies the modulation method used
to convert a digital stream of information to I(t) and Q(t) variations:
■
■
FILTER
ph=0 and baseband I(t) is generated
ph=-90 and baseband q(t) is generated
Otherwise, s ( t ) = I ( t ) cos ( φ0 ) – Q ( t ) sin ( φ0 )
BPSK (binary phase shift keying)
QPSK (quadrature phase shift keying)
One of the following keywords identifies the method used to filter the
I and Q signals before modulating the RF carrier signal:
■
■
COS (raised cosine Nyquist filter)
RECT (rectangular filtering)
FILCOEF
Filter parameter for the COS filter: 0 ≤filpar ≤1
RATE
Bit rate for modulation (bits per second).
■
For BPSK modulation, the data rate and the symbol rate are the
same.
■
For QPSK modulation, the symbol rate is half the data rate.
The Rb value must be greater than zero.
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Parameter
Description
BITSTREAM A binary (b) or hexadecimal (h) string that represents an input data
stream.
Valid data string characters are:
■
0 or 1 for binary (b) mode.
0, 1, 2, 3, 4, 5, 6, 7, 8, 9, A, B, C, D, E, F, a, b, c, d, e, or f for
hexadecimal (h) mode.
For example:
■
■
■
01010011b (binary)
0F647A30E9h (hexadecimal)
You can also use the standard V source and I source options for non-transient
simulations (such as DC=val and AC=mag,ph) a with the VMRF source.
Example
BITSTREAM=01010010011100b
data
1/dr
BPSK I and Q Signals
.707
1/dr
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QPSK I Signal
.707
1/dr
QPSK Q Signal
.707
1/dr
The Rb parameter represents the data rate. The associated symbol rate
represents how fast the I and Q data streams change. The period for each bit of
data is:
Equation 14
1T b = ----Rb
The symbol rate depends on whether you select BPSK or QPSK modulation:
■
For BPSK, the symbol rate is the same as the data rate:
S
■
R
BPSK
= Rb
For QPSK modulation, two bits are used to create each symbol so the
symbol rate is half the data rate.
R
S
QPSK
R
= -----b2
The period for each symbol is computed as:
Equation 15
1T s = ---Rs
This value is necessary for establishing the characteristics of Nyquist filters.
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The following equation calculates the raised cosine (COS) filter response:
1–α
f ≤-----------2T s
Equation 16
H rc ( f ) =
∫
Ts
0
2 πT s
1–α
T s cos -------- ⎛ f – ------------⎞
2α ⎝
2T s ⎠
1–α
1+α
------------ ≤ f ≤-----------2T s
2T s
1+a
f > -----------2T s
The VMRF signal source is designed primarily for TRAN and HB analyses, and
can generate baseband signals. You can also specify DC and AC values as
with any other HSPICE signal source:
■
In DC analysis, the VMRF source is a constant DC source.
■
In AC analysis, the source is a short or an open, unless you specify an AC
value.
■
In HB analysis, you must specify .OPTION TRANFORHB on the source
statement line. The TRANFORHB option supports the VMRF signal source as
well as the SIN, PULSE, and PWL sources.
The VMRF quadrature signal source typically involves an HF carrier signal that
is modulated with a baseband signal on a much different time scale. You must
set source and simulation control parameters appropriately to avoid timeconsuming simulations in both the time and frequency domains.
E, F, G, and H Element Statements
For E, F, G, and H elements, you can use the VMRF function to modulate I(t)
and Q(t) signals with a RF carrier signal. The I and Q signal are driven by PWL
sources that might be generated by an external tool, such as MATLAB. The
PWL source accepts a text file containing time and voltage (or current) pairs.
When the VMRF function is used with controlled sources, it is anticipated that
the in-phase (I) and quadrature (Q) signals are not digital, but continuous-time
analog signals. The VMRF function therefore includes no filtering, and merely
serves to create the complex modulation on the RF carrier.
Exxx n+ n- <VCVS> VMRF <(> Iin+ Iin- Qin+ Qin- FREQ=fc
+ PHASE=ph <SCALE=A> <)>
Fxxx n+ n- <CCCS> VMRF <(> VI VQ FREQ=fc PHASE=ph
+ <SCALE=A> <)>
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Gxxx n+ n- <VCCS> VMRF <(> Iin+ Iin- Qin+ Qin- FREQ=fc
+ PHASE=ph <SCALE=A> <)>
Hxxx n+ n- <CCVS> VMRF <(> VI VQ FREQ=fc PHASE=ph
+ <SCALE=A> <)>
Parameter
Description
Exxx
Voltage-controlled voltage source.
Fxxx
Current-controlled current source.
Gxxx
Voltage-controlled current source.
Hxxx
Current-controlled current source.
VCVS
Keyword for voltage-controlled voltage source.
CCCS
Keyword for current-controlled current source.
VCCS
Keyword for voltage-controlled current source.
CCVS
Keyword for current-controlled current source.
n+ n-
Positive and negative controlled source connecting nodes.
VMRF
Keyword that identifies and activates the vector-modulated RF signal
source.
Iin+ Iin-
Node names for input I(t) signal.
Qin+ Qin-
Node names for input Q(t) signal.
VI VQ
FREQ
Carrier frequency in Hertz. Set fc=0.0 to generate baseband I/Q
signals.
PHASE
Carrier phase (in degrees). If fc=0.0,
■
■
SCALE
166
ph=0 and baseband I(t) is generated
ph=-90 and baseband Q(t) is generated
Unit-less amplitude scaling parameter.
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Example
Emod1 inp1 inn1 VMRF It_plus It_neg Qt_plus Qt_neg
+ freq=1g phase=0 scale=1.5
File-Driven PWL Source
Vxxx n1 n2 PWL PWLFILE='filename' <col1, <col2>> <R=repeat>
+ <TD=delay> <options>
Ixxx n1 n2 PWL PWLFILE='filename' <col1, <col2>> <R=repeat>
+ <TD=delay> <options>
Parameter
Description
Vxxx
Independent voltage source.
Ixxx
Independent current source.
n1 n2
Positive and negative terminal node names.
PWL
Keyword for piecewise linear.
PWLFILE
Text file containing the PWL data consisting of time and voltage (or
current) pairs. This file should not contain a header row, unless it is a
comment. The PWL source data is obtained by extracting col1 and
col2 from the file.
col1, <col2>
Time values are in col1 and voltage (or current) values are in col2. By
default, col1=1 and col2=2.
R
Repeat function. When an argument is not specified, the source
repeats from the beginning of the function. The argument repeated is
the time, in seconds, which specifies the start point of the waveform
being repeat. The repeat time must be less than the greatest time point
in the file.
TD
Time delay, in seconds, of the PWL function.
options
Any standard V or I source options.
Example
Vit It_plus It_neg PWL PWLFILE=’Imod.dat’
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SWEEPBLOCK in Sweep Analyses
SWEEPBLOCK in Sweep Analyses
You can use the .SWEEPBLOCK statement to specify complicated sweeps.
Sweeps affect:
■
DC sweep analysis
■
Parameter sweeps around TRAN, AC, or HB analyses
■
Frequency values used in AC or HBAC analyses
Currently, HSPICE supports the following types of sweeps:
■
■
■
Linear sweeps: sweeps a variable over an interval with a constant
increment. The syntax is one of the following:
•
variable start stop increment
•
variable lin npoints start stop
Logarithmic sweeps: sweeps a variable over an interval. To obtain each
point, this sweep multiplies the previous point by a constant factor. You can
specify the factor as a number of points per decade or octave as in:
•
variable dec npoints start stop
•
variable oct npoints start stop
Point sweeps: a variable takes on specific values that you specify as a list.
The syntax is:
variable poi npoints p1 p2 …
■
Data sweeps: a .DATA statement identifies the swept variables and their
values. The syntax is:
data=dataname
You can use the SWEEPBLOCK feature to combine linear, logarithmic, and point
sweeps, which creates more complicated sets of values over which a variable is
swept.
The .TRAN, .AC, .DC, and .HB commands can specify
SWEEPBLOCK=blockname as a sweep instead of LIN, DEC, OCT, and so forth.
Also, you can use SWEEPBLOCK for frequency sweeps with
the .AC, .HBAC, .PHASENOISE, and .HBNOISE commands.
All commands that can use SWEEPBLOCK must refer to the SWEEPBLOCK
sweep type. In addition, you must specify SWEEPBLOCK as one of the syntax
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types allowed for frequency sweeps with the .HBAC, .PHASENOISE,
and .HBNOISE commands.
Input Syntax
The SWEEPBLOCK feature creates a sweep whose set of values is the union of
a set of linear, logarithmic, and point sweeps. To specify the set of values in the
SWEEPBLOCK, use the .SWEEPBLOCK command. This command also assigns
a name to the SWEEPBLOCK. For example,
.SWEEPBLOCK swblockname sweepspec [sweepspec
+ [sweepspec […]]]]
You can use SWEEPBLOCK to specify DC sweeps, parameter sweeps, AC and
HBAC frequency sweeps, or wherever HSPICE accepts sweeps.
You can specify an unlimited number of sweepspec parameters. Each
sweepspec can specify a linear, logarithmic, or point sweep by using one of
the following forms:
start stop increment
lin npoints start stop
dec npoints start stop
oct npoints start stop
poi npoints p1 p2 …
Example
The following example specifies a logarithmic sweep from 1 to 1e9 with more
resolution from 1e6 to 1e7:
.sweepblock freqsweep dec 10 1 1g dec 1000 1meg 10meg
Using SWEEPBLOCK in a DC Parameter Sweep
To use the sweepblock in a DC parameter sweep, use the following syntax:
.DC sweepspec [sweepspec [sweepspec]]
Each sweepspec can be a linear, logarithmic, point, or data sweep, or it can be
in the form:
variable SWEEPBLOCK=swblockname
The SWEEPBLOCK syntax sweeps the specified variable over the values
contained in the SWEEPBLOCK.
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Clock Source with Random Jitter
Example
.dc vin1 0 5 0.1 vin2 sweepblock=vin2vals
Using in Parameter Sweeps in TRAN, AC, and HB Analyses
To use the sweepblock in parameter sweeps on .TRAN, .AC, and .HB
commands, and any other commands that allow parameter sweeps, use the
following syntax:
variable sweepblock=swblockname
Example 1
.tran 1n 100n sweep rout sweepblock=rvals
AC and HBAC analysis frequency sweeps can use
sweepblock=swblockname to specify the frequency values.
Example 2
.ac sweepblock=freqsweep
Limitations
■
You cannot use recursive SWEEPBLOCK specifications. That is,
a .SWEEPBLOCK command cannot refer to another SWEEPBLOCK to build its
list of values.
■
You cannot include data sweeps in a .SWEEPBLOCK statement.
Clock Source with Random Jitter
In many applications involving signal integrity, RF, analog, and mixed-signal
design, it is desirable to have an ideal signal source, such as a sine wave or
square wave, that also includes a non-ideal random drift in phase (jitter). Such
a source is useful for representing non-ideal clock sources during time-domain
transient simulation. Modeling jitter in this way can be used to examine eyediagram behavior or study how jitter may propagate through a circuit or system.
A source with jitter is useful for representing non-ideal clock sources during
time-domain transient simulation.
The PERJITTER option allows you to add periodic jitter to SIN, COS and
PULSE time domain sources.
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Syntax of SIN, COS, and Pulse Sources
The syntax of SIN source is:
Vxxx n+ n- SIN <(> vo va <freq <td <q <j >>>> <)>
+ <PERJITTER=val SEED=val>>
Ixxx n+ n- SIN <(> vo va <freq <td <q <j >>>> <)>
+ <PERJITTER=val SEED=val>>
Parameter
Description
Vxxx
Independent voltage source.
Ixxx
Independent current source.
PERJITTER
Period jitter
PWL
Keyword for piecewise linear.
PWLFILE
Text file containing the PWL data consisting of time and voltage (or
current) pairs. This file should not contain a header row, unless it is a
comment. The PWL source data is obtained by extracting col1 and
col2 from the file.
col1, <col2>
Time values are in col1 and voltage (or current) values are in col2. By
default, col1=1 and col2=2.
R
Repeat function. When an argument is not specified, the source
repeats from the beginning of the function. The argument repeated is
the time, in seconds, which specifies the start point of the waveform
being repeat. The repeat time must be less than the greatest time point
in the file.
TD
Time delay, in seconds, of the PWL function.
options
Any standard V or I source options.
The sine wave behavior following the td time delay now becomes
Equation 17
V(t) + e
– ( t – td ) ⋅ θ
= V0 + Va ⋅
π
sin 2πf 0 ( t – t d ) + --------- ϕ + φt – d d
180
The Syntax of COS source is:
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Clock Source with Random Jitter
Vxxx n+ n- COS <(> vo va <freq <td <q <j >>>> <)>
<PERJITTER=val <SEED=val>>
Ixxx n+ n- COS <(> vo va <freq <td <q <j >>>> <)>
<PERJITTER=val <SEED=val>>
The new cosine wave becomes
Equation 18
V(t) + e
– ( t – td ) ⋅ θ
= V0 + Va ⋅
π
cos 2 πf 0 ( t – t d + x ( t ) ) + --------- ϕ
180
The syntax for the PULSE source is:
Vxxx n+ n- PU<LSE> <(>v1 v2 <td <tr <tf <pw <per>>>>> <)>
+ <PERJITTER=val SEED=val>>
Ixxx n+ n- PU<LSE> <(>v1 v2 <td <tr <tf <pw <per>>>>> <)>
+ <PERJITTER=val SEED=val>>
The effect of jitter on the PULSE source results in random shifts of the rise and
fall transitions that takes place at
RISE edge: td + n ⋅ T 0 ≤t ≤td + tr + n ⋅ T 0
FALL edge: td + pw + n ⋅ T 0 ≤t ≤td + pw + tf + n ⋅ T 0
The jitter effect is equivalent to introducing random shifts in the period T 0
consistent with the 1st order jitter model based on Period Jitter.
A Gaussian random number generator computes the time deviation x ( t ) after
each leading edge of the clock sources. For flexibility, the SEED parameter
(integer) is supported for generating different random number sequences when
different SEED integers are used for initialization. SEED does not set a fixed
time deviation. It only changes the sequence of random samples. By HSPICE
(Monte Carlo) convention, the default value for SEED is 1.
An interpretation of PERJITTER is to view it as causing each period of the
PULSE/SIN/COS to be a random variable T j , where period T j will have a
Gaussian distribution about the (mean) given period value of T 0 . The standard
deviation of this Gaussian is the PERJITTER value (it is considered RMS
period jitter), which results in a bell curve distribution centered about period T 0 .
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Apply the following considerations when using PERJITTER:
■
T j should be forced to be between: 0 < T j < 2 ⋅ T 0 , since period cannot go
negative, and the curve should be symmetrical.
■
It is reasonable to require that 2 ⋅ PERJITTER < T 0 . Otherwise, the jitter
would result in very large period changes, and many would be T j < 0 .
■
To establish a waveform reference, the first period should be T 0 (i.e., no jitter
in the first period). This helps to establish good eye diagrams.
Example
As an alternative to using a Verilog-A module, you can generate a pseudorandom binary sequence (PRBS) using the following steps:
1. Construct your usual linear feedback shift register (LFSR) generator.
2. Construct a matching (T,tr,tf) PULSE source as a clock, but add jitter to it
with the PERJITTER keyword.
3. Use the PULSE source to gate (buffer) the LFSR output (through an ideal
AND gate, VCCS, and so forth).
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References
References
[1] L.J. Greenstein and M.Shafi, Microwave Digital Radio, IEEE Press, 1988.
[2] N. Sheikholeslami and P. Kabal, “A Family of Nyquist Filters Based on
Generalized Raised-Cosine Spectra,” Proceedings of the 19th Biennial
Symposium on Communications (Kingston, Ontario), pages 131-135, June
1998.
[3] IEEE Standard Definitions of Physical Quantities for Fundamental
Frequency and Time Metrology - Random Instabilities, IEEE Std. 11391999.
[4] A. van der Ziel, Noise in Solid State Devices and Circuits, John Wiley &
Sons, © 1986.
[5] A. Demir, A. Mehrotra, and J. Roychowdhury, “Phase noise in oscillators: A
unifying theory and numerical methods for characterization,” IEEE Trans.
Circuits Syst. I, vol. 47, pp. 655-674, May 2000.
[6] A. Hajimiri, S. Limotyrakis, and T.H. Lee, “Jitter and phase noise in ring
oscillators,” IEEE J. Solid-State Circuits, vol. 34, no. 6, pp. 790-804, June
1999.
[7] Jitter Analysis Techniques for High Data Rates, Application Note 1432,
Agilent Technologies, Feb. 2003.[6] Characterization of Clocks and
Oscillators, NIST Technical Note 1337, National Institute of Standards and
Technology.
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7
7
Steady-State Harmonic Balance Analysis
Describes how to use harmonic balance analysis for frequency-driven, steadystate analysis.
HSPICE RF provides several analyses that support the simulation and analysis
of radio-frequency integrated circuits (RFICs). These analyses provide
simulation capabilities that are either much more difficult to perform, or are not
practically possible by using standard HSPICE analyses. The RF analyses
include:
■
Harmonic Balance (HB) for frequency-domain, steady-state analysis, see
Harmonic Balance Analysis on page 176.
■
Shooting Newton (SN) for frequency or time domain steady state analysis,
see Chapter 8, Steady-State Shooting Newton Analysis plus spectrum
analysis specific to the SN analysis (see Shooting Newton with Fourier
Transform (.SNFT).
■
Harmonic Balance oscillator analysis (HBOSC), see Harmonic Balance
Oscillator Analysis (.HBOSC) on page 210.
■
Shooting Newton oscillator analysis (SNOSC), see Oscillator Analysis
Using Shooting Newton (.SNOSC) on page 224.
■
Harmonic Balance AC (HBAC) for periodic AC analysis, see Chapter 10,
Large Signal Periodic AC, Transfer Function, and Noise Analyses, Multitone
Harmonic Balance AC Analysis (.HBAC) on page 255.
■
Shooting Newton AC analysis, see Shooting Newton AC Analysis (.SNAC)
on page 261.
■
Harmonic Balance Noise (HBNOISE) for periodic, time-varying AC noise
analysis (see Chapter 10, Large Signal Periodic AC, Transfer Function, and
Noise Analyses).
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Harmonic Balance Analysis
■
Shooting Newton noise analysis, see Shooting Newton Noise Analysis
(.SNNOISE) on page 274.
■
Harmonic balance transfer functions, see Multitone Harmonic Balance
Transfer Function Analysis (.HBXF) on page 291.
■
Shooting Newton transfer functions, see Shooting Newton Transfer Function
Analysis (.SNXF) on page 294
■
Frequency translation S-parameter extraction for describing N-port circuits
that exhibit frequency translation effects (see Frequency Translation SParameter (HBLIN) Extraction on page 300).
■
Envelope Analysis (ENV) (see Chapter 12, Envelope Analysis).
You can use steady-state analysis on a circuit if it contains only DC and
periodic sources. These analyses assume that all “start-up” transients have
completely died out with only the steady-state response remaining. Sources
that are not periodic or DC are treated as zero-valued in these analyses.
Harmonic Balance Analysis
Harmonic balance analysis (HB) is a frequency-domain, steady-state analysis
technique. In HSPICE RF, you can use this analysis technique on a circuit that
is excited by DC and periodic sources of one or more fundamental tones. The
solution that HB finds is a set of phasors for each harmonic signal in the circuit.
You can think of this solution as a set of truncated Fourier series. HSPICE RF
allows you to specify the solution spectrum to use in an analysis. HB analysis
then finds the set of phasors at these frequencies that describes the circuit
response. The result is a set of complex valued Fourier series coefficients that
represent the waveforms at each node in the circuit.
Linear circuit elements are evaluated in the frequency domain, while nonlinear
elements are evaluated in the time domain. The nonlinear response is then
transformed to the frequency domain where it is added to (or “balanced” with)
the linear response. The resulting composite response satisfies KCL and KVL
(Kirchoff's current and voltage laws) when the circuit solution is found.
Typical applications include performing intermodulation analysis, oscillator
analysis, and gain compression analysis, on amplifiers and mixers. HB analysis
also serves as a starting point for periodic AC and noise analyses.
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Harmonic Balance Analysis
Harmonic Balance Equations
We can write Kirchoff's current law in the time domain as:
t
Equation 19
f ( v, t ) = i ( v ( t ) ) + d q ( v ( t ) ) + ∫ y ( t – τ )v ( τ ) dτ + i s ( t ) = 0
dt
–∞
■
i(v(t)) represents the resistive currents from nonlinear devices
■
q represents the charges from nonlinear devices
■
y represents the admittance of the linear devices in the circuit
■
is represents the vector of independent current sources
■
v is a variable that represents the circuit unknowns, both node voltages and
branch currents, and f(v,t) is an error term that goes to zero when Kirchoff's
current law is satisfied.
Transforming this equation to the frequency domain results in:
Equation 20
F ( V ) = I ( V ) + ΩQ ( V ) + Y ( ω)V + I s = 0
Note:
Time-differentiation is transformed to multiplication by jω terms (which make
up the Ω matrix) in the frequency domain. The convolution integral is
transformed to a simple multiplication. The Y matrix is the circuit’s modified
nodal admittance matrix.
All terms above are vectors, representing the circuit response at each analysis
frequency.
The following equation shows the vector of (complex-valued) unknowns in the
frequency domain for a circuit with K analysis frequencies and N unknowns.
Equation 21
V = V ( 1,
0)
V ( 1,
1)
…V ( 1,
K – 1)
V ( 2,
0)
…V ( N,
K – 1)
HSPICE RF finds the unknown vector (V), which satisfies the system of
nonlinear equations shown in the equation above. This is done via the NewtonRaphson technique by using either a direct solver to factor the Jacobian matrix,
or an indirect solver. The indirect solver available in HSPICE RF is the
Generalized Minimum Residual (GMRES) Solver, a Krylov technique, and uses
a matrix-implicit algorithm.
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Chapter 7: Steady-State Harmonic Balance Analysis
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Features Supported
HB supports the following features:
■
All existing HSPICE RF models.
■
Unlimited number of independent input tones.
■
Sources with multiple HB specifications.
■
SIN, PULSE, VMRF, and PWL sources with TRANFORHB=1.
Prerequisites and Limitations
The following prerequisites and limitations apply to HB:
■
Requires one .HB statement.
■
Treats sources without a DC, HB, or TRANFORHB description as a zerovalue for HB unless the sources have a transient description, in which case,
the time=0 value is used as a DC value.
Input Syntax
Without SS_TONE
.HB TONES=<F1> [<F2> <...> <FN>] [SUBHARMS=SH]
+ <NHARMS=<H1>, <H2> <...> <HN>> <INTMODMAX=n>
+ [SWEEP parameter_sweep]
With SS_TONE
.HB TONES=<F1> [<F2> <...> <FN>]
+ <NHARMS=<H1>, <H2> <...> <HN>> <INTMODMAX=n>
+ <SS_TONE=n> [SWEEP parameter_sweep]
178
Parameter
Description
TONES
Fundamental frequencies.
SUBHARMS
Allows subharmonics in the analysis spectrum. The minimum nonDC frequency in the analysis spectrum is f/subharms, where f is the
frequency of oscillation.
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Harmonic Balance Analysis
Parameter
Description
NHARMS
Number of harmonics to use for each tone. Must have the same
number of entries as TONES. You must specify NHARMS,
INTMODMAX, or both.
INTMODMAX
INTMODMAX is the maximum intermodulation product order that
you can specify in the analysis spectrum. You must specify
NHARMS, INTMODMAX, or both.
SS_TONE
Small-signal tone number for HBLIN analysis. The value must be an
integer number. The default value is 0, indicating that no small
signal tone is specified. For additional information, see Frequency
Translation S-Parameter (HBLIN) Extraction on page 300.
SWEEP
Type of sweep. You can sweep up to three variables. You can specify
either LIN, DEC, OCT, POI, SWEEPBLOCK, DATA, OPTIMIZE, or
MONTE. Specify the nsteps, start, and stop frequencies using the
following syntax for each type of sweep:
■
■
■
■
■
■
■
■
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA=dataname
OPTIMIZE=OPTxxx
MONTE=val
HB Analysis Spectrum
The NHARMS and INTMODMAX input parameters define the spectrum.
■
If INTMODMAX=N, the spectrum consists of all f=a*f1 + b*f2 + ... + n*fn
frequencies so that f>=0 and |a|+|b|+...+|n|<=N. The a,b,...,n coefficients are
integers with absolute value <=N.
■
If you do not specify INTMODMAX, it defaults to the largest value in the
NHARMS list.
■
If entries in the NHARMS list are > INTMODMAX, HSPICE RF adds the m*fk
frequencies to the spectrum, where fk is the corresponding tone, and m is a
value <= the NHARMS entry.
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Example 1
.hb tones=f1, f2 intmodmax=1
The resulting HB analysis spectrum={dc, f1, f2}
Example 2
.hb tones=f1, f2 intmodmax=2
The resulting HB analysis spectrum={dc, f1, f2, f1+f2, f1-f2, 2*f1, 2*f2}
Example 3
.hb tones=f1, f2 intmodmax=3
The resulting HB analysis spectrum={dc, f1, f2, f1+f2, f1-f2, 2*f1, 2*f2, 2*f1+f2,
2*f1-f2, 2*f2+f1, 2*f2-f1, 3*f1, 3*f2}
Example 4
.hb tones=f1, f2 nharms=2,2
The resulting HB analysis spectrum={dc, f1, f2, f1+f2, f1-f2, 2*f1, 2*f2}
Example 5
hb tones=f1, f2 nharms=2,2 intmodmax=3
The resulting HB analysis spectrum={dc, f1, f2, f1+f2, f1-f2, 2*f1, 2*f2, 2*f1-f2,
2*f1+f2, 2*f2-f1, 2*f2+f1}
Example 6
.hb tones=f1, f2 nharms=5,5 intmodmax=3
The resulting HB analysis spectrum={dc, f1, f2, f1+f2, f1-f2, 2*f1, 2*f2, 2*f1-f2,
2*f1+f2, 2*f2-f1, 2*f2+f1, 3*f1, 3*f2, 4*f1, 4*f2, 5*f1, 5*f2}
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HB Analysis Options
The following table lists the .OPTION command options specific to HB
analysis.
Table 15
HB Analysis Options
Option
Description
HBCONTINUE
Specifies whether to use the sweep solution from the
previous simulation as the initial guess for the present
simulation.
■
■
HBJREUSE
HBCONTINUE=1 (default): Use solution from previous
simulation as the initial guess.
HBCONTINUE=0: Start each simulation in a sweep from
the DC solution.
Controls when to recalculate the Jacobian matrix:
■
HBJREUSE=0 recalculates the Jacobian matrix at each
iteration.
■
HBJREUSE=1 reuses the Jacobian matrix for several
iterations, if the error is sufficiently reduced.
The default is 0 if HBSOLVER=1 or 2, or 1 if HBSOLVER=0.
HBJREUSETOL
Determines when to recalculate Jacobian matrix (if
HBJREUSE=1). The percentage by which HSPICE RF must
reduce the error from the last iteration so you can use the
Jacobian matrix for the next iteration. Must be a real number,
between 0 and 1. The default is 0.05.
HBKRYLOVDIM
Dimension of the Krylov subspace that the Krylov solver
uses. Must be an integer, greater than zero. Default is 40.
HBKRYLOVTOL
The error tolerance for the Krylov solver. Must be a real
number, greater than zero. The default is 0.01.
HBLINESEARCHFAC The line search factor. If Newton iteration produces a new
vector of HB unknowns with a higher error than the last
iteration, then scale the update step by
HBLINESEARCHFAC, and try again. Must be a real number,
between 0 and 1. The default is 0.35.
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Table 15
HB Analysis Options (Continued)
Option
Description
HBMAXITER
Specifies the maximum number of Newton-Raphson
iterations that the HB engine performs. Analysis stops when
the number of iterations reaches this value. The default is
10000.
HBSOLVER
Specifies a preconditioner to solve nonlinear circuits.
■
■
■
182
HBSOLVER=0: invokes the direct solver.
HBSOLVER=1 (default): invokes the matrix-free Krylov
solver.
HBSOLVER=2: invokes the two-level hybrid timefrequency domain solver.
HBTOL
The absolute error tolerance for determining convergence.
Must be a real number that is greater than zero. The default
is 1.e-9.
LOADHB
LOADHB=’filename’ loads the state variable information
contained in the specified file. These values are used to
initialize the HB simulation.
SAVEHB
SAVEHB=’filename’ saves the final state (that is, the no
sweep point or the steady state of the first sweep point)
variable values from a HB simulation in the specified file. This
file can be loaded as the starting point for another simulation
by using a LOADHB option.
TRANFORHB
■
TRANFORHB=1: forces HB to recognize V/I sources that
include SIN, PULSE, VMRF, and PWL transient
descriptions, and to use them in analysis. However, if the
source also has an HB description, analysis uses the HB
description instead.
■
TRANFORHB=0: forces HB to ignore transient
descriptions of V/I sources, and to use only HB
descriptions.
To override this option, specify TRANFORHB in the source
description.
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Harmonic Balance Analysis
Harmonic Balance Output Measurements
This section explains the harmonic balance output measurements you receive
after HSPICE runs an HB simulation.
Harmonic Balance Signal Representation
The HB cosine sources can be interpreted in real/imaginary and polar formats
according to:
v ( t ) = A cos ( αt + φ) = Re {Ae
j ( αt + φ)
jφ jωt
}= Re {Ae e }
jφ
Equation 22
= Re {Ae [ cos ( αt ) + j sin ( αt ) ] }
= Re {[ V R + jV I ] [ cos ( αt ) + j sin ( αt ) ] }
= V R cos ( αt ) – V I sin ( at )
= A cos ( φ) cos ( αt ) – A sin ( φ) sin ( αt )
Note that real/imaginary and polar formats are related with the standard
convention:
Equation 23
V R + jV I = Ae
jφ
V R = A cos ( φ)
V I = A sin ( φ)
A =
2
2
VR + VI
V
tan φ = -----IVR
The result of HB analysis is a complex voltage (current) spectrum at each
circuit node (or specified branch). Let a[i] be the real part and b[i] be the
imaginary part of the complex voltage at the ith frequency index. Conversion to
a steady-state time-domain waveform is given by the Fourier series expansion:
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Chapter 7: Steady-State Harmonic Balance Analysis
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Equation 24
v(t) = a[0] + a[1]*cos(2πf[1]*t) – b[1]*sin(2πf[1]*t)
+ a[2]*cos(2πf[2]*t) – b[2]*sin(2πf[2]*t)
+ a[3]*cos(2πf[3]*t) – b[3]*sin(2πf[3]*t)
+...
+ a[N]*cos(2πf[N]*t) – b[N]*sin(2πf[N]*t)
Where:
■
v(t) is the resulting time domain waveform.
■
N+1 is the total number of harmonics (including DC) in the frequency
domain spectrum of the *.hb0 file (the zero-th data point represents DC).
■
a[i] is the real component at the ith frequency
■
b[i] is the imaginary component at the ith frequency
■
f[i] is the ith frequency value (with i=0 representing the zero frequency DC
term). These frequencies need not be harmonically related.
This frequency domain (Fourier coefficient) representation can be converted
into a steady-state time domain waveform output representation by using
the .PRINT or .PROBE HBTRAN output option or by invoking the To Time
Domain function on complex spectra within CosmosScope.
Output Syntax
This section describes the syntax for the HB .PRINT and .PROBE statements.
.PRINT and .PROBE Statements
.PRINT HB TYPE(NODES or ELEM)[INDICES]
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.PROBE HB TYPE(NODES or ELEM)[INDICES]
Parameter
Description
TYPE(NODES or ELEM) Specifies a harmonic type node or element.
TYPE can be one of the following:
■
■
■
■
Voltage type –
V = voltage magnitude and phase in degrees
VR = real component
VI = imaginary component
VM = magnitude
VP - Phase in degrees
VPD - Phase in degrees
VPR - Phase in radians
VDB - dB units
VDBM - dB relative to 1 mV
Current type –
I = current magnitude and phase in degrees
IR = real component
II = imaginary component
IM = magnitude
IP - Phase in degrees
IPD - Phase in degrees
IPR - Phase in radians
IDB - dB units
IDBM - dB relative to 1 mV
Power type – P
Frequency type –
‘HERTZ[i]’ (for single tone analysis), ‘HERTZ[i][j]’ (for
two-tone analysis) , ‘HERTZ[i][j][k]’ (for 3-tone analysis),
etc.
You must specify the harmonic index integer for the
HERTZ keyword. The frequency of the specified
harmonics is dumped.
NODES or ELEM can be one of the following:
■
■
■
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Voltage type – a single node name (n1), or a pair of node
names, (n1,n2)
Current type – an element name (elemname)
Power type – a resistor (resistorname) or port
(portname) element name.
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Parameter
Description
INDICES
Index to tones in the form [n1, n2, ..., nN], where nj is the
index of the HB tone and the HB statement contains N
tones. If INDICES is used, then wildcards are not
supported.
HB data can be transformed into the time domain and output using the
following syntax:
.PRINT hbtran ov1 <ov2 ... >
.PROBE hbtran ov1 <ov2 ... >
Where ov1 ... are the output variables to print or probe.
Outputting Phase Noise Source as ASCII Data Files Using
*.printpn0
HB phase noise and phase noise analyses can output simulation results as
ASCII data in *.printpn0 files for HBOSC and HBNOISE. By extending the E
and G voltage-controlled source syntax, the phase noise data in ASCII phase
noise files can used as input for specifying behavioral noise sources
Usage Model
The syntax for the voltage controlled voltage (E) or current (G) source is as
follows:
Exxx node1 node2 noise file='filename' <mname='measname'>
Gxxx node1 node2 noise file='filename' <mname='measname'>
Where,file='filename' is the name of the ASCII phase noise data file. The
file name is typically designated as 'design.printpn0', for a .HBOSC phase
noise analysis or .HBNOISE analysis.
mname='measname' is used to select the appropriate noise measurement
name to be taken from the *.printpn0 file.
measname can be one of the following:
186
■
NLP_L(f) selects the nlp_L(f) phase noise data in units of dBc/Hz
■
PAC_L(f) - selects the pac_l(f) phase noise data in units of dBc/Hz
■
BPN_L(f) - selects the bpn_l(f) phase noise data in units of dBc/Hz
■
ONOISE - selects the onoise data based on .HBNOISE or .SNNOISE
analysis
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Harmonic Balance Analysis
Calculating Power Measurements After HB Analyses
Two types of power measurements are available: dissipated power in resistors
and delivered power to port elements. The following subtle differences between
these two measurements are described in this section.
Power Dissipated in a Resistor
All power calculations make use of the fundamental phasor power relationship
given as the following equation, where voltage V and current I are complex
phasors given in peak values (not rms, nor peak-to-peak):
Equation 25
1
P rms = --- Re {VI∗ }
2
In the case of a simple resistor, its current and voltage are related according to
Vn=InR. The power dissipated in a resistor of (real) value R at frequency index
n is then given by:
2
Equation 26
Vn
P rms ( resistor ) [ n ] = ---------2R
Power Delivered to a Port Element
The port element can be either a source or sink for power. You can use a
special calculation that computes the power flowing into a port element even if
the port element itself is the source of that power. In the following figure is a
port element connected to a circuit (the port element may or may not include a
voltage source).
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Harmonic Balance Analysis
Zo
In
+
+
Vs
Port
Element
Remainder
Of
Circuit
Vn
-
Figure 23
Port Element
Let Vn be the (peak) voltage across the terminals of the port element (at
frequency index n). Let In be the (peak) current into the (1st) terminal of the port
element (at frequency index n). Let Zo be the impedance value of the z0 port
element. Then, the power wave flowing into the terminals of the port element
(at frequency index n) can be computed according to:
Equation 27
1 Vn + Zo In
P in [ n ] = --- ---------------------2 2 Z o
2
This power expression remains valid whether or not the port element includes
an internal voltage source at the same frequency. If the port element includes a
voltage source at the same frequency, you can use this power calculation to
compute the magnitude of the related large-signal scattering parameters.
If you expand the preceding formula, the power delivered to a port element with
(real) impedance Zo is given by
2
Equation 28
2
2
⎫
1 ⎧ Vn + Zo In
1
P rms ( port ) [ n ] = --- ⎨ ----------------------------------- + --- Re {V n I∗n }⎬
2⎩
2
4Z o
⎭
This power value represents the power incident upon and delivered to the port
element's load impedance (Zo) due to other power sources in the circuit, and
due to reflections of its own generated power.
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If the port element is used as a load resistor (no internal source), the preceding
equation reduces to that for the simple resistor.
If you used the port element as a power source (with non-zero available power,
i.e. a non-zero Vs) and it is terminated in a matched load (Zo), the port power
measurement returns 0 W, because no power is reflected.
You can request power measurements in the form of complete spectra or in the
form of scalar quantities that represent power at a particular element. To
request a complete power spectrum, use the following syntax.
.PRINT HB P(Elem)
.PROBE HB P(Elem)
To request a power value at a particular frequency tone, use the following
syntax:
.PRINT HB P(Elem)[<n1<,n2<n3,...>>>]
.PROBE HB P(Elem)[<n1<,n2<,n3,...>>>]
The Elem is the name of either a Resistor (R) or Port (P) element, and n1,n2,
and n3 are integer indices used for selecting a particular frequency in the
Harmonic Balance output spectrum.
Example 1
Prints a table of the RMS power (spectrum) dissipated by resistor R1.
.PRINT HB P(R1)
Example 2
Outputs the RMS power dissipated by resistor R1 at the fundamental HB
analysis frequency following a one-tone analysis.
.PROBE HB P(R1)[1]
Example 3
Prints the power dissipated by resistor R1 at DC following a one-tone analysis.
.PRINT HB P(R1)[0]
Example 4
Outputs the RMS power dissipated by resistor R1 at the (low-side) 3rd order
intermodulation product following an HB two-tone analysis.
.PROBE HB P(R1)[2,-1]
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Example 5
Prints the RMS power dissipated by resistor R1 at the (high-side) 3rd order
intermodulation product following an HB two-tone analysis.
.PRINT HB P(R1)[-1,2]
Example 6
Outputs the RMS power (spectrum) delivered to port element Pload.
.PROBE HB P(Pload)
Example 7
Prints the RMS power delivered to port element Pload at the fundamental HB
analysis frequency following a one-tone analysis.
.PRINT HB P(Pload)[1] $
Example 8
Outputs the RMS power delivered to port element Pload at the (low-side) 3rd
order intermodulation product following an HB two-tone analysis.
.PROBE HB P(Pload)[2,-1]
Calculations for Time-Domain Output
In addition to a frequency-domain output, HB analysis also supports a timedomain output. The equivalent time-domain waveform is generated according
to the Fourier series expansion given by
Equation 29
V ( n1 )@time t = SUM OVERm ( REALV ( n1 ) [ m ] ) • cos "" ( Ω[ m ] • t ) – IMAG(V ( n1 ) [ m ] sin Ω[ m • t ] • t
Where m starts from 0 to the number of frequency points in the HB simulation.
The output syntax is
.PRINT [HBTRAN | HBTR] V(n1)
.PROBE [HBTRAN | HBTR] V(n1)
The output time ranges from 0 to twice the period of the smallest frequency in
the HB spectra.
Minimizing Gibbs Phenomenon
You can use the HB_GIBBS option for HBTRAN output to minimize Gibbs’
phenomenon that may occur in transforming a square-wave signal from the
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frequency domain to the time domain. The syntax is
.OPTION HB_GIBBS = < n > (defaults to zero, which is equivalent to not
using it at all).
N
The result is that the HBTRAN waveforms are filtered by a ( sin c ( x ) ) function
before being transformed to the time domain via FFT. This option applies only
to single-tone output. For example:
.option hb_gibbs = 2
...
.print hbtran v(2)
Figure 24
Upper square-wave signal shows HB_GIBBS = 2, while the lower
shows the option = 0
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Output Examples
.PRINT HB P(rload)
.PROBE
.PRINT
.PROBE
.PRINT
.PROBE
$ RMS power (spectrum)
$ dissipated at the rload resistor
HB V(n1,v2)
$ Differential voltage (spectrum)
$ between the n1,n2 nodes
HB VP(out)[1]
$ Phase of voltage at the out
$ node, at the fundamental
$ frequency
HB P(Pout)[2,-1] $ RMS power delivered to the Pout
$ port, at third-order intermod
HBTRAN V(n1)
$ Voltage at n1 in time domain
HBTRAN V(n1<,n2>) $ Differential voltages between n1
$ and n2 node in time domain.
Using .MEASURE with .HB Analyses
■
For transient analysis (TRAN), the independent variable for
calculating .MEASURE is time.
■
For AC analysis, the independent variable for calculating .MEASURE is
frequency.
■
However, as with DC analysis, the use of a .MEASURE command is peculiar
for HB analysis, because it has no obvious independent variable.
In HSPICE RF, the independent variable for HB .MEASURE analysis is the first
swept variable specified in the .HB simulation control statement. This variable
can be anything: frequency, power, voltage, current, a component value, and so
on.
Example 1
For the following .HB simulation control statement, the independent variable is
the swept tone frequency, and the .MEASURE command values return results
based on this frequency sweep:
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* HARMONIC BALANCE tone-frequency sweep for amplifier
.param freq1=1.91e9 power=1e-3
.HB tones=freq1 nharms=10 sweep freq1 LIN 10 1.91e9 2.0e9
.MEASURE HB Patf0 FIND P(Rload)[1] AT=1.95e9 $ Power at
+ f0=1.95Ghz
.MEASURE HB Frq1W WHEN P(Rload)[1]=1. $ freq1 @ 1 Watt
.MEASURE HB BW1W TRIG AT=1.92e9 TARG P(Rload)[1] VAL=1.
+ CROSS=2 $ 1 Watt bandwidth
.MEASURE HB MaxPwr MAX P(Rload)[1] FROM=1.91e9 TO=2.0e9
+ $ Finds max output power
.MEASURE HB MinPwr MIN P(Rload)[1] FROM=1.91e9 TO=2.0e9
+ $ Finds min output power
Example 2
In the following example, the independent variable is the power variable, and
the .MEASURE values return results based on the power sweep. Units are in
Watts.
* HARMONIC BALANCE power sweep for amplifier
.param freq1=1.91e9 power=1e-3
.HB tones=freq1 nharms=10 sweep power DEC 10 1e-6 1e-3
.MEASURE HB Pat1uW FIND P(Rload)[1] AT=1e-6 $ Pout at 1uW
.MEASURE HB Pin1W WHEN P(Rload)[1]=1. $ Pin @ 1 Watt Pout
.MEASURE HB Prange1W TRIG AT=1.92e9 TARG P(Rload)[1] VAL=1.
+ CROSS=2
$ 1W oper. range
.MEASURE HB ssGain DERIV P(Rload)[1] AT=1e-5
+ $ relative power gain at 10uW input
.MEASURE HB Gain3rd DERIV P(Rload)[3] AT=1e-5
+ $ 3rd harmonic gain at 10uW input
.MEASURE HB PAE1W FIND ‘(P(Rload)[1]-power)/P(Vdc)[0]’
+ WHEN P(Rload)[1]=1 $ PAE at 1 Watt output
Example 3
In this example, the independent variable is again the power variable, and
the .MEASURE values return results based on the power sweep. This is a twotone sweep, where both input frequency sources are at the same power level in
Watts.
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HB Output Data Files
* HARMONIC BALANCE two-tone sweep for amplifier
* An IP3 calculation is made at 10uW in the sweep
.param freq1=1.91e9 freq2=1.91e9 power=1e-3
.HB tones=freq1,freq2 nharms=6,6 sweep power DEC 10 1e-6 1e-3
.MEASURE HB Pf1dBm FIND ’10.*LOG(P(Rload)[1,0]/1.e-3)’
+ AT=1e-5 $ P(f1) at 10uW input
.MEASURE HB P2f1_f2dBm FIND ’10.*LOG(P(Rload)[2,-1]/1.e-3)’
+ AT=1e-5 $ P(2f1-f2) at 10uW input
.MEASURE HB OIP3dBm PARAM = ‘0.5*(3.*Pf1dBm-P2f1_f2dBm)’
.MEASURE HB IIP3dBm PARAM = ‘OIP3dBm-Pf1dBm+20.0’
.MEASURE HB AM2PM DERIV VP(outp,outn)[1] AT=1e-5
+ $ AM to PM Conversion in Deg/Watt
If you do not specify an HB sweep, then .MEASURE assumes a single-valued
independent variable sweep.
You can apply the measurements to current, voltage, and power waveforms.
The independent variable for measurements is the swept variable (such as
power), not the frequency axis corresponding to a single HB steady state point.
HSPICE RF also supports the .MEASURE [HBTRAN | HBTR] ... syntax.
Similar to the .PROBE and .PRINT HBTR statements in the section
Calculations for Time-Domain Output on page 190, a .MEASURE HBTR
statement is applied on the signals obtained in the same way. Moreover, like a
.MEASURE statement in transient analysis, the independent variable in a
.MEASURE HBTR statement is time.
HSPICE RF optimization can read the data from .MEASURE HB and .MEASURE
HBTR statements. The optimization syntax in HSPICE RF is identical to that in
the HSPICE (for details, see Statistical Analysis in the HSPICE User Guide:
Simulation and Analysis). Due to the difference in the independent variable
between the .MEASURE HB and .MEASURE HBTR statements, these two types
of measurements cannot be mixed in a HSPICE RF optimization. But a
.MEASURE HBTR statement can be combined with a .MEASURE PHASENOISE
statement (see Measuring Phase Noise with .MEASURE PHASENOISE on
page 238) and a .MEASURE HBNOISE statement (see Measuring HBNOISE
Analyses with .MEASURE on page 271) in a HSPICE RF optimization flow.
HB Output Data Files
The results of an HB analysis are complex spectral components at each
frequency point. The a[i] is the real part, and b[i] is the imaginary part of the
complex voltage at frequency index i. The conversion to a steady state timedomain is then given by the Fourier series expansion.
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HB Output Data Files
An HB analysis produces these output data files:
■
Output from the .PRINT HB statement is written to a .printhb# file.
•
The header contains the large signal fundamental frequencies.
•
The columns of data are labeled as HERTZ, followed by frequency
indices, and then the output variable names.
•
The sum of the frequency indices, multiplied by the corresponding
fundamental frequencies, add up to the frequency in the first column.
■
Output from the .PROBE HB statement is written to a .hb# file. It is in the
same format as the HSPICE transient analysis .tr# file. Besides the output
waveform, it contains the information of harmonic indices and basic tone
frequencies.
■
Output from the .PRINT HBTRAN statement is written to a .printhr# file. The
format is identical to a .print# file.
■
Output from the .PROBE HBTRAN statement is written to a .hr# file. The
format is identical to a .tr# file.
■
Reported performance log statistics are written to a .lis file:
•
Name of HB data file.
•
Simulation time:
DC operating point (op) time
HB time
Total simulation time
•
Memory used
•
Size of matrix (nodes * harmonics)
•
Final HB residual error
Errors and Warnings
Table 16 lists the errors messages and Table 17 on page 197 lists the warning
messages.
Table 16
HB Analysis Error Messages
File
Description
HB_ERR.1
Harmonic numbers must be positive non-zero.
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HB Output Data Files
Table 16
196
HB Analysis Error Messages (Continued)
File
Description
HB_ERR.2
No .hb frequencies given.
HB_ERR.3
Negative frequency given.
HB_ERR.4
Number of harmonics should be greater than zero.
HB_ERR.5
Different number of tones, nharms.
HB_ERR.6
Bad probe node format for oscillator analysis.
HB_ERR.7
Bad format for FSPTS.
HB_ERR.8
Bad .hb keyword.
HB_ERR.9
Tones must be specified for .hb analysis.
HB_ERR.10
Nharms or intmodmax must be specified for .hb analysis.
HB_ERR.11
Source harmonic out of range.
HB_ERR.12
Source named in the tones list is not defined.
HB_ERR.13
Source named in the tones list does not have TRANFORHB
specified.
HB_ERR.14
Source named in the tones list has no transient description.
HB_ERR.15
Source named in the tones list must be HB, SIN, PULSE, PWL,
or VMRF.
HB_ERR.16
Tone specification for the source is inconsistent with its
frequency.
HB_ERR.17
HB oscillator analysis has reached the NULL solution.
HB_ERR.18
Bad subharms format.
HB_ERR.19
Modtone may not be set to the same value as tone.
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HB Output Data Files
Table 17
HB Analysis Warning Messages
File
Description
HB_WARN.1
.hb multiply defined. Last one will be used.
HB_WARN.2
Tone specified for V/I source not specified in .HB command.
HB_WARN.3
HB convergence not achieved.
HB_WARN.4
Source specifies both HB and transient description. HB
description will be used.
HB_WARN.5
Source specifies exponential decay. HB will ignore it.
HB_WARN.6
Source specifies a non-positive frequency.
HB_WARN.7
Source does not fit the HB spectrum.
HB_WARN.8
Source cannot be used with the TRANFORHB option.
HB_WARN.9
Frequency not found from transient analysis
HB_WARN.10
.hb/.hbosc will be ignored due to .env/.envosc.
HB_WARN.11
HBTRANINIT does not support more than one input tone.
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References
References
[1] S. Maas, Nonlinear Microwave Circuits, Chapter 3, IEEE Press, 1997.
[2] R. Gilmore and M.B. Steer, “Nonlinear Circuit Analysis Using the Method of
Harmonic Balance - A Review of the Art, Part I, Introductory Concepts.”
International Journal of Microwave and Millimeter-wave Computer-Aided
Engineering, Volume 1, No. 1, pages 22-37, 1991.
[3] R. Gilmore and M.B. Steer, “Nonlinear Circuit Analysis Using the Method of
Harmonic Balance - A Review of the Art. Part II. Advanced Concepts.”
International Journal of Microwave and Millimeter-wave Computer-Aided
Engineering, Volume 1, No. 2, pages 159-180, 1991.
[4] V. Rizzoli, F. Mastri, F. Sgallari, G. Spaletta, “Harmonic-Balance Simulation
of Strongly Nonlinear Very Large-Size Microwave Circuits by Inexact
Newton Methods,” MTT-S Digest, pages 1357-1360, 1996.
[5] S. Skaggs, Efficient Harmonic Balance Modeling of Large Microwave
Circuits, Ph.D. thesis, North Carolina State University, 1999.
[6] R.S. Carson, High-Frequency Amplifiers, 2nd Edition, John Wiley & Sons,
1982
[7] S.Y. Liao, Microwave Circuit Analysis and Amplifier Design, Prentice-Hall,
1987.
[8] J. Roychowdhury, D. Long, P. Feldmann, “Cyclostationary Noise Analysis of
Large RF Circuits with Multitone Excitations”, IEEE JSCC, volume 33,
number 3, March 1998.
[9] Y. Saad, Iterative Methods for Sparse Linear Systems, PWS Publishing
Company, 1995.
[10] J. Roychowdhury, D. Long, and P. Feldmann, “Cyclostationary Noise
Analysis of Large RF Circuits with Multitone Excitations,” IEEE Journal of
Solid-State Circuits, volume 33, pages 324–336, March 1998.
[11] K. Kurakawa, “Power waves and the Scattering Matrix,” IEEE Trans.
Microwave Theory Tech., vol. MTT-13, pp. 194-202, March 1965.
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8
Steady-State Shooting Newton Analysis
8
Describes HSPICE RF steady-state time domain analysis based on a
Shooting-Newton algorithm.
These topics are covered in the following sections:
■
SN Steady-State Time Domain Analysis
■
SN Analysis Syntax
■
SN Analysis Output
■
Shooting Newton with Fourier Transform (.SNFT)
SN Steady-State Time Domain Analysis
An advanced Shooting Newton (SN) algorithm provides additional performance
and functionality to HSPICE RF for time-domain, steady-state analysis.
Shooting-Newton adds analysis capabilities for PLL components, digital
circuits/logic, such as ring oscillators, frequency dividers, phase/frequency
detectors (PFDs), and for other digital logic circuits and RF components that
require steady-state analysis, but operate with waveforms that are more square
wave than sinusoidal.
The Shooting-Newton algorithm effectively analyzes applications including:
■
Ring oscillators (see Chapter 9, Oscillator and Phase Noise Analysis)
■
Frequency dividers (prescalers)
■
Mixer conversion gain
■
Phase-frequency detectors (PFDs)
■
Mixer noise figure
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SN Analysis Syntax
Functionality includes:
■
Both driven and oscillator (autonomous) analyses
■
Time Domain or Frequency analysis based on advanced Shooting Newton
algorithm
■
Spectrum analysis specific to the SN analysis (see Shooting Newton with
Fourier Transform (.SNFT) on page 205)
■
Shooting Newton-based AC analysis (SNAC) (see Shooting Newton AC
Analysis (.SNAC) on page 261)
■
Shooting Newton-based noise analysis (SNNOISE) (see Oscillator Analysis
Using Shooting Newton (.SNOSC) on page 224)
■
Shooting Newton-based phase noise analysis (PHASENOISE) (see
Oscillator and Phase Noise Analysis on page 209)
SN Analysis Syntax
Shooting Newton provides two syntaxes. Syntax #1 is recommended when you
are using/making Time Domain sources and measurements (for example,
going from .TRAN to .SN). Syntax #2 effectively supports Frequency Domain
sources and measurements (and should be used, for example, when going
from .HB to .SN).
Syntax #1
.SN TRES=<Tr> PERIOD=<T> [TRINIT=<Ti>]
+ [SWEEP parameter_sweep] [MAXTRINITCYCLES=<integer>]
or, Syntax #2
.SN TONE=<F1> NHARMS=<N> [TRINIT=<Ti]>
+ [SWEEP parameter_sweep] [MAXTRINITCYCLES=<integer>]
where
200
Parameter
Description
TRES
The time resolution to be computed for the steady-state
waveforms (in seconds).
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SN Analysis Syntax
Parameter
Description
PERIOD
The expected period T (seconds) of the steady-state
waveforms. Enter an approximate value when using for
oscillator analysis. The period of the steady-state waveform
may be entered either as PERIOD or its reciprocal, TONE.
TONE
The fundamental frequency (in Hz).
NHARMS
Specifies the number of high-frequency harmonic
components to include in the analysis. NHARMS defaults to
PERIOD/TRES rounded to nearest integer. NHARMS is
required to run subsequent SNAC, SNNOISE, SNXF, and
PHASENOISE analyses. When using Syntax #1, NHARMS is
computed automatically as NHARMS=Round(PERIOD/
TRES).
TRINIT
This is the transient initialization time. If not specified, the
transient initialization time will be equal to the period (for
Syntax 1) or the reciprocal of the tone (for Syntax 2).
SWEEP
Specifies the parameter sweep. As in all main analyses in
HSPICE RF such as .TRAN, .HB, etc., you can specify LIN,
DEC, OCT, POI, SWEEPBLOCK, DATA, MONTE, or
OPTIMIZE.
MAXTRINITCYCLES
Stops SN stabilization simulation and frequency detection
when the simulator detects that maxtrinitcycles have been
reached in the oscnode signal, or when time=trinit, whichever
comes first. Minimum cycles is 1.
Options
In addition to all .TRAN options, .SN analysis supports the following options.
Option
Default Description
.OPTION SNMAXITER=<integer>
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SN Analysis Output
Option
Default Description
.OPTION SNACCURACY=<integer> 10
Similar to the sim_accuracy definition in
.TRAN, i.e., larger values of snaccuracy
result in a more accurate solution but
may require more time points. Because
Shooting-Newton must store derivative
information at every time point, the
memory requirements may be
significant if the number of time points
is very large.
The maximum integer value is 50.
.OPTION LOADSNINIT=”filename”
Loads the operating point saved at the
end of SN initialization which is used as
initial conditions for the ShootingNewton method.
.OPTION SAVESNINIT=”filename”
Saves the operating point at the end of
SN initialization (sninit).
SN Analysis Output
The output from .SN analysis is generated in both time and frequency domains.
The time domain output variables are the same as for standard transient
analysis:
■
individual nodal voltages: V(n1 [,n2])
■
branch currents: I(Vxx)
■
element power dissipation: In(element)
It is also possible to output the results from Shooting Newton analysis in terms
of complex, frequency-domain output variables. This output format is activated
by using the “SNFD” keyword in the output syntax.
For output in the frequency domain, the syntax is identical to the Harmonic
Balance output syntax:
.PRINT SNFD TYPE(NODES | ELEM)[INDICES]
.PROBE SNFD TYPE(NODES | ELEM)[INDICES]
Parameter Description
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SN Analysis Output
TYPE Specifies a harmonic type node or element. TYPE can be one of the
following:
■
Voltage type –
■
V = voltage magnitude and phase in degrees
■
VR = real component
■
VI = imaginary component
■
VM = magnitude
■
VP - Phase in degrees
■
VPD - Phase in degrees
■
VPR - Phase in radians
■
VDB - dB units
■
VDBM - dB relative to 1 mV
■
Current type –
■
I = current magnitude and phase in degrees
■
IR = real component
■
II = imaginary component
■
IM = magnitude
■
IP - Phase in degrees
■
IPD - Phase in degrees
■
IPR - Phase in radians
■
IDB - dB units
■
IDBM - dB relative to 1 mV
■
Power type – P
■
Frequency type –
■
hertz[index], hertz[index1, index2, ...]
You must specify the harmonic index for the hertz variable. The frequency of
the specified harmonics is dumped.
Parameter Description
NODES | ELEM can be any of the following:
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SN Analysis Output
■
Voltage type – a single node name (n1), or a pair of node names, (n1,n2)
■
Current type – an element name (elemname)
■
Power type – a resistor (resistorname) or port (portname) element name
■
INDEX n1, is the harmonic index of the SNFD tone. Index is limited to the
single tone associated with the SN analysis.
Output Files
The time domain data are output to printsn0 and .sn0 files. Frequency domain
data are output to .printsnf0 and .snf0 files.
Output Format
The format for time domain output is the same as standard transient analysis.
For frequency domain output, the format is similar to HB. The main difference is
that Shooting Newton output in the frequency domain is single tone only.
The results of an SN analysis are complex spectral components at each
frequency point. The a[i] is the real part, and b[i] is the imaginary part of the
complex voltage at frequency index i. The conversion to a steady state timedomain is then given by the Fourier series expansion.
An SN analysis produces these output data files:
■
Output from the .PRINT SN statement is written to a .printsn# file.
•
The header contains the large signal fundamental frequencies.
•
The columns of data are labeled as HERTZ, followed by frequency
indices, and then the output variable names.
•
The sum of the frequency indices, multiplied by the corresponding
fundamental frequencies, add up to the frequency in the first column.
■
Output from the .PROBE SN statement is written to a .sn# file in the same
format as the HSPICE transient analysis .tr# file. It contains the information
of harmonic indices and basic tone frequencies plus the output waveform.
■
Reported performance log statistics are written to a .lis file:
•
Name of SN data file.
•
Simulation time:
DC operating point (op) time
SN time
Total simulation time
•
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Memory used
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Chapter 8: Steady-State Shooting Newton Analysis
Shooting Newton with Fourier Transform (.SNFT)
•
Size of matrix (nodes * harmonics)
•
Final SN residual error
Shooting Newton with Fourier Transform (.SNFT)
The .SNFT command is to the .SN analysis what .FFT is to the TRAN analysis,
a means to provide spectrum analysis. Spectrum analysis represents a timedomain signal, within the frequency domain. .SNFT uses the Fourier transform:
a Discrete Fourier Transform (DFT) uses sequences of time values to
determine the frequency content of analog signals, in circuit simulation.
The .SNFT statement uses the internal time point values.
By default, the .SNFT statement uses a second-order interpolation to obtain
waveform samples, based on the number of points that you specify.
You can use windowing functions to reduce the effects of waveform truncation
on the spectral content. You can also use the .SNFT command to specify:
■
output format
■
frequency
■
number of harmonics
■
total harmonic distortion (THD)
.SNFT Input Syntax
The .SNFT command an take arguments with either alphanumeric or numerics
and expressions.
Syntax # 1 Alphanumeric input
.SNFT <output_var> <START=value> <STOP=value>
+ <NP=value> <FORMAT=keyword>
+ <WINDOW=keyword> <ALFA=value>
+ <FREQ=value> <FMIN=value> <FMAX=value>
Syntax #2 Numerics and expressions
.SNFT <output_var> <START=param_expr1> <STOP=param_expr2>
+ <NP=param_expr3> <FORMAT=keyword>
+ <WINDOW=keyword> <ALFA=param_expr4>
+ <FREQ=param_expr5> <FMIN=param_expr6> <FMAX=param_expr7>
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Chapter 8: Steady-State Shooting Newton Analysis
Shooting Newton with Fourier Transform (.SNFT)
Arguments
Argument
Description
output_var
Can be any valid output variable, such as voltage, current, or power.
START
Start of the output variable waveform to analyze. Defaults to the START
value in the .SN statement, which defaults to 0.
FROM
An alias for START in .SNFT statements.
STOP
End of the output variable waveform to analyze. Defaults to the TSTOP
value in the .SN statement.
TO
An alias for STOP, in .SNFT statements.
NP
Number of points to use in the SNFT analysis. NP must be a power of
2. If NP is not a power of 2, HSPICE automatically adjusts it to the
closest higher number that is a power of 2. The default is 1024.
FORMAT
Specifies the output format:
■
■
WINDOW
Specifies the window type to use:
■
■
■
■
■
■
■
■
ALFA
NORM= normalized magnitude (default)
UNORM=unnormalized magnitude
RECT=simple rectangular truncation window (default).
BART=Bartlett (triangular) window.
HANN=Hanning window.
HAMM=Hamming window.
BLACK=Blackman window.
HARRIS=Blackman-Harris window.
GAUSS=Gaussian window.
KAISER=Kaiser-Bessel window.
Parameter to use in GAUSS and KAISER windows to control the
highest side-lobe level, bandwidth, and so on.
1.0 <= ALFA <= 20.0
The default is 3.0
FREQ
206
Frequency to analyze. If FREQ is non-zero, the output lists only the
harmonics of this frequency, based on FMIN and FMAX. HSPICE also
prints the THD for these harmonics. The default is 0.0 (Hz).
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Chapter 8: Steady-State Shooting Newton Analysis
Shooting Newton with Fourier Transform (.SNFT)
Argument
Description
FMIN
Minimum frequency for which HSPICE prints SNFT output into the
listing file. THD calculations also use this frequency.
T=(STOP-START)
The default is 1.0/T (Hz).
FMAX
Maximum frequency for which HSPICE prints SNFT output into the
listing file. THD calculations also use this frequency. The default is
0.5*NP*FM IN (Hz).
Example 1
.SNFT v(1)
.SNFT v(1,2) np=1024 start=0.3m stop=0.5m freq=5.0k
+ window=kaiser alfa=2.5
.SNFT I(rload) start=0m to=2.0m fmin=100k fmax=120k
+ format=unorm
.SNFT par(‘v(1) + v(2)’) from=0.2u stop=1.2u
+ window=harris
Example 2
.SNFT v(1) np=1024
.SNFT v(2) np=1024
This example generates an .snft0 file for the SNFT of v(1) and an .snft1 file for
the SNFT of v(2).
.SN Signal Sources
.SN analysis assumes that all stimuli are periodic with period T. If the circuit is
driven with more than one periodic stimulus, then the frequencies must be all
co-periodic and T must match the common period or some integer multiple of it.
The .SN analysis only supports .tran (time-domain) periodic signal sources.
(Refer to the .tran analysis for a detailed documentation on transient signal
sources).
.SN Reported Performance Log Statistics
The following performance statistics are displayed:
■
DC operating time
■
Initial transient time (including user's time for circuit stabilization)
■
Total simulation time
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Chapter 8: Steady-State Shooting Newton Analysis
Shooting Newton with Fourier Transform (.SNFT)
■
SN time
■
Total simulation time
■
Memory used
■
Final SN convergence residual error
■
The value of the computed frequency if the circuit is autonomous
Errors/Warnings
Error messages are displayed with convergence recommendations in cases of
non-convergence within the maximum number of Shooting-Newton iterations.
Error messages are displayed for software errors such as segmentation
violations, and abort conditions such as:
■
unrecognized format, i.e., unrecognized V/I source
■
faulty input values, i.e., wrong sign, out of range value
■
unspecified values, i.e., unspecified tone
■
inconsistent values, i.e., non-commensurable tones
■
duplicate values, i.e., same entry, given more than one, the last one is
always taken
Limitations and Assumptions
True distributed components (such as ideal delays or transmission lines) are
not supported; components with hidden states are not supported.
Example
For a demonstration of using Shooting Newton analysis you can run the
pdfcpGain.sp file shipped with the HSPICE RF distribution, located in the
directory $<installdir>/hspicerf/examples. This example performs analysis on a
D-flipflop phase frequency divider with charge pump, implemented in 50nm
technology. The example is configured to measure the gain (volts per degree)
of the DFF PFD and tri-state output combination.
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9
Oscillator and Phase Noise Analysis
9
Describes how to use HSPICE RF to perform oscillator and phase noise
analysis on oscillator circuits.
These topics are covered in the following sections:
■
Harmonic Balance or Shooting Newton for Oscillator Analysis
■
Harmonic Balance Oscillator Analysis (.HBOSC)
■
Input Syntax for Harmonic Balance Oscillator Analysis
■
HB Simulation of Ring Oscillators
■
HBOSC Analysis Using Transient Initialization
■
Oscillator Analysis Using Shooting Newton (.SNOSC)
■
Phase Noise Analysis (.PHASENOISE)
■
Accumulated Jitter Measurement for Closed Loop PLL Analysis
■
Small-Signal Phase-Domain Noise Analysis (.ACPHASENOISE)
Harmonic Balance or Shooting Newton for Oscillator Analysis
Oscillator Classification
Oscillators can be divided into two main categories:
1. Ring oscillators: These oscillators tend to have low Q and operate based on
delay of digital cells such as inverters. Ring oscillators have strong nonlinear
behavior and output signals are often square-wave-like. Ring oscillators can
be analyzed in either the frequency domain using Harmonic Balance
analysis or in the time domain using Shooting Newton analysis.
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Chapter 9: Oscillator and Phase Noise Analysis
Harmonic Balance Oscillator Analysis (.HBOSC)
2. Harmonic oscillators: Common harmonic oscillators are LC and crystal
oscillators. These oscillators tend to have a high Q, making it difficult to find
the oscillation frequency. Their behavior tends to be only mildly nonlinear
and their output signals tend to be close to purely sinusoidal. Harmonic
Balance analysis is most effective for analyzing harmonic oscillators.
HSPICE RF includes special analysis algorithms for finding the steady-state
solution for oscillator circuits. In oscillators, there are no driving sources that set
the frequencies of operation, but rather the fundamental oscillation frequency is
one of the unknowns that is being solved by the simulator. HSPICE RF
provides two approaches: either harmonic balance or analysis based on the
Shooting Newton algorithm.
The following sections are presented in this chapter:
■
Harmonic Balance Oscillator Analysis (.HBOSC)
■
Oscillator Analysis Using Shooting Newton (.SNOSC)
■
Phase Noise Analysis (.PHASENOISE)
Harmonic Balance Oscillator Analysis (.HBOSC)
Because the frequency of oscillation is not determined by the frequencies of
driving sources, a slightly different set of nonlinear equations are solved during
the simulation and are as shown in the following equation:
Equation 30
F ( V, ω0 ) = I ( V, ω0 ) + ΩQ ( V, ω0 ) + Y ( ω0 )V + I s
HSPICE harmonic balance oscillator analysis (.HBOSC) adds the fundamental
frequency of oscillation to the list of unknown circuit quantities. To
accommodate the extra unknown, the phase (or equivalently, the imaginary
part) of one unknown variable (generally a node voltage) is set to zero. The
phases of all circuit quantities are relative to the phase, at this reference node
(referred to as the “PROBENODE”).
Additionally, the HBOSC analysis tries to avoid the “degenerate solution,”
where all non-DC quantities are zero. Although this is a valid solution of the
above equation (it is the correct solution, if the circuit does not oscillate),
HBOSC analysis might find this solution incorrectly, if the algorithm starts from
a bad initial solution.
The HBOSC analysis follows a technique similar to that described by Ngoya, et
al, which uses an internally-applied voltage probe to find the oscillation voltage
and frequency. The source resistance of this probe is a short circuit at the
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Chapter 9: Oscillator and Phase Noise Analysis
Harmonic Balance Oscillator Analysis (.HBOSC)
oscillation frequency, and an open circuit otherwise. HSPICE RF uses a twotier Newton approach to find a non-zero probe voltage, which results in zero
probe current.
The DC solution is used as a starting point for the HB analysis of the oscillator
circuit. In addition to the DC solution, initial values for both the oscillation
frequency and the probe voltage are needed. HBOSC analysis calculates the
small-signal admittance that the voltage probe sees over a range of
frequencies in an attempt to find potential oscillation frequencies. Oscillation is
likely to occur where the real part of the probe current is negative, and the
imaginary part is zero. You can use the FSPTS parameter to specify the
frequency search. You must also supply an initial guess for the large signal
probe voltage. A value of one-half the supply voltage is often a good starting
point.
Input Syntax for Harmonic Balance Oscillator Analysis
The input syntax for HBOSC analysis supports two different formats,
depending on whether the PROBENODE location is specified using a circuit
element (current source) or using the HBOSC PROBENODE parameters:
Syntax #1
.HBOSC TONE=F1 NHARMS=H1
+ PROBENODE=N1,N2,VP
+[FSPTS=NUM, MIN, MAX] [STABILITY=(-2|-1|0|1|2)]
+[SWEEP PARAMETER_SWEEP] [SUBHARMS=I]
Syntax #2 (Uses current source to set PROBENODE)
ISRC N1 N2 HBOSCVPROBE=VP
.HBOSC TONE=F1 NHARMS=H1
+[FSPTS=NUM, MIN, MAX] [STABILITY=(-2|-1|0|1|2)]
+[SWEEP PARAMETER_SWEEP] [SUBHARMS=I]
Parameter
Description
TONE
Approximate value for oscillation frequency (Hz). The search for an exact
oscillation frequency begins from this value, unless you specify an FSPTS
range or transient initialization (see HB Simulation of Ring Oscillators on
page 216 for more information).
NHARMS
Number of harmonics to use for oscillator HB analysis.
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Chapter 9: Oscillator and Phase Noise Analysis
Harmonic Balance Oscillator Analysis (.HBOSC)
Parameter
Description
PROBENODE
Circuit nodes that are probed for oscillation conditions.
■
N1 and N2 are the positive and negative nodes for a voltage probe
inserted in the circuit to search for oscillation conditions.
■
VP is the initial probe voltage value (one-half the supply voltage is a
suggested value).
The phase of the probe voltage is forced to zero; all other phases are relative
to the probe phase. HSPICE RF uses this probe to calculate small-signal
admittance for the initial frequency estimates. It should be connected near
the “heart” of the oscillator (near resonators, inside the ring of a ring
oscillator, etc.).
Note: The PROBENODE pins and approximate voltage value can also be set
by using a zero amp current source that uses the HBOSCVPROBE keyword.
HBOSCVPROBE= Sets PROBENODE parameters with a separate current source element. If a
VP
current source with HBOSCVPROBE is used, the PROBENODE parameter
and its values need not be included in the .HBOSC command.
FSPTS
Specifies the frequency search points that HSPICE RF uses in its initial
small-signal frequency search to find an oscillation frequency. Optional, but
recommended for high-Q and most LC oscillators. If the circuit is a ring
oscillator, see HB Simulation of Ring Oscillators on page 216 for more
information on how to use the HBTRANINIT option.
■
NUM is an integer.
MIN and MAX are frequency values in units of Hz.
If the FSPTS analysis finds an approximate oscillation frequency, the TONE
parameter may be ignored.
■
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Harmonic Balance Oscillator Analysis (.HBOSC)
Parameter
Description
STABILITY
When used with FSPTS, activates the additional oscillator stability analyses
depending on the following values:
■
■
■
■
■
0: A single point oscillator frequency-search stability analysis is
performed. The FSPTS search is executed, and the first successful linear
oscillation frequency value found is used as the starting point for the twotier Newton nonlinear oscillator analysis. The probenode vp value
specified is used as the starting amplitude for the Newton solver.
1: (default) A single point oscillator frequency-search stability analysis,
plus an estimate of oscillator amplitude, is performed. The FSPTS search
is executed, and the first successful linear oscillation frequency value
found is used as the starting point for the two-tier Newton nonlinear
oscillator analysis. An additional analysis for automatically estimating the
probenode amplitude is also performed, and this value is used as the
starting amplitude for the two-tier Newton solver.
–1: A single point oscillator frequency-search stability analysis, plus an
estimate of oscillator amplitude, is performed. The FSPTS search is
executed, and the first successful linear oscillation frequency value found
is accurately computed and reported. An additional analysis for
automatically estimating the probenode amplitude is also performed, and
this value is also reported. The analysis aborts without attempting the
two-tier Newton nonlinear oscillator analysis. By using STABILITY=–1, a
check can be made if any linear oscillation conditions are found, before
attempting the nonlinear oscillator analysis.
2: A multi-point frequency-search stability analysis is performed. The
FSPTS search is executed, and all successful linear oscillation frequency
values found over the entire FSPTS search range are reported. For each
potential oscillation frequency found, an additional analysis for estimating
the probenode amplitude is also performed. All frequency and amplitude
values are reported. The frequency value that has the largest predicted
amplitude is used as the starting point for the two-tier Newton nonlinear
oscillator analysis.
–2: A multi-point frequency-search stability analysis is performed. The
FSPTS search is executed, and all successful linear oscillation frequency
values found over the entire FSPTS search range are reported. For each
potential oscillation frequency found, an additional analysis for estimating
the probenode amplitude is also performed. All frequency and amplitude
values are reported. The analysis aborts without attempting the two-tier
Newton nonlinear oscillator analysis. By using STABILITY=–2, a check
can be made if any linear oscillation conditions are found, before
attempting the nonlinear oscillator analysis.
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Harmonic Balance Oscillator Analysis (.HBOSC)
Parameter
Description
SWEEP
Specifies the type of sweep. You can sweep up to three variables. You can
specify either LIN, DEC, OCT, POI, SWEEPBLOCK, DATA, OPTIMIZE, or
MONTE. Specify the nsteps, start, and stop frequencies using the following
syntax for each type of sweep:
■
■
■
■
■
■
■
■
SUBHARMS
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA=dataname
OPTIMIZE=OPTxxx
MONTE=val
Allows subharmonics in the analysis spectrum. The minimum non-DC
frequency in the analysis spectrum is f/subharms, where f is the frequency
of oscillation. Use this option if your oscillator circuit includes a divider or
prescaler that will result in frequency terms that are subharmonics of the
fundamental oscillation frequency
Note:
You can specify either .OPTION HBTRANPTS or .OPTION HBTRANSTEP,
but not both.
Simulation Strategies for Harmonic Oscillators
Since harmonic oscillators tend to have high Q, they are more sensitive than
ring oscillators to frequency and amplitude guess. The high Q also means that
HBTRANINIT is not usually helpful, because it takes too long for transient
simulation to settle close enough to steady state. For these oscillators, FSPTS
tends to work well, because the behavior is close to linear.
The recommended setup for harmonic oscillators is:
214
■
Set up .HBOSC with FSPTS. Higher Q oscillators may need more FSPTS
points.
■
Choose a node directly connected to the oscillator or crystal as the
PROBENODE.
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Chapter 9: Oscillator and Phase Noise Analysis
Harmonic Balance Oscillator Analysis (.HBOSC)
■
The number of harmonics specified by nharms can be fairly small, perhaps
10, unless some digital circuitry is included in the simulation.
■
Do not use HBTRANINIT.
Because the challenges related to ring oscillators and harmonic oscillators
are so different, we approach them with separate simulation strategies.
.HBOSC Examples
Example 1
.HBOSC tone=900MEG nharms=9 probenode=gate,gnd,0.65
Performs an oscillator analysis, searching for frequencies in the vicinity of 900
MHz. This example uses nine harmonics with the probe inserted between the
gate and gnd nodes. The probe voltage estimate is 0.65 V.
Example 2
.HBOSC tone=2400MEG nharms=11
+ probenode=drainP,drainN,1.0 fspts=20,2100MEG,2700MEG
Performs an oscillator analysis, searching for frequencies in the vicinity of 2.4
GHz. This example uses 11 harmonics with the probe inserted between the
drainP and drainN nodes. The probe voltage estimate is 1.0 V.
Example 3
Another means to define the probenode information is through a zero-current
source. The following two methods define an equivalent .HBOSC command:
■
Method 1:
.HBOSC tone = 2.4G nharms = 10
+ probenode = drainP, drainN, 1.0
+ fspts = 20, 2.1G, 2.7G
■
Method 2:
ISRC drainP drainN 0 HBOSCVPROBE = 1.0
.HBOSC tone = 2.4G nharms = 10
+ fspts = 20, 2.1G, 2.7G
In Method 2, the PROBENODE information is defined by a current source in the
circuit. Only one such current source is needed, and its current must be 0.0
with the HBOSC PROBENODE voltage defined through its HBOSCVPROBE
property.
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Chapter 9: Oscillator and Phase Noise Analysis
Harmonic Balance Oscillator Analysis (.HBOSC)
HB Simulation of Ring Oscillators
Ring oscillators require a slightly different simulation approach in HB. Since
their oscillation is due to the inherent delay in the inverters of the ring, they are
best modeled in the time domain and not in the frequency domain.
In addition, ring oscillator waveforms frequently approach square waves, which
require a large number of harmonics to be described in the frequency domain.
An accurate initial guess is important if they are going to be simulated
accurately with HB.
The HSPICE RF HBOSC analysis typically starts from the DC solution and
looks for potential resonances in the linear portion of the circuit to determine
the initial guess for the oscillation frequency. However, these resonances
generally do not exist in ring oscillators, which do not contain linear resonant
elements.
HB analysis provides a second method of obtaining a good initial guess for the
oscillation frequency, which is specifically intended for ring oscillators. Instead
of starting from the results of a DC analysis, this method starts from the result
of a transient analysis. This method is called Transient Initialization and also
provides a good initial guess for all the voltages and currents in the circuit.
The recommended setup for ring oscillators is therefore:
216
■
Set up .HBOSC without FSPTS.
■
Choose one of the nodes in the ring as the PROBENODE.
■
Since ring oscillators tend to have square-wave-like output signals which
have significant high frequency content, a relatively large value, perhaps 50,
for nharms is recommended. Ring oscillators with more stages tend to need
more harmonics.
■
Set HBTRANINIT to a value that represents ~5-10 oscillator periods, and
make sure that you include an .ic command or other transient analysis setup
to start the oscillator in transient simulation. Longer HBTRANINIT times may
result in faster HBOSC convergence, at the expense of additional CPU time
spent on HBTRANINIT.
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Chapter 9: Oscillator and Phase Noise Analysis
HBOSC Analysis Using Transient Initialization
HBOSC Analysis Using Transient Initialization
To perform an HBOSC analysis, use the following options in your HSPICE RF
netlist.
Table 18
HBOSC Analysis Options for Transient Initialization
Keyword
Description
HBTRANINIT = time
Tells HB to use transient analysis to initialize all
state variables. time is when the circuit has
reached (or is near) steady-state. Default = 0.
HBTRANPTS = npts
npts specifies the number of points per period for
converting the time-domain data results from
transient analysis, into the frequency domain.
npts must be an integer greater than 0. The units
are in nharms (nh). Default=4*nh.
This option is relevant only if you set .OPTION
HBTRANINIT.
HBTRANSTEP = stepsize
stepsize specifies the step size for the transient
analysis.
The default is 1/(4*nh*f0), where nh is the
nharms value and f0 is the oscillation frequency.
This option is relevant only if you set .OPTION
HBTRANINIT.
HBTRANFREQSEARCH = 1|0
If HBTRANFREQSEARCH=1 (default), then HB
analysis calculates the oscillation frequency
from the transient analysis.
Otherwise, HB analysis assumes that the period
is 1/f, where f is the frequency specified in the
tones description.
You must also either specify the initial conditions or add a PWL or PULSE
source to start the oscillator for transient analysis. This source should provide a
brief stimulus, and then return to zero. HB analysis effectively ignores this type
of source, treating it as zero-valued.
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Chapter 9: Oscillator and Phase Noise Analysis
HBOSC Analysis Using Transient Initialization
This method does the following:
1. If HBTRANFREQSEARCH=1, transient analysis runs for several periods,
attempting to determine the oscillation frequency from the probe voltage
signal.
2. Transient analysis continues until the time specified in HBTRANINIT.
3. Stores the values of all state variables over the last period of the transient
analysis.
4. Transforms the state variables to the frequency domain by using a Fast
Fourier Transform (FFT) to establish an initial guess for HB oscillator
analysis.
5. Starts the standard HB oscillator analysis.
Additional .HBOSC Analysis Options
Oscillator analysis will make use of all standard HB analysis options as listed in
the following table. In addition, the following options are specifically for
oscillator applications.
Table 19
218
HBOSC Analysis Options for Oscillator Applications
Parameter
Description
HBFREQABSTOL
An additional convergence criterion for oscillator analysis.
HBFREQABSTOL is the maximum absolute change in
frequency between solver iterations for convergence.
Default is 1 Hz.
HBFREQRELTOL
An additional convergence criterion for oscillator analysis.
HBFREQRELTOL is the maximum relative change in
frequency between solver iterations for convergence.
Default is 1.e-9.
HBPROBETOL
HBOSC analysis tries to find a probe voltage at which the
probe current is less than HBPROBETOL. This option
defaults to the value of HBTOL, which defaults to 1.e-9.
HBMAXOSCITER
Maximum number of outer-loop iterations for HBOSC
analysis. It defaults to 10000.
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Chapter 9: Oscillator and Phase Noise Analysis
HBOSC Analysis Using Transient Initialization
.HBOSC Output Syntax
The output syntax for .HBOSC analysis is identical to that for HB analysis (see
Chapter 7, Steady-State Harmonic Balance Analysis). To output the final
frequency of oscillation, use the HERTZ keyword. For example, HERTZ[1]
identifies the fundamental frequency of oscillation.
Note:
For PROBENODE = n1 n2 vp, where vp is a voltage, units must be given
in volts.
See also Outputting Phase Noise Source as ASCII Data Files Using *.printpn0.
Troubleshooting Convergence Problems
This section lists the most common causes of convergence problems, how to
recognize them, and resolve them.
The HSPICE RF harmonic balance oscillator analysis is a two-tier iterative
analysis, consisting of “inner loop” and “outer loop” iterations. In the outer loop
iteration, HBOSC will iterate to reduce the “probe error” which is reported for
each outer loop iteration. Each outer loop iteration involves a non-autonomous
Harmonic Balance (HB) circuit solution; this non-autonomous solve is referred
to as the inner loop iteration.
If HBOSC has inner loop convergence problems, the simulation may get stuck
on the first outer loop iteration or you may see warning messages such as:
Warning: HB_WARN.3: Final HB residual value > HB_TOL.
Rank of HB Jacobian = 155
Warning: HB_WARN.3: HB convergence failure in non-autonomous HB.
For each outer loop iteration, the probe voltage and probe frequency are listed.
If an outer loop convergence problem occurs, you may see the following:
■
Decreasing probe voltage values.
■
Wildly fluctuating values of probe frequency.
Osc probe : voltage = 0.218234 frequency =
6.240794122744832e+09
■
A warning message which indicates that the oscillator simulation has
reached a non-oscillating DC solution.
Warning: HB_ERR.18: HB oscillator analysis has reached the
NULL solution.
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Chapter 9: Oscillator and Phase Noise Analysis
HBOSC Analysis Using Transient Initialization
General Convergence Issues
Probe Node Location
Since convergence is sensitive to the probe node location, convergence
problems can often be tracked to this setting.
A common scenario is that the oscillator's output signal is passed through one
or more buffers, and the designer may think of the buffer output as the oscillator
output. A frequent mistake is to place the probe node at the output of the buffer,
but in fact, this will cause HB convergence problems. In this case, the probe
node should be moved to the oscillator part of the circuit. Often, it is necessary
to select an internal node of a subcircuit to achieve this.
The most typical symptom of this problem is inner loop convergence failure. As
mentioned above, for ring oscillators, always choose a node that is part of the
ring, i.e., connecting two stages of the ring; for harmonic oscillators, choose a
node close to the oscillator.
Incorrect Source Values
If the original netlist was set up to simulate the oscillator in transient analysis,
some voltage or current sources may have transient descriptions (e.g., PWL) in
order to start the oscillator. For example, a voltage supply may be ramped to
simulate a power-up to start the oscillator:
Vvdd vdd 0 PWL (0 0 1n 3)
In this case, the user would like HBOSC to use 3 as the voltage source value,
but HBOSC will use 0 because the explicit DC value of the source is used for
Harmonic Balance. HSPICE RF tries to interpret your sources intelligently but,
in some cases it may not be able to determine what you intended.
For the above example, there are a few ways to ensure that HSPICE RF
correctly interprets the source.
■
Remove the explicit DC value. If only a transient description is given, HB will
use the time=infinity value of the source.
■
Add TRANFORHB=1
Vvdd vdd 0 PWL (0 0 1n 3) TRANFORHB=1
The TRANFORHB=1 keyword will cause HB to use the transient analysis
description in HB and HBOSC.
■
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Add an explicit HB value
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HBOSC Analysis Using Transient Initialization
Vvdd vdd 0 PWL (0 0 1n 3) HB 3 0 0
This causes HB to treat the source as a 3V DC source (0th harmonic is
specified). If a HB value is given, HSPICE RF will ignore the PWL
description and use “HB 3 0 0” (amplitude=3, phase=0, harmonic=0)
instead. The PWL description will still be used for HBTRANINIT.
Incorrect source values usually result in the following:
■
High residual value after HBTRANINT. Usually, HBTRANINIT should
produce a good starting point for HB or HBOSC. Typical residuals after
HBTRANINIT are 1e-4 or 1e-5. If the initial residual printed immediately
after HBTRANINIT is done is high, there may be a source problem. In the
VDD ramping example above, you might see a residual value of 3.Outer
loop may converge to DC solution because incorrect source values result in
a non-oscillatory circuit:
Warning: HB_ER.18: HB oscillator analysis has reached the NULL
solution.
■
In some cases, inner loop non-convergence may occur.
GMRES Convergence
When the default value for .option HBSOLVER (=1) is set, HSPICE RF uses a
GMRES iterative solver to solve the linear systems that arise on each inner
loop Newton-Raphson step. If GMRES does not solve the linear systems
accurately enough, then the inner loop may not converge.
The GMRES solver is controlled by two options:
■
HBKRYLOVTOL: relative tolerance for GMRES solver. Default is 0.01, or
1%. For some circuits, setting this option will help inner loop convergence:
.option HBKRYLOVTOL=1e-3
■
HBKRYLOVDIM: dimension of Krylov subspace used in GMRES iteration.
Also controls maximum number of GMRES iterations. In HSPICE RF's .lis
file, the number of GMRES iterations taken for each Newton-Raphson step
is listed. If that number is equal to HBKRYLOVDIM, convergence may be
improved by increasing HBKRYLOVDIM. Example:
.option HBKRYLOVDIM=80
The symptom for GMRES convergence difficulty is always inner loop
convergence failure, or slow inner loop convergence. If this problem occurs, the
inner loop convergence is often good until the residual reaches a fairly low
value like 1e-8 or 1e-7, and then stagnates.
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HBOSC Analysis Using Transient Initialization
Accuracy of Initial Guess
Both inner loop and outer loop convergence improves significantly if the
starting point or initial guess of the iterative method is good.
Outer Loop Convergence
For outer loop convergence, the initial guess consists simply of the oscillation
frequency and first harmonic amplitude at the probe node location. If inner loop
convergence is successful but outer loop convergence is not, then a better
frequency or amplitude guess may be needed.
If HBTRANINIT is being used, then the accuracy of the initial guess can be
improved by one of the following methods:
■
Increase the HBTRANINIT time, simply by increasing the value of the
HBTRANINIT option.
■
Increase the HBTRANINIT accuracy. You can increase the transient
analysis accuracy by setting .option DELMAX or .option SIM_ACCURACY.
For example, you may set
.option SIM_ACCURACY=10 HBTOL=1e-8
Note that SIM_ACCURACY will simultaneously tighten transient and HB
accuracy tolerances. If you want HB accuracy to remain unaffected, you
may also want to set HBTOL as in the example above.
■
Increase accuracy of time domain to frequency domain conversion of
HBTRANINIT results, by increasing HBTRANPTS or equivalently,
decreasing HBTRANSTEP. For example:
.option HBTRANSTEP=1p
If FSPTS is being used, you can increase the number of points. Sometimes, it
is best to supply a guess manually by removing FSPTS and adjusting the
TONES value.
If HBTRANINIT is not used, you may be able to improve convergence by
manually adjusting the PROBENODE amplitude guess.
To evaluate the effectiveness of your option settings, look at the “probe error”
reported after the first outer loop iteration:
Iteration 1
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HBOSC Analysis Using Transient Initialization
Osc probe : voltage = 0.2 frequency = 5.980000000000000e+09
hb residual = 7.628260e-10
Rank of HB Jacobian = 9102
Probe error = 0.000154462
dv = -0.0411324 df =-2.30464430e+08
A smaller probe error value indicates a better initial guess.
Inner Loop Convergence
If inner loop convergence is a problem, it may be because the initial voltage
waveform values are not close to the solution. The only way to improve the
voltage values is by using HBTRANINIT. While this will not work well for
harmonic oscillators, it does work well for ring oscillators. You can improve the
accuracy of HBTRANINIT as described in the outer loop convergence section
above.
If the initial residual is large after HBTRANINIT, you may want to check to make
sure that the voltage and current sources are consistent between HB and
transient analysis.
Insufficient Number of Harmonics
If the number of harmonics specified is too small to represent the signals
present in the circuit, you may see either convergence problems in either the
inner or outer loop, or the solution may converge to an unreasonable frequency
value.
It is difficult to know when the number of harmonics is insufficient, but if
suspected, it is a simple experiment to increase the value of NHARMS. If
convergence was achieved and the number of harmonics is large enough, then
the magnitude of the spectral data for all signals should significantly decay with
increasing frequency. If the spectral data for node voltages has not decayed at
the highest harmonics included in the simulation, an increase in the value of
NHARMS is recommended.
Presence of Frequency Divider
If a frequency divider is present and not accounted for by the SUBHARMS
setting, convergence is not possible because the Harmonic Balance spectrum
does not include the necessary low frequency components. As a result, you will
encounter inner loop convergence failure. When debugging HBOSC
convergence problems, it is necessary to rule out the possibility of presence of
frequency dividers early in the process.
If a frequency divider is present, you can simulate the circuit if you set
SUBHARMS to the largest frequency division present in the circuit. If a
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Chapter 9: Oscillator and Phase Noise Analysis
Oscillator Analysis Using Shooting Newton (.SNOSC)
frequency divider is present, it is almost always necessary to use the
HBTRANINIT option to achieve convergence.
To get optimal performance, it is recommend that you set
.option HBSOLVER=2
This activates a hybrid time/frequency-domain preconditioner which is
particularly effective on frequency dividers.
Oscillator Analysis Using Shooting Newton (.SNOSC)
The analysis described in Chapter 8, Steady-State Shooting Newton Analysis
also provides a very effective means for finding the steady-state for oscillator
circuits.
Ring oscillators are best suited for time domain analysis using Shooting
Newton because they tend to:
■
have a low Q
■
operate based on digital delays
■
have strongly nonlinear behavior
■
output signals that are piece-wise-linear or square-wave-like
HBOSC is superior for sinusoidal waveforms. As with the Harmonic Balance
approach, the goal is to solve for the additional unknown oscillation frequency.
This is accomplished in Shooting Newton by considering the period of the
waveform as an additional unknown, and solving the boundary conditions at
the waveform endpoints that coincide with steady-state operation. As with
regular Shooting Newton analysis, input may be specified in terms of time or
frequency values.
Syntax #1
.SNOSC TONE=F1 NHARMS=H1 [TRINIT=Ti] OSCNODE=N1
+[MAXTRINITCYCLES=N][SWEEP PARAMETER_SWEEP]
Syntax #2
.SNOSC TRES=Tr PERIOD=Tp [TRINIT=Tr] OSCNODE=N1
+[MAXTRINITCYCLES=I] SWEEP PARAMETER_SWEEP
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Oscillator Analysis Using Shooting Newton (.SNOSC)
Parameter
Description
TONE
Approximate value for oscillation frequency (Hz). The search
for an exact oscillation frequency begins from this value.
NHARMS
Number of harmonics to be used for oscillator SN analysis.
OSCNODE
Node used to probe for oscillation conditions. This node is
automatically analyzed to search for periodic behavior near
the TONE or PERIOD value specified.
TRINIT
This the transient initialization time. If not specified, the
transient initialization time will be equal to the period (for
Syntax 1) or the reciprocal of the tone (for Syntax 2). For
oscillators, we recommend specifying a transient initialization
time since the default initialization time is usually too short to
effectively stabilize the circuit.
MAXTRINITCYCLES
Stops SN stabilization simulation and frequency detection
when the simulator detects that MAXTRINITCYCLES have
been reached in the oscnode signal, or when time=trinit,
whichever comes first. Minimum cycles is 1. The
MAXTRINITCYCLES parameter is optional.
TRES
TRES is the time resolution to be computed for the steadystate waveforms (in seconds). The period of the steady-state
waveform may be entered either as PERIOD or its reciprocal,
TONE.
PERIOD
PERIOD is the expected period T (seconds) of the steadystate waveforms. Enter an approximate value when using for
oscillator analysis.
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Oscillator Analysis Using Shooting Newton (.SNOSC)
Parameter
Description
SWEEP
Specifies the type of sweep. You can sweep up to three
variables. You can specify either LIN, DEC, OCT, POI,
SWEEPBLOCK, DATA, OPTIMIZE, or MONTE. SWEEP is an
optional parameter. Specify the nsteps, start, and stop
frequencies using the following syntax for each type of sweep:
■
■
■
■
■
■
■
■
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA=dataname
OPTIMIZE=OPTxxx
MONTE=val
Example 1
.SNOSC tone=900MEG nharms=9 trinit=10n oscnode=gate
Performs an oscillator analysis, searching for periodic behavior after an initial
transient analysis of 10 ns. This example uses nine harmonics while searching
for an oscillation at the gate node.
Example 2
.SNOSC tone=2400MEG nharms=11 trinit=20n oscnode=drainP
Performs an oscillator analysis, searching for frequencies in the vicinity of 2.4
GHz. This example uses 11 harmonics and a search at the drainP.
.SNOSC Output Syntax
The output syntax for .SNOSC analysis is identical to that for SN analysis (see
Chapter 8, Steady-State Shooting Newton Analysis). To output the final
frequency of oscillation, use the HERTZ keyword. For example, HERTZ[1]
identifies the fundamental frequency of oscillation.
See also Using Noise Analysis Results as Input Noise Sources.
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Chapter 9: Oscillator and Phase Noise Analysis
Phase Noise Analysis (.PHASENOISE)
Phase Noise Analysis (.PHASENOISE)
The following topics are covered in this section:
■
Phase Noise Analysis Overview
■
Identifying Phase Noise Spurious Signals
■
PHASENOISE Input Syntax
■
Phase Noise Algorithms
■
PHASENOISE Output Syntax
■
Phase Noise Analysis Options
■
Measuring Phase Noise with .MEASURE PHASENOISE
■
Amplitude Modulation/Phase Modulation Separation
Phase Noise Analysis Overview
Phase Noise analysis requires first running either harmonic balance (HBOSC)
or Shooting Newton (SNOSC) analysis, and then PHASENOISE analysis. The
PHASENOISE analysis itself is identical whether you run SNOSC or HBOSC.
Figure 25 on page 227 shows a simple free-running oscillator, which includes a
port with injected current.
+
in
v(t)
-
Figure 25
Oscillator with Injected Current
An ideal oscillator would be insensitive to perturbations with a fixed amplitude,
frequency, and phase represented by:
Equation 31
v ( t ) = A cos [ ω0 t + φ0 ]
A noisy oscillator has amplitude and phase fluctuations we can write as:
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Chapter 9: Oscillator and Phase Noise Analysis
Phase Noise Analysis (.PHASENOISE)
Equation 32
v ( t ) = A ( t ) cos [ ω0 t + φ( t ) ]
In the preceding equation:
■
A(t) is the time varying amplitude for the noisy oscillator.
■
φ( t ) is the time varying phase for the noisy oscillator.
■
ω0 is the frequency of oscillation.
In most applications, the phase noise is of particular interest, because it
represents frequency fluctuations about the fundamental, which you cannot
remove. These fluctuations are random processes, and are typically expressed
in terms of their power spectral density. For most oscillators, the phase noise is
a low-frequency modulation that creates sidebands in the oscillator’s spectrum,
about ω0 .
For example, the following equation represents a simple sinusoidal variation in
the phase:
Equation 33
v ( t ) = A cos [ ω0 t + θ p sin ωm t ]
■
θ p is the peak phase deviation, specified as θ p = Δω ⁄ ωm
■
Δω is the peak angular frequency deviation.
For θ p « 1 , the following equation approximates the output:
Equation 34
θ
⎧
⎫
v ( t ) = A ⎨ cos ( ω0 t ) – ----p- [ cos ( ω0 + ωp ) t – cos ( ω0 + ωm ) t ] ⎬
2
⎩
⎭
That is, when the peak phase deviation is small, the result is frequency
components on each side of the fundamental with amplitude θ p ⁄ 2 .
The Single-Sideband Phase Noise L ( f m ) is the ratio of noise power to carrier
power in a 1Hz bandwidth, at offset ωm = 2πf m , which in this case can be
written as:
Equation 35
V sb
L ( f m ) = ⎛ --------⎞
⎝ A⎠
2
2
2
θp
θrms
= ---------- = --------------4
2
This model for oscillator noise shows that sidebands about the fundamental,
due to noise, are directly related to the spectrum of the phase fluctuations
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Chapter 9: Oscillator and Phase Noise Analysis
Phase Noise Analysis (.PHASENOISE)
θ ( t ) . The more general definition of phase noise relates it to the spectral
density of phase fluctuations, i.
Equation 36
θ2
S φ( ωm ) = -----p- = 2L ( f )
m
2
HSPICE RF uses several sophisticated analysis techniques for computing the
power spectrum of the phase variations to yield the phase noise response. This
information can be used to predict the spectrum of the oscillator about the
fundamental frequency, and also used to predict its random jitter
characteristics.
Any .PHASENOISE analysis will result in the calculation of a curve fit for a
power-law model according to:
Equation 37
⎧
⎫
L ( f ) = 10 ⋅ log ⎨ a3
------ + a2
------ + a1
------ + a0 ⎬
f
⎩ f3 f2
⎭
The coefficients a3 , a2 , a1 , and a0 are reported in the .lis file.
The .lis file includes a table which models the phase noise, including the
behavioral model fit and its fit error.
|-------------------------------------------------------------|
| L(f) = 10*log10( a3/f^(2+ef) + a2/f^2 + a1/f^(ef) + a0 ) dBc/Hz|
| a3 = 0.000000e+00
|
| a2 = 3.165111e-02
|
| a1 = 0.000000e+00
|
| a0 = 0.000000e+00
|
| ef = 1.000000e+00
|
| Average fit error = 1.6185e+00 dB
|
| Maximum fit error = 8.4862e+00 dB @ 1.0000e+07 Hz
|
Identifying Phase Noise Spurious Signals
Realistic phase noise responses include spurs. Spurs are contributions to the
phase noise that result from deterministic signals present within the circuit. In
most cases, the spurs are very small signals and do not interfere with the
steady-state operation of the oscillator, but do add energy to the output
spectrum of the oscillator. The energy that the spurs adds may need to be
included in jitter measurements. The phase noise spurs feature adds an
additional analysis option that can predict the spurious contributions to the
jitter.
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Chapter 9: Oscillator and Phase Noise Analysis
Phase Noise Analysis (.PHASENOISE)
To activate the new phase noise spur analysis, use the SPURIOUS keyword in
the .PHASENOISE command. An additional .HBAC analysis is performed that
predicts the spurious contributions to the phase noise.
A voltage or current source can be used to add spurious signals to an oscillator
circuit. The keyword SPUR identifies the spurious signal.
Syntax
Vxxxx n1 n2 … [SPUR mag phase freq] …
Ixxxx n1 n2 … [SPUR mag phase freq] …
Where,
■
mag is the amplitude in volts or amps
■
phase is the phase in degrees
■
freq is the frequency in Hz
The source is equivalent to a steady-state sinusoidal source at the specified
amplitude, phase, and frequency values. It is only used for the spurious
analysis and is ignored by all other analyses. The SPUR keyword is can be
combined into a source used for other analyses. It is recommended, however,
that SPUR sources be added as separate sources.
PHASENOISE Input Syntax
.PHASENOISE output frequency_sweep [method=0|1|2]
+ [carrierindex=int] [listfreq=(frequencies|none|all)]
+ [listcount=val] [listfloor=val] [listsources=on|off]
+ [spurious=0|1]
230
Parameter
Description
output
An output node, pair of nodes, or 2-terminal element. HSPICE RF
references phase noise calculations to this node (or pair of nodes).
Specify a pair of nodes as V(n+,n-). If you specify only one node, V(n+),
then HSPICE RF assumes that the second node is ground. You can
also specify a 2-terminal element.
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Phase Noise Analysis (.PHASENOISE)
Parameter
Description
frequency_sweep
A sweep of type LIN, OCT, DEC, POI, or SWEEPBLOCK. Specify the
type, nsteps, and start and stop time for each sweep type, where:
■
type = Frequency sweep type, such as OCT, DEC, or LIN.
nsteps = Number of steps per decade or total number of steps.
■
start = Starting frequency.
■
stop = Ending frequency.
The four parameters determine the offset frequency sweep about the
carrier used for the phase noise analysis.
■
LIN type nsteps start stopOCT type nsteps start stopDEC type nsteps
start stopPOI type nsteps start stopSWEEPBLOCK freq1 freq2 ... freqn
■
method
METHOD=0 (default) selects the Nonlinear Perturbation (NLP)
algorithm, which is used for low-offset frequencies.
■
METHOD=1 selects the Periodic AC (PAC) algorithm, which is used
for high-offset frequencies.
■
METHOD=2 selects the Broadband Phase Noise (BPN) algorithm,
which you can use to span low and high offset frequencies.
You can use METHOD to specify any single method. See the section on
Phasenoise Algorithms below for a more detailed discussion on using
the METHOD parameter.
carrierindex
Optional. Specifies the harmonic index of the carrier at which HSPICE
RF computes the phase noise. The phase noise output is normalized to
this carrier harmonic. Default=1.
listfreq
Dumps the element phase noise value to the .lis file. You can specify
which frequencies the element phase noise value dumps. The
frequencies must match the sweep_frequency values defined in the
parameter_sweep, otherwise they are ignored.
In the element phase noise output, the elements that contribute the
largest phase noise are dumped first. The frequency values can be
specified with the NONE or ALL keyword, which either dumps no
frequencies or every frequency defined in the parameter_sweep.
Frequency values must be enclosed in parentheses. For example:
listfreq=(none)
listfreq=(all)
listfreq=(1.0G)
listfreq=(1.0G, 2.0G)
The default value is the first frequency value.
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Phase Noise Analysis (.PHASENOISE)
Parameter
Description
listcount
Dumps the element phase noise value to the .lis file, which is sorted
from the largest to smallest value. You do not need to dump every noise
element; instead, you can define listcount to dump the number of
element phase-noise frequencies. For example, listcount=5 means that
only the top 5 noise contributors are dumped. The default value is 20.
listfloor
Dumps the element phase noise value to the .lis file and defines a
minimum meaningful noise value (in dBc/Hz units). Only those
elements with phase-noise values larger than the listfloor value are
dumped. For example, listfloor=-200 means that all noise values below
-200 (dBc/Hz) are not dumped. The default value is -300 dBc/Hz.
listsources
Dumps the element phase-noise value to the .lis file. When the element
has multiple noise sources, such as a level 54 MOSFET, which contains
the thermal, shot, and 1/f noise sources. When dumping the element
phase-noise value, you can decide if you need to dump the contribution
from each noise source. You can specify either ON or OFF: ON dumps
the contribution from each noise source and OFF does not. The default
value is OFF.
spurious
Selects phase noise spur analysis
0 = No spurious analysis (default)
1 = Activates additional SPUR source analysis
Phase Noise Algorithms
HSPICE RF provides three algorithms for oscillator phasenoise: nonlinear
perturbation, periodic AC, and broadband calculations. These algorithms are
selected by setting the METHOD parameter to 0, 1, or 2, respectively.
Each algorithm has their regions of validity and computational efficiency, so
some thought is necessary to obtain meaningful results from a PHASENOISE
simulation. For each algorithm, the region of validity depends on the particular
circuit being simulated. However, there are some general rules that can be
applied to oscillator types (that is, ring or harmonic) so that a valid region can
be identified. And there are techniques that can be used to check validity of
your simulation results.
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Phase Noise Analysis (.PHASENOISE)
Nonlinear Perturbation Algorithm
The nonlinear perturbation (NLP) algorithm, which is the default selection, is
typically the fastest computation, but is valid only in a region close to the carrier.
Generally, you will want to use this algorithm if you interested in phasenoise
close to the carrier and do not need to determine a noise floor. NLP
computation time is almost independent of the number of frequency points in
the phasenoise frequency sweep.
Periodic AC Algorithm
The periodic AC (PAC) algorithm is valid in a region away from the carrier and is
slower than the NLP algorithm. The PAC algorithm is used for getting
phasenoise in the far carrier region and when you need to determine a noise
floor.
The computation time for the PAC algorithm is approximately linearly
dependent on the number of frequency points in the phasenoise frequency
sweep. If you are using the PAC algorithm, you should try to minimize the
number of points in the sweep.
Another issue is that the PAC algorithm becomes more ill-conditioned as you
approach the carrier. This means that you may have to generate a steady-state
solution with more harmonics to get an accurate simulation as you get closer to
the carrier. So, if you find that the PAC is rolling off at close-in frequencies, you
should rerun HB analysis with a larger number of harmonics. Although,
typically, you will not see improvements in PAC accuracy beyond more than
about 100-200 harmonics.
Early in your testing, the best way to verify that NLP and PAC are giving
accurate results is to run both algorithms over a broad frequency range and
check that the curves have some range in frequency where they overlap.
Typically, you will see the NLP curve rolling off at 20 to 30 dB/decade as
frequency increases, characteristic of white noise or 1/f noise behavior. Also,
the PAC curve will at first be flat or even noisy close to the carrier. At some point
though, you will see this curve match the NLP roll-off.
The lowest frequency at which the curves overlap defines the point, fPAC above
which the PAC algorithm is valid. Sometimes, by increasing the number of HB
harmonics, it is possible to move fPAC to lower frequencies. The highest
frequency at which the curves overlap defines the point, fNLP below which the
NLP algorithm is valid. A rough rule of thumb is that fPAC = fo/Q, where fo is the
carrier frequency and Q is the oscillator Q-value. This implies that for high-Q
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Chapter 9: Oscillator and Phase Noise Analysis
Phase Noise Analysis (.PHASENOISE)
oscillators, such as crystal and some harmonic oscillators, that PAC will be
accurate to values quite close to the carrier.
Broadband Phase Noise Algorithm
The Broadband Phase Noise (BPN) algorithm allows phase noise simulation
over a broad frequency range. The BPN algorithm runs both the NLP and PAC
algorithms and then connects them in the overlap region to generate a single
phase noise curve. This algorithm is ideal for verifying the NLP and PAC
accuracy regions and when you require a phase noise response over a broad
frequency range.
PHASENOISE Output Syntax
HSPICE RF supports the output of the phase noise as well as the phase noise
due to a specified element. In addition, you can output phase noise due to the
specified noise source types. In addition, you can use specialized keywords to
output phase noise due to the specified noise source types, as described
below.
Specified Element
.PRINT PHASENOISE phnoise phnoise(element_name)
.PROBE PHASENOISE phnoise phnoise(element_name)
In this syntax, phnoise is the phase noise parameter.
The .PHASENOISE statement outputs raw data to the *.pn# and *.printpn#
files. HSPICE RF outputs the phnoise data in decibels, relative to the carrier
signal, per hertz, across the output nodes in the .PHASENOISE statement. The
data plot is a function of the offset frequency. Units are in dBc/Hz.
■
If you use the NLP algorithm (METHOD=0) default, HSPICE RF calculates
only the phase noise component.
■
If you use the PAC algorithm (METHOD=1), HSPICE RF sums both the phase
and amplitude noise components to show the total noise at the output.
■
If you use the BPN algorithm (METHOD=2), HSPICE RF adds both the phase
and amplitude noise components together to show the total noise at the
output. HSPICE RF outputs phnoise to the .pn# file if you set .OPTION
POST.
Element phase noise can also be analyzed through the .PRINT and .PROBE
statements, which the previous syntax shows. A single phnoise keyword
specifies the phase noise for the whole circuit, and the
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Phase Noise Analysis (.PHASENOISE)
phnoise(element_name) specifies the phase-noise value of the specified
element.
Example 1
.HBOSC TONE=900MEG NHARMS=9
+ PROBENODE=gate,gnd,0.65
.PHASENOISE V(gate,gnd) DEC 10 100 1.0e7
+ METHOD=0 CARRIERINDEX=1 $use NLP algorithm
This example performs an oscillator analysis, searching for frequencies in the
vicinity of 900 MHz, followed by a phase noise analysis at frequency offsets
from 100 Hz to 10 MHz.
Example 2
.HBOSC TONE=2400MEG NHARMS=11
+ PROBENODE=drainP,drainN,1.0
+ FSPTS=20,2100MEG,2700MEG
+ SWEEP Vtune 0.0 5.0 0.2
.PHASENOISE V(drainP,drainN) DEC 10 100 1.0e7
+ METHOD=0 CARRIERINDEX=1 $use NLP algorithm
This example performs a VCO analysis, searching for frequencies in the vicinity
of 2.4 GHz. This example uses eleven harmonics, and sweeps the VCO tuning
voltage from 0 to 5 V. HSPICE RF uses the nonlinear perturbation (NLP)
algorithm to perform a phase noise analysis about the fundamental frequency
for each tuning voltage value.
Frequency-Dependent and Frequency-Independent Sources
The phnoise_fdep keyword variable will collect all frequency-dependent
noise sources' contributions to the phase noise.
The phnoise_findep keyword variable will collect all frequency independent
noise sources' contributions.
.print phasenoise phnoise_fdep
.print phasenoise phnoise_findep
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Phase Noise Analysis (.PHASENOISE)
Frequency and Bias Dependencies
■
The following syntax is frequency-independent and bias-dependent:
.print phasenoise phnoise_cyclo
Also acceptable is:
.print phasenoise phnoise_cyclostationary
Where:
cyclo or cyclostationary means anything bias-dependent.
■
The following syntax is frequency-independent and bias-independent:
.print phasenoise phnoise_stationary
■
The following syntax is bias-independent and frequency-dependent:
.print phasenoise phnoise_flicker
■
The following syntax is frequency-dependent and bias-dependent:
.print phasenoise phnoise_cycloflicker
Also acceptable is:
.print phasenoise phnoise_cyclostationaryflicker
■
The phnoise_fdep is a combination of phnoise_flicker and
phnoise_cycloflicker.
■
The phnoise_findep is a combination of phnoise_stationary and
phnoise_cyclostationary.
A Noise type suffix can be added to each of these noise terms, phnoise, la,
ltotal, onoise, to select specific noise-type components:
Table 20
236
Summary of Noise Type Dependences
Noise type
frequency-dependent
bias-dependent
phnoise_stationary
No
No
phnoise_cyclostationary
No
Yes
phnoise_flicker
Yes
No
phnoise_cycloflicker
Yes
Yes
phnoise_fdep
is the union of phnoise_Flicker and
phnoise_cycloflicker noise types
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Chapter 9: Oscillator and Phase Noise Analysis
Phase Noise Analysis (.PHASENOISE)
Table 20
Summary of Noise Type Dependences
Noise type
frequency-dependent
bias-dependent
phnoise_findep
is the union of phnoise_stationary and
phnoise_cyclostationary noise types
Where: Noise_term can be phnoise, la, ltotal, onoise (see Amplitude
Modulation/Phase Modulation Separation later in this chapter).
See also Using Noise Analysis Results as Input Noise Sources.
Phase Noise Analysis Options
Table 21 lists the control options specific to PHASENOISE applications.
Table 21
PHASENOISE Analysis Options
Parameter
Description
BPNMATCHTOL=val
Determines the minimum required match between the
NLP and PAC phase noise algorithms. An acceptable
range is 0.05dB to 5dB. The default is 0.5dB.
PHASENOISEKRYLOVDIM
Specifies the dimension of the Krylov subspace that
the Krylov solver uses. This must be an integer greater
than zero. The default is 500.
PHASENOISEKRYLOVITER Specifies the maximum number of Krylov iterations
that the phase noise Krylov solver takes. Analysis
stops when the number of iterations reaches this
value. The default is 1000.
PHASENOISETOL
HSPICE User Guide: RF Analysis
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Specifies the error tolerance for the phase noise
solver. This must be a real number greater than zero.
The default is 1e-8.
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Phase Noise Analysis (.PHASENOISE)
Table 21
PHASENOISE Analysis Options (Continued)
Parameter
Description
PHNOISELORENTZ=val
Turns on a Lorentzian model for the phase noise
analysis.
■
■
■
■
PHNOISEAMPM=1
val=0: uses a linear approximation to a Lorentzian
model
val=1 (default): applies a lorentzian model to all
noise sources
val=2: applies a Lorentzian model to all nonfrequency dependent noise sources
val=3: Lorentzian model applied to white noise
source, Gaussian model applied to flicker noise
sources.
Turns on amplitude modulation/phase modulation
separation. See Amplitude Modulation/Phase
Modulation Separation for details.
Measuring Phase Noise with .MEASURE PHASENOISE
The HSPICE RF optimization flow can read the measured data from a
.MEASURE PHASENOISE analysis. This flow can be combined in the HSPICE
RF optimization routine with a .MEASURE HBTR analysis (see
Using .MEASURE with .HB Analyses on page 192) and a .MEASURE HBNOISE
analysis (see Measuring HBNOISE Analyses with .MEASURE on page 271).
The .MEASURE PHASENOISE syntax supports the following measurements:
■
FIND
.MEASURE PHASENOISE result FIND phnoise at = IFB_value
— yields the result of a variable value at a specific input frequency band
(IFB) point. For example:
.MEASURE PHASENOISE np1 find PHNOISE at=100K
■
WHEN
.MEASURE PHASENOISE result WHEN phnoise=value
—yields the input frequency point at a specific phnoise value. For example:
.MEASURE PHASENOISE fcorn1 when PHNOISE=-120
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Phase Noise Analysis (.PHASENOISE)
■
RMS, average, min, max, and peak-to-peak
.MEASURE PHASENOISE result func phnoise
+ [FROM = IFB1] [TO = IFB2]
—yields the average, RMS, minimum, maximum, or peak-to-peak value of
the phase noise from frequency IFB1 to frequency IFB2, where the value
of func can be RMS, AVG, MIN, MAX or PP. If FROM and TO are not
specified, the value will be calculated over the frequency range specified in
the .PHASENOISE command. For example:
.measure PHASENOISE agn1 AVG phnoise from=100k to=10meg
■
Integral evaluation
.MEASURE PHASENOISE result INTEGRAL phnoise
+ [FROM = IFB1] [TO = IFB2]
—integrates the phase noise value from the IFB1 frequency to the IFB2
frequency. For example:
.MEASURE PHASENOISE rns1 INTEGRAL phnoise from=50k to 500k
■
Derivative evaluation
.MEASURE PHASENOISE result DERIVATIVE phnoise AT = IFB1
—finds the derivative of phase noise at the IFB1 frequency point. For
example:
.MEASURE PHASENOISE fdn1 DERIVATIVE phnoise at=10meg
Note:
.MEASURE PHASENOISE cannot contain an expression that uses a
phasenoise variable as an argument. You also cannot use .MEASURE
PHASENOISE for error measurement and expression evaluation of the
.PHASENOISE command.
See also, the .MEASURE PHASENOISE command in the HSPICE Reference
Manual: Commands and Control Options.
Amplitude Modulation/Phase Modulation Separation
Amplitude Modulation (AM) and Phase Modulation (PM) components of the
total noise can be separated by calculating components in-phase (AM
component) and in quadrature (PM component) with the carrier using PAC and
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Phase Noise Analysis (.PHASENOISE)
BPN PHASENOISE analysis. Output and measure syntax is used to separate
AM/PM noise.
This feature is turned on by setting the .OPTION PHNOISEAMPM=1
(see .OPTION PHNOISEAMPM in the HSPICE Reference Manual: Commands
and Control Options. See also, Important Note for AM/PM Users at the end of
this section.
Keywords for AM/PM separations are:
■
Phase Modulation term only: phnoise
■
Amplitude Modulation term only: la
■
Total phase noise: ltotal
■
Voltage noise: onoise
.PROBE PHASENOISE phnoise [la] [ltotal] [onoise]
.PRINT PHASENOISE phnoise [la] [ltotal] [onoise]
A Noise type suffix can be added to each of these noise terms, phnoise, la,
ltotal, onoise, to select specific noise-type components:
Table 22
Summary of Noise_term
Noise type
frequency-dependent bias-dependent
Noise_term_phnoise_stationary
No
No
Noise_term_phnoise_cyclostationary
No
Yes
Noise_term_phnoise_flicker
Yes
No
Noise_term_phnoise_cycloflicker
Yes
Yes
Noise_term_phnoise_fdep
The union of phnoise_Flicker and
phnoise_cycloflicker noise
types
Noise_term_phnoise_findep
The union of phnoise_stationary
and phnoise_cyclostationary
noise types
Where: Noise_term can be phnoise, la, ltotal, onoise.
Example
.PROBE PHASENOISE phnoise_cyclostationary
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Phase Noise Analysis (.PHASENOISE)
This example outputs the phase modulation noise associated only with
Cyclostationary sources (i.e., sources that are bias dependent, but not
frequency dependent).
Noise Element output is of the form Noise_term (element_name), where
Noise_term can be phnoise, la, ltotal, onoise, and element_name is a
valid netlist element name.
Output File Format
■
File *.printpn#: Writes output from the .PRINT statement when using HB to
obtain the steady state solution.
■
File *.pn#: Writes output from the .PROBE statement when using HB to
obtain the steady state solution.
■
File *.printsnpn#: Writes output from the .PRINT statement when using SN
to obtain the steady state solution.
■
File *.snpn#: Writes output from the .PROBE statement when using SN to
obtain the steady state solution.
Noise source contributions are listed sequentially and are controlled by the
.PHASENOISE command line parameters Listfreq, ListCount, Listfloor,
Listsources. A noise list block will be generated for each output parameter
specified in the .PRINT/.PROBE statement e.g., phnoise, la, ltotal,
onoise.
.MEASURE Syntax and File Format
.MEASURE PHASENOISE extends output variables to the set: am[noise]
pm[noise]
Measure File Format
■
File *.mpn#: Writes output from the .MEASURE statement when using HB
to obtain the steady state solution.
■
File *.msnpn#: Writes output from the .MEASURE statement when using
SN to obtain the steady state solution.
Interpreting Phase Noise Analysis Results
A typical phase noise plot consists of a line, which drops off as a function of
frequency, at a slope of -20dbc/decade where white noise dominates, or 30dbc/decade where flicker noise dominates. At very low offset frequencies,
the phase noise rolls off at according to a Lorentzian shape, such that it never
exceeds 0 dbc/Hz even for very low offset frequencies. The 0 dbc/Hz value
represents the power of the carrier oscillation, at 0 offset frequency. At very
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Phase Noise Analysis (.PHASENOISE)
high offset frequencies, the slope can deviate from -20 dbc/decade due to the
existence of a noise floor or a circuit feedback effect.
Numerical methods for phase noise analysis have limitations. The main
limitation in the PAC phase noise algorithm is that it rolls off too quickly at low
offset frequencies. In the low frequency region, NLP should be trusted. The
main limitation of the NLP algorithm is that it does not cover all high frequency
effects, so PAC should be trusted in the high frequency region.
The BPN algorithm attempts to combine the NLP and PAC results to generate a
single result that is valid for all offset frequencies. It may fail if it cannot identify
an overlap region where NLP and PAC results match. If no overlap region can
be found, the user should attempt to increase nharms on the .HBOSC
command, as this will increase the accuracy of both algorithms, especially PAC.
PAC accuracy is more sensitive to nharms than NLP.
If the phase noise results are suspected to be inaccurate, check the following:
1. Is the .HBOSC steady state solution fully converged?
Explanation: The NLP or PAC small-signal noise analysis requires a highly
accurate steady state solution.
2. Is the phase noise analysis fully converged?
Explanation: Phase noise analysis uses a GMRES iterative linear solver.
If this iterative solver reaches its iteration limit before it is fully converged, the
results are not reliable. Check the number of Krylov iterations that the phase
noise analysis required. If it took the maximum number of iterations (as set
by PHASENOISEKRYLOVITR, default=1000), then the results may not be
fully converged and should not be trusted.
The options PHASENOISEKRYLOVDIM, PHASENOISEKRYLOVTOL, and
PHASENOISEKRYLOVITR can be used to control the GMRES solver. You
can increase PHASENOISEKRYLOVDIM to improve convergence rate at
the expense of memory, or increase PHASENOISEKRYLOVITR to allow
more iterations.
Important Note for AM/PM Users
There are discrepancies that may occur between this feature and the traditional
PAC phase noise analysis in HSPICE RF. Total phase noise (i.e., ltotal) is
the sum of two terms, the amplitude modulation (am) and phase modulation
(phnoise). Traditionally, PAC phase noise reports the phnoise (phase
modulation component) and ltotal (total phase noise) terms as identical,
with the assumption that the am term (amplitude modulation component) was
zero.
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Accumulated Jitter Measurement for Closed Loop PLL Analysis
The PAC phase noise am/pm feature described in this section separately
calculates the am and phnoise components. All phase noise measurements
involving either PAC or BPN will be affected. In most cases the differences
between PHNOISEAMPM=1 and =0 will be small unless you expect a significant
AM component. You may see a slight decrease in the new phase noise (phase
modulation) component compared to the old calculation.
For example, the random jitter calculations HSPICE RF uses are accurate only
when they involve the PM component of phase noise. A small error may be
introduced when they are based on ltotal, or AM noise is included.
NLP phase noise (method=0) only calculates the phnoise component and is
not affected by the am/pm option.
BPN phase noise (method=2) is affected in that the far side component is
derived from the PAC phase noise.
Workaround: Make sure that.OPTION PHNOISEAMPM=0 (the default) to
assure that Periodic AC phase noise amplitude modulation (AM) component is
set to zero to maintain backward compatibility and traditional results for phnoise
and jitter, for example.
Accumulated Jitter Measurement for Closed Loop PLL Analysis
Enhancements to HSPICE RF include considerable support for a variety of
jitter measurements. Many of these are important in a PLL flow, where the
HBOSC or SNOSC analyses are used to compute a phase noise response for
an oscillator or VCO, and the resulting random jitter is derived from phase
noise. In the PLL methodology, other HSPICE RF analyses are used to
compute phase noise contributions from the other PLL building block. A closed
loop analysis is then performed, using phase-domain models for both signal
and noise responses, that takes into account the noise contributions from all
such blocks. To complete this flow is the ability to compute “Accumulated Jitter”
or “Timing Jitter” for the closed loop PLL. The accumulated jitter response is
essentially an integral transformation of the closed-loop PLL response. The
following sections show how accumulated jitter also be measured directly from
the phase noise output of an open loop oscillator analysis.
In the PLL flow, the closed loop phase noise must be interpreted from the
results of a linear HSPICE RF .AC/.NOISE analysis. The following sections
describe a capability that allows direct measurement of accumulated jitter from
the results of this closed loop noise analysis, without any special interpretation
of the results.
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Accumulated Jitter Measurement for Closed Loop PLL Analysis
Jitter Measurements from Phase Noise
These topics are discussed in the following sections:
■
Jitter Definitions
■
Jitter Output Syntax
■
.MEASURE Statements for Jitter
■
Peak-to-Peak Jitter
Jitter Definitions
HSPICE RF provides several random jitter (RJ) measurements. This section
defines, describes, and compares the various jitter measurements provided.
Random jitter measurements are derived from the results of an HSPICE RF
phase noise analysis. The relationships between phase noise and the random
jitter measurements are presented here, and their means for calculation. The
types of random jitter measurements include: Timing, Phase, Period, Tracking,
Long-Term, and Cycle-to-Cycle Jitter.
Timing jitter is a measurement of oscillator uncertainty in the time domain. For
clock applications, time domain measurements are preferable, since most
specifications of concern involve time domain values.
Timing jitter is the standard deviation of the timing uncertainty, which is a
function of the auto-correlation function in the power spectrum of the phase
variations. Timing Jitter is the square root of the variance (standard deviation
squared) of the timing uncertainty between two clock edges separated by an
interval given by τ = N ⋅ T o , where T o is the ideal clock period. It can be
written as a function of the auto-correlation function of the power spectrum of
phase variations as:
Equation 38
2
2
σTIE ( τ ) = ------[ R φ( 0 ) – R φ( τ ) ]
2
ωo
where TIE refers to the Time Interval Error. This measurement is known as
Timing Jitter, Accumulated Jitter, or N-Cycle Jitter, since it represents the jitter
that may accumulate over an interval of many periods.
The Weiner-Khintchine Theorem [1] relates the auto correlation function to the
power spectrum of phase variations as in the following equation:
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∞
Equation 39
∞
1
jωτ
R φ( τ ) = ------ ∫ S φ( ω)e dω= 2 ∫ L ( f ) cos ( 2πfτ ) df
2π
–∞
0
where S φ( ω) is the double-sided power spectrum of phase variations, and L ( f )
is the single-sideband phase noise. The auto-correlation for τ = 0 is given by
∞
Equation 40
R φ( 0 ) ≡
2
φrms
= 2 ∫ L ( f ) df
0
which defines φrms in HSPICE RF known as RMS Phase Jitter.
2
Using the identity 2 sin α = 1 – cos 2α we can then write:
∞
Equation 41
2
σTIE
8
2
( τ ) = ------L ( f ) sin ( πfτ ) df
2 ∫
ωo
0
to enable currently supported HSPICE RF jitter measurements to be written as:
∞
Equation 42
φrms = σph ⋅ ω0 =
∞
2 ∫ L ( f ) df
“RMS Phase Jitter”
0
2
2
σTIE ( τ ) = ----- 2 ∫ L ( f )sin ( πfτ )df “Timing (Time Interval Error) Jitter”
ω0
0
From these definitions, several other key jitter measurements can be derived,
including Period Jitter, Tracking Jitter, Long-Term Jitter, and Cycle-to-Cycle
Jitter.
Period Jitter is equivalent to the value for Timing Jitter for a one period interval.
We therefore have:
∞
Equation 43
σPER
2
2
= σTIE ( T 0 ) = ----- 2 ∫ L ( f )sin ( πfT 0 )df
ω0
"Period Jitter"
0
Tracking Jitter is equivalent to the value (in units of seconds) for RMS Phase
Jitter, or:
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∞
Equation 44
σtr = σph
φrms
1
= --------= ----- 2 ∫ L ( f )df “Tracking Jitter”
ω0
ω0
0
Long-Term Jitter is equivalent to
Equation 45
2 times the Tracking Jitter, i.e.:
σΔT →∞ = σTIE ( τ →∞) =
φrms
2
2 --------- = ----ω0
ω0
∞
∫
L ( f ) df
“Long-Term Jitter”
0
Cycle-to-Cycle Jitter is based on the difference between adjacent Period Jitter
measurements. It is given by:
Equation 46
σCTC =
2
2
4σPER – σTIE ( 2T 0 )
“Cycle-to-Cycle Jitter”
In general, each of the above calculations must be performed carefully over
limits of integration to accurately calculate jitter expressions based on the finite
frequency limits provided for the phase noise analysis. Linear interpolation is
used, but the phase noise generally follows more of a power law expansion.
Jitter Output Syntax
The timing jitter calculations are derived from the results of phase noise
analysis. The phase noise output syntax supports the JITTER keyword as an
output keyword in addition to the PHNOISE keyword.
.PRINT PHASENOISE PHNOISE JITTER
.PROBE PHASENOISE PHNOISE JITTER
If the JITTER keyword is present, the .PHASENOISE statement also outputs
the raw jitter data to *.jt0 and *.printjt0 data files. The PHNOISE data is given in
units of dBc/Hz, i.e., dB relative to the carrier, per Hz, across the output nodes
specified by the PHASENOISE statement. The data plot is a function of offset
frequency. If the JITTER keyword is present, .PHASENOISE outputs the TIE
Timing Jitter (Accumulated Jitter) data to *.jt0 and *.printjt0 data files. These
data are plotted as a function of time in units of seconds. The jitter calculations
make use of some of the parameters given in the .PHASENOISE syntax (see
PHASENOISE Input Syntax for the syntax and examples.).
The time samples for timing jitter output make use of the same number of
points as the phase noise frequency sweep specification.
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The output of timing jitter information uses a corresponding time sampling
derived via:
Equation 47
1
2
N
τ 1 = ------, τ 2 = ------, ...,τ N = -----T0
T0
T0
.MEASURE Statements for Jitter
The jitter-specific .MEASURE statements specify the jitter keywords as follows.
(For discussion of the BER parameter, see below.)
.MEASURE PHASENOISE Jname PERJITTER phnoise
+ [UNITS=(sec|rad|UI)] [BER=val]
.MEASURE PHASENOISE Jname CTCJITTER phnoise
+ [UNITS=(sec|rad|UI)] [BER=val]
.MEASURE PHASENOISE Jname RMSJITTER phnoise
+ [FROM start_frequency] [TO end_frequency]
+ [UNITS=(sec|rad|UI)] [BER=val]
.MEASURE PHASENOISE Jname PHJITTER phnoise
+ [FROM start_frequency] [TO end_frequency]
+ [UNITS=(sec|rad|UI)] [BER=val]
.MEASURE PHASENOISE Jname TRJITTER phnoise
+ [FROM start_frequency] [TO end_frequency]
+ [UNITS=(sec|rad|UI)] [BER=val]
.MEASURE PHASENOISE Jname LTJITTER phnoise
+ [FROM start_frequency] [TO end_frequency]
+ [UNITS=(sec|rad|UI)] [BER=val]
RMSJITTER, PHJITTER, and TRJITTER are synonymous measurements, all
based on the calculations described related to the RMS Phase Jitter value in
units of seconds given by σph = φrms ⁄ ω0 . These measurements allow control
of the integration range using the FROM and TO parameters. The
measurements for PERJITTER, and CTCJITTER use the full offset frequency
sweep range given for the phase noise analysis to compute values (the FROM
and TO parameters are ignored if entered).
As given currently in HSPICE RF, the frequency intervals can be modified for
these jitter calculations (if desired, although not recommended), and UNITS
can be selected between seconds, radians, and Unit Intervals. The following
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table specifies the calculation used for units=seconds for each jitter
measurement.
MEASURE name
Calculation used (Units=sec)
RMSJITTER
σph = φrms ⁄ ω0
PHJITTER
σph = φrms ⁄ ω0
TRJITTER
σph = φrms ⁄ ω0
PERJITTER
σPER
LTJITTER
σΔT →∞ =
CTCJITTER
σCTC
2φrms ⁄ ωo
Example
.meas phasenoise rj RMSJITTER phnoise from 1K to 100K
+ units = rad
Peak-to-Peak Jitter
As noted in .MEASURE Statements for Jitter, an additional BER (Bit Error
Rate) parameter is supported. This parameter allows you to convert any jitter
value from an RMS value into a Peak-to-Peak value. The RMS jitter values
correspond to a 1-sigma standard deviation value for the Gaussian distribution
of the jitter. Peak-to-peak values represent the full span of the Gaussian
distribution. Since this span is theoretically unbounded for truly random
distributions, the conversion to peak-to-peak values has to be interpreted as
spanning some number of sigma values. You can arrive at this number (i.e.,
“sigma multiplier”) by specifying a corresponding Bit Error Rate.
The term “BER” corresponds to the unitless Bit Error Rate that allows for this
conversion. The following table shows some example conversions from various
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BER values into a “sigma multiplier” value which corresponds to the number of
sigma standard deviations in converting from RMS to peak-to-peak values:
Bit Error Rate
Sigma Multiplier
10-3
6.180
10-4
7.438
10-5
8.530
10-6
9.507
10-7
10.399
10-8
11.224
10-9
11.996
10-10
12.723
10-11
13.412
10-12
14.069
10-13
14.698
10-14
15.301
10-15
15.883
10-16
16.444
These conversions are done in accordance with the relationship:
Equation 48
1--α
erfc ----------------- = BER
2
2⋅ 2
where, erfc is the complementary error function, and α is the Sigma Multiplier.
Support for peak-to-peak conversions is included for a continuous range of
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Small-Signal Phase-Domain Noise Analysis (.ACPHASENOISE)
BER values from 10
range).
– 16
–3
≤ α ≤ 10 (and some values extrapolated outside this
Specification of the BER parameter results in the output of the Peak to Peak
jitter value, and not the RMS value. Labels for the measurements show
appropriate “rms” and “p-p” labels. A BER parameter set to BER=0 is
equivalent to having no parameter, and only results in the RMS calculation.
Errors/Warnings
Error handling and recovery is exercised to capture obvious errors in input
specifications. The following error checking is performed:
■
Calculations are be performed if oscillator or phase noise analysis fails.
■
ERROR if L(f) > 1 over any part of the frequency sweep (non-dB form).
■
ERROR if L(f) < 0 over any part of the frequency sweep (non-dB form).
■
Error if any time or frequency samples are negative values.
■
ERROR if BER < 0 for any Jitter measurement.
■
WARNING if BER > 1 for any Jitter measurement.
■
WARNING if f0 < 10 Hz. Message: “Jitter calculations may be ineffective for
offset frequencies under 10 Hz.”
Small-Signal Phase-Domain Noise Analysis (.ACPHASENOISE)
To see the influence that oscillator or VCO phase noise can have on a system
where it is present, it is necessary to perform a phase-domain analysis where
the circuit variables are phase, and the input noise stimuli are phase noise. This
is the purpose of the .ACPHASENOISE analysis in HSPICE RF.
This type of analysis is critical, for example, in analyzing the effects of noise in
a phase-locked loop (PLL). In a PLL design flow, the HBOSC or SNOSC
analyses are used to compute a phase noise response for an oscillator or VCO.
HSPICE RF analyses can be used to compute phase noise contributions from
the other PLL building blocks. A closed loop PLL analysis can then be
performed by using phase-domain models for both signal and noise responses,
where the noise contributions from all blocks are input as phase noise stimuli.
Such an analysis can be performed to determine the PLL closed-loop phase
noise, based on the contributions of each block, determined by an open loop
analysis.
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Small-Signal Phase-Domain Noise Analysis (.ACPHASENOISE)
AC Phase Noise Analysis Syntax
A small-signal phase-domain analysis, such as that useful for PLL noise
analyses, can use a methodology based on small-signal .AC and .NOISE
analyses, where the circuit analyzed is a phase-domain modeled circuit (not
voltage/current domain), and the results are therefore interpreted accordingly
as phase variables. The variables of the associated noise analysis must
therefore interpreted as phase noise variables. A normal .AC analysis
command can be used to activate the small-signal analysis and specify its
LIN/OCT/DEC/POI or SWEEPBLOCK frequency sweep. However, to properly
interpret signal and noise quantities as phase variables, the following command
properly formats output in terms of phase noise variables. The syntax is:
.ACPHASENOISE output input [interval] carrier=freq
+ [listfreq=(frequencies|none|all)]
+ [listcount=val] [listfloor=val]
+ [listsources=(1|0)]
where:
■
output
Node voltage or branch current output variable
■
input
Independent source used as input reference
■
interval Number of intervals for which to dump jitter and noise summary
information
■
freq
Frequency (in Hz) of the fundamental carrier upon which the
noise transformations are based
■
list…
List parameters usage is consistent with other HSPICE noise
analyses
ACPHASENOISE Analysis .PRINT/.PROBE Syntax
The unique aspect of the .ACPHASENOISE analysis is that it allows the small
signal noise calculation results to be interpreted as phase noise values. The
available .print/.probe measurements reflect this. The .print/.probe
output syntax are the “JITTER” and “PHNOISE” keywords consistent with the
HSPICE RF .phasenoise analysis, namely:
.PRINT ACPHASENOISE PHNOISE JITTER
.PROBE ACPHASENOISE PHNOISE JITTER
As with the .PHASENOISE analysis, the .ACPHASENOISE analysis outputs raw
data to *.pn0 and *.printpn0 files. The PHNOISE data is given in units of dBc/Hz,
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Small-Signal Phase-Domain Noise Analysis (.ACPHASENOISE)
i.e., dB relative to the carrier, per Hz, across the output nodes specified by the
.ACPHASENOISE statement. The data plot is a function of offset frequency. If
the “JITTER” keyword is present, .ACPHASENOISE also outputs the
accumulated TIE jitter data to *.jt0 and *.printjt0 data files. These data are
plotted as a function of time in units of seconds. The Timing Jitter data itself has
units of seconds. The timing jitter calculations make use of the parameters
given in the .ACPHASENOISE syntax, such as “freq” and “interval”.
Se also Using Noise Analysis Results as Input Noise Sources.
.MEASURE Support for ACPHASENOISE
Single valued jitter measurements are available from .MEASURE statements.
Examples include period jitter, cycle-to-cycle jitter, and phase jitter
measurements, respectively, as shown below:
.MEASURE ACPHASENOISE Jname PERJITTER phnoise
+ [UNITS=(sec|rad|UI)] [BER=val]
.MEASURE ACPHASENOISE Jname CTCJITTER phnoise
+ [UNITS=(sec|rad|UI)] [BER=val
.MEASURE ACPHASENOISE Jname PHJITTER phnoise
+ [FROM start_frequency [TO end_frequency]
+ [UNITS=(sec|rad|UI)] [BER=val]
Errors and Warnings
Error checking of values are consistent with that used for the .PHASENOISE
analysis.
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Chapter 9: Oscillator and Phase Noise Analysis
References
References
[1] E. Ngoya, A. Suarez, R. Sommet, R. Quere, “Steady State Analysis of Free
or Forced Oscillators by Harmonic Balance and Stability Investigation of
Periodic and Quasi-Periodic Regimes,” International Journal of Microwave
and Millimeter-Wave Computer-Aided Engineering, Volume 5, Number 3,
pages 210-223 (1995)
[2] C.R. Chang, M.B. Steer, S. Martin, E. Reese, “Computer-Aided Analysis of
Free-Running Microwave Oscillators,” IEEE Trans. on Microwave Theory
and Techniques, Volume 39, No. 10, pages 1735-1745, October 1991.
[3] G.D. Vendelin, Design of Amplifiers and Oscillators by the S-Parameter
Method, John Wiley & Sons, 1982
[4] A. Demir, A. Mehrotra, J. Roychowdhury, “Phase Noise in Oscillators: A
Unifying Theory and Numerical Methods for Characterization” in Proc. IEEE
DAC, pages 26-31, June 1998.
[5] A. Demir, A. Mehrotra, and J. Roychowdhury, “Phase Noise in Oscillators: A
Unifying Theory and Numerical Methods for Characterization,” IEEE Trans.
Circuits System I, Volume 47, pages 655–674, May 2000.
[6] A. van der Ziel, Noise in Solid State Devices and Circuits, John Wiley &
Sons, 1986.
[7] A. Hajimiri, S. Limotyrakis, and T.H. Lee, “Jitter and phase noise in ring
oscillators,” IEEE J. Solid-State Circuits, vol. 34, no. 6, pp. 790-804, June
1999.
[8] Jitter Analysis Techniques for High Data Rates, Application Note 1432,
Agilent Technologies, Feb. 2003.
[9] Characterization of Clocks and Oscillators, NIST Technical Note 1337,
National Institute of Standards and Technology.
[10] G.V. Klimovitch, “Near-carrier oscillator spectrum due to flicker and white
noise,” Proc. ISCAS 2000 (Geneva), May 2000.
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References
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10
Large Signal Periodic AC, Transfer Function, and Noise
Analyses
10
Describes how to use both harmonic balance-based and Shooting Newtonbased AC, and transfer function analyses, as well as nonlinear, steady-state
noise analysis.
The following topics are presented in this section:
■
Multitone Harmonic Balance AC Analysis (.HBAC)
■
Shooting Newton AC Analysis (.SNAC)
■
Multitone Harmonic Balance Noise (.HBNOISE)
■
Shooting Newton Noise Analysis (.SNNOISE)
■
Periodic Time-Dependent Noise Analysis (.PTDNOISE)
■
Multitone Harmonic Balance Transfer Function Analysis (.HBXF)
■
Shooting Newton Transfer Function Analysis (.SNXF)
Multitone Harmonic Balance AC Analysis (.HBAC)
You use the .HBAC (Harmonic Balance AC) statement for analyzing linear
behavior in large-signal periodic systems. The .HBAC statement uses a
periodic AC (PAC) algorithm to perform linear analysis of autonomous
(oscillator) or nonautonomous (driven) circuits, where the linear coefficients are
modulated by a periodic, steady-state signal.
Multitone HBAC analysis extends single-tone HBAC to quasi-periodic systems
with more than one periodic, steady-state tone. One application of multitone
HBAC is to more efficiently determine mixer conversion gain under the
influence of a strong interfering signal than is possible by running a swept
three-tone HB simulation.
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Multitone Harmonic Balance AC Analysis (.HBAC)
Prerequisites and Limitations
The following prerequisites and limitations apply to HBAC:
■
Requires one and only one .HBAC statement. If you use multiple .HBAC
statements, HSPICE RF uses only the last .HBAC statement.
■
Requires one and only one .HB statement.
■
Supports arbitrary number of tones.
■
Requires placing the parameter sweep in the .HB statement.
■
Requires at least one HB source.
■
Requires at least one HBAC source.
■
Supports unlimited number of HB and HBAC sources.
■
The requested maximum harmonic in a .PROBE or .PRINT statement must
be less than or equal to half the number of harmonics specified in harmonic
balance (that is, max_harm <= num_hb_harms / 2).
Input Syntax
.HBAC <frequency_sweep>
Parameter
Description
frequency_sweep
Frequency sweep range for the input signal (also referred to
as the input frequency band (IFB) or fin). You can specify LIN,
DEC, OCT, POI, or SWEEPBLOCK. Specify the nsteps, start,
and stop frequencies using the following syntax for each type
of sweep:
■
■
■
■
■
■
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA=dataname
HBAC Analysis Options
The following options directly relate to a HBAC analysis and override the
corresponding PAC options if specified in the netlist:
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■
.OPTION HBACTOL, default = 1x10-8, Range = 1x10-14 to Infinity
■
.OPTION HBACKRYLOVDIM, default = 300, Range = 1 to Infinity
■
.OPTION HBACKRYLOVITR, default = 1000, Range = 1 to Infinity
If these parameters are not specified, then the following conditions apply:
■
If HBACTOL > HBTOL, then HBACTOL = HBTOL
■
If HBACKRYLOVDIM < HBKRYLOVDIM, then HBACKRYLOVDIM =
HBKRYLOVDIM
Output Syntax
This section describes the syntax for the HBAC .PRINT and .PROBE
statements. These statements are similar to those used for HB analysis.
.PRINT and .PROBE Statements
.PRINT HB TYPE(NODES | ELEM)[INDICES]
.PROBE HB TYPE(NODES | ELEM)[INDICES]
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Multitone Harmonic Balance AC Analysis (.HBAC)
Parameter
Description
TYPE
Specifies a harmonic type node or element.
TYPE can be one of the following:
■
■
■
■
NODES |
ELEM
NODES or ELEM can be one of the following:
■
■
■
■
258
Voltage type –
V = voltage magnitude and phase in degrees
VR = real component
VI = imaginary component
VM = magnitude
VP - Phase in degrees
VPD - Phase in degrees
VPR - Phase in radians
VDB - dB units
VDBM - dB relative to 1 mV
Current type –
I = current magnitude and phase in degrees
IR = real component
II = imaginary component
IM = magnitude
IP - Phase in degrees
IPD - Phase in degrees
IPR - Phase in radians
IDB - dB units
IDBM - dB relative to 1 mV
Power type – P
Frequency type – hertz[index], hertz[index1, index2, ...] You must specify the
harmonic index for the hertz variable. The frequency of the specified harmonics
is dumped.
Voltage type – a single node name (n1), or a pair of node names, (n1,n2)
Current type – an element name (elemname)
Power type – a resistor (resistorname) or port (portname) element name
Frequency type – the harmonic index for the hertz variable. The frequency of
the specified harmonics is dumped.
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Multitone Harmonic Balance AC Analysis (.HBAC)
Parameter
Description
INDICES
Index to tones in the form [n1, n2, ..., nK, +/-1].
■
nj is the index of the j-th HB tone and the .HB statement contains K tones
+/-1 is the index of the HBAC tone
Wildcards are not supported if this parameter is used.
■
You can transform HB data into the time domain and output by using the following
syntax:.PRINT HBTRAN ov1 [ov2 ... ovN].PROBE HBTRAN ov1 [ov2 ... ovN]. See
TYPE above for voltage and current type definitions.
HBAC Output Data Files
An HBAC analysis produces these output data files:
■
Output from the .PRINT statement is written to a .printhb# file. This data is
against the IFB points.
•
The header contains the large-signal fundamental and the range of
small-signal frequencies.
•
The columns of data are labeled as F(Hz), followed by the output
variable names. Each variable name has the associated mixing pair
value appended.
All N variable names and all M mixing pair values are printed for each
swept small-signal frequency value (a total of N*M for each frequency
value).
■
Output from the .PROBE statement is written to a .hb# file. This data is
against the IFB points.
■
Reported performance log statistics are written to a .lis file:
•
Number of nodes
•
Number of FFT points
•
Number of equations
•
Memory in use
•
CPU time
•
Maximum Krylov iterations
•
Maximum Krylov dimension
•
Target GMRES residual
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Multitone Harmonic Balance AC Analysis (.HBAC)
•
GMRES residual
•
Actual Krylov iterations taken
•
Frequency (swept input frequency values).
Errors and Warnings
The following error and warning messages are used when HSPICE encounters
a problem with a HBAC analysis.
Error Messages
HBAC frequency sweep includes negative frequencies. HBAC allows only
frequencies that are greater than or equal to zero.
No HB statement is specified (error at parser). HBAC requires an HB statement
to generate the steady-state solution.
Warning Messages
More than one HBAC statement (warning at parser). HSPICE RF uses only the
last HBAC statement in the netlist.
No HBAC sources are specified (error at parser). HBAC requires at least one
HBAC source.
GMRES Convergence Failure. When GMRES (Generalized Minimum
Residual) reaches the maximum number of iterations and the residual is
greater than the specified tolerance. The HBAC analysis generates a warning
and then continue as if the data were valid. This warning reports the following
information:
■
Final GMRES Residual
■
Target GMRES Residual
■
Maximum Krylov Iterations
■
Actual Krylov Iterations taken
HBAC Example
The following example is shipped with the HSPICE RF distribution as
mix_hbac.sp and is available in directory:
$<installdir>/demo/hspicerf/examples.
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Shooting Newton AC Analysis (.SNAC)
* Test HBAC: ideal I,Q mixer
.OPTIONS PROBE
.OPTIONS POST=2
vlo lo 0 1.0 cos(1.0 0.5 1g) TRANFORHB=1 $ Periodic, Large-Signal
Input
rlo lo 0 1.0
rrf rf 0 1.0 $ Noise source
rrf1 rf1 rf 1.0 $ Noise source
g1 0 if cur='1.0*v(lo)*v(rf)' $ mixer element
c1 0 if q='1.0e-9*v(lo)*v(rf)' $ mixer element
rout if ifg 1.0
vctrl ifg 0 0.0
h1 out 0 vctrl 1.0
rh1 out 0 1.0
vrf rf1 0 hbac .001 0 $ Small signal source
.hb tones=1.0g nharms=3
.hbac lin 1 0.8g 0.8g
.print hb v(rf1) v(lo) v(out)
.probe hb v(rf1) v(lo) v(out)
.measure hb vout1 find v(out)[1,-1] at=0.8g
.end
Shooting Newton AC Analysis (.SNAC)
You use the Shooting Newton AC (.SNAC) statement for analyzing linear
behavior in large-signal periodic systems. The .SNAC statement uses a
periodic AC (PAC) and Shooting Newton algorithm to perform linear analysis of
nonautonomous (driven) circuits, where the linear coefficients are modulated
by a periodic, steady-state signal.
The following section describes the periodic AC analysis based on a Shooting
Newton algorithm. This functionality is similar to the Harmonic Balance (HBAC)
for periodic AC analysis.
Prerequisites and Limitations
The following prerequisites and limitations apply to SNAC:
■
Requires one and only one .SNAC statement. If you use multiple .SNAC
statements, HSPICE RF uses only the last .SNAC statement.
■
Requires one and only one .SN statement.
■
Requires placing the parameter sweep in the .SN statement.
■
Requires at least one Periodic source.
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Shooting Newton AC Analysis (.SNAC)
■
Limited to simulations that can be reduced to a single tone SN analysis.
■
Supports unlimited number of sources.
■
The requested maximum harmonic in a .PROBE or .PRINT statement must
be less than or equal to half the number of harmonics specified in the SN
statement (that is, max_harm ≤ nharms / 2).
Input Syntax
.SNAC <frequency_sweep>
Parameter
Description
frequency_sweep
Frequency sweep range for the input signal (also referred to as the input
frequency band (IFB) or fin). You can specify LIN, DEC, OCT, POI, or
SWEEPBLOCK. Specify the nsteps, start, and stop frequencies using the
following syntax for each type of sweep:
■
■
■
■
■
■
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA=dataname
Output Syntax
This section describes the syntax for the SNAC .PRINT and .PROBE
statements. These statements are similar to those used for HB analysis.
.PRINT and .PROBE Statements
.PRINT SN TYPE(NODES | ELEM)[INDICES]
.PROBE SN TYPE(NODES | ELEM)[INDICES]
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Parameter
Description
TYPE
Specifies a harmonic type node or element.
TYPE can be one of the following:
■
■
■
■
NODES |
ELEM
Voltage type –
V = voltage magnitude and phase in degrees
VR = real component
VI = imaginary component
VM = magnitude
VP - Phase in degrees
VPD - Phase in degrees
VPR - Phase in radians
VDB - dB units
VDBM - dB relative to 1 mV
Current type –
I = current magnitude and phase in degrees
IR = real component
II = imaginary component
IM = magnitude
IP - Phase in degrees
IPD - Phase in degrees
IPR - Phase in radians
IDB - dB units
IDBM - dB relative to 1 mV
Power type – P
Frequency type – hertz[index], hertz[index1, index2, ...] You must specify the
harmonic index for the hertz variable. The frequency of the specified
harmonics is dumped.
NODES or ELEM can be one of the following:
■
■
■
■
Voltage type – a single node name (n1), or a pair of node names, (n1,n2)
Current type – an element name (elemname)
Power type – a resistor (resistorname) or port (portname) element name
Frequency type – the harmonic index for the hertz variable. The frequency of
the specified harmonics is dumped.
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Parameter
Description
INDICES
Index to tones in the form [n1, +/-1].
■
n1 is the index of the SN tone
+/-1 is the index of the SNAC tone
Wildcards are not supported if this parameter is used.
■
You can transform SN data into the time domain and output by using the following
syntax:.PRINT SNTRAN ov1 [ov2 ... ovN]. PROBE SNTRAN ov1 [ov2 ... ovN].
See TYPE above for voltage and current type definitions.
SNAC Output Data Files
A SNAC analysis produces these output data files:
■
Output from the .PRINT statement is written to a .printsnac# file.
■
This data is against the IFB points.
■
The header contains the large-signal fundamental and the range of smallsignal frequencies.
■
The columns of data are labeled as F(Hz), followed by the output variable
names. Each variable name has the associated mixing pair value
appended. All N variable names and all M mixing pair values are printed for
each swept small-signal frequency value (a total of N*M for each frequency
value).
■
Output from the .PROBE statement is written to a .snac# file.
Reported performance log statistics are written to a .lis file:
264
■
Number of nodes
■
Number of FFT points
■
Number of equations
■
Memory in use
■
CPU time
■
Maximum Krylov iterations
■
Maximum Krylov dimension
■
Target GMRES residual
■
GMRES residual
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■
Actual Krylov iterations taken
■
Frequency (swept input frequency values)
Errors and Warnings
The following error and warning messages are used when HSPICE encounters
a problem with a SNAC analysis.
Error Messages
SNAC frequency sweep includes negative frequencies. SNAC allows only
frequencies that are greater than or equal to zero.
No SN statement is specified (error at parser). SNAC requires an SN statement
to generate the steady-state solution.
Warning Messages
More than one SNAC statement (warning at parser). HSPICE RF uses only the
last SNAC statement in the netlist.
No SNAC sources are specified (error at parser). SNAC requires at least one
SNAC source.
GMRES Convergence Failure. When GMRES (Generalized Minimum
Residual) reaches the maximum number of iterations and the residual is
greater than the specified tolerance. The SNAC analysis generates a warning
and then continue as if the data were valid. This warning reports the following
information:
■
Final GMRES Residual
■
Target GMRES Residual
■
Maximum Krylov Iterations
■
Actual Krylov Iterations taken
SNAC Example
The following example is shipped with the HSPICE RF distribution as
mix_snac.sp and is available in directory:
$<installdir>/demo/hspicerf/examples.
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Multitone Harmonic Balance Noise (.HBNOISE)
* Test SNAC: ideal I,Q mixer -rrd
.OPTIONS PROBE
.OPTIONS POST=2
$.OPTIONS snmaxiter=100
.OPTIONS SNACCURACY=5
vlo lo 0 1.0 cos(1.0 0.5 1g) $ Periodic, Large-Signal SN Input
rlo lo 0 1.0
rrf rf 0 1.0 $ Noise source
rrf1 rf1 rf 1.0 $ Noise source
g1 0 if cur='1.0*v(lo)*v(rf)' $ mixer element
c1 0 if q='1.0e-9*v(lo)*v(rf)' $ mixer element
rout if ifg 1.0
vctrl ifg 0 0.0
h1 out 0 vctrl 1.0
rh1 out 0 1.0
vrf rf1 0 snac .001 24.0 $ Small signal for SNAC with 1-tone
SN Input
.sn tones=1.0g nharms=3
.snac lin 1 0.8g 0.8g
.print sn v(rf1) v(lo) v(out)
.print snfd v(rf1) v(lo) v(out)
.print snac v(rf1) v(lo) v(out)
.measure snac vout1 find v(out)[1,-1] at=0.8g
.measure snac vout2 find v(out)[0,1] at=0.8g
.measure snac vout3 find v(out)[1,1] at=0.8g
.measure sn vlo1 find v(lo) at=0.5n
.measure sn vlo2 find v(lo) at=1n
.measure snfd vlo3 find v(lo)[1] at=1
.end
Multitone Harmonic Balance Noise (.HBNOISE)
An HBNOISE (Harmonic Balance noise) analysis simulates the noise behavior
in periodic systems. It uses a Periodic AC (PAC) algorithm to perform noise
analysis of nonautonomous (driven) circuits under periodic, steady-state tone
conditions. This can be extended to quasi-periodic systems having more than
one periodic, steady-state tone. One application for a multitone HBNOISE
analysis is determining mixer noise figures under the influence of a strong
interfering signal.
The PAC method simulates noise assuming that the stationary noise sources
and/or the transfer function from the noise source to a specific output are
periodically modulated.
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Multitone Harmonic Balance Noise (.HBNOISE)
■
The modulated noise source (thermal, shot, or flicker) is modeled as a
cyclostationary noise source.
■
A PAC algorithm solves the modulated transfer function.
■
You can also use the HBNOISE PAC method with correlated noise sources,
including the MOSFET level 9 and level 11 models, and the behavioral noise
source in the G-element (Voltage Dependent Current Source).
You use the .HBNOISE statement to perform a Periodic Noise Analysis.
Supported Features
HBNOISE supports the following features:
■
All existing HSPICE RF noise models.
■
Uses more than one single-tone, harmonic balance to generate the steadystate solution.
■
Unlimited number of HB sources (using the same tone, possibly multiple
harmonics).
■
Includes stationary, cyclostationary, frequency-dependent, and correlated
noise effects.
■
Swept parameter analysis.
■
Results are independent of the number of HBAC sources in the netlist.
Prerequisites and Limitations
The following prerequisites and limitations apply to HBNOISE:
■
Requires one .HB statement (which determines the steady-state solution).
■
Requires at least one HB source or one TRANFORHB source.
■
Requires placing the parameter sweep in the .HB statement.
■
The requested maximum harmonic in .HBNOISE must be less than or equal
to half the number of harmonics used in harmonic balance (that is,
max_harm <= num_hb_harms/2).
Input Syntax
.HBNOISE [output] [insrc] [parameter_sweep]
+ [n1, n2, ..., nk,+/-1]
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+ [listfreq=(frequencies|none|all)] [listcount=val]
+ [listfloor=val] [listsources=on|off]
Parameter
Description
output
Output node, pair of nodes, or 2-terminal element. HSPICE RF
references equivalent noise output to this node (or pair of nodes).
Specify a pair of nodes as V(n+,n-). If you specify only one node,
V(n+), then HSPICE RF assumes that the second node is ground.
You can also specify a 2-terminal element name that refers to an
existing element in the netlist.
insrc
An input source. If this is a resistor, HSPICE RF uses it as a
reference noise source to determine the noise figure. If the
resistance value is 0, the result is an infinite noise figure.
parameter_sweep Frequency sweep range for the input signal. Also referred to as the
input frequency band (IFB) or fin). You can specify LIN, DEC, OCT,
POI, SWEEPBLOCK, DATA, MONTE, or OPTIMIZE sweeps.
Specify the nsteps, start, and stop frequencies using the following
syntax for each type of sweep:
■
■
■
■
■
n1,n2,...,nk,
+/-1
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
Index term defining the output frequency band (OFB or fout) at
which the noise is evaluated. Generally,
fout=ABS(n1*f+n2*f2+...+nk*fk+/-fin)
Where:
■
f1,f2,...,fk are the first through k-th steady-state tones
determined from the harmonic balance solution
■
fin is the IFB defined by parameter_sweep.
The default index term is [1,1,...1,-1]. For a single tone analysis, the
default mode is consistent with simulating a low-side, down
conversion mixer where the RF signal is specified by the IFB and
the noise is measured at a down-converted frequency that the OFB
specifies. In general, you can use the [n1,n2,...,nk,+/-1] index term
to specify an arbitrary offset. The noise figure measurement is also
dependent on this index term.
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Parameter
Description
listfreq
Prints the element noise value to the .lis file. You can specify at
which frequencies the element noise value is printed. The
frequencies must match the sweep_frequency values defined in
the parameter_sweep, otherwise they are ignored.
In the element noise output, the elements that contribute the
largest noise are printed first. The frequency values can be
specified with the NONE or ALL keyword, which either prints no
frequencies or every frequency defined in parameter_sweep.
Frequency values must be enclosed in parentheses. For
example:listfreq=(none)
listfreq=(all)
listfreq=(1.0G)
listfreq=(1.0G, 2.0G)The default value is NONE.
listcount
Prints the element noise value to the .lis file, which is sorted from
the largest to smallest value. You do not need to print every noise
element; instead, you can define listcount to print the number
of element noise frequencies. For example, listcount=5 means
that only the top 5 noise contributors are printed. The default value
is 1.
listfloor
Prints the element noise value to the .lis file and defines a
minimum meaningful noise value (in V/Hz1/2 units). Only those
elements with noise values larger than listfloor are printed.
The default value is 1.0e-14 V/Hz1/2.
listsources
Prints the element noise value to the .lis file when the element has
multiple noise sources, such as a FET, which contains the thermal,
shot, and 1/f noise sources. You can specify either ON or OFF: ON
Prints the contribution from each noise source and OFF does not.
The default value is OFF.
Output Syntax
The HSPICE RF HB and SN noise analyses can output the output noise
(onoise), noise figures (NF, SSNF and DSNF) and, the input referred noise
(inoise). This section describes the syntax for the HBNOISE .PRINT and
.PROBE statements.
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.PRINT and .PROBE Statements
.PRINT HBNOISE <ONOISE> <NF> <SSNF> <DSNF> <INOISE>
.PROBE HBNOISE <ONOISE> <NF> <SSNF> <DSNF> <INOISE>
Parameter
Description
ONOISE
Outputs the voltage noise at the output frequency band (OFB) across
the output nodes in the .HBNOISE statement. The data is plotted as a
function of the input frequency band (IFB) points. Units are in V/Hz1/2.
Simulation ignores ONOISE when applied to autonomous circuits.
NF
SSNF
NF and SSNF both output a single-side band noise figure as a function
of the IFB points:
NF = SSNF = 10 Log(SSF)
Single side-band noise factor, SSF = {(Total Noise at output, at OFB,
originating from all frequencies) - (Load Noise originating from OFB)} /
(Input Source Noise originating from IFB).
DSNF
DSNF outputs a double side-band noise figure as a function of the IFB
points.
DSNF = 10 Log(DSF)
Double side-band noise factor, DSF = {(Total Noise at output, at the
OFB, originating from all frequencies) - (Load Noise originating from the
OFB)} / (Input Source Noise originating from the IFB and from the
image of IFB).
INOISE
Outputs input referred noise which can be printed, probed, or
measured.
Output Data Files
An HBNOISE analysis produces these output data files:
■
Output from the .PRINT statement is written to a .printpn# file.
■
Output from the .PROBE statement is written to a .pn# file.
Both the *.printpn# and *.pn# files output data against the input frequency
band points.
■
Standard output information is written to a .lis file:
•
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•
HBNOISE linear solver method
•
HBNOISE simulation time
•
total simulation time
See also Using Noise Analysis Results as Input Noise Sources.
Measuring HBNOISE Analyses with .MEASURE
Note:
A .MEASURE HBNOISE statement cannot contain an expression that uses a
HBNOISE variable as an argument. Also, you cannot use a .MEASURE
HBNOISE statement for error measurement and expression evaluation of
HBNOISE.
The .MEASURE HBNOISE syntax supports several types of measurements:
■
Find-when
.MEASURE HBNOISE result FIND out_var1
+ AT = Input_Frequency_Band value
The previous measurement yields the result of a variable value at a specific
IFB point.
.MEASURE HBNOISE result FIND out_var1
+ WHEN out_var2 = out_var3
The previous measurement yields the result at the input frequency point
when out_var2 == out_var3.
.MEASURE HBNOISE result WHEN out_var2 = out_var3
The previous measurement yields the input frequency point when out_var2
== out_var3.
■
Average, RMS, min, max, and peak-to-peak
.MEASURE HBNOISE result <RMS> out_var < FROM = IFB1 >
+ < TO = IFB2 >
■
Integral evaluation
.MEASURE HBNOISE result INTEGRAL out_var
+ < FROM = IFB1 > < TO = IFB2 >
This measurement integrates the out_var value from the IFB1 frequency to
the IFB2 frequency.
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■
Derivative evaluation
.MEASURE HBNOISE result DERIVATIVE out_var AT = IFB1
This measurement finds the derivative of out_var at the IFB1 frequency
point.
Note:
.MEASURE HBNOISE cannot contain an expression that uses an
hbnoise variable as an argument. You also cannot use .MEASURE
HBNOISE for error measurement and expression evaluation of
HBNOISE.
■
Input referred noise
.MEASURE <HBNOISE|SNNOISE> result FIND inoise
+ AT = <IFB_value>
This measurement yields the result of the input referred noise at a specific
input frequency band point.
.MEASURE <HBNOISE|SNNOISE> result FIND inoise
+ WHEN out_var2 = out_var3
This measurement yields the result at the input frequency point when
out_var2 == out_var3.
.MEASURE HBNOISE result func inoise [FROM = <IFB1>]
+ [TO = <IFB2>]
Where func is one of the following measurement types:
272
•
AVG (average): Calculates the area under the inoise curve, divided by
the periods of interest.
•
MAX (maximum): Reports the maximum value of inoise over the
specified interval.
•
MIN (minimum): Reports the minimum value of inoise over the specified
interval.
•
PP (peak-to-peak): Reports the maximum value, minus the minimum
value of inoise over the specified interval.
•
RMS (root mean squared): Calculates the square root of the area under
the inoise curve, divided by the period of interest.
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.MEASURE HBNOISE result INTEGRAL inoise
+ [FROM =<IFB1>] [TO = <IFB2>]
This measurement integrates the inoise value from the IFB1 frequency to
the IFB2 frequency.
.MEASURE HBNOISE result DERIVATIVE inoise AT = <IFB1>
This measurement finds the derivative of inoise at the IFB1 frequency point.
The HSPICE RF optimization flow can read the measured data from a
.MEASURE HBNOISE analysis. This flow can be combined in the HSPICE RF
optimization routine with a .MEASURE HBTR analysis (see Using .MEASURE
with .HB Analyses on page 192) and a .MEASURE PHASENOISE analysis (see
Measuring Phase Noise with .MEASURE PHASENOISE on page 238).
Errors and Warnings
HBNOISE Errors
See the list of HBAC Errors and Warnings on page 260.
HBNOISE Example
This example performs an HB analysis, then runs an HBNOISE analysis over a
range of frequencies, from 9.0e8 to 9.2e8 Hz. Simulation outputs the output
noise at V(out) and the single side-band noise figure versus IFB, from 1e8 to
1.2e8 Hz, to the *.pn0 file. The netlist for this example is shown immediately
following.
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Shooting Newton Noise Analysis (.SNNOISE)
$$*-Ideal mixer + noise source
$ prints total noise at the output (1.57156p V/sqrt-Hz),
$ single-sideband noise figure, (3.01 dB)
$ double-sideband noise figure. (0 dB)
.OPTION PROBE
.OPTION POST=2
vlo lo 0 0.0 hb 1.0 0 1 1$ Periodic, HB Input
Ilo lo 0 0
rsrc rfin rf1 1.0$ Noise source
c1 0 if q='1.0e-9*v(lo)*v(rfin)' $ mixer element
g1 0 if cur=’1.0*v(lo)*v(rfin)’ $ mixer element
rout if 0 1.0
vrf rf1 0 $ hbac 2.0 0.0
.hb tones=1.0g nharms=4 $ sweep mval 1 2 1
.HBNOISE rout rsrc lin 11 0.90g 0.92g
.print HBNOISE onoise ssnf dsnf
.probe HBNOISE onoise ssnf dsnf
.end
Shooting Newton Noise Analysis (.SNNOISE)
A SNNOISE (Shooting Newton noise) analysis simulates the noise behavior in
periodic systems. It uses a Periodic AC (PAC) algorithm to perform noise
analysis of nonautonomous (driven) circuits under periodic, steady-state tone
conditions. SNNOISE is similar to the HBNOISE analysis.
The PAC method simulates noise assuming that the stationary noise sources
and/or the transfer function from the noise source to a specific output are
periodically modulated.
■
The modulated noise source (thermal, shot, or flicker) is modeled as a
cyclostationary noise source.
■
A PAC algorithm solves the modulated transfer function.
■
You can also use the SNNOISE PAC method with correlated noise sources,
including the MOSFET Level 9 and Level 11 models, and the behavioral
noise source in the G-element (Voltage Dependent Current Source).
You use the .SNNOISE statement to perform a Periodic Noise Analysis.
Supported Features
SNNOISE supports the following features:
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■
All existing HSPICE RF noise models.
■
Uses Shooting Newton to generate the steady-state solution.
■
Unlimited number of sources.
■
Includes stationary, cyclostationary, frequency-dependent, and correlated
noise effects.
■
Swept parameter analysis.
■
Results are independent of the number of SNAC sources in the netlist.
Prerequisites and Limitations
The following prerequisites and limitations apply to SNNOISE:
■
Requires one .SN statement (which determines the steady-state solution).
■
Requires at least one Periodic source. Does not recognize HB sources.
■
Requires placing the parameter sweep in the .SN statement.
Input Syntax
.SNNOISE [output] [insrc] [parameter_sweep]
+ <[n1+/-1]>
+ <listfreq=(frequencies|none|all)> <listcount=val>
+ <listfloor=val> <listsources=on|off>
Parameter
Description
output
Can be an output node, pair of nodes, or 2-terminal element. HSPICE RF
references the equivalent noise output to this node (or pair of nodes). Specify
a pair of nodes as V(n+,n-). If you specify only one node, V(n+), then HSPICE
RF assumes the second node is ground. You can also specify a 2-terminal
element name that refers to an existing element in the netlist.
insrc
An input source. If this is a resistor, HSPICE RF uses it as a reference noise
source to determine the noise figure. If the resistance value is 0, the result is
an infinite noise figure.
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Parameter
Description
parameter_sweep Frequency sweep range for the input signal. Also referred to as the input
frequency band (IFB) or fin). You can specify LIN, DEC, OCT, POI,
SWEEPBLOCK, DATA, MONTE, or OPTIMIZE sweeps. Specify the nsteps,
start, and stop frequencies using the following syntax for each type of sweep:
■
■
■
■
■
n1,+/-1
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
Index term defining the output frequency band (OFB or fout) at which the
noise is evaluated. Generally,
fout=ABS(n1*f1+/-fin)
Where:
f1 is the fundamental harmonic (tone) determined in the Shooting Newton
analysis
n1 is the associated harmonic multiplier
n1,n2,...,nk are the associated harmonic multipliers; n1 can be any nonnegative integer ≤ nharm defined in the .SN statement; +/-1 is fixed, either
+1 or -1
fin is the IFB defined by parameter_sweep.
The default index term is [1,-1]. For a single tone analysis, the default mode
is consistent with simulating a low-side, down conversion mixer where the RF
signal is specified by the IFB and the noise is measured at a down-converted
frequency that the OFB specifies. In general, you can use the [n1,+/-1] index
term to specify an arbitrary offset. The noise figure measurement is also
dependent on this index term. See Specifying Variant Indicesbelow. (See
alsoMeasuring SNNOISE Analyses with .MEASURE.)
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Parameter
Description
listfreq
Prints the element noise value to the .lis file. You can specify at which
frequencies the element noise value is printed. The frequencies must match
the sweep_frequency values defined in the parameter_sweep, otherwise
they are ignored.
In the element noise output, the elements that contribute the largest noise are
printed first. The frequency values can be specified with the NONE or ALL
keyword, which either prints no frequencies or every frequency defined in
parameter_sweep. Frequency values must be enclosed in parentheses. For
example:listfreq=(none)
listfreq=(all)
listfreq=(1.0G)
listfreq=(1.0G, 2.0G)
The default value is NONE.
listcount
Prints the element noise value to the .lis file, which is sorted from the largest
to smallest value. You do not need to print every noise element; instead, you
can define listcount to print the number of element noise frequencies. For
example, listcount=5 means that only the top 5 noise contributors are
printed. The default value is 1.
listfloor
Prints the element noise value to the .lis file and defines a minimum
meaningful noise value (in V/Hz1/2 units). Only those elements with noise
values larger than listfloor are printed. The default value is 1.0e-14 V/
Hz1/2.
listsources
Prints the element noise value to the .lis file when the element has multiple
noise sources, such as a FET, which contains the thermal, shot, and 1/f noise
sources. You can specify either ON or OFF: ON Prints the contribution from
each noise source and OFF does not. The default value is OFF.
Specifying Variant Indices
.SNNOISE is the appropriate HSPICE RF analysis for the computation of noise
at the output of a sample and hold circuit. When using .SNNOISE, you need to
specify the indices as [0,1] instead of the default [1,-1]. When you specify the
indices as [0,1], you get results that are “what you see is what you get” with
respect to the frequency sweep specified in the .SNNOISE command. There
are two more important things to consider when using .SNNOISE. You must
use enough harmonics to resolve the clock edge and you will want a high
density of SN time points. It is recommended that the number of time points be
between 2 and 20 times the number of harmonics used. To increase the density
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of the time points during the .SN analysis, you can use the option DELMAX to
specify a maximum time step.
For example, you can use the following settings:
.OPTION SNACCURACY=50
.OPTION DELMAX=5p
.SN TONES=5e6 nharms=2000 trinit=800n
.SNNOISE v(vo) vsin [0,1] dec 100 1e6 1e9
This example uses 100 points per decade for the frequency sweep instead of
1000 points per decade. This is to speed up the simulation. It does not affect
the accuracy of the results.
Output Syntax
This section describes the syntax for the SNNOISE .PRINT and .PROBE
statements.
.PRINT and .PROBE Statements
.PRINT SNNOISE <ONOISE> <NF> <SSNF> <DSNF> <INOISE>
.PROBE SNNOISE <ONOISE> <NF> <SSNF> <DSNF> <INOISE>
Parameter
Description
ONOISE
Outputs the voltage noise at the output frequency band (OFB) across
the output nodes in the .SNNOISE statement. The data is plotted as a
function of the input frequency band (IFB) points. Units are in V/Hz1/2.
Simulation ignores ONOISE when applied to autonomous circuits.
NF
SSNF
NF and SSNF both output a single-side band noise figure as a function
of the IFB points:
NF = SSNF = 10 Log(SSF)
Single side-band noise factor, SSF = {(Total Noise at output, at OFB,
originating from all frequencies) - (Load Noise originating from OFB)} /
(Input Source Noise originating from IFB).
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Parameter
Description
DSNF
DSNF outputs a double side-band noise figure as a function of the IFB
points.
DSNF = 10 Log(DSF)
Double side-band noise factor, DSF = {(Total Noise at output, at the
OFB, originating from all frequencies) - (Load Noise originating from the
OFB)} / (Input Source Noise originating from the IFB and from the
image of IFB).
INOISE
Outputs input referred noise which can be printed, probed, or
measured.
Output Data Files
An SNNOISE analysis produces these output data files:
■
Output from the .PRINT statement is written to a .printsnpn# file.
■
Output from the .PROBE statement is written to a .snpn# file.
Both the *.printsnpn# and *.pn# files output data against the input frequency
band points.
■
Standard output information is written to a .lis file:
•
simulation time
•
SNNOISE linear solver method
•
SNNOISE simulation time
•
total simulation time
See also Using Noise Analysis Results as Input Noise Sources.
Measuring SNNOISE Analyses with .MEASURE
Note:
A .MEASURE SNNOISE statement cannot contain an expression that uses a
SNNOISE variable as an argument. Also, you cannot use a .MEASURE
SNNOISE statement for error measurement and expression evaluation of
SNNOISE.
The .MEASURE SNNOISE syntax supports four types of measurements:
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■
Find-when
.MEASURE SNNOISE result FIND out_var1
+ At = Input_Frequency_Band value
The previous measurement yields the result of a variable value at a specific
IFB point.
.MEASURE SNNOISE result FIND out_var1
+ WHEN out_var2 = out_var3
The previous measurement yields the result at the input frequency point
when out_var2 == out_var3.
.MEASURE SNNOISE result WHEN out_var2 = out_var3
The previous measurement yields the input frequency point when out_var2
== out_var3.
■
Average, RMS, min, max, and peak-to-peak
.MEASURE SNNOISE result <RMS> out_var < FROM = IFB1 >
+ < TO = IFB2 >
■
Integral evaluation
.MEASURE SNNOISE result INTEGRAL out_var
+ < FROM = IFB1 > < TO = IFB2 >
This measurement integrates the out_var value from the IFB1 frequency to
the IFB2 frequency.
■
Derivative evaluation
.MEASURE SNNOISE result DERIVATIVE out_var AT = IFB1
This measurement finds the derivative of out_var at the IFB1 frequency
point.
Note:
.MEASURE SNNOISE cannot contain an expression that uses an
hbnoise variable as an argument. You also cannot use .MEASURE
SNNOISE for error measurement and expression evaluation of
SNNOISE.
■
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.MEASURE SNNOISE result FIND inoise
+ AT = <IFB_value>
This measurement yields the result of the input referred noise at a specific
input frequency band point.
.MEASURE SNNOISE result FIND inoise
+ WHEN out_var2 = out_var3
This measurement yields the result at the input frequency point when
out_var2 == out_var3.
.MEASURE SNNOISE result func inoise [FROM = <IFB1>]
+ [TO = <IFB2>]
Where func is one of the following measurement types:
•
AVG (average): Calculates the area under the inoise curve, divided by
the periods of interest.
•
MAX (maximum): Reports the maximum value of inoise over the
specified interval.
•
MIN (minimum): Reports the minimum value of inoise over the specified
interval.
•
PP (peak-to-peak): Reports the maximum value, minus the minimum
value of inoise over the specified interval.
•
RMS (root mean squared): Calculates the square root of the area under
the inoise curve, divided by the period of interest.
.MEASURE SNNOISE result INTEGRAL inoise
+ [FROM =<IFB1>] [TO = <IFB2>]
This measurement integrates the inoise value from the IFB1 frequency to
the IFB2 frequency.
.MEASURE SNNOISE result DERIVATIVE inoise AT = <IFB1>
This measurement finds the derivative of inoise at the IFB1 frequency point.
SNNOISE Analysis Example
This example performs an SN analysis, then runs an SNNOISE analysis over a
range of frequencies, from 9.0e8 to 9.2e8 Hz. Simulation outputs the output
noise at V(out) and the single side-band noise figure versus IFB, from 9.0e8 to
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Periodic Time-Dependent Noise Analysis (.PTDNOISE)
9.2e8 Hz, to the *.pn0 file. The netlist for this example is shown immediately
following.
*
$$*-Ideal mixer + noise source
$ prints total noise PSD at the output (2.47e-20 V^2) when q=0
$ single-sideband noise figure, (3.01 dB)
$ double-sideband noise figure. (0 dB)
.OPTION PROBE
.OPTION POST=2
vlo lo 0 0.0 cos (0 1.0 1.0g 0 0 0)
Ilo lo 0 0
rsrc rfin rf1 1.0$ Noise source
g1 0 if cur='1.0*v(lo)*v(rfin)' $ mixer element
c1 0 if q='1.0e-9*v(lo)*v(rfin)' $ mixer element
rout if 0 1.0
vrf rf1 0 $ hbac 2.0 0.0
.option delmax=0.002n
.SN tones=1G
nharms=4 trstab=10n
.SNNOISE rout rsrc lin 11 0.90g 0.92g
.probe SNNOISE onoise ssnf dsnf
.print SNNOISE onoise ssnf dsnf
.end
Periodic Time-Dependent Noise Analysis (.PTDNOISE)
While HBNOISE and SNNOISE calculate a time-averaged power spectral
density, there are applications where a characterization of the timedependence of the noise is required. These applications include computation of
jitter associated with a noisy signal crossing a threshold and computation of the
noise associated with discretization of an analog signal, which computes the
noise in a periodically driven circuit at a point in time. Periodic Time-Dependent
noise analysis (PTDNOISE) calculates the noise spectrum and the total noise
at a point in time. Jitter in a digital threshold circuit can then be determined from
the total noise and the digital signal slew rate.
Circuits driven by large periodic signals produce cyclostationary noise, that is,
the noise characteristics are periodic in time. Cyclostationary noise can be
characterized in several ways, with the particular application determining which
is appropriate.[9] The time-average power spectral density (PSD) ignores
frequency correlations in the noise, but is adequate when the fundamental
frequency of the cyclostationary noise is much larger than the bandwidth of
interest. The time-average PSD is calculated in the HBNOISE/SNNOISE
analyses. [10]
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The harmonic power spectral density (HPSD) or equivalently, the autocorrelation function, R(t1,t2), contains the correlation information between
noise sidebands that is necessary to build behavioral cyclostationary noise
sources and to separate the amplitude modulation (AM) and phase modulation
(PM) noise components. (See Amplitude Modulation/Phase Modulation
Separation for more information.
The time-dependent power spectral density (TDPSD) can be integrated over
frequency to yield the time-dependent noise (TDN). TDN can then be used to
determine jitter associated with a noisy signal crossing a threshold. PTDNOISE
analysis allows the calculation of TDPSD, TDN, and jitter. In addition, you can
calculate both the time-domain power spectral density (TDSN) and the
integrated noise (time-dependent noise, TDN) at multiple time points.
By measuring the jitter associated with a noisy signal crossing a threshold, jitter
is modeled by displacing the time in a noise free signal v(t) with a stochastic
process j(t).
Equation 49
V jitter ( t ) = v ( t + j ( t ) )
We can also determine the voltage at this node including the time-dependent
noise n(t):
Equation 50
Vn ( t ) = v ( t ) + n ( t )
by equating these two representations, expanding in a Taylor series, and
dropping higher order terms, as follows:
Equation 51
V ( t ) + n ( t ) = v ( t + j ( t ) ) = v ( t ) + dv ( t ) ⁄ dt ⋅ j ( t ) + …
Equation 52
N ( t ) = dv ( t ) ⁄ dt ⋅ j ( t )
In terms of variances, jitter is then defined as:
Equation 53
2
Var ( j ( t ) ) = n ( t ) ⁄ ( dv ( t ) ⁄ dt )
2
PTDNOISE Input Syntax
.PTDNOISE output TIME =<val|meas|sweep>
+ frequency_sweep
+ [listtime=-<(time_points|none|all)>]
+ [<listfreq=<(frequencies|none|all)>]
+ [listcount=<val>] [<listfloor=val>]
+ [<listsources=on|off>]
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Parameter
Description
output
Is an output node, pair of nodes, or 2-terminal elements. HSPICE RF
references the equivalent noise output to this node (or pair of nodes). Specify
a pair of nodes as V(n+,n-); only one node as V(n+, n-). If you specify only one
node, V(n+), then HSPICE RF assumes the second node is ground. You can
also specify a 2-terminal element name that refers to an existing element in
the netlist.
TIME
Time point, time points, or measure value where the time domain noise is
evaluated
■
■
■
TIME=<val> - a single time point at which time domain noise is measured.
The value can be a numeric value or a parameter.
TIME=<meas> - the result name of a .MEASURE statement located in the
netlist. The time point or points generated from the .MEASURE statement
are used to evaluate the noise characteristics. This is useful if you want to
evaluate noise or jitter when a signal reaches some threshold value. If
TIME=<meas> is specified and a strobed jitter measurement is specified a
warning statement will be generated indicating that the measurement will
be ignored.
TIME=<sweep> - a set of time points at which time domain noise is
measured. The sweep can be specified as a LIN, DEC, OCT, POI,
SWEEPBLOCK or DATA sweep. If TIME = <sweep> is specified, all strobed
jitter measurements will be ignored. Specify the number of steps (nsteps),
start, and stop time using the following syntax for each type of sweep:
- LIN nsteps start stop
- DEC nsteps start stop
- OCT nsteps start stop
- POI nvalues freq_values
- SWEEPBLOCK sweepblock_name
- DATA data name
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Parameter
Description
frequency_sweep Frequency sweep range for the output noise spectrum. The upper and lower
limits also specify the integral range in calculating the integrated noise value.
Specify LIN,DEC, OCT, POI, SWEEPBLOCK, DATA sweeps. Specify the
nsteps, start, and stop frequencies using the following syntax for each type of
sweep:
■
■
■
■
■
listtime
LIN nsteps start stop
DECnsteps start stop
OCT nsteps start stopPOI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA data_name
Prints the element noise spectral density value to the .lis file. This information
is only printed if a noise spectrum is requested in a .PRINT or .PROBE
statement. You can specify at which time points the element noise value is
printed. The frequencies must match the time values defined by the TIME
keyword, otherwise they will be ignored. The time values can be specified with
the NONE or ALL keyword, which either prints no time points or all time points
defined in by the TIME keyword. Time values must be enclosed in
parentheses. For example:
listtime=(none)
listtime=(all)
listtime=(1n)
listtime=(1n, 2n)
The default value is the first time value.
Note: The noise listing is printed in frequency blocks for each time point. See
the remaining list parameters, listfreq, listcount and listsources to determine
how each frequency block is constructed.
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Periodic Time-Dependent Noise Analysis (.PTDNOISE)
Parameter
Description
listfreq
Prints the element noise value to the .lis file. This information is only printed if
a noise spectrum is requested in a PRINT or PROBE statement. (See
PTDNOISE Output Syntax and File Format.) You can specify which
frequencies the element noise is printed. The frequencies must match the
sweep_frequency values defined in the frequency_sweep, otherwise they are
ignored.
In the element noise output, the elements that contribute the largest noise are
printed first. The frequency values can be specified with the NONE or ALL
keyword, which either prints no frequencies or every frequency defined in
frequency_sweep. Frequency values must be enclosed in parentheses. For
example:
listfreq=(none)
listfreq=(all)
listfreq=(1.0G)
■
listfreq=(1.0G, 2.0G)
The default value is NONE.
listcount
Prints the element noise value to the .lis file, which is sorted from the largest
to smallest value. You do not need to print every noise element; instead, you
can define listcount to print the number of element noise frequencies. For
example, listcount=5 means that only the top 5 noise contributors are
printed. The default value is 1.
listfloor
Prints the element noise value to the .lis file and defines a minimum
meaningful noise value (in V/Hz1/2 units). Only those elements with noise
values larger than listfloor are printed. The default value is 1.0e-14 V/Hz1/
2
.
listsources
Prints the element noise value to the .lis file when the element has multiple
noise sources, such as a MOSFET, which contains the thermal, shot, and 1/f
noise sources. You can specify either ON or OFF: ON prints the contribution
from each noise source and OFF does not. The default value is OFF.
PTDNOISE Output Syntax and File Format
PTDNOISE output syntax allows for the output of a single parameter: onoise,
where, onoise is the noise voltage spectral density at each frequency point
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Periodic Time-Dependent Noise Analysis (.PTDNOISE)
specified by the frequency_sweep keyword for the time points specified by the
TIME keyword. The units are V Hz .
.PROBE PTDNOISE <onoise>
.PRINT PTDNOISE <onoise>
Parameter Units
Description
onoise
Noise voltage spectral density at each frequency point specified
by frequency_sweep at the time point specified by time_value
V Hz
Output File Format
The following PTDNOISE output files are generated depending on the user
input:
File
Description
*.printptn#
Writes output from the .PRINT statement when using HB to obtain the
steady state solution
*.ptn#
Writes output from the .PROBE statement when using HB to obtain
the steady state solution
*.printsnptn#
Reports output from the .PRINT statement when using SN to obtain
the steady state solution.
*.snptn#
Writes output from the .PROBE statement when using SN to obtain
the steady state solution.
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Periodic Time-Dependent Noise Analysis (.PTDNOISE)
File
Description
*.lis
Standard output file *.lis contains the following information:
Performance Statistics Log
Number of Nodes
Number of FFT Points
Number of Equations
Memory in use
Maximum Krylov iterations
Maximum Krylov Dimension
Target GMRES Residual
Gmres Residual
Actual Krylov Iterations taken
Frequency (swept input frequency values)
Noise source contributions are listed sequentially and are controlled
by the PTDNOISE command line parameters: listtime, listfreq,
listcount, listfloor, and listsources.
.MEASURE Syntax and File Format
The syntax for .MEASURE PTDNOISE is:
.MEASURE PTDNOISE result <integnoise|jitter|slewrate>
.MEASURE PTDNOISE allows for the measurement of these parameters:
integnoise, time-point, tdelta-value, slewrate, and strobed jitter.
288
Parameter
Units Description
integnoise
V
slewrate
v/sec Output signal slewrate at the time point specified by
TIME=val.
jitter
sec
Voltage noise integrated over a frequency range specified
by frequency_range at the time point specified by
TIME=<val>.
Calculated from the noise voltage (integrated over the
frequency range specified by frequency_range), divided by
the slew rate at the same node(s), at the time point specified
by TIME=<val>.
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Periodic Time-Dependent Noise Analysis (.PTDNOISE)
Note:
.MEASURE PTDNOISE will be ignored when a TIME=sweep is specified in
the netlist. A warning statement will be issued indicating that the measure
statement will be ignored in the simulation.
Measure File Format
File
Description
*.msnptn#
Contains output from the .MEASURE statement when using .SN to
obtain the steady state solution.
Error Handling and Warnings
Error messages are generated under the following circumstances:
■
PTDNOISE frequency sweep includes negative frequencies. PTDNOISE
allows only frequencies that are greater than or equal to zero.
■
PTDNOISE time sweep includes negative times. PTDNOISE allows only
time points that are greater than or equal to zero.
■
No SN statement is specified (error at parser). PTDNOISE requires an SN
statement to generate the steady-state solution.
■
Incorrect match to .MEASURE statement.
A warning is issued for a PTDNOISE convergence failure. When the gmres
solver reaches the maximum number of iterations and the residual is greater
than the specified tolerance, PTDNOISE generates a warning and then
continue as if the data were valid. The Warning reports the following
information:
■
Final GMRES Residual
■
Target GMRES Residual
■
Maximum Krylov Iterations
■
Actual Krylov Iterations taken
Usage Example
The following test case illustrates the PTDNOISE analysis for a simple inverter.
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Periodic Time-Dependent Noise Analysis (.PTDNOISE)
* Simple RC + Inverter - rcInvPTDNoise.sp
* rrd Jan 03, 2007
* Simulates PSD(t,f) of a simple inverter
* sweep time points
.param f0 = 5.0e8
.sn tones=f0 nharms=4 trinit=10n
.PTDNOISE v(out1) TIME=lin 3 0 2n TDELTA=.1n dec 5 1e5 1e10
+ listfreq=(1e6,1e8)
+ listcount=1
+ listsources=ON
.MEASURE PTDNOISE strobejit STROBEJITTER onoise FROM = 1e4 TO =
1e10
$.measure SN t1 trig AT=0 targ v(out1) val=1.5 fall=1
.opt post
.probe ptdnoise onoise
.print ptdnoise onoise
.probe sn v(out1)
vd
vdd 0
3.0
.global vdd
vgate in0 0
COS(1.5 1.4 'f0'
rin
in0 in1 50
rout out1 0
.1g
0 0 0)
xo1 in1 out1 inv
.subckt inv in out
m1 out in 0 0 n l=350e-9 w=4.5e-6
m2 out in vdd vdd p l=350e-9 w=4.5e-6
.ends
.MODEL N NMOS
+Level= 49 Tnom=27.0 version =3.1 TLEVC= 1
*
***
.MODEL P PMOS
+Level= 49 Tnom=27.0 version =3.1 TLEVC= 1
.end
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Multitone Harmonic Balance Transfer Function Analysis (.HBXF)
Multitone Harmonic Balance Transfer Function Analysis (.HBXF)
The .HBXF command calculates the transfer function from a given source in
the circuit to a designated output. Frequency conversion is calculated from the
input frequencies to a single output frequency that is specified with the
command. The relationship between the .HBXF command and the input/output
is expressed in the following equation:
Equation 54
Y m ( jω0 ) =
∑ HBXFm,n ( jω0 ,j ( ω + Δω) ) ⋅
X n ( j ( ω + Δω) )
ωε W
Where:
■
HBXF m ,n ( jω0 ,j ( ω + Δω) ) is the transfer function from input port n to the
output port m
■
W is the set of all possible harmonics
■
ω + Δω is the input frequency
■
Δω is the offset frequency
■
m is the output node number
■
n is the input node number
■
ω0 is the output frequency
■
Y is the output (voltage or current)
■
X is the input (voltage or current)
Supported Features
The .HBXF command supports the following features:
■
All existing HSPICE RF models and elements
■
Sweep parameter analysis
■
Unlimited number of HB sources
Prerequisites and Limitations
The following prerequisites and limitations apply to the .HBXF command:
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Multitone Harmonic Balance Transfer Function Analysis (.HBXF)
■
Only one .HBXF statement is required. If you use multiple .HBXF
statements, HSPICE RF only uses the last .HBXF statement.
■
At least one .HB statement is required, which determines the steady-state
solution.
■
Parameter sweeps must be placed in .HB statements.
Input Syntax
.HBXF out_var freq_sweep
Parameter
Description
out_var
Specify i(2_port_elem) or V(n1<,n2>)
freq_sweep
Frequency sweep range for the input signal (also referred to as the
input frequency band (IFB or fin)). A sweep of type LIN, DEC, OCT,
POI, or SWEEPBLOCK. Specify the nsteps, start, and stop
frequencies using the following syntax for each type of sweep:
■
LIN nsteps start stop
DEC nsteps start stop
■
OCT nsteps start stop
■
POI nsteps freq_values
■
SWEEPBLOCK = BlockName
Specify the frequency sweep range for the output signal. HSPICE
RF determines the offset frequency in the input sidebands; for
example,
■
f1 = abs(fout - k*f0) s.t. f1<=f0/2
The f0 is the steady-state fundamental tone, and f1 is the input
frequency.
Output Syntax
This section describes the syntax for the HBXF .PRINT and .PROBE
statements.
.PRINT and .PROBE Statements
.PRINT HBXF TYPE(NODES | ELEM)
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Multitone Harmonic Balance Transfer Function Analysis (.HBXF)
.PROBE HBXF TYPE(NODES | ELEM)
Parameter
Description
TYPE
TYPE can be one of the following:
■
TFV = existing source
TFI = placeholder value for the current source
attached to the given node.
The transfer function is computed on the output
variables and input current or voltage.
■
NODES | ELEM
NODES or ELEM can be one of the following:
■
■
■
Voltage type – a single node name (n1), or a pair
of node names, (n1,n2)
Current type – an element name (elemname)
Power type – a resistor (resistorname) or port
(portname) element name.
Output Data Files
An HBXF calculation produces these output data files:
■
Output from the .PRINT statement is written to a .printxf# file.
•
The output is in ohms, siemens, or undesignated units, and the header
in the output file is Z(..). Y(..) or GAIN(..).
■
Output from the .PROBE statement is written to an .xf# file.
■
Reported performance log statistics are written to a .lis file:
•
HBXF CPU time
•
HBXF peak memory usage
Example
Based on the HB analysis, the following example computes the transimpedance from isrc to v(1).
.hb tones=1e9 nharms=4
.hbxf v(1) lin 10 1e8 1.2e8
.print hbxf tfv(isrc) tfi(n3)
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Shooting Newton Transfer Function Analysis (.SNXF)
HBXF Test Listing
* Test HBXF: nonlinear order-2 poly equation
.OPTIONS PROBE
.OPTIONS POST=2
vlo lo 0 cos(0 1.0 1g 0 0) tranforhb=1
rlo lo 0 50
vrf1 rf1 0 0
rrf1 rf1 0 50
E1 out 0 POLY(2) lo 0 rf1 0 0 1 1 1 10 1
rout out 0 50
.hb tones=1g nharms=5
.hbxf v(out) lin 2 100meg 200meg
.print hb v(out) v(rf1) v(lo)
.print hbxf tfv(vrf1) tfv(vlo)
.end
Shooting Newton Transfer Function Analysis (.SNXF)
The .SNXF command calculates transfer functions from an arbitrary number of
small signal sources to a designated output in a circuit under periodic steady
state conditions. Frequency conversion is calculated from multiple input
frequencies to a single output at a single frequency that is specified on the
command line.
Prerequisites and Limitations
The following prerequisites and limitations apply to the .SNXF command:
■
Only one .SNXF statement is required. If you use multiple .SNXF
statements, HSPICE RF only uses the last one issued.
■
At least one .SN statement is required, which determines the steady-state
solution.
■
Parameter sweeps must be placed in .SN statements.
Input Syntax
.SNXF out_var <freq_sweep>
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Parameter Description
Parameter
Description
out_var
Specify i(2_port_elem) or V(n1<,n2>)
freq_sweep
Frequency sweep range for the input signal (also referred to as the
input frequency band (IFB or fin)). A sweep of type LIN, DEC, OCT,
POI, or SWEEPBLOCK. Specify the nsteps, start, and stop
frequencies using the following syntax for each type of sweep:
■
LIN nsteps start stop
DEC nsteps start stop
■
OCT nsteps start stop
■
POI nsteps freq_values
■
SWEEPBLOCK = BlockName
Specify the frequency sweep range for the output signal. HSPICE
RF determines the offset frequency in the input sidebands Fin,
where Fin = abs(n*F0 +/- Fout). F0 is the steady-state fundamental
tone, and Fout is the output frequency. SNXF then generates the
transfer functions from all of the input sidebands (the Fin values) to
the output frequency Fout.
■
Output Syntax
This section describes the syntax for the SNXF .PRINT and .PROBE
statements.
.PRINT and .PROBE Statements
.PRINT SNXF TYPE(NODES | ELEM)
.PROBE SNXF TYPE(NODES | ELEM)
Parameter Description
TYPE can be one of the following:
■
TFV = existing source
■
TFI = placeholder value for the current source attached to the given node.
The transfer function is computed on the output variables and input current or
voltage.NODES | ELEM NODES or ELEM can be one of the following:
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Shooting Newton Transfer Function Analysis (.SNXF)
■
Voltage type – a single node name (n1), or a pair of node names, (n1,n2)
■
Current type – an element name (elemname)
■
Power type – a resistor (resistorname) or port (portname) element name
Output Data Files
An SNXF calculation produces these output data files:
■
Output from the .PRINT statement is written to a .printsnxf# file. The output
is in ohms, siemens, or undesignated units, and the header in the output file
is Z(..). Y(..) or GAIN(..).
■
Output from the .PROBE statement is written to a .snxf# file.
Reported performance log statistics are written to a .lis file:
■
SNXF CPU time
■
SNXF peak memory usage
Example
Based on the SN analysis, the following example computes the
transimpedance from isrc to v(1).
.SN tones=1e9 nharms=4
.SNXF v(1) lin 10 1e8 1.2e8
print SNXF TFV(isrc) TFI(n3)
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Shooting Newton Transfer Function Analysis (.SNXF)
SNXF Test Listing
* Test SNXF: nonlinear order-2 poly equation
.OPTIONS PROBE
.OPTIONS POST=2
vlo lo 0 cos(0 1.0 1g 0 0)
rlo lo 0 50
vrf1 rf1 0 0
rrf1 rf1 0 50
E1 out 0 POLY(2) lo 0 rf1 0 0 1 1 1 10 1
rout out 0 50
.opt delmax=.01n
.sn tones=1g nharms=5
.snxf v(out) lin 2 100meg 200meg
.print sn v(out) v(rf1) v(lo)
.print snxf tfv(vrf1) tfv(vlo)
.end
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References
References
[1] S. Maas, Nonlinear Microwave Circuits, Chapter 3, IEEE Press, 1997.
[2] R. Gilmore and M.B. Steer, “Nonlinear Circuit Analysis Using the Method of
Harmonic Balance - A Review of the Art, Part I, Introductory Concepts.”
International Journal of Microwave and Millimeter-wave Computer-Aided
Engineering, Volume 1, No. 1, pages 22-37, 1991.
[3] R. Gilmore and M.B. Steer, “Nonlinear Circuit Analysis Using the Method of
Harmonic Balance - A Review of the Art. Part II. Advanced Concepts.”
International Journal of Microwave and Millimeter-wave Computer-Aided
Engineering, Volume 1, No. 2, pages 159-180, 1991.
[4] V. Rizzoli, F. Mastri, F. Sgallari, G. Spaletta, “Harmonic-Balance Simulation
of Strongly Nonlinear Very Large-Size Microwave Circuits by Inexact
Newton Methods,” MTT-S Digest, pages 1357-1360, 1996.
[5] S. Skaggs, Efficient Harmonic Balance Modeling of Large Microwave
Circuits, Ph.D. thesis, North Carolina State University, 1999.
[6] R.S. Carson, High-Frequency Amplifiers, 2nd Edition, John Wiley & Sons,
1982
[7] S.Y. Liao, Microwave Circuit Analysis and Amplifier Design, Prentice-Hall,
1987.
[8] Y. Saad, Iterative Methods for Sparse Linear Systems, PWS Publishing
Company, 1995.
[9] J. Roychowdhury, D. Long, and P. Feldmann, “Cyclostationary Noise
Analysis of Large RF Circuits with Multitone Excitations,” IEEE Journal of
Solid-State Circuits, volume 33, pages 324–336, March 1998.
[10] A. Demir, A. Sangiovanni-Vincentelli, “Analysis and Simulation of Noise in
Nonlinear Electronic Circuits and Systems”, Kluwer Academic, 1998.
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11
S-parameter Analyses
11
Describes how to do frequency translation and large-signal
S-parameter extraction, as well as noise parameter calculation.
These topics are covered in the following sections:
■
Frequency Translation S-Parameter (HBLIN) Extraction on page 300
■
Large-Signal S-parameter (HBLSP) Analysis on page 307
This chapter discusses various techniques supported in HSPICE RF for
extracting circuit scattering parameters. Since RF circuits can operate under
large-signal and small-signal conditions, there are several types of scattering
parameters that are useful to measure.
Linear small-signal scattering parameters represent the RF frequency-domain
transfer characteristics for a circuit that is operating at its DC bias condition, but
the stimulus and response signals are sufficiently small that they do not
influence the operating point. This type of analysis is performed using the .LIN
analysis, which is supported in both HSPICE and HSPICE RF. For information
on doing small-signal S-parameter analysis (.LIN), please see (Linear Network
Parameter Analysis) in the HSPICE User Guide: Simulation and Analysis.
In the case of RF mixers and receiver front-ends, some of the input and output
frequencies of interest involve a frequency translation. This translation is
intentional and caused by nonlinear mixing in the circuit due to devices being
driven by large-signal periodic waveforms. This type of scattering parameter
analysis therefore must begin by solving the large-signal periodic response,
and then finding the small-signal behavior about this large-signal operating
point. This capability is provided by the .HBLIN analysis, which has setup and
analysis control options similar to .LIN, but is capable of extracting
S-parameters about a large-signal periodic steady-state operating point.
In the case of circuits such as power amplifiers, the extraction of scattering
parameters is also important, but the circuit stimulus and response signals may
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Frequency Translation S-Parameter (HBLIN) Extraction
themselves be large-signal periodic waveforms. And, it can be important to
analyze how these S-parameter vary as a function of input power levels. This
capability is provided by the .HBLSP Large-Signal S-parameter analysis, which
uses large-signal stimulus signals for the S-parameter extractions.
Frequency Translation S-Parameter (HBLIN) Extraction
Frequency translation scattering parameter (S-parameter) extraction is used to
describe N-port circuits that exhibit frequency translation effects, such as
mixers. The analysis is similar to the existing LIN analysis, except that the
circuit is first linearized about a periodically varying operating point instead of a
simple DC operating point. After the linearization, the S-parameters between
circuit ports that convert signals from one frequency band to another are
calculated.
You use the .HBLIN statement to extract frequency translation S-parameters
and noise figures.
Frequency translation S-parameter describes the capability of a periodically
linear time varying systems to shift signals in frequency. The S-parameters for
a frequency translation system are similar to the S-parameters of a linear-timevarying system, it is defined as:
b = S⋅ a
Equation 55
b i,m ( ω)
S i,j;m,n ( ω) = ----------------a j,n ( ω)
a k ≠ j, p ≠ n ( ω) = 0
The incident waves, a i ,n ( ω) , and reflected waves, b i ,n ( ω) , are defined by using
these equations:
Equation 56
300
V i ( ω + nω0 ) + Z 0i I i ( ω + nω0 )
a i ,n ( ω) = ---------------------------------------------------------------------2 Z 0i
V i ( ω + nω0 ) – Z 0i I i ( ω + nω0 )
b i ,n ( ω) = --------------------------------------------------------------------2 Z 0i
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Frequency Translation S-Parameter (HBLIN) Extraction
Where,
■
ω0 is the fundamental frequency (tone).
■
n is a signed integer.
■
i is the port number.
■
a i ,n ( ω) is the input wave at the frequency ω + nω0 on the ith port.
■
b i ,n ( ω) is the reflected wave at the frequency ω + nω0 on the ith port.
■
V i ( ω + nω0 ) is the Fourier coefficient at the frequency ω + nω0 of the voltage
at port i.
■
I i ( ω + ωn 0 ) is the Fourier coefficient at the frequency ω + nω0 of the current
at port i.
■
Z 0i is the reference impedance at port i.
■
V and I definitions are Fourier coefficients rather than phasors.
For a multi-tone analysis, it can be expressed as:
b = S⋅ a
Si
Equation 57
b i ,m ,m ...m ( ω)
1
2
N
( ω) = ------------------------------------,j;m 1 ...m N ,n 1 ,n 2 ...n N
a j ,n ,n ...n ( ω)
1
2
N
a k ,p ,p ...p k ≠ j ,∇ p ≠ n ( ω) = 0
1 2
N
q
q
Equation 58
Where,
■
ωj is the ith tone.
The frequency translate S-parameters are calculated by applying different
n j ( j = 1 ∼N ) to different ports.
Limitations
The HBLIN analysis has these known limitations:
■
Noise parameters are not calculated for mixed-mode operation.
■
Only the S-parameters corresponding to the set of frequencies specified at
each port are extracted.
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N
N
∑
∑
⎛
⎞
⎛
⎞
⎜
⎟
⎜
n j ωj + Z 0i I i ω +
n j ωj⎟
Vi ω +
⎜
⎟
⎜
⎟
⎝
⎠
⎝
⎠
j=1
j=1
a i ,n ,n ...n ( ω) = ---------------------------------------------------------------------------------------------1 2
N
2 Z oi
N
N
∑
∑
⎛
⎞
⎛
⎞
⎜
⎟
⎜
n j ωj – Z 0i I i ω +
n j ωj⎟
Vi ω +
⎜
⎟
⎜
⎟
⎝
⎠
⎝
⎠
j=1
j=1
b i ,n ,n ...n ( ω) = ------------------------------------------------------------------------------------------1 2
N
2 Z oi
■
Multiple small-signal tones are not supported.
■
The port (P) element impedance cannot be specified as complex.
HB Analysis
An HB analysis is required prior to an HBLIN analysis. To extract the frequency
translation S-parameters, a sweep of the small-signal tone is necessary. You
can identify the small-signal tone sweep in the .HBLIN command or in the .HB
command together with a SS_TONE specification.
For additional information regarding HB analysis, see Harmonic Balance
Analysis on page 176.
Port-element
You must use a port (P) element as the termination at each port of the system.
To indicate the frequency band that the S-parameters are extracted from, it is
necessary to specify a harmonic index for each P-element.
Port Element Syntax
Without SS_TONE
Pxxx p n <n_ref> <PORT=portnumber >
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+ <HBLIN = [H1, H2, ... HN, +/-1]> ...
With SS_TONE
Pxxx p n <n_ref> <PORT=portnumber >
+ <HBLIN = [H1, H2, ... +/-1 ... HN]> ...
Parameter
Description
n_ref
Reference node used when a mixed-mode port is specified.
PORT
The port number. Numbered sequentially beginning with 1 with no
shared port numbers.
HBLIN
Integer vector that specifies the harmonic index corresponding to the
tones defined in the .HB command. The +/-1 term corresponds to the
small-signal tone specified by SS_TONE in the .HB command. If there
is no SS_TONE in the .HB command, the +/-1 term must be at the last
entry of HBLIN vector.
HBLIN Analysis
You use the .HBLIN statement to extract frequency translation S-parameters
and noise figures.
Input Syntax
Without SS_TONE
.HBLIN <frequency_sweep>
+ <NOISECALC = [1|0|yes|no]> <FILENAME=file_name>
+ <DATAFORMAT = [ri|ma|db]>
+ <MIXEDMODE2PORT = [dd|cc|cd|dc|sd|sc|cs|ds]>
With SS_TONE
.HBLIN <NOISECALC = [1|0|yes|no]> <FILENAME=file_name>
+ <DATAFORMAT = [ri|ma|db]>
+ <MIXEDMODE2PORT = [dd|cc|cd|dc|sd|sc|cs|ds]>
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Parameter
Description
frequency_sweep
Frequency sweep range for the input signal (also referred to as
the input frequency band (IFB) or fin). You can specify LIN,
DEC, OCT, POI, or SWEEPBLOCK. Specify the nsteps, start,
and stop frequencies using the following syntax for each type of
sweep:
■
■
■
■
■
■
NOISECALC
Enables calculating the noise figure. The default is no (0).
FILENAME
Specifies the output file name for the extracted S-parameters or
the object name after the -o command-line option. The default
is the netlist file name.
DATAFORMAT
Specifies the format of the output data file.
■
■
■
304
LIN nsteps start stop
DEC nsteps start stop
OCT nsteps start stop
POI nsteps freq_values
SWEEPBLOCK nsteps freq1 freq2 ... freqn
DATA=dataname
dataformat=RI, real-imaginary.
dataformat=MA, magnitude-phase. This is the default format
for Touchstone files.
dataformat=DB, DB(magnitude)-phase.
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Frequency Translation S-Parameter (HBLIN) Extraction
Parameter
Description
MIXEDMODE2PORT Describes the mixed-mode data map of output mixed mode Sparameter matrix. The availability and default value for this
keyword depends on the first two port (P element) configuration
as follows:
■
■
■
■
case 1: p1=p2=single-ended (standard-mode P element)
available: ss
default: ss
case 2: p1=p2=balanced (mixed-mode P element)
available: dd, cd, dc, cc
default: dd
case 3: p1=balanced p2=single-ended
available: ds, cs
default: ds
case 4: p1=single p2=balanced
available: sd, sc
default: sd
Example 1
Single-tone analysis with frequency translation. In this example, the 2-port Sparameters from RF (1G-del_f) to IF (del_f) are extracted. The LO signal is
specified by normal voltage source Vlo. The frequency on port 1 is in the RF
band, 1G-del_f, and the frequency on port 2 is in the IF band, del_f. The IF
band is swept from 0- to 100-MHz. The results are output to file ex1.s2p.
p1 RFin gnd port=1 HBLIN=(1,-1)
p2 IFout gnd port=2 HBLIN=(0,1)
Vlo LOin gnd DC 0 HB 2.5 0 1 1
.HB tones=1G harms=5
.HBLIN lin 5 0 100meg noisecalc=no filename=ex1
+ dataformat=ma
Example 2
Another single-tone analysis with frequency translation example. In this
example, the 3-port S-parameters are extracted. Port 3 provides the periodic
large signal. The frequency on port 1 is del_f, the frequency on port 2 is
1G*2-del_f, and the frequency on port 3 is 1G*1+del_f. The small-signal
frequency is swept from 0 to 100MHz. HBNOISE calculation is required. The
results are output to file ex2.s3p.
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p1 1 0 port=1 HBLIN=(0, 1)
p2 2 0 port=2 HBLIN=(2, -1)
p3 3 0 port=3 hb 0.5 0 1 1 HBLIN=(1, 1)
.HB tones=1G harms=5
.HBLIN lin 5 0 100meg noisecalc=yes filename=ex2
Output Syntax
This section describes the syntax for the HBLIN .PRINT and .PROBE
statements.
.PRINT and .PROBE Statements
.PRINT
.PROBE
.PRINT
.PROBE
.PRINT
.PROBE
HBLIN Smn | Smn(TYPE) | S(m,n) | S(m,n)(TYPE)
HBLIN Smn | Smn(TYPE) | S(m, n) | S(m, n)(TYPE)
HBLIN SXYmn | SXYmn(TYPE) | SXY(m,n) | SXY(m,n)(TYPE)
HBLIN SXYmn | SXYmn(TYPE) | SXY(m, n) | SXY(m, n)(TYPE)
HBLIN <NF> <SSNF> <DSNF>
HBLIN <NF> <SSNF> <DSNF>
Parameter
Description
Smn | Smn(TYPE) |
Complex 2-port parameters. Where:
S(m,n) | S(m,n)(TYPE)
■
m = 1 or 2
SXYmn | SXYmn(TYPE) |
■
n = 1 or 2
SXY(m,n) | SXY(m,n)(TYPE) ■ X and Y are used for mixed-mode S-parameter
output. The values for X and Y can be D
(differential), C (common), or S (single-end).
■
TYPE = R, I, M, P, PD, D, DB, or DBM
R = real
I = imaginary
M = magnitude
P = PD = phase in degrees
D = DB = decibels
DBM = decibels per 1.0e-3
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Parameter
Description
NF
SSNF
NF and SSNF both output a single-side band noise
figure as a function of the IFB points:
NF = SSNF = 10 Log(SSF)
Single side-band noise factor, SSF = {(Total Noise at
output, at OFB, originating from all frequencies) - (Load
Noise originating from OFB)} / (Input Source Noise
originating from IFB).
DSNF
DSNF outputs a double side-band noise figure as a
function of the IFB points.
DSNF = 10 Log(DSF)
Double side-band noise factor, DSF = {(Total Noise at
output, at the OFB, originating from all frequencies) (Load Noise originating from the OFB)} / (Input Source
Noise originating from the IFB and from the image of
IFB).
Output Data Files
An HBLIN analysis produces these output data files:
■
The S-parameters from the .PRINT statement are written to a .printhl# file.
■
The extracted S-parameters from the .PROBE statement are written to
a .hl# file.
Large-Signal S-parameter (HBLSP) Analysis
An HBLSP analysis provides three kinds of analyses for periodically-driven
nonlinear circuits, such as those that employ power amplifiers and filters:
■
Two-port power-dependant (large-signal) S-parameter extraction
■
Two-port small-signal S-parameter extraction
■
Two-port small-signal noise parameter calculation
Unlike small-signal S-parameters, which are based on linear analysis, powerdependent S-parameters are based on harmonic balance simulation. Its
solution accounts for nonlinear effects such as compression and variation in
power levels.
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The definition for power-dependent S-parameters is similar to that for smallsignal parameters. Power-dependent S-parameters are defined as the ratio of
reflected and incident waves by using this equation:
b=S*a ;
S[i, j]=b[i,n]/a[j,n]
when a[k,n](k!=j)=0
The incident waves, a[i, n], and reflected waves, b[i, n], are defined by using
these equations:
a[i, n] = (V[i](n*W0) + Zo[i] * I[i](n*W0)) / (2 * sqrt(Zo[i]))
b[i, n] = (V[i](n*W0) - Zo[i] * I[i](n*W0)) / (2 * sqrt(Zo[i]))
Where:
■
W0 is the fundamental frequency (tone).
■
n is a signed integer.
■
i is the port number.
■
a[i, n] is the input wave at the frequency n*W0 on the ith port.
■
b[i, n] is the reflected wave at the frequency n*W0 on the ith port.
■
V[i](n*W0) is the Fourier coefficient at the frequency n*W0 of the voltage at
port i.
■
I[i](n*W0) is the Fourier coefficient at the frequency n*W0 of the current at
port i.
■
Zo[i] is the reference impedance at port i.
An HBLSP analysis only extracts the S-parameters on the first harmonic (that
is, n=1).
Limitations
The HBLSP analysis has these known limitations:
308
■
Power-dependent S-parameter extraction is a 2-port analysis only. Multiport
power-dependent S-parameters are not currently supported.
■
The intermodulation data block (IMTDATA) in the .p2d# file is not supported.
■
The internal impedance of the P (port) Element can only be a real value.
Complex impedance values are not supported.
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Large-Signal S-parameter (HBLSP) Analysis
Input Syntax
.HBLSP NHARMS=nh <POWERUNIT=[dbm | watt]>
+ <SSPCALC=[1|0|YES|NO]> <NOISECALC=[1|0|YES|NO]>
+ <FILENAME=file_name> <DATAFORMAT=[ri | ma | db]>
+ FREQSWEEP freq_sweep POWERSWEEP power_sweep
Parameter
Description
NHARMS
Number of harmonics in the HB analysis triggered by the .HBLSP
statement.
POWERUNIT
Power unit. Default is watt.
SSPCALC
Extract small-signal S-parameters. Default is 0 (NO).
NOISECALC
Perform small-signal 2-port noise analysis. Default is 0 (NO).
FILENAME
Output data .p2d# filename. Default is the netlist name or the object
name after the -o command-line option.
DATAFORMAT
Format of the output data file. Default is ma (magnitude, angle).
FREQSWEEP
Frequency sweep specification. A sweep of type LIN, DEC, OCT,
POI, or SWEEPBLOCK. Specify the nsteps, start, and stop times
using the following syntax for each type of sweep:
■
LIN nsteps start stop
DEC nsteps start stop
■
OCT nsteps start stop
■
POI nsteps freq_values
■
SWEEPBLOCK=blockname
This keyword must appear before the POWERSWEEP keyword.
■
POWERSWEEP Power sweep specification. A sweep of type LIN, DEC, OCT,POI, or
SWEEPBLOCK. Specify the nsteps, start, and stop frequencies
using the following syntax for each type of sweep:
■
LIN nsteps start stop
DEC nsteps start stop
■
OCT nsteps start stop
■
POI nsteps power_values
■
SWEEPBLOCK=blockname
This keyword must follow the FREQSWEEP keyword.
■
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Note:
The FREQSWEEP and POWERSWEEP keywords must appear at the end of an
.HBLSP statement.
Examples
Example 1does 2-port single-tone, power-dependent S-parameter extraction,
without frequency translation:
■
Frequency sweep: The fundamental tone is swept from 0 to 1G
■
Power sweep: The power input at port 1 is swept from 6 to 10 Watts.
■
Five harmonics are required for the HB analysis. Large-signal S-parameters
are extracted on the first harmonic.
■
Five harmonics are required in the HBLSP triggered HB analysis.
■
The DC value in p1 statement is used to set DC bias, which is used to
perform small-signal analyses.
■
Small-signal S-parameters are required extracted.
■
Small-signal two-port noise analysis is required.
■
The data will be output to the ex1.p2d file.
Example 1
2-Port, Single Tone
p1 1 0 port=1 dc=1v
p2 2 0 port=2
.hblsp nharms=5 powerunit = watt
+ sspcalc=1 noisecalc=1 filename=ex1
+ freqsweep lin 5 0 1G powersweep lin 5 6 10
Example 2 generates large scale S-parameters as a function of input for a
differential equalizer.
Example 2
4-Port Network
* hblsp example
.opt post
p1 n1 0 port=1 ac=1
p2 n2 0 port=2
*** put your DUT
R1 n1 n2 10***
.hblsp nharms=5
+ freqsweep lin 4 1k 10k
+ powersweep lin 2 5 10
.end
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Output Syntax
This section describes the syntax for the HBLSP .PRINT and .PROBE
statements. These statements only support S and noise parameter outputs.
Node voltage, branch current, and all other parameters are not supported in
HBLSP .PRINT and .PROBE statements.
.PRINT and .PROBE Statements
.PRINT HBLSP Smn | Smn(TYPE) |
+ ...small signal 2-port noise
.PROBE HBLSP Smn | Smn(TYPE) |
+ ...small signal 2-port noise
S(m, n) | S(m, n)(TYPE)
params...
S(m, n) | S(m, n)(TYPE)
params...
Parameter
Description
Smn | Smn(TYPE) |
S(m,n) | S(m,n)(TYPE)
Complex 2-port parameters. Where:
■
■
■
... small signal 2-port noise
parameters ...
m = 1 or 2
n = 1 or 2
TYPE = R, I, M, P, PD, D, DB, or DBM
R = real
I = imaginary
M = magnitude
P = PD = phase in degrees
D = DB = decibels
DBM = decibels per 1.0e-3
G_AS | NF | RN | YOPT | GAMMA_OPT | NFMIN |
VN2 | ZCOR | GN | RHON | YCOR | ZOPT | IN2
For a description of these parameters, see Linear
Network Parameter Analysis in the HSPICE User
Guide: Simulation and Analysis.
Output Data Files
An HBLSP analysis produces these output data files:
■
The large-signal S-parameters from the .PRINT statement are written to
a .printls# file.
■
The small-signal S-parameters from the .PRINT statement are written to
a .printss# file.
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■
The large-signal S-parameters from the .PROBE statement are written to
a .ls# file.
■
The small-signal S-parameters from the .PROBE statement are written to
a .ss# file.
■
The extracted large- and small-signal S and noise parameters are written to
a .p2d# file.
The large- and small-signal S-parameters from the .PROBE statement are
viewable in CosmosScope.
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12
Envelope Analysis
Describes how to use envelope simulation.
These topics are covered in the following sections:
■
Envelope Simulation
■
Envelope Analysis Commands
■
Nonautonomous Form
■
Oscillator Analysis Form
■
Fast Fourier Transform Form
■
Output Syntax
Envelope Simulation
Envelope simulation combines features of time- and frequency-domain
analysis. Harmonic Balance (HB) solves for a static set of phasors for all the
circuit state variables, as shown in this equation:
N
Equation 59
v ( t ) = a0 +
∑ [ ai cos ωi t + bi sin ωi t ]
i=1
In contrast, envelope analysis finds a dynamic, time-dependent set of phasors,
as this equation shows:
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Equation 60
v ( t ) = a 0 ( t̂ ) +
N
∑ [ ai ( t̂ ) cos ωi t + bi ( t̂ ) sin ωi t ]
i=1
Thus, in envelope simulation, each signal is described by the evolving
spectrum. Envelope analysis is generally used on circuits excited by signals
with significantly different timescales. An HB simulation is performed at each
point in time of the slower-moving ( t̂ ) timescale. In this way, for example, a 2tone HB simulation can be converted into a series of related 1-tone simulations
where the transient analysis proceeds on the ( t̂ ) timescale, and 1-tone HB
simulations are performed with the higher frequency tone as the fundamental
frequency.
In HSPICE RF, any voltage or current source identified as a HB source either in
a V or I element statement, or by an .OPTION TRANFORHB command, is used
for HB simulations at each point in t̂ time. All other sources are associated with
the transient timescale. Also, the input waveforms can be represented in the
frequency domain as RF carriers modulated by an envelope by identifying a
VMRF signal source in a V or I element statement. The amplitude and phase
values of the sampled envelope are used as the input signal for HB analysis.
Some typical applications for envelope simulation are amplifier spectral
regrowth, adjacent channel power ration (ACPR), and oscillator startup and
shutdown analyses.
Envelope Analysis Commands
This section describes those commands specific to envelope analysis. These
commands are:
■
Standard envelope simulation (.ENV)
■
Oscillator simulation, both startup and shutdown (.ENVOSC)
■
Envelope Fast Fourier Transform (.ENVFFT)
Nonautonomous Form
.ENV TONES=f1<f2...fn> NHARMS=h1<h2...hn>
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+ ENV_STEP=tstep ENV_STOP=tstop
Parameter
Description
TONES
Carrier frequencies, in hertz.
NHARMS
Number of harmonics.
ENV_STEP
Envelope step size, in seconds.
ENV_STOP
Envelope stop time, in seconds.
Description
You use the .ENV command to do standard envelope simulation. The
simulation proceeds just as it does in standard transient simulation, starting at
time=0 and continuing until time=env_stop. An HB analysis is performed at
each step in time. You can use Backward-Euler (BE), trapezoidal (TRAP), or
level-2 Gear (GEAR) integration.
Recommended option settings are:
■
For BE integration, set .OPTION SIM_ORDER=1.
■
For TRAP, set .OPTION SIM_ORDER=2 (default) METHOD=TRAP (default).
■
For GEAR, set .OPTION SIM_ORDER=2 (default) METHOD=GEAR.
Example
.env tones=1e9 nharms=6 env_step=10n env_stop=1u
Oscillator Analysis Form
.ENVOSC TONE=f1 NHARMS=h1 ENV_STEP=tstep ENV_STOP=tstop
+ PROBENODE=n1,n2,vosc <FSPTS=num, min, max>
Parameter
Description
TONE
Carrier frequencies, in hertz.
NHARMS
Number of harmonics.
ENV_STEP
Envelope step size, in seconds.
ENV_STOP
Envelope stop time, in seconds.
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Parameter
Description
PROBENODE
Defines the nodes used for oscillator conditions and the initial
probe voltage value.
FSPTS
Specifies the frequency search points used in the initial smallsignal frequency search. Usage depends on oscillator type.
Description
You use the .ENVOSC command to do envelope simulation for oscillator startup
or shutdown.
Oscillator startup or shutdown analysis with this command must be helped
along by converting a bias source from a DC description to a PWL description
that either:
■
Starts at a low value that supports oscillation and ramps up to a final value
(startup simulation)
■
Starts at the DC value and ramps down to zero (shutdown simulation).
In addition to solving for the state variables at each envelope time point, the
.ENVOSC command also solves for the frequency. This command is intended to
be applied to high-Q oscillators that take a long time to reach steady-state. For
these circuits, standard transient analysis is too costly. Low-Q oscillators, such
as typical ring oscillators, are more efficiently simulated with standard transient
analysis.
Example
.envosc tone=250Meg nharms=10 env_step=20n env_stop=10u
+ probenode=v5,0,1.25
Fast Fourier Transform Form
.ENVFFT <output_var> <NP=value> <FORMAT=keyword>
+ <WINDOW=keyword> <ALFA=value>
316
Parameter
Description
output_var
Any valid output variable.
NP
The number of points to use in the FFT analysis. NP must be a
power of 2. If not a power of 2, then it is automatically adjusted to
the closest higher number that is a power of 2. The default is 1024.
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Parameter
Description
FORMAT
Specifies the output format:
NORM= normalized magnitude
UNORM=unnormalized magnitude (default)
WINDOW
Specifies the window type to use:
RECT=simple rectangular truncation window (default)
BART=Bartlett (triangular) window
HANN=Hanning window
HAMM=Hamming window
BLACK=Blackman window
HARRIS=Blackman-Harris window
GAUSS=Gaussian window
KAISER=Kaiser-Bessel window
ALFA
Controls the highest side-lobe level and bandwidth for GAUSS and
KAISER windows. The default is 3.0.
Description
You use the .ENVFFT command to perform Fast fourier Transform (FFT) on
envelope output. This command is similar to the .FFT command. The only
difference is that transformation is performed on real data with the .FFT
command, and with the .ENVFFT command, the data being transformed is
complex. You usually want to do this for a specific harmonic of a voltage,
current, or power signal.
Example
.envfft v(out)[1]
Output Syntax
The results from envelope simulation can be made available through the
.PRINT, .PROBE, and .MEASURE commands. This section describes the basic
syntax you can use for this purpose.
.PRINT or .PROBE
You can print or probe envelope simulation results by using the following
commands:
.PRINT ENV ov1 <ov2... >
.PROBE ENV ov1 <ov2... >
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Where ov1... are the output variables to print or probe.
.MEASURE
In HSPICE RF, the independent variable for envelope simulation is the first
tone. Otherwise and except for the analysis type, the .MEASURE statement
syntax is the same as the syntax for HB; for example,
.MEASURE ENV result ...
Envelope Output Data File Format
The results of envelope simulations are written to *.ev# data files by the
.PROBE statement. The format of an *.ev# data file is equivalent to an *.hb#
data file with the addition of one fundamental parameter sweep that represents
the slowly-varying time-envelope variation t̂ of the Fourier coefficients and
frequencies. You can recognize this swept parameter” in the *.ev# file by the
keyword env_time.
Each row in the tabulated data of an *.ev# file includes values for identifying
frequency information, the complex data for the output variables, and
information on the envelope time sweep. For example, the header for a data file
dump for output variables v(in) and v(out) that follow a 2-tone envelope
analysis, have entries for:
hertz
v(in)
v(out)
n0
f0
n1
f1
sweep
env_time
$&%#
Which result in data blocks with floating point values following:
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env_time[0]
f[0] a[0]{v(in)} b[0] {v(in)} a[0] {v(out)} b[0] {v(out)} n0
f0 n1 f1
f[1] a[1]{v(in)} b[1] {v(in)} a[1] {v(out)} b[1] {v(out)} n0
f0 n1 f1
...
f[N] a[N]{v(in)} b[N] {v(in)} a[N] {v(out)} b[N] {v(out)} n0
f0 n1 f1
env_time[1]
f[0] a[0]{v(in)} b[0] {v(in)} a[0] {v(out)} b[0] {v(out)} n0
f0 n1 f1
f[1] a[1]{v(in)} b[1] {v(in)} a[1] {v(out)} b[1] {v(out)} n0
f0 n1 f1
...
f[N] a[N]{v(in)} b[N] {v(in)} a[N] {v(out)} b[N] {v(out)} n0
f0 n1 f1
...
env_time[M-1]
f[0] a[0]{v(in)} b[0] {v(in)} a[0] {v(out)} b[0] {v(out)} n0
f0 n1 f1
f[1] a[1]{v(in)} b[1] {v(in)} a[1] {v(out)} b[1] {v(out)} n0
f0 n1 f1
...
f[N] a[N]{v(in)} b[N] {v(in)} a[N] {v(out)} b[N] {v(out)} n0
f0 n1 f1
Where there are M data blocks corresponding to M envelope time points, with
each block containing N+1 rows for the frequency data. The units for the
env_time sweep are seconds.
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Post-Layout Analysis
13
Describes the post-layout analysis flow, including post-layout back-annotation,
DSPF and SPEF files, linear acceleration, check statements, and power
analysis.
These topics are covered in the following sections:
■
Post-Layout Back-Annotation
■
Linear Acceleration Control Options Summary
Post-Layout Back-Annotation
A traditional, straightforward, “brute-force” flow runs an RC extraction tool that
produces a detailed standard parasitic format (DSPF) file. DSPF is the
standard format for transferring RC parasitic information. This traditional flow
then feeds this DSPF file into the circuit simulation tool for post-layout
simulation.
A key problem is that the DSPF file is flat. Accurately simulating a complete
design, such as an SRAM or an on-chip cache, is a waste of workstation
memory, disc space usage, and simulation runtime. Because this DSPF file is
flat, control and analysis are limited.
■
How do you set different options for different blocks for better trade-off
between speed and accuracy?
■
How do you perform a power analysis on a flat netlist to check the power
consumption?
■
This traditional flow flattens all nodes after extraction so it is more difficult to
compare the delay before and after extraction.
■
This traditional flow can also stress the limits of an extraction tool so
reliability also becomes an issue.
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HSPICE RF provides a flow that solves all of these problems.
■
Star-RCXT generates a hierarchical Layout Versus Schematic (LVS) ideal
netlist, and flat information about RC parasitics in a DSPF or (standard
parasitic exchange format (SPEF) file.
■
HSPICE RF uses the hybrid flat-hierarchical approach to back-annotate the
RC parasitics, from the DSPF or SPEF file, into the hierarchical LVS ideal
netlist.
Using the hierarchical LVS ideal netlist cuts simulation runtime and CPU
memory usage. Because HSPICE RF uses the hierarchical LVS ideal netlist as
the top-level netlist, you can fully control the netlist. For example:
■
You can set different modes to different blocks for better accuracy and speed
trade-off.
■
You can run power analysis, based on the hierarchical LVS ideal netlist, to
determine the power consumption of each block. If you use the hierarchical
LVS ideal netlist, you can reuse all post-processing statements from the prelayout simulation for the post-layout simulation. This saves time, and the
capacity of the verification tool is not stressed so reliability is higher.
HSPICE RF supports only the XREF:COMPLETE flow and the XREF:NO flow
from Star-RCXT. Refer to the Star-RCXT User Guide for more information
about the XREF flow.
To generate a hierarchical LVS ideal netlist with Star-RCXT, include the
following options in the Star-RCXT command file.
*** for XREF:NO flow ***
NETLIST_IDEAL_SPICE_FILE: ideal_spice_netlist.sp
NETLIST_IDEAL_SPICE_TYPE: layout
NETLIST_IDEAL_SPICE_HIER:YES
*** for XREF:COMPLETE flow ***
NETLIST_IDEAL_SPICE_FILE: ideal_spice_netlist.sp
NETLIST_IDEAL_SPICE_TYPE: schematic
NETLIST_IDEAL_SPICE_HIER:YES
Note:
Before version 2002.2, Star-RCXT used
NETLIST_IDEAL_SPICE_SKIP_CELLS to generate the hierarchical ideal
SPICE netlist. HSPICE RF can still simulate post-layout designs using the
brute-force flow, but the post-layout flow is preferable in HSPICE RF.
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HSPICE RF supports these post-layout flows to address your post-layout
simulation needs.
■
Standard Post-Layout Flow
■
Selective Post-Layout Flow
■
Additional Post-Layout Options
Standard Post-Layout Flow
Use this flow mainly for analog or mixed signal design, and high-coverage
verification runs when you need to back-annotate RC parasitics into the
hierarchical LVS ideal netlist. In this flow, HSPICE RF expands all nets from the
DSPF or SPEF file. To expand only selected nets, use see Selective PostLayout Flow on page 326.
Extraction Tool
Ideal Netlist
DSPF
SPEF
HSPICE RF
Back-annotation
.html
.lis
Figure 26
Standard Post-Layout Flow
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Standard Post-Layout Flow Control Options
The standard post-layout flow options are SIM_DSPF and SIM_SPEF. Include
one of these options in your netlist. For example,
.OPTION SIM_DSPF=“[scope] dspf_filename”
.OPTION SIM_SPEF=“spec_filename”
In the SIM_DSPF syntax, scope can be a subcircuit definition or an instance. If
you do not specify scope, it defaults to the top-level definition. HSPICE RF
requires both a DSPF file and an ideal netlist. Only flat DSPF files are
supported; hierarchy statements, such as .SUBCKT and .x1, are ignored.
Very large circuits generate very large DSPF files; this is when using either the
SIM_DSPF or the SIM_DSPF_ACTIVE option can really improve performance.
You can specify a DSPF file in the SIM_SPEF option, or a SPEF file in the
SIM_DSPF option. The scope function is not supported in the SPEF format.
For descriptions and usage examples, see .OPTION SIM_DSPF and .OPTION
SIM_SPEF in the HSPICE Reference Manual: Commands and Control
Options.
Example
$ models
.MODEL p pmos
.MODEL n nmos
.INCLUDE add4.dspf
.OPTION SIM_DSPF=“add4.dspf”
.VEC “dspf_adder.vec”
.TRAN 1n 5u
vdd vdd 0 3.3
.OPTION POST
.END
SIM_DSPF With SIM_LA Option
The SIM_DSPF option accelerates the simulation by more than 100%. By using
the SIM_LA option at the same time, you can further reduce the total CPU time:
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$ models
.MODEL p pmos
.MODEL n nmos
.INCLUDE add4.dspf
.OPTION SIM_DSPF="add4.dspf"
.OPTION SIM_LA=PACT
.VEC "dspf_adder.vec"
.TRAN 1n 5u
vdd vdd 0 3.3
.OPTION POST
.END
To expand only active nodes, such as those that move, include the
SIM_DSPF_ACTIVE option in your netlist. For example:
.OPTION SIM_DSPF_ACTIVE=“active_net_filename”
This option is most effective when used with a large design—for example, over
5K transistors. Smaller designs lose some of the performance gain, due to
internal overhead processing.
For syntax and description of SIM_DSPF_ACTIVE option, see .OPTION
SIM_DSPF_ACTIVE in the HSPICE Reference Manual: Commands and
Control Options.
When you have included the appropriate control option, run HSPICE RF, using
the ideal netlist.
The structure of a DSPF file is:
*|DSPF 1.0
*|DESIGN “demo”
*|Date “October 6, 1998”
...
.SUBCKT < name > < pins >
* Net Section
C1 ...
R1 ...
...
* Instance Section
...
.ENDS
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Selective Post-Layout Flow
Extraction Tool
DSPF
SPEF
Ideal Netlist
HSPICE RF
Active Nodes
Back-annotation
HSPICE RF
.html
.lis
Figure 27
Selective Post-Layout Flow
You can use the selective post-layout flow to simulate a post-layout design for a
memory or digital circuit, and for a corner-point verification run. Instead of backannotating all RC parasitics into the ideal netlist, the selective post-layout flow
automatically detects and back-annotates only active parasitics, into the
hierarchical LVS ideal netlist. For a high-latency design, the selective postlayout flow is an order of magnitude faster than the standard post-layout flow.
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Note:
The selective post-layout flow applies only to RF transient analyses and
cannot be used with other analyses such as DC, AC, or HB.
Selective Post-Layout Flow Control Options
To invoke the selective post-layout flow, include one of the options listed in
Table 23 in your netlist.
Table 23
Selective Post-Layout Flow Options
Syntax
Description
SIM_DSPF_ACTIVE
-orSIM_SPEF_ACTIVE
HSPICE RF performs a preliminary verification run to
determine the activity of the nodes and generates two ASCII
files: active_node.rc and active_node.rcxt. These files save
all active node information in both Star-RC format and StarRCXT format.
By default, a node is considered active if the voltage varies
by more than 0.1V. To change this value, use the
SIM_DSPF_VTOL or SIM_SPEF_VTOL option.
For descriptions and usage examples, see OPTION
SIM_DSPF_ACTIVE and .OPTION SIM_SPEF_ACTIVE in
the HSPICE Reference Manual: Commands and Control
Options.
SIM_DSPF_VTOL
-orSIM_SPEF_VTOL
HSPICE RF performs a second simulation run by using the
active_node file, the DSPF or SPEF file, and the hierarchical
LVS ideal netlist to back-annotate only active portions of the
circuit. If a net is latent, then HSPICE RF does not expand
the net. This saves simulation runtime and memory.
■
value is the tolerance of the voltage change.
scopen can be a subcircuit definition (which has an @
prefix), or a subcircuit instance.
By default, HSPICE RF performs only one iteration of the
second simulation run. Use the SIM_DSPF_MAX_ITER or
SIM_SPEF_MAX_ITER option to change it.
■
For descriptions and usage examples, see .OPTION
SIM_DSPF_VTOL and .OPTION SIM_SPEF_VTOL in the
HSPICE Reference Manual: Commands and Control
Options.
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Table 23
Selective Post-Layout Flow Options (Continued)
Syntax
Description
SIM_DSPF_MAX_ITER value is the maximum number of iterations for the second
-orsimulation run.
SIM_SPEF_MAX_ITER Some of the latent nets might turn active after the first
iteration of the second run. In this case:
■
Resimulate the netlist to ensure the accuracy of the postlayout simulation.
■
Use SIM_DSPF_MAX_ITER or SIM_SPEF_MAX_ITER
to set the maximum number of iterations for the second
run. If the active_node remains the same after the second
simulation run, HSPICE RF ignores these options.
For descriptions and usage examples, see .OPTION
SIM_DSPF_MAX_ITER and .OPTION
SIM_SPEF_MAX_ITER in the HSPICE Reference Manual:
Commands and Control Options.
Additional Post-Layout Options
Other post-layout options are listed in Table 24.
Table 24
Additional Post-Layout Options
Syntax
Description
SIM_DSPF_RAIL
-orSIM_SPEF_RAIL
By default, HSPICE RF does not back-annotate parasitics
of the power-net. To back-annotate power-net parasitics,
include one of these options in the netlist.
Default=OFF. ON expands nets in a power rail as it expands
all nets.
SIM_DSPF_SCALER
SIM_SPEF_SCALER
-orSIM_DSPF_SCALEC
SIM_SPEF_SCALEC
328
Scales the resistance or capacitance values.
■
■
scaleR is the scale factor for resistance
scaleC is the scale factor for capacitance.
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Table 24
Additional Post-Layout Options (Continued)
Syntax
Description
SIM_DSPF_LUMPCAPS If HSPICE RF cannot back-annotate an instance in a net
-orbecause one or more instances are missing in the
SIM_SPEF_LUMPCAPS hierarchical LVS ideal netlist, then by default HSPICE RF
does not evaluate the net. Instead of ignoring all parasitic
information for this net, HSPICE RF includes these options
to connect a lumped capacitor with a value equal to the net
capacitance to this net.
Default = ON adds lumped capacitance; ignores other net
contents.
SIM_DSPF_INSERROR HSPICE RF supports options to skip the unmatched
-orinstance, and continue the evaluation of the next instance.
SIM_SPEF_INSERROR The default is OFF. ON skips unmatched instances and
continues the evaluation.
SIM_SPEF_PARVALUE
This option affects only values in a SPEF file that have
triplet format: float:float:float, which this option interprets as
best:average:worst.
In such cases:
■
■
■
If SIM_SPEF_PARVALUE=1, HSPICE RF uses best.
If SIM_SPEF_PARVALUE=2 (default), HSPICE RF uses
average.
If SIM_SPEF_PARVALUE=3, HSPICE RF uses worst.
Unsupported SPEF Options
HSPICE RF does not yet support the following IEEE-481 SPEF options:
■
Hierarchical SPEF definition (multiple SPEF files connected with a
hierarchical definition):
■
*DEFINE and *PDEFINE
■
*R_NET and *R_PNET definition
■
*D_PNET definition.
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Selective Extraction Flow
Use the selective extraction flow if disk space is limited. Especially use this
option when simulating a full-chip post-layout design, where block latency is
high. HSPICE RF feedbacks the active net information to Star-RCXT to extract
only the active parasitic.
The major advantage of this flow is a smaller DSPF or SPEF file, which saves
disk space.
Star-RCXT
DSPF/SPEF
Post-Layout Flow
Ideal Netlist
OR
HSPICE RF
Active Nodes
Star-RCXT
DSPF/SPEF
Post-Layout Flow
Figure 28
330
Selective Extraction Flow
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Note:
HSPICE RF generates an active node file in both Star-RC and Star-RCXT
format. It then expands the active node file to the Star-RCXT command file
to extract only active parasitics.
Overview of DSPF Files
In general, an SPF (Standard Parasitic Format) file describes interconnect
delay and loading, due to parasitic resistance and capacitance. DSPF (Detailed
Standard Parasitic Format) is a specific type of SPF file that describes the
actual parasitic resistance and capacitance components of a net. DSPF is a
standard output format commonly used in many parasitic extraction tools,
including Star-RCXT. The HSPICE RF circuit simulator can read DSPF files.
DSPF File Structure
The DSPF standard is published by Open Verilog International (OVI). For
information about how to obtain the complete DSPF specification, or any other
documents from OVI, see:
http://www.ovi.org/document.html
The OVI DSPF specification requires the following file structure in a DSPF file.
Parameters in {braces} are optional:
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DSPF_file : :=
*|DSPF{version}
{*|DESIGN design_name}
{*|DATE date}
{*|VENDOR vendor}
{*|PROGRAM program_name}
{*|VERSION program_version}
{*|DIVIDER divider}
{*|DELIMITER delimiter}
.SUBCKT
*|GROUND_NET
{path divider} net_name
*|NET {path divider} net_name ||
{path divider} instance_name ||
pin_name
net_capacitance
*|P (pin_name pin_type
pinCap
{resistance {unit} {O}
capacitance {unit} {F}}
{x_coordinate y_coordinate})
||
*|I {path divider} instance_name
delimiter pin_name
{path divider} instance_name
pin_name pin_type
pinCap
{resistance {unit} {O}
capacitance {unit}{F}}
{x_coordinate y_coordinate}
*|S ({path divider} net_name ||
{path divider} instance_name
delimiter pin_name ||
pin_name
instance_number
{x_coordinate y_coordinate})
capacitor_statements
resistor_statements
subcircuit_call_statements
.ENDS
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{.END}
Table 25
DSPF Parameters
Parameter
Definition
*|DSPF
Specifies that the file is in DSPF format.
{version}
Version number of the DSPF specification (optional).
*|
Words that start with *| are keywords.
||
Or (use the option either preceding or following ||). For
example, *|P || *I means you can use either the *|P option or
the *|I option.
design_name
Name of your circuit design (optional).
date
Date and time when a parasitic extraction tool (such as StarRCXT) generated the DSPF file (optional).
vendor
Name of the vendor (such as Synopsys) whose tools you
used to generate the DSPF file (optional).
program_name
Name of the program (such as Star-RCXT) that generated
the DSPF file (optional).
program_version
Version number of the program that generated the DSPF file
(optional).
divider
Character that divides levels of hierarchy in a circuit path
(optional). If you do not define this parameter, the default
hierarchy divider is a slash (/). For example, X1/X2 indicates
that X2 is a subcircuit of the X1 circuit.
delimiter
Character used to separate the name of an instance and a
pin in a concatenated instance pin name, or a net name and
a sub-node number in a concatenated sub-node name. If you
do not define this parameter, the default delimiter is a colon
(:).
path
Hierarchical path to a net, instance, or pin, within a circuit.
net_name
Name of a net in a circuit or subcircuit.
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Table 25
DSPF Parameters (Continued)
Parameter
Definition
instance_name
Name of an instance of a subcircuit.
pin_name
Name of a pin on an instance of a subcircuit.
pinCap
Capacitance of a pin.
pin_type
■
■
■
■
■
■
resistance
I (input)
O (output)
B (bidirectional)
X (don’t care)
S (switch)
J (jumper)
Resistance on a pin in ohms for input (I), output (O), or
bidirectional (B) pins. You can use resistance-capacitance
(RC) pairs to model pin characteristics by using a higherorder equivalent RC ladder circuit than a single capacitor
model. For example: C0 {R1 C1 R2 C2...}. Attaching RC pairs
increases the order of the equivalent circuit from the first (C0)
order. For X, S, and J pin types, simulation ignores this
generalized capacitance value, but you should insert a 0
value as a place-holder for format integrity.
The resistance value can be a real number or an exponent
(optionally followed by a real number). You can enter an O
(ohms) after the value.
capacitance
Capacitance on a pin in farads for input (I), output (O), or
bidirectional (B) pins. Use as part of a resistancecapacitance (RC) pair. Optionally enter an F (farads) after the
value.
unit
■
■
■
■
■
■
334
K (kilo)
M (milli)
U (micro)
N (nano)
P (pico)
F (femto)
x_coordinate
Location of a pin relative to the x (horizontal) axis.
y_coordinate
Location of a pin relative to the y (vertical) axis.
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Table 25
DSPF Parameters (Continued)
Parameter
Definition
capacitor_ statements
SPICE-type statements that define capacitors in the
subcircuit.
resistor_ statements
SPICE-type statements that define resistors in the subcircuit.
subcircuit_call_
statements
Statements that call the subcircuit from higher-level circuits.
.END
Marks the end of the file (optional).
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DSPF File Example
*|DSPF 1.0
*|DESIGN "my_circuit"
*|DATE June 15, 2002 14:12:43
*|VENDOR "Synopsys"
*|PROGRAM "Star-RC"
*|VERSION "Star-RCXT 2002.2"
*|DIVIDER /
*|DELIMITER :
.SUBCKT BUFFER OUT IN
* Description of Nets
*GROUND_NET VSS
*|NET IN 1.221451PF
*|P(IN 1 0.0 0 10)
*|I(DF1:A DF1 A I 0.0PF 10.0 10.0)
*|I(DF1:B DF1 B I 0.0PF 10 0 20.0)
*|S(IN:1 5.0 10.0)(IN:2 5.0 20.0)
C1 IN VSS 0.117763PF
C2 IN:1 VSS 0.276325PF
C3 IN:2 VSS 0.286325PF
C4 DF1:A VSS 0.270519PF
C5 DF1:B VSS 0.270519PF
R20 IN N:1 1.70333E00
R21 IN:1 DF1:A 1.29167E-01
R22 IN:1 IN:2 1.29167E-01
R23 IN:2 DF1:B 1.70333E-01
*|NET BF 0.287069PF
*|I(DF1:C DF1 C O 0.0PF 12.0 15.0)
*|I(INV1:IN INV1 IN I 0.0PF 30.0 15.0)
C6 DF1:C VSS 0.208719PF
C7 INV1:IN VSS 0.783500PF
R24 DF1:C INV1:IN 1.80833E-01
*|NET OUT 0.148478PF
*|S(OUT:1 45.0 15.0)
*|P(OUT O 0.0PF 50.0 5.0)
*|I(INV1:OUT INV1 OUT O 0.0PF 40.0 15.0)
C8 INV1:OUT VSS 0.147069PF
C9 OUT:1 VSS 0.632813PF
C10 OUT VSS 0.776250PF
R25 INV1:OUT OUT:1 3.11000E00
R26 OUT:1 OUT 3.03333E00
* Description of Instances
XDF1 DF1:A DF1:B DF1:C DFF
XINV1 INV1:IN INV1:OUT INV
.ENDS
.END
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Overview of SPEF Files
The Standard Parasitics Exchange Format (SPEF) file structure is described in
IEEE standard IEEE-1481. For information about how to obtain the complete
SPEC (IEEE-1481) specification, or any other documents from IEEE, see:
http://www.ieee.org/products/onlinepubs/stand/standards.html
SPEF File Structure
The IEEE-1481 specification requires the following file structure in a SPEF file.
Parameters in [brackets] are optional:
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SPEF_file : :=
*SPEF version
*DESIGN design_name
*DATE date
*VENDOR vendor
*PROGRAM program_name
*VERSION program_version
*DESIGN_FLOW flow_type {flow_type}
*DIVIDER divider
*DELIMITER delimiter
*BUS_DELIMITER bus_prefix bus_suffix
*T_UNIT time_unit NS|PS
*C_UNIT capacitance_unit FF|PF
*R_UNIT resistance_unit OHM|KOHM
*L_UNIT inductance_unit HENRY|MH|UH
[*NAME_MAP name_index name_id|bit|path|name|physical_ref]
[*POWER_NETS logical_power_net physical_power_net ...]
[*GROUND_NETS ground_net ...]
[*PORTS logical_port I|B|O
*C coordinate ...
*L par_value
*S rising_slew falling_slew [low_threshold high_threshold]
*D cell_type]
[*PHYSICAL_PORTS [physical_instance delimiter]
physical_port I|B|O
*C coordinate ...
*L par_value
*S rising_slew falling_slew [low_threshold high_threshold]
*D cell_type]
[*DEFINE logical_instance design_name |
*PDEFINE physical_instance design_name]
*D_NET net_path total_capacitance
[*V routing_confidence]
[*CONN
*P [logical_instance delimiter] logical_port|physical_port
I|B|O
*C coordinate ...
*L par_value
*S rising_slew falling_slew
[low_threshold high_threshold]
*D cell_type
|
*I [physical_instance delimiter] logical_pin|physical_node
I|B|O
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*C coordinate ...
*L par_value
*S rising_slew falling_slew
[low_threshold high_threshold]
*D cell_type
*N net_name delimiter net_number coordinate
[*CAP cap_id node1 [node2] capacitance]
[*RES res_id node1 node2 resistance]
[*INDUC induc_id node1 node2 inductance]
*END
Table 26
SPEF Parameters
Parameter
Definition
*SPEF
Specifies that the file is in SPEF format.
{version}
Version number of the SPEF specification, such as “IEEE 14811998”.
*
Words that start with an asterisk (*) are keywords.
|
Or. For example, NS|PS means choose either nanoseconds or
picoseconds as the time units.
design_name
Name of your circuit design.
date
Date and time when a parasitic extraction tool (such as Star-RCXT)
generated the SPEF file.
vendor
Name of the vendor (such as Synopsys) whose tools you used to
generate the SPEF file (optional).
program_name
Name of the program (such as Star-RCXT) that generated the
SPEF file.
program_version
Version number of the program that generated the SPEF file.
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Table 26
SPEF Parameters (Continued)
Parameter
Definition
flow_type
One or more of the following flow types:
■
■
■
■
■
■
■
340
EXTERNAL_LOADS: The SPEF file defines all external loads (if
any). If you do not specify this flow type, then some or all external
loads are not defined in this SPEF file. If HSPICE RF cannot find
external load data outside the SPEF file, it reports an error.
EXTERNAL_SLEWS: The SPEF file defines all external slews (if
any). If you do not specify this flow type, then some or all external
slews are not defined in this SPEF file. If HSPICE RF cannot find
external slew data outside the SPEF file, it reports an error.
FULL_CONNECTIVITY: A SPEF file defines all net connectivity.
If you do not specify this flow type, then some or all net
connectivity is not defined in this SPEF file. If HSPICE RF cannot
find connectivity data outside the SPEF file, it issues an error.
This flow does not look for presence or absence of power and
ground nets, or any other nets that do not correspond to the
logical netlist. If a SPEC file includes FULL_CONNECTIVITY
and MISSING_NETS, HSPICE RF reports an error.
MISSING_NETS: If any logical nets are not defined in the netlist,
HSPICE RF merges missing parasitic data from another source.
If it does not find another source, HSPICE RF rereads the netlist
and estimates the missing parasitics. This flow does not look for
presence or absence of power and ground nets, or any other
nets that do not correspond to the logical netlist. If you use
FULL_CONNECTIVITY and MISSING_NETS in the same SPEF
file, HSPICE RF reports an error.
NETLIST_TYPE_VERILOG, NETLIST_TYPE_VHDL87,
NETLIST_TYPE_VHDL93, or NETLIST_TYPE_EDIF: Specifies
the type of naming conventions used in the SPEF file. If you
specify more than one format in one SPEF file, HSPICE RF
reports an error.
ROUTING_CONFIDENCE positive_integer: Specifies a default
routing confidence value for all nets in the SPEF file.
ROUTING_CONFIDENCE_ENTRY positive_integer
character_string: Specifies one or more characters that
represent additional routing confidence values, which you can
assign to nets in the SPEF file.
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Table 26
SPEF Parameters (Continued)
Parameter
Definition
flow_type
(continued)
■
■
■
divider
NAME_SCOPE LOCAL|FLAT: Specifies whether paths in the
SPEF file are LOCAL (relative to the current SPEF file) or FLAT
(relative to the top level of your circuit design).
SLEW_THRESHOLDS low high: Specifies low and high default
input slew thresholds for your circuit design as a percentage of
the voltage level for the input pin.
PIN_CAP NONE|INPUT_OUTPUT|INPUT_ONLY: Specifies the
type of pin capacitance to include when calculating the total
capacitance for all nets in the SPEF file, either no capacitance,
all input and output capacitances, or only input capacitances.
Character used to divide levels of hierarchy in a circuit path name.
Must be one of the following characters: . / : |
For example, X1/X2 means that X2 is a subcircuit of the X1 circuit.
delimiter
Character used to separate the name of an instance and a pin in a
concatenated instance pin name. Must be one of these characters:
./:|
bus_prefix
bus_suffix
Delimiter characters that precede and follow a bus bit or an arrayed
instance number. If these characters are not matching pairs,
HSPICE RF reports an error. Valid bus delimiter prefix and suffix
character pairs are brackets “[ ]”, braces “{ }”, parentheses “( )”, or
angle brackets “< >”>
time_unit
A positive number. For example, 10 PS means use time units of 10
picoseconds. 5 NS means use time units of 5 nanoseconds.
capacitance_unit
A positive number. For example, 10 PF means capacitance units of
10 picofarads. 5 FF means use capacitance units of 5
femtoseconds.
resistance_unit
Positive number. For example, 10 OHM sets resistance units to 10
ohms. 5 KOHM sets resistance units to 5 kilo ohms.
inductance_unit
A positive number. For example, 10 HENRY means use inductance
units of 10 henries. 5 MH means use inductance units of 5
millihenries. 2 UH means use inductance units of 2 micro-henries.
name_index
Name used throughout a SPEF file. To reduce file space, you can
map other names to this name.
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Table 26
SPEF Parameters (Continued)
Parameter
Definition
name_id|bit|path|name|
physical_ref
A name identifier, bit, path, name, or physical reference to map to
the name_index.
logical_power_net
Logical path (or logical path index) to a power net.
physical_power_net
Physical path (or physical path index) to a power net. You can
specify multiple logical_power_net physical_power_net pairs.
ground_net
Name of a net to use as a ground net. You can specify multiple
ground net names.
logical_port
Logical name of an input, output, or bidirectional port.
coordinate
Geometric location of a logical or physical port.
par_value
Either a single float value, or a triplet in float:float:float form.
rising_slew
Rising slew of the waveform for the port. T_UNIT defines the time
unit for the waveform.
falling_slew
Rising slew of the waveform for the port. T_UNIT defines the time
unit for the waveform.
low_threshold
Low voltage threshold as a percentage of the port’s input voltage.
Can bed one float value or a triplet in float:float:float form.
high_threshold
High voltage threshold as a percentage of the input voltage for the
port. Either a single float value or a triplet in float:float:float form.
cell_type
Type of cell that drives the port. If you do not know the cell type, use
the reserved word UNKNOWN_DRIVER as the cell type.
physical_port
Physical name of an input, output, or bidirectional port.
logical_instance
Logical name of a subcircuit in your design_name circuit design.
You can specify more than one logical_instance. Whenever you
specify a logical instance name, you must set NAME_SCOPE to
FLAT. If you connect a logical net to a physical port, HSPICE RF
reports an error.
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Table 26
SPEF Parameters (Continued)
Parameter
Definition
physical_instance
Physical name of a subcircuit in your design_name circuit design.
You can specify more than one physical_instance. Whenever you
specify a physical instance name, you must set NAME_SCOPE to
FLAT. If you connect a physical net to a logical port, HSPICE RF
reports an error.
routing_confidence
One of the following positive integers:
■
■
■
■
■
■
■
■
■
■
10: Statistical wire load model.
20: Physical wire load model.
30: Physical partitions with locations, no cell placement.
40: Estimated cell placement with Steiner tree-based route.
50: Estimated cell placement with global route.
60: Final cell placement with Steiner route.
70: Final cell placement with global route.
80: Final cell placement, final route, 2d extraction.
90: Final cell placement, final route, 2.5d extraction.
100: Final cell placement, final route, 3d extraction.
logical_pin
Logical name of a pin.
physical_node
Physical name of a node.
net_name
Name of a net in a circuit or subcircuit.
cap_id
Unique identifier for capacitance between two specific nodes.
res_id
Unique identifier for resistance between two specific nodes.
induc_id
Unique identifier for inductance between two specific nodes.
node1
First of two nodes, between which you are specifying a capacitance,
resistance, or inductance value.
node2
Second of two nodes, between which you are specifying a
capacitance, resistance, or inductance value. For a capacitance
value, if you do not specify a second node name, HSPICE RF
assumes that the second node is ground.
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Table 26
SPEF Parameters (Continued)
Parameter
Definition
capacitance
Specifies the capacitance value assigned to a cap_id identifier.
capacitance_unit defines the units of capacitance. For example, if
you set capacitance to 5 and capacitance_unit to 10 PF, then the
actual capacitance value is 50 picoFarads.
resistance
Specifies the resistance value assigned to a res_id identifier.
resistance_unit defines the units of resistance. For example, if you
set resistance to 5 and resistance_unit to 5 KOHM, then the actual
resistance value is 25 kilo ohms.
inductance
Specifies the resistance value assigned to an induc_id identifier.
inductance_unit defines the units of inductance. For example, if you
set inductance to 6 and inductance_unit to 2 UH, then the actual
inductance value is 12 microhenries.
SPEF File Example
*SPEF "IEEE 1481-1998"
*DESIGN
"My_design"
*DATE
"11:26:34 Friday June 28, 2002"
*VENDOR
"Synopsys, Inc."
*PROGRAM
"Star-RCXT"
*VERSION
"2002.2."
*DESIGN_FLOW
"EXTERNAL_LOADS" "EXTERNAL_SLEWS" "MISSING_NETS"
*DIVIDER
/
*DELIMITER
:
*BUS_DELIMITER
[ ]
*T_UNIT
1 NS
*C_UNIT
1 PF
*R_UNIT
1 OHM
*L_UNIT
1 HENRY
*POWER_NETS
VDD
*GND_NETS
VSS
*PORTS
CONTROL O *L 30 *S 0 0
FARLOAD O *L 30 *S 0 0
INVX1FNTC_IN I *L 30 *S 5 5
NEARLOAD O *L 30 *S 0 0
TREE O *L 30 *S 0 0
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If you use triplet format, the above section would look like this:
*PORTS
CONTROL O *L 30:30:30 *S 0:0:0 0:0:0
FARLOAD O *L 30:30:30 *S 0:0:0 0:0:0
INVX1FNTC_IN I *L 30:30:30 *S 5:5:5 5:5:5
NEARLOAD O *L 30:30:30 *S 0:0:0 0:0:0
TREE O *L 30:30:30 *S 0:0:0 0:0:0
This triplet formatting principle applies to the rest of this example.
*D_NET INVX1FNTC_IN 0.033
*CONN
*P INVX1FNTC_IN I
*I FL_1281:A *L 0.033
*END
*D_NET INVX1FNTC 2.033341
*CONN
*I FL_1281:X
*I I1184:A I
*I FL_1000:A
*I NL_1000:A
*I TR_1000:A
O *L 0.0
*L 0.343
I *L 0.343
I *L 0.343
I *L 0.343
*CAP
216 FL_1000:A 0.346393
217 I1184:A 0.344053
218 INVX1FNTC_IN 0
219 INVX1FNTC_IN:10 0.154198
220 INVX1FNTC_IN:11 0.117827
221 INVX1FNTC_IN:12 0.463063
222 INVX1FNTC_IN:13 0.0384381
223 INVX1FNTC_IN:14 0.00246845
224 INVX1FNTC_IN:15 0.00350198
225 INVX1FNTC_IN:16 0.00226712
226 INVX1FNTC_IN:17 0.0426184
227 INVX1FNTC_IN:18 0.0209701
228 INVX1FNTC_IN:2 0.0699292
229 INVX1FNTC_IN:20 0.019987
230 INVX1FNTC_IN:21 0.0110279
231 INVX1FNTC_IN:24 0.0192603
232 INVX1FNTC_IN:25 0.0141824
233 INVX1FNTC_IN:3 0.0520437
234 INVX1FNTC_IN:4 0.0527105
235 INVX1FNTC_IN:5 0.1184749
236 INVX1FNTC_IN:6 0.0468458
237 INVX1FNTC_IN:7 0.0391578
238 INVX1FNTC_IN:8 0.0113856
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Chapter 13: Post-Layout Analysis
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239 INVX1FNTC_IN:9 0.0142528
240 NL_1000:A 0.344804
241 TR_000:A 0.34506
*RES
152 INVX1FNTC_IN INVX1FNTC_IN:18 8.39117
153 INVX1FNTC_IN INVX1FNTC_IN:5 25.1397
154 INVX1FNTC_IN:11 INVX1FNTC_IN:20 4.59517
155 INVX1FNTC_IN:12 INVX1FNTC_IN:13 3.688
156 INVX1FNTC_IN:13 INVX1FNTC_IN:17 25.102
157 INVX1FNTC_IN:14 INVX1FNTC_IN:16 0.0856444
158 INVX1FNTC_IN:14 NL_1000:A 0.804
159 INVX1FNTC_IN:15 INVX1FNTC_IN:16 1.73764
160 INVX1FNTC_IN:15 INVX1FNTC_IN:24 0.307175
161 INVX1FNTC_IN:17 INVX1FNTC_IN:25 5.65517
162 INVX1FNTC_IN:18 FL_1000:A 1/36317
163 INVX1FNTC_IN:2 INVX1FNTC_IN:4 6.95371
164 INVX1FNTC_IN:2 INVX1FNTC_IN:5 50.9942
165 INVX1FNTC_IN: INVX1FNTC_IN:21 4.71035
166 INVX1FNTC_IN: I1184:A 0.403175
167 INVX1FNTC_IN: TR_1000:A 0.923175
168 INVX1FNTC_IN: INVX1FNTC_IN:12 31.7256
169 INVX1FNTC_IN: INVX1FNTC_IN:4 11.9254
170 INVX1FNTC_IN: INVX1FNTC_IN:7 25.3618
171 INVX1FNTC_IN: INVX1FNTC_IN:6 23.3057
172 INVX1FNTC_IN: INVX1FNTC_IN:24 8.64717
173 INVX1FNTC_IN: INVX1FNTC_IN:8 7.46529
174 INVX1FNTC_IN: INVX1FNTC_IN:10 2.04729
175 INVX1FNTC_IN: INVX1FNTC_IN:10 10.8533
176 INVX1FNTC_IN: INVX1FNTC_IN:11 1.05164
*END
*D_NET NE_794 1.98538
*CONN
*I NL_1039:X O *L 0 *D INVX
*I NL_2039:A I *L 0.343
*I NL_1040:A I *L 0.343
*CAP
3387
3388
3389
3390
3391
3392
3393
346
NE_794 0
NE_794:1 0.0792492
NE_794:10 0.0789158
NE_794:11 0.0789991
NE_794:12 0.0789991
NE_794:13 0.0792992
NE_794:14 0.00093352
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Chapter 13: Post-Layout Analysis
Post-Layout Back-Annotation
3394
3395
3396
3397
3398
3399
3400
3401
3402
3403
3404
3405
3406
3407
3408
3409
3410
3411
3412
3413
3414
NE_794:15 0.00063346
NE_794:16 0.0792992
NE_794:17 0.80116
NE_794:18 0.80116
NE_794:19 0.00125452
NE_794:2 0.0789158
NE_794:20 0.00336991
NE_794:21 0.00668512
NE_794:23 0.00294932
NE_794:25 0.00259882
NE_794:26 0.00184653
NE_794:3 0.0789158
NE_794:4 0.0796826
NE_794:5 0.0796826
NE_794:6 0.0789991
NE_794:7 0.0789991
NE_794:8 0.0793992
NE_794:9 0.0789158
NL_1039:X 0.00871972
NL_1040:A 0.344453
NL_2039:A 0.343427
*RES
2879
2880
2881
2882
2883
2884
2885
2886
2887
2888
2889
2890
2891
2892
2893
2894
2895
2896
2897
2898
2899
2900
2901
2902
2903
NE_794:1 NE_794:13 66.1953
NE_794:1 NE_794:2 0.311289
NE_794:11 NE_794:12 0.311289
NE_794:13 NE_794:14 0.353289
NE_794:14 NE_794:19 0.365644
NE_794:15 NE_794:16 0.227289
NE_794:15 NE_794:20 0.239644
NE_794:17 NE_794:18 0.14
NE_794:19 NE_794:21 0.0511746
NE_794:2 NE_794:9 65.9153
NE_794:20 NE_794:23 1.15117
NE_794:21 NL_1039:X 3.01917
NE_794:25 NE_794:26 0.166349
NE_794:26 NL_1040:A 0.651175
NE_794:3 NE_794:10 65.9153
NE_794:3 NE_794:4 0.311289
NE_794:4 NE_794:17 66.5437
NE_794:5 NE_794:18 66.5437
NE_794:5 NE_794:6 0.311289
NE_794:6 NE_794:11 65.98853
NE_794:7 NE_794:12 65.9853
NE_794:7 NE_794:8 0.311289
NE_794:8 NE_794:16 66.3213
NE_794:9 NE_794:10 0.311289
NL_1039:X NE_794:25 1.00317
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Chapter 13: Post-Layout Analysis
Linear Acceleration
2904 NL_2039:A NE_794:23 0.171175
*END
Linear Acceleration
Linear acceleration, by using the SIM_LA option, accelerates the simulation of
circuits that include large linear RC networks. To achieve this acceleration,
HSPICE RF reduces all matrices that represent RC networks. The result is a
smaller matrix that maintains the original port behavior, yet achieves significant
savings in memory and computation. Thus, the SIM_LA option is ideal for
circuits with large numbers of resistors and capacitors, such as clock trees,
power lines, or substrate networks.
In general, the RC elements are separated into their own network. The nodes
shared by both main circuit elements (including .PRINT, .PROBE,
and .MEASURE statements), and RC elements. are the port nodes of the RC
network,. All other RC nodes are internal nodes. The currents flowing into the
port nodes are a frequency-dependent function of the voltages at those nodes.
The multiport admittance of a network represents this relationship.
348
■
The SIM_LA option formulates matrices to represent multiport admittance.
■
Then, to eliminate as many internal nodes as possible, it reduces the size of
these matrices, while preserving the admittance, otherwise known as port
node behavior.
■
The amount of reduction depends on the f0 upper frequency, the threshold
frequency where SIM_LA preserves the admittance. This is shown
graphically in Figure 29.
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Chapter 13: Post-Layout Analysis
Linear Acceleration
admittance
nce
itta
m
d
a
ual
approx
act
f0
Figure 29
frequency
Multiport Admittance vs. Frequency
The SIM_LA option is very effective for post-layout simulation, because of the
volume of parasitics. For frequencies below f0, the approx signal matches that
of the original admittance. Above f0, the two waveforms diverge, but
presumably the higher frequencies are not of interest. The lower the f0
frequency, the greater the amount of reduction.
For the syntax and description of this control option, see .OPTION SIM_LA in
the HSPICE Reference Manual: Commands and Control Options.
You can choose one of two algorithms, explained in the following sections:
■
PACT Algorithm
■
PI Algorithm
PACT Algorithm
The PACT (Pole Analysis via Congruence Transforms) algorithm reduces the
RC networks in a well-conditioned manner, while preserving network stability.
■
The transform preserves the first two moments of admittance at DC (slope
and offset), so that DC behavior is correct (see Figure 30).
■
The algorithm preserves enough low-frequency poles from the original
network to maintain the circuit behavior up to a specified maximum
frequency f0, within the specified tolerance.
This approach is the most accurate of the two algorithms, and is the default.
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Chapter 13: Post-Layout Analysis
Linear Acceleration
admittance
ed
erv
s
re
tp
e
s
off
nd
ce
a
tan
e
t
i
p
dm
slo
al a
u
t
ac
PACT: poles added
f0
Figure 30
frequency
PACT Algorithm
PI Algorithm
This algorithm creates a pi model of the RC network.
■
For a two-port, the pi model reduced network consists of:
•
a resistor connecting the two ports, and
•
a capacitor connecting each port to ground
The result resembles the Greek letter pi.
■
For a general multiport, SIM_LA preserves the DC admittance between the
ports, and the total capacitance that connects the ports to ground. All
floating capacitances are lumped to ground.
Linear Acceleration Control Options Summary
In addition to .OPTION SIM_LA, other options are available to control the
maximum resistance and minimum capacitance values to preserve, and to limit
the operating parameters of the PACT algorithm. Table 27 on page 351
contains a summary of these control options. For the syntax and descriptions of
these options, see the respective sections in the HSPICE and RF Netlist
Simulation Control Options in the HSPICE Reference Manual: Commands and
Control Options.
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Linear Acceleration
Table 27
PACT Options
Syntax
Description
.OPTION SIM_LA=PACT | PI
Activates linear matrix reduction and selects
between two methods.
.OPTION SIM_LA_FREQ=<value>
Upper frequency where you need accuracy
preserved. value is the upper frequency for
which the PACT algorithm preserves accuracy. If
value is 0, PACT drops all capacitors, because
only DC is of interest. The maximum frequency
required for accurate reduction depends on both
the technology of the circuit and the time scale
of interest. In general, the faster the circuit, the
higher the maximum frequency. The default is
1GHz.
.OPTION SIM_LA_MAXR=<value>
Maximum resistance for linear matrix reduction.
value is the maximum resistance preserved in
the reduction. SIM_LA assumes that any
resistor greater than value has an infinite
resistance, and drops the resistor after reduction
finishes. The default is 1e15 ohms.
.OPTION SIM_LA_MINC=<value>
Minimum capacitance for linear matrix
reduction. value is the minimum capacitance
preserved in the reduction. After reduction
completes, SIM_LA lumps any capacitor smaller
than value to ground. The default is 1e-16
farads.
.OPTION SIM_LA_MINMODE=
ON|OFF
Reduces the number of nodes instead of the
number of elements.
.OPTION SIM_LA_TIME=<value>
Minimum time for which accuracy must be
preserved. value is the minimum switching time
for which the PACT algorithm preserves
accuracy. HSPICE RF does not accurately
represent waveforms that occur more rapidly
than this time. SIM_LA_TIME is simply the dual
of SIM_LA_FREQ. The default is equivalent to
setting LA_FREQ=1 GHz. The default is 1ns.
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Chapter 13: Post-Layout Analysis
Linear Acceleration
Table 27
PACT Options (Continued)
Syntax
Description
.OPTION SIM_LA_TOL=<value>
Error tolerance for the PACT algorithm. value is
the error tolerance for the PACT algorithm, is
between 0.0 and 1.0. The default is 0.05.
Example
In this example, the circuit has a typical risetime of 1ns. Set the maximum
frequency to 1 GHz, or set the minimum switching time to 1ns.
.OPTION SIM_LA_FREQ = 1GHz
-or.OPTION SIM_LA_TIME = 1ns
However, if spikes occur in 0.1ns, HSPICE will not accurately simulate them. To
capture the behavior of the spikes, use:
.OPTION SIM_LA_FREQ = 10GHz
-or.OPTION SIM_LA_TIME = 0.1ns
Note:
Higher frequencies (smaller times) increase accuracy, but only up to the
minimum time step used in HSPICE.
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14
Statistical and Monte Carlo Analysis
14
Describes the features available in HSPICE RF for statistical analysis.
Described in this chapter are the features available in HSPICE RF for statistical
analysis. These features are supported for HSPICE RF and differ from the
enhanced statistical analysis features available for HSPICE (described in the
chapter Monte Carlo Analysis Using the Variation Block Flow in the HSPICE
User Guide: Simulation and Analysis.)
The following subjects are described in this chapter:
■
Application of Statistical Analysis
■
Analytical Model Types
■
Simulating Circuit and Model Temperatures
■
Worst Case Analysis
■
Getting Started with Traditional Monte Carlo Simulations
■
Monte Carlo Analysis
■
Worst Case and Monte Carlo Sweep Example
■
Simulating the Effects of Global and Local Variations with Monte Carlo
Application of Statistical Analysis
When you design an electrical circuit, it must meet tolerances for the specific
manufacturing process. The electrical yield is the number of parts that meet the
electrical test specifications. Overall process efficiency requires maximum
yield. To analyze and optimize the yield, HSPICE RF supports statistical
techniques and observes the effects of variations in element and model
parameters.
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Chapter 14: Statistical and Monte Carlo Analysis
Analytical Model Types
Analytical Model Types
To model parametric and statistical variation in circuit behavior, use:
■
.PARAM statement to investigate the performance of a circuit as you change
circuit parameters. For details about the .PARAM statement, see
the .PARAM statement in the HSPICE Reference Manual: Commands and
Control Options
■
Temperature variation analysis to vary the circuit and component
temperatures, and compare the circuit responses. You can study the
temperature-dependent effects of the circuit, in detail.
■
Monte Carlo analysis when you know the statistical standard deviations of
component values to center a design. This provides maximum process
yield, and determines component tolerances.
■
Worst-case corner analysis when you know the component value limit to
automate quality assurance for:
■
•
basic circuit function
•
process extremes
•
quick estimation of speed and power tradeoffs
•
best-case and worst-case model selection
•
parameter corners
•
library files
Data-driven analysis for cell characterization, response surface, or Taguchi
analysis. See Performing Digital Cell Characterization in the HSPICE User
Guide: Simulation and Analysis. Automates characterization of cells and
calculates the coefficient of polynomial delay for timing simulation. You can
simultaneously vary any number of parameters and perform an unlimited
number of analyses. This analysis uses an ASCII file format so HSPICE RF
can automatically generate parameter values. This analysis can replace
hundreds or thousands of HSPICE RF simulation runs.
Use yield analyses to modify:
354
■
DC operating points
■
DC sweeps
■
AC sweeps
■
Transient analysis.
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Simulating Circuit and Model Temperatures
CosmosScope can generate scatter plots from the operating point analysis or a
family of curve plots for DC, AC, and transient analysis.
Use .MEASURE statements to save results for delay times, power, or any other
characteristic extracted in a .MEASURE statement. HSPICE RF generates a
table of results in an .mt# file in ASCII format. You can analyze the numbers
directly or read this file into CosmosScope to view the distributions. Also, if you
use .MEASURE statements in a Monte Carlo or data-driven analysis, then the
HSPICE RF output file includes the following statistical results in the listing:
x 1 + x 2 + …+ x n
Mean -------------------------------------N
( x 1 – Mean ) 2 + …x
( n – Mean ) 2
---------------------------------------------------------------------------Variance
N–1
Sigma
Variance
x 1 – Mean + …+ x n – Mean
Average Deviation -------------------------------------------------------------------------N–1
Simulating Circuit and Model Temperatures
Temperature affects all electrical circuits. Figure 31 shows the key temperature
parameters associated with circuit simulation:
■
Model reference temperature – you can model different models at different
temperatures. Each model has a TREF (temperature reference) parameter.
■
Element junction temperature – each resistor, transistor, or other element
generates heat so an element is hotter than the ambient temperature.
■
Part temperature – at the system level each part has its own temperature.
■
System temperature – a collection of parts form a system, which has a local
temperature.
■
Ambient temperature – the ambient temperature is the air temperature of
the system.
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Chapter 14: Statistical and Monte Carlo Analysis
Simulating Circuit and Model Temperatures
Ambient Temperature
System Temperature
source
drain
gate
Model Junction Temperature
Figure 31
Part Temperature
source
drain
gate
Part Junction Temperature
Part Junction Temperature Sets System Performance
HSPICE RF calculates temperatures as differences from the ambient
temperature:
Equation 61
Tambient + Δsystem + Δpart + Δjunction = Tjunction
Equation 62
Ids = f ( Tjunction, Tmodel )
Every element includes a DTEMP keyword, which defines the difference
between junction and ambient temperature.
Example
The following example uses DTEMP in a MOSFET element statement:
M1 drain gate source bulk Model_name W=10u L=1u DTEMP=+20
Temperature Analysis
You can specify three temperatures:
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Simulating Circuit and Model Temperatures
■
Model reference temperature specified in a .MODEL statement. The
temperature parameter is usually TREF, but can be TEMP or TNOM in some
models. This parameter specifies the temperature, in ° C, at which
HSPICE RF measures and extracts the model parameters. Set the value of
TNOM in an .OPTION statement. Its default value is 25° C.
■
Circuit temperature that you specify using a .TEMP statement or the TEMP
parameter. This is the temperature, in ° C, at which HSPICE RF simulates
all elements. To modify the temperature for a particular element, use the
DTEMP parameter. The default circuit temperature is the value of TNOM.
■
Individual element temperature, which is the circuit temperature, plus an
optional amount that you specify in the DTEMP parameter.
To specify the temperature of a circuit in a simulation run, use either the .TEMP
statement, or the TEMP parameter in the .DC, .AC, or .TRAN statements.
HSPICE RF compares the circuit simulation temperature that one of these
statements sets against the reference temperature that the TNOM option sets.
TNOM defaults to 25° C, unless you use the SPICE option, which defaults to
27° C. To calculate the derating of component values and model parameters,
HSPICE RF uses the difference between the circuit simulation temperature,
and the TNOM reference temperature.
Elements and models within a circuit can operate at different temperatures. For
example, a high-speed input/output buffer that switches at 50 MHz is much
hotter than a low-drive NAND gate that switches at 1 MHz). To simulate this
temperature difference, specify both an element temperature parameter
(DTEMP), and a model reference parameter (TREF). If you specify DTEMP in an
element statement, the element temperature for the simulation is:
element temperature=circuit temperature + DTEMP
Specify the DTEMP value in the element statement (resistor, capacitor, inductor,
diode, BJT, JFET, or MOSFET statement), or in a subcircuit element. Assign a
parameter to DTEMP, then use the .DC statement to sweep the parameter. The
DTEMP value defaults to zero.
If you specify TREF in the model statement, the model reference temperature
changes (TREF overrides TNOM). Derating the model parameters is based on
the difference between circuit simulator temperature and TREF (instead of
TNOM).
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Chapter 14: Statistical and Monte Carlo Analysis
Worst Case Analysis
.TEMP Statement
To specify the temperature of a circuit for a HSPICE RF simulation, use
the .TEMP statement.
Worst Case Analysis
Circuit designers often use worst-case analysis when designing and analyzing
MOS and BJT IC circuits. To simulate the worst case, set all variables to their 2or 3-sigma worst-case values. Because several independent variables rarely
attain their worst-case values simultaneously, this technique tends to be overly
pessimistic and can lead to over-designing the circuit. However, this analysis is
useful as a fast check.
Model Skew Parameters
The HSPICE RF device models include physically-measurable model
parameters. The circuit simulator uses parameter variations to predict how an
actual circuit responds to extremes in the manufacturing process. Physicallymeasurable model parameters are called skew parameters, because they skew
from a statistical mean to obtain predicted performance variations.
Examples of skew parameters are the difference between the drawn and
physical dimension of metal, postillion, or active layers, on an integrated circuit.
Generally, you specify skew parameters independently of each other, so you
can use combinations of skew parameters to represent worst cases. Typical
skew parameters for CMOS technology include:
■
XL – polysilicon CD (critical dimension of the poly layer, representing the
difference between drawn and actual size).
■
XWn, XWp – active CD (critical dimension of the active layer, representing the
difference between drawn and actual size).
■
TOX – thickness of the gate oxide.
■
RSHn, RSHp – resistivity of the active layer.
■
DELVTOn, DELVTOp– variation in threshold voltage.
You can use these parameters in any level of MOS model, within the HSPICE
RF device models. The DELVTO parameter shifts the threshold value. HSPICE
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RF adds this value to VTO for the Level 3 model, and adds or subtracts it from
VFB0 for the BSIM model. Table 28 shows whether HSPICE RF adds or
subtracts deviations from the average.
Table 28
Sigma Deviations
Type
Parameter
Slow
Fast
NMOS
XL
+
-
RSH
+
-
DELVTO
+
-
TOX
+
-
XW
-
+
XL
+
-
RSH
+
-
DELVTO
-
+
TOX
+
-
XW
-
+
PMOS
HSPICE RF selects skew parameters based on the available historical data
that it collects either during fabrication or electrical test. For example,
HSPICE RF collects the XL skew parameter for poly CD during fabrication. This
parameter is usually the most important skew parameter for a MOS process.
Figure 32 is an example of data that historical records produce.
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3 sigma
2 sigma
Fab Database
1 sigma
Run# PolyCD
Mean
101 +0.04u
102 -0.06u
pop.#
103 +0.03u
...
XL value
Figure 32
Historical Records for Skew Parameters in a MOS Process
Using Skew Parameters
Figure 33 shows how to create a worst-case corners library file for a CMOS
process model. Specify the physically-measured parameter variations so that
their proper minimum and maximum values are consistent with measured
current (IDS) variations. For example, HSPICE can generate a 3-sigma
variation in IDS from a 2-sigma variation in physically-measured parameters.
SS
Slow Corner Skew Parameters
EE
Extracted Skew Parameters
TT
Typical Corner Skew Parameters + Gaussian
FF
Fast Corner Skew Parameters
pop.
IDS
Figure 33
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The .LIB (library) statement, and the .INCLUDE (include file) statement,
access the models and skew. The library contains parameters that
modify .MODEL statements. The following example of .LIB features both
worst-case and statistical-distribution data by using model skew parameters. In
statistical distribution, the median value is the default for all non-Monte Carlo
analysis.
Example
.LIB TT
$TYPICAL P-CHANNEL AND N-CHANNEL CMOS LIBRARY DATE:3/4/91
$ PROCESS: 1.0U CMOS, FAB22, STATISTICS COLLECTED 3/90-2/91
$ following distributions are 3 sigma ABSOLUTE GAUSSIAN
.PARAM
$ polysilicon Critical Dimensions
+ polycd=agauss(0,0.06u,1) xl=’polycd-sigma*0.06u’
$ Active layer Critical Dimensions
+ nactcd=agauss(0,0.3u,1) xwn=’nactcd+sigma*0.3u’
+ pactcd=agauss(0,0.3u,1) xwp=’pactcd+sigma*0.3u’
$ Gate Oxide Critical Dimensions (200 angstrom +/- 10a at 1
$ sigma)
+ toxcd=agauss(200,10,1) tox=’toxcd-sigma*10’
$ Threshold voltage variation
+ vtoncd=agauss(0,0.05v,1) delvton=’vtoncd-sigma*0.05’
+ vtopcd=agauss(0,0.05v,1) delvtop=’vtopcd+sigma*0.05’
.INC ‘/usr/meta/lib/cmos1_mod.dat’ $ model include file
.ENDL TT
.LIB FF
$HIGH GAIN P-CH AND N-CH CMOS LIBRARY 3SIGMA VALUES
.PARAM TOX=230 XL=-0.18u DELVTON=-.15V DELVTOP= 0.15V
.INC ‘/usr/meta/lib/cmos1_mod.dat’ $ model include file
.ENDL FF
The /usr/meta/lib/cmos1_mod.dat include file contains the model.
.MODEL NCH NMOS LEVEL=2 XL=XL TOX=TOX DELVTO=DELVTON . .
.MODEL PCH PMOS LEVEL=2 XL=XL TOX=TOX DELVTO=DELVTOP . .
Note:
The model keyname (left) equals the skew parameter (right). Model keys
and skew parameters can use the same names.
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Skew File Interface to Device Models
Skew parameters are model parameters for transistor models or passive
components. A typical device model set includes:
■
MOSFET models for all device sizes by using an automatic model selector.
■
RC wire models for polysilicon, metal1, and metal2 layers in the drawn
dimension. Models include temperature coefficients and fringe capacitance.
■
Single-diode and distributed-diode models for N+, P+, and well (includes
temperature, leakage, and capacitance based on the drawn dimension).
■
BJT models for parasitic bipolar transistors. You can also use these for any
special BJTs, such as a BiCMOS for ECL BJT process (includes current and
capacitance as a function of temperature).
■
Metal1 and metal2 transmission line models for long metal lines.
■
Models must accept elements. Sizes are based on a drawn dimension. If
you draw a cell at 2μ dimension and shrink it to 1μ, the physical size is 0.9μ.
The effective electrical size is 0.8μ. Account for the four dimension levels:
•
drawn size
•
shrunken size
•
physical size
•
electrical size
Most simulator models scale directly from drawn to electrical size. HSPICE
MOS models support all four size levels as in Figure 34.
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Drawn Size
Shrunken Size
2m
1m
LMLT
WMLT
XL
XW
Electrical Size
source
Physical Size
source
drain
drain
gate
gate
LD
WD
0.8 m
Figure 34
0.9 m
Device Model from Drawn to Electrical Size
Getting Started with Traditional Monte Carlo Simulations
The following is a high-level overview of HSPICE traditional Monte Carlo
analysis. The sections that follow provide more in-depth information.
Note:
Traditional Monte Carlo analysis is available to HSPICE RF users. For the
most effective and efficient methods it is suggested that you explore
Variation Block-based Monte Carlo simulation, available in HSPICE, only
(see Monte Carlo Analysis Using the Variation Block Flow in the HSPICE
User Guide: Simulation and Analysis).
The basic premise of a Monte Carlo analysis is that you are going to
parameterize one or more circuit variables, vary those values by a randomized
amount from the norm, and run HSPICE/HSPICE RF a pre-determined number
of times. Each run is called a sweep which generates tabular or plot data as
specified by the user. Measurements are also typically used to look at circuit
operating conditions from run to run.
You can randomize anything that can be set with a parameter or variable.
Examples include things as diverse as a simple resistor value, a model
parameter for a MOSFET, or the length of a transmission line.
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Values can be varied using three basic statistical variations; uniform, limit, and
Gaussian. Using those methods, you choose the nominal value and the
absolute or relative variation. You can optionally supply the standard deviation
and a multiplier.
Basic Syntax
The basic syntax of a Monte Carlo analysis includes three elements:
1. Defining a parameter with one of the distribution keywords
2. Using the parameter in your netlist as the value for an element or model
parameter
3. Including the keywords SWEEP and MONTE in the analysis statement
Consider the following example. In this simple RC charging circuit, the value of
r1 has a nominal value of 1K and is varied by 400 ohms for 10 iterations.
RC charging circuit
.option post probe
*define a parameter called "resval" with an absolute, uniform
distribution
.param resval=aunif(1000,400)
vsrc_one 1 0 5v
r_one 1 2 resval
c_one 2 0 1u
.ic 2=0
*specify 10 Monte Carlo iterations
.tran 1e-5 5e-3 sweep monte=10
*measure to find when 1 time constant (.632*vdd) occurs
.meas tran tc when v(2)='.632*5'
*create plots of the charging curve and resistor values
.probe v(2) par(resval)
.end
The resulting waveforms are called “multi-member”. Plotting the one signal
displays the curves from all the runs.
Local and Global Parameter Variation
A common source of confusion is local and global parameter variation. The key
is that each time you use a parameter, it gets assigned a new random value.
Take the following examples:
.param resval=aunif(1000,400)
r_one 1 2 resval
r_two 2 3 resval
r_three 3 4 resval
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In this case, all three resistors get unique, random values. If you want to set a
group of components to the same random value, assign an intermediate
parameter first:
.param resval=aunif(1000,400)
.param my_resval=resval
r_one 1 2 my_resval
r_two 2 3 my_resval
r_three 3 4 my_resval
In the second example, the assignment of a random value is only done once,
then used three times. The exception to this rule is for model parameters.
Exception for Model Parameters
Since a model definition is only done once, the behavior described above
would assign the same parameter value to all devices referencing that model.
To overcome this, .OPTION MODMONTE lets the user decide if all instances of
a device should get the same or unique model parameters.
Starting Values and Seeds
Another source of confusion is the starting value. If you run the same Monte
Carlo simulation twice, the results will be identical. Why? HSPICE/HSPICE RF
always uses the same “seed” value for the first run. If it randomized the seed by
default, it would be hard if not impossible to tell whether changes you made to
the circuit and topology were the result of your changes or the new random
values. You can specify a seed or tell HSPICE to pick a random seed
with .OPTION SEED if that behavior is desired.
Other Monte Carlo Control Options
.OPTION MONTECON—some random parameter assignments can cause
HSPICE not to converge. This parameter is used to decide whether to
terminate a simulation or move to the next run if convergence fails.
■
■
.OPTION RANDGEN—use this option to specify the type of random number
generator used.
■
.OPTION MCBRIEF—controls how HSPICE outputs Monte Carlo
parameters.
Monte Carlo Analysis
Monte Carlo analysis uses a random number generator to create the following
types of functions.
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■
■
■
Gaussian parameter distribution
•
Relative variation—variation is a ratio of the average.
•
Absolute variation—adds variation to the average.
•
Bimodal–multiplies distribution to statistically reduce nominal
parameters.
Uniform parameter distribution
•
Relative variation—variation is a ratio of the average.
•
Absolute variation—adds variation to the average.
•
Bimodal–multiplies distribution to statistically reduce nominal
parameters.
Random limit parameter distribution
•
Absolute variation—adds variation to the average.
•
Monte Carlo analysis randomly selects the min or max variation.
The value of the MONTE analysis keyword determines how many times to
perform operating point, DC sweep, AC sweep, or transient analysis.
Uniform Distribution
Gaussian Distribution
Population
Population
Abs
variation
3 Sigma
Nom_value
Abs
variation
Nom_value
Rel_variation=Abs_variation/Nom_value
Figure 35
Monte Carlo Distribution
Monte Carlo Setup
To set up a Monte Carlo analysis, use the following HSPICE statements:
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■
.PARAM statement—sets a model or element parameter to a Gaussian,
Uniform, or Limit function distribution.
■
.DC, .AC, or .TRAN analysis—enables MONTE.
■
.MEASURE statement—calculates the output mean, variance, sigma, and
standard deviation.
■
.MODEL statement—sets model parameters to a Gaussian, Uniform, or
Limit function distribution.
Select the type of analysis to run, such as operating point, DC sweep, AC
sweep, or TRAN sweep.
Operating Point
.DC MONTE=<firstrun=num1>
-or.DC MONTE=list <(> <num1:num2> <num3> <num5:num6> <num7> <)>
DC Sweep
.DC vin 1 5 0.25 sweep MONTE=val <firstrun=num1>
-or.DC vin 1 5 0.25 sweep MONTE=list<(> <num1:num2> <num3>
+ <num5:num6> <num7> <)>
AC Sweep
.AC dec 10 100 1meg sweep MONTE=val <firstrun=num1>
-or.AC dec 10 100 1meg sweep MONTE=list<(> <num1:num2>
+ <num3> <num5:num6> <num7> <)>
TRAN Sweep
.TRAN 1n 10n sweep MONTE=val <firstrun=num1>
-or.TRAN 1n 10n sweep MONTE=list<(> <num1:num2> <num3>
+ <num5:num6> <num7> <)>
The val value specifies the number of Monte Carlo iterations to perform. A
reasonable number is 30. The statistical significance of 30 iterations is quite
high. If the circuit operates correctly for all 30 iterations, there is a 99%
probability that over 80% of all possible component values operate correctly.
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The relative error of a quantity, determined through Monte Carlo analysis, is
proportional to val-1/2.
The firstrun values specify the desired number of iterations. HSPICE RF runs
from num1 to num1+val-1. The number after firstrun can be a parameter. You
can write only one number after list. The colon represents “from ... to ...".
Specifying only one number makes HSPICE RF runs only a the one specified
point.
Example 1
In this example, HSPICE RF runs from the 90th to 99th Monte Carlo iterations:
.tran 1n 10 sweep monte=10 firstrun=90
You can write more than one number after list. The colon represents “from ... to
...". Specifying only one number makes HSPICE RF run only at that single
point.
Example 1
In this example, HSPICE RF begins running at the 10th iteration, then
continues from the 20th to the 30th, at the 40th, and finally from the 46th to
72nd Monte Carlo iteration. The numbers after list can not be parameter.
.tran 1n 10n sweep monte=list(10 20:30 40 46:72)
Monte Carlo Output
368
■
.MEASURE statements are the most convenient way to summarize the
results.
■
.PRINT statements generate tabular results, and print the values of all
Monte Carlo parameters.
■
.MCBRIEF determines the output types of the random parameters during
Monte Carlo analysis to improve output performance.
■
If one iteration is out of specification, you can obtain the component values
from the tabular listing. A detailed re-simulation of that iteration might help
identify the problem.
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.PARAM Distribution Function
This section describes how to use assign a .PARAM parameter in Monte Carlo
analysis. For a general description of the .PARAM statement, see the .PARAM
command in the HSPICE and HSPICE RF Command Reference.
You can assign a .PARAM parameter to the keywords of elements and models,
and assign a distribution function to each .PARAM parameter. HSPICE RF
recalculates the distribution function each time that and element or model
keyword uses a parameter. When you use this feature, Monte Carlo analysis
can use a parameterized schematic netlist without additional modifications.
Syntax
.PARAM xx=UNIF(nominal_val, rel_variation
+ <, multiplier>)
.PARAM xx=AUNIF(nominal_val, abs_variation <,
+ multiplier>)
.PARAM xx=GAUSS(nominal_val, rel_variation, sigma <,
+ multiplier>)
.PARAM xx=AGAUSS(nominal_val, abs_variation, sigma <,
+ multiplier>)
.PARAM xx=LIMIT(nominal_val, abs_variation)
Argument
Description
xx
Distribution function calculates the value of this parameter.
UNIF
Uniform distribution function by using relative variation.
AUNIF
Uniform distribution function by using absolute variation.
GAUSS
Gaussian distribution function by using relative variation.
AGAUSS
Gaussian distribution function by using absolute variation
LIMIT
Random-limit distribution function by using absolute variation.
Adds +/- abs_variation to nominal_val based on whether the
random outcome of a -1 to 1 distribution is greater than or less
than 0.
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Argument
Description
nominal_val
Nominal value in Monte Carlo analysis and default value in all
other analyses.
abs_variation
AUNIF and AGAUSS vary the nominal_val by +/- abs_variation.
rel_variation
UNIF and GAUSS vary the nominal_val by +/- (nominal_val ⋅
rel_variation).
sigma
Specifies abs_variation or rel_variation at the sigma level. For
example, if sigma=3, then the standard deviation is abs_variation
divided by 3.
multiplier
If you do not specify a multiplier, the default is 1. HSPICE RF
recalculates many times and saves the largest deviation. The
resulting parameter value might be greater than or less than
nominal_val. The resulting distribution is bimodal.
Example 1
In this example, each R has an unique variation.
.param mc_var=agauss(0,1,3)
.param val='1000*(1+mc_var)'
v_vin vin 0 dc=1 ac=.1
r1 vin 0 '1000*(1+mc_var)'
r2 vin 0 '1000*(1+mc_var)'
$ +/- 20% swing
Example 2
In this example, each R has an identical variation.
.param mc_var=agauss(0,1,3)
.param val='1+mc_var'
v_vin vin 0 dc=1 ac=.1
r1 vin 0 '1000*val'
r2 vin 0 '1000*val'
$ +/- 20% swing
Example 3
In this example, local variations to an instance parameter are applied by
assigning randomly-generated variations directly to each instance parameter.
Each resistor r1 through r3 receives randomly different resistance values
during each Monte Carlo run.
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.param
r1 1 2
r2 3 4
r3 5 6
r_local=agauss(...)
r=r_local
r=r_local
r=r_local
Example 4
In this example, global variations to an instance parameter are applied by
assigning the variation to an intermediate parameter before assigning it to each
instance parameter. Each resistor r1 through r3 receives the same random
resistance value during each Monte Carlo run.
.param
.param
r1 1 2
r2 3 4
r3 5 6
r_random=agauss(...)
r_global=r_random
r=r_global
r=r_global
r=r_global
Monte Carlo Parameter Distribution
Each time you use a parameter, Monte Carlo calculates a new random variable.
■
If you do not specify a Monte Carlo distribution, then HSPICE RF assumes
the nominal value.
■
If you specify a Monte Carlo distribution for only one analysis, HSPICE RF
uses the nominal value for all other analyses.
You can assign a Monte Carlo distribution to all elements that share a common
model. The actual element value varies according to the element distribution. If
you assign a Monte Carlo distribution to a model keyword, then all elements
that share the model, use the same keyword value. You can use this feature to
create double element and model distributions.
For example, the MOSFET channel length varies from transistor to transistor by
a small amount that corresponds to the die distribution. The die distribution is
responsible for offset voltages in operational amplifiers, and for the tendency of
flip-flops to settle into random states. However, all transistors on a die site vary
according to the wafer or fabrication run distribution. This value is much larger
than the die distribution, but affects all transistors the same way. You can
specify the wafer distribution in the MOSFET model to set the speed and power
dissipation characteristics.
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Monte Carlo Examples
Gaussian, Uniform, and Limit Functions
You can find the sample netlist for this example in the following directory:
$installdir/demo/hspice/apps/mont1.sp
119.182
MONT1.SP TEST OF MONTE CARLO, GAUSSIAN, UNIFORM, AND LIMIT FUNCTIONS
May 15 2003 11:41:23
MONT1_SV0
RUNIF_1
110.0
VOLT [LIN]
100.0
90.0
80.1384
120.0
MONT1_SV0
RUNIF_10
110.0
100.0
90.0
80.0402
1.0
10.0
20.0
30.0
40.0
50.0
60.0
MONTE CARLO [LIN]
Figure 36
372
Uniform Functions
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115.0
MONT1.SP TEST OF MONTE CARLO, GAUSSIAN, UNIFORM, AND LIMIT FUNCTIONS
May 15 2003 11:41:23
MONT1_SV
RGAUSS_1
110.0
VOLT [LIN]
105.0
100.0
95.0
90.0
MONT1_SV
RGAUSS_1
118.375
110.0
100.0
90.0
80.9998
1.0
10.0
20.0
30.0
40.0
50.0
60.0
MONTE CARLO [LIN]
Figure 37
Gaussian Functions
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MONT1.SP TEST OF MONTE CARLO, GAUSSIAN, UNIFORM, AND LIMIT FUNCTIONS
May 15 2003 11:41:23
MONT1.SV0
LIMIT
120.0
115.0
110.0
VOLT [LIN]
105.0
100.0
95.0
90.0
85.0
80.0
1.0
10.0
20.0
30.0
40.0
50.0
60.0
MONTE CARLO [LIN]
Figure 38
Limit Functions
Major and Minor Distribution
In MOS IC processes, manufacturing tolerance parameters have both a major
and a minor statistical distribution.
374
■
The major distribution is the wafer-to-wafer and run-to-run variation. It
determines electrical yield.
■
The minor distribution is the transistor-to-transistor process variation. It is
responsible for critical second-order effects, such as amplifier offset voltage
and flip-flop preference.
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major distribution
minor distribution
pop.#
XL
(polysilicon linewidth variation)
Figure 39
Major and Minor Distribution of Manufacturing Variations
The following example is a Monte Carlo analysis of a DC sweep in HSPICE RF.
Monte Carlo sweeps the VDD supply voltage from 4.5 volts to 5.5 volts.
You can find the sample netlist for this example in the following directory:
$installdir/demo/hspice/apps/mondc_a.sp
■
The M1 through M4 transistors form two inverters.
■
The nominal value of the LENGTH parameter sets the channel lengths for the
MOSFETs, which are set to 1u in this example.
■
All transistors are on the same integrated circuit die. The LEFF parameter
specifies the distribution—for example, a ±5% distribution in channel length
variation at the ±3-sigma level.
■
Each MOSFET has an independent random Gaussian value.
The PHOTO parameter controls the difference between the physical gate length
and the drawn gate length. Because both n-channel and p-channel transistors
use the same layer for the gates, Monte Carlo analysis sets XPHOTO
distribution to the PHOTO local parameter.
XPHOTO controls PHOTO lithography for both NMOS and PMOS devices, which
is consistent with the physics of manufacturing.
RC Time Constant
This simple example shows uniform distribution for resistance and capacitance.
It also shows the resulting transient waveforms for 10 different random values.
You can find the sample netlist for this example in the following directory:
$installdir/demo/hspice/apps/rc_monte.sp
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*FILE: NOM1.SP WITH UNIFORM DISTRIBUTION
May 15 2003 12:38:49
MONT1.SV0
1
992.750N
900.0N
VOLT [LIN]
800.0N
700.0N
600.0N
10
500.0N
9
8
400.0N
7
5
26
3
1
300.0N
0
200.0N
400.0N
600.0N
800.0N
1.0
TIME [LIN]
Figure 40
Monte Carlo Analysis of RC Time Constant
Switched Capacitor Filter Design
Capacitors used in switched-capacitor filters consist of parallel connections of a
basic cell. Use Monte Carlo techniques in HSPICE RF to estimate the variation
in total capacitance. The capacitance calculation uses two distributions:
376
■
Minor (element) distribution of cell capacitance from cell-to-cell on a single
die.
■
Major (model) distribution of the capacitance from wafer-to-wafer or from
manufacturing run-to-run.
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cap-to-cap
(element)
C1a
C1b
C1a
C1b
C1c
C1d
C1c
C1d
run-to-run
(model)
Figure 41
Monte Carlo Distribution
You can approach this problem from physical or electrical levels.
■
The physical level relies on physical distributions, such as oxide thickness
and polysilicon line width control.
■
The electrical level relies on actual capacitor measurements.
Physical Approach:
1. Since oxide thickness control is excellent for small areas on a single wafer,
you can use a local variation in polysilicon to control the variation in
capacitance for adjacent cells.
2. Next, define a local poly line-width variation and a global (model-level) poly
line-width variation. In this example:
•
The local polysilicon line width control for a line 10 m wide,
manufactured with process A, is ±0.02 m for a 1-sigma distribution.
•
The global (model level) polysilicon line-width control is much wider; use
0.1 m for this example.
3. The global oxide thickness is 200 angstroms with a ±5 angstrom variation at
1 sigma.
4. The cap element is square with local poly variation in both directions.
5. The cap model has two distributions:
•
poly line-width distribution
•
oxide thickness distribution.
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The effective length is:
Leff=Ldrawn - 2 ⋅
DEL
The model poly distribution is half the physical per-side values:
C1a 1 0 CMOD W=ELPOLY L=ELPOLY
C1b 1 0 CMOD W=ELPOLY L=ELPOLY
C1C 1 0 CMOD W=ELPOLY L=ELPOLY
C1D 1 0 CMOD W=ELPOLY L=ELPOLY
$ 10U POLYWIDTH,0.05U=1SIGMA
$ CAP MODEL USES 2*MODPOLY .05u= 1 sigma
$ 5angstrom oxide thickness AT 1SIGMA
.PARAM ELPOLY=AGAUSS(10U,0.02U,1)
+ MODPOLY=AGAUSS(0,.05U,1)
+ POLYCAP=AGAUSS(200e-10,5e-10,1)
.MODEL CMOD C THICK=POLYCAP DEL=MODPOLY
Electrical Approach:
The electrical approach assumes no physical interpretation, but requires a local
(element) distribution and a global (model) distribution. In this example:
■
You can match the capacitors to ±1% for the 2-sigma population.
■
The process can maintain a ±10% variation from run to run for a 2-sigma
distribution.
C1a 1 0 CMOD SCALE=ELCAP
C1b 1 0 CMOD SCALE=ELCAP
C1C 1 0 CMOD SCALE=ELCAP
C1D 1 0 CMOD SCALE=ELCAP
.PARAM ELCAP=Gauss(1,.01,2) $ 1% at 2 sigma
+ MODCAP=Gauss(.25p,.1,2) $10% at 2 sigma
.MODEL CMOD C CAP=MODCAP
Worst Case and Monte Carlo Sweep Example
The following example measures the delay and the power consumption of two
inverters. Additional inverters buffer the input and load the output.
This netlist contains commands for two sets of transient analysis: parameter
sweep from -3 to +3-sigma, and a Monte Carlo analysis. It creates one set of
output files (mt0 and tr0) for the sigma sweep, and one set (mt1 and tr1) for
Monte Carlo.
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$ inv.sp sweep mosfet -3 sigma to +3 sigma, use measure output
.param vref=2.5 sigma=0
.global 1
vcc 1 0 5.0
vin in 0 pwl 0,0 0.2n,5
x1 in 2 inv
x2 2 3 inv
x3 3 out inv
x4 out 4 inv
.macro inv in out
mn out in 0 0 nch w=10u l=1u
mp out in 1 1 pch w=10u l=1u
.eom
.param mult1=1
+ polycd=agauss(0,0.06u,1)
xl='polycd-sigma*0.06u'
+ nactcd=agauss(0,0.3u,1) xwn='nactcd+sigma*0.3u'
+ pactcd=agauss(0,0.3u,1) xwp='pactcd+sigma*0.3u'
+ toxcd=agauss(200,10,1)
tox='toxcd-sigma*10'
+ vtoncd=agauss(0,0.05v,1) delvton='vtoncd-sigma*0.05'
+ vtopcd=agauss(0,0.05v,1) delvtop='vtoncd+sigma*0.05'
+ rshncd=agauss(50,8,1)
rshn='rshncd-sigma*8'
+ rshpcd=agauss(150,20,1)
rshp='rshpcd-sigma*20'
* level=28 example model
.model nch nmos
+ level=28 lmlt=mult1 wmlt=mult1 wref=22u lref=4.4u
+ xl=xl xw=xwn tox=tox delvto=delvton rsh=rshn
...
.model pch pmos
+ level=28 lmlt=mult1 wmlt=mult1 wref=22u lref=4.4u
+ xl=xl xw=xwp tox=tox delvto=delvtop rsh=rshp
+ ld=0.08u wd=0.2u acm=2 ldif=0 hdif=2.5u
+ rs=0 rd=0 rdc=0 rsc=0 rsh=rshp js=3e-04 jsw=9e-10
...
* transient with sweep
.tran 20p 1.0n
sweep sigma -3 3 .5
.meas s_delay trig v(2) val=vref fall=1
+
targ v(out) val=vref fall=1
.meas s_power rms power
* transient with Monte Carlo
.tran 20p 1.0n
sweep monte=100
.meas m_delay trig v(2) val=vref fall=1
+
targ v(out) val=vref fall=1
.meas m_power rms power
.probe tran v(in) v(1) v(2) v(3) v(4)
.end
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Transient Sigma Sweep Results
The plot in Figure 42 shows the family of transient analysis curves for the
transient sweep of the sigma parameter from -3 to +3 from the file inv.tr0. In the
sweep, HSPICE RF uses the values of sigma to update the skew parameters,
which in turn modify the actual NMOS and PMOS models.
Operating-Point Results in Transient Analysis
If you want to get OP results after every Monte Carlo simulation in transient
analysis, you can add the option opfile to the netlist. OP results will all output
to the file *.dp0.
Figure 42
Sweep of Skew Parameters from -3 Sigma to +3 Sigma
To view the measured results, plot the inv.mt0 output file. The plot in Figure 43
shows the measured pair delay and the total dissipative power, as a function of
the parameter sigma.
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Figure 43
Sweep MOS Inverter, Pair Delay and Power: -3 Sigma to 3 Sigma
Monte Carlo Results
This section describes the output of the Monte Carlo analysis in HSPICE RF.
The plot in Figure 44 shows that the relationship between TOX against XL
(polysilicon width=transistor length)) is completely random, as set up in the
input file.
To generate this plot in CosmosScope:
1. Read in the file inv.mt1.
2. Open the Calculator, select TOX (left mouse button), transfer to calculator
(middle mouse button), and then select and transfer XL.
3. On the WAVE pulldown in the calculator, select f(x), and then click the plot
icon.
4. Using the right mouse button on the plotted waveform, select Attributes to
change from the line plot to symbols.
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Figure 44
Scatter Plot, XL and TOX
The next graph (see Figure 45) is a standard scatter plot showing the
measured delay for the inverter pair against the Monte Carlo index number.
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Figure 45
Scatter Plot of Inverter Pair Delay
If a particular result looks interesting; for example, if the simulation 68 (monte
carlo index=68) produces the smallest delay, then you can obtain the Monte
Carlo parameters for that simulation.
*** monte carlo index =
68 ***
MONTE CARLO PARAMETER DEFINITIONS
polycd xl
= -1.6245E-07
nactcd xwn
= 3.4997E-08
pactcd xwp
= 3.6255E-08
toxcd
tox
=
191.0
vtoncd delvton
= -2.2821E-02
delvtop
= 4.1776E-02
vtopcd
rshncd rshn
=
45.16
rshpcd rshp
=
166.2
m_delay= 1.7929E-10 targ= 3.4539E-10
m_power= 6.6384E-03 from= 0.0000E+00
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to=
1.6610E-10
1.0000E-09
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Worst Case and Monte Carlo Sweep Example
In the preceding listing, the m_delay value of 1.79e-10 seconds is the fastest
pair delay. You can also examine the Monte Carlo parameters that produced
this result.
The information on shortest delay and so forth is also available from the
statistics section at the end of the output listing. While this information is useful
to determine whether the circuit meets specification, it is often desirable to
understand the relationship of the parameters to circuit performance. Plotting
the results against the Monte Carlo index number does not help for this
purpose. You need to generate plots that display a Monte Carlo result as a
function of a parameter. For example, Figure 46 shows the inverter pair delay to
channel as a function of poly width, which relates directly to device length.
Figure 46
Delay as a function of Poly width (XL)
Figure 47 shows the pair delay against the TOX parameter. The scatter plot
shows no obvious dependence, which means that the effect of TOX is much
smaller than XL. To explore this in more detail, set the XL skew parameter to a
constant and run a simulation.
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Figure 47
Sensitivity of Delay with TOX
The plot in Figure 48 overlays the skew result with the ones from Monte Carlo.
The skew simulation traverses the design space with all parameters changing
in parallel and then produces a relationship between power and delay, which
shows as a single line. Monte Carlo exercises a variety of independent
parameter combinations, and shows that there is no simple relationship
between the two results. Since the distributions were defined as Gaussian in
the netlist, parameter values close to the nominal are more often exercised
than the ones far away. With the relatively small number of samples, the chance
of hitting a combination at the extremes is very small. In other words, designing
for 3-sigma extreme for every parameter is probably not a good solution from
the point of view of economy.
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Figure 48
Superimposing Sigma Sweep Over Monte Carlo
Figure 49 superimposes the required part grades for product sales onto the
Monte Carlo plot. This example uses a 250 ps delay and 6.0 mW power
dissipation to determine the four binning grades.
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Figure 49
Speed/Power Yield Estimation
Sorting the results from inv.mt1 yields:
■
Bin1 - 18%
■
Bin2 - 30%
■
Bin3 - 31%
■
Bin4 - 21%
If this circuit is representative of the entire chip, then the present yield should
be 18% for the premium Bin 1 parts, assuming variations in process
parameters as specified in the netlist. Of course this example only shows the
principle on how to analyze the Monte Carlo results; there is no market for a
device with two of these inverters.
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Simulating the Effects of Global and Local Variations with Monte Carlo
Simulating the Effects of Global and Local Variations with Monte Carlo
Monte Carlo analysis is dependent on a method to describe variability. Four
different approaches are available in HSPICE RF:
■
specify distributions on parameters and apply these to instance parameters
■
specify distributions on parameters and apply these to model parameters
■
specify distributions on model parameters using DEV/LOT construct
These three methods are still supported in HSPICE RF.
In the following sections, the three methods are described. The description
relies on test cases, which can be found in the tar file monte_test.tar in
directory $<installdir>/demo/hspice/apps.
Variations Specified on Geometrical Instance Parameters
This method consists of defining parameters with variation using the
distribution functions UNIF, AUINF, GAUSS, AGAUSS, and LIMIT. These
parameters are then used to generate dependent parameters or in the place of
instance parameters. In a Monte Carlo simulation, at the beginning of each
sample, new random values are calculated for these parameters. For each
reference, a new random value is generated; however, no new value is
generated for a derived parameter. Therefore, it is possible to apply
independent variations to parameters of different devices, as well as the same
variation to parameters of a group of devices. Parameters that describe
distributions can be used in expressions, thus it is possible to create
combinations of variations (correlations).
These concepts are best explained with circuit examples. In the three following
examples, variation is defined on the width of a physical resistor, which has a
model. If this device was a polysilicon resistor for example, then the variations
describe essentially the effects of photoresist exposure and etching on the
width of the poly layer.
■
388
test1.sp has a distribution parameter defined called globw. A parameter
called globwidth is assigned the value of globw. The parameter globwidth is
assigned a different random value for each Monte Carlo sample. The
parameter globwidth is used to define the width of the physical resistors r1,
r2, r3, and r4, with model “resistor”. Since parameter globwidth does not
have its own distribution defined, but rather gets its value from the parameter
globw, the value for globwidth is the same wherever it is used; thus the
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resistors have the same width for each Monte Carlo sample, and therefore
the same resistance. When plotting the simulation results v1, v2, v3, and v4
from the .meas file, the waveforms overlay perfectly. This type of setup is
typically used to model global variations, which means variations that affect
all devices the same way.
■
test2.sp has a distribution parameter defined called locwidth. This
parameter is used to define the width of the physical resistors r1, r2, r3, and
r4, with model “resistor”. Since the parameter has its own distribution
defined, its value will be different for each reference, and of course for each
Monte Carlo sample. Therefore, the resistors will always have different
values, and the voltages will be different. This type of setup is typically used
to model local variations, which means variations that affect devices in a
different way.
■
test3.sp has two kinds of distributions defined: globw/globwidth as in the first
example, and locwidth as in the second example. The sum of the two is used
to define the width of the resistors. Therefore, the resistors will always have
different widths: a common variation due to globwidth and a separate
variation due to locwidth. In the example, the distribution for locwidth was
chosen as narrower than for globwidth. When overlaying the measurement
results, the large common variation can easily be seen; however, all
voltages are different.
In summary, each reference to a parameter with a specified distribution causes
a new random variable to be generated for each Monte Carlo sample. When
referencing the parameter on an instance, the effect of a local variation is
created. When referencing the parameter on an expression for a second
parameter and using the second parameter on an instance, then the effect of a
global variation is created.
Variations Specified in the Context of Subcircuits
The concept explained in the previous section applies also to subcircuits as
instances, and instances within subcircuits. Here we again use the example of
a physical resistor, with variation of its width.
■
test4.sp uses a subcircuit for each resistor instead of the top-level resistors
in test3.sp. On each subcircuit, a parameter “width” is assigned a value by
an expression, which is the same for all of them. This value is then passed
into the subcircuit and the resistor width gets this value. Because the
expression is the same for all subcircuits, the value of parameter “width” will
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be the same for all subcircuits, thus it expresses a global variation.
Therefore all resistors have the same width, and the terminal voltages are
the same.
■
In test5.sp, if a different “width” is used for the subcircuits, then the
expressions are treated separately, get local variation assigned, and
different values are passed into the subcircuit. In test5.sp, the differences
inside of the expressions are kept numerically very small, thus the
differences from the different values of “locwidth” are dominant and the
results look almost identical to the ones from test3.sp.
■
In test6.sp, the resistor width is assigned inside of the subcircuit. The
variations get picked up from the top level. Because each subcircuit is a
separate entity, the parameter “w” is treated as a separate reference, thus
each resistor will have its own value, partly defined through the common
value of “globwidth” and partly through the separate value of “locwidth”.
■
test7.sp has two resistors in the subcircuit. Each device in each subcircuit
has a separate reference to the variation, therefore each device gets its own
value.
■
In test8.sp, the variation definition for “locwidth” has been moved from the
top level into the subcircuit. Each resistor has a common global variation
and its own local variation.
■
test9.sp assigns the top level variation to a local parameter, which in turn is
applied to the width definition of the resistor. This happens independently
within each subcircuit, thus we end up with the same values for the resistor
pair in each subcircuit, but different values for the different pairs. This
technique can be applied to long resistors when a middle terminal is
required for connecting capacitance to the substrate. The resulting two
resistor pieces will have the same resistance, but it will be different from
other resistor pairs.
In summary, each subcircuit has its own parameter space, therefore it is
possible to put groups of identical components into a subcircuit, and within
each group all devices have the same parameter values, but between the
groups, parameters are different. When specifying variations on these
parameters, the effects of local variations between the groups are created.
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Variations on a Model Parameter Using a Local Model in
Subcircuit
If a model is specified within a subcircuit, then the specified parameter values
apply only to the devices in the same subcircuit. Therefore, it is possible to
calculate the value of a model parameter within the subcircuit; for example, as
a function of geometry information.
When specifying variations on these parameters, the effects of local variations
between subcircuits are created. If this method is used at the extreme with one
device per subcircuit, then each device has its own model. This approach leads
to a substantial overhead in the simulator and is therefore not recommended.
Indirect Variations on a Model Parameter
In sections Variations Specified on Geometrical Instance Parameters and
Variations Specified in the Context of Subcircuits, variations on geometrical
parameters were presented. If we want to specify variations on a model
parameter; for example, the threshold of a MOS device, then the approach
explained in the previous section with one model per device in a subcircuit
could be used. However, this is impractical because the netlist needs to be
created to call each device as a subcircuit, and because of the overhead. Since
variations are of interest only on a few model parameters, an indirect method of
varying model parameters can be used. Some special instance parameters are
available for this purpose. For example, for MOS devices, the parameter delvt0
defines a shift in threshold.
Referencing a parameter with a distribution as value for delvt0 creates the
effect of local threshold variations. A significant number of parameters of this
type are available in HSPICE RF for BSIM3 and BSIM4 models. The variations
can be tailored for each device depending on its size for example. A
disadvantage of this method is that the netlist needs to be parameterized
properly to get the correct variations. The process of preparing a basic netlist
for Monte Carlo simulations with this approach is tedious and error prone,
therefore it is best handled with scripts.
Bsim3 supports the following instance parameters:
L, w, ad, as, pd, ps, nrd, nrs, rdc, rsc, off, ic, dtemp, delvto, geo, sa, sb, sd, nf,
stimod, sa1, sa2, sa3, sa4, sa5, sa6, sa7, sa8, sa9, sa10, sb1, sb2, sb3, sb4,
sb5, sb6, sb7, sb8, sb9, sb10, sw1, sw2, sw3, sw4, sw5, sw6, sw7, sw8, sw9,
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sw10, mulu0, mulua, mulub, tnodeout, rth0, cth0, deltox, delk1, delnfct, and
acnqsmod.
Bsim4 supports the following instance parameters:
L, w, ad, as, pd, ps, nrd, nrs, rdc, rsc, off, ic, dtemp, delvto, geo, rbsb, rbdb,
rbpb, rbps, rbpd, trnqsmod, acnqsmod, rbodymod, rgatemod, geomod,
rgeomod, nf, min, mulu0, delk1, delnfct, deltox, sa, sb, sd, stimod, sa1, sa2,
sa3, sa4, sa5, sa6, sa7, sa8, sa9, sa10, sb1, sb2, sb3, sb4, sb5, sb6, sb7, sb8,
sb9, sb10, sw1, sw2, sw3, sw4, sw5, sw6, sw7, sw8, sw9, sw10, xgw, ngcon,
sca, scb, scc, sc, delk2, delxj, mulngate, delrsh, delrshg, dellpe0, deldvt0, and
mulvsat. (The mulvsat instance parameter is supplied to facilitate efficient
modeling of mismatch, local variation, and mechanical stress and proximity
effects.)
Variations Specified on Model Parameters
In this section, we investigate the method of specifying distributions on
parameters and using these parameters to define values of model parameters.
With this approach, the netlist does not have to be parameterized. The
modmonte option can be used to distinguish between global variations (all
devices of a particular model have the same parameter set) or local variations
(every device has a unique random value for the specified parameters).
■
test10.sp shows a simple case where the model parameter for sheet
resistivity is assigned a distribution defined on the parameter rsheet. The
results show that all resistors have the same value for each Monte Carlo
sample, but a different one for different samples. This setup is useful for
studying global variations.
■
test11.sp has .option modmonte=1 added. Now every resistor has a
different value.
Note that .option modmonte has no effect on any other approach presented
here.
In summary, assigning parameters with specified distributions to model
parameters allows for investigating the effects of global or local variations, but
not both. The possibility of selecting one or the other with a simple option is
misleading in the sense that the underlying definitions for global and local
variations are not the same for a realistic semiconductor technology.
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Variations Specified Using DEV and LOT
The two limitations of the approach described in section Variations Specified on
Model Parameters are resolved in this method by specifying global and local
variations directly on a model parameter with the syntax:
parameterName=parameterValue LOT/distribution LotDist
+ DEV/distribution DevDist
Where,
LOT keyword for global distribution
DEV keyword for local distribution
distribution is as explained in section Variations Specified on Geometrical
Instance Parameters
LotDist, DevDist characteristic number for the distribution. 3-sigma
value for Gaussian distributions.
■
test12.sp has large global and small local variation, similar to the setup in
the file test3.sp The result shows four different curves, with a large common
part and small separate parts. The amount of variation defined in the two
files is the same. The curves look different from the test3.sp results,
because different random sequences are used. However the statistical
results (sigma) converge for a large number of samples.
There is no option available to select only local or only global variations. This
can be an obstacle if the file is read-only or encrypted.
Combinations of Variation Specifications
Specifying distributions on parameters and applying them to model parameters
can be used on some models and the DEV/LOT approach on others in the same
simulation.
■
test13.sp has DEV/LOT specified for model res1, and the parameter “width”
for model res2. The values for the resistors with model res1 are different,
and the values for resistors with model res2 are the same.
■
test14.sp is similar to test7.sp and has modmonte=1 specified. All four
resistors have different values. However, note that in reality, the sigma for
width would be different when simulating local or global variations.
■
test15.sp has instance parameter variations specified on two resistors and
DEV/LOT on two others. From the waveforms, v3 and v4 form a first pair, and
v1 and v2 a second pair.
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It is also possible to mix variations on instance parameters and model
parameters in the same setup.
■
test16.sp has small instance parameter variations specified on width and
relatively large model parameter variations on the sheet resistivity, rsh. The
results show four different waveforms, with a common behavior.
■
test17.sp shows instance and model parameter variations as in the previous
test case, but .option modmonte is set to 1, thus the model variations
affect every device in a different way. The results show completely
independent behavior of all four resistors.
If an instance parameter or instance parameter variations and model parameter
variations are specified on the same parameter, then the instance parameter
always overrides the model parameter. Because only few parameters can be
used in both domains, this case is rather seldom, but it needs to be considered
to avoid unexpected results.
■
test18.sp has model variation specified on width with a parameter. Two
resistors have width also defined on instance. The resistors with instance
parameter do not vary at all. The other two resistors vary independently, as
expected because .option modmonte is set to 1.
■
test19.sp is similar to test18.sp with .option modmonte set to 0. The two
resistors that do not have width defined on the instance line vary together.
■
test20.sp has DEV/LOT specified. Instance parameters override variations
on selected resistors.
The DEV/LOT approach has no mechanism to describe variation as a function
of an element parameter.
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15
Using HSPICE with HSPICE RF
Describes how various analysis features differ in HSPICE RF as compared to
standard HSPICE.
This first section of this chapter describes topics related to transient analysis
and the other section describe other differences between HSPICE and
HSPICE RF.
These topics are covered in the following sections
■
RF Numerical Integration Algorithm Control
■
RF Transient Analysis Accuracy Control
■
RF Transient Analysis Output File Formats
■
Compressing Analog Files
RF Numerical Integration Algorithm Control
In HSPICE RF, you can select either the Backward-Euler or Trapezoidal
integration algorithm. Each of these algorithms has its own advantages and
disadvantages for specific circuit types. For pre-charging simulation or timing
critical simulation, the Trapezoidal algorithm usually improves accuracy.
You use the SIM_ORDER option to control the amount of Backward-Euler (BE)
to mix with the Trapezoidal (TRAP) method for hybrid integration. For example,
.OPTION SIM_ORDER=x
Setting SIM_ORDER to its lowest value selects Backward-Euler integration
algorithm, and setting it to its highest value selects Trapezoidal integration.
For the syntax and description of this control option, see .OPTION
SIM_ORDER in the HSPICE Reference Manual: Commands and Control
Options.
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RF Transient Analysis Accuracy Control
RF Transient Analysis Accuracy Control
The default time step method in HSPICE RF mixes timestep algorithms
Trapezoidal and second-order Gear (Gear-2). This yields a more accurate
scheme than Trapezoidal or Backward-Euler. Also, detection of numerical
oscillations inserts fewer Backward-Euler steps than in previous HSPICE
versions.
.OPTION SIM_ACCURACY
You use the SIM_ACCURACY option to modify the size of timesteps in HSPICE
RF. For example,
.OPTION SIM_ACCURACY=<value>
A timestep is a time interval at which you evaluate a signal. HSPICE RF
discretely expresses the time continuum as a series of points. At each point or
timestep, a circuit simulator evaluates the corresponding voltage or current
value of a signal. Thus, a resulting signal waveform is a series of individual data
points; connecting these points results in a smooth curve.
You can apply different accuracy settings to different blocks or time intervals.
The syntax to set accuracy on a block, instance, or time interval is similar to the
settings used for a power supply.
Note:
An .OPTION SIM_ACCURACY takes precedence over an.OPTION
ACCURATE.
For the syntax and description of this control option, see .OPTION
SIM_ACCURACY in the HSPICE Reference Manual: Commands and Control
Options.
Algorithm Control
In HSPICE RF, you can select the Backward-Euler, Trapezoidal, Gear, or hybrid
method algorithms. Each of these algorithms has its own advantages and
disadvantages for specific circuit types. These methods have tradeoffs related
to accuracy, avoidance of numerical oscillations, and numerical damping of
circuit oscillations. For pre-charging simulation or timing critical simulations, the
Trapezoidal algorithm usually improves accuracy.
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RF Transient Analysis Accuracy Control
.OPTION METHOD
You use the METHOD option to select a numeric integration method for a
transient analysis.
HSPICE RF supports three basic timestep algorithms: Trapezoidal (TRAP),
second-order Gear (Gear-2), and Backward-Euler (BE). Backward-Euler is the
same as first-order Gear. Also, HSPICE RF supports a hybrid algorithm
(TRAPGEAR), which is a mixture of the three basic algorithms.
HSPICE RF contains an algorithm for auto-detection of numerical oscillations
commonly encountered with trapezoidal integration. If HSPICE RF detects
such oscillations, it inserts BE steps, but not more than one BE step for every
10 time steps. To turn off auto-detection, use the PURETP option.
The TRAPGEAR method, combining 90% trapezoidal with 10% Gear-2. HSPICE
RF inserts BE steps, when the simulator encounters a breakpoint, or when the
auto-detection algorithm finds numerical oscillations.
For the syntax and description of this control option, see .OPTION METHOD in
the HSPICE Reference Manual: Commands and Control Options.
.OPTION MAXORD
You use the MAXORD option to select the maximum order of integration for the
GEAR method. Either the first-order Gear (Backward-Euler), or the secondorder Gear (Gear-2) integration method.
For the syntax and description of this control option, see .OPTION MAXORD in
the HSPICE Reference Manual: Commands and Control Options.
.OPTION SIM_ORDER
You use the SIM_ORDER option to control the amount of Backward-Euler (BE)
to mix with the Trapezoidal method for hybrid integration. This option affects
time stepping when you set .OPTION METHOD to TRAP or TRAPGEAR.
For the syntax and description of this control option, see .OPTION
SIM_ORDER in the HSPICE Reference Manual: Commands and Control
Options.
.OPTION SIM_TG_THETA
You use the SIM_TG_THETA option to control the amount of Gear-2 method to
mix with trapezoidal integration for the hybrid TRAPGEAR method.
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For the syntax and description of this control option, see .OPTION
SIM_TG_THETA in the HSPICE Reference Manual: Commands and Control
Options.
.OPTION SIM_TRAP
You use the SIM_TRAP option to change the default SIM_TG_THETA to 0, so
that method=trapgear acts like METHOD=TRAP.
For the syntax and description of this control option, see .OPTION SIM_TRAP
in the HSPICE Reference Manual: Commands and Control Options.
.OPTION PURETP
You use the PURETP option to turn off insertion of Backward-Euler (BE) steps
due to auto-detection of numerical oscillations.
For the syntax and description of this control option, see .OPTION PURETP in
the HSPICE Reference Manual: Commands and Control Options.
.OPTION SIM_OSC_DETECT_TOL
You use the SIM_OSC_DETECT_TOL option to specify the tolerance for
detecting numerical oscillations. If HSPICE RF detects numerical oscillations, it
inserts Backward-Euler (BE) steps. Smaller values of this tolerance result in
fewer BE steps.
For the syntax and description of this control option, see .OPTION
SIM_OSC_DETECT_TOL in the HSPICE Reference Manual: Commands and
Control Options.
RF Transient Analysis Output File Formats
The default output format for transient analysis in HSPICE RF is the same as in
HSPICE: the .tr0 file format. See Transient Analysis in the HSPICE User Guide:
Simulation and Analysis. HSPICE RF supports these output formats, which are
described in this section:
398
■
Tabulated Data Output
■
WDB Output Format
■
NW Output Format
■
VCD Output Format
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■
turboWave Output Format (tw)
■
Undertow Output Format (ut)
■
CSDF Output Format
If your netlist includes an unsupported output format, HSPICE RF prints a
warning message, indicating that the selected format is unsupported. HSPICE
RF then automatically defaults the output to TR0 format.
Tabulated Data Output
HSPICE RF outputs all analog waveforms specified in a .PRINT statement.
HSPICE RF saves these waveforms as ASCII tabulated data, into a file with
the .PRINT extension.
To display waveforms graphically, CosmosScope can directly read the
tabulated data. For more information about CosmosScope, see the
CosmosScope User’s Guide and Reference.
Note:
Tabulated data excludes waveforms specified in .PROBE statements.
WDB Output Format
You can use the waveform database (WDB) output format in .OPTION POST. It
was developed for maximum efficiency. The output file is *.wdb#. For example,
to output to a *.wdb# file, enter:
.OPTION POST=wdba
Signals across multiple hierarchies, that map to the same node, are named
together. They also share the same waveform data.
You can also set up the database so that CosmosScope extracts one signal at
a time. This means that CosmosScope does not need to read the entire output
file to display a single waveform.
The WDB format was designed to make accessing waveform data faster and
more efficient. It is a true database so the waveform browser does not have to
load the complete waveform file for you to view a single signal. This feature is
especially useful if the size of the waveform file is several gigabytes.
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Furthermore, the WDB format is usually more compact than XP and NW
(described later in this section). However, if the NW file is already very small,
then WDB offers little advantage in size or speed.
You can compress WDB files. For additional information, see Compressing
Analog Files on page 401.
TR Output Format
HSPICE RF stores simulation results for analysis by using the CosmosScope
graphical interface method. For example, these commands output a *.tr# file in
TR format:
■
.OPTION POST=1 saves the results in binary format
■
.OPTION POST=2 saves the results in ASCII format.
NW Output Format
HSPICE RF outputs the NW format to a file with the .nw# extension. You need
a Synopsys waveform display tool to process a file in NW format. For example,
to output to a *.nw# file, enter:
.OPTION POST=nw
You can compress NW files. For additional information, see Compressing
Analog Files on page 401.
Note:
The NW format is currently not supported by Monte Carlo analysis.
VCD Output Format
To output your waveforms from HSPICE RF in VCD (Value Change Dump)
format, set the VCD option in conjunction with the .LPRINT statement. For
example,
.OPTION VCD
.LPRINT (0.5 4.5) v(0) v(2) v(6)
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Compressing Analog Files
.LPRINT Statement
You use the .LPRINT statement to produce output in VCD file format from
transient analysis. For example,
.LPRINT (v1,v2) output_varable_list
For additional information, see .LPRINT in the HSPICE Reference Manual:
Commands and Control Options.
turboWave Output Format
To use turboWave output format TW, enter:
.OPTION POST=tw
This format supports analog compression as described in Compressing Analog
Files on page 401.
Undertow Output Format
To use Veritools Undertow output format UT, enter:
.OPTION POST=ut
This format supports analog compression as described in Compressing Analog
Files on page 401.
The waveform list in UT format now displays in a hierarchical structure, rather
than one flat level as in previous versions.
CSDF Output Format
To use CSDF output format CSDF, enter:
.OPTION POST=csdf
.OPTION csdf [overrides .OPTION POST setting]
Compressing Analog Files
Analog compression eliminates unnecessary data points from a HSPICE RF
voltage or current waveform to reduce the size of the waveform file.
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Compressing Analog Files
Eliminating Voltage Datapoints
You use the SIM_DELTAV option to determine the selection criteria for HSPICE
RF voltage waveforms in WDB or NW format. For example,
.OPTION SIM_DELTAV=<value>
During simulation, HSPICE RF checks whether the value of the X signal at the
n timestep changes by more than the SIM_DELTAV option, from its previous
value at the n-1 timestep.
■
If yes, then HSPICE RF saves the new data point.
■
Otherwise, this new data point is lost.
Typically such an algorithm yields a reduced file size with minimal resolution
loss as long as you set an appropriate SIM_DELTAV value. If a value for the
SIM_DELTAV option is too large, the waveform degrades.
NW and WDB both eliminate these data points,
which are within DELTAV or DELTAI of the previous
data point, and are not ON the plotted waveform line.
Figure 50
NW retains these data
points that are ON the line,
plotting 3 segments. But
WDB eliminates these data
points, plotting only ONE
segment for this line.
Analog Compression Formats
For a additional information, see .OPTION SIM_DELTAV in the HSPICE
Reference Manual: Commands and Control Options.
Eliminating Current Datapoints
You use the SIM_DELTAI option to determine the selection criteria for HSPICE
RF current waveforms in WDB or NW format. For example,
.OPTION SIM_DELTAI=<value>
For a additional information, see .OPTION SIM_DELTAI in the HSPICE
Reference Manual: Commands and Control Options.
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16
Advanced Features
Describes how to invoke HSPICE RF and how to perform advanced tasks,
including redirecting input and output.
HSPICE RF accepts a netlist file from standard input and delivers the ASCII
text simulation results to an HTML file or to stdout. Error and warning
messages are forwarded to standard error output.
This chapter describes how to do this as well as how to invoke HSPICE RF and
redirect input and output.
These topics are covered in the following sections:
■
Creating a Configuration File
■
Using Wildcards in HSPICE RF
■
Limiting Output Data Size
■
Probing Subcircuit Currents
■
Generating Measurement Output Files
■
Optimization
■
Using CHECK Statements
■
POWER DC Analysis
■
Detecting and Reporting Surge Currents
Creating a Configuration File
You can create a configuration file, called .hspicerf, to customize your HSPICE
RF simulation. HSPICE RF first searches for .hspicerf in your current working
directory, then in your home directory as defined by $HOME. The configuration
options listed in Table 29 are available for your use.
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Table 29
Configuration File Options
Keyword
Description
Example
flush_waveform
Flushes a waveform. If you do not specify a
percentage, then the default value is 20%.
flush_waveform
percent%
ground_floating_
node
Uses .IC statements to set floating nodes in a
ground_floating_
circuit to ground. You can select three options for node 1
grounding floating nodes:
■
■
■
If set to 1, grounds only floating nodes
(gates, bulk, control nodes, non-rail bulk) that
are included in the .IC set.
If set to 2, adds unconnected terminals to this
set.
If set to 3, uses .IC statements to ground all
floating nodes, including dangling terminals.
hier_delimiter
Changes the delimiter for subcircuit hierarchies hier_delimiter /
from “.” to the specified symbol.
html
Stores all HSPICE RF output in HTML format.
htmlhspicerf
test
This example creates
a file named test.html
in the current directory.
integer_node
Removes leading zeros from node names. For
example, HSPICE RF considers 0002 and 2 to
be the same node.
integer_node
Without this keyword, 0002 and 2 are two
separate nodes.
max_waveform_size
Automatically limits the waveform file size.
■
■
■
404
max_waveform_
If the number is less than 5000, HSPICE RF size 2000000000
resets it to 2G.
If you do not set the number, HSPICE RF
uses the default, 2G.
If you do not set the line, the file size has no
limit.
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Table 29
Configuration File Options (Continued)
Keyword
Description
Example
negative_td
Allows negative time delay input in pwl
(piecewise linear with repeat), pl (piecewise
linear), exp (exponential, rising time delay only),
sin (damped sinusoidal), pulse (trapezoidal
pulse), and am (amplitude modulation) formats.
If you do not set
negative_td, a
negative time delay
defaults to zero.
port_element_
voltage_ matchload
Allows the alternate Port element definition. A
Port element consists of a voltage source in
series with a resistor.
port_element_
voltage_
matchload
For the explanation that follows, let the userspecified DC, AC, or transient value of the Port
element be V, and let the voltage across the
overall port element be Vp.
By default, HSPICE RF will set the internal
voltage source value to V. The value of Vp will be
lower than V, depending on the internal
impedance and the network's input impedance.
With the alternate definition, the internal voltage
source value is adjusted to 2*V, so that Vp=V
when the Port element's impedance is matched
with the network input impedance. The actual
value of Vp will still depend on the port and
network impedances.
rcxt_divider
Defines the hierarchy delimiter in the active
nodes file in RCXT format.
skip_nrd_nrs
Directs HSPICE RF to consider transistors with skip_nrd_nrs
matching geometries (except for NRD and NRS)
as identical for pre-characterization purposes.
unit_atto
Activates detection of the “atto.” unit. Otherwise, unit_atto
HSPICE RF assumes that “a” represents
“amperes.”
v_supply
Changes the default voltage supply range for
characterization.
v_supply 3
wildcard_left_range
Begins range expression.
wildcard_left_
range [
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Table 29
Configuration File Options (Continued)
Keyword
Description
Example
wildcard_match_all
Matches any group of characters.
wildcard_match_
all *
wildcard_match_one
Matches any single character.
wildcard_match_
one ?
wildcard_right_range Ends range expression.
wildcard_right_
range ]
Note:
For more information about wildcards, see Using Wildcards in HSPICE RF
on page 406.
Inserting Comments in a .hspice File
To insert comments into your .hspicerf file, include a number sign character (#)
as the first character in a line. For example, this configuration file shows how to
use comments in a .hspicerf file:
# sample configuration file
# the next line of code changes the delimiter
# for subcircuit hierarchies from "," to "^"
hier_delimiter ^
# the next line of code matches any groups of "*" characters
wildcard_match_all *
# the next line of code matches one "?" character
wildcard_match_one ?
# the next line of code begins the range expression with
# the "[" character
wildcard_left_range [
# the next line of code ends the range expression with
# the "]" character
wildcard_right_range ]
Using Wildcards in HSPICE RF
You can use wildcards to match node names. HSPICE RF uses wildcards
somewhat differently than standard HSPICE.
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Limiting Output Data Size
Before using wildcards, you must define the wildcard configuration in
a .hspicerf file. For example, you can define the following wildcards in
a .hspicerf file:
file .hspicerf
wildcard_match_one
wildcard_match_all
wildcard_left_range
wildcard_right_range
?
*
[
]
The .PRINT, .PROBE, .LPRINT, and .CHECK statements support wildcards in
HSPICE RF.
For more information about using wildcards in an HSPICE configuration file,
see Using Wildcards in PRINT and PROBE Statements in the HSPICE User
Guide: Simulation and Analysis.
Limiting Output Data Size
For multi-million transistor simulations, an unrestricted waveform file can grow
to several gigabytes in size. The file becomes unreadable in some waveform
viewers, and requires excessive space on the hard drive.
This section describes options that limit the number of nodes output to the
waveform file to reduce the file size. HSPICE RF supports the following options
to control the output:
■
SIM_POSTTOP Option
■
SIM_POSTSKIP Option
■
SIM_POSTAT Option
■
SIM_POSTDOWN Option
■
SIM_POSTSCOPE Option
SIM_POSTTOP Option
You use the SIM_POSTTOP option to limit the data written to your waveform file
to data from only the top n level nodes. This option outputs instances up to n
levels deep. For example,
.OPTION SIM_POSTTOP=<n>
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Note:
To enable the waveform display interface, you also need the POST option.
For additional information, see .OPTION SIM_POSTTOP in the HSPICE and
HSPICE RF Command Reference.
SIM_POSTSKIP Option
You use the SIM_POSTSKIP to have the SIM_POSTTOP option skip any
instances and their children that the subckt_definition defines. For example,
.OPTION SIM_POSTSKIP=<subckt_definition>
For additional information, see .OPTION SIM_POSTSKIP in the HSPICE and
HSPICE RF Command Reference.
SIM_POSTAT Option
You use the SIM_POSTAT
option to limit the waveform output to only the nodes in the specified subcircuit
instance. For example,
.OPTION SIM_POSTAT=<instance>
This option can be used in conjunction with the SIM_POSTTOP option and
when present, has precedence over the SIM_POSTSKIP option.
For additional information, see .OPTION SIM_POSTAT in the HSPICE and
HSPICE RF Command Reference.
SIM_POSTDOWN Option
You use the SIM_POSTDOWN option to include an instance and all children of
that instance in the output. For example,
.OPTION SIM_POSTDOWN=<instance>
It can be used in conjunction with the SIM_POSTTOP option and when present,
has precedence over the SIM_POSTSKIP option.
For additional information, see .OPTION SIM_POSTDOWN in the HSPICE and
HSPICE RF Command Reference.
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Probing Subcircuit Currents
SIM_POSTSCOPE Option
You use the SIM_POSTSCOPE option to specify the signal types to probe from
within a scope. For example,
.OPTION SIM_POSTSCOPE=net|port|all
For additional information, see .OPTION SIM_POSTSCOPE in the HSPICE
and HSPICE RF Command Reference.
Probing Subcircuit Currents
To provide subcircuit power probing utilities, HSPICE RF uses the X() and X0()
extended output variables. You can use these X variables in .PROBE, .PRINT,
or .MEASURE statements.
The following syntax is for the output variable X():
X (subcircuit_node_path)
X0 (subcircuit_node_path)
subcircuit_node_path specifies the subcircuit path and the subcircuit node
name definition. The node must be either an external node in a subcircuit
definition or a global node.
X() returns the total current flowing into a subcircuit branch, including all lower
subcircuit hierarchies. X0() returns only current flowing into a subcircuit branch,
minus any current flowing into lower subcircuit hierarchies. Figure 51 on
page 410 illustrates the difference between the X() and X0 () variables.
The dotted line boxes represent subcircuits, and the black circles are the
external nodes. The X(X1.vc1) path returns the current of the X1subcircuit,
through the vc1 node, including the current to the X1.X1 and X1.X2 subcircuits
as represented by the white (black outlined) arrows. In contrast, X0(X1.vc2)
returns only the current flowing through vc2 to the top level of the X1 subcircuit
as shown by the black arrows.
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VDD1
VDD2
X0(X1.vc2)
X(X1.vc1)
vc1
X(X2.vd2)
vc2
X1
vd1
vd2
X2
X1.X1
X1.X2
Figure 51
Probing Subcircuit Currents
Example 1
In this example, the first five lines constitute the definition of the sb1 subcircuit
with external nodes named node1, node2, and clr. The line beginning with
X1 is an instance of sb1 with nodes named;
■
11 (references node1)
■
12 (references node2)
■
0 (references clr)
.subckt sb1 node1 node2 clr
* subckt elements
R1 node1 node2 1K
C1 clr node1 1U
.ends
* subcircuit instance
X1 11 12 0 sb1
.PRINT X(X1.node1) ‘X(X1.clr) + I(X1.R1)’
To find the current flowing into node 11 of the X1 subcircuit instance, this
example uses the X() variable. HSPICE RF maps node 11 to the node1
external node as shown in the first part of the .PRINT statement.
The latter half of the .PRINT statement illustrates that you can combine the
X() variable with I() variables.
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Generating Measurement Output Files
Example 2
In this example, the X() variable finds the current through the in node of the S1
subcircuit.
.subckt S1 in out
R1 in inp 1K
C1 inp 0 1u
R2 in out 1K
.PROBE X(in)
.ends
Generating Measurement Output Files
You can make all of the same measurements with the .MEASURE statement in
HSPICE RF as you can in HSPICE.
The results of the .MEASURE statements appear in a file with one of the
following filename extensions:
■
.mt# for measurements in transient analysis
■
.ms# for measurements in DC analysis
■
.ma# for measurements in AC analysis
■
.mb# for measurements in HB analysis
■
.mp# for measurements in HBNOISE analysis
For more information about .MEASURE statements, see HSPICE and HSPICE
RF Netlist Commands in the HSPICE Reference Manual: Commands and
Control Options.
Optimization
Like HSPICE, HSPICE RF employs an incremental optimization technique.
This technique solves the DC parameters first, then the AC parameters, and
finally the transient parameters.
To perform optimization, create an input netlist file that specifies:
■
Optimization parameters with upper and lower boundary values along with
an initial guess.
■
An AC, DC, TRAN, HB, or HBOSC optimization statement.
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Optimization
■
An optimization model statement.
■
Optimization measurement statements for optimization parameters.
If you provide the input netlist file, optimization specifications, limits, and initial
guess, then the optimizer reiterates the simulation until it finds an optimized
solution.
Usage Notes and Examples
■
Optimization works for TRAN, AC, DC, HB, HBOSC, and HBAC analyses.
■
You can add the GOAL options in every meaningful .MEASURE statement,
like FIND-WHEN, FIND-AT, and so forth.
■
A data sweep is not required to be defined in the .HB statement for HB
optimization to use the measured result from .MEASURE HBNOISE,
PHASENOISE, or HBTRAN statements. Therefore, parameter sweep is not
supported for this type of optimization.
■
Optimize multiple parameters with multiple goals by selecting .MODEL OPT
LEVEL=0 (modified Lavenberg-Marquardt method).
■
Optimize single parameters in single measurement situations by selecting
.MODEL OPT LEVEL=1 (bisection method).
■
Examples
•
Setting optimization parameters
.param W=opt1(231u, 100u, 800u)
.param Rs=opt1(10,8,20)
•
Optimization analysis statement
.HB tones=2.25g 2.5g nharms=6,3
+ sweep Pin:dbm -30 0 2
+ sweep optimize = opt1
+ results = gain $measure result to tune the parameters
+ model= optmod1
•
Selecting an optimization model
.model optmod1 opt level=1 $Bisection method
+ itropt=40 relin=1e-4 relout=1e-6 $ accuracy settings
•
Measurement statements to tune the optimization parameters
.measure HB vif find vdb(if+)[-1,1] at 10e-6
.measure HB vrf find vdb(rf+)[0,1] at 10e-6
.measure HB gain=param('vif-vrf') goal=-2
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Optimization
•
Measurement statement to find the fundamental frequency from HB
analysis
.measure HB frequency_max FIND ‘HERTZ[1]’ at=0
Optimizing AC, DC. and TRAN Analyses
The HSPICE syntax is followed for optimizing AC, DC. and TRAN analyses.
The required statements are:
■
Optimization .PARAM statement
.PARAM <ParamName>=OPTxxx(<Init>,<LoLim>,<HiLim>)
■
Optimizing .TRAN statement
.TRAN tincr1 tstop1 <tincr2 tstop2 ... tincrN tstopN>
+ SWEEP OPTIMIZE=OPTxxx RESTULTS=measname MODEL=optmod
■
Optimizing .MODEL statement
.MODEL mname OPT LEVEL=[0|1]
Where:
•
0 specifies the Modified Levenberg-Marquardt method. You would use
this setting with multiple optimization parameters and goals.
•
1 specifies the Bisection method. You would use this setting with one
optimization parameter.
Optimizing HB Analysis
There are two types of optimizations with HB analyses:
■
Optimization with only HB measurements
■
Optimization with HBNOISE, PHASENOISE, or HBTRAN measurements
Optimization With HB Measurements
The required statements are:
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Optimization
■
Analysis statement
.HB TONES=<f1>[<f2> ... <fn>] <NHARMS=<h1>,<h2> ... <hn>>
+ SWEEP parameter_sweep OPTIMIZE=OPTxxx RESULT=measname
+ MODEL=mname
■
Measure statement
.MEASURE HB measname FIND out_var1 AT=val GOAL=val
Optimization With HBNOISE, PHASENOISE, or HBTRAN
Measurements
The required statements are:
■
Analysis statement
.HB TONES=<f1>[<f2> ... <fn>] <NHARMS=<h1>,<h2> ... <hn>>
+ SWEEP OPTIMIZE=OPTxxx RESULT=measname MODEL=mname
For example,
.HBOSC tones=1g nharms = 5 optimize = opt1
+ result = y1, y2 model = m1
.model m1 opt level=0
.PHASENOISE dec 1 1k 1g
.meas phasenoise y1 find phnoise at 10k goal = -150dbc
.meas phasenoise y2 RMSJITTER phnoise units = sec goal =
1.0e-12
■
Measure statement
.MEASURE HBNOISE measname FIND out_var1 AT=val GOAL=val
.MEASURE PHASENOISE measname FIND out_var1 AT=val
+ GOAL=val
.MEASURE HBTRAN measname FIND out_var1 AT=val GOAL=val
Optimizing HBOSC Analysis
There are two types of optimizations with .HBOSC analyses:
■
Optimization with only HB measurements
■
Optimization with HBNOISE, PHASENOISE, or HBTRAN measurements
Optimization With HB Measurements
The required statements are:
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Using CHECK Statements
■
Analysis statement
.HBOSC TONES=<f1>[<f2> ... <fn>] <NHARMS=<h1>,<h2> ... <hn>>
+ SWEEP parameter_sweep OPTIMIZE=OPTxxx RESULT=measname
+ MODEL=mname
■
Measure statement
.MEASURE HB measname FIND out_var1 AT=val GOAL=val
Optimization With HBNOISE, PHASENOISE, or HBTRAN
Measurements
The required statements are:
■
Analysis statement
.HBOSC TONES=<f1>[<f2> ... <fn>] <NHARMS=<h1>,<h2> ... <hn>>
+ SWEEP OPTIMIZE=OPTxxx RESULT=measname MODEL=mname
For example,
.HBOSC tones=1g nharms = 5 sweep x 1 5 1 optimize = opt1
+ result = y1, y2 model = m1
.model m1 opt level=0
.PHASENOISE dec 1 1k 1g
.meas phasenoise y1 find phnoise at 10k goal = -150dbc
.meas phasenoise y2 RMSJITTER phnoise units = sec goal =
1.0e-12
Measure statement—
.MEASURE HBNOISE measname FIND out_var1 AT=val GOAL=val
.MEASURE PHASENOISE measname FIND out_var1 AT=val
+ GOAL=val
.MEASURE HBTRAN measname FIND out_var1 AT=val GOAL=val
Optimization with HBNOISE, PHASENOISE or HBTRAN measurements must
not be used in combination with HB measurement optimization as shown in
Optimization With HB Measurements.
Using CHECK Statements
The CHECK statements in HSPICE RF offer the following instrumentation:
■
Setting Global Hi/Lo Levels
■
Slew, Rise, and Fall Conditions
■
Edge Timing Verification
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■
Setup and Hold Verification
■
IR Drop Detection
The results of these statements appear in a file with an .err extension. To
prevent creating unwieldy files, HSPICE RF reports only the first 10 violations
for a particular check in the .err file.
Setting Global Hi/Lo Levels
You use the .CHECK GLOBAL_LEVEL statement to globally set the desired
high and low definitions for all CHECK statements. For example,
.CHECK GLOBAL_LEVEL (hi lo hi_th lo_th)
Values for hi, lo, and the thresholds are defined by using this statement.
For syntax and description of this statement, see .CHECK GLOBAL_LEVEL in
the HSPICE Reference Manual: Commands and Control Options.
Slew, Rise, and Fall Conditions
You use the .CHECK SLEW statement to verify that a slew rate occurs within
the specified window of time. For example,
.CHECK SLEW (min max) node1 <node2 ...> <(hi lo hi_th lo_th)
3.3
2.6
0.7
0.0
1ns < t < 3ns
Figure 52
SLEW Example
For syntax and description of this statement, see .CHECK SLEW in the
HSPICE Reference Manual: Commands and Control Options.
You use the .CHECK RISE statement to verify that a rise time occurs within the
specified window of time. For example,
416
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Chapter 16: Advanced Features
Using CHECK Statements
.CHECK RISE (min max) node1 <node2 ...> <(hi lo hi_th lo_th)>
HI
HI_thresh
LO_thresh
LO
1.5 ns < t < 2.2 ns
Figure 53
RISE Time Example
For syntax and description of this statement, see .CHECK RISE in the HSPICE
and HSPICE RF Command Reference.
You use the .CHECK FALL statement to verify that a fall time occurs within the
specified window of time. For example,
.CHECK FALL (min max) node1 <node2 ...> <(hi lo hi_th lo_th)>
For syntax and description of this statement, see .CHECK FALL in the HSPICE
Reference Manual: Commands and Control Options.
Edge Timing Verification
The edge condition verifies that a triggering event provokes an appropriate
RISE or FALL action, within the specified time window. You use the .CHECK
EDGE statement to verify this condition. For example,
.CHECK EDGE (ref RISE|FALL min max RISE|FALL)
+ node1 < node2 . . . > < (hi lo hi_th low_th) >
HSPICE User Guide: RF Analysis
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Chapter 16: Advanced Features
Using CHECK Statements
voutA
CLK
HI
HI_thresh
LO_thresh
LO
1ns < t < 3 ns
Figure 54
EDGE Example
For syntax and description of this statement, see .CHECK EDGE in the
HSPICE and HSPICE RF Command Reference.
Setup and Hold Verification
You use the .CHECK SETUP and .CHECK HOLD statements to ensure that
specified signals do not switch for a specified period of time. For example,
.CHECK SETUP (ref RISE|FALL duration RISE|FALL) node1
+< node2 . . . > < (hi lo hi_th low_th) >
.CHECK HOLD (ref RISE|FALL duration RISE|FALL) node1
+< node2 . . . > < (hi lo hi_th low_th) >
■
For a SETUP condition, this is the minimum time before the triggering event,
during which the specified nodes cannot rise or fall.
nodeA
v1
HI
HI_thresh
LO_thresh
LO
t >=2ns
Figure 55
418
SETUP Example
HSPICE User Guide: RF Analysis
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Chapter 16: Advanced Features
Using CHECK Statements
For syntax and description of this statement, see .CHECK SETUP in the
HSPICE Reference Manual: Commands and Control Options.
■
For a HOLD condition, this is minimum time required after the triggering
event, before the specified nodes can rise or fall.
vin*
nodeA
HI
HI_thresh
LO_thresh
LO
t >=2ns
Figure 56
HOLD Example
For syntax and description of this statement, see .CHECK HOLD in the
HSPICE and HSPICE RF Command Reference.
IR Drop Detection
You use the .CHECK IRDROP statement to verify that the IR drop does not
exceed, or does not fall below, a specified value for a specified duration. For
example,
.CHECK IRDROP ( volt_val time ) node1 < node2 . . . >
+ < ( hi lo hi_th low_th ) >
v1
-2 volts
t <=1ns
Figure 57
IR Drop Example
For syntax and description of this statement, see .CHECK IRDROP in the
HSPICE Reference Manual: Commands and Control Options.
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419
Chapter 16: Advanced Features
POWER DC Analysis
POWER DC Analysis
You use the .POWERDC (standby current) statement to calculate the DC
leakage current of a design hierarchy. For example,
.POWERDC <keyword> <subckt_name1...>
This statement creates a table that lists the measurements of the AVG, MAX,
and MIN values for the current of every instance in the subcircuit. This table
also lists the sum of the power of each port in the subcircuit.
You use the SIM_POWERDC_HSPICE option to increase the accuracy of
operating point (OP) calculations.
Or for even higher accuracy in operating point calculations, you use the
SIM_POWERDC_ACCURACY option.
For syntax and description of this statement and options, see .POWERDC,
.OPTION SIM_POWERDC_ACCURACY, or .OPTION
SIM_POWERDC_HSPICE in the HSPICE Reference Manual: Commands and
Control Options.
Power DC Analysis Output Format
*** Leakage Current Result ***
Subckt Name=XXX
Instance Name
Port
Max(A)
Min(A)
Avg(A)
.....
Total Power
Max(W)
Min(W)
Avg(W)
NOTE:
Power=Sum{Ii * Vi}
Subckt Name=XXX
Instance Name
Port
Max(A)
Min(A)
Avg(A)
.....
Total Power
Max(W)
Min(W)
Avg(W)
Example
.global vdd vss
.powerdc all
x1 in1 mid1 inv
x2 mid1 out1 inv
.subckt inv in out
mn out in vss vss nch
mp vdd in out vdd pch
.ends
.end
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Chapter 16: Advanced Features
POWER Analysis
(Output)
*** Leakage Current Result ***
Subckt Name=Top Level
Instance Name
Port
Max(A)
x1
in
.......
x1
out
.......
x2
in
.......
x2
out
.......
Total Power
.......
Subckt Name=inv
Instance Name
Port
Max(A)
mn
d
.......
mn
g
.......
mn
s
.......
mn
b
.......
mp
d
.......
mp
g
.......
mp
s
.......
mp
b
.......
Total Power
.......
Min(A)
Avg(A)
Min(A)
Avg(A)
POWER Analysis
The .POWER statement in HSPICE RF creates a table, which by default
contains the measurements for AVG, RMS, MAX, and MIN for every signal
specified. For example,
.POWER <signals> <REF=vname FROM=start_time TO=end_time>
By default, the scope of these measurements are set from 0 to the maximum
timepoint specified in the .TRAN statement.
For syntax and description of .POWER statement, see .POWER in the HSPICE
and HSPICE RF Command Reference.
Example 1
In this example, no simulation start and stop time is specified for the x1.in
signal, so the simulation scope for this signal runs from the start (0ps) to the
last .tran time (100ps).
.power x1.in
.tran 4ps 100ps
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Chapter 16: Advanced Features
POWER Analysis
Example 2
You can use the FROM and TO times to specify a separate measurement start
and stop time for each signal. In this example:
■
The scope for simulating the x2.in signal is from 20ps to 80ps.
■
The scope for simulating the x0.in signal is from 30ps to 70ps.
.param myendtime=80ps
.power x2.in REF=a123 from=20ps to=80ps
.power x0.in REF=abc from=30ps to=’myendtime - 10ps’
Setting Default Start and Stop Times
In addition to using FROM and TO times in a .POWER statement, you can also
use the SIM_POWERSTART and SIM_POWERSTOP options with .POWER
statements to specify default start and stop times for measuring signals during
simulation. These times apply to all signals that do not have their own defined
FROM and TO measurement times. For example,
.OPTION SIM_POWERSTART=<time>
.OPTION SIM_POWERSTOP=<time>
These options control the power measurement scope; the default is for the
entire run.
For syntax and description of these options, see .OPTION
SIM_POWERSTART or .OPTION SIM_POWERSTOP in the HSPICE and
HSPICE RF Command Reference.
Controlling Power Analysis Waveform Dumps
You use the SIM_POWERPOST option to control power analysis waveform
dumping. For example,
.OPTION SIM_POWERPOST=ON|OFF
Considering the potentially enormous number of signals, there is no waveform
dumping by default for the signals in the .POWER statement. Setting
SIM_POWERPOST=ON turns on power analysis waveform dumping.
422
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Chapter 16: Advanced Features
Detecting and Reporting Surge Currents
Detecting and Reporting Surge Currents
The .SURGE statement in HSPICE RF automatically detects and reports a
current surge that exceeds the specified surge tolerance. For example,
.SURGE surge_threshold surge_width node1 < node2 .... noden >
This statement reports any current surge that is greater than surge_threshold
for a duration of more than surge_width.
For additional information, see .SURGE in the HSPICE Reference Manual:
Commands and Control Options.
HSPICE User Guide: RF Analysis
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Chapter 16: Advanced Features
Detecting and Reporting Surge Currents
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Index
A
B
abs(x) function 85
absolute
power function 85
value function 85
.AC statement 357
accuracy control 396
acos(x) function 85
AGAUSS keyword 369
algebraic
expressions 83
algorithm
linear acceleration 349
nonlinear perturbation 233
numerical integration 395, 396
periodic AC 233
amplifier 15, 20
amplifier, IP3 24
amplitude modulation (AM) 239
AM-PM separation 239
analysis
data driven 354, 355
Monte Carlo 355, 365, 365–387
phase noise 227
statistical 358–387
Taguchi 354
temperature 354, 356
time domain steady-state 199
worst case 354, 358–387
yield 354
arccos(x) function 85
arcsin(x) function 84
arctan(x) function 85
arithmetic operators 84
ASIC libraries 76
asin(x) function 84
atan(x) function 85
AUNIF keyword 369
average deviation 355
backslash continuation character 84
Backward-Euler
algorithm 395, 396
integration 395, 396
Behavioral resistors 102
BJTs
elements, names 140
block elements 122
broadband phasenoise 234
broadband phasenoise algorithm 234
C
C Element (capacitor) 107
capacitance
manufacturing variations 376
capacitor
charge-based 107
element 104, 107
frequency-dependent 108
linear 107
cell characterization 354
charge-based capacitor 107
.CHECK EDGE statement 417
.CHECK FALL statement 417
.CHECK GLOBAL_LEVEL statement 416
.CHECK HOLD statement 418
.CHECK IRDROP statement 419
.CHECK RISE statement 416
.CHECK SETUP statement 418
.CHECK SLEW statement 416
choke elements 122
circuit description syntax 10
circuits
reusable 71
temperature 357
clock source, random jitter 170
CMOS GPS VCO 33
Colpitts oscillator 29
425
Index
D
.command
.PRINT ENV 317
command
.PROBE ENV 317
commands
hspicerf 9
PTDNOISE 282
comparing results 44
compression of input files 59
config file
hspicerf 403
configuration file 403
example 406
configuration options
flush_waveform 404
ground_floating_ node 404
hier_delimiter 404
html 404
integer_node 404
max_waveform_size 404
negative_td 405
port_element_ voltage_ matchload 405
rcxt_divider 405
unit_atto 405
v_supply 405
wildcard_left_range 405
wildcard_match_all 406
wildcard_match_one 406
wildcard_right_range 406
continuation character, parameter strings 84
cos(x) function 84
cosh(x) function 85
Cosmos-Scope 12
coupled inductor element 118
D
data-driven analysis 354, 355
db(x) function 86
DC block elements 122
.DC statement 357
DDL 75
DDLPATH environment variable 75
decibel function 86
DEFW option 92
.DEL LIB statement 70
DELVTO model parameter 359
demo files 58
426
58
demonstration files, RF 58
demonstration input files 58
Detailed Standard Parasitic Format See DSPF
deviation, average 355
device model cards 28
diodes
junction 138
models 138
polysilicon capacitor length 138
DSPF
expansion 331
file structure 325
DTEMP parameter 356, 357
E
edge condition 417
element
active
BJTs 140
diodes 137
JFETs 142
MESFETs 142
MOSFETs 144
C (capacitor) 107
identifiers 66
L (inductor) 114
markers, mutual inductors 112
passive 98
capacitors 104
inductor 109
mutual inductor 112
R (resistor) 101
statements 75
temperature 357
transmission line 124
element parameters
BJTs 140
capacitors 104
DTEMP 356
inductors 109–111
JFETs and MESFETs 142–143
linear inductors 109, 121
MOSFETs 144–146
mutual inductors, Kxxx 112
resistors 99–100
transmission lines
W Element 124–125, 126
Index
F
elements
coupled inductor 118
.END statement
missing 60
.ENV statement 314
Envelope Analysis (ENV) 313
envelope simulation 313
.ENVFFT 316
.ENVFFT statement 316
environment variables 75
.ENVOSC 315
.ENVOSC statement 315
errors
missing .END statement 60
example
configuration file 406
Monte Carlo 372, 378
worst case 378
examples, RF tutorials 15
exp(x) function 85
exponential function 85
expressions, algebraic 83
Extended output variables 409
external data files 70
F
fall time
verification 417
files
external data 70
.hl# 307
hspice.ini 75
hspicerf 403
include files 70
.ls# 312
.p2d# 312
.printhl# 307
.printls# 311
.printss# 311
.ss# 312
files, output 10
first character descriptions 65
flags 403
flush_waveform configuration option 404
format
output
DSPF 325
format, output
NW 400
WDB 399
Foster pole-residue form
E element 157
G element 157
frequency
variable 88
frequency-dependent
capacitor 108
inductor 114
resistor 103
functions
built-in 84–88
table 84
G
GAUSS
functions 373
keyword 369
parameter distribution 366
generating output 10
global parameters 89
ground_floating_ node configuration option 404
H
Harmonic Balance (HB) 175
analysis spectrum 179
equations 177
errors 195
options 181
oscillator analysis 209
output 184
syntax 178
warnings 195
.HB
for HBLIN 302
HB analysis
IP3 amplifier 24
power amplifier 20
HB_GIBBS option 190
HBAC 43, 255
errors 260, 265
example 43
output 257, 317
output data files 259
syntax 256
427
Index
I
warnings 260, 265
HBAC analysis
mixer 41
.HBLIN 300, 303
limitations 301
output syntax 306
.HBLSP 307
example 310
input syntax 309
limitations 308
output data files 307, 311
output syntax 311
.HBOSC
options 218
HBOSC analysis
Colpitts oscillator 29
VCO 33
.HBOSC statement 209
HBXF
command 291
hertz variable 88
hier_delimiter configuration option 404
hierarchical designs, flattened 70
.hl# file 307
hold time verification 418
hspice.ini file 75
hspicerf command 9
hspicerf file 403
hspicerf test 404
html configuration option 404
I
ideal transformer 121
.INCLUDE statement 70, 76, 77
individual element temperature 357
inductor
coupled 118
frequency-dependent 114
inductors
element 109
node names 109, 122
input
files
character case 60
compression 59
netlist 59
structure 70
428
table of components 70
input files demonstration 58
input files, demo examples 58
int(x) function 86
integer function 86
integer_node configuration option 404
invoking HSPICE RF 9
IR drop
checking 419
J
JFETs
elements 142
length 142
width 142
jitter
random, with clock source 170
jitter, random, clock source 170
K
keywords
DTEMP 356
MONTE 366
PAR 84
L
L Element (inductor) 114
large-signal S parameter extraction 307
LENGTH model parameter 375
.LIB
statement 70, 77
libraries
ASIC cells 76
configuring 92
creating parameters 90
DDL 75
duplicated parameter names 90
integrity 89
search 76
subcircuits 77
vendor 76
LIMIT keyword 369
LIN analysis 15
linear
acceleration 348
capacitor 107
Index
M
inductor 114
matrix reduction 348
resistor 101
linear elements
elements, linear 123
local parameters 89
log(x) function 85
log10(x) function 85
logarithm function 85
low noise amplifier 15
.LPRINT statement 401
.ls# file 312
M
manufacturing tolerances 374
max(x,y) function 86
max_waveform_size configuration option 404
mean, statistical 355
.measure 318
.MEASURE ENV command 318
.MEASURE statement parameters 83
MESFETs 142
min(x,y) function 86
mixer 41
model cards 28
model parameters
capacitance distribution 376
DELVTO 359
DTEMP 357
LENGTH 375
manufacturing tolerances 374
PHOTO 375
RSH 359
sigma deviations, worst case analysis 359
skew 358
TEMP 357
temperature analysis 357
TOX 359
TREF 355, 357
XPHOTO 375
.MODEL statement 357
models
Monte Carlo analysis 365, 371, 378
reference temperature 357
specifying 76
typical set 362
Monte Carlo
analysis 354, 355, 378–387
distribution options 369–370
Monte Carlo analaysis
operating-point results in transient analysis 380
MONTE keyword 366
MOSFETs
drain diffusion area 144
elements 144
initial conditions 145
node names 144
perimeter 145
source 145, 146
squares 145
temperature differential 145
zero-bias voltage threshold shift 146
multiply parameter 72, 99
multi-tone HB analysis
mixer 41
mutual inductor 112
N
natural log function 85
negative_td configuration option 405
netlist 70
flat 70
input files 59
schematic 70
noise
.HBNOISE 266, 274
noise parameter extraction
small-signal 307
nonlinear perturbation algorithm 233
numerical integration 395, 396
NW output format 400
O
operators 84
optimization 411
syntax 411
.OPTION
MAXORD 397
PURETP 398
SIM_ACCURACY 396
SIM_DELTAI 402
SIM_DELTAV 402
SIM_DSPF 324
429
Index
P
SIM_DSPF_ACTIVE 324, 327
SIM_DSPF_INSERROR 329
SIM_DSPF_LUMPCAPS 329
SIM_DSPF_MAX_ITER 328
SIM_DSPF_RAIL 328
SIM_DSPF_SCALEC 328
SIM_DSPF_SCALER 328
SIM_DSPF_VTOL 327
SIM_LA 324, 325, 348, 351
SIM_LA_FREQ 351
SIM_LA_MAXR 351
SIM_LA_MINC 351
SIM_LA_MINMODE 351
SIM_LA_TIME 351
SIM_LA_TOL 352
SIM_ORDER 395, 397
SIM_OSC_DETECT_TOL 398
SIM_POSTAT 408
SIM_POSTDOWN 408
SIM_POSTSCOPE 409
SIM_POSTSKIP 407, 408
SIM_POWERDC_ACCURACY 420
SIM_POWERDC_HSPICE 420
SIM_POWERPOST 422
SIM_POWERSTART 422
SIM_SPEF 324
SIM_SPEF_ACTIVE 327
SIM_SPEF_INSERROR 329
SIM_SPEF_LUMPCAPS 329
SIM_SPEF_MAX_ITER 328
SIM_SPEF_PARVALUE 329
SIM_SPEF_RAIL 328
SIM_SPEF_SCALEC 328
SIM_SPEF_SCALER 328
SIM_SPEF_VTOL 327
SIM_TG_THETA 397
SIM_TRAP 398
.OPTION HB_GIBBS 190
options, configuration file 404
oscillator
HB analysis 209
phase noise 227
oscillator example 29
output
files 10
format
DSPF 331
NW 400
430
tabulated data 399
WDB 399
generating 10
restricting 407
variables
function 87
P
.p2d# file 312
packed input files 59
PAR keyword 84
.PARAM statement 354
parameters
algebraic 83, 84
analysis 83
assignment 81
cell geometry 89
constants 82
data type 81
defaults 92
defining 79, 90
evaluation order 81
hierarchical 72, 88
inheritance 91, 92
input netlist file 69
libraries 90–92
M 72
measurement 83
optimization 89
overriding 90, 92
PARHIER option 92
passing 88–95
order 81
problems 95
Release 95.1 and earlier 95
scope 88–89, 95
simple 82
subcircuit 72
user-defined 82
PARHIER option 92
passive element 98
periodic AC algorithm 233
periodic pime-dependent noise analysis 282
phase modulation (PM) 239
phase noise 227
phase noise analysis 227
PHASENOISE 227, 230
Index
Q
PHASENOISE algorithms 232
PHOTO model parameter 375
PI (linear acceleration) algorithm 350
PLL, jitter measurements 243
port_element_voltage_matchload configuration
option 405
pow(x,y) function 85
power amplifier 20
power amplifier IP3 24
.POWER statement 421
power, function 85
.POWERDC statement 420
.PRINT ENV command 317
.printhl# file 307
.printls# file 311
.printss# file 311
.PROBE command 317
Probing Subcircuit currents 409
PTDNOISE
input syntax 283
.MEASURE 288
output file format 287
output syntax 286
overview 282
syntax 283
PTDNOISE command 282
pwr(x,y) function 85
Q
quality assurance 354
R
R Element (resistor) 101
rcells, reusing 90
rcxt_divider configuration option 405
reference temperature 357
reluctors 118
resistor
frequency-dependent 103
length parameter 100
linear 101
model name 99
node to bulk capacitance 100
width parameter 100
restricting output 407
results 44
reusing simulation output 401, 420, 421
RF
demo files 58
tutorial examples 15
rise time
example 417
verify 416
RSH model parameter 359
S
S parameter extraction
large-signal 307
power-dependent 299
small-signal 307
saturable core
elements 112
models 112
scale factors 68
SCALE parameter 99
schematic
netlists 70
schematic netlists 70
scope of parameters 89
SEARCH option 77
separating AM-PM noise 239
SETUP time verification 418
sgn(x) function 86
Shooting Newton
driven phase frequency circuit example 46
overview 199
ring oscillator example 53
sign function 86
signed power function 85
SIM_ACCURACY option 396
SIM_ACTIVE option 324, 327, 328, 329
SIM_DELTAI option 402
SIM_DELTAV option 402
SIM_DSPF option 324, 395, 396, 402
SIM_DSPF_ACTIVE option 324, 327
SIM_DSPF_INSERROR option 329
SIM_DSPF_LUMPCAPS option 329
SIM_DSPF_MAX_ITER option 328
SIM_DSPF_RAIL option 328
SIM_DSPF_SCALEC option 328
SIM_DSPF_SCALER option 328
431
Index
S
SIM_DSPF_VTOL option 327
SIM_LA option 324, 325, 348, 351
SIM_LA_FREQ option 351
SIM_LA_MAXR option 351
SIM_LA_MINC option 351
SIM_LA_MINMODE option 351
SIM_LA_TIME option 351
SIM_LA_TOL option 352
SIM_ORDER option 395
SIM_POSTAT option 408
SIM_POSTDOWN option 408
SIM_POSTSCOPE option 409
SIM_POSTSKIP option 407, 408
SIM_POWERDC_ACCURACY option 420
SIM_POWERED_HSPICE option 420
SIM_POWERPOST option 422
SIM_POWERSTART option 422
SIM_SPEF option 324
SIM_SPEF_ACTIVE option 327
SIM_SPEF_INSERROR option 329
SIM_SPEF_LUMPCAPS option 329
SIM_SPEF_MAX_ITER option 328
SIM_SPEF_PARVALUE option 329
SIM_SPEF_RAIL option 328
SIM_SPEF_SCALEC option 328
SIM_SPEF_SCALER option 328
SIM_SPEF_VTOL option 327
simulation engine 1
sin(x) function 84
sinh(x) function 85
skew
file 362
parameters 358
skip_nrd_nrs configuration option
configuration options
skip_nrd_nrs 405
slew rate
example 416
verification 416
small-signal noise parameter extraction 307
small-signal S parameter extraction 307
SN steady-state time domain analysis 199
SNAC
input syntax 261
output data files 264
SNFT 205
432
SNNOISE 274
input syntax 275
output data files 279
SNOSC 224
SNXF 294
spur (spurious) signals 229
sqrt(x) function 85
square root function 85
.ss# file 312
starting
hspicerf 9
statement 315, 316
.ENV 314
.HBOSC 209
statements
.AC 357
.CHECK EDGE 417
.CHECK FALL 417
.CHECK GLOBAL_LEVEL 416
.CHECK HOLD 418
.CHECK IRDROP 419
.CHECK RISE 416
.CHECK SETUP 418
.CHECK SLEW 416
.DC 357
.HBXF 291
.LPRINT 401
.MODEL 357
.POWER 421
.POWERDC 420
.PTDNOISE 282
.SURGE 423
.TEMP 357, 358
.TRAN 357, 421
statistical analysis 358–387
statistics, calculations 355
steady state time domain analysis, Shooting
Newton 199
strobejitter 288
subcircuit
probing currents 409
subcircuits
creating reusable circuits 71
hierarchical parameters 72
library structure 77
multiplying 73
.PRINT and .PLOT statements 75
.SURGE statement 423
Index
T
T
tabulated data output 399
Taguchi analysis 354
tan(x) function 84
tanh(x) function 85
TEMP
model parameter 357
.TEMP statement 357, 358
temper variable 88
temperature
circuit 355, 357
coefficients 99
derating 357
element 357
reference 357
variable 88
Temperature Variation Analysis 354
time domain steady state analysis 199
time variable 88
TNOM option 357
TOX model parameter 359
.TRAN statement 357, 421
transfer sign function 86
transformer, ideal 121
Trapezoidal (TRAP) integration algorithm 395, 396
TREF model parameter 357
tutorial 15
overview 1
simulation engine 1
two-tone HB 42
U
UNIF keyword 369
uniform parameter distribution 366
unit_atto configuration option 405
V
variance, statistical 355
VCD format 399
VCO 33
vector-modualted RF 159
vector-modulated RF
E element 165
F element 165
G element 165
H element 165
I element 161
implementation 159
V element 161
vendor libraries 76
Verilog-A support 5
VMRF, See vector-modulated RF 159
W
waveform display 12
WDB format 399
W-elements 124
wildcard uses 406
wildcard_left_range configuration option 405
wildcard_match_all configuration option 406
wildcard_match_one configuration option 406
wildcard_right_range configuration option 406
worst case analysis 358, 378, 387
Worst Case Corners Analysis 354
X
X() variable 409
XL model parameter 359
XPHOTO model parameter 375
XW model parameter 359
Y
yield analysis 354
v_supply configuration option 405
variables
HSPICE-specific 88
433
Index
Y
434
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