User manual | welding power supply with improved power quality

VOL. 12, NO. 16, AUGUST 2017
ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
www.arpnjournals.com
WELDING POWER SUPPLY WITH IMPROVED POWER QUALITY
Srividya A. S., S. Malathi and J. Jayachandran
Department of Electrical and Electronics Engineering, SASTRA University, India
ABSTRACT
A switched mode power supply for Manual Metal Arc Welding (MMAW) is proposed in this paper. This is done
as a comparative study, using two different converters at the front end - namely, Zeta converter and Canonical Switching
Cell converter. Both the converters operate in discontinuous inductor current mode (DICM) to accomplish inherent Power
Factor Correction (PFC). This mode of operation reduces the intricacy of control and provides considerable dc-voltage
regulation. A pulse width modulated (PWM) isolated full bridge dc-dc converter is used at the load side, for both the
designs to provide high frequency isolation. A closed loop control is being envisaged to incorporate dc voltage regulation
at the output and to provide over current protection, so that the designs are suitable for the intended application. The
designs have been simulated and the results obtained show how these two designs satisfy the requirement of the power
supply for arc welding process. The performance of the two power supplies have been evaluated on the basis of power
supply current, dynamic characteristics, power factor and voltage regulation.
Keywords: cell converter, canonical switching, voltage regulation, pulse width modulation.
1. INTRODUCTION
Welding is one of the most important process in
manufacturing industries. There are several welding
techniques, among which, arc welding is the most
commonly used in fabrication processes. Several
industries like the automotive, construction, chemical,
power generation etc immensely depend on this process
and a major amount of electrical power is consumed by
these power supplies. There are two major forms of arc
welding power supply - one with dc output, and another
with dc output. A dc power supply has current and voltage
at a steady polarity. This gives a stable arc and a smooth
weld in comparison to the power supply with an ac output.
An ac output power supply does not have a steady polarity
voltage or current, which is suitable only for aluminium
welding [1].
The quality of the weld essentially relies on the
power supply deployed. The power supplies for the
Welding process can be designed using several
configurations of circuits. A general dc Arc Welding
Power Supply (AWPS) employs an uncontrolled diode
bridge rectifier, which is trailed by a large dc-link
capacitor, at the front end. The power quality indices were
measured for the conventional power supply (power
factor, total harmonic distortion of the input current) and it
was observed that the existing power supplies have a low
power factor (PF) and large harmonic currents. This
causes increased losses in the utility side. Subsequent
works led to development of Power Factor Correction
based AWPS with better power quality [2], [3]. The
Welding Power Supplies must yield a low value of THD
and a high PF as per IEC 61000-3-2 and IEEE 519-1992
standards [4]. As far as the AWPS is concerned, fast
dynamic response and short circuit/over current protection
are the most mandatory features.
The AWPS is designed based on it overcurrent
response, arc stability, etc. For satisfactory weld
performance, a welding power supply with a constant arc
length, constant output voltage and current is chosen, as
mentioned above [5]. However, the foremost feature of the
welding power supply is to provide output voltage
regulation and to limit the output current, so as to provide
a high quality weld. Several topologies can be used for
designing a power supply for welding application.
A power supply designed with a boost converter
has certain drawbacks, like high current at start-up. It also
has not output current limiting capability. The key factors
of welding power supply are not complied. A buck
converter is also proposed in [6], but again here, the
capacitors used are of large value. The switches used in
these conventional converters are switched at a high
frequency, causing voltage stress across the switch and
also increases switching losses. Moreover, it does not
match the high frequency commutation current. This
drawback hampers the use of power supplies designed
based on half/full bridge inverters. Next up, buck-boost
converters offer better performance than the above
mentioned converters. Various single stage isolated PFC
configurations can be compared with two stage converters.
Among the several topologies, the Zeta and CSC offer
better power factor correction.
The above mentioned problems are taken care of
in the proposed configuration. In case the inductor is
designed to operate continuously, i.e., in Continuous
Conduction Mode (CCM) as presented in [7], an
additional input current sensing is required to imbibe
power factor correction. Moreover, operating the
converters in DICM mode has eradicated the extra current
sensing feature, which adds to the complexity of the
design. To offer a wide range of operation, the DICM is
found to be suitable by using reduced number of
components and sensing circuitry etc. Further, a feedback
circuitry has been incorporated to handle overcurrent in
the power supply. This helps in achieving an improved
weld quality. In short, the converter can be used to control
several parameters, like the welding current, voltage,
trigger pulse duty cycle, limit of overload current, and so
on. The performance of the two converters has been
verified by modelling the design in MATLAB/Simulink
environment. The simulated results are found to confirm
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VOL. 12, NO. 16, AUGUST 2017
ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
www.arpnjournals.com
the practical usability of the converter, with the various
power quality factors, conforming to the standards.
2. ZETA CONVERTER BASED AWPS
CIRCUIT CONFIGURATION
Zeta Converter
DBR
is
D1
D3
Sz
Filter
Cz1
Lz2
Lf
Vz
Io
Lo1
Do1
Sz1
Ro
Cb1
Vs
Cf
D4
+
Vd
Lz1
N2
Dz
Co
N1
‐
Cb2
D2
N3
Do1
Sz2
Lo2
Io
Vz
Voltage
Controller
PWM
Generator
Control unit for Zeta Converter
Current
Controller
Gating pulses to
Sz1 Sz2
Ioc
Vze
Vref
Ioe
PWM
Generator
Iref
Cmin Comparat‐
tor
Cv Voltage
Controller
Vo
Voe
Vref
Control unit for Full Bridge Converter
Figure-1. Schematic of Zeta converter based AWPS.
A Diode Bridge Rectifier (DBR) with a LC filter
is connected to the input side of a non-isolated Zeta
converter. This contains two inductors Lz1 and Lz2, an
intermediate capacitor C1, a high frequency switch Sz and
a diode Dz. The voltage obtained at the converter is
terminated at the isolated converter, containing two
capacitors of equal value, two transistor switches and a
high frequency transformer (HFT). The voltage output
from the HFT is connected to LC filters before it
terminated to electrical load.
The AC mains input voltage is rectified using the
diode bridge rectifier as indicated in the above figure. The
DC voltage is connected to a LC filter circuit to narrow
down the ripples. The PFC Zeta converter receives the
filtered DC voltage, which regulates the obtained DC
output voltage and makes the circuit to draw a sinusoidal
current from the AC mains at unity power factor. The
three different operating conditions - DCM conditions
(input inductor operating in DCM, capacitor operating in
DCM and the output inductor operating in DCM) are
analyzed to fix the best operating mode for the PFC
converter at the front-end. When the output inductor is
operating in DCM, it is observed that the THD obtained is
lowest. The DC output voltage so obtained from the Zeta
converter is connected to the isolated converter to achieve
a DC voltage at the output. Two transistor switches
present in the isolated converter are operated at a high
frequency, so that an AC voltage is obtained, which is then
given to the high frequency transformer (HFT). A full
wave bridge configuration is configured for best core
utilization. LC Filters are provided in each output winding
to decrease the ripples present in the output voltage and
output current. The isolated converter is operated in CCM,
to reduce the component stresses.
OPERATING PRINCIPLE
To understand the whole operation of the power
supply, the different Discontinuous Conduction Modes
(DCM)of operations are deliberated, i.e., the input
inductor (L1) operating in DCM, the intermediate
capacitor (C1) operating in DCM and the output inductor
(L2) operating in DCM.
G
Ipv
+
Vpv
L2
+ Vc1 ‐
Q
Cin
C1
L1
Dz
i L1
+
+
Vc2
C2
Vout
‐
‐
‐
Zeta Converter
Figure-2. Zeta converter.
The Zeta converter is analysed for Discontinuous
Conduction Mode in all the three components (input
inductor L1, intermediate capacitor C1 and output inductor
L2). The converter has better performance when the
inductor L2 has discontinuous current compared to that
when the rest two components have discontinuous current.
The circuit is analysed in this scenario and is explained
below.
The operation of the PFC converter with output
inductor in working in DCM. The waveforms of the
various components of the Zeta converter during the
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VOL. 12, NO. 16, AUGUST 2017
ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
www.arpnjournals.com
output inductor operating in DCM are indicated for one
switching cycle.
L z 2 min 
Gating pulses
Sz ON
Sz OFF

i Lz1
i Lz1
VCz1
VCz1
Output Inductor
in DCM
i Lz2
Sz on
i Lz2 peak
DESIGN
The design of the circuit is based on the switch
on and switch off period, i.e., change in the inductor
current in this phase. The characteristics of the transistor
switches and the diodes are considered to be ideal. The
transistor switching frequency is much higher than the AC
power line frequency.
Duty ratio of the PFC Zeta converter is given by,
Vdc
.
Vin (t ) V dc
(1)
a. Selection of the value of input inductor
The critical inductance value is given by
L z1 min 
D(t )TVin (t ) Vin (t )T

i in (t )
i in (t )
1.414 170  50S
0.5  2.05 1.414
2

Vd TV dc 
Vdc


2Vin (t ) Pin Vin (t )  V dc 
(153.1) 2  50S  300 
L z 2 min 

300
 1.15mH
2  350 1.414 170  (1.414 170)  300 
(3)
where Vdc and Idc are the output voltage and
current of the PFC Zeta converter, respectively.
The value of the inductor is estimated to be 1.15
mH and the value is also proportional to the rms value of
supply voltage.
c. Selection of the value of intermediate capacitor
The value of the capacitor for CCM operation is
given by,


V dc


(
)

V
t
V
dc 
 in


300
(2)
 (1.414 170)  300   4.6mH


where D(t) is the duty ratio, Vin and Iin are the
input voltage and current drawn respectively, from the AC
mains supply and T is the total switching time in one
switching cycle. The value of the inductor is estimated to
be 0.92 mH and is proportional to the rms value of AC
supply voltage.
Minimum value of inductor for DCM operation = 0.2 mH
Minimum value of inductor for CCM operation = 4.6 mH
D(t )TVdc Rdc

2Vc1
C1 

Vdc

Vdc 2  Vdc  Vin (t )
2 (Vdc  Vin (t ))

 Pin 
TVdc
TPin
2 (Vdc  Vin (t ))   (Vdc  Vin (t ))
C1 
where, D(t) is the duty ratio of the Zeta converter,
Vdc is the output voltage of the Zeta converter and Vin is
the input voltage from AC power lines.
L z1 min 
(1  D (t ))TV dc (t ) V dc DT RinV dc D (t )T


2 I dc
2 I in (t )
2Vin (t )
Minimum value of the output inductor for DCM operation
= 0.7 mH
Minimum value of the output inductor for CCM operation
= 5.75 mH
Sz off
Figure-3. Waveforms of various components.
(Lz2 in DCM).
D(t ) 
b. Selection of the value of output inductor
The critical inductance value estimated by,
50S  350
(4)
 0.0628F
20.3(300  270(1.414))  (300  270(1.414))
where Δ is the permissible ripple in the voltage
across the intermediate capacitor, Pin is the input power
and VC1 is the voltage across the intermediate capacitor.
The value of the capacitor is estimated to be
0.0628 µF.
Hence, the practical value of the Capacitor value
selected = 0.066 µF.
d. Selection of the output capacitor
The working of the output capacitor is inline with
the operation of the input capacitor of the isolated
converter. The value of the output capacitor is selected in a
manner so as to eradicate the second order harmonic
voltage.
Ch 
I dc
2Vdc
(5)
For a 9 V ripple, the value estimated = 400 µF.
e. Design of filter
The filter is important in keeping the harmonic
distortion at the AC input mains power supply at a low
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VOL. 12, NO. 16, AUGUST 2017
ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
www.arpnjournals.com
value. The value of the inductor and capacitor constituting
the filer section, are calculated as,
C max 
I p tan 
2fV p
(6)
where θ is assumed as 1° for maintaining a high
power factor, Vp and Ip are the peak input voltage and
input current. The maximum value of capacitor is
estimated as 0.4 µF.
Ld 
1
4 f c C d
2
2
(7)
Here fc is the cut-off frequency. The value of the
same is selected such that it is much more than the AC
power supply line frequency (f = 50Hz) and less than the
transistor switching frequency (fs = 20kHz). Cut-off
frequency is considered as 2 kHz. The corresponding
value of inductor is estimated to be 3.1 mH.
The transistor switches present at the primary
side of the HFT are operated at 60 kHz.
voltage varies, the output voltage Vco also changes to
adjust the duty cycle. Thus, the on and off time (width) of
PWM pulses changes suitably to maintain the output DC
voltage at a constant value.
b. Control of the isolated converter
To control the output voltages of the isolated
converter, average current control method is used. For
control, the output voltage obtained Vo is measured and
compared to a reference voltage Vref, to generate a voltage
error which is the input to the PI controller.
The output of the PI controller is compared to a
saw-tooth wave signal to produce the switching pulses to
make both the transistor switches on and off alternately in
each half cycle of one PWM period with sufficient dead
time to avoid shoot-through. The width of the pulses
changes as per the voltage error output of the comparator.
By modifying the duty cycle of the PWM pulses, the
control of the isolated converter is able to respond to other
output voltage variations.
The estimated values of all the components are
tabulated in Table-1.
Table-1. Specifications for Zeta converter based Awps.
CONTROL
There are two self-regulating controllers provided
to control the output voltages of the Zeta converter and the
isolated converter. The front-end (Zeta) converter is
controlled using the voltage follower approach and the
back end (isolated) converter uses the average current
control technique.
V e ( n )  V dcref  V dc ( n )
Selected
value
5 mH
Inductor, Lz2
1.15 mH
0.7 mH
Capacitor, Cz
0.062 µF
66 nF
Capacitor, Cb1, Cb2
630 µF
660 µF
SIMULATION RESULTS
Supply Voltage
V o lt a g e (V )
400
200
0
-200
-400
Rectified Voltage
400
V o lt a g e (V )
a. Control of the PFC converter
The control circuit of the front-end converter
produces pulses in accordance with the output voltage
error. The error is the difference in voltage between the
desired level of voltage and the measured voltage.
The voltage error at an instant n is given by the
expression,
Inductor, Lz1
Calculated
value
4.6 mH
Component
200
(8)
0
Filter Output Voltage
The voltage error Ve is supplied to the
proportional-integral (PI) controller to produce a
controlled output voltage (Vco).
V o lt a g e (V )
300
200
100
0
0
0.5
1.5
2
2.5
3
3.5
4
Time (sec)
V co ( n)  V co ( n  1)  k p V e ( n)  V e (n  1)  k i V e ( n) (9)
where kp and ki are the proportional gain and integral gain
to be set for the PI controller.
The output signal from the PI controller is then
compared to a high-frequency saw-tooth signal (St) to
produce the PWM pulses required to trigger the transistor
switches.
When St< Vco, then S = ON, else S = OFF, S
represents the switching signal output from the PWM
generator, to trigger for the transistor switch. If the output
1
Figure-4. Input voltage - 220V, rectified voltage
and the filtered input voltage.
Intermediate Output Voltage
500
0
-500
0
0.5
1
1.5
2
2.5
3
3.5
Time (sec)
Figure-5. Voltage at the primary side of HFT.
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VOL. 12, NO. 16, AUGUST 2017
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ARPN Journal of Engineering and Applied Sciences
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Output Voltage
50
Mag (% of Fundamental)
Fundamental (50Hz) = 56.19 , THD= 5.370%
40
30
20
10
0
-10
0
0.5
1
1.5
2
2.5
3
3.5
4
30
20
10
0
0
100
200
300
Time (sec)
Figure-6. Load voltage - 45V.
Output Current
0.5
1
1.5
2
2.5
3
3.5
4
Input Power Factor
0.5
0
0.5
1
1.5
2
2.5
900
1000
a) all semiconductor devices have been considered ideal;
b) the capacitors Cbo and Co were of large value to
maintain constant output voltages Vco and Vo without
any ripple for one switching period;
c) as the switching frequency (fs) >> the line frequency
(f), the supply voltage has been considered constant
for one switching frequency cycle.
Figure-7. Load current - 75 A.
0
800
OPERATION
The operating principle of the converters has
been elaborately explained in this section. Certain
assumptions have been made to explain the working of the
welding power supply:
Time (Sec)
1
700
Figure-9. Input current THD - 5.37%.
80
60
40
20
0
0
400
500
600
Frequency (Hz)
3
3.5
3. CSC CONVERTER BASED AWPS
4
Time (sec)
Figure-8. Input power factor - 0.7.
S1
Do1
S2
Lo1
Io
N2
N1
S3
DBR
is
D1
D3
Filter
Ci
Lf
S1
Sb
N2
Co
N1
Cb
S3
Lb
D2
Do3
Vb
Cf
D4
Ro
S2
Db
Vd
Vs
N3
Do2
S4
Gating pulses to
S1 S2 S3 S4
N3
Do4
S4
ma2
PWM
Generator
Cmin
Control unit for
Full Bridge
Converter
Comparat‐
tor
Cv
Control unit for CSC Converter
Vb
ma1
PWM
Generator
Voltage
Controller
S1
Do5
Vbe
N2
Vb *
S3
S4
Isc *
Ioc
S2
Current
Controller
Lo2
Voltage
Controller
Vo *
Ioe
Io
N1
N3
Do6
Figure-10. Schematic of CSC converter based AWPS.
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MODES OF OPERATION OF CSC CONVERTER
Ci Capacitor Ci
Discharging
D1
D3
Vs
Sb
Db
Cf
D4
Cb
D2
Isolated
Full bridge
converter
Lf
be large as to maintain a continuous voltage during this
interval.
Mode II: This mode begins when the switch is
turned off and the diode is forward biased. Here, the
energy stored in the inductor Lc is transferred to the
capacitor Cc and the dc-link capacitor Cco. These
capacitors start charging and the voltages across them
starts increasing. However, the inductor current decreases.
This stage is shown in Figure-11b.
Lb
Inductor Lb
Charging
Capacitor Cb
Charging
(a) Mode-I
Ci Capacitor Ci
Charging
D1
D3
Sb
Vs
Db
Cf
D4
Cb
D2
Isolated
Full bridge
converter
Lf
Lb
Inductor Lb
Discharging
Capacitor Cb
Charging
(b) Mode-II
Ci Capacitor Ci
Charging
D1
D3
Vs
Sb
Db
Cf
D4
Cb
D2
Isolated
Full bridge
converter
Lf
Lb
Inductor Lb
Discharging
Capacitor Cb
Discharging
(c) Mode-III
Figure-11. Operating modes of CSC converter.
The CSC converter operates in a similar manner
as that of a standard Cuk converter, though the latter uses
two inductors as compared to one in the former. The
operating modes have been shown in Figure-11. The input
to the CSC converter is given from the supply through the
Diode Bridge Rectifier. When the CSC converter operates
in DICM, three intermediate operating modes are
described for every switching cycle, which involves the
charging/discharging of the capacitors/inductor, due to the
turn on and off of the switch and the diode. This is
discussed in detail:
Mode I: In this mode, the switch Sc is on and the
diode Dc is in reverse biased state. During this period, the
inductor Lc is charged through both the supply and the
capacitor Cc. This is shown in Figure-11a. This causes the
capacitor Cc to discharge through the inductor Lb and the
dc-link capacitor Cb, which leads to decrease in voltage
across the capacitor Cc. Hence, the capacitor value should
Mode III: The DICM mode of operation of the
converter starts in this mode, when the inductor current
iLco becomes zero. The capacitor keeps charging through
the supply. The current to the converter is given by the dclink capacitor, as the diode is reverse biased. This is
shown in Figure-11c. This completes one switching cycle.
The inductor current remains zero until the switch is
turned on to continue the switching sequence.
MODES OF OPERATION OF FULL BRIDGE
CONVERTER
The isolated full bridge converter is fed by the
controlled output of the CSC converter. It is designed as a
buck converter, as it is required to step down the dc-link
voltage to a required level. To reduce the switching device
rating and to extend the range of output power rating, three
FB converters are connected in parallel. The FB converters
operate in Continuous Conduction Mode. The operating
modes are explained.
Mode I: In this mode, the switches S1 and S4 are
triggered and the dc-link capacitor voltage appears across
the primary winding of the high-frequency transformer
(HFT). The diodes connected to the first half of the
secondary winding are forward biased. The inductors
connected at the output (Lo1 and Lo2) start storing energy,
causing the inductor current to increase. However, the
output filter capacitor Co discharges through the load.
Mode II: This mode is begins when all the
switches S1, S2, S3 and S4 are turned off and all the
diodes connected on the secondary winding of the
transformer act as freewheeling diodes. The energy in the
inductors is discharged to the output capacitor Co and the
load. This causes the current of the inductor to decrease
linearly.
Mode III: This mode is similar to Mode I,
switch pair S2 and S3 is triggered to shift the energy to the
load. Again, the diodes Do1, Do3 and Do5 remain reverse
biased and the output capacitor Co discharges through the
load.
Mode IV: Modes II and IV are analogous as
well. None of the switch pairs conduct and all the output
rectifier diodes freewheel the energy piled up in the
inductors. The output capacitor Co is charged by the
energy released by the inductors. This mode ceases when
the switches S1 and S4 are triggered again and the
operation repeats in successive switching cycles.
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VOL. 12, NO. 16, AUGUST 2017
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ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
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DESIGN OF CSC CONVERTER BASED AWPS
The AWPS has been simulated for a power rating
of 1.5kW. It has been seen that the power supply is highly
effective for welding purposes. The switching frequency
of both the front end and isolated converters is much
higher than the line frequency. Hence, the supply voltage
is supposed to be a constant during one switching cycle.
A) Design of input filter
The output obtained at the DBR is given to the
filter circuit consisting of an inductor and capacitor. This
smoothens the input current harmonics of higher order.
The filter components have been designed to obtain lower
harmonic content in the current. The capacitance value is
calculated as,
C max 
I p tan 
(10)
2fV p
where Vp and Ip are the peak ac input voltage and
current, respectively, and f is the fundamental frequency.
Θ is assumed as 1°, so as to attain high power factor. The
maximum value of the capacitance is approximated to 0.4
µF.
The inductor is also designed to obtain a lower
harmonic content in the input current, whose value is
estimated as,
Lf 
1
2
4 f c C f
(11)
2
Dco  2 M K a
(14)
where
M
Vc 360

 1.157
Vm 311
(15)
and Ka is the conduction parameter.
For DICM operation,
Ka 
1
2( M  | sin t |) 2
(16)
The conduction parameter for the CSC converter
to operate in Discontinuous Conduction Mode is,
K a |t 90 
1
 0.107
2( M  1) 2
(17)
The value of duty ratio Dcn for the CSC converter
is obtained by substituting these values in (14).
Dcn=0.271
A. Design of input inductors (Lc): The inductor
in the CSC converter is designed to have discontinuous
inductor current in every switching cycle. The nominal
value of Lc is,
Vd Dcn
198  0.271

 145.89H
2 f sc I d 2  30000  6.13
where fc is the cut-off frequency (here, assumed as 3
kHz).
Lcn 
B) Design of CSC converter
The CSC converter is intended to supply a dclink voltage of 360V at a switching frequency of 30 kHz.
The inductor Lco has been discussed to be operating in
DICM, so the current is discontinuous during a switching
period. However, the capacitor voltage Cc remains
continuous, unlike in the inductor. The input supply
voltage (Vs) being assumed as 220 V, the rectified output
of DBR (Vr) is estimated as,
where fsc is the switching frequency of the CSC
converter. The value of the inductor Lco must be less than
the critical value for proper operation of the inductor in
Discontinuous Conduction Mode.
Vr 
2 2V s


2 2 (220)

 198V
(12)
The output voltage Vc of the CSC converter is
expressed by,
Vc 
Dc
Vr
1  Dc
(13)
where Dc is the duty ratio of the switch Sc.
The duty ratio is chosen at an optimal value for
Discontinuous Operating Mode, which is given by,
(18)
B. Design of capacitor (Cc): This capacitor is
designed such that the voltage across it is continuous
throughout a switching period. The voltage ripple limit
that is permitted is assumed to be 10% of VCc.
Ci 
Vco Dbn
Pi Dbn
1500  0.271


 0.672 F
VCi f sb Rb VCi f sbVco 56  30000  360
(19)
C. Design of DC-Link capacitor (Cco): The
value of this capacitor relies on the voltage ripple ΔVco in
the dc-link voltage. Assuming the output voltage ripple to
be 5% of the dc-link voltage, the capacitor value is
calculated as,
Cb 
Po
1500

 736.8F
2f LVco Vco 2  50  360  18
(20)
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ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
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where fL is the line frequency, and ΔVco is the
output voltage ripple. The capacitor value is chosen as
940µF to reduce output voltage ripple.
C) Design of full bridge buck converter
Full bridge converters are used in the second
stage of the AWPS. Since the required voltage required at
the output is much less than that obtained at the CSC
converter, the full bridge converter is a buck converter,
connected in a modular manner. Hence, the devices used
in each of them are of lower power rating. The input to
this full bridge buck converter is the output of the CSC
converter. This second stage achieves isolation by
operating at a switching frequency of 100 kHz and it
signifies the regular welding power supply. For simple
explanation, a single module of the FB converter is
considered here for analysis.
A. Design of turns ratio of HFT: The net
change in the inductor current over one switching cycle is
zero, for an isolated full bridge buck converter. The ratio
of input to output voltage is given by,
  N2
Vd 
  N1
 t

  Vo  ON

 Ts
   t OFF
  Vo 
   Ts

  0

VoVd
Vo
 N1  .


 N2 
(22)
The FB buck converter operates in CCM. Thus
the turns ratio is calculated to obtain an output voltage of
60V, by reorganizing the above equation.
 N1

 N2
 2 D f Vd 2  0.4  360
 

 4 .8
60
Vo

(23)
B. Design of output inductors (Lo1 and Lo2):
The converter in the second stage of the power supply is
designed to operate in CCM. Thus the currents through
these inductors must be continuous in one switching cycle.
To satisfy this condition, the inductor values must be large
enough to maintain continuous current. The allowed ripple
in the current ΔiLo, is 10% of Io. Hence, the output inductor
values are expressed by,
Lo1 
V o ( 0 .5  D f )
f s ( i Lo1 )

60  (0.5  0.4)
 12 H
100000  5
Vo (1  2D f )
2
(24)
C. Design of output capacitor (Co): This
capacitor is deliberated to alleviate the ripples in the
output voltage, which can be calculated by,
.
(25)
32 f s Lo1Co
Assuming the output voltage ripple to be 10%,
the value of capacitor is given by,
Co 
Vo (1  D f )
2
8 f s Lo1 (Vo )

60  (1  0.4)
 5F
8  100000 2  15  6
(26)
The above calculated values for the design of the
power supply, along with the component requirement is
tabulated in Table-2.
Table-2. Design specifications for the proposed welding
power supply.
Converter
Component
Inductor, L
CSC
Converter
(21)
where Ts=1/fs indicated the switching period of
the FB converter. For the rated output voltage Vo, the duty
ratio is calculated by,
Df 
Vo 
FB Buck
Converter
Capacitor, C
DC-link
capacitor
Turns ratio,
N1/N2
Output Inductors,
Lo1 and Lo2
Output
Capacitor, Co
Calculated
value
145.89µH
Opted
value
90µH
0.672µF
0.66µF
736.8µF
940µF
4.8
5
12µH
15µH
5µF
7µF
CONTROL OF THE WELDING POWER SUPPLY
The power supply for welding application is
suitably designed with two separate controllers for each
converter. Here, a voltage follower approach is
implemented for the first converter, i.e., the front end CSC
converter, so as to achieve near unity power factor at the
utility side. For the second stage converter however, a
dual-loop control scheme is implemented. A concise
description of both the control strategies is described.
a) Control of the CSC converter
The CSC converter is designed to operate in
DICM; the voltage follower method is used. The DICM
operation of the inductor in the CSC converter inherently
helps in achieving power factor correction by employing a
voltage feedback control loop. This regulates the dc-link
voltage Vco in spite of output current and output voltage
variations. Here, the dc-link voltage is taken as feedback
and is compared to a reference voltage Vcref. The error
voltage produced Vce at any instant is expressed as,
Vce (n)  Vcref (n)  Vc (n)
(27)
Subsequently, this error to fed to a proportional
and integral (PI) controller to get a controlled output
voltage.
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ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
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b) Control of the FB buck converter
This full bridge Buck converter is anticipated to
deliver a constant dc output voltage. As an adjunct
function, the converter should also have current limiting
function during overload conditions, so as to guarantee a
high-quality weld over a wide range of voltage variation.
This is one of the most indispensable criterions, which
must be incorporated in the design of power supply for
welding purposes. A dual loop control scheme is adopted
here. In addition to tracking the output voltage, the output
current is also tracked. The voltage loop is used to control
the output voltage, whereas the current loop limits the
output load current during overload conditions. The output
voltage Vo is measured and then compared to a reference
voltage Vref to generate an error signal, Ve. This error
signal is then fed to the PI controller, as done in the
control of CSC converter, to sustain the output voltage at
the required value. The output of the PI voltage controller
is given by,
CV ( k )  C V ( k  1)  k ' pv {V e ( k )  V e ( k  1)}  k ' ivV e ( k ) (29)
where k’pv and k’iv are the proportional and
integral gains of the voltage controller, respectively.
In the second loop, the output current Io is taken
as feedback signal and compared to a current limit Ilim. It
comprises of a PI controller, which processes the error
signal to confine the output current within limits.
Supply Voltage
V o lta g e (V )
400
200
0
-200
-400
Rectified Voltage
V o lt a g e (V )
400
200
0
Filter Output Voltage
300
V o lta g e (V )
where kpv and kiv indicate the proportional and
integral gains of the PI controller respectively.
Thereafter, the output of the PI voltage controller
is compared to a high frequency saw-tooth waveform to
derive the trigger pulses for the switch in the CSC
converter.
If the magnitude of the saw tooth wave is less
than the magnitude of the controlled output voltage Vco at
any instant, then the switch Sc will be turned on; and if the
magnitude of the saw tooth wave is greater than the
magnitude of the controlled output voltage Vco, then the
switch Sc will be turned off.
SIMULATION RESULTS
200
100
0
0
0.5
1
1.5
2
2.5
3
3.5
4
Time (sec)
Figure-12. Supply voltage - 220V, Rectified voltage
after DBR, Filtered output voltage.
Output Voltage of CSC converter
400
Voltage (V)
Vco ( n )  Vco ( n  1)  k pv {Vce ( n )  Vce ( n  1)}  k ivVce ( n ) (28)
300
200
100
0
0
0.5
1
1.5
2
2.5
3
3.5
4
Time (sec)
Figure-13. Voltage of the CSC converter.
Output Voltage
60
40
20
0
0
0.5
1
1.5
2
2.5
3
3.5
4
Time (Sec)
Figure-14. Load voltage - 45V.
Output Current
80
60
40
20
0
0
0.5
1
1.5
2
2.5
3
3.5
4
3.5
4
Time (Sec)
Figure-15. Load current - 75 A.
Power Factor
1
0.5
C i (k )  Ci (k  1)  k ' pi {I e (k )  I e (k  1)}  k ' ii I e (k ) (30)
0
0
0.5
1
1.5
2
2.5
3
Time (Sec)
Figure-16. Input power factor - 0.9.
Fundamental (50Hz) = 55.98 , THD= 4.208%
30
M a g (% o f F u n d a m e n t a l)
where k’pi and k’ii represent the proportional and
integral gain constants of the current controller,
respectively. Thereafter, the comparator compares the
outputs of both the current and voltage controllers, and the
signal with the lower amplitude is fed to the pulse widthmodulated generator. It produces the gating pulses
required to trigger the devices and hence the duty cycle of
the switches correspond to the changes in the output
voltage and current.
20
10
0
0
500
1000
1500
Frequency (Hz)
Figure-17. Input current THD = 4.20%.
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ARPN Journal of Engineering and Applied Sciences
©2006-2017 Asian Research Publishing Network (ARPN). All rights reserved.
www.arpnjournals.com
welding can be replaced with the CSC converter based
power supply.
4. VALIDATION OF THE AWPS WITH
SIMULATION RESULTS
Table-3. Comparison of performances of the designed
welding power supplies.
Input Voltage
Zeta based
AWPS
220V
CSC based
AWPS
220V
Output Voltage
45V
45V
Parameter
Output Current
75V
75V
Input Power Factor
0.75
0.9
Input Current THD
5.37%
4.20%
A. Steady state performance
The welding power supply is rated at 220 V
supply voltage. The corresponding output voltage, output
current is shown in the Figures 14, 15. The transitional
results - like the CSC converter dc voltage and the current
is presented in Figure-13. The output voltage is regulated
to 45 V, while the dc voltage is retained at 300V. The
voltage follower approach maintains a sinusoidal input
current, which is in phase with the input voltage. The CSC
inductor working in DICM and the continuous voltage
across the intermediate capacitor is shown.
B. Comparative analysis
A comparative study was made by simulating
both the designed converters and the results have been
presented in Table-3. Clearly, from the tabulation, the Zeta
converter has a low power factor. The power factor can be
improvised by modifying the configuration, but the CSC
converter provides satisfactory operation without any kind
of modification. The initial cost and size of the Zeta
converter is quite high compared to the CSC converter
AWPS. On the basis of the observed results, it can be
confirmed that the designed can be validated for the
welding application and the CSC converter based AWPS
achieves a better performance than the conventional
welding power supplies and the Zeta converter based
AWPS and can be used as a feasible solution for AWPS.
REFERENCES
[1] K. Weman. 2003. Welding Process Handbook.
Cambridge, MA, USA: Woodhead. pp. 13-25.
[2] B. Singh, S. Singh, A. Chandra, and K. Al-Haddad.
2011. Comprehensive study of single phase AC-DC
power factor corrected converters with high frequency
isolation. IEEE Trans. Ind. Informat. 7(4): 540-556.
[3] J.-M. Wang and S.-T. Wu. 2015. A novel inverter for
arc welding machines. IEEE Trans. Ind. Electron.
62(3): 1431-1439.
[4] 20004. Limits for Harmonic Emissions. International
Electro Technical Commission Standard 61000-3-2.
[5] Q. F. Teng, W. Z. Zhang, J. G. Zhu, and Y.G. Guo.
2011. Modeling of arc welding power supply. in Proc.
Int. Conf. Appl. Supercond. Electromagn. Devices.
pp. 228-231.
[6] K. Mahmoodi, M. Jafari and Z. Malekjamshidi. 2012.
Operation of a Fuzzy controlled half-bridge DCconverter as a welding current-source. Indonesian J.
Elect. Eng. 10(1): 17-24.
[7] D. Murthy-Bellur and M. K. Kazimierczuk. 2011.
Isolated two-transistor Zeta converter with reduced
transistor voltage stress. IEEE Trans. Circuits Syst. I,
Reg. Papers. 58(1): 41-45.
5. CONCLUSIONS
Two different configurations have been simulated
in MATLAB/Simulink for the requirement of a power
factor corrected power supply for welding application. The
first configuration is one with a Zeta Converter in the front
end. It was found to give appreciable performance in
comparison with the conventional welding power suppl.
The distortion in the input current has drastically
decreased, along with a good input power factor. The
performance was still improved with another
configuration, namely a Canonical Switching Cell
converter. This uses one inductor unlike the former and
has been found to be performing better. The results
obtained from the simulation prove the same. Better
power quality is attained-with power quality indices with
in specified limits. Thus the conventional power supply for
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