Texas Instruments | Quad, Unity-Gain, Low-Noise, Voltage-Feedback Operational Amplifier (Rev. D) | Datasheet | Texas Instruments Quad, Unity-Gain, Low-Noise, Voltage-Feedback Operational Amplifier (Rev. D) Datasheet

Texas Instruments Quad, Unity-Gain, Low-Noise, Voltage-Feedback Operational Amplifier (Rev. D) Datasheet
OPA4820
SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
Quad, Unity-Gain Stable, Low-Noise,
Voltage-Feedback Operational Amplifier
FEATURES
D
D
D
D
D
D
DESCRIPTION
HIGH BANDWIDTH: 220MHz (G = +2)
HIGH OUTPUT CURRENT: ±85mA
LOW INPUT NOISE: 2.5nV/√Hz
LOW SUPPLY CURRENT: 5.7mA/ch
FLEXIBLE SUPPLY VOLTAGE:
±2V to ±6.3V Dual Supply
+4V to +12.6V Single Supply
EXCELLENT DC ACCURACY:
Maximum 25°C Input Offset Voltage = 0.8mV
Maximum 25°C Input Offset Current = 500nA
The OPA4820 provides a wideband, unity-gain stable,
voltage-feedback amplifier with a very low input noise
voltage and high output current using a low 5.7mA/ch
supply current. At unity-gain, the OPA4820 gives
> 600MHz bandwidth with < 1 dB peaking. The OPA4820
complements this high-speed operation with excellent DC
precision in a low-power device. A worst-case input offset
voltage of ±0.8mV and an offset current of ±500nA give
excellent absolute DC precision for pulse amplifier
applications.
Minimal input and output voltage swing headroom allow
the OPA4820 to operate on a single +5V supply with
> 2VPP output swing. While not a rail-to-rail (RR) output,
this swing will support most emerging analog-to-digital
converter (ADC) input ranges with lower power and noise
than typical RR output op amps.
APPLICATIONS
D
D
D
D
D
D
D
D
D
LOW-COST VIDEO LINE DRIVERS
ADC PREAMPS
ACTIVE FILTERS
LOW-NOISE INTEGRATORS
PORTABLE TEST EQUIPMENT
OPTICAL CHANNEL AMPLIFIERS
LOW-POWER, BASEBAND AMPLIFIERS
CCD IMAGING CHANNEL AMPLIFIERS
OPA4650 UPGRADE
Exceptionally low dG/dP (0.01%/0.03°) supports low-cost
composite video line driver applications. Existing designs
can use the industry-standard quad pinout SO-14 package
while emerging high-density portable applications can use
the TSSOP-14.
RELATED PRODUCTS
+12V
1/4
OPA4820
10kΩ
402Ω
Transmit
Filter
50Ω
VCM
133Ω
100Ω
Line
14VPP
402Ω
DUALS
TRIPLES
QUADS
OPA354
OPA690
—
—
OPA820
OPA2354
OPA2690
OPA2652
OPA2822
—
—
OPA3690
—
—
—
OPA4354
—
—
—
—
FEATURES
CMOS RR Output
High Slew Rate
SOT23-8
Low-Noise
Single Channel
1:1
+6V
2VPP
SINGLES
50Ω
10dBm
3.5 Crest Factor
10kΩ
1/4
OPA4820
800Ω
AFE
800Ω
402Ω
402Ω
VCM
1/4
OPA4820
402Ω
Receiver
Filter
402Ω
1/4
OPA4820
VCM
Low-Noise Transceiver Interface
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright  2004−2008, Texas Instruments Incorporated
! ! www.ti.com
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
This integrated circuit can be damaged by ESD. Texas
Instruments recommends that all integrated circuits be
handled with appropriate precautions. Failure to observe
proper handling and installation procedures can cause damage.
ABSOLUTE MAXIMUM RATINGS(1)
Power Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±6.5V
Internal Power Dissipation . . . . . . . . . See Thermal Characteristics
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±1.2V
Input Common-Mode Voltage Range . . . . . . . . . . . . . . . . . . . . ±VS
Storage Temperature Range . . . . . . . . . . . . . . . . . −65°C to +125°C
Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300°C
Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
ESD Rating:
Human Body Model (HBM) . . . . . . . . . . . . . . . . . . . . . . . 2000V
Charge Device Model (CDM) . . . . . . . . . . . . . . . . . . . . . 1000V
Machine Model (MM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200V
(1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods
may degrade device reliability. These are stress ratings only, and
functional operation of the device at these or any other conditions
beyond those specified is not supported.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could
cause the device not to meet its published specifications.
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
OPA4820
″
SO-14
D
−45°C to +85°C
OPA4820
OPA4820ID
Rails, 58
″
″
″
″
OPA4820IDR
Tape and Reel, 2500
OPA4820
″
TSSOP-14
PW
−45°C to +85°C
OPA4820
OPA4820IPWT
Tape and Reel, 250
″
″
″
″
OPA4820IPWR
Tape and Reel, 2500
ORDERING
NUMBER
TRANSPORT MEDIA,
QUANTITY
(1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet or see the
TI website at www.ti.com.
PIN CONFIGURATION
TOP VIEW
SO, TSSOP
Output A
1
−Input A
2
A
Output D
13
−Input D
D
+Input A
3
12
+Input D
+V
4
11
−V
+Input B
5
10
+Input C
B
C
−Input B
6
9
−Input C
Output B
7
8
Output C
OPA4820
2
14
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
At RF = 402Ω, RL = 100Ω, and GND = +2, unless otherwise noted.
OPA4820ID, IPW
TYP
PARAMETER
AC PERFORMANCE
Small-Signal Bandwidth
Gain-Bandwidth Product
Peaking at a Gain of 1
Bandwidth for 0.1dB Gain Flatness
Large-Signal Bandwidth
Slew Rate
Rise Time and Fall TIme
Settling Time to 0.02%
Settling Time to 0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Noninverting Input Current Noise
Differential Gain
Differential Phase
All Hostile Crosstalk, Input-Referred
DC PERFORMANCE(4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Input Offset Voltage Drift
Input Bias Current
Average Input Bias Current Drift
Input Offset Current
Inverting Input Bias Current Drift
INPUT
Common-Mode Input Range (CMIR)(5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance, Differential Mode
Input Impedance, Common-Mode
OUTPUT
Output Voltage Swing
Output Current
Short-Circuit Output Current
Closed-Loop Output Impedance
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage
Minimum Operating Voltage
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (−PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IPW
Thermal Resistance, qJA
D
SO-14
PW
TSSOP-14
CONDITIONS
+25°C
G = +1, VO = 0.1VPP, RF = 25Ω
G = +2, VO = 0.1VPP
G = +10, VO = 0.1VPP
G ≥ 20
VO = 0.1VPP, RF = 25Ω
G = +2, VO = 0.1VPP
G = +2, 2VPP
G = +2, 2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 1MHz, VO = 2VPP
RL = 200Ω
RL ≥ 500Ω
RL = 200Ω
RL ≥ 500Ω
f > 100kHz
f > 100kHz
G = +2, NTSC, VO = 1.4VPP, RL = 150Ω
G = +2, NTSC, VO = 1.4VPP, RL = 150Ω
3 Channels Driven at 5MHz, 1VPP
4th Channel Measured
650
220
27
250
1
33
80
240
1.5
22
18
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
−84
−90
−92
−105
2.5
1.7
0.003
0.06
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C(2)
−40°C to
+85°C(2)
150
21
200
140
19
185
135
18
180
190
184
178
−80
−85
−87
−100
2.7
2.6
−79
−83
−86
−97
2.8
2.8
−78
−81
−85
−95
2.9
3.0
−61
66
±0.25
62
±0.8
−9
−20
±100
±500
VCM = 0V, Input-Referred
VCM = 0V
VCM = 0V
±4.0
85
18 0.8
6 1.0
RL ≥ 500Ω
RL = 100Ω
VO = 0
Output Shorted to Ground
G = +2, f ≤ 100kHz
±3.7
±3.6
±85
±110
0.04
Junction-to-Ambient
Junction-to-Ambient
MIN/
MAX
TEST
LEVEL(3)
MHz
MHz
MHz
MHz
dB
MHz
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
C
B
C
C
C
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
%
°
max
max
max
max
max
max
typ
typ
B
B
B
B
B
B
C
C
dB
typ
C
61
±1.2
8
−22
30
±800
5
60
±1.5
10
−26
50
±900
5
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
±3.8
76
±3.7
75
±3.6
73
V
dB
kΩ  pF
MΩ  pF
min
min
typ
typ
A
A
C
C
±3.5
±3.5
±70
±3.45
±3.45
±65
±3.4
±3.4
±60
V
V
mA
mA
Ω
min
min
min
typ
typ
A
A
A
C
C
±6.3
±6.3
±6.3
23.4
21.8
64
25
20.2
63
25.8
19.4
62
V
V
V
mA
mA
dB
typ
max
typ
max
min
min
C
A
C
A
A
A
−40 to +85
°C
typ
C
100
110
°C/W
°C/W
typ
typ
C
C
±5
VS = ±5V
VS = ±5V
Input-Referred
UNITS
±2
22.6
22.6
72
(1)
Junction temperature = ambient for +25°C specifications.
Junction temperature = ambient at low temperature limits; junction temperature = ambient +28°C at high temperature limit for over temperature specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only
for information.
(4) Current is considered positive out of pin.
(5) Tested < 3dB below minimum specified CMRR at ± CMIR limits.
(2)
3
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
At RF = 402Ω, RL = 100Ω to 2.5V, and G = +2, unless otherwise noted.
OPA4820ID, IPW
TYP
PARAMETER
AC PERFORMANCE
Small-Signal Bandwidth
Gain-Bandwidth Product
Peaking at a Gain of 1
Large-Signal Bandwidth
Slew Rate
Rise Time and Fall Time
Settling Time to 0.02%
Settling Time to 0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Noninverting Input Current Noise
All Hostile Crosstalk, Input-Referred
DC PERFORMANCE(4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Input Offset Voltage Drift
Input Bias Current
Average Input Bias Current Drift
Input Offset Current
Inverting Input Bias Current Drift
INPUT
Least Positive Input Voltage
Most Positive Input Voltage
Common-Mode Rejection Ratio (CMRR)(5)
Input Impedance, Differential-Mode
Input Impedance, Common-Mode
OUTPUT
Least Positive Output Voltage
Most Negative Output Voltage
Output Current
Short-Circuit Output Current
Closed-Loop Output Impedance
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage
Minimum Operating Voltage
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (+PSRR)
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C(2)
−40°C to
+85°C(2)
148
18
180
135
17
170
130
16
160
135
130
125
−75
−79
−87
−95
2.8
2.5
−74
−77
−86
−93
2.9
2.7
−73
−75
−85
−92
3.0
2.9
CONDITIONS
+25°C
G = +1, VO = 0.1VPP, RF = 25Ω
G = +2, VO = 0.1VPP
G = +10, VO = 0.1VPP
G ≥ 20
VO = 0.1VPP, RF = 25Ω
G = +2, VO = 2VPP
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 1MHz, VO = 2VPP
RL = 200Ω
RL ≥ 500Ω
RL = 200Ω
RL ≥ 500Ω
f > 100kHz
f > 100kHz
3 Channels Driven at 5MHz, 1VPP
4th Channel Measured
520
210
25
230
2
67
190
1.8
25
22
VO = 2.5V, RL = 100Ω
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
65
±0.35
60
±1.3
−8
−18
±100
±500
VCM = 2.5V, Input-Referred
VCM = 2.5V
VCM = 2.5V
0.9
4.4
83
15 1
5 1.3
RL ≥ 500Ω to 2.5V
RL = 100Ω to 2.5V
RL ≥ 500Ω to 2.5V
RL = 100Ω to 2.5V
VO = 2.5V
Output Shorted to Ground
G = +2, f ≤ 100kHz
3.9
3.8
1.2
1.2
±75
±105
0.04
−79
−83
−94
−98
2.5
1.6
−61
+4
20.4
20.4
68
UNITS
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
B
C
C
C
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
max
max
max
max
max
max
B
B
B
B
B
B
dB
typ
C
(3)
59
±1.7
8
−20
30
±800
5
58
±2.0
10
−24
50
±900
5
dB
mV
µV/°C
µA
nA/°C
µA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
1.1
4.1
74
1.2
4.0
73
1.3
3.9
72
V
V
dB
kΩ  pF
MΩ  pF
min
max
min
typ
typ
A
A
A
C
C
3.8
3.7
1.3
1.3
±60
3.75
3.65
1.35
1.35
±55
3.7
3.6
1.4
1.4
±50
V
V
V
V
mA
mA
Ω
min
min
min
min
min
typ
typ
A
A
A
A
A
C
C
+12.6
+12.6
+12.6
22.2
18.0
22.6
17.4
23.0
16.8
V
V
V
mA
mA
dB
typ
max
typ
max
min
typ
C
A
C
A
A
C
+5
VS = +5V, 4 Channels
VS = +5V, 4 Channels
Input-Referred
TEST
LEVEL
MIN/
MAX
THERMAL CHARACTERISTICS
Specification: ID, IPW
−40 to +85
°C
typ
C
Thermal Resistance, qJA
D
SO-14
Junction-to-Ambient
100
°C/W
typ
C
PW
TSSOP-14
Junction-to-Ambient
110
°C/W
typ
C
(1) Junction temperature = ambient for +25°C specifications.
(2) Junction temperature = ambient at low temperature limits; junction temperature = ambient +13°C at high temperature limit for over temperature.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only
for information.
(4) Current considered positive out of pin.
(5) Tested < 3dB below minimum specified CMRR at ± CMIR limits.
4
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±5V
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted.
NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE
3
INVERTING SMALL−SIGNAL FREQUENCY RESPONSE
3
G = +1, RF = 25Ω
G = −1
0
Normalized Gain (dB)
−3
G = +5
G = +2
−6
G = +10
−9
−12
VO = 0.1VPP
RL = 100Ω
See Figure 1
−15
−18
1M
−3
−6
G = −10
−9
−12
VO = 0.1VPP
RL = 100Ω
See Figure 2
−15
−18
10M
100M
1G
1
10
Frequency (Hz)
100
NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE
INVERTING LARGE−SIGNAL FREQUENCY RESPONSE
3
VO = 0.5VPP
6
0
3
−3
VO = 1VPP
0
Gain (dB)
Gain (dB)
VO = 0.5VPP
VO = 2VPP
−3
VO = 4VPP
−6
−12
1
VO = 1VPP
VO = 2VPP
−6
VO = 4VPP
−9
−12
G = +2V/V
RL = 100Ω
See Figure 1
−9
G = −1V/V
RL = 100Ω
See Figure 2
−15
−18
10
100
500
1
10
Frequency (MHz)
0
1.5
1.0
Small−Signal ± 100mV
Left Scale
0.5
0
−0.1
−0.5
−0.2
−1.0
−0.3
−1.5
−0.4
−2.0
Time (10ns/div)
Small−Signal Output Voltage (100mV/div)
0.2
Large−Signal Output Voltage (500mV/div)
Small−Signal Output Voltage (100mV/div)
Large−Signal ± 1V
Right Scale
500
INVERTING PULSE RESPONSE
2.0
G = +2V/V
See Figure 1
100
Frequency (MHz)
NONINVERTING PULSE RESPONSE
0.4
0.1
500
Frequency (MHz)
9
0.3
G = −2
G = −5
0.4
0.3
2.0
G = −1V/V
See Figure 2
1.5
0.2
1.0
0.1
0
0.5
Small−Signal ± 100mV
Left Scale
−0.1
−0.2
−0.3
0
−0.5
Large−Signal ± 1V
Right Scale
−0.4
−1.0
−1.5
−2.0
Large−Signal Output Voltage (500mV/div)
Normalized Gain (dB)
0
Time (10ns/div)
5
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted.
1MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
−60
−65
−65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
HARMONIC DISTORTION vs LOAD RESISTANCE
−60
−70
−75
2nd−Harmonic
−80
−85
−90
G = +2V/V
f = 1MHz
VO = 2VPP
−95
−100
3rd−Harmonic
G = +2V/V
RL = 200Ω
VO = 2VPP
See Figure 1
−70
−75
−80
2nd−Harmonic
−85
−90
3rd−Harmonic
−95
−100
−105
−105
100
2.5
1000
3.0
3.5
−70
2nd−Harmonic
−75
−80
−85
−90
3rd−Harmonic
−95
6.0
−70
−75
−80
2nd−Harmonic
−85
−90
3rd−Harmonic
−95
−105
−105
0.1
1
0.1
10
1
HARMONIC DISTORTION vs NONINVERTING GAIN
f = 1MHz
RL = 200Ω
VO = 2VPP
See Figure 1
−75
−80
HARMONIC DISTORTION vs INVERTING GAIN
−70
2nd−Harmonic
−75
Harmonic Distortion (dBc)
−70
10
Output Voltage Swing (VPP)
Frequency (MHz)
Harmonic Distortion (dBc)
5.5
−100
−100
−85
−90
3rd−Harmonic
−95
−100
−105
2nd−Harmonic
−80
−85
3rd−Harmonic
−90
−95
f = 1MHz
RL = 200Ω
VO = 2VPP
See Figure 2
−100
−105
−110
−110
1
10
Gain (V/V)
6
5.0
G = +2V/V
f = 1MHz
RL = 200Ω
See Figure 1
−65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
G = +2V/V
VO = 2VPP
RL = 200Ω
See Figure 1
−65
4.5
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY
−60
4.0
Supply Voltage (±VS)
Resistance (Ω)
1
10
Gain ( V/V )
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted.
TWO−TONE, 3RD−ORDER
INTERMODULATION INTERCEPT
INPUT VOLTAGE AND CURRENT NOISE
100
45
Intercept Point (+dBm)
Voltage Noise (nV/√Hz)
Current Noise (pA/√Hz)
PI
1 /4
50Ω
40
10
Voltage Noise (2.5nV/√Hz)
PO
OP A4820
200Ω
402 Ω
35
402 Ω
30
25
20
Current Noise (1.7pA/√Hz)
1
100
1k
10k
100k
1M
15
10M
0
5
10
Frequency (Hz)
Normalized Gain to Capacitive Load (dB)
RS (Ω )
0.1dB Peaking Targeted
10
1
10
100
8
5
CL = 47pF
4
CL = 100pF
3
2
RS
VI
1
50Ω
VO
OPA4820
CL
0
1kΩ(1)
40 2Ω
−1
402Ω
−2
NOTE: (1 ) 1kΩ is optional.
−3
1
1000
10
100
400
Frequency (MHz)
OPEN−LOOP GAIN AND PHASE
80
CMRR
80
70
70
60
Open−Loop Gain (dB)
Common−Mode Rejection Ratio (dB)
Power−Supply Rejection Ratio (dB)
30
CL = 22pF
6
CMRR AND PSRR vs FREQUENCY
60
25
CL = 10pF
7
Capacitive Load (pF)
90
20
FREQUENCY RESPONSE vs CAPACITIVE LOAD
RECOMMENDED RS vs CAPACITIVE LOAD
100
1
15
Frequency (MHz)
+PSRR
50
40
0
−20
20 log (AOL)
−40
−60
50
−80
40
∠AOL
30
−100
20
−120
20
10
−140
10
0
−160
0
−10
100
30
−PSRR
1k
10k
100k
1M
Frequency (Hz)
10M
100M
1k
10k
100k
1M
Frequency (Hz)
10M
100M
1G
Open−Loop Phase (_)
10
−180
7
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted.
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
Output Current
Power Limit
Limit
3
RL = 100Ω
2
VO (V)
10
1
RL = 25Ω
0
RL = 50Ω
−1
−2
−3
Output Current
−4
Single Channel
−100
0.1
1W Internal
Limit
−5
−150
1
Power Limit
0.01
−50
0
50
100
150
1k
10k
100k
IO (mA)
4
5
6
3
4
4
2
1
Output Left Scale
0
0
−2
−1
−2
RL = 100Ω
G = +2V/V
See Figure 1
−3
−8
Input/Output Voltage (1V/div)
2
Input Voltage (1V/div)
Output Voltage (2V/div)
Input Right Scale
−6
3
Input
1
0
−1
−2
−3
Output
RL = 100Ω
G = −1V/V
See Figure 2
−5
Time (40ns/div)
Time (40ns/div)
TYPICAL DC DRIFT OVER TEMPERATURE
COMPOSITE VIDEO dG/dP
0.14
0.28
0.12
0.24
0.10
0.20
0.08
0.16
dP Negative Video
0.12
0.04
0.08
0.02
Input Offset Voltage (VOS)
Left Scale
0.5
0
10x Input Offset Current (IOS)
Right Scale
−0.5
1
2
3
Video Loads
0
4
10
0
−10
Input Bias Current (IB)
Right Scale
0.04
dG Positive Video
0
8
Input Offset Voltage (mV)
0.32
dP Positive Video
20
0.36
Differential Phase (_)
Differential Gain (%)
dG Negative Video
0.16
0.06
1.0
0.40
G = +2V/V
0.18
100M
2
−4
−4
0.20
10M
INVERTING OVERDRIVE RECOVERY
NONINVERTING OVERDRIVE RECOVERY
8
−4
1M
Frequency (Hz)
−1.0
−50
−25
0
25
50
75
Ambient Temperature (_C)
100
−20
125
Input Bias and Offset Current (µA)
4
CLOSED−LOOP OUTPUT IMPEDANCE vs FREQUENCY
1W Internal
Output Impedance (Ω )
5
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted.
COMMON−MODE INPUT RANGE AND OUTPUT SWING
vs SUPPLY VOLTAGE
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
26
24
Total Supply Current
Right Scale
Sourcing Output Current
Left Scale
90
22
80
20
Sinking Output Current
Left Scale
70
−50
−25
0
25
+VIN
4
−VIN
3
+VOUT
2
−VOUT
1
50
75
100
18
125
0
2.5
Ambient Temperature (_ C)
3.5
4.0
4.5
5.0
5.5
6.0
CROSSTALK vs FREQUENCY
−10
Common−Mode Input Impedance
−20
1M
Crosstalk (dB)
Input Impedance (Ω )
3.0
Supply Voltage (±VS)
COMMON−MODE AND DIFFERENTIAL
INPUT IMPEDANCE
10M
RL ≥ 500Ω
5
Voltage Range (V)
100
6
Supply Current (2mA/div)
Output Current (10mA/div)
110
100k
Differential Input Impedance
All Hostile Crosstalk
1VPP Output, 3 Channels, 100Ω Load
−30
−40
−50
−60
10k
Channel−to−Channel Crosstalk
1VPP Output, 1 Channel, 100ΩLoad
−70
−80
1k
100
1k
10k
100k
Frequency (Hz)
1M
10M
100M
1
10
100
Frequency (MHz)
9
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = +5V
RF = 402Ω, RL = 100Ω to VS/2, and G = +2, unless otherwise noted.
NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE
3
INVERTING SMALL−SIGNAL FREQUENCY RESPONSE
3
G = +1, RF = 25Ω
G = −1
0
−3
G = +5
−6
Normalized Gain (dB)
G = +2
G = +10
−9
−12
VO = 0.1VPP
RL = 100Ω
See Figure 3
−15
−18
1
−3
G = −10
−9
−12
VO = 0.1VPP
RL = 100Ω
See Figure 4
−15
−18
10
100
600
1
10
Frequency (MHz)
NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE
VO = 1VPP
VO = 1VPP
−3
VO = 2VPP
Gain (dB)
Gain (dB)
VO = 0.5VPP
0
3
0
VO = 3VPP
−6
VO = 2VPP
−6
VO = 3VPP
−9
−12
G = +2V/V
RL = 100Ω
See Figure 3
−9
−12
1
G = −1V/V
RL = 100Ω
See Figure 4
−15
−18
10
100
500
1
10
Frequency (MHz)
0
1.0
Small−Signal ± 100mV
Left Scale
0.5
0
−0.1
−0.5
−0.2
−1.0
−0.3
−1.5
−0.4
−2.0
Time (10ns/div)
10
1.5
Small−Signal Output Voltage (100mV/div)
0.2
Large−Signal Output Voltage (500mV/div)
Small−Signal Output Voltage (100mV/div)
Large−Signal ± 1V
Right Scale
500
INVERTING PULSE RESPONSE
2.0
G = +2V/V
See Figure 3
100
Frequency (MHz)
NONINVERTING PULSE RESPONSE
0.4
0.1
500
INVERTING LARGE−SIGNAL FREQUENCY RESPONSE
3
VO = 0.5VPP
6
0.3
100
Frequency (MHz)
9
−3
G = −2
G = −5
−6
0.4
0.3
2.0
G = −1V/V
See Figure 4
1.5
0.2
1.0
0.1
0.5
0
−0.1
−0.2
−0.3
Small−Signal ± 100mV
Left Scale
Large−Signal ± 1V
Right Scale
−0.4
0
−0.5
−1.0
−1.5
−2.0
Time (10ns/div)
Large−Signal Output Voltage (500mV/div)
Normalized Gain (dB)
0
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = +5V (continued)
RF = 402Ω, RL = 100Ω to VS/2, and G = +2, unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs LOAD RESISTANCE
−60
−60
G = +2V/V
f = 1MHz
VO = 2VPP
−70
−75
2nd−Harmonic
−80
−85
−90
3rd−Harmonic
−95
G = +2V/V
RL = 200Ω
VO = 2VPP
−65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−65
−70
2nd−Harmonic
−75
−80
−85
−90
3rd−Harmonic
−95
−100
−100
−105
−105
100
0.1
1000
1
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−60
−70
−75
−80
2nd−Harmonic
−85
−90
−95
3rd−Harmonic
−100
1
−70
−75
2nd−Harmonic
−80
−85
−90
3rd−Harmonic
−95
−100
−105
0.1
f = 1MHz
RL = 200Ω
VO = 2VPP
See Figure 3
−65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
HARMONIC DISTORTION vs NONINVERTING GAIN
−60
G = +2V/V
f = 1MHz
RL = 200Ω
See Figure 3
−65
10
−105
1
10
Output Voltage Swing (VPP)
Gain (V/V)
TWO−TONE, 3RD−ORDER
INTERMODULATION INTERCEPT
HARMONIC DISTORTION vs INVERTING GAIN
−65
Harmonic Distortion (dBc)
40
f = 1MHz
RL = 200Ω
VO = 2VPP
See Figure 4
−70
+5V
2nd−Harmonic
0.01µF
Intercept Point (+dBm)
−60
10
Frequency (MHz)
Resistance (Ω)
−75
−80
−85
3rd−Harmonic
−90
−95
35
806Ω
PI
57.6Ω
1/4
O PA4820
806Ω
PO
200Ω
402Ω
0.1µF
30
402Ω
0.1µF
25
20
−100
−105
1
10
Gain ( V/V )
15
0
5
10
15
20
25
30
Frequency (MHz)
11
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = +5V (continued)
RF = 402Ω, RL = 100Ω to VS/2, and G = +2, unless otherwise noted.
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
Normalized Gain to Capacitive Load (dB)
100
10
1
1
10
100
1000
C L = 10pF
7
6
5
CL = 22pF
4
CL = 47pF
3
CL = 100pF
2
+5V
0.01µF
1
806Ω
RS
VI
57.6Ω
0
806Ω
CL
−1
402Ω
−2
−3
1
10
Input Offset Voltage (VOS)
Left Scale
0
−0.5
−5
−10
Input Bias Current (IB)
Right Scale
−25
0
25
50
75
100
−15
125
Output Current (5mA/div)
5
Imput Bias and Offset Current (µA)
Input Offset Voltage (mV)
10
10x Input Offset Current (IOS)
Right Scale
Ambient Temperature (_C)
12
90
15
1.0
−1.5
−50
100
300
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
TYPICAL DC DRIFT OVER TEMPERATURE
−1.0
NOTE: (1) 1kΩ is optional.
0.01µF
Frequency (MHz)
1.5
0
1kΩ(1)
402Ω
Capacitive Load (pF)
0.5
VO
OPA820
23
85
Total Supply Current
Right Scale
80
75
70
21
20
Sinking Output Current
Left Scale
65
60
−50
22
19
Sourcing Output Current
Left Scale
−25
0
25
50
75
Ambient Temperature (_ C)
100
18
17
125
Supply Current (1mA/div)
RS (Ω)
0.1dB Peaking Targeted
8
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
APPLICATIONS INFORMATION
+5V
WIDEBAND VOLTAGE-FEEDBACK
OPERATION
Figure 1 shows the gain of +2 configuration used as the
basis for most of the Typical Characteristics. Most of the
curves were characterized using signal sources with 50Ω
driving impedance and with measurement equipment
presenting 50Ω load impedance. In Figure 1, the 50Ω
shunt resistor at the VI terminal matches the source
impedance of the test generator while the 50Ω series
resistor at the VO terminal provides a matching resistor for
the measurement equipment load. Generally, data sheet
specifications refer to the voltage swings at the output pin
(VO in Figure 1). The 100Ω load, combined with the 804Ω
total feedback network load, presents the OPA4820 with
an effective load of approximately 90Ω in Figure 1.
+5V
+VS
+
0.1µF
2.2µF
50Ω Source
VIN
50Ω
RS 50Ω Load
50Ω
VO
1/4
OPA4820
RF
402Ω
0.1µF
+
RG
402Ω
2.2µF
−VS
−5V
Figure 1. Gain of +2, High-Frequency Application
and Characterization Circuit
WIDEBAND INVERTING OPERATION
Operating the OPA4820 as an inverting amplifier has
several benefits and is particularly useful when a matched
50Ω source and input impedance is required. Figure 2
shows the inverting gain of −1 circuit used as the basis of
the inverting mode Typical Characteristics.
0.1µF
RT
205Ω
0.01µF
50Ω Source
1/4
OPA4820
RG
402Ω
2.2µF
50Ω
VO
50Ω Load
RF
402Ω
VI
RM
57.6Ω
0.1µF
+
Proper printed circuit board (PCB) layout and careful
component selection will maximize the performance of the
OPA4820 in all applications, as discussed in the following
sections of this data sheet.
+
The combination of speed and dynamic range offered by
the OPA4820 is easily achieved in a wide variety of
application circuits, providing that simple principles of
good design are observed. For example, good
power-supply decoupling, as shown in Figure 1, is
essential to achieve the lowest possible harmonic
distortion and smooth frequency response.
2.2µF
−5V
Figure 2. Inverting G = −1 Specifications and Test
Circuit
In the inverting case, just the feedback resistor appears as
part of the total output load in parallel with the actual load.
For the 100Ω load used in the Typical Characteristics, this
gives a total load of 80Ω in this inverting configuration. The
gain resistor is set to get the desired gain (in this case
402Ω for a gain of −1) while an additional input matching
resistor (RM) can be used to set the total input impedance
equal to the source if desired. In this case, RM = 57.6Ω in
parallel with the 402Ω gain setting resistor gives a
matched input impedance of 50Ω. This matching is only
needed when the input needs to be matched to a source
impedance, as in the characterization testing done using
the circuit of Figure 2.
The OPA4820 offers extremely good DC accuracy as well as
low noise and distortion. To take full advantage of that DC
precision, the total DC impedance looking out of each of the
input nodes must be matched to get bias current
cancellation. For the circuit of Figure 2, this requires the
205Ω resistor shown to ground on the noninverting input. The
calculation for this resistor includes a DC-coupled 50Ω
source impedance along with RG and RM. Although this
resistor will provide cancellation for the bias current, it must
be well decoupled (0.01µF in Figure 2) to filter the noise
contribution of the resistor and the input current noise.
As the required RG resistor approaches 50Ω at higher
gains, the bandwidth for the circuit in Figure 2 will far
exceed the bandwidth at that same gain magnitude for the
noninverting circuit of Figure 1. This occurs due to the
lower noise gain for the circuit of Figure 2 when the 50Ω
source impedance is included in the analysis. For
instance, at a signal gain of −10 (RG = 50Ω, RM = open,
RF = 499Ω) the noise gain for the circuit of Figure 2 will
be 1 + 499Ω/(50Ω + 50Ω) = 6 as a result of adding the 50Ω
source in the noise gain equation. This gives considerable
higher bandwidth than the noninverting gain of +10. Using
the 240MHz gain bandwidth product for the OPA4820, an
inverting gain of −10 from a 50Ω source to a 50Ω RG gives
42MHz bandwidth, whereas the noninverting gain of +10
gives 27MHz.
13
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
WIDEBAND SINGLE-SUPPLY OPERATION
while delivering more than 60mA output current giving
2.4V output swing into 100Ω (5.6dBm maximum at a
matched 50Ω load).
Figure 3 shows the AC-coupled, single +5V supply, gain of
+2V/V circuit configuration used as a basis for the +5V only
Electrical and Typical Characteristics. The key requirement for single-supply operation is to maintain input and
output signal swings within the useable voltage ranges at
both the input and the output. The circuit of Figure 3
establishes an input midpoint bias using a simple resistive
divider from the +5V supply (two 806Ω resistors) to the
noninverting input. The input signal is then AC-coupled
into this midpoint voltage bias. The input voltage can swing
to within 0.9V of the negative supply and 0.6V of the
positive supply, giving a 3.5VPP input signal range. The
input impedance matching resistor (57.6Ω) used in
Figure 3 is adjusted to give a 50Ω input match when the
parallel combination of the biasing divider network is
included. The gain resistor (RG) is AC-coupled, giving the
circuit a DC gain of +1. This puts the input DC bias voltage
(2.5V) on the output as well. On a single +5V supply, the
output voltage can swing to within 1.3V of either supply pin
Figure 4 shows the AC-coupled, single +5V supply, gain of
−1V/V circuit configuration used as a basis for the +5V only
Typical Characteristic curves. In this case, the midpoint
DC bias on the noninverting input is also decoupled with
an additional 0.01µF decoupling capacitor. This reduces
the source impedance at higher frequencies for the
noninverting input bias current noise. This 2.5V bias on the
noninverting input pin appears on the inverting input pin
and, since RG is DC blocked by the input capacitor, will
also appear at the output pin.
The single-supply test circuits of Figure 3 and Figure 4
show +5V operation. These same circuits can be used
over a single-supply range of +4V to +12.6V. Operating on
a single +12V supply, with the Absolute Maximum Supply
voltage specification of +13V, gives adequate design
margin for the typical ±5% supply tolerance.
+5V
+VS
+
0.1µF
50Ω Source
0.01µF
6.8µF
806Ω
DIS
VI
57.6Ω
VO
1/4
OPA4820
806Ω
100Ω
VS/2
RF
402Ω
RG
402Ω
0.01µF
Figure 3. AC-Coupled, G = +2V/V, Single-Supply Specifications and Test Circuit
+5V
+VS
+
0.1µF
6.8µF
806Ω
DIS
0.01µF
806Ω
RG
0.01µF 402Ω
1/4
OPA4820
VO
100Ω
VS/2
RF
402Ω
VI
Figure 4. AC-Coupled, G = −1V/V, Single-Supply Specifications and Test Circuit
14
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
DIFFERENTIAL INTERFACE APPLICATIONS
Dual and quad op amps are particularly suitable to
differential input to differential output applications.
Typically, these fall into either ADC input interface or line
driver applications. Two basic approaches to differential
I/O are noninverting or inverting configurations. Since the
output is differential, the signal polarity is somewhat
meaningless—the noninverting and inverting terminology
applies here to where the input is brought into the
OPA4820. Each has its advantages and disadvantages.
Figure 5 shows a basic starting point for noninverting
differential I/O applications.
Figure 6 shows a differential I/O stage configured as an
inverting amplifier. In this case, the gain resistors (RG)
become the input resistance for the source. This provides
a better noise performance than the noninverting
configuration, but does limit the flexibility in setting the
input impedance separately from the gain.
+VCC
VCM
1/4
OPA4820
+VCC
1/4
OPA4820
RG
RF
402Ω
RF
402Ω
VI
RG
RF
402Ω
VI
VO
VO
RG
RF
402Ω
1/4
OPA4820
1/4
OPA4820
−VCC
Figure 5. Noninverting Differential I/O Amplifier
This approach provides for a source termination
impedance that is independent of the signal gain. For
instance, simple differential filters may be included in the
signal path right up to the noninverting inputs without
interacting with the amplifier gain. The differential signal
gain for the circuit of Figure 5 is:
VO
+ AD + 1 ) 2
VI
RF
RG
(1)
Figure 5 shows the recommended value of 402Ω.
However, the gain may be adjusted using just the RG
resistor.
Various combinations of single-supply or AC-coupled
gains can also be delivered using the basic circuit of
Figure 5. Common-mode bias voltages on the two
noninverting inputs pass on to the output with a gain of 1
since an equal DC voltage at each inverting node creates
no current through RG, giving that voltage a commonmode gain of 1 to the output.
VCM
−VCC
Figure 6. Inverting Differential I/O Amplifier
The two noninverting inputs provide an easy
common-mode control input. This is particularly useful if
the source is AC-coupled through either blocking caps or
a transformer. In either case, the common-mode input
voltages on the two noninverting inputs again have a gain
of 1 to the output pins, giving an easy common-mode
control for single-supply operation. The input resistors
may be adjusted to the desired gain but will also be
changing the input impedance as well. The differential gain
for this circuit is:
VO
R
+* F
VI
RG
(2)
15
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
DC-COUPLED SINGLE-TO-DIFFERENTIAL
CONVERSION
stage, creating no current flow and placing the desired
VCM at the output of this stage as well. Both the positive
and negative output are shown in Figure 8.
The previous differential output circuits were set up to
receive a differential input as well as provide a differential
output. Figure 7 shows one way to provide a single to
differential conversion, with DC coupling, and independent
output common-mode control using a quad op amp.
The circuit of Figure 7 provides several useful features for
isolating the input signal from the final outputs. Using the
first amplifier as a simple noninverting stage gives an
independent adjustment on RI (to set the source loading)
while the gain can be easily adjusting in this stage using
the RG resistor. The next stage allows a separate output
common-mode level to be set up. The desired output
common-mode voltage, VCM, is cut in half and applied to
the noninverting input of the 2nd stage. The signal path in
this stage sees a gain of −1 while this (1/2 × VCM) voltage
sees a gain of +2. The output of this 2nd stage is then the
original common-mode voltage plus the inverted signal
from the output of the first stage. The 2nd stage output
appears directly at the output of the noninverting final
stage. The inverting node of the inverting output stage is
also biased to the common-mode voltage, equal to the
common-mode voltage appearing at the output of the 2nd
LOW-POWER, DIFFERENTIAL I/O,
4th-ORDER ACTIVE FILTER
The OPA4820 can give a very capable gain block for active
filters. The quad design lends itself very well to differential
active filters. Where the filter topology is looking for a
simple gain function to implement the filter, the
noninverting configuration is preferred to isolate the filter
elements from the gain elements in the design. Figure 9
shows an example of a 10MHz, 4th-order Butterworth,
low-pass Sallen-Key filter. The design places the higher Q
stage first to allow the lower Q 2nd stage to roll off the
peaked noise of the first stage. The resistor values have
been adjusted slightly to account for the amplifier group
delay.
While this circuit is bipolar, using ±5V supplies, it can easily
be adapted to single-supply operation. This will add two
real zeroes in the response, transforming this circuit into a
bandpass. The frequency response for the filter of Figure 9
is shown in Figure 10.
+5V
1.5V
VCM
200Ω
1/4
OPA4820
750Ω VCM
2
VI
RI
1/4
OPA4820
402Ω
RG
0.1µF
750Ω
750Ω
1/4
OPA4820
750Ω
402Ω
402Ω
+VOUT = VCM + VI (1 +
50Ω
200Ω
1/4
OPA4820
−VOUT = VCM − VI (1 +
250Ω
−5V
Figure 7. Precision, DC-Coupled, Single-to-Differential Conversion
16
402Ω
)
RG
402Ω
)
RG
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
3.0
VIN = 0V to 0.5V
RG = 402Ω
+VOUT
2.5
Voltage (V)
2.0
1.5
1.0
−VOUT
0.5
0
Time (10ns/div)
Figure 8. Pulse Response for Figure 7 Schematic
100pF
100pF
+5V
161Ω
77Ω
121Ω
294Ω
VO/VI = 4V/V
1/4
OPA4820
1/4
OPA4820
250Ω
VI
50pF
500Ω
250Ω
161Ω
VO
250Ω
1/4
OPA4820
121Ω
PD = 225mW
250Ω
500Ω
50pF
f−3dB = 10MHz
77Ω
100pF
294Ω
1/4
OPA4820
100pF
GD = 2, ωO = 2π 10MHz, Q = 1.31
−5V
GD = 2, ωO = 2π 10MHz, Q = 0.54
Figure 9. Low-Power, Differential I/O, 4th-Order Butterworth Active Filter
LOW-POWER xDSL TRANSCEIVER
INTERFACE
15
Differential Gain (dB)
12
9
6
3
0
−3
−6
−9
1
10
Frequency (MHz)
Figure 10. Differential 4th-Order, 10MHz
Butterworth Filter
100
With four amplifiers available, the quad OPA4820 can
meet the needs for both differential driver and receiver in
a low-power xDSL line interface design. A simplified
design example is shown on the front page. Two amplifiers
are used as a noninverting differential driver while the
other two implement the driver echo cancellation and
receiver amplifier function. This example shows a single
+12V design where the drive side is taking a 2VPP
maximum input from the transmit filter and providing a
differential gain of 7, giving a maximum 14VPP differential
output swing. This is coupled through 50Ω matching
resistors and a 1:1 transformer to give a maximum 7VPP
on a 100Ω line. This 7VPP corresponds to a 10dBm line
power with a 3.5 crest factor.
17
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
The differential receiver is configured as an inverting
summing stage where the outputs of the driver are
cancelled prior to appearing at the output of the receive
amplifiers. This is done by summing the output voltages for
the drive amplifiers and their attenuated and inverted
levels (at the transformer input) into the inverting inputs of
each receiver amplifier. The resistor values are set (see
the circuit on the front page) to give perfect drive signal
cancellation if the drive signal is attenuated by 1/2, going
from the drive amplifier outputs to the transformer input.
The signal received through the transformer has a gain of
1 through the receive amplifiers. Higher gain could easily
be provided by scaling the resistors summing into the
inverting inputs of the receiver amplifiers down while
keeping the same ratio between them.
DUAL-CHANNEL, DIFFERENTIAL ADC DRIVER
Where a low-noise, single-supply, interface to a differential
input +5V ADC is required, the circuit of Figure 11 can
provide a high dynamic range, medium gain interface for
dual high-performance ADCs. The circuit of Figure 11
uses two amplifiers in the differential inverting
configuration. The common-mode voltage is set on the
noninverting inputs to the supply midscale. In this
example, the input signal is coupled in through a 1:2
transformer. This provides both signal gain, single to
differential conversion, and a reduction in noise figure. To
show a 50Ω input impedance at the input to the
transformer, two 200Ω resistors are required on the
transformer secondary. These two resistors are also the
amplifier gain elements. Since the same DC voltage
appears on both inverting nodes in the circuit of Figure 11,
no DC current will flow through the transformer, giving a
DC gain of 1 to the output for this common-mode voltage,
VCM.
The circuit of Figure 11 is particularly suitable for a
moderate resolution dual ADC used as I/Q samplers. The
optional 500Ω resistors to ground on each amplifier output
can be added to improve the 2nd- and 3rd-harmonic
distortion by > 15dB if higher dynamic range is required.
18
The 5mA added output stage current significantly
improves linearity if that is required. The measured
2nd-harmonic distortion is consistently lower than the
3rd-harmonics for this balanced differential design. It is
particularly helpful for this low-power design if there are no
grounds in the signal path after the low-level signal at the
transformer input. The two pull-down resistors do show a
signal path ground and should be connected at the same
physical point to ground, in order to eliminate imbalanced
ground return currents from degrading 2nd-harmonic
distortion.
VIDEO LINE DRIVING
Most video distribution systems are designed with 75Ω
series resistors to drive a matched 75Ω cable. In order to
deliver a net gain of 1 to the 75Ω matched load, the
amplifier is typically set up for a voltage gain of +2,
compensating for the 6dB attenuation of the voltage
divider formed by the series and shunt 75Ω resistors at
either end of the cable.
The circuit of Figure 1 applies to this requirement if all
references to 50Ω resistors are replaced by 75Ω values.
Often, the amplifier gain is further increased to 2.2, which
recovers the additional DC loss of a typical long cable run.
This change would require the gain resistor (RG) in
Figure 1 to be reduced from 402Ω to 335Ω. In either case,
both the gain flatness and the differential gain/phase
performance of the OPA4820 will provide exceptional
results in video distribution applications. Differential gain
and phase measure the change in overall small-signal
gain and phase for the color sub-carrier frequency
(3.58MHz in NTSC systems) versus changes in the
large-signal output level (which represents luminance
information in a composite video signal). The OPA4820,
with the typical 150Ω load of a single matched video cable,
shows less than 0.003%/0.06° differential gain/phase
errors over the standard luminance range for a positive
video (negative sync) signal. Similar performance would
be observed for multiple video signals (see Figure 12).
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
+5V
1kΩ
VCM
0.1µF
1kΩ
1/4
OPA4820
Dual ADC
500Ω
1:2
50Ω
Source
200Ω
800Ω
RS
200Ω
800Ω
RS
16.7dB
Noise Figure
CL
1 of 2
Channels
1/4
OPA4820
Gain = 8V/V
18dB
500Ω
VCM
Figure 11. Single-Supply Differential ADC Driver (1 of 2 channels)
335Ω
402Ω
75Ω Transmission Line
75Ω
1/4
OPA4820
VOUT
Video
Input
75Ω
75Ω
75Ω
VOUT
75Ω
75Ω
High output current drive capability allows three
back−terminated 75Ω transmission lines to be simultaneously driven.
VOUT
75Ω
Figure 12. Video Distribution Amplifier
SINGLE OP AMP DIFFERENTIAL AMPLIFIER
The voltage-feedback architecture of the OPA4820, with
its high common-mode rejection ratio (CMRR), will provide
exceptional performance in differential amplifier configurations. Figure 13 shows a typical configuration. The starting
point for this design is the selection of the RF value in the
range of 200Ω to 2kΩ. Lower values reduce the required
RG, increasing the load on the V2 source and on the
OPA4820 output. Higher values increase output noise as
well as the effects of parasitic board and device
capacitances. Following the selection of RF, RG must be
set to achieve the desired inverting gain for V2. Remember
that the bandwidth will be set approximately by the gain
bandwidth product (GBP) divided by the noise gain
(1 + RF/RG). For accurate differential operation (that is,
good CMRR), the ratio R2/R1 must be set equal to RF/RG.
+5V
Power−supply decoupling not shown.
R1
V1
R2
1/4
OPA4820
RG
RF
V2
50Ω
VO =
when
RF
(V − V2)
RG 1
R2 R F
=
R1 R G
−5V
Figure 13. High-Speed, Single Differential
Amplifier
19
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Usually, it is best to set the absolute values of R2 and R1
equal to RF and RG, respectively; this equalizes the divider
resistances and cancels the effect of input bias currents.
However, it is sometimes useful to scale the values of R2
and R1 in order to adjust the loading on the driving source,
V1. In most cases, the achievable low-frequency CMRR
will be limited by the accuracy of the resistor values. The
85dB CMRR of the OPA4820 itself will not determine the
overall circuit CMRR unless the resistor ratios are
matched to better than 0.003%. If it is necessary to trim the
CMRR, then R2 is the suggested adjustment point.
4-CHANNEL DAC TRANSIMPEDANCE
AMPLIFIER
High-frequency Digital-to-Analog Converters (DACs)
require a low-distortion output amplifier to retain their
SFDR performance into real-world loads. See Figure 14
for a single-ended output drive implementation. In this
circuit, only one side of the complementary output drive
signal is used. The diagram shows the signal output
current connected into the virtual ground-summing
junction of the OPA4820, which is set up as a
transimpedance stage or I-V converter. The unused
current output of the DAC is connected to ground. If the
DAC requires its outputs to be terminated to a compliance
voltage other than ground for operation, then the
appropriate voltage level may be applied to the
noninverting input of the OPA4820.
1/4
OPA4820
High−Speed
DAC
VO = ID RF
RF
peaking. To achieve a flat transimpedance frequency
response, this pole in the feedback network should be set
to:
1
+
2pR FCF
GBP
Ǹ4pR
C
F
(3)
D
which will give a corner frequency f−3dB of approximately:
f *3dB +
GBP
Ǹ2pR
C
F
(4)
D
ACTIVE FILTERS
Most active filter topologies will have exceptional
performance using the broad bandwidth and unity-gain
stability of the OPA4820. Topologies employing capacitive
feedback require a unity-gain stable, voltage-feedback op
amp. Sallen-Key filters simply use the op amp as a
noninverting gain stage inside an RC network. Either
current- or voltage-feedback op amps may be used in
Sallen-Key implementations.
Figure 15 shows an example Sallen-Key low-pass filter, in
which the OPA4820 is set up to deliver a low-frequency
gain of +2. The filter component values have been
selected to achieve a maximally-flat Butterworth response
with a 5MHz, −3dB bandwidth. The resistor values have
been slightly adjusted to compensate for the effects of the
240MHz bandwidth provided by the OPA4820 in this
configuration. This filter may be combined with the ADC
driver suggestions to provide moderate (2-pole) Nyquist
filtering, limiting noise, and out-of-band harmonics into the
input of an ADC. This filter will deliver the exceptionally low
harmonic distortion required by high SFDR ADCs such as
the ADS850 (14-bit, 10MSPS, 82dB SFDR).
CF
ID
CD
C1
150pF
GBP →Gain Bandwidth
Product (Hz) for the OPA4820.
+5V
R1
124Ω
ID
R2
505Ω
V1
C2
100pF
Figure 14. Wideband, Low-Distortion DAC
Transimpedance Amplifier
The DC gain for this circuit is equal to RF. At high
frequencies, the DAC output capacitance (CD) will
produce a zero in the noise gain for the OPA4820 that may
cause peaking in the closed-loop frequency response. CF
is added across RF to compensate for this noise-gain
20
1/4
OPA4820
VO
RF
402Ω
Power−supply
decoupling not shown.
−5V
RG
402Ω
Figure 15. 5MHz Butterworth Low-Pass Active
Filter
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SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
Another type of filter, a high-Q bandpass filter, is shown in
Figure 16. The transfer function for this filter is:
R )R
s R 3R C4
V OUT
1 4 1
+
1
V IN
2
s ) sR C ) R
1 1
with
and
R3
(5)
2R 4R 5C 1C 2
R3
w O2 +
R 2R 4R 5C 1C 2
wO
+ 1
Q
R 1C 1
(6)
and
DEMONSTRATION FIXTURES
Two printed circuit boards (PCBs) are available to assist
in the initial evaluation of circuit performance using the
OPA4820 in its two package options. Both of these are
offered free of charge as unpopulated PCBs, delivered
with a user’s guide. The summary information for these
fixtures is shown in the table below.
(7)
PRODUCT
For the values chosen in Figure 16:
wO
] 1MHz
2p
fO +
DESIGN-IN TOOLS
(8)
PACKAGE
ORDERING NUMBER
LITERATURE
NUMBER
OPA4820ID
SO-14
DEM-OPA-SO-4A
SBOU016
OPA4820IPW
TSSOP-14
DEM-OPA-TSSOP-4A
SBOU017
Q = 100
See Figure 17 for the frequency response of the filter
shown in Figure 16.
MACROMODELS AND APPLICATIONS
SUPPORT
R3
500Ω
1/4
OPA4820
C2
1000pF
R1
15.8kΩ
R2
158Ω
R4
500Ω
R5
158Ω
1/4
OPA4820
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA4820
product folder.
VOUT
VIN
C1
1000pF
Figure 16. High-Q 1MHz Bandpass Filter
Computer simulation of circuit performance using SPICE
is often a quick way to analyze the performance of the
OPA4820 and its circuit designs. This is particularly true for
video and RF amplifier circuits where parasitic capacitance and inductance can play a major role on circuit
performance. A SPICE model for the OPA4820 is
available through the TI web page (www.ti.com). The
applications department is also available for design
assistance. These models predict typical small-signal AC,
transient steps, DC performance, and noise under a wide
variety of operating conditions. The models include the
noise terms found in the electrical specifications of the
data sheet. These models do not attempt to distinguish
between the package types in their small-signal AC
performance.
Gain (dB)
OPERATING SUGGESTIONS
OPTIMIZING RESISTOR VALUES
6
0
−6
−12
−18
−24
−30
−36
−42
−48
−54
−60
−66
−72
100k
1M
10M
Frequency (Hz)
Figure 17. High-Q 1MHz Bandpass Filter
Frequency Response
100M
Since the OPA4820 is a unity-gain stable, voltage-feedback op amp, a wide range of resistor values may be used
for the feedback and gain-setting resistors. The primary
limits on these values are set by dynamic range (noise and
distortion) and parasitic capacitance considerations. Usually, the feedback resistor value should be between 200Ω
and 1kΩ. Below 200Ω, the feedback network will present
additional output loading which can degrade the harmonic
distortion performance of the OPA4820. Above 1kΩ, the
typical parasitic capacitance (approximately 0.2pF)
across the feedback resistor may cause unintentional
band limiting in the amplifier response. A 25Ω feedback
resistor is suggested for AV = +1V/V.
A good rule of thumb is to target the parallel combination
of RF and RG (see Figure 1) to be less than about 200Ω.
The combined impedance RF || RG interacts with the
inverting input capacitance, placing an additional pole in
the feedback network, and thus a zero in the forward
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response. Assuming a 2pF total parasitic on the inverting
node, holding RF || RG < 200Ω will keep this pole above
400MHz. By itself, this constraint implies that the feedback
resistor RF can increase to several kΩ at high gains. This
is acceptable as long as the pole formed by RF and any
parasitic capacitance appearing in parallel is kept out of
the frequency range of interest.
In the inverting configuration, an additional design
consideration must be noted. RG becomes the input
resistor and therefore the load impedance to the driving
source. If impedance matching is desired, RG may be set
equal to the required termination value. However, at low
inverting gains, the resulting feedback resistor value can
present a significant load to the amplifier output. For
example, an inverting gain of 2 with a 50Ω input matching
resistor (= RG) would require a 100Ω feedback resistor,
which would contribute to output loading in parallel with the
external load. In such a case, it would be preferable to
increase both the RF and RG values, and then achieve the
input matching impedance with a third resistor to ground
(see Figure 2). The total input impedance becomes the
parallel combination of RG and the additional shunt
resistor.
BANDWIDTH vs GAIN
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the GBP shown in the
specifications. Ideally, dividing GBP by the noninverting
signal gain (also called the noise gain, or NG) will predict
the closed-loop bandwidth. In practice, this only holds true
when the phase margin approaches 90°, as it does in
high-gain configurations. At low signal gains, most
amplifiers will exhibit a more complex response with lower
phase margin. The OPA4820 is optimized to give a
maximally-flat, 2nd-order Butterworth response in a gain
of 2. In this configuration, the OPA4820 has approximately
64° of phase margin and will show a typical −3dB
bandwidth of 240MHz. When the phase margin is 64°, the
closed-loop bandwidth is approximately √2 greater than
the value predicted by dividing GBP by the noise gain.
Increasing the gain will cause the phase margin to
approach 90° and the bandwidth to more closely approach
the predicted value of (GBP/NG). At a gain of +10, the
27MHz bandwidth shown in the Electrical Characteristics
agrees with that predicted using the simple formula and
the typical GBP of 250MHz.
OUTPUT DRIVE CAPABILITY
The OPA4820 has been optimized to drive the demanding
load of a doubly-terminated transmission line. When a 50Ω
line is driven, a series 50Ω into the cable and a terminating
50Ω load at the end of the cable are used. Under these
conditions, the cable impedance will appear resistive over
a wide frequency range, and the total effective load on the
OPA4820 is 100Ω in parallel with the resistance of the
feedback network. The electrical characteristics show a
22
±3.6V swing into this load—which will then be reduced to
a ±1.8V swing at the termination resistor. The ±75mA
output drive over temperature provides adequate current
drive margin for this load. Higher voltage swings (and
lower distortion) are achievable when driving higher
impedance loads.
A single video load typically appears as a 150Ω load (using
standard 75Ω cables) to the driving amplifier. The
OPA4820 provides adequate voltage and current drive to
support up to three parallel video loads (50Ω total load) for
an NTSC signal. With only one load, the OPA4820
achieves an exceptionally low 0.01%/0.03° dG/dP error.
DRIVING CAPACITIVE LOADS
One of the most demanding, and yet very common, load
conditions for an op amp is capacitive loading. A
high-speed, high open-loop gain amplifier like the
OPA4820 can be very susceptible to decreased stability
and closed-loop response peaking when a capacitive load
is placed directly on the output pin. In simple terms, the
capacitive load reacts with the open-loop output
resistance of the amplifier to introduce an additional pole
into the loop and thereby decrease the phase margin. This
issue has become a popular topic of application notes and
articles, and several external solutions to this problem
have been suggested. When the primary considerations
are frequency response flatness, pulse response fidelity,
and/or distortion, the simplest and most effective solution
is to isolate the capacitive load from the feedback loop by
inserting a series isolation resistor between the amplifier
output and the capacitive load. This does not eliminate the
pole from the loop response, but rather shifts it and adds
a zero at a higher frequency. The additional zero acts to
cancel the phase lag from the capacitive load pole, thus
increasing the phase margin and improving stability.
The Typical Characteristics show the recommended RS vs
Capacitive Load and the resulting frequency response at
the load. The criterion for setting the recommended
resistor is maximum bandwidth, flat frequency response at
the load. Since there is now a passive low-pass filter
between the output pin and the load capacitance, the
response at the output pin itself is typically somewhat
peaked, and becomes flat after the roll-off action of the RC
network. This is not a concern in most applications, but can
cause clipping if the desired signal swing at the load is very
close to the amplifier’s swing limit. Such clipping would be
most likely to occur in pulse response applications where
the frequency peaking is manifested as an overshoot in the
step response.
Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA4820. Long PCB
traces, unmatched cables, and connections to multiple
devices can easily cause this value to be exceeded.
Always consider this effect carefully, and add the
recommended series resistor as close as possible to the
OPA4820 output pin (see the Board Layout section).
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DISTORTION PERFORMANCE
The OPA4820 is capable of delivering an exceptionally low
distortion signal at high frequencies and low gains. The
distortion plots in the Typical Characteristics show the
typical distortion under a wide variety of conditions. Most
of these plots are limited to 100dB dynamic range. The
OPA4820 distortion does not rise above −90dBc until
either the signal level exceeds 0.9V and/or the fundamental frequency exceeds 500kHz. Distortion in the audio
band is ≤ −100dBc.
Generally, until the fundamental signal reaches very high
frequencies or powers, the 2nd-harmonic will dominate the
distortion with a negligible 3rd-harmonic component.
Focusing then on the 2nd-harmonic, increasing the load
impedance improves distortion directly. Remember that
the total load includes the feedback network—in the
noninverting configuration this is the sum of RF + RG,
whereas in the inverting configuration this is just RF (see
Figure 1). Increasing the output voltage swing increases
harmonic distortion directly. Increasing the signal gain will
also increase the 2nd-harmonic distortion. Again, a 6dB
increase in gain will increase the 2nd- and 3rd-harmonic by
6dB even with a constant output power and frequency.
Finally, the distortion increases as the fundamental
frequency increases because of the roll-off in the loop gain
with frequency. Conversely, the distortion will improve
going to lower frequencies down to the dominant
open-loop pole at approximately 100kHz. Starting from the
−85dBc 2nd-harmonic for 2VPP into 200Ω, G = +2
distortion at 1MHz (from the Typical Characteristics), the
2nd-harmonic distortion will not show any improvement
below 100kHz and will then be:
−85dB − 20log (1MHz/100kHz) = −105dBc
NOISE PERFORMANCE
The OPA4820 complements its low harmonic distortion
with low input noise terms. Both the input-referred voltage
noise and the two input-referred current noise terms
combine to give a low output noise under a wide variety of
operating conditions. Figure 18 shows the op amp noise
analysis model with all the noise terms included. In this
model, all the noise terms are taken to be noise voltage or
current density terms in either nV/√Hz or pA/√Hz.
The total output spot noise voltage is computed as the
square root of the squared contributing terms to the output
noise voltage. This computation is adding all the contributing noise powers at the output by superposition, then
taking the square root to get back to a spot noise voltage.
Equation 9 shows the general form for this output noise
voltage using the terms presented in Figure 18.
EO +
Ǹƪ
ƫ
E2NI ) ǒI BNR SǓ ) 4kTR S NG 2 ) ǒI BIR FǓ ) 4kTR FNG
2
2
(9)
ENI
1/4
OPA4820
RS
EO
IBN
ERS
RF
√4kTRS
RG
4kT
RG
√4kTRF
I BI
4kT = 1.6E − 20J
at 290_ K
Figure 18. Op Amp Noise Analysis Model
Dividing this expression by the noise gain (NG = 1 +
RF/RG) will give the equivalent input referred spot noise
voltage at the noninverting input, as shown in Equation 10.
EN +
Ǹ
E2NI ) ǒIBNRSǓ ) 4kTR S )
2
ǒ Ǔ
IBIRF
NG
2
)
4kTR F
NG
(10)
Evaluating these two equations for the OPA4820 circuit
presented in Figure 1 will give a total output spot noise
voltage of 6.44nV/√Hz and an equivalent input spot noise
voltage of 3.22nV/√Hz.
DC OFFSET CONTROL
The OPA4820 can provide excellent DC signal accuracy
because of its high open-loop gain, high common-mode
rejection, high power-supply rejection, and low input offset
voltage and bias current offset errors. To take full
advantage of this low input offset voltage, careful attention
to input bias current cancellation is also required. The
high-speed input stage for the OPA4820 has a moderately
high input bias current (9µA typ into the pins) but with a
very
close
match
between
the
two
input
currents—typically 100nA input offset current. The total
output offset voltage may be considerably reduced by
matching the source impedances looking out of the two
inputs. For example, one way to add bias current
cancellation to the circuit of Figure 1 would be to insert a
175Ω series resistor into the noninverting input from the
50Ω terminating resistor. When the 50Ω source resistor is
DC-coupled, this will increase the source impedance for
the noninverting input bias current to 200Ω. Since this is
now equal to the impedance looking out of the inverting
input (RF || RG), the circuit will cancel the gains for the bias
currents to the output leaving only the offset current times
the feedback resistor as a residual DC error term at the
output. Using a 402Ω feedback resistor, this output error
will now be less than ±0.5µA × 402Ω = ±208µV at 25°C.
23
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THERMAL ANALYSIS
The OPA4820 will not require heatsinking or airflow in
most applications. Maximum desired junction temperature
would set the maximum allowed internal power dissipation
as described below. In no case should the maximum
junction temperature be allowed to exceed +150°C.
Operating junction temperature (TJ) is given by
TA + PD × qJA. The total internal power dissipation (PD)
is the sum of quiescent power (PDQ) and additional power
dissipated in the output stage (PDL) to deliver load power.
Quiescent power is simply the specified no-load supply
current times the total supply voltage across the part. PDL
will depend on the required output signal and load but
would, for a grounded resistive load, be at a maximum
when the output is fixed at a voltage equal to 1/2 of either
supply voltage (for equal bipolar supplies). Under this
worst-case condition, PDL = VS2/(4 × RL), where RL
includes feedback network loading.
Note that it is the power in the output stage and not in the
load that determines internal power dissipation.
As a worst-case example, compute the maximum TJ using
all channels of an OPA4820IPW (TSSOP-14 package) in
the circuit of Figure 1 operating at the maximum specified
ambient temperature of +85°C.
PD = 10V(25.8mA) + 4 × [52/(4 × (100Ω || 800Ω))] = 539mW
Maximum TJ = +85°C + (539mW × 110°C/W) = 144°C
This maximum operating junction temperature is below the
absolute maximum junction temperature. Most junction
temperatures in applications will be lower since an
absolute worst-case output stage power was assumed in
this calculation.
BOARD LAYOUT
Achieving optimum performance with a high-frequency
amplifier such as the OPA4820 requires careful attention
to board layout parasitics and external component types.
Recommendations that will optimize performance include:
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output and inverting input pins can cause instability: on the
noninverting input, it can react with the source impedance
to cause unintentional bandlimiting. To reduce unwanted
capacitance, a window around the signal I/O pins should
be opened in all of the ground and power planes around
those pins. Otherwise, ground and power planes should
be unbroken elsewhere on the board.
24
b) Minimize the distance (< 0.25”) from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane
layout should not be in close proximity to the signal I/O
pins. Avoid narrow power and ground traces to minimize
inductance between the pins and the decoupling
capacitors. The power-supply connections should always
be decoupled with these capacitors. Larger (2.2µF to
6.8µF) decoupling capacitors, effective at lower frequency,
should also be used on the main supply pins. These may
be placed somewhat farther from the device and may be
shared among several devices in the same area of the
PCB.
c) Careful selection and placement of external
components will preserve the high-frequency performance of the OPA4820. Resistors should be a very low
reactance type. Surface-mount resistors work best and
allow a tighter overall layout. Metal-film and carbon
composition, axially leaded resistors can also provide
good high-frequency performance. Again, keep their leads
and PCB trace length as short as possible. Never use
wire-wound type resistors in a high-frequency application.
Since the output pin and inverting input pin are the most
sensitive to parasitic capacitance, always position the
feedback and series output resistor, if any, as close as
possible to the output pin. Other network components,
such as noninverting input termination resistors, should
also be placed close to the package. Where double-side
component mounting is allowed, place the feedback
resistor directly under the package on the other side of the
board between the output and inverting input pins. Even
with a low parasitic capacitance shunting the external
resistors, excessively high resistor values can create
significant time constants that can degrade performance.
Good axial metal-film or surface-mount resistors have
approximately 0.2pF in shunt with the resistor. For resistor
values > 1.5kΩ, this parasitic capacitance can add a pole
and/or a zero below 500MHz that can effect circuit
operation. Keep resistor values as low as possible
consistent with load-driving considerations. It has been
suggested here that a good starting point for design would
be to set RG || RF = 200Ω. Using this setting will
automatically keep the resistor noise terms low, and
minimize the effect of their parasitic capacitance.
"#$%
www.ti.com
SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
d) Connections to other wideband devices on the
board may be made with short direct traces or through
onboard transmission lines. For short connections,
consider the trace and the input to the next device as a
lumped capacitive load. Relatively wide traces (50mils to
100mils) should be used, preferably with ground and
power planes opened up around them. Estimate the total
capacitive load and set RS from the plot of Recommended
RS vs Capacitive Load. Low parasitic capacitive loads
(< 5pF) may not need an RS since the OPA4820 is
nominally compensated to operate with a 2pF parasitic
load. Higher parasitic capacitive loads without an RS are
allowed as the signal gain increases (increasing the
unloaded phase margin). If a long trace is required, and the
6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance
transmission line using microstrip or stripline techniques
(consult an ECL design handbook for microstrip and
stripline layout techniques). A 50Ω environment is
normally not necessary onboard, and in fact, a higher
impedance environment will improve distortion as shown
in the distortion versus load plots. With a characteristic
board trace impedance defined based on board material
and trace dimensions, a matching series resistor into the
trace from the output of the OPA4820 is used as well as a
terminating shunt resistor at the input of the destination
device. Remember also that the terminating impedance
will be the parallel combination of the shunt resistor and
input impedance of the destination device; this total
effective impedance should be set to match the trace
impedance. If the 6dB attenuation of a doubly-terminated
transmission line is unacceptable, a long trace can be
series-terminated at the source end only. Treat the trace as
a capacitive load in this case and set the series resistor
value as shown in the plot of RS vs Capacitive Load. This
will not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of the
destination device is low, there will be some signal
attenuation due to the voltage divider formed by the series
output into the terminating impedance.
e) Socketing a high-speed part like the OPA4820 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an
extremely troublesome parasitic network, which can make
it almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the
OPA4820 onto the board.
INPUT AND ESD PROTECTION
The OPA4820 is built using a very high-speed
complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very small
geometry devices. These breakdowns are reflected in the
Absolute Maximum Ratings table. All device pins are
protected with internal ESD protection diodes to the power
supplies, as shown in Figure 19.
+VCC
External
Pin
−VCC
Figure 19. Internal ESD Protection
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA continuous
current. Where higher currents are possible (for example,
in systems with ±15V supply parts driving into the
OPA4820), current-limiting series resistors should be
added into the two inputs. Keep these resistor values as
low as possible since high values degrade both noise
performance and frequency response. Figure 20 shows
an example protection circuit for I/O voltages that may
exceed the supplies.
+5V
50Ω Source
Power−supply
decoupling not shown.
174Ω
V1
50Ω
50Ω D1
D2
1/4
OPA4820
RF
301Ω
50Ω
RG
301Ω
VO
−5V
D1 = D2 IN5911 (or equivalent)
Figure 20. Gain of +2 with Input Protection
25
www.ti.com
SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008
Revision History
DATE
REV
PAGE
SECTION
8/08
D
2
Absolute Maximum Ratings
21
Design-In Tools
6/06
C
26
Application Information
DESCRIPTION
Changed Storage Temperature minimum value from −40°C to −65°C.
Demonstration fixture numbers changed.
Added Revision History table.
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
26
PACKAGE OPTION ADDENDUM
www.ti.com
24-Aug-2018
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
OPA4820ID
ACTIVE
SOIC
D
14
50
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA4820
OPA4820IDG4
ACTIVE
SOIC
D
14
50
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA4820
OPA4820IDR
ACTIVE
SOIC
D
14
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA4820
OPA4820IPWR
ACTIVE
TSSOP
PW
14
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
4820
OPA4820IPWT
ACTIVE
TSSOP
PW
14
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
4820
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
24-Aug-2018
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
OPA4820IDR
SOIC
D
14
2500
330.0
16.4
6.5
9.0
2.1
8.0
16.0
Q1
OPA4820IPWR
TSSOP
PW
14
2500
330.0
12.4
6.9
5.6
1.6
8.0
12.0
Q1
OPA4820IPWT
TSSOP
PW
14
250
180.0
12.4
6.9
5.6
1.6
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
OPA4820IDR
SOIC
D
14
2500
367.0
367.0
38.0
OPA4820IPWR
TSSOP
PW
14
2500
367.0
367.0
35.0
OPA4820IPWT
TSSOP
PW
14
250
210.0
185.0
35.0
Pack Materials-Page 2
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