Texas Instruments | OA-16 Wideband AGC Amplifier as a Differential Amplifier (Rev. B) | Application notes | Texas Instruments OA-16 Wideband AGC Amplifier as a Differential Amplifier (Rev. B) Application notes

Texas Instruments OA-16 Wideband AGC Amplifier as a Differential Amplifier (Rev. B) Application notes
OBSOLETE
Application Report
SNOA393B – January 1993 – Revised April 2013
OA-16 Wideband AGC Amplifier as a Differential Amplifier
ABSTRACT
This application report discusses the use of a wideband AGC amplifier as a differential amplifier.
1
2
3
4
Contents
Introduction .................................................................................................................. 2
Application Hints ............................................................................................................ 6
2.1
Improving CMRR ................................................................................................... 6
2.2
Setting the Differential Gain ...................................................................................... 9
2.3
Using the Gain Adjust Pin ........................................................................................ 9
2.4
Input Noise ........................................................................................................ 10
Application Suggestion ................................................................................................... 10
3.1
Wideband Differential Coax Line Receiver .................................................................... 10
3.2
VIdeo Loop-Through Amplifier .................................................................................. 11
3.3
Very Wideband Pulse-Differencing Amplifier ................................................................. 12
3.4
Alternative Wideband Differential Amplifiers .................................................................. 12
Conclusion .................................................................................................................. 13
List of Figures
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
..........................................................................................
CLC520 Fixed-Gain Differential Amplifier Configuration ...............................................................
Single-Ended Gain and Phase ............................................................................................
Input-Referred Common-Mode Error Model .............................................................................
Differential Gain Test Circuit...............................................................................................
Differential Gain and Phase ...............................................................................................
Common-Mode Rejection Ratio of the Circuit in .......................................................................
Differential Amplifier with Inverting Response Compensation ........................................................
Single-Ended Gains with Inverting Path Compensation ...............................................................
Improved CMRR with Better Response-Match Over Frequency .....................................................
Input Noise Voltage .......................................................................................................
Differential Coax Line Receiver ..........................................................................................
Video Loop-Through Connection Using A Wideband Differential Amplifier........................................
Very Wideband Single Input Pulse Response .........................................................................
Single Amplifier Differential Amplifier With Input Buffering...........................................................
CLC520 Internal Block Diagram
2
3
3
4
5
5
6
7
9
9
10
10
11
12
12
All trademarks are the property of their respective owners.
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
1
OBSOLETE
Introduction
1
www.ti.com
Introduction
The CLC520 is a very flexible DC-coupled Automatic Gain Control amplifier (AGC). Unique features
include two closely-matched differential inputs, a wideband gain control channel (100 MHz), and a ground
referenced DC-coupled output signal driven from a low output impedance amplifier. Figure 1 illustrates the
internal block diagram and pin assignments of the CLC520.
As shown in Figure 1, two unity-gain closed-loop input buffers on pins 3 and 6 are used to force the two
input voltages to appear across the external resistor, Rg. The differential voltage across Rg generates a
signal current which is amplified by a factor of 1.85 and fed into a two quadrant multiplier stage. The gainadjustment voltage on pin 2 determines how much of this signal current makes it through the multiplier
stage, with the remainder of the signal current being shunted to ground. The multiplier's output current
then flows through the transimpedance amplifier formed by the external feedback resistor, Rf, and the
internal amplifier. If the non-inverting input of this output amplifier, VREF, is tied to ground then a groundreferenced DC-coupled replica of the differential voltage across Rg appears at the output of the op amp.
The values of Rf and Rg, along with the gain-adjust voltage, determine the gain. Refer to the CLC520 data
sheet for a more complete operational and performance discussion.
Figure 1. CLC520 Internal Block Diagram
2
OA-16 Wideband AGC Amplifier as a Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
Copyright © 1993–2013, Texas Instruments Incorporated
OBSOLETE
Introduction
www.ti.com
In order to implement a fixed-gain differential amplifier, the CLC520 will rely on R's very well-matched
input buffers and it's differential-to-single-ended voltage conversion. For the purposes of this discussion,
the gain-control input will be held at a fixed level to yield the maximum gain given by 1.85 × Rf/Rg. Thus,
the differential signal gain depends only on the ratio of two external resistors and the internal currentmirror gain. Both Rf and Rg can be adjusted to yield a wide range of differential gains. As an example, the
circuit of Figure 2 is used to demonstrate the performance of the CLC520 in a fixed-gain differential
amplifier configuration.
Figure 2. CLC520 Fixed-Gain Differential Amplifier Configuration
To demonstrate this application, the CLC520 is set up for a gain of 4.08V/V. The 50Ω impedancematching resistor at the output effectively halves the differential gain to 2.04V/V (6.2 dB) at the 50Ω load.
Figure 3 shows the single-ended gain and phase response for both inputs on a linear frequency scale
through 200 MHz. Note the 180° phase offset for the inverting-signal gain, indicating signal inversion. The
slightly quicker roll-off of the inverting-gain response is consistent from part to part. This broadband
performance is maintained as the part is operated at higher gain settings. It is the close, broadband, gain
match of the inputs that allows the CLC520 to provide this wideband differential amplifier with very good
common-mode signal rejection.
Figure 3. Single-Ended Gain and Phase
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
3
OBSOLETE
Introduction
www.ti.com
One measure of a good differential amplifier is its ability to reject common-mode signals. The common
approach in describing this rejection is as a Common Mode Rejection Ratio (CMRR). The definition of
CMRR is structured to allow the common-mode input signal to be placed in series with one of the
differential inputs, (divided by CMRR), as an equivalent error term. With the following definition of CMRR,
an equivalent input error term is placed at one of the inputs as shown in Figure 4.
Figure 4. Input-Referred Common-Mode Error Model
CMRR = 20log(Ad) − 20log(Ac)
(1)
(2)
where:
Ad = Diff gain
Ac = Common-mode gain
where:
V+, V− = pure differential signals
VC = Common mode signal element
This definition of CMRR essentially input refers an output signal due to a common-mode input signal that
effectively holds the common-mode gain constant as the differential gain is changed. In computing the
actual input-to-output signal gain due to a common-mode input voltage, simply use Ac. Note, with Ac << 1,
the logarithmic form of CMRR yields a large positive value. However, in computing the output commonmode signal, as shown in Figure 4, a linear (V/V) gain must be used and the error must be considered
bipolar.
To measure the CMRR as defined in Figure 4, a measure of the pure differential gain must first be made.
This measurement can be accomplished with the circuit of Figure 5. This circuit uses a transformer with a
center tapped secondary to generate a pure differential input signal. The center tap also provides a DC
path to ground supplying a DC-bias current to each of the inputs. It is necessary, in all cases, to carefully
consider the source of these DC-bias currents. The transformer's frequency response was normalized
prior to the gain and phase response measurement. Although using this transformer effectively AC
couples the differential gain, it is important to recognize that the CLC520 is a truly DC-coupled device. The
measured gain and phase for the circuit of Figure 5 are shown in Figure 6. In order to maintain
compatibility with the common-mode gain measurement, this figure is represented with a logarithmic
frequency sweep from 100 kHz to 100 MHz. This circuit offers an exceptional gain-flatness with only 0.5
dB roll-off to 100 MHz.
4
OA-16 Wideband AGC Amplifier as a Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
Copyright © 1993–2013, Texas Instruments Incorporated
OBSOLETE
Introduction
www.ti.com
Figure 5. Differential Gain Test Circuit
Figure 6. Differential Gain and Phase
The common-mode gain is measured by replacing each of the 50Ω input resistors of Figure 2 with 100Ω
while connecting the two inputs together. Tying the inputs together forces the input signals to be exactly
the same while the resistor replacement retains the 50Ω input impedance match. In an actual application,
connecting the two inputs together is impractical. In most cases the common-mode gain is not set by the
amplifier, but by the mismatch of signal attenuations arising from each signal-source's impedance into the
single-ended input impedance of each of the differential amplifier's inputs. A careful attention to the signalsource impedance match is necessary in order for the CMRR performance to be dominated by the
amplifier and not by the deleterious effects of signal-source impedance mismatches. The common-mode
gain measurement made here sidesteps those issues by simply tying the two inputs together. Figure 7
shows the CMRR using the measured differential gain, the measured common-mode gain and the
logarithmic form of CMRR (Equation 1).
The upper limit of CMRR at low-frequencies (below 100 kHz) is approximately 70 dB. This limit is set by
the differential-to-single-ended conversion that takes place internal to the CLC520. At higher frequencies,
the divergence in single-ended gains results in a 40 dB roll-off of CMRR at 10 MHz (shown in Figure 3).
The CLC520's two high-impedance inputs with its internal wideband differential-to-single ended conversion
combine to form a very wideband, high CMRR, differential amplifier.
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
5
OBSOLETE
Application Hints
www.ti.com
Figure 7. Common-Mode Rejection Ratio of the Circuit in Figure 2
2
Application Hints
2.1
Improving CMRR
Several elements combine to set the frequency response of the CLC520. On the input side, parasitic
capacitance to ground on either of the buffer outputs (pins 4 and 5) can cause high-frequency peaking. It
is essential to keep the PC trace capacitance small and balanced when connecting Rg. For the tests
shown here, Rg was soldered directly across the pins of the DIP while those pins were lifted from the
board. On the output side, Rf will determine the frequency response of the output amplifier. Since this
amplifier uses the current-feedback topology, Rf is the dominant element determining Rs frequency
response. Increasing the value of Rf can be used to roll-off any peaking caused by parasitic capacitance
on the output of the input buffers. However, it is preferable (from a noise standpoint) to minimize this
parasitic on pins 4 and 5 and use lower values of Rf (and therefore lower values of Rg for any particular
gain). The CLC520 is designed for use with a 1 kW feedback resistor. Decreasing this value will cause the
frequency response to peak, while increasing it will roll the response off. Most designs should start by first
selecting a value for Rf and then determine the required Rg using the design equations found in the
CLC520 data sheet. An additional constraint on lower values of Rg for good linear operation is that the
maximum current supplied by the buffers through Rg should be kept within ±1.35 mA. This will set a
maximum differential input voltage based on this current limit and the value of Rg.
Once the parasitic capacitance to ground on pins 4 and 5 has been minimized, a frequency response
similar to that shown in Figure 3 can be achieved for each of the two inputs separately. It is possible to
take advantage of a parasitic gain imbalance in order to bring the inverting gain, at higher frequencies, into
a closer match with the non-inverting gain. A closer gain match over a wider frequency range will improve
the CMRR at high frequencies.
Although the equivalent circuit of Figure 1 shows an output that depends only on the current through Rg,
any additional current driven in to or out of the buffers will also generate an output signal. Therefore, by
adding an AG coupled path to ground on the output of the inverting buffer, its response can be matched to
that of the noninverting buffer. The circuit of Figure 8 shows the original test circuit with the addition of this
frequency-response matching network (RT and CT).
6
OA-16 Wideband AGC Amplifier as a Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
Copyright © 1993–2013, Texas Instruments Incorporated
OBSOLETE
Application Hints
www.ti.com
Figure 8. Differential Amplifier with Inverting Response Compensation
The single-ended frequency responses shown in Figure 3 show a lower bandwidth for the inverting gain
path vs. the non-inverting. This bandwidth mismatch is consistent from part to part and is set by the
internal gain path. The buffer bandwidths are considerably higher and do not play a role determining this
response. The following analysis will show how to select the appropriate values for RT and CT such that
the frequency response of the inverting gain path can be matched to that of the non-inverting gain path.
ω+ = Non-inverting response pole
ω− = Inverting response pole
Non-inverting frequency response:
(3)
Inverting frequency:
(4)
Assuming:
ω− < ω+
Compensate A− to achieve the following:
(5)
The single-ended gain response of either input may be analyzed by grounding one input in order to
determine the current generated at the output of the active buffer channel. Adding the RT-CT series
combination will then provide a means of canceling the internal inverting-path pole with a zero, and
replacing it with a pole that matches that seen by the single-ended non-inverting gain path. Note, adding
this network will not impact the non-inverting response as long as it is assumed the buffers have zero
output-impedance. The following analysis provides a method for computing the required values of RT and
CT given Rg and the initial single-ended frequency response of each input as shown in Figure 3. Note: the
input-to-output gain from the current produced in the compensation path is 1/2 that of the gain of the
current produced through Rg.
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
7
OBSOLETE
Application Hints
www.ti.com
The output voltage due to an inverting input voltage is:
(6)
Solving this for the gain to the output:
(7)
The non-inverting path has a gain of:
(8)
−
Equating these two gains requires a cancelling of the ω pole with the zero developed by the RTCT
network while placing the RTCT pole at ω+.
Solving for RT and CT:
(9)
+
−
Estimating ω and ω from the −1 dB roll-off frequencies of Figure 3 and using:
ω−3 dB = 1.97 ω−1 dB for a 1-pole response roll-off
ω− = 2π (176 MHz)
ω+ = 2π (240 MHz)
RT and CT therefore,
(10)
Figure 9 and Figure 10 show the resulting single-ended frequency responses and the CMRR achieved
through this compensation. Comparing Figure 9 to Figure 3 shows a much closer match over frequency. A
significant improvement in the high-frequency CMRR has been achieved with this simple approach. CT
should be tuned for best CMRR at these higher frequencies. Note that when using different values of Rf
and Rg, a remeasurement of the single-ended gains is required in order to provide the single ended gain
poles necessary for this compensation analysis.
8
OA-16 Wideband AGC Amplifier as a Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
Copyright © 1993–2013, Texas Instruments Incorporated
OBSOLETE
Application Hints
www.ti.com
Figure 9. Single-Ended Gains with Inverting Path
Compensation
2.2
Figure 10. Improved CMRR with Better ResponseMatch Over Frequency
Setting the Differential Gain
To use the CLC520 at a fixed gain, it is best (from a temperature stability standpoint) to operate at its
maximum gain, determined by Rf and Rg. The adjustable portion of the CLC520's gain is set by a twotransistor internal differential stage which compares the voltage seen on pin 2 to an internal reference
voltage developed as a resistor divider from the positive supply to ground. With approximately 750W
internally to ground on pin 2, the 400Ω external resistor shown on the circuits above will develop
approximately 3.3V at pin 2, insuring the internal gain stage is fully switched to maximum gain.
Note that the signal gain is also dependent on an internal current-mirror gain from the current developed in
Rg to the multiplier stage. This nominal 1.85 factor will show some part-to-part tolerance and a slight
temperature dependence. A ±3% part-to-part tolerance in this current gain along with a +80 ppm/°C
temperature drift over 0°C–70°C may by used in the design of the CLC520 circuits.
2.3
Using the Gain Adjust Pin
The fixed-gain differential amplifiers shown above can also be disabled with an open-collector pull-down
device on pin 2. Once pin 2 is pulled below 0.4V, the gain will be attenuated by greater than 60 dB. Again,
refer to the CLC520 data sheet for a full discussion of signal attenuation vs. gain-adjust voltage. Although
the forward path can be shut down in this fashion, the output pin remains a low impedance driver: it will
not be tri-stated. However, when driving several of these differential stages into an n:l MUX, shutting down
the CLC520's gain will significantly improve the overall signal isolation at the MUX output.
An adjustable-gain differential amplifier can also be implemented with the CLC520. As discussed in the
data sheet, the CLC520's gain adjustment is intended for operation inside an AGC loop. The gain-adjust
accuracy and temperature stability of the CLC520 does not support open loop operation. A companion
part, the CLC522, should be used if absolute gain accuracy and gain temperature stability is desired in an
open loop (no feedback to the gain adjust pin), adjustable-gain differential-amplifier application.
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
9
OBSOLETE
Application Suggestion
2.4
www.ti.com
Input Noise
The equivalent input noise of the CLC520 is set largely by the value of Rg. As shown in the data sheet, a
model for the input noise voltage due to Rg is simply Rg × l 8pA/√Hz. For any given gain setting, scaling
down the values of Rf and Rg will reduce this input noise. Since Rf controls the output-amplifier stability, it
cannot be made too small. For a fixed Rf, decreasing Rg will increase the signal gain. Since the input noise
decreases at the same rate as the gain increases, the output noise remains nearly constant as Rg is
decreased.
Figure 11 shows the measured input-referred spot noise voltage for the differential amplifier circuit of
Figure 5.
Figure 11. Input Noise Voltage
3
Application Suggestion
3.1
Wideband Differential Coax Line Receiver
It is often necessary to transfer high-speed signals from point to point via a matched-impedance coaxial
line. Figure 12 illustrates one receiver implementation using the CLC520 at a fixed gain. Since both
buffers have high impedance inputs, a simple termination across the center conductor and properly
terminate the cable allowing the differential signal to be picked-off and amplified by the CLC520. This
circuit ties the coax shield into the local ground through a high-frequency blocking ferrite bead. This will
help prevent coupling of high-frequency common-mode noise from the coaxial line onto the local ground,
while at the same time setting the DC voltage and current operating point for the CLC520 inputs. This will
also act to break high-frequency ground loops between different pieces of equipment.
Figure 12. Differential Coax Line Receiver
10
OA-16 Wideband AGC Amplifier as a Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
Copyright © 1993–2013, Texas Instruments Incorporated
OBSOLETE
Application Suggestion
www.ti.com
3.2
VIdeo Loop-Through Amplifier
The loop-through connection is one alternative to the impedance-matching approach of high-speed
signals. For this approach, a high-input-impedance differential amplifier is simply placed across the center
conductor and shield with minimal loading and no characteristic impedance-matching. Good highfrequency common mode rejection and good wideband differential amplification are essential for this
application. The final destination of this daisy-chained connection terminates the cable in it's characteristic
impedance.
An implementation of this loop-through connection using the CLC520 is shown in Figure 13. This circuit is
a replication the circuit of Figure 2 with some additional input resistors and a shutdown control gate.
The 20Ω resistors to ground will insure a DC-bias path for the input-stage bias currents. If it is absolutely
certain that a DC path through both the center conductor and the shield will be maintained, the 20kΩ
resistors can be eliminated with an overall improvement of VSWR. With only the 20kΩ termination, the
CLC520's input offset current drift will generate a nominal input offset-voltage drift of 100mV/°C. It is
desirable considering common mode rejection and offset-current drift, to keep these input termination
resistors as large as possible. Ideally, the termination resistors should be eliminated if the bias current can
be supplied by the cable. Remember, any mis-match in the single-ended attenuations from the center
conductor's and shields source impedances into the CLC520's input impedances will degrade the CMRR.
The two series 37.5Ω resistors into pins 3 and 6 act to isolate the inputs from the cable reactance helping
to maintain high-frequency input stability. These resistors, included with the parasitic input-capacitance to
ground, will also form a matched-impedance termination for the cable at very high frequencies (>500
MHz): well beyond the signal frequencies of interest. The full signal level would be available to
downstream stages using this wideband differential amplifier as a loop-through connection.
Figure 13. Video Loop-Through Connection Using A Wideband Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
11
OBSOLETE
Application Suggestion
3.3
www.ti.com
Very Wideband Pulse-Differencing Amplifier
With the addition of several frequency-response trims, the basic circuit Figure 2 can be used to implement
a very wideband pulse-differencing amplifier. Targeting a gain of +1V/V into a matched 50Ω load,
bandwidths in excess of 300 MHz are achievable. Figure 14 shows a typical single-sided pulse response.
The input rise time for this test is approximately 800 ps. With a 1.15 ns output rise time and a 08 ns input
rise time, the amplifiers actual rise time is approximately 0.8 ns for this 0.9V step at the load. Very similar
and well-matched results can be achieved for both the inverting and non-inverting inputs.
Figure 14. Very Wideband Single Input Pulse Response
3.4
Alternative Wideband Differential Amplifiers
Although the classical single op amp differential amplifier has found wide usage, several intrinsic problems
limit it's performance. Both signal inputs are looking into relatively low and not necessarily well-matched
impedances causing unbalanced signal-source attenuation, having the effect of degraded CMRR. Most
simplified analysis assume a 0W source impedance in order to circumvent this problem. Furthermore,
resistor inaccuracies, instead of the amplifier itself, will typically dominate the CMRR. These resistors and
the amplifier's open-loop gain will determine the differential-to-single ended conversion carried out so well
by the CLC520.
However, a classical single-amp differential amplifier combined with a pair of wideband, low-outputimpedance buffers can be made to approach the performance of the CLC520. This approach may be
preferred if lower input noise, lower power dissipation and improved DC-drift characteristics are worth a
higher number of parts, lower differential bandwidth and the necessary precise resistor matching.
Figure 15 provides an example of a single-amp differential amplifier using two buffers from a CLC114
quad buffer and a low-gain op amp with a differential gain of +1V/V.
Figure 15. Single Amplifier Differential Amplifier With Input Buffering
12
OA-16 Wideband AGC Amplifier as a Differential Amplifier
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
Copyright © 1993–2013, Texas Instruments Incorporated
OBSOLETE
Conclusion
www.ti.com
The two input-barriers provide many of the same advantages found with the CLC520 inputs. Any of the
input terminations described for the CLC520 may be used here as well. The optional 500Ω resistor to
ground on the output of the non-inverting buffer provides a means of matching the loads seen by both
buffer outputs. This load matching will improve the high frequency response-match. The four 250Ω
resistors should be matched as closely as possible since any mis-match will degrade the CMRR. The
recommended low-gain differencing amplifier may be chosen from the following selection of Texas
Instruments wideband low-gain amplifiers:
CLC402— Low-gain high-accuracy current-feed back amplifier
Lower CMRR than the CLC420 with wider bandwidth and better fine-scale, pulse-settling accuracy.
CLC409— Very wideband, low-gain, current-feed back amplifier.
CLC410— Intermediate performance, low-gain, current-feedback amplifier. This part also includes a
shutdown feature and provides the best dG/d_ for composite video applications.
CLC420— Unity-gain stable voltage feedback amplifier. This part will provide the best CMRR and DC
accuracy
CLC502— Similar to the CLC402 but with an output-clipping feature
All of these parts are optimized for the 250Ω feedback resistor shown in the circuit of Figure 15.
4
Conclusion
As operating speeds have increased, the need for a wide-bandwidth high-CMRR differential amplifiers has
increased. Texas Instruments CLC520 and CLC522 provide all of the required building blocks integrated
into one part. Although intended for adjustable gain requirements, operating the CLC520 at a fixed gain is
perfectly acceptable and preferable in a differential receiver application. Signal bandwidths in excess of
150 MHz over a wide range of gains, along with CMRR exceeding 60 dB through 10 MHz, and two
matched high-impedance inputs provide all the essential requirements for wideband differential
amplification. In some applications using wideband, low power buffers and a standard single op amp
differential amplifier topology offers certain advantages over the CLC520 approach.
The circuits included in this application rerport have been tested with Texas Instruments parts that may
have been obsoleted and/or replaced with newer products. Please refer to the CLC to LMH conversion
table to find the appropriate replacement part for the obsolete device.
SNOA393B – January 1993 – Revised April 2013
Submit Documentation Feedback
OA-16 Wideband AGC Amplifier as a Differential Amplifier
Copyright © 1993–2013, Texas Instruments Incorporated
13
IMPORTANT NOTICE
Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other
changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest
issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and
complete. All semiconductor products (also referred to herein as “components”) are sold subject to TI’s terms and conditions of sale
supplied at the time of order acknowledgment.
TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms
and conditions of sale of semiconductor products. Testing and other quality control techniques are used to the extent TI deems necessary
to support this warranty. Except where mandated by applicable law, testing of all parameters of each component is not necessarily
performed.
TI assumes no liability for applications assistance or the design of Buyers’ products. Buyers are responsible for their products and
applications using TI components. To minimize the risks associated with Buyers’ products and applications, Buyers should provide
adequate design and operating safeguards.
TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or
other intellectual property right relating to any combination, machine, or process in which TI components or services are used. Information
published by TI regarding third-party products or services does not constitute a license to use such products or services or a warranty or
endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the
third party, or a license from TI under the patents or other intellectual property of TI.
Reproduction of significant portions of TI information in TI data books or data sheets is permissible only if reproduction is without alteration
and is accompanied by all associated warranties, conditions, limitations, and notices. TI is not responsible or liable for such altered
documentation. Information of third parties may be subject to additional restrictions.
Resale of TI components or services with statements different from or beyond the parameters stated by TI for that component or service
voids all express and any implied warranties for the associated TI component or service and is an unfair and deceptive business practice.
TI is not responsible or liable for any such statements.
Buyer acknowledges and agrees that it is solely responsible for compliance with all legal, regulatory and safety-related requirements
concerning its products, and any use of TI components in its applications, notwithstanding any applications-related information or support
that may be provided by TI. Buyer represents and agrees that it has all the necessary expertise to create and implement safeguards which
anticipate dangerous consequences of failures, monitor failures and their consequences, lessen the likelihood of failures that might cause
harm and take appropriate remedial actions. Buyer will fully indemnify TI and its representatives against any damages arising out of the use
of any TI components in safety-critical applications.
In some cases, TI components may be promoted specifically to facilitate safety-related applications. With such components, TI’s goal is to
help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and
requirements. Nonetheless, such components are subject to these terms.
No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties
have executed a special agreement specifically governing such use.
Only those TI components which TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in
military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components
which have not been so designated is solely at the Buyer's risk, and that Buyer is solely responsible for compliance with all legal and
regulatory requirements in connection with such use.
TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of
non-designated products, TI will not be responsible for any failure to meet ISO/TS16949.
Products
Applications
Audio
www.ti.com/audio
Automotive and Transportation
www.ti.com/automotive
Amplifiers
amplifier.ti.com
Communications and Telecom
www.ti.com/communications
Data Converters
dataconverter.ti.com
Computers and Peripherals
www.ti.com/computers
DLP® Products
www.dlp.com
Consumer Electronics
www.ti.com/consumer-apps
DSP
dsp.ti.com
Energy and Lighting
www.ti.com/energy
Clocks and Timers
www.ti.com/clocks
Industrial
www.ti.com/industrial
Interface
interface.ti.com
Medical
www.ti.com/medical
Logic
logic.ti.com
Security
www.ti.com/security
Power Mgmt
power.ti.com
Space, Avionics and Defense
www.ti.com/space-avionics-defense
Microcontrollers
microcontroller.ti.com
Video and Imaging
www.ti.com/video
RFID
www.ti-rfid.com
OMAP Applications Processors
www.ti.com/omap
TI E2E Community
e2e.ti.com
Wireless Connectivity
www.ti.com/wirelessconnectivity
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2013, Texas Instruments Incorporated
Was this manual useful for you? yes no
Thank you for your participation!

* Your assessment is very important for improving the work of artificial intelligence, which forms the content of this project

Related manuals

Download PDF

advertising