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Texas Instruments Large-Signal Specifications for High-Voltage Line Drivers Application notes
Application Report
SBOA126 – June 2011
Large-Signal Specifications for High-Voltage Line Drivers
Kristoffer Flores, Xavier Ramus
............................................................................. High-Speed Products
ABSTRACT
Output voltage swing, output current, and slew rate interact with many other specifications in operational
amplifiers. This application note develops the definitions for each parameter and the relationship between
important ac parameters such as large-signal bandwidth and distortion.
1
2
3
4
5
6
7
Contents
Introduction ..................................................................................................................
Slew Rate ....................................................................................................................
Slew Rate and Large-Signal Bandwidth .................................................................................
A Warning about Spice Simulations ......................................................................................
Output Voltage Swing and Output Current ..............................................................................
Conclusion ...................................................................................................................
References ...................................................................................................................
1
3
5
8
8
9
9
List of Figures
1
THS3091 10-MHz Pulse Response, Gain of 5-V/V Configuration, 100-Ω Load .................................... 4
2
THS3091 Pulse Response Rising Edge, Gain of 5-V/V Configuration, 100-Ω Load ............................... 4
3
Slew Rate Representation for a Sinewave .............................................................................. 5
4
Theoretical Slew-Limited Output for Three Different Slew Rates ..................................................... 6
5
THS4631 Large-Signal Frequency Response, Gain = 2 V/V
7
6
THS3091 Large-Signal Frequency Response, Gain = 5 V/V
7
7
8
1
.........................................................
.........................................................
THS3091 Output Voltage Swing versus Load Resistance ............................................................
OPA2673 Output Voltage versus Output Current ......................................................................
8
9
Introduction
Bandwidth and distortion performance of an operational amplifier do not remain constant over the entire
output voltage swing range of the amp. Some of the device parameters that affect both bandwidth and
distortion are what is normally considered dc parameters (such as output current and output voltage), or
parameters defined when the amplifier is saturating (such as slew rate). This application note defines each
of these parameters and provides guidelines for designers to follow during the selection process in order
to achieve maximum performance.
In the high-speed amplifier market, there are two primary competing architectures. Each approach offers
its own set of advantages. It is appropriate to begin, then, by understanding how to identify a good
standalone op amp for a high-speed, high voltage swing application.
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Introduction
1.1
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Voltage-Feedback or Current-Feedback Op Amps
In applications where a high voltage swing output is required, the output signal conditioning stage (or
stages) typically requires gain. The signal may originate from a 3.3-V or 5-V digital-to-analog converter
(DAC) with a 1-VPP output voltage range, while the required signal is a single-ended 10 VPP or higher.
Depending on the bandwidth and slew rate requirements, the gain and output swing requirements
generally determine the use of either a current feedback (CFB) amplifier or a voltage feedback (VFB)
amplifier.
Voltage Feedback
VFB op amps are typically constrained by the gain-bandwidth product (GBWP) parameter. GBWP theory
dictates that if an op amp has a bandwidth of 10 MHz at a gain of 10 V/V, then it has 5-MHz bandwidth at
a gain of 20 V/V, 20-MHz bandwidth at a gain of 5 V/V, and so on. The product of the gain times the
bandwidth is constant. High-speed VFB amplifiers do not strictly follow this rule at lower gains because of
less compensated designs and package parasitics that affect unity-gain bandwidth (see Ref. 1). However,
the general inverse relationship between gain and bandwidth is evident in most VFB amplifiers. As an
example, consider the OPA820 and the THS4631, two high-speed VFB amplifiers.
The OPA820 is a high-speed, 12-V capable, bipolar voltage feedback op amp. Table 1 shows the
small-signal bandwidth specifications from the OPA820 product data sheet.
Table 1. OPA820 Small-Signal Bandwidth and Gain-Bandwidth Product
OPA820ID, IDBV
TYP
MIN/MAX OVER
TEMPERATURE
CONDITIONS
+25°C
+25°C
0°C to
70°C
–40°C
to
+85°C
G = +1, VO = 0.1VPP, RF = 0Ω
800
G = +2, VO = 0.1VPP
240
170
160
155
MHz
PARAMETER
UNIT
AC PERFORMANCE
Small-Signal Bandwidth
MHz
G = +10, VO = 0.1VPP
30
23
21
20
MHz
G ≥ 20
280
220
204
200
MHz
Gain-Bandwidth Product
The GBWP is specified as 280 MHz and all small-signal bandwidths adhere closely to the expected
results. At a gain of 10 V/V, the small-signal bandwidth is 30 MHz, which is close to the 280-MHz GBWP.
At 2-V/V gain, the small-signal bandwidth is 240 MHz, which already exceeds the GBWP. Notice that at
low gains, the small-signal bandwidth times the gain is actually much larger than the GBWP.
Table 2 presents the same performance data for the THS4631, a high-voltage, high slew rate, wideband
FET op amp. These data (from the THS4631 product data sheet) indicate that the GBWP relationship
holds more steady at lower gains than it does with the OPA820. In the same manner for the lowest gain,
however, the relationship between gain and bandwidth breaks down.
Table 2. THS4631 Small-Signal Bandwidth and Gain-Bandwidth Product
THS4631
TYP
MIN/MAX OVER
TEMPERATURE
CONDITIONS
+25°C
0°C to
70°C
G = 1, RF = 0 Ω, VO = 200 mVPP
325
MHz
G = 2, RF = 499 Ω, VO = 200 mVPP
105
MHz
G = 5, RF = 499 Ω, VO = 200 mVPP
55
MHz
G = 10, RF = 499 Ω, VO = 200 mVPP
25
MHz
G > 20
210
MHz
PARAMETER
+25°C
–40°C
to
+85°C
UNIT
AC PERFORMANCE
Small-Signal Bandwidth, –3 dB
Gain-Bandwidth Product
2
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Slew Rate
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Current Feedback
In contrast with VFB op amps, CFB op amps do not have a gain-bandwidth product. While the feedback
factor that determines compensation in VFB op amps involves both gain-setting resistors (that is, the
noninverting gain or the noise gain), the feedback factor in CFB op amps only involves the feedback
resistor. This architecture ultimately results in a relative independence between the gain and bandwidth in
CFB op amps. An in-depth discussion of the VFB versus CFB topology can be found in the application
note, Voltage Feedback vs Current Feedback Op Amps (see Ref. 2).
Looking at the relationship between gain and bandwidth for current-feedback amplifiers, the THS3091
high-speed, 30-V current-feedback op amp shows an example of the relative independence between gain
and bandwidth. This effect is illustrated in Table 3 (from the THS3091 product data sheet).
Table 3. THS3091 Small-Signal Bandwidths for Different Gains
THS3091
TYP
MIN/MAX OVER
TEMPERATURE
CONDITIONS
+25°C
0°C to
70°C
G = 1, RF = 1.78 kΩ, VO = 200 mVPP
235
MHz
G = 2, RF = 1.21 kΩ, VO = 200 mVPP
210
MHz
G = 5, RF = 1 kΩ, VO = 200 mVPP
190
MHz
G = 10, RF = 866 Ω, VO = 200 mVPP
180
MHz
PARAMETER
+25°C
–40°C
to
+85°C
UNIT
AC PERFORMANCE
Small-Signal Bandwidth
While the THS3091 at unity-gain has almost 100 MHz lower bandwidth than the THS4631, the THS3091
is able to maintain a much higher bandwidth than the THS4631 as the gain increases. By the time the
gain of both amplifiers is set at a 10-V/V configuration, the THS3091 has more than seven times the
bandwidth of the THS4631. The bandwidth advantage at higher gain configurations of the THS3091 CFB
amplifier compared to the THS4631 VFB amplifier also suggests that the THS3091 has more loop-gain
margin available at higher frequencies (translating to better distortion performance at higher frequencies)
than the THS4631. This characteristic is a direct consequence of the device architecture: the CFB
compensation is the feedback resistance and the inverting input resistance times the noise gain
(RF + rinv. NG), instead of the noise gain (NG) alone for VFB op amps. As the gain increases, the feedback
resistance decreases to maintain optimum compensation; in other words, the term (RF + rinv. NG) remains
constant.
Why review the small-signal bandwidth advantage of CFB amplifiers when the signals of concern have
large swings? As this report discusses later, slew rate is the major factor that determines large-signal
bandwidth. Small-signal bandwidth typically sets the upper limit on large-signal bandwidth, while slew rate
determines how much bandwidth can actually be realized with large signals. Furthermore, the CFB
architecture that allows for higher bandwidth at higher gains also allows the CFB amplifier to achieve
much higher slew rates than do traditional, differential-pair input VFB amplifiers. This CFB architectural
advantage has been implemented in slew-boosted VFB architecture, thereby minimizing the slew rate
differences. These slew-boosted VFB op amps, however, are still limited by the GBWP characteristic.
2
Slew Rate
Slew rate is the measure of the maximum rate of change that the operational amplifier output can achieve.
This specification can be expressed as:
dV
SR =
dt
Slew rate is a large-signal parameter and should not be confused with rise and fall times, two small-signal
parameters.
Slew rate is specified in units of volts per unit of time; typically, volts per microsecond (V/μs). For example,
the THS3091 is rated for a typical slew rate of 7300 V/μs at a gain configuration of 5 V/V for a 20-V output
step, meaning that the output can swing 20 V in about 2.7 ns (20 V / 7300 V/μs = 2.7 ns). The time it
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Slew Rate
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actually takes for the output to swing 20 V is greater than 2.7 ns because of the fact that the slew rate is
measured from 25% to 75% (or 10% to 90%, depending on the definition of the specification that is used)
of the step voltage, and there is no slewing in the first and last 25% (or 10%) of the voltage step. Figure 1
shows the pulse response of the THS3091 configured at a gain of 5 V/V and driven with a 4-VPP square
wave input. Figure 2 shows the same pulse response but is enlarged to illustrate the slew rate at the rising
edge.
THS3091
LARGE-SIGNAL PULSE RESPONSE
15
Output Voltage (V)
10
5
0
-5
-10
-15
0
10
20
30
40
50
60
70
80
90
100
Time (ns)
Figure 1. THS3091 10-MHz Pulse Response, Gain of 5-V/V Configuration, 100-Ω Load
THS3091
PULSE RESPONSE RISING EDGE
15
Output Voltage (V)
10
Dt
5
DV
25% to 75%
0
-5
-10
Slew Rate =
V
DV
= 7300
ms
Dt
-15
15
17
19
21
23
25
27
29
31
33
35
Time (ns)
Figure 2. THS3091 Pulse Response Rising Edge, Gain of 5-V/V Configuration, 100-Ω Load
In both VFB and CFB amplifiers, the slew rate is determined by the available current to charge and
discharge the internal dominant pole compensation capacitor of the op amp. In a standard VFB amplifier,
the current consists of the tail current to the input differential pair. Given a particular device, this tail
current is fixed, thus establishing a ceiling to the possible slew rate.
On the other hand, in a CFB amplifier, there is no input differential pair; instead, the noninverting input pin
voltage is buffered to the inverting pin, and the resulting error current out of (or into) the inverting input is
current-mirrored to charge or discharge the dominant pole compensation capacitor. This error current
consists of the current from the op amp output through the feedback resistor and current through the gain
setting resistor, which is much greater than the available tail current in a standard VFB amplifier. In this
way, a CFB amplifier is capable of having much higher slew rates than a standard VFB amplifier (see Ref.
3).
4
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Slew Rate and Large-Signal Bandwidth
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3
Slew Rate and Large-Signal Bandwidth
For an amplifier with a sine wave input (and ideally, a scaled replica sinewave output centered around
0 V), the maximum slope or rate of change occurs at the zero crossings. For the sine wave signal shown
in Figure 3:
V(t) = VP·sin(2p·f·t)
The maximum slope magnitudes occur every half-period zero crossing at:
T = 0,
T
T
…n
2
2
where n is any integer and T is the period of the sine wave (or 1/f).
T = 1/f
Voltage (V)
VP
0
dV
dt
?
DV
Dt
-VP
0
T/2
T
3T/2
2T
Time
Figure 3. Slew Rate Representation for a Sinewave
The maximum slope can be interpreted as the required slew rate to accurately generate the sine wave.
This slew rate can be determined by calculating the first derivative of the waveform with respect to time,
and then evaluating the derivative at a zero crossing where the slope is at a maximum.
For the sinusoid V(t), the first time derivative is:
dV(t) = 2pf·V ·cos(2p·f·t)
P
dt
Evaluating the derivative at t = 0 results in:
dV(0) = 2pf·V
P
dt
Replacing dV/dt in the equation with SR (slew rate) gives a useful relationship that helps determine the
required slew rate to pass a sine wave of a desired frequency and amplitude, as shown in Equation 1.
SR = 2pf·VP
(1)
Rearranging Equation 1 yields the inverse equation (Equation 2) that relates an op amp driver with a given
slew rate and desired output sine wave amplitude (VP) to the maximum sine wave frequency, fmax, that the
driver can successfully pass without being slew-rate limited (see Ref. 4).
SR
fMAX =
2pVP
(2)
At frequencies above fmax, the output is limited by the slew rate such that the output can no longer track
the ideal sine wave output. The op amp enters an open loop when the output becomes slew-limited
because the error signal cannot be minimized. For this reason, fmax can be considered the non-slew-limited
bandwidth of the amplifier.
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Slew Rate and Large-Signal Bandwidth
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Figure 4 shows three simulated output waveforms for three drivers with different slew rates versus an
ideal 20-VPP, 100-MHz sine wave. Using Equation 1 with a 100-MHz maximum flatband frequency and
10 V for peak amplitude gives a slew rate requirement of 6283 V/μs. In Figure 4, the three drivers are
slew-rate limited at 5 kV/μs, 4 kV/μs, and 2.5 kV/μs; thus, the respective outputs are distorted into
triangular waves with a slope equal to the slew rate.
20
100-MHz Sine
5000 V/ms
4000 V/ms
2500 V/ms
Voltage (V)
10
0
-10
-20
0
5
10
15
20
Time
Figure 4. Theoretical Slew-Limited Output for Three Different Slew Rates
As the slew rate decreases to the point where the output reduces to a triangle wave, the peak amplitude
of the signal is also reduced, though the fundamental period remains the same. It can be seen from
Figure 4 that the peak of a triangular, slew-limited output is the slew rate multiplied by 1/4 of the
fundamental period.
It is important to note that the non-slew-limited bandwidth, fmax, predicted by Equation 2 is not the –3-dB
bandwidth. Theoretically, an amplifier with a given slew rate (SR) is able to output a sine wave of
amplitude VP up to a frequency, fmax, with no loss in amplitude as a result of slew-limiting. As the signal
frequency rises above fmax, odd-order harmonic distortion becomes drastically degraded as the output
becomes more and more triangular because of slew-limiting. In practice, however, the fmax point is not a
hard limit, and there is visible distortion (that is, distortion visible on an oscilloscope) well below the
non-slew-limited bandwidth.
How well, then, does Equation 2 predict the non-slew-limited bandwidth? In Figure 5 and Figure 6, the
measured, non-slew-limited bandwidth for the THS4631 and THS3091 amplifiers are respectively
compared against the predicted bandwidths. The calculated non-slew-limited bandwidth using Equation 2
does not correspond to a well-defined frequency response threshold in the same manner that a –3-dB
bandwidth does; therefore, Equation 1 can be modified to reflect the loss in signal at the –3-dB point by
dividing the amplitude by √2 (see Ref. 5), as shown in Equation 3.
VP
SR-3dB = 2pf
2
(3)
The slew rate normally covers only 80% of the signal from 10% to 90%. Consequently, instead of being
the derivative of a sine wave, this factor is compensated by multiplying Equation 3 by 0.8, which gives
Equation 4:
VP
SR-3dB, 10% - 90% = 0.8·2pf-3dB
2
(4)
The THS4631 product data sheet reports a 10-V step slew rate of 900 V/μs at a gain of 2-V/V
configuration. Using Equation 4, the estimated –3-dB bandwidth for a 10-VPP output is 50.6 MHz.
V
900
ms
= 50.6 MHz
f-3dB =
VP
0.8·2p
2
6
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Figure 5 shows the measured large-signal frequency response taken with a network analyzer for the
THS4631. As expected, the –3-dB bandwidth is reduced for a 10-VPP output and matches the calculated
value.
THS4631
LARGE-SIGNAL FREQUENCY RESPONSE
9
6
Gain (dB)
3
0
-3
-6
2 VPP
10 VPP
-9
0.1
1
10
100
1000
Frequency (MHz)
Figure 5. THS4631 Large-Signal Frequency Response, Gain = 2 V/V
Applying Equation 4 to the THS3091 produces a result with some contradiction: 7300 V/μs for a gain of
5 V/V with a 20-V step indicates that the amplifier can support a 200-MHz, 20-VPP large-signal bandwidth,
and 5000 V/μs for a gain of 2 V/V with a 10-V step indicates that a 280-MHz large-signal bandwidth is
possible. Both of the large-signal bandwidth measurements in Figure 6 show a –3-dB bandwidth in the
range of 120 MHz to 150 MHz.
THS3091
LARGE-SIGNAL FREQUENCY RESPONSE
17
Gain (dB)
14
11
8
2 VPP
10 VPP
20 VPP
5
1
10
100
1000
Frequency (MHz)
Figure 6. THS3091 Large-Signal Frequency Response, Gain = 5 V/V
This result indicates that the slew rate measurement made is incorrect because the amplifier may not
have been slew-limited, or that the 25% to 75% measurement cannot be used when comparing the slew
rate to large-signal bandwidth.
Therefore, while Equation 4 can provide a good indicator of the actual slew rate capability of an amplifier,
it does not always align with the actual large-signal behavior of a given op amp. As mentioned before, the
distortion performance begins to degrade well before the non-slew-limited bandwidth is reached, and
choosing an op amp with an excess slew rate (that is, a rate greater than actually required) is
recommended. To achieve an 80-dBc distortion level, it is recommended to have the amplifier slew 20
times faster than the maximum signal.
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A Warning about Spice Simulations
4
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A Warning about Spice Simulations
Spice simulation tools are useful to verify the first-order expectation of the behavior and performance of a
circuit. To make maximum use of the model, though, it is important to understand the limitations of
simulations and device models.
A Spice ac simulation reduces the simulation circuit to an idealized linear model and calculates the circuit
frequency response based on infinitely small signals around the operating point. Therefore, an ac sweep
is not able to demonstrate any large-signal frequency response of an op amp circuit. A time-domain or
transient analysis simulation gives a better indication of large-signal behavior, provided that the op amp
slew rate has been modeled in the macromodel. The Spice macro header information for most newer TI
high-speed op amp device models specify the device parameters that have been modeled.
5
Output Voltage Swing and Output Current
The last important considerations when selecting an op amp for driving large output signals are output
voltage swing and output current drive. The first consideration here is that the output transistors in an op
amp output stage require headroom from the supply voltage, thus limiting the output swing. A
high-voltage, high output current op amp typically uses a push-pull emitter output stage that requires 1 V
to 2 V of headroom from the positive supply and from the negative supply with no output load. This
headroom requirement depends on the operating supply voltage of the amplifier as well as the output
drive capability. A ±15 V amplifier normally has a higher headroom requirement than a ±5-V supply
device. For example, the THS3091 output voltage swing versus load resistance graph shown in Figure 7
illustrates that with a 1-kΩ resistive load, the output voltage swing with ±15-V supplies is approximately
±13.2 V.
THS3091
OUTPUT VOLTAGE vs
LOAD RESISTANCE
16
Output Voltage, VO (V)
12
8
4
VS = ±15 V
TA = -40°C to +85°C
0
-4
-8
-12
-16
10
100
1000
Load Resistance, RL (W)
Figure 7. THS3091 Output Voltage Swing versus Load Resistance
The output voltage swing is also affected by the output current ability of the amplifier. Figure 7 shows the
significance of an amplifier output current sinking and sourcing capability. As the load resistance
decreases below 100 Ω, the output voltage swing range of the THS3091 becomes limited as a result of
the finite current output to drive the load (V = IR).
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The latest line driver amplifier data sheets also include an output voltage versus output current graph that
can be used to quickly evaluate the op amp voltage and current limitations. Figure 8 shows such a graph
for the OPA2673 that also provides a useful 2-W internal power dissipation boundary.
6
Output Voltage (V)
4
2W Internal
Power Dissipation
Single Channel
50W
Load Line
2
0
100W Load Line
10W Load Line
-2
2W Internal
Power Dissipation
Single Channel
-4
25W Load Line
-6
-800
-600
-400
-200
0
200
400
600
800
Output Current (mA)
Figure 8. OPA2673 Output Voltage versus Output Current
6
Conclusion
Slew rate, output voltage swing, and output current drive are strongly related to bandwidth and distortion.
The parameters and architectural concepts developed in this application note can be used during the
selection process to refine component selection.
7
References
Unless otherwise indicated, the following documents are available for download at www.ti.com.
1. Ramus, X. (2009). Making the most of a low-power, high-speed operational amplifier. Texas
Instruments application report SBOA121.
2. Karki, J. (1998). Voltage feedback vs current feedback op amps. Texas Instruments application report
SLVA051.
3. Ramus, X. (2009). Voltage feedback vs. current feedback amplifiers: Advantages and limitations.
Presentation at IEEE Long Island Section.
4. Predicting op amp slew rate limited response. LB-19, 1972 National Semiconductor. Available at
http://www.national.com/ms/LB/LB-19.pdf.
5. Steffes, M. (2008). Avoiding performance pitfalls for the pulse response of high-speed op amps. Article
on EN-Genius Network.
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www.ti-rfid.com
Wireless
www.ti.com/wireless-apps
RF/IF and ZigBee® Solutions
www.ti.com/lprf
TI E2E Community Home Page
e2e.ti.com
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