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Texas Instruments Making the Most of a Low-Power, High-Speed Operational Amplifier Application notes
Application Report
SBOA121 – November 2009
Making the Most of a Low-Power, High-Speed Operational
Amplifier
Xavier Ramus
.................................................................................................... High-Speed Products
ABSTRACT
High-speed, high-performance operational amplifiers tend to be associated with high power dissipation.
This application note compares the relative performance of several low-power, high-speed operational
amplifiers and describes trade-offs to balance performance with low quiescent power dissipation.
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Contents
Introduction .................................................................................................................. 2
Bandwidth Performance Comparison .................................................................................... 2
Harmonic Distortion Comparison ......................................................................................... 4
Low-Power Filtering ......................................................................................................... 8
Multiplying DAC Transimpedance Amplifier ............................................................................ 10
Conclusion .................................................................................................................. 12
List of Figures
...................................................................
1
OPA890 Open-Loop Gain and Phase Performance
2
Video Summing Amplifier Circuit .......................................................................................... 3
3
VFA Loop Gain (Typical Performance)
4
5
6
7
8
9
10
11
12
13
14
15
16
..................................................................................
CFA Loop Gain (Typical Performance) ..................................................................................
OPA2889 Low-Frequency Harmonic Distortion (Gain of –1V/V) .....................................................
OPA683 Low-Frequency Harmonic Distortion (Gain of –1V/V) .......................................................
OPA683 Open-Loop Transimpedance Gain Performance ............................................................
OPA2889 Open-Loop Gain Performance ................................................................................
OPA847 Low-Frequency Harmonic Distortion (Gain of –10V/V) .....................................................
OPA683 Low-Frequency Harmonic Distortion (Gain of –10V/V) .....................................................
MFB Filter Topology Using the OPA2889 ...............................................................................
MFB Filter Frequency Response .........................................................................................
Single-Supply Sallen-Key Filter Implementation ........................................................................
Butterworth Low-Pass Filter Frequency Response ...................................................................
Multiplying DAC Transimpedance Driver ...............................................................................
Multiplying DAC Frequency Response Using the OPA890 as Output Stage......................................
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1
Introduction
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Introduction
Achieving undeniably portable instrumentation requires the use of a low power dissipation operational
amplifier. It is extremely difficult to balance various low-power application circuits while maintaining high
system performance. This difficulty is partly the result of the ac performance degradation, such as slew
rate and bandwidth, as the amplifier quiescent current is reduced. In this application report, we first start
with bandwidth performance comparison, then move to harmonic distortion, and see what actual losses
we must address with low quiescent current devices. This document concludes with the analysis of
several application circuits as well as reviewing the advantages of each amplifier topology for each
application.
2
Bandwidth Performance Comparison
A useful practice for voltage-feedback amplifiers (VFAs) is to do a side-by-side comparison of the gain
bandwidth product (GBP) and the quiescent current for a given op amp.
Note that for different op amp architectures such as the current-feedback amplifier (CFA), this approach
does not work because a CFA does not have a GBP. Instead, the bandwidth of the highest gain shown in
the product data sheet is used to mirror the GBP definition of the VFA.
The method used to derive the GBP from the data sheet is provided below for a high-speed VFA. It is
extremely rare to have the unity-gain bandwidth be equal to the gain bandwidth product. This rarity is as
much the result of the practice of not having overcompensated amplifiers as it is the effects of package
parasitics that extend the unity-gain bandwidth. For high-speed amplifiers, you must measure the GBP at
gain of 40dB or greater in the open-loop gain graph. The example performance graph shown in Figure 1 is
taken from the OPA890 data sheet.
80
180
160
Open-Loop Gain
60
140
50
120
Open-Loop Phase
40
100
30
80
20
60
10
40
0
20
-10
Open-Loop Phase (° )
Open-Loop Gain (dB)
70
0
100
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Figure 1. OPA890 Open-Loop Gain and Phase Performance
In Figure 1, at the intersection of the red line and the blue line, you measure the GBP; in the case of the
OPA890, this measurement results in a bandwidth of 1.3MHz for a 100V/V gain (40dB), or 130MHz GBP.
This number is consistent with the number reported in the ±5V electrical specifications table of the device
data sheet.
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Bandwidth Performance Comparison
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Table 1 summarizes the bandwidth and quiescent current information for various low-power, high-speed
VFAs and CFAs.
Table 1. VFA and CFA Comparison
(1)
VFA Device
Gain Bandwidth
Product (MHz)
Quiescent Current
(mA)
OPA890
130
1.1
OPA2889
75
0.46
THS4281
38
0.8
CFA Device
Bandwidth (MHz)
Gain (1) (V/V)
Quiescent Current
(mA)
OPA684
71
100
1.7
OPA683
35
100
0.94
Highest gain recommended for this device.
By comparing the OPA890 with the OPA683, we can see the clear advantage of using a current-feedback
amplifier for high-gain applications. If we tried to use the OPA890 in a 100V/V application, we would
achieve a 1.3MHz, –3dB bandwidth. This performance pales in comparison with the 35MHz bandwidth of
the OPA683, or for slightly more quiescent power dissipation, the 71MHz bandwidth of the OPA684.
Note that all of these devices are capable of operating on a +3V supply; at this low supply voltage, though,
only the THS4281 has the full output voltage swing because it is a rail-to-rail input/output (RRIO) op amp.
The application described in Figure 2 takes advantage of the low-power CFA OPA684 for a video
summing amplifier. In this configuration, the OPA684 provides greater than 120MHz bandwidth. If a VFA
(such as the OPA890) were used instead, the maximum bandwidth would be limited to 15.5MHz as a
result of the gain bandwidth product dependency for the VFA architecture. To achieve the same
bandwidth as the OPA684, a 1GHz GBP VFA would be required. Although such devices are readily
available, the quiescent current would have to be increased tenfold.
+5V
500W
DIS
IN1
Supply decoupling not shown.
75W
88.9W
OPA684
75W Load
500W
-2(IN1 + IN2 + IN3 + IN4)
IN2
1kW
88.9W
500W
IN3
-5V
88.9W
500W
IN4
88.9W
Figure 2. Video Summing Amplifier Circuit
Figure 2 shows a typical inverting summing application where four sources are summed through 500Ω
gain resistors while also including an 88.9Ω terminating impedance, to present a 75Ω input impedance to
each source. The gain for each channel is –2V/V to the output pin and –1V/V to the matched load. For the
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Harmonic Distortion Comparison
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OPA684, the extremely low inverting input impedance ensures non-interactive summing for all of the
channels. The amplifier bandwidth is largely independent of variations in the gain setting elements, and
instead depends primarily on the feedback resistor value. This type of circuit may be used to sum
numerous signals together or, where the earlier stages can be disabled, to allow multiple channels to be
brought together with only the active channel passing on to the output.
3
Harmonic Distortion Comparison
Harmonic distortion for both VFAs and CFAs depends on the loop gain. For a VFA, the loop gain is the
difference between the noise gain of the closed-loop amplifier and the open-loop gain; for a CFA, it is the
difference between the compensation element and the open-loop transimpedance gain. Figure 3 shows
the loop gain performance for a VFA, while Figure 4 illustrates the loop gain performance for a CFA.
80
AOL
Noise Gain (dB)
Open-Loop Gain (dB)
70
60
50
Loop
Gain
40
30
20
Noise Gain = 6dB
10
0
-10
100
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Open-Loop Transimpedance Gain (dBW)
Figure 3. VFA Loop Gain (Typical Performance)
120
100
Loop
Gain
80
20log (1kW)
60
40
20
0
100
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Figure 4. CFA Loop Gain (Typical Performance)
By comparing Figure 3 and Figure 4, we can see that a CFA has less loop gain than a VFA for low-gain
operation if comparing dB to dBΩ directly. This characteristic indicates that a CFA may maintain better
distortion at higher gains than does a VFA if only the gain resistor is changed. Conversely, at low gains,
the VFA achieves much better distortion than the CFA if the gain is adjusted using only the gain resistor.
Note that for both architectures, loop gain varies with frequency. The CFA maintains a higher loop gain
with higher frequency than does a VFA.
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Harmonic Distortion Comparison
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To the first order, the loop-gain of a CFA depends on the feedback resistor. To the second order, it
depends on the noise gain and the inverting input resistance, as can be seen in the following equation.
RCOMP = RF + rI × NG
With:
• RCOMP: Total compensation of a CFA
• RF: Feedback resistor
• rI: Inverting input resistance
• NG: Noise Gain
As the NG increases, RF can be reduced to optimize the bandwidth.
As an example, the relationship between distortion and the amplifier architecture for low gain performance
is shown in the harmonic distortion graphs for the OPA2889 and OPA683. These plots, tested under the
same conditions to 1MHz, are shown in Figure 5 and Figure 6, respectively.
LOW-FREQUENCY HARMONIC DISTORTION
GAIN = -1V/V
Harmonic Distortion (dBc)
-85
RL = 500W
VO = 2VPP
VS = ±5V
-90
-95
-100
-105
-110
-115
HD2 (dBc)
HD3 (dBc)
-120
1k
10k
100k
1M
Frequency (Hz)
Figure 5. OPA2889 Low-Frequency Harmonic Distortion (Gain of –1V/V)
LOW-FREQUENCY HARMONIC DISTORTION
GAIN = -1V/V
-85
RL = 500W
VO = 2VPP
VS = +4V
Harmonic Distortion (dBc)
-90
-95
-100
-105
-110
-115
-120
HD2 (dBc)
HD3 (dBc)
-125
-130
1k
10k
100k
1M
Frequency (Hz)
Figure 6. OPA683 Low-Frequency Harmonic Distortion (Gain of –1V/V)
Note that as the open-loop gain for a VFA and the open-loop transimpedance gain for a CFA decrease,
the harmonic distortion deteriorates.
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Harmonic Distortion Comparison
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The open-loop transimpedance gain of the OPA683 is shown in Figure 7. Here, the roll-off frequency of
the open-loop transimpedance gain matches the increase in distortion in Figure 6. The same relation
between roll-off of the open-loop gain and corner frequency at which the distortion increases can be
observed for the OPA2889, as Figure 8 illustrates.
OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE
0
20log (ZOL)
100
-30
80
Open-Loop Phase (°)
Open-Loop Transimpedance Gain (dB)
120
-60
60
-90
Ð ZOL
40
-120
20
-150
0
1k
10k
100k
1M
10M
-180
100M
Frequency (Hz)
Figure 7. OPA683 Open-Loop Transimpedance Gain Performance
OPEN-LOOP GAIN AND PHASE
120
0
Open-Loop Phase
80
-30
-60
Open-Loop Gain
60
-90
40
-120
20
-150
0
-180
-20
Open-Loop Phase (°)
Open-Loop Gain (dB)
100
-210
100
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Figure 8. OPA2889 Open-Loop Gain Performance
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Harmonic Distortion Comparison
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Keeping in mind that CFAs do have an advantage for high-gain circuits, how does the OPA683 compare
to a high-bandwidth, decompensated VFA? Figure 9 and Figure 10, respectively, compare the OPA847 to
the OPA683.
LOW-FREQUENCY HARMONIC DISTORTION
GAIN = -10V/V
-85
RL = 1kW
VO = 2VPP
Harmonic Distortion (dBc)
-90
-95
-100
-105
-110
HD2 (dBc)
HD3 (dBc)
-115
-120
-125
-130
-135
1k
10k
100k
1M
Frequency (Hz)
Figure 9. OPA847 Low-Frequency Harmonic Distortion (Gain of –10V/V)
LOW-FREQUENCY HARMONIC DISTORTION
GAIN = -10V/V
-90
RL = 1kW
VO = 2VPP
Harmonic Distortion (dBc)
-95
-100
-105
-110
-115
-120
HD2 (dBc)
HD3 (dBc)
-125
-130
1k
10k
100k
1M
Frequency (Hz)
Figure 10. OPA683 Low-Frequency Harmonic Distortion (Gain of –10V/V)
Recall that at a gain of 100V/V, the OPA683 continues to have a bandwidth of 35MHz. The OPA847, with
a GBP of 3900MHz, would achieve 39MHz for the same gain. Distortion, as shown in Figure 9, is very
close at 100kHz to that of the OPA683; but the OPA847 requires 18.1mA to achieve this level of
performance, whereas the OPA683 requires only 0.94mA.
Why then would we use the OPA847?
This amplifier, with its much higher quiescent current, also offers many specifications that the OPA683
cannot achieve—such as very low noise and (for a high-speed amplifier) relatively good dc precision. On
the other hand, the OPA683 has more drive capability.
All in all, the final application dictates the amplifier requirements; but low-power devices should not
necessarily be excluded in favor of higher dissipation amplifiers when they have adequate performance for
the end application.
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Low-Power Filtering
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Low-Power Filtering
There are several common filtering architectures, among which two active-filter approaches are prominent:
the multiple-feedback filter and the Sallen-Key filter.
4.1
MFB Filters
The multiple-feedback filter (MFB) is also called an infinite gain filter because the operational amplifier
operates as an integrator. A typical fully-differential MFB filter is shown in Figure 11 with a 2MHz
Butterworth filter frequency response shown in Figure 12.
+12V
6kW
VCM
50W
1/2
OPA2889
1000pF
6kW
R3A
232W
R1A
402W
C1A
129pF
R2A
402W
C2
257pF
VIN
R2B
402W
C1B
129pF
R3B
232W
R1B
402W
50W
VOUT
1/2
OPA2889
VCM
Figure 11. MFB Filter Topology Using the OPA2889
3
Gain (dB)
0
-3
-6
-9
-12
10k
100k
1M
10M
Frequency (Hz)
Figure 12. MFB Filter Frequency Response
In the circuit shown in Figure 11, the noninverting inputs are used to set the common-mode voltage (VCM)
at the output of the differential circuit. In this case, it is generated by two 6kΩ resistors that set the
reference at the mid-supply, and bypassed by a 1nF capacitor. This capacitor eliminates the
high-frequency noise contribution of the 6kΩ resistors. A 50Ω series resistance on the noninverting input
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helps isolate any LC parasitic elements and avoids potential oscillations. The common-mode gain for each
of the amplifier is 1V/V. Consequently, a unity-gain stable amplifier is required. A non-unity-gain stable
amplifier may be used in this application to provide better distortion, if necessary, at the expense of
additional components in order to provide the adequate noise gain shaping to the common-mode gain of
the amplifier.
Note that because of the capacitance across the feedback path, a VFA is required for this filter. CFA use
is not recommended with this filter architecture. The filter design here is a Butterworth filter and has been
implemented using the following component ratios:
1
fo =
2pRC
•
• R1 = R2 = 0.65 • R
• R3 = 0.375 • R
• C1 = C
• C2 = 2 • C
One advantage of this type of filter is the independent setting of characteristic pulsation and the quality
coefficient ωo and Q. This topology is normally used in filters that have high Qs and require a high gain.
4.2
Sallen-Key Filters
A Sallen-Key topology is normally used to set the gain independently of the filter. In a unity-gain
configuration, this filter is usually applied in filters with high-gain accuracy and low Q. A typical Sallen-Key
filter circuit, as Figure 13 illustrates, shows a single-supply, low-pass filter with a 4V/V gain. The dc noise
gain is set to 1V/V because of the series capacitor in the gain. With this dc isolation for the gain, the bias
can be easily set by a resistor divider at the filter input. This technique is accomplished here with two
12.5kΩ resistors. The filter resistors and capacitors have been adjusted to provide a Butterworth (Q =
0.707) response with a ω0 = 2π • 5MHz. This approach gives a flat passband response with a –3dB cutoff
at 5MHz.
+5V
Supply Decoupling
Not Shown
100pF
12.5kW
0.1mF
157W
446W
VIN
0.1mF
150pF
VOUT
OPA683
12.5kW
1kW
1.4kW
467W
0.1mF
Figure 13. Single-Supply Sallen-Key Filter Implementation
The OPA683 provides an exceptionally capable gain block for implementing Sallen-Key-type filters.
Typically, the bandwidth interaction with gain settings for low-power amplifiers constrain these filters to
using unity-gain amplifiers. Because the OPA683 holds very high bandwidth to high gains, however,
applications that provide signal gain as well as the desired filter shape are easily implemented.
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Multiplying DAC Transimpedance Amplifier
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Figure 14 shows an example of a 5MHz, second-order, low-pass filter where the amplifier provides a
voltage gain of 4V/V. This single-supply implementation (applicable to single +12V operation as well)
consumes only 5.1mW quiescent power.
LOW-POWER, 5MHz LOW-PASS ACTIVE FILTER
15
12
Gain (dB)
9
6
3
0
-3
-6
-9
1k
10k
100k
1M
10M 20M
Frequency (Hz)
Figure 14. Butterworth Low-Pass Filter Frequency Response
5
Multiplying DAC Transimpedance Amplifier
Multiplying digital-to-analog converters (DACs), such as the DAC7822, can make good use of the
low-power, high slew rate VFA. A typical circuit, shown in Figure 15, shows the OPA890 used as the
output transimpedance driver. Note that a CFA can be used in transimpedance applications; however,
because the feedback element is also the compensation element, a CFA would generally provide very
little flexibility in a design where bandwidth is important. Additionally, in a VFA transimpedance application,
a feedback capacitance controls the frequency response peaking and achieves unconditional stability.
Such a technique is not recommended for CFAs because it may lead to instability under the best case
conditions.
+5V
VDD
GND
DB0
DB1
VREF
DB2
½
DB3
R1
DB4 DAC7822 R
FB
DB5
DB6
IOUT1
DB7
IOUT2
DB8
DB9
R2
DB10
R2_3
DB11
R3
-5V
2.5pF
+7.5V
VOUT
OPA890
0V £ VOUT £ 5V
5.56kW
0.1mF -2.5V
Figure 15. Multiplying DAC Transimpedance Driver
In the schematic illustrated in Figure 15, the transimpedance gain is set by a resistor internal to the
DAC7822. The 2.5pF capacitance was selected to achieve a flat frequency response. On the inverting
node, a 5.56kΩ resistor is used to minimize dc offset error. A 0.1µF capacitor was placed in parallel to this
resistance to minimize the noise contribution for frequency above 300Hz.
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Note that in order to achieve maximum amplitude, the supply voltages were set at +7.5V and –2.5V for the
amplifier, allowing the output voltage to swing between 0V and 5V. Figure 16 shows the frequency
response of this circuit.
DAC FREQUENCY RESPONSE
83
77
Gain (dB)
71
65
59
53
47
41
100k
1M
10M
100M
Frequency (Hz)
Figure 16. Multiplying DAC Frequency Response Using the OPA890 as Output Stage
Now look more closely at the interaction of the OPA890 with the DAC7822. We can see that the output
impedance out of the DAC7822 is not high impedance, but varies between 4.5kΩ and 22.1kΩ (excluding
code 000h) for a 10kΩ nominal VREF input resistance. The IOUT resistance changes are directly related to
the code change. This relatively low impedance has multiple effects when an operational amplifier is used.
Some of the effects that apply to all amplifier technologies are:
• the noise gain of the amplifier changes for each code;
• the output offset voltage of the amplifier changes for each code because of the input offset voltage;
• the input offset current cannot be cancelled. The effects of the input bias current can be reduced, but
not eliminated, by selecting a CMOS amplifier instead of a bipolar amplifier, thus affecting the total
output offset voltage of the amplifier with each code.
• The noninverting pin of the amplifier must be tied to ground and cannot be used to create a dc offset to
center the output voltage to any desired value.
The following analysis excludes the input offset current. The total output offset voltage variations, as a
result of the code changing internally to the DAC, can be expressed as:
ΔVOSO = +ΔNG • { [ (RF || ROUT1) – RS] + VOS}
where:
4.5kΩ ≤ ROUT1 ≤ 22.1kΩ
RF = 10kΩ
Using this value, the variation of the parallel combination of RF and ROUT1 can be constrained to:
4.19kΩ ≤ (RF || ROUT1) ≤ 6.88kΩ
In order to optimize the bias current cancellation, RS is selected to be the average of those limiting
numbers, or:
(6.88kW + 4.19kW)
RS =
= 5.56kW
2
Looking at the variation for each code, the total error (when including all codes) is ~3.9 LSB for the
OPA890.
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Conclusion
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Notice that most of the error occurs at the first few codes (0, 1, 2). Excluding these codes from the
analysis then yields the result shown in Table 2.
Table 2. DC Accuracy vs Code
Codes
Total Error Because of
VOS and IB
All codes
3.9 LSB
Excluding code 0
2.5 LSB
Excluding code 0 and 1
2 LSB
Excluding code 0, 1 and 2
1.83 LSB
Note that 1LSB = 1.221mV in the circuit shown in Figure 13.
Eliminating a few codes increases the resolution of this high-speed, low-power, multiplying DAC
transimpedance amplifier solution.
6
Conclusion
Relatively high bandwidth and high slew rate devices are now available with low power dissipation. This
report has discussed the some of the trade-offs between various architectures as well as offered a better
understanding of the specification, along with suggestions and solutions for low-power applications.
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specifically designated by TI as military-grade or "enhanced plastic." Only products designated by TI as military-grade meet military
specifications. Buyers acknowledge and agree that any such use of TI products which TI has not designated as military-grade is solely at
the Buyer's risk, and that they are solely responsible for compliance with all legal and regulatory requirements in connection with such use.
TI products are neither designed nor intended for use in automotive applications or environments unless the specific TI products are
designated by TI as compliant with ISO/TS 16949 requirements. Buyers acknowledge and agree that, if they use any non-designated
products in automotive applications, TI will not be responsible for any failure to meet such requirements.
Following are URLs where you can obtain information on other Texas Instruments products and application solutions:
Products
Amplifiers
Data Converters
DLP® Products
DSP
Clocks and Timers
Interface
Logic
Power Mgmt
Microcontrollers
RFID
RF/IF and ZigBee® Solutions
amplifier.ti.com
dataconverter.ti.com
www.dlp.com
dsp.ti.com
www.ti.com/clocks
interface.ti.com
logic.ti.com
power.ti.com
microcontroller.ti.com
www.ti-rfid.com
www.ti.com/lprf
Applications
Audio
Automotive
Broadband
Digital Control
Medical
Military
Optical Networking
Security
Telephony
Video & Imaging
Wireless
www.ti.com/audio
www.ti.com/automotive
www.ti.com/broadband
www.ti.com/digitalcontrol
www.ti.com/medical
www.ti.com/military
www.ti.com/opticalnetwork
www.ti.com/security
www.ti.com/telephony
www.ti.com/video
www.ti.com/wireless
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2009, Texas Instruments Incorporated
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