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Texas Instruments AN-1002 ADC16071/ADC16471 Analog Layout and Interface Design Considerations Application notes
LM833
AN-1002 ADC16071/ADC16471 Analog Layout and Interface Design Considerations
Literature Number: SNOA350
National Semiconductor
Application Note 1002
Mark Seiders
September 1995
INTRODUCTION
The ADC16071 and the ADC16471 are 16-bit oversampling
delta-sigma (DR) analog to digital converters that are capable of delivering high performance signal conversions at
data output rates up to 192 kSps (kilo samples per second).
The ADC16071/ADC16471’s ultimate performance is dependent on the analog interface, the digital interface, and
the printed circuit board on which it is placed. While the
board design and layout can sometimes be taken for granted in lower resolution ADC applications, it is critical in obtaining best performance from high-resolution ADCs. Extracting all of the performance that the ADC16071/
ADC16471 is capable of delivering requires special attention to such areas as board layout, ground planes, power
supply bypassing, power supply routing, socketing, clock
generation, signal routing, and analog signal conditioning.
VINb that has a common mode voltage at or below VMID,
where VMID is an output pin on the ADC16071/ADC16471
with a potential equal to one half of the analog supply
(VA a /2). The ADC16471 has an internal 2.5V bandgap reference that sets VREF a e VMID a 1 .25V and VREFb e
VMID b 1.25V. The ADC16071 requires an externally applied reference whose range (VREF a b VREFb) can be
varied from 1V to VA a . See Reference Voltage Generation
for the ADC16071 for examples of driving the ADC16071’s
reference inputs.
ANALOG INPUT RANGE
The ADC16071/ADC16471 produces a 16-bit, twos compliment output according to the following equation:
32768 # (VIN a b VINb)
(VREF a b VREFb)
The signals applied to VIN a and VINb must have potentials
between the analog supply (VA a ) and analog ground
(AGND). For accurate conversions, the absolute difference
between VIN a and VINb should be less than the difference
between VREF a and VREFb. Best harmonic performance
will result when a balanced voltage is applied to VIN a and
output e
ANTI-ALIASING FILTER CONSIDERATIONS
One of the biggest advantages of oversampling DR ADCs is
their relaxed requirements for anti-aliasing filters. With any
ADC, aliasing will not occur provided that no frequencies
greater than one half the sampling rate are present at the
analog input pins. To prevent aliasing and maximize bandwidth with a Nyquist rate (non-oversampling) ADC, the analog anti-aliasing filter typically must have a flat response up
to about 0.45 # fS and attenuate all frequencies above 0.5 #
fS to levels below the noise floor. Designing a filter with
such a sharp drop-off can be difficult and expensive, requiring precision components and additional board space. Furthermore, analog ‘‘brick wall’’ filters usually have a non-linear phase response. If phase distortion is a concern, the
implementation of such a filter can be even more difficult, if
not impossible. Figure 1 compares the anti-aliasing filter requirements for a Nyquist rate ADC to those of an oversampling ADC such as the ADC16071/ADC16471.
ADC16071/ADC16471 Analog Layout and Interface Design Considerations
ADC16071/ADC16471
Analog Layout and
Interface Design
Considerations
TL/H/12482 – 1
M e Oversampling Ratio
fS e Output Data Rate
fBB e Desired Frequency (Base Band)
FIGURE 1. AAF Requirements for Nyquist Rate ADC vs Oversampling ADC
AN-1002
C1995 National Semiconductor Corporation
TL/H/12482
RRD-B30M115/Printed in U. S. A.
are those above M # fS b fBB, which are aliased into the
baseband. Thus the external anti-aliasing filter for the
ADC16071/ADC16471 need only cut off frequencies above
M # fS b fBB. The ADC16071/ADC16471 has an oversampling ratio of 64 (M e 64). This ratio allows the ADC16071/
ADC16471’s anti-aliasing filter’s critical point of attenuation
to be pushed out 127 times (63.5 # fS vs 0.5 # fS) higher
than what it would need to be for a Nyquist rate converter
with equivalent output bandwidth!
The ADC164071/ADC16471’s modulator samples the analog input at a rate equal to fCLK/2, where fCLK is the frequency of the clock applied to the ADC16071/ADC16471’s
CLK pin. The output data rate (fS) is equal to (/64 (the oversampling ratio) of the modulator’s sample rate, or fCLK/128.
The analog baseband (fBB) is equal to one half of the data
output rate, or fCLK/256.
By oversampling the analog input at 64 times the Nyquist
rate (fS) for the desired analog baseband, the ADC16071/
ADC16471 pushes out the point at which aliasing occurs.
This dramatically relaxes the performance requirements for
the anti-aliasing filter. The critical point of attenuation for an
oversampling ADC’s anti-aliasing filter is typically pushed
out even further because of on-chip digital filtering. The
ADC16071/ADC16471 contains a 246 tap internal, linear
phase, finite impulse response (FlR) filter that cuts off all
frequencies above the analog baseband (fBB).
Aliased frequencies are mirrored about half the sampling
rate of the modulator, (M # fS)/2. Therefore, any frequencies between (M # fS)/2 and M # fS are aliased into the
range between (M # fS)/2 and DC. Since all frequencies
greater than the baseband (fBB) are filtered out by the onchip digital filters, the only potentially damaging frequencies
PASSIVE RC ANTI-ALIASING FILTER NETWORK
A recommended, simple anti-aliasing input network is the
first-order, passive, low-pass RC filter shown in Figure 2.
This network has a flat frequency and linear phase response in the analog baseband, and eliminates analog frequency components above M # fS b fBB that may cause
aliasing. In addition, C1, C2, and C3 provide a charge reservoir for the ADC16071/ADC16471 modulator’s input capacitors (see Analog Interface Amplifier Considerations ). The
filter’s b3 dB cutoff frequency is:
1
fc e
6q RC
where R e R1 e R2 and C e C1 e C2 e C3
TL/H/12482 – 2
FIGURE 2. Simple, Passive, Low-Pass Input Network
2
To ensure that the filter’s frequency response is flat in the
baseband and that it provides sufficient attenuation to frequencies above M # fS b fBB, the values of R and C should
be chosen so that the filter’s 3 dB cutoff is between fCLK/
250 and fCLK/100. With an fCLK of 24.576 MHz (192 kHz
data output rate), typical values for R and C are 100X and
3300 pF, respectively. These values result in a 3 dB cutoff
equal to approximately 160 kHz, or fCLK/150 and an attenuation of about 40 dB at M # fS b fBB.
LEVEL SHIFTING THE INPUT SIGNAL
For best conversion performance, the signal applied to the
ADC16071/ADC16471’s analog input pins, VIN a and VINb,
should be a balanced AC signal with a common mode voltage at or below one-half of the ADC’s supply voltage (VMID).
The simplest way to do this is to capacitively couple the
applied input signal and connect the VMID output to VIN a
and VINb through 4.7 kX resistors (Figure 3).
TL/H/12482 – 3
FIGURE 3. Capacitively Coupling and Level Shifting a Balanced Input Signal
3
the modulator is filtered by the internal, brick wall FIR. See
Appendix: Noise Shaping in Delta Sigma Modulators for further discussion.
RELATION BETWEEN CAPACITOR DIELECTRIC AND
SIGNAL DISTORTION.
For any capacitors connected to the ADC16071/
ADC16471’s analog inputs, the dielectric plays an important
role in determining the amount of distortion generated in the
input signal. The dielectric must have low dielectric absorption. This requirement is fulfilled by using capacitors that
have film dielectrics. Of these, polypropylene and polystyrene are the best. These are followed by polycarbonate and
mylar. If ceramic capacitors are chosen, use only capacitors
with NPO dielectrics.
Due to overload in the modulator’s comparators, as the analog input amplitude approaches full scale, the modulator’s
feedback coefficients begin to change. This tends to reduce
the cutoff frequency of the modulator’s noise shaping characteristic, allowing more quantization noise to pass in the
analog baseband. Since anything passed in the analog
baseband won’t be filtered by the FIR, added quantization
noise will be present in the output of the ADC16071/
ADC16471. When examining an output spectrum from the
ADC16071/ADC16471, this additional quantization noise
can be seen as a slight raising of the noise floor toward the
upper end of the analog baseband and increased odd harmonic distortion. Figures 4 and 5 show output spectra from
the same ADC16071 with input amplitudes of b3 dB and
b 0.8 dB below full scale (dBFS), respectively. The raised
noise floor and additional odd harmonic distortion are visually noticeable with a b0.8 dB input.
INPUT SIGNAL MAGNITUDE AND OVERLOAD
Following the switched capacitor input of the ADC16071/
ADC16471, the analog input and reference voltages are fed
into a pseudo fourth order, MASH (Multistage noise Shaping) delta sigma modulator. The modulator is designed to
act as a high-pass filter to the quantization noise introduced
by its comparators. This high-pass noise shaping characteristic minimizes the amount of quantization noise present in
the baseband at the output of the modulator. The higher
frequency quantization noise that is present at the output of
TL/H/12482 – 4
FIGURE 4. Output Spectrum with a b3 dB Input Amplitude
TL/H/12482 – 5
FIGURE 5. Output Spectrum with a b0.8 dB Input Amplitude
4
TL/H/12482 – 6
FIGURE 6. Dynamic Performance Degradation Due to Modulator Overload
to recover quickly from the transient current requirements of
the switched capacitor input. The capacitors used in the
recommended anti-aliasing filter configuration (Figure 2)
help by acting as charge reservoirs for these current spikes,
but it is still recommended that amplifiers be used that have
a minimum gain bandwidth of one half the frequency of the
clock. For example, when the clock frequency is 24.576
MHz, the gain-bandwidth of any op-amps driving the inputs
of the ADC16071/ADC16471 should be at least 13 MHz.
The LM6218 and the LM833 are good choices for buffering
or amplifying signals applied to the ADC16071/ADC16471.
These amplifiers have sufficient bandwidth and slew rate
and produce sufficiently low distortion and noise. Additionally, they are available in a dual package, saving board space
and component count.
To help source amplifiers settle faster, a series resistance
(50X to 100X) may be placed between the amplifier’s output and the ADC16071/ADC16471’s inputs. This is already
accomplished when the passive low-pass network as
shown in Figure 2 is connected between the amplifier’s output and the ADC16071/ADC16471’s inputs.
At room temperature, the ADC16071/ADC16471 performs
well (meets its published specifications) with input amplitudes up to b3 dB FS. As the input amplitude exceeds
b 3 dB FS, performance begins to degrade. At b 2 dB FS,
the SNR is about 2 dB worse than with a b3 dB FS input.
With a b1.4 dB FS input, the SNR is about 6 dB worse. With
a b0.66 dB FS input, the SNR drops by more than 10 dB
from the b3 dB FS input case. Figure 6 illustrates the typical degradation in the dynamic performance of the
ADC16071/ADC16471 as the input amplitude approaches
full scale. At higher temperatures, the nonlinearities may be
a factor at slightly lower input amplitudes, but overload
noise and distortion shouldn’t be experienced over the entire b40§ C to a 85§ C temperature range with input ampIitudes of b6 dB FS or Iess.
ANALOG INTERFACE AMPLIFIER CONSIDERATIONS
The input impedance of the ADC16071/ADC16471, due to
the effective resistance of the switched capacitor input, varies as follows:
1012
2.35 # (fCLK/2)
where fCLK is the frequency of the clock applied to the
ADC16071/ADC16471’s CLK pin.
The current required during the act of switching, or connecting, the input sampling capacitors between the source circuitry and the ADC16071/ADC16471’s modulator input can
cause momentary instability in amplifiers with limited gainbandwidth. To overcome this problem, amplifiers used to
drive the inputs of the ADC16071/ADC16471 must be able
ZIN e
SINGLE-ENDED BIPOLAR INPUT TO BALANCED
UNIPOLAR OUTPUT BUFFER
The ADC16071/ADC16471 exhibits the best distortion performance when a balanced AC signal is applied to its analog
inputs that has a common mode offset at or below VMID.
The circuit in Figure 7 can be used to convert a single-ended, bipolar signal centered about ground to a balanced signal centered about VMID.
5
TL/H/12482 – 7
FIGURE 7. Unbalanced-to-Balanced Buffer
source will have to sink an average of about 270 mA and up
to a peak of about 540 mA. An alternate approach is to
connect a coupling capacitor between the output of the signal source and the input to the circuit in Figure 7.
When the ADC16071/ADC16471 is driven by a balanced
signal, the conversion process will cancel out common
mode noise and reduce harmonic distortion. Figure 8 shows
an output spectrum from an ADC16071 with a single-ended
input signal centered around VMID. Figure 9 shows the output spectrum from the same ADC16071 with same signal
source (without the VMID offset) after it has been converted
to a balanced signal using the circuit in Figure 7. The distortion performance (THD) improves by more than 25 dB when
the input is converted to a balanced signal.
This circuit’s level shifting is accomplished using the
ADC16071/ADC16471’s on-chip one-half supply voltage
output, VMID. The VMID output voltage is divided in half and
applied to the non-inverting input of the circuit’s first inverting buffer. VMID is divided in half because the difference
between the DC offset at the input to the circuit (0V) and the
voltage at the non-inverting input of the first buffer (VMID/2)
will see a gain of two. This results in an offset voltage equal
to VMID at the output of the first inverting buffer. VMID is also
applied to the non-inverting input of the circuit’s second inverting buffer. The outputs are two 180§ out-of-phase signals (VIN a and VINb) that swing above and below the VMID
voltage.
It is important to note that because of the difference in potential between the inverting input of the first buffer and the
common mode output of the signal source, the signal
6
TL/H/12482 – 8
FIGURE 8. Output Spectrum with Single-Ended Input
TL/H/12482 – 9
FIGURE 9. Output Spectrum with Balanced Input
half and applied to the non-inverting inputs of each of the
inverting buffers. To maintain the input signal’s original
phase, the positive inverting buffer’s output is applied to
VINb and the negative inverting buffer output is applied to
VIN a .
BALANCED BIPOLAR INPUT TO BALANCED UNIPOLAR
OUTPUT BUFFER
The circuit shown in Figure 10 simply buffers and level shifts
a balanced analog signal centered about ground. The
ADC16071/ADC16471’s VMID output voltage is divided in
TL/H/12482 – 10
FIGURE 10. Balanced Bipolar to Unipolar Buffer
7
POWER SUPPLY VOLTAGES FOR IMPROVED
PERFORMANCE
REFERENCE VOLTAGE GENERATION FOR THE
ADC16071
The ADC16071 requires an external reference voltage
source. It must have low output noise and be stable. A suggested circuit that generates a stable reference voltage that
can be adjusted between 1.9V and 2.6V is shown in Figure
11. It uses the LM4041-ADJ adjustable shunt bandgap reference. The potentiometer, RP, adjusts the output between
1.9V and 2.6V. If a fixed output is desired, replace the R1,
RP, and R2 resistor string with the fixed resistor string
shown in Figure 12. Use the equation in Figure 12 to determine the fixed resistor values.
While adequate performance will be achieved by operating
the ADC16071/ADC16471 with a 5V connected to VA a ,
VM a and VD a , dynamic performance, as indicated by
SINAD, can be further enhanced by changing VD a to a voltage lower than VA a and VM a . By setting VD a to 3.5V and
VA a and VM a to 5.5V, improvements of up to 5 dB will be
seen in both noise floor and harmonic performance. The
improved performance can be attributed to the reduction of
digital switching noise due to the lower digital supply voltage.
TL/H/12482 – 11
FIGURE 11. 1.9V to 2.6V Adjustable Reference for the ADC16071
TL/H/12482 – 12
From the LM4041-ADJ datasheet:
R2
VOUT
e
b1
R1
1.24 b (1.3 c 10-3) VOUT
FIGURE 12. 2.0V Fixed Reference for the ADC16071
8
shoot of no more than 100 mVPP) and has rise and fall
times in the range of 3 ns – 10 ns (10% – 90%). The Ecliptek
(EC1100 series) and SaRonix (NCH060 and NCH080 series) are recommended crystal clock oscillators for driving
the CLK input of the ADC16071/ADC16471. Both of these
families use HCMOS logic circuitry for fast rise and fall
times.
Overshoot and ringing on the clock-signal edge that a converter uses to internally clock its operation will result in increased noise and distortion. The effects of overshoot and
ringing can be minimized by using a series damping resistor
between the output of the clock-signal source and the
ADC16071/ADC16471’s CLK pin. The value of the resistor
used is dependent on the board layout, and usually ranges
from 25X to 150X. A typical starting value is 50X.
PRINTED CIRCUIT BOARD CONSIDERATIONS
Ground Planes and Signal Trace Layers
Analog and digital ground planes are essential in extracting
the best performance from high-resolution delta-sigma converters. Ground planes reduce ground return impedances to
low levels, ensuring that power supply bypass capacitors
have the lowest AC-resistance path possible. The
ADC16071/ADC16471’s conversion performance is optimized using separate analog and digital ground planes. The
ground planes should be connected together at a single
point, the power supply ground connection.
Best performance is achieved by ensuring that the trace/
ground plane association integrity is maintained. All analog
and digital traces are placed over, or within, their associated
ground plane.
In a multilayer printed circuit board with separate ground
and trace layers, the supply and signal trace layers should
be ‘‘sandwiched’’ between the analog and digital ground
plane layers (Figure 13). The outer ground plane layers act
as shields, attenuating noise from external sources and
from internal digital switching.
Analog signal, digital control signal, and power supply
traces should be separated from each other. If the physical
board layout prevents adequate separation of the digital,
analog, and power supply traces, they should be placed on
different circuit board layers and cross at right angles.
SOCKET CONSIDERATIONS FOR IMPROVED POWER
SUPPLY BYPASSING
The ADC16071/ADC16471 is clocked at very high frequencies. This high frequency clocking produces high frequency
current spikes and glitches on the power supply lines. If not
attenuated, these power supply perturbations will degrade
the ADC16071/ADC16471’s conversion performance.
For all integrated circuits, the power supply inputs should
always be viewed as signal inputs. The internal circuit will
treat any AC signal appearing on the power supply voltage
as another input signal.
The ADC16071/ADC16471’s power supply rejection (PSR)
is high at low frequencies and usually decreases as frequency increases. Thus, at the high frequencies used to clock
the ADC16071/ADC16471, the PSR is low. Therefore, external power supply bypass capacitors are needed to provide the ADC16071/ADC16471’s transient current requirements and to improve the PSR by attenuating the high frequency noise created by high speed digital switching.
INPUT NETWORK LAYOUT AND ROUTING
Careful consideration must be observed concerning the layout and placement of the input network connected to the
two balanced inputs, VIN a and VINb. The layout should be
balanced and symmetrical with respect to the VIN a and
VINb pins. All associated traces should have equal trace
length and width dimensions. This symmetry should be extended back to the outputs of circuitry that drives VIN a and
VINb.
CLOCK SIGNAL GENERATION AND ROUTING
The ADC16071/ADC16471 requires a low jitter clock signal
applied to its CLK pin that is free of ringing (over/under-
TL/H/12482 – 13
FIGURE 13. PCB Layout with ‘‘Sandwiched’’ Power Supply and Trace Layers
9
rectly under the package and between the pins using the
shortest possible trace lengths (Figure 14). This ‘‘socket’’ is
created by using individual machined socket-pins. These
pins require a hole size of 58 mils. This hole size ensures
that only the topmost portion of the pin remains above the
circuit board. These ‘‘socket’’ pins will tightly grip the
ADC16071/ADC16471 plastic package’s pins, further reducing a possible source of performance degradation
caused by loose fitting sockets.
Suggested power-supply bypassing consists of surfacemount 0.1 mF monolithic ceramic and 10 mF tantalum capacitors. When using the ADC16071/ADC16471 in the DIP
package, the bypass capacitors’ size is limited by the distance between the DIP package pins. When placed under
an ADC16071/ADC16471 DIP package using a modified
‘‘socket’’ as in Figure 14, the 0.1 mF SMD capacitor’s physical size is limited to package number 0805. The 10 mF SMD
capacitor’s physical size is limited to package number 1210.
As the distance between the ADC16071/ADC16471 and its
bypass capacitors increases, so do the bypass capacitor
lead inductances. Increased lead inductances result in decreased high frequency attenuation. At the frequencies
used to clock the ADC16071/ADC16471 (fCLK e 24.576
MHz), even a typical lead-length (bond wires, package lead,
and capacitors leads) of 10mm has an inductance of 20 nH
or 3X impedance. This impedance reduces the efficiency of
the bypass capacitors. Spikes and glitches riding on the DC
supply voltage are most efficiently attenuated when power
supply bypass capacitors are placed as close as possible to
the power supply pins.
Ideally the ADC16071/16471 should be soldered directly to
the printed circuit board. This minimizes lead length between power supply and ground pins and power supply bypass components. Even a lead-length increase of 0.125×
can degrade SINAD performance by 5 dB–15 dB.
When using the ADC16071/ADC16471 in the molded Dualin-Line Package (DIP), mounting the ADC in a modified
‘‘socket’’, allows surface-mount capacitors to be placed di-
TL/H/12482 – 14
FIGURE 14. ADC16071/ADC16471 PCB Mounting and Bypass Capacitor Positioning
10
APPENDIX
NOISE SHAPING IN DELTA SIGMA MODULATORS
TL/H/12482 – 15
fS e Output Data Rate
FIGURE A. Delta Sigma ADC Block Diagram
TL/H/12482 – 16
M e Oversampling Ratio
FIGURE B. Block Diagram of a 1-Bit, First Order Delta Sigma Modulator
For example, a modulator output of: 1,0,1,1,1,0,0,0,1,0, represents an analog input halfway between positive and negative full scale (5 out of a possible 10 ones).
Because of the crude approximation made by the comparator of a DR modulator (it is quantizing with only 1-bit of resolution), a large amount of quantization noise is introduced
into the system. But because of the noise ‘‘shaping’’ characteristic that is inherent to the design of DR modulators,
much of the quantization noise introduced by the modulator’s comparators is pushed beyond the frequency band of
interest (fBB), where it may be filtered digitally.
A delta-sigma converter consists of a DR modulator that is
essentially a high speed, low resolution ADC, and a DSP
block that trades time for resolution (i.e., 64 # fS with 1-bit to
fS with 16 bits) and filters the output of the modulator. The
DSP block typically consists of a comb filter, sometimes
called a decimator, and an FIR filter that has a ‘‘brick wall’’
low-pass characteristic.
Figure B is a block diagram of a 1-bit, first order modulator.
The difference (D) between the analog input and the comparator’s previous output is integrated (R) in such a manner
that the average of the digital output is equal to the analog
input.
The ones and zeros at the modulator’s output represent the
comparator’s positive and negative full scale, respectively.
11
DOUT e (VIN b DOUT) # H(s) a q
DOUT # (1 a H(s)) e VIN a q
VIN # H(s)
q
a
DOUT e
1 a H(s)
1 a H(s)
VIN
q#s
.
a
. . DOUT e
1as
1as
H(s) e
1
s
TL/H/12482 – 17
FIGURE C. Noise Shaping in Delta Sigma Modulator
If we make the approximation that the comparator of Figure
B can be treated as the addition of quantization noise (q)
that has a ‘‘white’’ spectral distribution (uniform energy at all
frequencies) and uncorrelated to the analog input, then the
substitution shown in the block diagram of Figure C may be
made. From this block diagram, the output of the modulator
may be equated to the difference between the quantized
output, DOUT, and the analog input, VIN, times the transfer
function of the integrator, H(s), plus the quantization noise,
q. From the resulting transfer function for DOUT in terms of
the analog input and the quantization noise, it can be shown
that quantization noise is filtered through a high-pass filter
q#s
,
sa1
while the input signal passes unattenuated at low frequencies (f k fBB kk M # fS). This high-pass function ‘‘shapes’’
the quantization noise out of the baseband, fBB to higher
frequencies where it will be cut off by the digital filtering
within the ADC.
#
By increasing the order of a DR modulator (adding more
integrators to the modulator), the noise shaping effect is
enhanced. Figure D’s curves show how the flat quantization
noise is ‘‘shaped’’ into first-, second-, and third-order modulator characteristics.
DR modulators further reduce the amount of quantization
noise in the baseband by oversampling the input signal. The
quantization noise is assumed to be spread out equally from
DC up to the sample rate of the modulator. As the oversampling ratio is increased, so is the range over which the quantization noise is spread. The total noise does not decrease,
but the density per frequency band does. With a first order
modulator, the theoretical maximum signal-to-quantizationnoise ratio in the baseband can be shown to increase by 9
dB with each doubling of the oversampling ratio.
J
TL/H/12482 – 18
FIGURE D. Noise Shaping Characteristic Curves
12
13
ADC16071/ADC16471 Analog Layout and Interface Design Considerations
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TI products are not authorized for use in safety-critical applications (such as life support) where a failure of the TI product would reasonably
be expected to cause severe personal injury or death, unless officers of the parties have executed an agreement specifically governing
such use. Buyers represent that they have all necessary expertise in the safety and regulatory ramifications of their applications, and
acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products
and any use of TI products in such safety-critical applications, notwithstanding any applications-related information or support that may be
provided by TI. Further, Buyers must fully indemnify TI and its representatives against any damages arising out of the use of TI products in
such safety-critical applications.
TI products are neither designed nor intended for use in military/aerospace applications or environments unless the TI products are
specifically designated by TI as military-grade or "enhanced plastic." Only products designated by TI as military-grade meet military
specifications. Buyers acknowledge and agree that any such use of TI products which TI has not designated as military-grade is solely at
the Buyer's risk, and that they are solely responsible for compliance with all legal and regulatory requirements in connection with such use.
TI products are neither designed nor intended for use in automotive applications or environments unless the specific TI products are
designated by TI as compliant with ISO/TS 16949 requirements. Buyers acknowledge and agree that, if they use any non-designated
products in automotive applications, TI will not be responsible for any failure to meet such requirements.
Following are URLs where you can obtain information on other Texas Instruments products and application solutions:
Products
Applications
Audio
www.ti.com/audio
Communications and Telecom www.ti.com/communications
Amplifiers
amplifier.ti.com
Computers and Peripherals
www.ti.com/computers
Data Converters
dataconverter.ti.com
Consumer Electronics
www.ti.com/consumer-apps
DLP® Products
www.dlp.com
Energy and Lighting
www.ti.com/energy
DSP
dsp.ti.com
Industrial
www.ti.com/industrial
Clocks and Timers
www.ti.com/clocks
Medical
www.ti.com/medical
Interface
interface.ti.com
Security
www.ti.com/security
Logic
logic.ti.com
Space, Avionics and Defense
www.ti.com/space-avionics-defense
Power Mgmt
power.ti.com
Transportation and Automotive www.ti.com/automotive
Microcontrollers
microcontroller.ti.com
Video and Imaging
RFID
www.ti-rfid.com
OMAP Mobile Processors
www.ti.com/omap
Wireless Connectivity
www.ti.com/wirelessconnectivity
TI E2E Community Home Page
www.ti.com/video
e2e.ti.com
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2011, Texas Instruments Incorporated
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