Texas Instruments | ADS114S0xB Low-Power, Low-Noise, Highly Integrated, 6- and 12-Channel, 4-kSPS, 16-Bit, Delta-Sigma ADC With PGA and Voltage Reference | Datasheet | Texas Instruments ADS114S0xB Low-Power, Low-Noise, Highly Integrated, 6- and 12-Channel, 4-kSPS, 16-Bit, Delta-Sigma ADC With PGA and Voltage Reference Datasheet

Texas Instruments ADS114S0xB Low-Power, Low-Noise, Highly Integrated, 6- and 12-Channel, 4-kSPS, 16-Bit, Delta-Sigma ADC With PGA and Voltage Reference Datasheet
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ADS114S06B, ADS114S08B
SBAS852 – AUGUST 2017
ADS114S0xB Low-Power, Low-Noise, Highly Integrated, 6- and 12-Channel,
4-kSPS, 16-Bit, Delta-Sigma ADC With PGA and Voltage Reference
1 Features
3 Description
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The ADS114S06B and ADS114S08B are precision,
16-bit, delta-sigma (ΔΣ), analog-to-digital converters
(ADCs) that offer low power consumption and many
integrated features to reduce system cost and
component count in applications measuring smallsignal sensors.
1
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•
•
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•
•
•
•
•
•
•
•
Low Power Consumption: As Low as 280 µA
Programmable Gain: 1 to 128
Programmable Data Rates: 2.5 SPS to 4 kSPS
Simultaneous 50-Hz and 60-Hz Rejection at
≤ 20 SPS With Low-Latency Digital Filter
Analog Multiplexer with 12 (ADS114S08B) or 6
(ADS114S06B) Independently Selectable Inputs
Dual-Matched Programmable Current Sources for
Sensor Excitation: 10 µA to 2000 µA
Internal Reference: 2.5 V, 8 ppm/°C (typ) Drift
Internal Oscillator: 4.096 MHz, 2% Accuracy
Internal Temperature Sensor
Self Offset and System Calibration
Four General-Purpose I/Os
SPI-Compatible Interface
Analog Supply: Unipolar (2.7 V to 5.25 V) or
Bipolar (±2.5 V)
Digital Supply: 2.7 V to 3.6 V
Operating Temperature: –40°C to +125°C
Pin-Compatible With the ADS114S0x
These ADCs feature a digital filter that offers lowlatency conversion results and 50-Hz or 60-Hz
rejection for noisy industrial environments. A lownoise, programmable gain amplifier (PGA) provides
gains ranging from 1 to 128 to amplify low-level
signals for resistive bridge or thermocouple
applications. Additionally, these devices integrate a
low-drift, 2.5-V reference that reduces printed circuit
board (PCB) area. Finally, two programmable
excitation current sources (IDACs) allow for easy and
accurate RTD biasing.
An input multiplexer supports 12 inputs for the
ADS114S08B and six inputs for the ADS114S06B
that can be connected to the ADC in any combination
for design flexibility. In addition, these devices include
features such as sensor burn-out detection, voltage
bias for thermocouples, system monitoring, and four
general-purpose I/Os.
The devices are offered in a leadless VQFN-32 or a
TQFP-32 package.
2 Applications
•
•
•
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Sensor Transducers and Transmitters:
Temperature, Pressure, Strain, Flow
PLC and DCS Analog Input Modules
Temperature Controllers
Climate Chambers, Industrial Ovens
Device Information
ORDER NUMBER
ADS114S0xB
PACKAGE (PIN)
BODY SIZE
TQFP (32)
5.0 mm × 5.0 mm
VQFN (32)
5.0 mm × 5.0 mm
Functional Block Diagram
AVDD
REFN0
Burnout
Detect
Excitation
Current
Sources
AINCOM
AIN0
REFP0
Reference
Mux
REFCOM REFOUT
DVDD
2.5-V
Reference
ADS114S06B
ADS114S08B
IOVDD
AIN1
AIN2
Reference
Buffers
VBIAS
AIN3
AIN4
AIN5
Input
Mux
PGA
AIN6 / REFP1
16-Bit û
ADC
START/SYNC
Configurable
Digital
Filter
Serial
Interface
and
Control
AIN7 / REFN1
SCLK
DIN
AIN8 / GPIO0
DOUT/DRDY
AIN9 / GPIO1
Power Supplies
AIN10 / GPIO2
Temperature
Sensor
AIN11 / GPIO3
ADS114S08B
Only
RESET
CS
Burnout
Detect
AVSS
System-, SelfCalibration
DRDY
4.096-MHz
Oscillator
CLK
DGND
Copyright © 2017, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
ADS114S06B, ADS114S08B
SBAS852 – AUGUST 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Device Family Comparison Table ........................
Pin Configuration and Functions .........................
Specifications.........................................................
7.1
7.2
7.3
7.4
7.5
7.6
7.7
7.8
9.6 Register Map........................................................... 54
1
1
1
2
3
3
5
10 Application and Implementation........................ 67
10.1 Application Information.......................................... 67
10.2 Typical Application ................................................ 72
10.3 Do's and Don'ts ..................................................... 77
11 Power Supply Recommendations ..................... 79
11.1
11.2
11.3
11.4
Absolute Maximum Ratings ..................................... 5
ESD Ratings.............................................................. 5
Recommended Operating Conditions....................... 6
Thermal Information .................................................. 6
Electrical Characteristics........................................... 7
Timing Characteristics............................................. 10
Switching Characteristics ........................................ 10
Typical Characteristics ............................................ 13
8
Parameter Measurement Information ................ 19
9
Detailed Description ............................................ 21
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
Programming...........................................................
79
79
79
79
12 Layout................................................................... 80
12.1 Layout Guidelines ................................................. 80
12.2 Layout Example .................................................... 81
13 Device and Documentation Support ................. 82
13.1
13.2
13.3
13.4
13.5
13.6
13.7
13.8
8.1 Noise Performance ................................................. 19
9.1
9.2
9.3
9.4
9.5
Power Supplies .....................................................
Power-Supply Sequencing....................................
Power-On Reset....................................................
Power-Supply Decoupling.....................................
21
22
23
42
46
Device Support......................................................
Documentation Support ........................................
Related Links ........................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
82
82
82
82
82
82
82
83
14 Mechanical, Packaging, and Orderable
Information ........................................................... 83
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
2
DATE
REVISION
NOTES
August 2017
*
Initial release.
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SBAS852 – AUGUST 2017
5 Device Family Comparison Table
PRODUCT
RESOLUTION (Bits)
NUMBER OF INPUTS
ADS114S08B
16
12 analog inputs
ADS114S06B
16
6 analog inputs
6 Pin Configuration and Functions
RHB Package
32-Pin VQFN
Top View
REFOUT
AIN4
3
22
GPIO0/AIN8
AIN3
4
21
AIN2
5
AIN1
AIN0
REFN0
AVSS
AVSS
AVDD
NC
27
26
25
GPIO0/AIN8
GPIO1/AIN9
AIN3
4
21
GPIO1/AIN9
20
GPIO2/AIN10
AIN2
5
20
GPIO2/AIN10
6
19
GPIO3/AIN11
AIN1
6
19
GPIO3/AIN11
7
18
RESET
AIN0
7
18
RESET
START/SYNC
8
17
CLK
DVDD
DGND
IOVDD
DRDY
CS
Not to scale
DOUT/DRDY
16
DVDD
CLK
9
15
13
DRDY
14
12
DOUT/DRDY
DGND
11
IOVDD
10
DIN
SCLK
17
9
8
SCLK
Pad
DIN
Thermal
16
22
15
3
14
REFOUT
AIN4
13
REFCOM
23
12
24
2
11
1
AIN5
10
AINCOM
CS
START/SYNC
28
NC
25
23
REFP0
AVDD
26
2
29
AVSS
27
AIN5
REFN1/AIN7
AVSS
28
REFCOM
30
REFN0
29
24
REFP1/AIN6
REFP0
30
1
31
REFN1/AIN7
31
AINCOM
32
REFP1/AIN6
32
PBS Package
32-Pin TQFP
Top View
Not to scale
NOTE: The analog input functions (AIN6–AIN11) are not available on pins 19 to 22, 31, and 32 for the ADS114S06B.
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Pin Functions
PIN
NO.
DESCRIPTION (1)
NAME
FUNCTION
1
AINCOM
Analog input
Common analog input for single-ended measurements
2
AIN5
Analog input
Analog input 5
3
AIN4
Analog input
Analog input 4
4
AIN3
Analog input
Analog input 3
5
AIN2
Analog input
Analog input 2
6
AIN1
Analog input
Analog input 1
7
AIN0
Analog input
Analog input 0
8
START/SYNC
Digital input
Start conversion
9
CS
Digital input
Chip select; active low
10
DIN
Digital input
Serial data input
11
SCLK
Digital input
Serial clock input
12
DOUT/DRDY
Digital output
Serial data output combined with data ready; active low
13
DRDY
Digital output
Data ready; active low
14
DGND
Digital ground
Digital ground
15
IOVDD
Digital supply
Digital I/O power supply. In case IOVDD is not tied to DVDD, connect a 100-nF (or larger) capacitor to
DGND.
16
DVDD
Digital supply
Digital core power supply. Connect a 100-nF (or larger) capacitor to DGND.
17
CLK
Digital input
External clock input. Connect to DGND to use the internal oscillator.
18
RESET
Digital input
Reset; active low
19
GPIO3/AIN11
Analog input/output
General-purpose I/O (2); analog input 11 (ADS114S08B only)
20
GPIO2/AIN10
Analog input/output
General-purpose I/O (2); analog input 10 (ADS114S08B only)
21
GPIO1/AIN9
Analog input/output
General-purpose I/O (2); analog input 9 (ADS114S08B only)
22
GPIO0/AIN8
Analog input/output
General-purpose I/O (2); analog input 8 (ADS114S08B only)
23
REFOUT
Analog output
Positive voltage reference output. Connect a 1-µF to 47-µF capacitor to REFCOM if the internal
voltage reference is enabled.
24
REFCOM
Analog output
Negative voltage reference output. Connect to AVSS.
25
NC
—
26
AVDD
Analog supply
Positive analog power supply. Connect a 330-nF (or larger) capacitor to AVSS.
27
AVSS
Analog supply
Negative analog power supply
28
AVSS
Analog supply
Negative analog power supply
29
REFN0
Analog input
Negative external reference input 0
30
REFP0
Analog input
Positive external reference input 0
31
REFN1/AIN7
Analog input
Negative external reference input 1; analog input 7 (ADS114S08B only)
32
REFP1/AIN6
Analog input
Positive external reference input 1; analog input 6 (ADS114S08B only)
Pad
Thermal Pad
—
(1)
(2)
4
Leave unconnected or connect to AVSS
RHB package only. Thermal power pad. Connect to AVSS.
See the Unused Inputs and Outputs section for details on how to connect unused pins.
General-purpose inputs and outputs use logic levels based on the analog supply.
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7 Specifications
7.1 Absolute Maximum Ratings (1)
Power-supply voltage
MIN
MAX
AVDD to AVSS
–0.3
5.5
AVSS to DGND
–2.8
0.3
DVDD to DGND
–0.3
3.9
IOVDD to DGND
UNIT
V
–0.3
5.5
Analog input voltage
AINx, GPIOx, REFPx, REFNx, REFCOM
AVSS – 0.3
AVDD + 0.3
V
Digital input voltage
CS, SCLK, DIN, DOUT/DRDY, DRDY,
START, RESET, CLK
DGND – 0.3
IOVDD + 0.3
V
Continuous, REFN0, REFOUT
–100
100
Continuous, all other pins except power-supply pins
–10
10
Input current
Temperature
(1)
Junction, TJ
mA
150
Storage, Tstg
–60
°C
150
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1)
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±2500
±1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
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7.3 Recommended Operating Conditions
over operating ambient temperature range (unless otherwise noted)
MIN
NOM
MAX
UNIT
POWER SUPPLY
Analog power supply
AVDD to AVSS
2.7
AVSS to DGND
–2.625
AVDD to DGND
1.5
Digital core power supply
DVDD to DGND
Digital IO power supply
IOVDD to DGND
5.25
0
0.05
V
5.25
2.7
3.6
V
DVDD
5.25
V
ANALOG INPUTS (1)
PGA bypassed
PGA enabled, gain = 1 to 16
Absolute input voltage (2)
V(AINx)
PGA enabled, gain = 32 to 128
VIN
Differential input voltage
VIN = VAINP – VAINN
AVSS – 0.05
AVDD + 0.05
AVSS + 0.15 +
|VINMAX|·(Gain – 1) / 2
AVDD – 0.15 –
|VINMAX|·(Gain –1) / 2
AVSS + 0.15 +
15.5·|VINMAX|
AVDD – 0.15 –
15.5·|VINMAX|
–VREF / Gain
VREF / Gain
V
0.5
AVDD – AVSS
V
V
VOLTAGE REFERENCE INPUTS (3)
VREF
Differential reference input
voltage
V(REFNx)
Absolute negative reference
voltage
Negative reference buffer disabled
AVSS – 0.05
V(REFPx) – 0.5
V
Negative reference buffer enabled
AVSS
V(REFPx) – 0.5
V
V(REFPx)
Absolute positive reference
voltage
Positive reference buffer disabled
V(REFNx) + 0.5
AVDD + 0.05
V
Positive reference buffer enabled
V(REFNx) + 0.5
AVDD
V
VREF = V(REFPx) – V(REFNx)
EXTERNAL CLOCK SOURCE (4)
fCLK
External clock frequency
Duty cycle
2
4.096
4.5
40%
50%
60%
MHz
GENERAL-PURPOSE INPUTS (GPIOs)
Input voltage
AVSS – 0.05
AVDD + 0.05
V
DGND
IOVDD
V
–40
125
°C
DIGITAL INPUTS (Other than GPIOs)
Input voltage
TEMPERATURE RANGE
TA
(1)
(2)
(3)
(4)
Operating ambient temperature
AINP and AINN denote the positive and negative inputs of the PGA. Any of the available analog inputs (AINx) can be selected as either
AINP or AINN by the input multiplexer.
VINMAX denotes the maximum differential input voltage, VIN, that is expected in the application. |VINMAX| can be smaller than VREF / Gain.
REFPx and REFNx denote one of the two available external differential reference input pairs.
An external clock is not required when the internal oscillator is used.
7.4 Thermal Information
ADS114S06B, ADS114S08B
THERMAL METRIC
(1)
VQFN (RHB)
TQFP (PBS)
32 PINS
32 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
45.2
75.5
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
28.3
17.1
°C/W
RθJB
Junction-to-board thermal resistance
15.8
28.5
°C/W
ψJT
Junction-to-top characterization parameter
0.4
0.4
°C/W
ψJB
Junction-to-board characterization parameter
15.7
28.3
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
2.3
n/a
°C/W
(1)
6
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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7.5 Electrical Characteristics
minimum and maximum specifications apply from TA = –40°C to +125°C; typical specifications are at TA = 25°C;
all specifications are at AVDD = 2.7 V to 5.25 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, all gains, internal reference, internal
oscillator, and all data rates (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ANALOG INPUTS
Absolute input current
Differential input current
PGA bypassed,
AVSS + 0.1 V ≤ V(AINx) ≤ AVDD – 0.1 V
±0.5
nA
PGA enabled, gain 1 to 128,
V(AINx)MIN ≤ V(AINx) ≤ V(AINx)MAX
–10
PGA bypassed,
VCM = AVDD / 2, –VREF ≤ VIN ≤ VREF
±0.1
10
±1
PGA enabled, gain 1 to 128,
VCM = AVDD / 2, –VREF / Gain ≤ VIN ≤ VREF / Gain
nA/V
±0.02
nA
PGA
1, 2, 4, 8, 16,
32, 64, 128
Gain settings
Startup time
Enabling the PGA in conversion mode
190
µs
SYSTEM PERFORMANCE
Resolution (no missing codes)
16
DR
Data rate
INL
Integral nonlinearity (best fit)
PGA bypassed, VCM = AVDD / 2
1
PGA enabled, gain = 1 to 128, VCM = AVDD / 2
2
PGA bypassed
Offset drift
Gain error (1)
Gain drift
(1)
NMRR
Normal-mode rejection ratio (2)
CMRR
Common-mode rejection ratio
PSRR
Power-supply rejection ratio
(1)
(2)
ppmFSR
2
PGA bypassed, after internal offset calibration
On the order of noisePP at the
set DR and gain
PGA enabled, gain = 1 to 128, after internal offset
calibration
On the order of noisePP at the
set DR and gain
PGA bypassed
10
PGA enabled, gain = 1 to 128
15
TA = 25°C, PGA bypassed
TA = 25°C, PGA enabled, gain = 1 to 128
PGA bypassed
µV
nV/°C
0.01%
0.1%
0.025%
0.2%
0.5
PGA enabled, gain = 1 to 128
Noise (input-referred)
25
20 / Gain
PGA enabled, gain = 16 to 128
Input offset voltage
SPS
20
PGA enabled, gain = 1 to 8
VIO
Bits
2.5, 5, 10, 16.6,
20, 50, 60, 100,
200, 400, 800,
1000, 2000, 4000
1
ppm/°C
See the Noise Performance section
fIN = 50 Hz or 60 Hz (±1 Hz), DR = 20 SPS
75
fIN = 50 Hz or 60 Hz (±1 Hz), DR = 20 SPS,
external fCLK = 4.096 MHz
95
95
dB
At dc
120
fCM = 50 Hz or 60 Hz (±1 Hz),
DR = 2.5 SPS, 5 SPS, 10 SPS, 20 SPS
125
AVDD at dc
105
AVDD at 50 Hz or 60 Hz
115
DVDD at dc
115
dB
dB
Excluding error of voltage reference.
See the 50-Hz and 60-Hz Line Cycle Rejection section for more information.
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Electrical Characteristics (continued)
minimum and maximum specifications apply from TA = –40°C to +125°C; typical specifications are at TA = 25°C;
all specifications are at AVDD = 2.7 V to 5.25 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, all gains, internal reference, internal
oscillator, and all data rates (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
VOLTAGE REFERENCE INPUTS
Absolute input current
Reference buffers disabled, external VREF = 2.5 V,
REFP1/REFN1 inputs
4
µA/V
Reference buffers enabled, external VREF = 2.5 V,
REFP1/REFN1 inputs
5
nA
INTERNAL VOLTAGE REFERENCE
VREF
Output voltage
Accuracy
2.5
TA = 25°C
–0.2%
Temperature drift
Output current
PSRR
0.2%
8
40
AVDD = 2.7 V to 3.3 V, sink and source
–5
5
AVDD = 3.3 V to 5.25 V, sink and source
–10
10
Short-circuit current limit
Sink and source
70
Power-supply rejection ratio
AVDD at dc
85
Load regulation
V
±0.01%
AVDD = 2.7 V to 3.3 V,
load current = –5 mA to 5 mA
8
AVDD = 3.3 V to 5.25 V,
load current = –10 mA to 10 mA
8
Startup time
1-µF capacitor on REFOUT, 0.001% settling
Capacitive load stability
Capacitor on REFOUT
Reference noise
f = 0.1 Hz to 10 Hz, 1-µF capacitor on REFOUT
100
ppm/°C
mA
mA
dB
µV/mA
5.9
1
ms
47
µF
9
µVPP
4.096
MHz
INTERNAL OSCILLATOR
fCLK
Frequency
Accuracy
–2%
2%
EXCITATION CURRENT SOURCES (IDACS)
10, 50, 100,
250, 500, 750,
1000, 1500, 2000
Current settings
Compliance voltage (3)
µA
10 µA to 750 µA, 0.1% deviation
AVSS
AVDD – 0.4
1 mA to 2 mA, 0.1% deviation
AVSS
AVDD – 0.6
–6%
±1%
V
Accuracy (each IDAC)
TA = 25°C, 10 µA to 2 mA
Current mismatch between
IDACs
6%
TA = 25°C, 10 µA to 2 mA
Temperature drift (each IDAC)
10 µA to 2 mA
100
ppm/°C
Temperature drift matching
between IDACs
10 µA to 2 mA
10
ppm/°C
Startup time
With internal reference already settled. From end of
WREG command to current flowing out of pin.
22
µs
0.2%
BIAS VOLTAGE
VBIAS
Output voltage
Output impedance
Startup time
Combined capacitive load on all selected analog
inputs CLOAD = 1 µF, 0.1% settling
(AVDD + AVSS) / 2
V
350
Ω
2.8
ms
0.2, 1, 10
µA
BURNOUT CURRENT SOURCES (BOCS)
Current settings
Accuracy
0.2 µA, sinking or sourcing
±8%
1 µA, sinking or sourcing
±4%
10 µA, sinking or sourcing
±2%
(AVDD – AVSS) / 4 monitor
±1%
DVDD / 4 monitor
±1%
SUPPLY VOLTAGE MONITORS
Accuracy
(3)
8
The IDAC current does not change by more than 0.1% from the nominal value when staying within the specified compliance voltage.
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Electrical Characteristics (continued)
minimum and maximum specifications apply from TA = –40°C to +125°C; typical specifications are at TA = 25°C;
all specifications are at AVDD = 2.7 V to 5.25 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, all gains, internal reference, internal
oscillator, and all data rates (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEMPERATURE SENSOR
Output voltage
TA = 25°C
Temperature coefficient
129
mV
403
µV/°C
GENERAL-PURPOSE INPUT/OUTPUTS (GPIOs)
VIL
Logic input level, low
VIH
Logic input level, high
VOL
Logic output level, low
VOH
Logic output level, high
AVSS – 0.05
0.3 AVDD
V
0.7 AVDD
AVDD + 0.05
V
IOL = 1 mA
AVSS
0.2 AVDD
V
IOH = 1 mA
0.8 AVDD
AVDD
V
DIGITAL INPUT/OUTPUTS
VIL
Logic input level, low
VIH
Logic input level, high
DGND
0.3 IOVDD
V
0.7 IOVDD
IOVDD
VOL
Logic output level, low
V
IOL = 1 mA
DGND
0.2 IOVDD
VOH
V
Logic output level, high
IOH = 1 mA
0.8 IOVDD
IOVDD
V
Input current
DGND ≤ VDigital Input ≤ IOVDD
–1
1
µA
ANALOG SUPPLY CURRENT (AVDD = 3.3 V, External Reference, Internal Reference Disabled, Reference Buffers Disabled, IDACs Disabled, VBIAS
Disabled, Internal Oscillator, All Data Rates, VIN = 0 V)
IAVDD
Analog supply current
Power-down mode
0.1
Standby mode, PGA bypassed
70
Conversion mode, PGA bypassed
85
Conversion mode, PGA enabled, gain = 1, 2
120
Conversion mode, PGA enabled, gain = 4, 8
140
Conversion mode, PGA enabled, gain = 16, 32
165
Conversion mode, PGA enabled, gain = 64
200
Conversion mode, PGA enabled, gain = 128
250
µA
ADDITIONAL ANALOG SUPPLY CURRENTS PER FUNCTION (AVDD = 3.3 V)
Internal 2.5-V reference, no external load
IAVDD
Analog supply current
185
Positive reference buffer
35
Negative reference buffer
25
VBIAS buffer, no external load
10
IDAC overhead, 10 µA to 250 µA
20
IDAC overhead, 500 µA to 750 µA
30
IDAC overhead, 1 mA
40
IDAC overhead, 1.5 mA
50
IDAC overhead, 2 mA
65
µA
DIGITAL SUPPLY CURRENT (DVDD = IOVDD = 3.3 V, All Data Rates, SPI Not Active)
IDVDD +
IIOVDD
Digital supply current
Power-down mode, internal oscillator
0.1
Standby mode, internal oscillator
185
Conversion mode, internal oscillator
225
Conversion mode, external fCLK = 4.096 MHz
195
µA
POWER DISSIPATION (AVDD = DVDD = IOVDD = 3.3 V, Internal Reference Enabled, Reference Buffers Disabled, IDACs Disabled, VBIAS Disabled,
Internal Oscillator, All Data Rates, VIN = 0 V, SPI Not Active)
PD
Power dissipation
Conversion mode, PGA enabled, gain = 1
1.75
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7.6 Timing Characteristics
over operating ambient temperature range, DVDD = 2.7 V to 3.6 V, IOVDD = DVDD to 5.25 V, and
DOUT/DRDY load = 20 pF || 100 kΩ to DGND (unless otherwise noted)
MIN
MAX
UNIT (1)
SERIAL INTERFACE
td(CSSC)
Delay time, first SCLK rising edge after CS falling edge
20
ns
td(SCCS)
Delay time, CS rising edge after final SCLK falling edge
20
ns
tw(CSH)
Pulse duration, CS high
30
ns
tc(SC)
SCLK period
100
ns
tw(SCH)
Pulse duration, SCLK high
40
ns
tw(SCL)
Pulse duration, SCLK low
40
ns
tsu(DI)
Setup time, DIN valid before SCLK falling edge
15
ns
th(DI)
Hold time, DIN valid after SCLK falling edge
20
ns
td(CMD)
Delay time, between bytes or commands
0
ns
tw(RSL)
Pulse duration, RESET low
4
tCLK
td(RSSC)
Delay time, first SCLK rising edge after RESET rising edge (or 7th SCLK
falling edge of RESET command)
4096
tCLK
RESET PIN
START/SYNC PIN
tw(STH)
Pulse duration, START/SYNC high
4
tCLK
tw(STL)
Pulse duration, START/SYNC low
4
tCLK
tsu(STDR)
Setup time, START/SYNC falling edge (or 7th SCLK falling edge of STOP
command) before DRDY falling edge to stop further conversions
(continuous conversion mode)
32
tCLK
(1)
tCLK = 1 / fCLK.
7.7 Switching Characteristics
over operating ambient temperature range, DVDD = 2.7 V to 3.6 V, IOVDD = DVDD to 5.25 V, and
DOUT/DRDY load = 20 pF || 100 kΩ to DGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT (1)
tp(CSDO)
Propagation delay time, CS falling edge to DOUT
driven
0
25
ns
tp(SCDO)
Propagation delay time, SCLK rising edge to valid
new DOUT
3
30
ns
tp(CSDOZ)
Propagation delay time, CS rising edge to DOUT high
impedance
0
25
ns
tp(STDR)
Propagation delay time, START/SYNC rising edge (or
first SCLK rising edge of any command or data read)
to DRDY rising edge
2
tCLK
tw(DRH)
Pulse duration, DRDY high
tp(GPIO)
Propagation delay time, last SCLK falling edge of
WREG command to GPIOx output valid
SPI timeout per 8 bits
(1)
(2)
10
24
(2)
3
15
2
tCLK
100
ns
tCLK
tCLK = 1 / fCLK
The SPI interface resets when an entire byte is not sent within the specified timeout time.
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tw(CSH)
CS
ttd(CSSC)t
ttc(SC)t
tw(SCH)
ttd(SCCS)t
SCLK
tsu(DI)
th(DI)
tw(SCL)
DIN
NOTE: Single-byte communication is shown. Actual communication can be multiple bytes.
Figure 1. Serial Interface Timing Requirements
CS
SCLK
tp(CSDO)
DOUT/DRDY
tp(SCDO)
tp(CSDOZ)
Hi-Z
Hi-Z
NOTE: Single-byte communication is shown. Actual communication can be multiple bytes.
Figure 2. Serial Interface Switching Characteristics
tw(RSL)
RESET
td(RSSC)
SCLK
DIN
RESET command
New command
Figure 3. RESET Pin and RESET Command Timing Requirements
tw(STL)
START/SYNC
DRDY
tw(STH)
tp(STDR)
tp(STDR)
tw(DRH)
tsu(STDR)
Figure 4. START/SYNC Pin Timing Requirements
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SCLK
DIN
DRDY
START command
STOP command
tp(STDR)
tp(STDR)
tsu(STDR)
Figure 5. START Command Timing Requirements
SCLK
DIN
WREG
WREG GPIODAT
01h
01h
01h
Write two registers
GPIO0 set as output
GPIO0 set high
GPIO0 enabled
GPIO0
tp(GPIO)
Figure 6. GPIO Switching Characteristics
12
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7.8 Typical Characteristics
at TA = 25°C, AVDD = 3.3 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, using internal VREF = 2.5 V, internal 4.096-MHz oscillator,
and PGA enabled (unless otherwise noted)
2000
2000
25qC
85qC
125qC
-40qC
1000
Absolute Input Current (pA)
Absolute Input Current (pA)
-40qC
0
-1000
-2000
-3000
-4000
125qC
0
-1000
-2000
-3000
0.5
1
1.5
2
V(AINx) (V)
2.5
3
3.5
0
PGA bypassed, DR = 20 SPS, VIN = 0 V
0.5
1
1.5
2
V(AINx) (V)
2.5
3
3.5
PGA bypassed, DR = 4 kSPS, VIN = 0 V
Figure 7. Absolute Input Current vs Absolute Input Voltage
Figure 8. Absolute Input Current vs Absolute Input Voltage
2000
2000
-40qC
25qC
85qC
125qC
-40qC
25qC
85qC
125qC
1500
Absolute Input Current (pA)
1500
Absolute Input Current (pA)
85qC
-4000
0
1000
500
0
-500
-1000
-1500
1000
500
0
-500
-1000
-1500
-2000
-2000
0
0.5
1
1.5
2
V(AINx) (V)
2.5
3
3.5
0
PGA enabled, gain = 1, DR = 20 SPS, VIN = 0 V
0.5
1
1.5
2
V(AINx) (V)
2.5
3
3.5
PGA enabled, gain = 1, DR = 4 kSPS, VIN = 0 V
Figure 9. Absolute Input Current vs Absolute Input Voltage
Figure 10. Absolute Input Current vs Absolute Input Voltage
2000
2000
-40qC
25qC
85qC
125qC
-40qC
25qC
85qC
125qC
1500
Differential Input Current (pA)
1500
Differential Input Current (pA)
25qC
1000
1000
500
0
-500
-1000
-1500
1000
500
0
-500
-1000
-1500
-2000
-2.5
-2
-1.5
-1
-0.5
0
0.5
VIN (V)
1
1.5
2
2.5
PGA bypassed, DR = 20 SPS, VCM = 1.65 V
Figure 11. Differential Input Current vs Differential Input
Voltage
-2000
-2.5
-2
-1.5
-1
-0.5
0
0.5
VIN (V)
1
1.5
2
2.5
PGA bypassed, DR = 4 kSPS, VCM = 1.65 V
Figure 12. Differential Input Current vs Differential Input
Voltage
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Typical Characteristics (continued)
at TA = 25°C, AVDD = 3.3 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, using internal VREF = 2.5 V, internal 4.096-MHz oscillator,
and PGA enabled (unless otherwise noted)
200
400
-40qC
25qC
85qC
125qC
-40qC
100
50
0
-50
-100
-150
-200
-2.5
-2
-1.5
-1
-0.5
0
0.5
VIN (V)
1
1.5
2
100
0
-100
-200
-400
-2.5
2.5
-2
-1.5
-1
-0.5
0
0.5
VIN (V)
1
1.5
2
2.5
PGA enabled, DR = 4 kSPS, VCM = 1.65 V
Figure 14. Differential Input Current vs Differential Input
Voltage
3
3
2
2
INL (ppm of FSR)
INL (ppm of FSR)
125qC
-300
Figure 13. Differential Input Current vs Differential Input
Voltage
1
0
-1
1
0
-1
-2
-2
-3
-100 -80
-60
-40
-20
0
20
VIN (% of FSR)
40
60
80
-3
-100
100
-80
-60
-40
-20
0
20
VIN (% of FSR)
40
60
80
100
PGA enabled, gain = 1
PGA bypassed, gain = 1
Figure 16. INL vs Differential Input Voltage
Figure 15. INL vs Differential Input Voltage
16
10
Gain = 1
Gain = 2
Gain = 4
Gain = 8
Gain = 16
Gain = 32
Gain = 64
Gain = 128
14
Gain = 1
Gain = 2
Gain = 4
Gain = 8
Gain = 16
Gain = 32
Gain = 64
Gain = 128
12
Offset Voltage (PV)
INL (ppm of FSR)
85qC
200
PGA enabled, DR = 20 SPS, VCM = 1.65 V
8
25qC
300
Differential Input Current (pA)
Differential Input Current (pA)
150
6
4
10
8
6
4
2
2
0
-50
-25
0
25
50
Temperature (qC)
75
Figure 17. INL vs Temperature
14
100
125
0
-50
-25
0
25
50
Temperature (qC)
75
100
125
Figure 18. Offset Voltage vs Temperature
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Typical Characteristics (continued)
at TA = 25°C, AVDD = 3.3 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, using internal VREF = 2.5 V, internal 4.096-MHz oscillator,
and PGA enabled (unless otherwise noted)
2.502
Internal Reference Voltage (V)
0.03
Gain Error (%)
0.025
0.02
0.015
0.01
0.005
PGA disabled
Gain = 1
Gain = 4
Gain = 32
2.501
2.5
2.499
2.498
Gain = 128
2.497
0
-50
-25
0
25
50
Temperature (qC)
75
100
-50
125
-25
0
25
50
Temperature (qC)
75
100
125
28 units, TQFP package
Figure 20. Internal Reference Voltage vs Temperature
Figure 19. Gain Error vs Temperature
2.5002
10
Internal Reference Noise (PV)
Internal Reference Voltage (V)
8
2.5001
2.5
2.4999
2.4998
6
4
2
0
-2
-4
-6
-8
2.4997
2.7
3
3.3
3.6
3.9
4.2
AVDD (V)
4.5
4.8
5.1
-10
5.4
Time (1 s/div)
Figure 22. Internal Reference Voltage Noise
Figure 21. Internal Reference Voltage vs AVDD
4.12
Internal Oscillator Frequency (MHz)
Number of Occurrences
250
200
150
100
4.106
4.104
4.102
4.1
4.098
4.096
4.094
4.092
0
4.09
50
4.11
4.1
4.09
4.08
4.07
4.06
-50
-25
25
50
Temperature (C)
75
100
125
28 units
Internal Oscillator Frequency (MHz)
Figure 23. Internal Oscillator Frequency Histogram
0
Figure 24. Internal Oscillator Frequency vs Temperature
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Typical Characteristics (continued)
0
0
-10
-1
IDAC Error (%)
IDAC Error (%)
at TA = 25°C, AVDD = 3.3 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, using internal VREF = 2.5 V, internal 4.096-MHz oscillator,
and PGA enabled (unless otherwise noted)
-20
-30
10 µA
50 µA
100 µA
250 µA
500 µA
-40
750 µA
1 mA
1.5 mA
2 mA
0.5
1
-3
10 µA
50 µA
100 µA
250 µA
500 µA
-4
-5
2.5
-50
0
-2
1.5
2
2.5
IDAC Output Voltage (V)
3
3.5
2.7
2.8
2.9
3
IDAC Output Voltage (V)
3.1
3.2
Figure 26. IDAC Accuracy vs Compliance Voltage
Figure 25. IDAC Accuracy vs Compliance Voltage
3
0.3
2
10 PA
50 PA
100 PA
250 PA
500 PA
0.25
IDAC Mismatch Error (%)
IDAC Error (%)
2.6
750 µA
1 mA
1.5 mA
2 mA
1
0
-1
10 PA
50 PA
100 PA
250 PA
500 PA
-2
-3
-50
-25
0
25
50
Temperature (qC)
750 PA
1 mA
1.5 mA
2 mA
75
100
0.2
750 PA
1 mA
1.5 mA
2 mA
0.15
0.1
0.05
0
-50
125
-25
0
25
50
Temperature (qC)
75
100
125
IDAC output voltage = 1.65 V
Figure 27. IDAC Accuracy vs Temperature
Figure 28. IDAC Matching vs Temperature
180
AVDD = 2.7 V
AVDD = 5.25 V
(AVDD - AVSS) / 2 Ratio
0.5005
0.5
0.4995
0.499
0.4985
0.498
-50
-25
0
25
50
Temperature (qC)
75
100
Figure 29. VBIAS Voltage [(AVDD – AVSS) / 2] vs
Temperature
16
125
Temperature Sensor Voltage (mV)
0.501
160
140
120
100
80
-50
-25
0
25
50
Temperature (qC)
75
100
125
Figure 30. Temperature Sensor Voltage vs Temperature
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Typical Characteristics (continued)
at TA = 25°C, AVDD = 3.3 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, using internal VREF = 2.5 V, internal 4.096-MHz oscillator,
and PGA enabled (unless otherwise noted)
0.6
3.3
3.2
GPIO Pin Output Voltage (V)
GPIO Pin Output Voltage (V)
-40qC
3.1
3
2.9
2.8
-40qC
25qC
85qC
25qC
0.5
0.4
0.3
0.2
0.1
0
0
1
2
3
4
5
Sourcing Current (mA)
6
7
0
8
1
2
AVDD = 3.3 V
0.6
3.2
0.5
Digital Pin Output Voltage (V)
-40qC
3.1
3
2.9
2.8
25qC
6
7
8
Figure 32. GPIO Pin Output Voltage vs Sinking Current
3.3
-40qC
3
4
5
Sinking Current (mA)
AVDD = 3.3 V
Figure 31. GPIO Pin Output Voltage vs Sourcing Current
Digital Pin Output Voltage (V)
125qC
125qC
2.7
85qC
25qC
85qC
125qC
0.4
0.3
0.2
0.1
125qC
0
2.7
0
1
2
3
4
5
Sourcing Current (mA)
6
7
0
8
1
2
3
4
5
Sinking Current (mA)
6
7
8
DVDD = 3.3 V
DVDD = 3.3 V
Figure 33. Digital Pin Output Voltage vs Sourcing Current
Figure 34. Digital Pin Output Voltage vs Sinking Current
400
400
Standby mode
PGA bypassed
Gain = 1
350
Gain = 4
Gain = 16
Gain = 64
Gain = 128
350
300
AVDD Current (PA)
AVDD Current (PA)
85qC
250
200
150
200
150
50
50
0
25
50
Temperature (qC)
75
100
125
Standby and conversion mode, external VREF
Figure 35. Analog Supply Current vs Temperature
Gain = 64
Gain = 128
250
100
-25
Gain = 4
Gain = 16
300
100
0
-50
PGA bypassed
Gain = 1
0
2.7
3
3.3
3.6
3.9
4.2
AVDD (V)
4.5
4.8
5.1
5.4
Conversion mode, external VREF
Figure 36. Analog Supply Current vs AVDD
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Typical Characteristics (continued)
at TA = 25°C, AVDD = 3.3 V, AVSS = 0 V, DVDD = IOVDD = 3.3 V, using internal VREF = 2.5 V, internal 4.096-MHz oscillator,
and PGA enabled (unless otherwise noted)
1
Internal Reference AVDD Current (PA)
260
AVDD Current (PA)
0.8
0.6
0.4
0.2
0
-50
-25
0
25
50
Temperature (qC)
75
100
240
220
200
180
160
140
120
-50
125
-25
0
25
50
Temperature (qC)
75
100
125
Power-down mode
Figure 38. Internal Reference AVDD Current vs Temperature
260
240
240
220
220
DVDD Current (PA)
DVDD Current (PA)
Figure 37. Analog Supply Current vs Temperature
260
200
180
160
140
-25
0
25
50
Temperature (qC)
75
100
180
160
140
Standby mode
Conversion mode
120
-50
200
125
120
2.7
2.8
2.9
Standby and conversion mode
3
3.1
3.2
DVDD (V)
3.3
3.4
3.5
3.6
Conversion mode
Figure 39. Digital Supply Current vs Temperature
Figure 40. Digital Supply Current vs DVDD
5
DVDD Current (PA)
4
3
2
1
0
-50
-25
0
25
50
Temperature (qC)
75
100
125
Power-down mode
Figure 41. Digital Supply Current vs Temperature
18
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8 Parameter Measurement Information
8.1 Noise Performance
Delta-sigma (ΔΣ) analog-to-digital converters (ADCs) are based on the principle of oversampling. The input
signal of a ΔΣ ADC is sampled at a high frequency (modulator frequency) and subsequently filtered and
decimated in the digital domain to yield a conversion result at the respective output data rate. The ratio between
modulator frequency and output data rate is called the oversampling ratio (OSR). By increasing the OSR, and
thus reducing the output data rate, the noise performance of the ADC can be optimized. In other words, the
input-referred noise drops when reducing the output data rate because more samples of the internal modulator
are averaged to yield one conversion result. Increasing the gain also reduces the input-referred noise, which is
particularly useful when measuring low-level signals.
Table 1 and Table 2 summarize the device noise performance. Data are representative of typical noise
performance at TA = 25°C using the internal 2.5-V reference. Data shown are based on 512 consecutive samples
from a single device with inputs internally shorted. Table 1 lists the input-referred root mean square noise in units
of μVRMS for the conditions shown. Peak-to-peak (µVPP) values are shown in parentheses. Table 2 lists the
corresponding data in effective resolution calculated from μVRMS values using Equation 1. Noise-free resolution is
calculated from µVPP values using Equation 2.
The input-referred noise (Table 1) only changes marginally when using an external low-noise reference, such as
the REF5025. To calculate effective resolution and noise-free resolution when using a reference voltage other
than 2.5 V, use Equation 1 and Equation 2:
Effective Resolution = ln[(2 · VREF / Gain) / VRMS-Noise] / ln(2)
Noise-Free Resolution= ln[(2 · VREF / Gain) / VPP-Noise] / ln(2)
(1)
(2)
Noise performance with the PGA bypassed are identical to the noise performance of the device with gain = 1.
Table 1. Noise in μVRMS (μVPP) at AVDD = 3.3 V, AVSS = 0 V,
PGA Enabled, and Internal 2.5-V Reference
DATA
RATE
(SPS)
GAIN
1
2
4
8
16
32
64
128
2.5
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.60)
5
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.60)
10
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.60)
16.6
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.60)
20
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.60)
50
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.60)
60
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.2)
0.60 (0.90)
100
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.4)
0.60 (1.3)
200
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.4)
1.2 (1.9)
0.60 (1.7)
400
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (2.8)
1.2 (2.9)
0.60 (2.3)
800
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (4.0)
1.2 (3.8)
0.60 (3.2)
1000
76.3 (76.3)
38.1 (38.1)
19.1 (19.1)
9.5 (9.5)
4.8 (4.8)
2.4 (5.1)
1.2 (4.3)
0.60 (3.8)
2000
76.3 (83)
38.1 (80)
19.1 (32)
9.5 (17)
4.8 (11)
2.4 (6.7)
1.2 (6.6)
1.0 (6.5)
4000
103 (629)
38.1 (404)
24 (160)
12 (70)
6.4 (39)
3.3 (21)
3.1 (21)
2.6 (20)
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Table 2. Effective Resolution From RMS Noise (Noise-Free Resolution From Peak-to-Peak Noise)
at AVDD = 3.3 V, AVSS = 0 V, PGA Enabled, and Internal 2.5-V Reference
20
DATA
RATE
(SPS)
GAIN
1
2
4
8
16
32
64
128
2.5
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
5
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
10
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16.6
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
20
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
50
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.5)
60
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.4)
100
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.8)
16 (14.9)
200
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.3)
16 (14.5)
400
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.8)
16 (14.7)
16 (14.0)
800
16 (16)
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.2)
16 (14.3)
16 (13.6)
1000
16 (16)
16 (16)
16 (16)
16 (16)
16 (15.8)
16 (14.9)
16 (14.2)
16 (13.3)
2000
16 (15.9)
16 (14.9)
16 (15.3)
16 (15.2)
16 (14.8)
16 (14.5)
16 (13.5)
15.2 (12.6)
4000
16 (13.0)
16 (12.6)
15.7 (12.9)
16 (13.1)
15.6 (13.0)
15.5 (12.9)
14.4 (11.9)
13.6 (10.9)
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9 Detailed Description
9.1 Overview
The ADS114S06B and ADS114S08B are precision 16-bit, delta-sigma (ΔΣ) ADCs with an integrated analog front
end (AFE) to simplify precision sensor connections. The ADC provides output data rates from 2.5 SPS to
4000 SPS for flexibility in resolution and data rates over a wide range of applications. The low-noise and low-drift
architecture make these devices suitable for precise measurement of low-voltage sensors, such as load cells and
temperature sensors.
The ADS114S0xB incorporate several features that simplify precision sensor measurements. Key integrated
features include:
• Low-noise, CMOS PGA
• Low-drift, 2.5-V voltage reference
• Two sets of buffered external reference inputs
• Dual, matched, sensor-excitation current sources (IDACs)
• Internal 4.096-MHz oscillator
• Temperature sensor
• Four general-purpose input/output pins (GPIOs)
As described in the Functional Block Diagram section, these devices provide 12 (ADS114S08B) or six
(ADS114S06B) analog inputs that are configurable as either single-ended inputs, differential inputs, or any
combination of the two. Many of the analog inputs have additional features as programmed by the user. The
analog inputs can be programmed to enable the following extended features:
• Two sensor excitation current sources: all analog input pins (and REFP1 and REFN1 on the ADS114S06B)
• Sensor biasing voltage (VBIAS): pins AIN0, AIN1, AIN2, AIN3, AIN4, AIN5, AINCOM
• Four GPIO pins: AIN8, AIN9, AIN10, AIN11 (ADS114S08B only, the ADS114S06B has dedicated GPIOs)
• Sensor burn-out current sources: analog input pins selected for ADC input
Following the input multiplexer (MUX), the ADC features a high input-impedance, low-noise, programmable gain
amplifier (PGA), eliminating the need for an external amplifier. The PGA gain is programmable from 1 to 128 in
binary steps. The PGA can be bypassed to allow the input range to extend 50 mV below ground or above
supply.
An inherently stable delta-sigma modulator measures the ratio of the input voltage to the reference voltage to
provide the ADC result. The ADC operates with the internal 2.5-V reference, or with up to two external reference
inputs. The REFOUT pin provides the buffered 2.5-V internal voltage reference output that can be used to bias
external circuitry.
The digital filter provides settled data with 50-Hz and 60-Hz line-cycle rejection at data rates of 2.5 SPS, 5 SPS,
10 SPS, and 20 SPS, 50-Hz rejection at data rates of 16.6 SPS and 50 SPS, and 60-Hz rejection at a data rate
of 60 SPS.
Two programmable excitation current sources provide bias to resistive sensors [such as resistance temperature
detectors (RTDs) or thermistors]. The ADC integrates several system monitors for read back, such as
temperature sensor and supply monitors. Four GPIO pins are available as either dedicated pins (ADS114S06B)
or combined with analog input pins (ADS114S08B).
The ADS114S0xB system clock is either provided by the internal low-drift, 4.096-MHz oscillator or an external
clock source on the CLK input.
The SPI-compatible serial interface is used to read the conversion data and also to configure and control the
ADC. The serial interface consists of four signals: CS, SCLK, DIN, and DOUT/DRDY. The dual function
DOUT/DRDY output indicates when conversion data are ready and also provides the data output. The serial
interface can be implemented with as little as three connections by tying CS low. Start ADC conversions with
either the START/SYNC pin or with commands. The ADC can be programmed for a continuous conversion mode
or to perform single-shot conversions.
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Overview (continued)
The AVDD analog supply operates with bipolar supplies from ±1.5 V to ±2.625 V or with a unipolar supply from
2.7 V to 5.25 V. For unipolar-supply operation, use the VBIAS voltage to bias isolated (floating) sensors. The
digital supplies operate with unipolar supplies only. The DVDD digital power supply operates from 2.7 V to 3.6 V
and the IOVDD supply operates from DVDD to 5.25 V.
9.2 Functional Block Diagram
AVDD
REFN0
Burnout
Detect
Excitation
Current
Sources
AINCOM
AIN0
REFP0
Reference
Mux
REFCOM REFOUT
DVDD
2.5-V
Reference
ADS114S06B
ADS114S08B
IOVDD
AIN1
AIN2
Reference
Buffers
VBIAS
AIN3
AIN4
AIN5
Input
Mux
PGA
AIN6 / REFP1
16-Bit û
ADC
START/SYNC
Configurable
Digital
Filter
Serial
Interface
and
Control
AIN7 / REFN1
SCLK
DIN
AIN8 / GPIO0
DOUT/DRDY
AIN9 / GPIO1
Power Supplies
AIN10 / GPIO2
System-, SelfCalibration
Temperature
Sensor
AIN11 / GPIO3
ADS114S08B
Only
RESET
CS
DRDY
4.096-MHz
Oscillator
Burnout
Detect
AVSS
CLK
DGND
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9.3 Feature Description
9.3.1 Multiplexer
The ADS114S0xB contains a flexible input multiplexer; see Figure 42. Select any of the six (ADS114S06B) or 12
(ADS114S08B) analog inputs as the positive or negative input for the PGA using the MUX_P[3:0] and
MUX_N[3:0] bits in the input multiplexer register (02h). In addition, AINCOM can be selected as the positive or
negative PGA input. AINCOM is treated as a regular analog input, as is AINx. Use AINCOM in single-ended
measurement applications as the common input for the other analog inputs.
The multiplexer also routes the excitation current sources to drive resistive sensors (bridges, RTDs, and
thermistors) and can provide bias voltages for unbiased sensors (unbiased thermocouples for example) to analog
input pins.
The ADS114S0xB also contain a set of system monitor functions measured through the multiplexer. The inputs
can be shorted together at mid-supply [(AVDD + AVSS) / 2] to measure and calibrate the input offset of the
analog front-end and the ADC. The system monitor also includes a temperature sensor that provides a
measurement of the device temperature. The system monitor can also measure the analog and digital supplies,
measuring [(AVDD – AVSS) / 4] for the analog supply or DVDD / 4 for the digital supply. Finally, the system
monitor contains a set of burn-out current sources that pull the inputs to either supply if the sensor has burned
out and has a high impedance so that the ADC measures a full-scale reading.
The multiplexer implements a break-before-make circuit. When changing the multiplexer channels using the
MUX_P[3:0] and MUX_N[3:0] bits, the device first disconnects the PGA inputs from the analog inputs and
connects them to mid-supply for 2 · tCLK. In the next step, the PGA inputs connect to the selected new analog
input channels. This break-before-make behavior ensures the ADC always starts from a known state and that the
analog inputs are not momentarily shorted together.
Electrostatic discharge (ESD) diodes to AVDD and AVSS protect the inputs. To prevent the ESD diodes from
turning on, the absolute voltage on any input must stay within the range provided by Equation 3:
AVSS – 0.3 V < V(AINx) < AVDD + 0.3 V
(3)
External Schottky clamp diodes or series resistors may be required to limit the input current to safe values (see
the Absolute Maximum Ratings table). Overdriving an unselected input on the device can affect conversions
taking place on other input pins.
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Feature Description (continued)
AVDD
AVDD
IDAC2
IDAC1
(1)
AVSS
AVDD
VBIAS
(AVDD + AVSS) / 2
AINCOM
VBIAS
AVSS
AVDD
AVSS
AVDD
AVSS
AVDD
AVSS
AVDD
AVSS
AVDD
AIN0
VBIAS
AIN1
VBIAS
AVDD
AVDD
AIN2
VBIAS
AIN3
VBIAS
Temperature
Diode
AIN4
(2)
VBIAS
AVSS
AVDD
(AVDD í AVSS) ‡ (5 / 8)
(AVDD í AVSS) ‡ (3 / 8)
AIN5
(3)
AVSS
AVDD
'9'' ‡ (4 / 12)
'9'' ‡ (1 / 12)
AIN6
AVSS
AVDD
AVDD
AIN7
AVSS
AVDD
AVSS
AVDD
Burn-Out Current Source
AIN8
AINP
PGA
To ADC
AINN
AIN9
AVSS
AVDD
Burn-Out Current Source
AIN10
AVSS
AVSS
AVDD
AIN11
ADS114S08B Only
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(1)
AINP and AINN are connected together to (AVDD + AVSS) / 2 for offset measurement.
(2)
Measurement for the analog supply equivalent to (AVDD – AVSS) / 4.
(3)
Measurement for the analog supply equivalent to DVDD / 4.
Figure 42. Analog Input Multiplexer
24
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Feature Description (continued)
9.3.2 Low-Noise Programmable Gain Amplifier
The ADS114S06B and ADS114S08B feature a low-drift, low-noise, high input impedance programmable gain
amplifier (PGA). Figure 43 shows a simplified diagram of the PGA. The PGA consists of two chopper-stabilized
amplifiers (A1 and A2) and a resistor feedback network that sets the gain of the PGA. The PGA input is equipped
with an electromagnetic interference (EMI) filter and an antialiasing filter on the output.
250
+
AINP
A1
16 pF
RF
320 pF
RG
RF
250
2.5 k
ADC
2.5 k
A2
+
AINN
16 pF
Figure 43. Simplified PGA Diagram
The PGA can be set to gains of 1, 2, 4, 8, 16, 32, 64, or 128 using the GAIN[2:0] bits in the gain setting register
(03h). Gain is changed inside the device using a variable resistor, RG. The differential full-scale input voltage
range (FSR) of the PGA is defined by the gain setting and the reference voltage used, as shown in Equation 4:
FSR = ±VREF / Gain
(4)
Table 3 shows the corresponding full-scale ranges when using the internal 2.5-V reference.
Table 3. PGA Full-Scale Range
GAIN SETTING
FSR
1
±2.5 V
2
±1.25 V
4
±0.625 V
8
±0.313 V
16
±0.156 V
32
±0.078 V
64
±0.039 V
128
±0.020 V
The PGA must be enabled with the PGA_EN[1:0] bits of the gain setting register (03h). Setting these bits to 00
powers down and bypasses the PGA. A setting of 01 enables the PGA. The 10 and 11 settings are reserved and
must not be written to the device.
With the PGA enabled, gains 64 and 128 are established in the digital domain. When the device is set to 64 or
128, the PGA is set to a gain of 32, and additional gain is established with digital scaling. The input-referred
noise does still improve compared to the gain = 32 setting because the PGA is biased with a higher supply
current to reduce noise.
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9.3.2.1 PGA Input-Voltage Requirements
As with many amplifiers, the PGA has an absolute input voltage range requirement that cannot be exceeded.
The maximum and minimum absolute input voltages are limited by the voltage swing capability of the PGA
output. The specified minimum and maximum absolute input voltages (VAINP and VAINN) depend on the PGA gain,
the maximum differential input voltage (VINMAX), and the tolerance of the analog power-supply voltages (AVDD
and AVSS). Use the maximum voltage expected in the application for VINMAX. The absolute positive and negative
input voltages must be within the specified range, as shown in Equation 5:
AVSS + 0.15 V + |VINMAX| · (Gain – 1) / 2 < VAINP, VAINN < AVDD – 0.15 V – |VINMAX| · (Gain – 1) / 2
where
•
•
VAINP, VAINN = absolute input voltage
VINMAX = VAINP – VAINN = maximum differential input voltage
(5)
As mentioned in the previous section, PGA gain settings of 64 and 128 are scaled in the digital domain and are
not implemented with the amplifier. When using the PGA in gains of 64 and 128, set the gain in Equation 5 to 32
to calculate the absolute input voltage range.
The relationship between the PGA input to the PGA output is shown graphically in Figure 44. The PGA output
voltages (VOUTP, VOUTN) depend on the PGA gain and the input voltage magnitudes. For linear operation, the
PGA output voltages must not exceed AVDD – 0.15 V or AVSS + 0.15 V. Note that the diagram depicts a
positive differential input voltage that results in a positive differential output voltage.
PGA Input
PGA Output
AVDD
AVDD ± 0.15 V
VOUTP = VAINP + VIN Â (Gain ± 1) / 2
VAINP
VIN = VAINP ± VAINN
VAINN
VOUTN = VAINN ± VIN Â (Gain ± 1) / 2
AVSS + 0.15 V
AVSS
Figure 44. PGA Input/Output Range
Download the ADS1x4S0x design calculator from www.ti.com. This calculator can be used to determine the input
voltage range of the PGA.
9.3.2.2 Bypassing the PGA
At a gain of 1, the device can be configured to disable and bypass the low-noise PGA. Disabling the PGA lowers
the overall power consumption and also removes the restrictions of Equation 5 for the input voltage range. If the
PGA is bypassed, the ADC absolute input voltage range extends beyond the AVDD and AVSS power supplies,
allowing input voltages at or below ground. The absolute input voltage range when the PGA is bypassed is
shown in Equation 6:
AVSS – 0.05 V < VAINP, VAINN < AVDD + 0.05 V
(6)
In order to measure single-ended signals that are referenced to AVSS (AINP = VIN, AINN = AVSS), the PGA must
be bypassed. The PGA is bypassed and powered down by setting the PGA_EN[1:0] bits to 00 in the gain setting
register (03h).
For signal sources with high output impedance, external buffering may still be necessary. Note that active buffers
introduce noise and also introduce offset and gain errors. Consider all of these factors in high-accuracy
applications.
26
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9.3.3 Voltage Reference
The devices require a reference voltage for operation. The ADS114S0xB offers an integrated low-drift 2.5-V
reference. For applications that require a different reference voltage value or a ratiometric measurement
approach, the ADS114S08B offers two differential reference input pairs (REFP0, REFN0 and REFP1, REFN1).
The differential reference inputs allow freedom in the reference common-mode voltage. REFP0 and REFN0 are
dedicated reference inputs, whereas REFP1 and REFN1 are shared with inputs AIN6 and AIN7 (respectively) on
the ADS114S08B. The specified external reference voltage range is 0.5 V to AVDD. The reference voltage is
shown in Equation 7, where V(REFPx) and V(REFNx) are the absolute positive and absolute negative reference
voltages.
VREF = V(REFPx) – V(REFNx)
(7)
The polarity of the reference voltage internal to the ADC must be positive. The magnitude of the reference
voltage together with the PGA gain establishes the ADC full-scale differential input range as defined by
FSR = ±VREF / Gain.
Figure 45 shows the block diagram of the reference multiplexer. The ADC reference multiplexer selects between
the internal reference and two external references (REF0 and REF1). The reference multiplexer is programmed
with the REFSEL[1:0] bits in the reference control register (05h). By default, the external reference pair REFP0,
REFN0 is selected.
REFSEL[1:0] bits of REF register
00 = REFP0, REFN0
01 = REFP1, REFN1
10 = Internal 2.5-V reference
11 = Reserved
REFP_BUF bit of REF register
0 = Enabled
1 = Disabled
00
REFP0
01
REFP1
10
REFOUT
VREFP
Internal
2.5-V
Reference
VREFN
1 PF (1)
00
REFN0
01
REFN1
10
REFCOM
REFCON[1:0] bits of REF register
00 = Internal reference off
01 = Internal reference on;
off in power-down mode
10 = Internal reference always on
11 = Reserved
(1)
ADC
REFN_BUF bit of REF register
0 = Enabled
1 = Disabled
The internal reference requires a minimum 1-µF capacitor connected from REFOUT to REFCOM.
Figure 45. Reference Multiplexer Block Diagram
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9.3.3.1 Internal Reference
The ADC integrates a precision, low-drift, 2.5-V reference. The internal reference is enabled by setting
REFCON[1:0] to 10 (reference is always on) or 01 (reference is on, but powers down in power-down mode) in
the reference control register (05h). By default, the internal voltage reference is powered down. To select the
internal reference for use with the ADC, set the REFSEL[1:0] bits to 10. The REFOUT pin provides a buffered
reference output voltage when the internal reference voltage is enabled. The negative reference output is the
REFCOM pin, as shown in Figure 45. Connect a capacitor in the range of 1 μF to 47 μF between REFOUT and
REFCOM. Larger capacitor values help filter more noise at the expense of a longer reference start-up time.
The capacitor is not required if the internal reference is not used. However, the internal reference must be
powered on if using the IDACs.
The internal reference requires a start-up time that must be accounted for before starting a conversion, as shown
in Table 4.
Table 4. Internal Reference Settling Time
REFOUT CAPACITOR
1 µF
10 µF
47 µF
SETTLING ERROR
SETTLING TIME (ms)
0.01%
4.5
0.001%
5.9
0.01%
4.9
0.001%
6.3
0.01%
5.5
0.001%
7.0
9.3.3.2 External Reference
The ADS114S0xB provides two external reference inputs selectable through the reference multiplexer. The
reference inputs are differential with independent positive and negative inputs. REFP0 and REFN0 or REFP1
and REFN1 can be selected as the ADC reference. REFP1 and REFN1 are shared inputs with analog pins AIN6
and AIN7 in the ADS114S08B.
Without buffering, the reference input impedance is approximately 250 kΩ. The reference input current can lead
to possible errors from either high reference source impedance or through reference input filtering. To reduce the
input current, use either internal or external reference buffers. In most applications external reference buffering is
not necessary.
Connect a bypass capacitor across the external reference input pins if an external reference is used. Follow the
specified absolute and differential reference voltage requirements.
9.3.3.3 Reference Buffers
The device has two individually selectable reference input buffers to lower the reference input current. Use the
REFP_BUF and REFN_BUF bits in the reference control register (05h) to enable or disable the positive and
negative reference buffers respectively. These bits are active low. Writing a 1 to REFP_BUF or REFN_BUF
disables the reference buffers.
The reference buffers are recommended to be disabled when the internal reference is selected for
measurements. The positive reference buffer is recommended to be disabled when REFPx is at AVDD and the
negative reference buffer is recommended to be disabled when REFNx is at AVSS.
28
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9.3.4 Clock Source
The ADS114S0xB system clock is either provided by the internal low-drift 4.096-MHz oscillator or an external
clock source on the CLK input. Use the CLK bit within the data rate register (04h) to select the internal
4.096-MHz oscillator or an external clock source.
The device defaults to using the internal oscillator. If the device is reset (from either the RESET pin, or the
RESET command), then the clock source returns to using the internal oscillator.
9.3.5 Delta-Sigma Modulator
A delta-sigma (ΔΣ) modulator is used in the devices to convert the analog input voltage into a pulse code
modulated (PCM) data stream. The modulator runs at a modulator clock frequency of fMOD = fCLK / 16, where fCLK
is either provided by the internal 4.096-MHz oscillator or the external clock source.
9.3.6 Digital Filter
The devices offer digital filter options for decimation of the digital data stream coming from the delta-sigma
modulator. The implementation of the digital filter is determined by the data rate setting. Figure 46 shows the
digital filter implementation.
The low-latency digital filter is a finite impulse response (FIR) filter that provides settled data, given that the
analog input signal has settled to the final value before the conversion is started. This digital filter implementation
is especially useful when multiple channels must be scanned in minimal time.
fCLK = 4.096 MHz
fCLK / 16
Low-Latency Filter
fMOD = 256 kHz
SINC3
Filter
LL2
Filter
LL1
Filter
SINC1
Filter
SINC1
Filter
20, 10, 5, 2.5 SPS
ADC
400, 200, 100, 60, 50, 16.6 SPS
ADC data
output
4000, 2000, 1000, 800 SPS
DR[3:0] bits of
DATARATE Register
0000 = 2.5 SPS
1000 = 200 SPS
0001 = 5 SPS
1001 = 400 SPS
0010 = 10 SPS
1010 = 800 SPS
0011 = 16.6 SPS
1011 = 1000 SPS
0100 = 20 SPS
1100 = 2000 SPS
0101 = 50 SPS
1101 = 4000 SPS
0110 = 60 SPS
1110 = 4000 SPS
0111 = 100 SPS
1111 = Reserved
NOTE: LL filter = low-latency filter.
Figure 46. Digital Filter Architecture
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The device requires a set number of modulator clocks to output a single ADC conversion data. This number is
known as the oversampling ratio (OSR). The OSR of the digital filter is set using the DR[3:0] bits in the data rate
register. The data rate is determined by Equation 8.
Data Rate = fMOD / OSR
(8)
The relationship between the data rate and oversampling ratio is shown in Table 5.
Table 5. ADC Data Rates and Digital Filter Oversampling Ratios
(1)
(2)
NOMINAL DATA RATE
(SPS) (1)
DATA RATE REGISTER
DR[3:0]
OVERSAMPLING
RATIO (2)
2.5
0000
102400
5
0001
51200
10
0010
25600
16.6
0011
15360
20
0100
12800
50
0101
5120
60
0110
4264
100
0111
2560
200
1000
1280
400
1001
640
800
1010
320
1000
1011
256
2000
1100
128
4000
1101
64
Valid for the internal oscillator or an external 4.096-MHz clock. The data rate scales with internal
oscillator or external clock frequency.
The oversampling ratio is fMOD divided by the data rate; fMOD = fCLK / 16.
9.3.6.1 Digital Filter Frequency Response
The digital filter provides many data rate options for rejecting 50-Hz and 60-Hz line cycle noise. At data rates of
2.5 SPS, 5 SPS, 10 SPS, and 20 SPS, the filter rejects both 50-Hz and 60-Hz line frequencies. At data rates of
16.6 SPS and 50 SPS, the filter has a notch at 50 Hz. At a 60-SPS data rate, the filter has a notch at 60 Hz.
For detailed frequency response plots showing line cycle noise rejection, download the ADS1x4S0x design
calculator from www.ti.com.
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0
0
-20
-20
Magnitude (dB)
Magnitude (dB)
Figure 47 to Figure 61 illustrate the frequency response of the digital filter for different data rates. Table 6 lists
the bandwidth of the digital filter for each data rate.
-40
-60
-80
-40
-60
-80
-100
-100
-120
-120
0
10
20
30
40
Frequency (Hz)
50
60
0
70
10
20
fCLK = 4.096 MHz
30
40
Frequency (Hz)
50
60
70
fCLK = 4.096 MHz
Figure 47. Digital Filter Frequency Response,
Data Rate = 2.5 SPS
Figure 48. Digital Filter Frequency Response,
Data Rate = 5 SPS
0
0
-40
Magnitude (dB)
Magnitude (dB)
-20
-60
-80
-20
-40
-100
-60
-120
0
10
20
30
40
50
60
Frequency (Hz)
70
80
90
0
100
20
40
fCLK = 4.096 MHz
120
140
160
fCLK = 4.096 MHz
Figure 49. Digital Filter Frequency Response,
Data Rate = 10 SPS
Figure 50. Digital Filter Frequency Response,
Data Rate = 16.6 SPS
0
0
-20
-20
-40
-40
Magnitude (dB)
Magnitude (dB)
60
80
100
Frequency (Hz)
-60
-80
-100
-60
-80
-100
-120
0
20
40
60
80 100 120
Frequency (Hz)
140
160
180
200
-120
40
45
fCLK = 4.096 MHz
50
55
60
Frequency (Hz)
65
70
fCLK = 4.096 MHz
Figure 51. Digital Filter Frequency Response,
Data Rate = 20 SPS
Figure 52. Digital Filter Frequency Response,
Data Rate = 20 SPS, Zoomed to 50 Hz and 60 Hz
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0
0
-20
-20
Magnitude (dB)
Magnitude (dB)
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-40
-60
-40
-60
0
50
100
150
200
250
Frequency (Hz)
300
350
400
0
100
fCLK = 4.096 MHz
0
Magnitude (dB)
Magnitude (dB)
400
Figure 54. Digital Filter Frequency Response,
Data Rate = 60 SPS
0
-20
-40
-20
-40
-60
-60
0
200
400
600
Frequency (Hz)
800
0
1000
400
800
1200
Frequency (Hz)
1600
2000
fCLK = 4.096 MHz
fCLK = 4.096 MHz
Figure 56. Digital Filter Frequency Response,
Data Rate = 200 SPS
Figure 55. Digital Filter Frequency Response,
Data Rate = 100 SPS
0
0
-20
-20
Magnitude (dB)
Magnitude (dB)
300
fCLK = 4.096 MHz
Figure 53. Digital Filter Frequency Response,
Data Rate = 50 SPS
-40
-60
-40
-60
-80
-80
0
500
1000
1500 2000 2500
Frequency (Hz)
3000
3500
4000
0
1000
2000
fCLK = 4.096 MHz
3000 4000 5000
Frequency (Hz)
6000
7000
8000
fCLK = 4.096 MHz
Figure 57. Digital Filter Frequency Response,
Data Rate = 400 SPS
32
200
Frequency (Hz)
Figure 58. Digital Filter Frequency Response,
Data Rate = 800 SPS
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0
0
-20
Magnitude (dB)
Magnitude (dB)
-20
-40
-60
-40
-60
-80
-100
-80
0
2000
4000
6000
Frequency (Hz)
8000
10000
0
4000
8000
12000
Frequency (Hz)
16000
20000
fCLK = 4.096 MHz
fCLK = 4.096 MHz
Figure 60. Digital Filter Frequency Response,
Data Rate = 2 kSPS
Figure 59. Digital Filter Frequency Response,
Data Rate = 1 kSPS
0
Magnitude (dB)
-20
-40
-60
-80
-100
0
8000
16000
24000
Frequency (Hz)
32000
40000
fCLK = 4.096 MHz
Figure 61. Digital Filter Frequency Response,
Data Rate = 4 kSPS
Table 6. Digital Filter Bandwidth
(1)
NOMINAL DATA RATE (SPS) (1)
–3-dB BANDWIDTH (Hz) (1)
2.5
1.1
5
2.2
10
4.7
16.6
7.4
20
13.2
50
22.1
60
26.6
100
44.4
200
89.9
400
190
800
574
1000
717
2000
1434
4000
2868
Valid for the internal oscillator or an external 4.096-MHz clock. Scales proportional with fCLK.
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The digital filter notches and output data rate scale proportionally with the clock frequency. For example, a notch
that appears at 20 Hz when using a 4.096-MHz clock appears at 10 Hz if a 2.048-MHz clock is used. The
internal oscillator can vary over temperature as specified in the Electrical Characteristics table. The data rate,
conversion time, and filter notches consequently vary by the same percentage. Consider using an external
precision clock source if a digital filter notch at a specific frequency with a tighter tolerance is required.
9.3.6.2 Data Conversion Time
The amount of time required to receive data from the ADC depends on more than just the nominal data rate of
the device. The data period also depends on the mode of operation and other configurations of the device. In
normal operation, the data settles in one data period. However, a small amount of latency exists to set up the
device, calculate the conversion data from the modulator samples, and other overhead that adds time to the
conversion. For this reason, the first conversion data takes longer than subsequent data conversions.
Table 7 shows the conversion times for the digital filter for each ADC data rate and various conversion modes.
Table 7. Data Conversion Time
NOMINAL
DATA RATE (1)
(SPS)
(1)
(2)
FIRST DATA
FOR CONTINUOUS CONVERSION MODE
OR SINGLE-SHOT CONVERSION MODE (2)
SECOND AND SUBSEQUENT
CONVERSIONS FOR CONTINUOUS
CONVERSION MODE
ms
NUMBER OF
tMOD PERIODS
ms
NUMBER OF
tMOD PERIODS
2.5
406.559
104079
400
102400
5
206.559
52879
200
51200
10
106.559
27279
100
25600
16.6
60.309
15439
60
15360
20
56.559
14479
50
12800
50
20.211
5174
20
5120
60
16.965
4343
16.66
4264
100
10.211
2614
10
2560
200
5.211
1334
5
1280
400
2.711
694
2.5
640
800
1.461
374
1.25
320
1000
1.211
310
1
256
2000
0.711
182
0.5
128
4000
0.461
118
0.25
64
Valid for the internal oscillator or an external 4.096-MHz clock. Scales proportional with fCLK.
Conversions start at the rising edge of the START/SYNC pin or on the seventh SCLK falling edge for a START command.
9.3.6.3 Note on Conversion Time
Each data period consists of time required for the modulator to sample the analog inputs. However, there is
additional time required before the samples become an ADC conversion result.
When a new conversion is started, there is a configuration delay time of 14 · tMOD (where tMOD = 16 · tCLK) that is
added before the conversion starts. This delay allows for additional settling time for external RC filters on the
analog inputs and for the antialiasing filter after the PGA. The configuration delay occurs at the start of a new
conversion after a START command is sent, the START/SYNC pin is taken high, or a WREG command is sent
to change any configuration register from address 03h to 07h (as described in the WREG section).
Also, overhead time is needed to convert the modulator samples into an ADC conversion result. This overhead
time includes any necessary offset or gain compensation after the digital filter accumulates a data result. The first
conversion when the device is in continuous conversion mode (just as in single-shot conversion mode) includes
the configuration delay, the modulator sampling time, and the overhead time. The second and subsequent
conversions are the normal data period (period as given by the inverse of the data rate).
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Figure 62 shows the time sequence for the ADC in both continuous conversion and single-shot conversion
modes.
Single-shot conversion mode
Conversion
start(1)
Data
ready
Configuration delay
Modulator sampling
x
x
Sampling for
first data
ADC overhead
x
DRDY
Continuous conversion mode
Conversion
start(1)
x
First
data ready
Second
data ready
Third data
ready
Sampling for
first data
Sampling for
second data
x
Continued
sampling
Sampling for
third data
DRDY
(1)
Conversions start at the rising edge of the START/SYNC pin or on the seventh SCLK falling edge for a START
command.
Figure 62. Single-Shot Conversion Mode and Continuous Conversion Mode Sequences
9.3.6.4 50-Hz and 60-Hz Line Cycle Rejection
If the ADC connection leads are in close proximity to industrial motors and conductors, coupling of 50-Hz and
60-Hz power line frequencies can occur. The coupled noise interferes with the signal voltage, and can lead to
inaccurate or unstable conversions. The digital filter provides enhanced rejection of power-line-coupled noise for
data rates of 60 SPS and less. Program the filter to tradeoff data rate and conversion latency versus the desired
level of line cycle rejection. Table 8 summarizes the ADC 50-Hz and 60-Hz line-cycle rejection based on ±1-Hz
and ±2-Hz tolerance of power-line frequency. The best possible power-line rejection is provided by using an
accurate ADC clock.
Table 8. 50-Hz and 60-Hz Line Cycle Rejection
DATA RATE (SPS) (1)
(1)
DIGITAL FILTER LINE CYCLE REJECTION (dB)
50 Hz ± 1 Hz
60 Hz ± 1 Hz
50 Hz ± 2 Hz
60 Hz ± 2 Hz
2.5
–113.7
–95.4
–97.7
–92.4
5
–111.9
–95.4
–87.6
–81.8
10
–111.5
–95.4
–85.7
–81.0
16.6
–33.8
–20.9
–27.8
–20.8
20
–95.4
–95.4
–75.5
–80.5
50
–33.8
–15.5
–27.6
–15.1
60
–13.4
–35.0
–12.6
–29.0
fCLK = 4.096 MHz.
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9.3.7 Excitation Current Sources (IDACs)
The ADS114S0xB incorporates two integrated, matched current sources (IDAC1, IDAC2). The current sources
provide excitation current to resistive temperature devices (RTDs), thermistors, diodes, and other resistive
sensors that require constant current biasing. The current sources are programmable to output values between
10 μA to 2000 μA using the IMAG[3:0] bits in the excitation current register 1 (06h). Each current source can be
connected to any of the analog inputs AINx as well as the REFP1 and REFN1 inputs for the ADS114S06B. Both
current sources can also be connected to the same pin. The routing of the IDACs is configured by the
I1MUX[3:0] and I2MUX[3:0] bits in the excitation current register 2 (07h). In three-wire RTD applications, the
matched current sources can be used to cancel errors caused by sensor lead resistance (see the Typical
Application section for more details). Figure 63 details the IDAC connection through the input multiplexer.
I1MUX[3:0] bits of the IDACMUX register.
IDAC routing to AIN8 ± AIN11 is available
only on the ADS114S08B.
AIN0
AIN0
0000
AIN1
AIN1
0001
AIN2
AIN2
0010
AIN3
AIN3
0011
AIN4
AIN4
0100
AIN5
0101
AIN6 / REFP1
AIN6/REFP1
0110
AIN7 / REFN1
AIN7/REFN1
0111
1000
AIN5
AIN8
AIN8
ADS114S08B AIN9
Only
AIN10
AIN9
1001
AIN10
1010
AIN11
AIN11
1011
AINCOM
1100
AINCOM
AVDD
IDAC1
Mux
IDAC1
IMAG[3:0] bits of the IDACMAG register.
0000 = Off
0001 = 10 µA
0010 = 50 µA
0011 = 100 µA
0100 = 250 µA
0101 = 500 µA
0110 = 750 µA
0111 = 1000 µA
1000 = 1500 µA
1001 = 2000 µA
No Connection 1101-1111
AIN0
0000
AIN1
0001
AIN2
0010
AIN3
0011
AIN4
0100
AIN5
0101
AIN6/REFP1
0110
AIN7/REFN1
0111
AIN8
1000
AIN9
1001
AIN10
1010
AIN11
1011
AINCOM
1100
AVDD
IDAC2
Mux
IDAC2
No Connection 1101-1111
I2MUX[3:0] bits of the IDACMUX register.
IDAC routing to AIN8 ± AIN11 is available
only on the ADS114S08B.
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Figure 63. IDAC Block Diagram
The internal reference must be enabled for IDAC operation. As any current source, the IDAC requires voltage
headroom to the positive supply to operate. This voltage headroom is the compliance voltage. When driving
resistive sensors and biasing resistors, take care not to exceed the compliance voltage of the IDACs, otherwise
the specified accuracy of the IDAC current may not be met. For IDAC compliance voltage specifications, see the
Electrical Characteristics table.
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9.3.8 Bias Voltage Generation
The ADS114S0xB provides an internal bias voltage generator, VBIAS, that is set to (AVDD + AVSS) / 2. The
bias voltage is internally buffered and can be established on the analog inputs AIN0 to AIN5 and AINCOM using
the VB_AINx bits in the sensor biasing register (08h). A typical use case for VBIAS is biasing unbiased
thermocouples to within the common-mode voltage range of the PGA. A block diagram of the VBIAS voltage
generator and connection diagram is shown in Figure 64.
AVDD
AIN0
AIN1
AIN2
Input
Mux
AIN3
AIN4
AIN5
R
AINCOM
(AVDD + AVSS) / 2
VB_AINx bits of VBIAS register
0 = VBIAS not connected to AINx
1 = VBIAS connected to AINx
R
AVSS
Figure 64. VBIAS Block Diagram
The start-up time of the VBIAS voltage depends on the pin load capacitance. The total capacitance includes any
capacitance connected from VBIAS to AVDD, AVSS, and ground. Table 9 lists the VBIAS voltage settling times
for various external load capacitances. Ensure the VBIAS voltage is fully settled before starting a conversion.
Table 9. VBIAS Settling Time
LOAD CAPACITANCE
SETTLING TIME
0.1 µF
280 µs
1 µF
2.8 ms
10 µF
28 ms
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9.3.9 System Monitor
The ADS114S0xB provides a set of system monitor functions. These functions measure the device temperature,
analog power supply, digital power supply, or use current sources to detect sensor malfunction. System monitor
functions are enabled through the SYS_MON[2:0] bits of the system control register (09h).
9.3.9.1 Internal Temperature Sensor
On-chip diodes provide temperature-sensing capability. Enable the internal temperature sensor by setting
SYS_MON[2:0] = 010 in the system control register (09h). The temperature sensor outputs a voltage proportional
to the device temperature as specified in the Electrical Characteristics table.
When measuring the internal temperature sensor, the analog inputs are disconnected from the ADC and the
output voltage of the temperature sensor is routed to the ADC for measurement using the selected PGA gain,
data rate, and voltage reference. If enabled, PGA gain must be limited to 4 for the temperature sensor
measurement to remain within the allowed absolute input voltage range of the PGA. As a result of the low device
junction-to-PCB thermal resistance (RθJB), the internal device temperature closely tracks the printed circuit board
(PCB) temperature.
9.3.9.2 Power Supply Monitors
The ADS114S0xB provides a means for monitoring both the analog and digital power supply (AVDD and DVDD).
The power-supply voltages are divided by a resistor network to reduce the voltages to within the ADC input
range. The reduced power-supply voltage is routed to the ADC input multiplexer. The analog (VANLMON) and
digital (VDIGMON) power-supply readings are scaled by Equation 9 and Equation 10, respectively:
VANLMON = (AVDD – AVSS) / 4
VDIGMON = (DVDD – DGND) / 4
(9)
(10)
Enable the supply voltage monitors using the SYS_MON[2:0] bits in the system control register (09h). Setting
SYS_MON[2:0] to 011 measures VANLMON, and setting SYS_MON[2:0] to 100 measures VDIGMON.
When the supply voltage monitor is enabled, the analog inputs are disconnected from the ADC and the PGA gain
is set to 1, regardless of the GAIN[2:0] bit values in the gain setting register (03h). Supply voltage monitor
measurements can be done with either the PGA enabled or PGA disabled via the PGA_EN[1:0] register. To
obtain valid power-supply monitor readings, the reference voltage must be larger than the power-supply
measurements shown in Equation 9 and Equation 10.
9.3.9.3 Burn-Out Current Sources
To help detect a possible sensor malfunction, the ADS114S0xB provides selectable current sources to function
as burn-out current sources (BOCS) using the SYS_MON[2:0] bits in the system control register (09h). Current
sources are set to values of 0.2 µA, 1 µA, and 10 µA with SYS_MON[2:0] settings of 101, 110, and 111,
respectively.
When enabled, one BOCS sources current to the selected positive analog input (AINP) and the other BOCS
sinks current from the selected negative analog input (AINN). With an open-circuit in a burned out sensor, these
BOCSs pull the positive input towards AVDD and the negative input towards AVSS, resulting in a full-scale
reading. A full-scale reading can also indicate that the sensor is overloaded or that the reference voltage is
absent. A near-zero reading can indicate a shorted sensor. Distinguishing a shorted sensor condition from a
normal reading can be difficult, especially if an RC filter is used at the inputs. The voltage drop across the
external filter resistance and the residual resistance of the multiplexer can cause the output to read a value
higher than zero.
The ADC readings of a functional sensor can be corrupted when the burn-out current sources are enabled. The
burn-out current sources are recommended to be disabled when performing the precision measurement, and are
recommended to be enabled only when testing for sensor fault conditions. If the global chop mode is enabled,
disable this mode before making a measurement with the burn-out current sources.
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9.3.10 Status Register
The ADS114S0xB has a one-byte status register (01h) that contains a POR flag to indicate if a start-up or poweron reset condition has occurred and a RDY flag to indicate when a device has started up and is ready for
communication.
The status register data field and field description are found in Figure 77 and Table 17.
9.3.10.1 POR Flag
After the power supplies are turned on, the ADC remains in reset until DVDD, IOVDD, and the analog power
supply (AVDD – AVSS) voltage exceed the respective power-on reset (POR) voltage thresholds. If a POR event
has occurred, the FL_POR flag (bit 7 of the status register) is set. This flag indicates that a POR event has
occurred and has not been cleared. This flag is cleared with a user register write to set the bit to 0. The power-on
reset is described further in the Power-On Reset section.
9.3.10.2 RDY Flag
The RDY flag indicates that the device has started up and is ready to receive a configuration change. During a
reset or POR event, the device is resetting the register map and may not be available. The RDY flag is shown
with bit 6 of the status register.
9.3.11 General-Purpose Inputs and Outputs (GPIOs)
The ADS114S06B offers four dedicated general-purpose input and output (GPIO) pins, and the ADS114S08B
offers four pins (AIN8 to AIN11) that serve a dual purpose as either analog inputs or GPIOs.
Two registers control the function of the GPIO pins. Use the CON[3:0] bits of the GPIO configuration register
(11h) to configure a pin as a GPIO pin. The upper four bits (DIR[3:0]) of the GPIO data register (10h) configure
the GPIO pin as either an input or an output. The lower four bits (DAT[3:0]) of the GPIO data register contain the
input or output GPIO data. If a GPIO pin is configured as an input, the respective DAT[x] bit reads the status of
the pin; if a GPIO pin is configured as an output, write the output status to the respective DAT[x] bit. For more
information about the use of GPIO pins, see the Configuration Registers section.
Figure 65 depicts a diagram of how these functions are combined onto a single pin. Note that when the pin is
configured as a GPIO, the corresponding logic is powered from AVDD and AVSS. When the devices are
operated with bipolar analog supplies, the GPIO outputs bipolar voltages. Care must be taken to not load the
GPIO pins when used as outputs because large currents can cause droop or noise on the analog supplies. GPIO
pins use Schmitt triggered inputs with hysteresis to make the input more resistance to noise; see the Electrical
Characteristics table for GPIO thresholds.
For connections of unused GPIO pins, see the Unused Inputs and Outputs section.
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AVDD
CON[3:0] bits of GPIOCON register
0 = no connect
1 = connect
DAT[3:0] bits of GPIODAT register
0 = VGPIO is low
1 = VGPIO is high
GPIO
1 of 4
AIN8
AIN9
Write
GPIO[0]
0
GPIO[1]
Read
0
1
GPIO[2]
AIN10
GPIO Read Select
GPIO[3]
AIN11
DIR[7:4] bits of GPIODAT register
0 = Output
1 = Input
GPIO logic is powered from
AVDD to AVSS
AVSS
Figure 65. GPIO Block Diagram
9.3.12 Calibration
The ADC incorporates offset and gain calibration commands, as well as user-offset and full-scale (gain)
calibration registers to calibrate the ADC. The ADC calibration registers are 16 bits wide. Use calibration to
correct internal ADC errors or overall system errors. Calibrate by sending calibration commands to the ADC, or
by direct user calibration. In user calibration, the user calculates and writes the correction values to the
calibration registers. The ADC performs self or system-offset calibration, or a system gain calibration. Perform
offset calibration before system gain calibration. After power-on, wait for the power supplies and reference
voltage to fully settle before calibrating.
As shown in Figure 66, the value of the offset calibration register is subtracted from the filter output and then
multiplied by the full-scale register value divided by 4000h. The data are then clipped to a 16-bit value to provide
the final output.
AINN
Digital
Filter
ADC
AINP
ADC
+
Output Data
Clipped to 16 bits
Final
Output
1/4000h
OFCAL[1:0] registers
(register addresses = 0Ch, 0Bh)
> 0000h: negative offset
= 0000h: no offset
< 0000h: positive offset
FSCAL[1:0] registers
(register addresses = 0Fh, 0Eh)
< 4000h: Gain > 1
= 4000h: Gain = 1
> 4000h: Gain < 1
Figure 66. ADC Calibration Block Diagram
Calibration commands cannot be used when the device is in standby mode (when the START/SYNC pin is low,
or when the STOP command is issued).
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9.3.12.1 Offset Calibration
The offset calibration word is 16 bits, consisting of two 8-bit registers, as shown in the two registers starting with
offset calibration register 1. The offset value is twos complement format with a maximum positive value equal to
7FFFh, and a maximum negative value equal to 8000h. This value is subtracted from each output reading as an
offset correction. A register value equal to 0000h has no offset correction. If global chop mode is enabled, the
offset calibration register is disabled. Table 10 shows example settings of the offset register.
Table 10. Offset Calibration Register Values
(1)
OFC REGISTER VALUE
OFFSET CALIBRATED OUTPUT CODE (1)
0001h
FFFFh
0000h
0000h
FFFFh
0001h
Ideal output code with shorted input, excluding ADC noise and offset voltage error.
The user can select how many samples (1, 4, 8, or 16) to average for self or system offset calibration using the
CAL_SAMP[1:0] bits in the system control register (09h). Fewer readings shorten the calibration time but also
provide less accuracy. Averaging more readings takes longer but yields a more accurate calibration result by
reducing the noise level.
Two commands can be used to perform offset calibration. SFOCAL is a self offset calibration that internally sets
the input to mid-scale using the SYS_MON[2:0] = 001 setting and takes a measurement of the offset. SYOCAL is
a system offset calibration where the user must input a null voltage to calibrate the system offset. After either
command is issued, the OFC register is updated.
After an offset calibration is performed, the device starts a new conversion and DRDY falls to indicate a new
conversion has completed.
9.3.12.2 Gain Calibration
The full-scale (gain) calibration word is 16 bits consisting of two 8-bit registers, as shown in the two registers
starting with gain calibration register 1. The gain calibration value is straight binary, normalized to a unity-gain
correction factor at a register value equal to 4000h. Table 11 shows register values for selected gain factors. Do
not exceed the PGA input range limits during gain calibration.
Table 11. Gain Calibration Register Values
FSC REGISTER VALUE
GAIN FACTOR
4333h
1.05
4000h
1.00
3CCCh
0.95
All gains of the ADS114S0xB are factory trimmed to meet the gain error specified in the Electrical Characteristics
table at TA = 25°C. When the gain drift of the devices over temperature is very low, there is typically no need for
self gain calibration.
The SYGCAL command initiates a system gain calibration, where the user sets the input to full-scale to remove
gain error. After the SYGCAL is issued, the FSC register is updated. As with the offset calibration, the
CAL_SAMP[1:0] bits determine the number of samples used for a gain calibration.
As with an offset calibration, the device starts a new conversion after a gain calibration and DRDY falls to
indicate a new conversion has completed.
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9.4 Device Functional Modes
The device operates in three different modes: power-down mode, standby mode, and conversion mode.
Figure 67 shows a flow chart of the different operating modes and how the device transitions from one mode to
another.
Power-On Reset or
RESET pin high or
RESET command?(1)
Power-down
Mode(2)
Reset device to
default settings
WAKEUP
Command?
No
Yes
Standby
Mode
No
Complete current
conversion(4)
START/SYNC
rising edge or START
Command?
Yes
Yes
Conversion
Mode
No
START/SYNC
pin low or STOP
Command?
Start new
conversion
1 = Single-Shot
conversion mode
Conversion
mode selection(3)
0 = Continuous
conversion mode
(1)
Any reset (power-on, command, or pin), immediately resets the device.
(2)
A POWERDOWN command aborts an ongoing conversion and immediately puts the device into power-down mode.
(3)
The conversion mode is selected with the MODE bit in the data rate register.
(4)
The rising edge of the START/SYNC pin or the START command starts a new conversion without completing the
current conversion.
Figure 67. Operating Flow Chart
9.4.1 Reset
The ADS114S0xB is reset in one of three ways:
• Power-on reset
• RESET pin
• RESET command
When a reset occurs, the configuration registers reset to default values and the device enters standby mode. The
device then waits for the rising edge of the START/SYNC pin or a START command to enter conversion mode. If
the device had been using an external clock, the reset sets the device to use the internal oscillator as a default
configuration. See the Timing Characteristics section for reset timing information.
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Device Functional Modes (continued)
9.4.1.1 Power-On Reset
The ADS114S0xB incorporates a power-on reset circuit that holds the device in reset until all supplies reach
approximately 1.65 V. The power-on reset also ensures that the device starts operating in a known state in case
a brown-out event occurs, when the supplies have dipped below the minimum operating voltages. When the
device completes a POR sequence, the FL_POR flag in the status register is set high to indicate that a POR has
occurred.
Begin communications with the device 2.2 ms after the power supplies reach minimum operating voltages. The
only exception is polling the status register for the RDY bit. If the user polls the RDY bit, then use an SCLK rate
of half the maximum-specified SCLK rate to get a proper reading when the device is making internal
configurations. This 2.2-ms POR time is required for the internal oscillator to start up and the device to properly
set internal configurations. After the internal configurations are set, the device sets the RDY bit in the device
status register (01h). When this bit is set to 0, user configurations can be programmed into the device. Figure 68
shows the power-on reset timing sequence for the device.
DVDD, IOVDD
VPOR
VPOR
All supplies reach
minimum operating voltage
AVDD - AVSS
Internal
Oscillator Startup
FL_POR bit of
STATUS register is set to 1
Internal
Configuration
Standby
Mode
RDY bit of
STATUS register is set to 0
2.2 ms
If polling for RDY during this period, SCLK
must be less than half maximum rate
Figure 68. Power-On Reset Timing Sequence
9.4.1.2 RESET Pin
Reset the ADC by taking the RESET pin low for a minimum of 4 · tCLK· cycles, and then returning the pin high.
After the rising edge of the RESET pin, a delay time of td(RSSC) is required before sending the first serial interface
command or starting a conversion. See the Timing Characteristics section for reset timing information.
9.4.1.3 Reset by Command
Reset the ADC by using the RESET command (06h or 07h). The command is decoded on the seventh SCLK
falling edge. After sending the RESET command, a delay time of td(RSSC) is required before sending the first serial
interface command or starting a conversion. See the Timing Characteristics section for reset timing information.
9.4.2 Power-Down Mode
Power-down mode is entered by sending the POWERDOWN command. In this mode, all analog and digital
circuitry is powered down for lowest power consumption regardless of the register settings. Only the internal
voltage reference can be configured to stay on during power-down mode in case a faster start-up time is
required. All register values retain the current settings during power-down mode. The configuration registers can
be read and written in power-down mode. A WAKEUP command must be issued in order to exit power-down
mode and to enter standby mode.
When the POWERDOWN command is issued, the device enters power-down mode 2 · tCLK after the seventh
SCLK falling edge of the command. For lowest power consumption (on DVDD and IOVDD), stop the external
clock when in power-down mode. TI recommends selecting the internal oscillator before sending the
POWERDOWN command to avoid issues with the command decoding.
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Device Functional Modes (continued)
To release the device from POWERDOWN, issue the WAKEUP command to enter standby mode. The device
then waits for the rising edge of the START/SYNC pin or a START command to go into conversion mode.
When in power-down mode, the device responds to the RREG, RDATA, and WAKEUP commands. The WREG
and RESET commands can also be sent, but are ignored until a WAKEUP command is sent and the internal
oscillator resumes operation.
9.4.3 Standby Mode
The device powers up in standby mode and automatically enters this mode whenever there is no ongoing
conversion. When the STOP command is sent (or the START/SYNC pin is taken low) in continuous conversion
mode, or when a conversion completes in single-shot conversion mode, the device enters standby mode.
Standby mode offers several different options and features to lower the power consumption:
• The PGA can be powered down by setting PGA_EN[1:0] to 00 in the gain setting register (03h).
• The internal voltage reference can be powered down by setting REFCON[1:0] to 00 in the reference control
register (05h). This setting also turns off the IDACs.
• The digital filter is held in reset state.
• The clock to the modulator and digital core is gated to decrease dynamic switching losses.
If powered down in standby mode, the PGA and internal reference can require extra time to power up. Extra
delay may be required between power up of the PGA or the internal reference, and the start of conversions. In
particular, the reference power up time is dependent on the capacitance between REFOUT and REFCOM.
Calibration commands are not decoded when the device is in standby mode.
9.4.4 Conversion Modes
The ADS114S0xB offers two conversion modes: continuous conversion and single-shot conversion mode.
Continuous-conversion mode converts indefinitely until stopped by the user. Single-shot conversion mode
performs one conversion after the START/SYNC pin is taken high or after the START command is sent. Use the
MODE bit in the data rate register (04h) to program the conversion mode. Figure 69 shows how the
START/SYNC pin and the START command are used to control ADC conversions.
(2)
(1)
(2)
DRDY
START/SYNC Pin
SCLK
START
Command
DIN
START (3)
Standby Mode
STOP
Conversion Mode
Standby Mode
(1)
DRDY rises at the first SCLK rising edge or the rising edge of the START/SYNC pin.
(2)
START and STOP commands take effect 2 · tCLK after the seventh SCLK falling edge. The conversion starts 2 · tCLK
after the START/SYNC rising edge.
(3)
To synchronize a conversion, the STOP command must be issued prior to the START command. STOP and START
commands can be issued without a delay between the commands.
Figure 69. Conversion Start and Stop Timing
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Device Functional Modes (continued)
ADC conversions are controlled by the START/SYNC pin or by serial commands. For the device to start
converting in continuous conversion or single-shot conversion mode, a START command must be sent or the
START/SYNC pin must be taken high. If using commands to control conversions, keep the START/SYNC pin low
to avoid possible contentions between the START/SYNC pin and commands.
Conversions can be synchronized to perform a conversion at a particular time. To synchronize the conversion
with the START/SYNC pin, take the pin low. The rising edge of the START/SYNC pin starts a new conversion.
Similarly, a conversion can be synchronized using the START command. If the device is in standby mode, issue
a START command. If the device is in conversion mode, issue a STOP command followed by a START
command. The STOP and START commands can be consecutive. A new conversion starts on the seventh
SCLK falling edge of the START command.
9.4.4.1 Continuous Conversion Mode
The device is configured for continuous conversion mode by setting the MODE bit to 0 in the data rate register
(04h). A START command must be sent or the START/SYNC pin must be taken high for the device to start
converting continuously. When controlling the device with commands, hold the START/SYNC pin low. Taking the
START/SYNC pin low or sending the STOP command stops the device from converting after the currently
ongoing conversion completes, indicated by the falling edge of DRDY. The device enters standby mode
thereafter.
For information on the exact timing of continuous conversion mode data, see Table 7.
9.4.4.2 Single-Shot Conversion Mode
The device is configured for single-shot conversion mode by setting the MODE bit to 1 in the data rate register
(04h). A START command must be sent or the START/SYNC pin must be taken high for the device to start a
single conversion. After the conversion completes, the device enters standby mode again. To start a new
conversion, the START command must be sent again or the START/SYNC pin must be taken low and then high
again.
For information on the exact timing of single-shot conversion mode data, see Table 7.
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9.5 Programming
9.5.1 Serial Interface
The ADC has an SPI-compatible, bidirectional serial interface that is used to read the conversion data as well as
to configure and control the ADC. Only SPI mode 1 (CPOL = 0, CPHA = 1) is supported. The serial interface
consists of five control lines: CS, SCLK, DIN, DOUT/DRDY, and DRDY but can be used with only four or even
three control signals. If the ADS114S08B or ADS114S06B is the only device connected to the SPI bus, then the
CS input can be tied low so that only SCLK, DIN, and DOUT/DRDY are required to communicate with the device.
9.5.1.1 Chip Select (CS)
The CS pin is an active low input that enables the ADC serial interface for communication and is useful when
multiple devices share the same serial bus. CS must be low during the entire data transaction. When CS is high,
the serial interface is reset, SCLK input activity is ignored (blocking input commands), and the DOUT/DRDY
output enters a high-impedance state. ADC conversions are not affected by the state of CS. In situations where
multiple devices are present on the bus, the dedicated DRDY pin can provide an uninterrupted monitor of the
conversion status and is not affected by CS. If the serial bus is not shared with another peripheral, CS can be
tied to DGND to permanently enable the ADC interface and DOUT/DRDY can be used to indicate conversion
status. These changes reduce the serial interface from five I/Os to three I/Os.
9.5.1.2 Serial Clock (SCLK)
The serial interface clock is a noise-filtered, Schmidt-triggered input used to clock data into and out of the ADC.
Input data to the ADC are latched on the falling SCLK edge and output data from the ADC are updated on the
rising SCLK edge. Return SCLK low after the data sequence is complete. Even though the SCLK input has
hysteresis, keep SCLK as clean as possible to prevent unintentional SCLK transitions. Avoid ringing and voltage
overshoot on the SCLK input. Place a series termination resistor at the SCLK drive pin to help reduce ringing.
9.5.1.3 Serial Data Input (DIN)
The serial data input pin (DIN) is used with SCLK to send data (commands and register data) to the device. The
device latches data on DIN on the SCLK falling edge. The device never drives the DIN pin. During data
readback, when no command is intended, keep DIN low.
9.5.1.4 Serial Data Output and Data Ready (DOUT/DRDY)
The DOUT/DRDY pin is a dual-function output. The pin functions as the digital data output and the ADC dataready indication.
First, this pin is used with SCLK to read conversion and register data from the device. Conversion or register
data are shifted out on DOUT/DRDY on the SCLK rising edge. DOUT/DRDY goes to a high-impedance state
when CS is high.
Second, the DOUT/DRDY pin indicates availability of new conversion data. DOUT/DRDY transitions low at the
same time that the DRDY pin goes low to indicate new conversion data are available. Both signals can be used
to detect if new data are ready. However, because DOUT/DRDY is disabled when CS is high, use the dedicated
DRDY pin when monitoring conversions on multiple devices on the SPI bus.
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Programming (continued)
9.5.1.5 Data Ready (DRDY)
The DRDY pin is an output that transitions low to indicate when conversion data are ready for retrieval. Initially,
DRDY is high at power-on. When converting, the state of DRDY depends on whether the conversion data are
retrieved or not. In continuous conversion mode after DRDY goes low, DRDY is driven high on the first SCLK
rising edge. If data are not read, DRDY remains low and then pulses high 24 · tCLK before the next DRDY falling
edge. The data must be retrieved before the next DRDY update, otherwise the data are overwritten by new data
and any previous data are lost. Figure 70 shows the DRDY operation without data retrieval. Figure 71 shows the
DRDY operation with data retrieval after each conversion completes.
DRDY
START/SYNC Pin
SCLK
START
Command
DIN
(1)
START
DRDY returns high with the rising edge of the first SCLK after a data ready indication.
Figure 70. DRDY Operation Without Data Retrieval
DRDY
START/SYNC
SCLK
START
Command
DIN
START (3)
DOUT/DRDY
RDATA
RDATA
'RQ¶W
Care
'RQ¶W
Care
Conversion
Data 1
(1)
Conversion
Data 2
DRDY returns high with the rising edge of the first SCLK after a data ready indication.
Figure 71. DRDY Operation With Data Retrieval
9.5.1.6 Timeout
The ADS114S0xB offers a serial interface timeout feature that is used to recover communication when a serial
interface transmission is interrupted. This feature is especially useful in applications where CS is permanently
tied low and is not used to frame a communication sequence. The SPI interface resets when no valid 8 bits are
received within 215 · tCLK. The timeout feature is enabled by setting the TIMEOUT bit to 1 in the system control
register (09h).
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Programming (continued)
9.5.2 Data Format
The devices provide 16 bits of data in binary twos complement format. The size of one code (LSB) is calculated
using Equation 11.
1 LSB = (2 · VREF / Gain) / 216 = +FS / 215
(11)
A positive full-scale input [VIN ≥ (+FS – 1 LSB) = (VREF / Gain – 1 LSB)] produces an output code of 7FFFh and a
negative full-scale input (VIN ≤ –FS = –VREF / Gain) produces an output code of 8000h. The output clips at these
codes for signals that exceed full-scale.
Table 12 summarizes the ideal output codes for different input signals.
Table 12. Ideal Output Code vs Input Signal
INPUT SIGNAL,
VIN = VAINP – VAINN
15
≥ FS (2
IDEAL OUTPUT CODE (1)
15
– 1) / 2
7FFFh
FS / 215
0001h
0
0000h
15
(1)
–FS / 2
FFFFh
≤ –FS
8000h
Excludes the effects of noise, INL, offset, and gain errors.
Mapping of the analog input signal to the output codes is shown in Figure 72.
7FFFh
0001h
0000h
FFFFh
¼
Output Code
¼
7FFEh
8001h
8000h
¼
-FS
2
15
FS
¼
-1
-FS
2
0
Input Voltage (VIN)
15
2
15
FS
2
-1
15
Figure 72. Code Transition Diagram
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9.5.3 Commands
Commands are used to control the ADC, access the configuration registers, and retrieve data. Many of the
commands are stand-alone (that is, single-byte). The register write and register read commands, however, are
multibyte, consisting of two command bytes plus the register data byte or bytes. The commands are listed in
Table 13.
Table 13. Command Definitions
COMMAND
DESCRIPTION
FIRST
COMMAND BYTE
SECOND
COMMAND BYTE
Control Commands
NOP
No operation
0000 0000 (00h)
—
WAKEUP
Wake-up from power-down mode
0000 001x (02h, 03h) (1)
—
POWERDOWN
Enter power-down mode
0000 010x (04h, 05h) (1)
—
(1)
—
RESET
Reset the device
0000 011x (06h, 07h)
START
Start conversions
0000 100x (08h, 09h) (1)
—
STOP
Stop conversions
0000 101x (0Ah, 0Bh) (1)
—
Calibration Commands
SYOCAL
System offset calibration
0001 0110 (16h)
—
SYGCAL
System gain calibration
0001 0111 (17h)
—
SFOCAL
Self offset calibration
0001 1001 (19h)
—
0001 001x (12h / 13h) (1)
—
Read nnnnn registers starting at address rrrrr
001r rrrr (2)
000n nnnn (3)
Write nnnnn registers starting at address rrrrr
(2)
000n nnnn (3)
Data Read Command
RDATA
Read data
Register Read and Write Commands
RREG
WREG
(1)
(2)
(3)
010r rrrr
x = don't care.
r rrrr = starting register address.
n nnnn = number of registers to read or write – 1.
Commands can be sent at any time, either during a conversion or when conversions are stopped. However, if
register read or write commands are in progress when conversion data are ready, the ADC blocks loading of
conversion data to the output shift register. The CS input pin can be taken high between commands; or held low
between consecutive commands. CS must stay low for the entire command sequence. Complete the command,
or terminate the command before completion by taking CS high. Only send the commands that are listed in
Table 13.
9.5.3.1 NOP
NOP is a no-operation command. The NOP command is used to clock out data without clocking in a command.
9.5.3.2 WAKEUP
Issue the WAKEUP command to exit power-down mode and to place the device into standby mode.
When running off the external clock, the external clock must be running before sending the WAKEUP command,
otherwise the command is not decoded.
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9.5.3.3 POWERDOWN
Sending the POWERDOWN command aborts a currently ongoing conversion and puts the device into powerdown mode. The device goes into power-down mode 2 · tCLK after the seventh SCLK falling edge of the
command.
For lowest power consumption on DVDD and IOVDD, stop the external clock when in power-down mode. The
device does not gate the external clock. When running off the external clock, provide at a minimum two
additional tCLKs after the POWERDOWN command is issued, otherwise the device does not enter power-down
mode. Alternatively, select the internal oscillator before sending the POWERDOWN command to avoid any
issues with decoding of the POWERDOWN and WAKEUP commands.
During power-down mode, the only commands that are available are RREG, RDATA, and WAKEUP.
9.5.3.4 RESET
The RESET command resets the digital filter and sets all configuration register values to default settings. A
RESET command also puts the device into standby mode. When in standby mode, the device waits for a rising
edge on the START/SYNC pin or a START command to resume conversions. After sending the RESET
command, a delay time of td(RSSC) is required before sending the first serial interface command or starting a
conversion. See the Timing Characteristics section for reset timing information.
If the device had been using an external clock, the reset sets the device to use the internal oscillator as a default
configuration.
9.5.3.5 START
When the device is configured for continuous conversion mode, issue the START command for the device to
start converting. Every time a conversion completes, the device automatically starts a new conversion until the
STOP command is sent.
In single-shot conversion mode, the START command is used to start a single conversion. After the conversion
completes, the device enters standby mode.
Tie the START/SYNC pin low when the device is controlled through the START and STOP commands. The
START command is not decoded if the START/SYNC pin is high. If the device is already in conversion mode, the
START command has no effect.
9.5.3.6 STOP
The STOP command is used in continuous conversion mode to stop the device from converting. The current
conversion is allowed to complete. After DRDY transitions low, the device enters standby mode. The STOP
command has no effect in single-shot conversion mode.
Hold the START/SYNC pin low when the device is controlled through START and STOP commands.
9.5.3.7 SYOCAL
The SYOCAL command initiates a system offset calibration. For a system offset calibration, the inputs must be
externally shorted to a voltage within the input range, ideally near the mid-supply voltage of (AVDD + AVSS) / 2.
The OFC registers are updated when the command completes. Calibration commands must be issued in
conversion mode.
9.5.3.8 SYGCAL
The SYGCAL command initiates the system gain calibration. For a system gain calibration, the input must be
externally set to full-scale. The FSC registers are updated after this operation. Calibration commands must be
issued in conversion mode.
9.5.3.9 SFOCAL
The SFOCAL command initiates a self offset calibration. The device internally shorts the inputs to mid-supply
and performs the calibration. The OFC registers are updated after this operation. Calibration commands must be
issued in conversion mode.
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9.5.3.10 RDATA
Read conversion data from the device with the RDATA command at any time. Figure 73 shows the read data
sequence. The MSB of the conversion data is output on the first SCLK rising edge after the command. An
RDATA command must be sent for each read operation. The ADC does not respond to commands until the read
operation is complete, or terminated by taking CS high. New data indicated by the DRDY and DOUT/DRDY will
not corrupt a read of conversion data with RDATA.
After all bytes are read, the data-byte sequence is repeated by continuing SCLK.
CS
(1)
9
1
17
SCLK
DIN
DOUT/DRDY
RDATA
HI-Z
(2)
'RQ¶W &DUH
Data 1
Data 2
ADC Data Bytes
(1)
CS can be tied low. If CS is low, DOUT/DRDY asserts low with DRDY.
(2)
DOUT/DRDY is driven low with DRDY. If a read operation occurs after the DRDY falling edge, then DOUT/DRDY can
be high or low.
Figure 73. Read Data Sequence
9.5.3.11 RREG
Use the RREG command to read the device register data. Read the register data one register at a time, or read
a block of register data. The starting register address can be any register in the register map. The RREG
command consists of two bytes. The first byte specifies the starting register address: 001r rrrr, where r rrrr is the
starting register address. The second command byte is the number of registers to read (minus 1): 000n nnnn,
where n nnnn is the number of registers to read minus 1.
After the read command is sent, the ADC responds with one or more register data bytes, most significant bit first.
If the byte count exceeds the last register address, the ADC begins to output zero data. During the register read
operation, any conversion data that becomes available is not loaded to the output shift register to avoid data
contention. However, the conversion data can be retrieved later by the RDATA command. After the register read
command has started, further commands are blocked until one of the following conditions are met:
• The read operation is completed
• The read operation is terminated by taking CS high
• The read operation is terminated by a serial interface timeout
• The ADC is reset by toggling the RESET pin
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Figure 74 shows a two-register read operation example. As shown, the commands required to read data from
two registers starting at register REF (address = 05h) are: command byte 1 = 25h and command byte 2 = 01h.
Keep DIN low after the two command bytes are sent.
(1)
CS
1
9
17
25
SCLK
'21¶7 &$5(
DOUT/DRDY
DIN
(1)
'21¶7 &$5(
0010 0101
REG DATA 1
REG DATA 2
0000 0001
CS can be set high or kept low between commands. If kept low, the command must be completed.
Figure 74. Read Register Sequence
9.5.3.12 WREG
Use the WREG command to write the device register data. The register data are written one register at a time or
as a block of register data. The starting register address is any register in the register map.
The WREG command consists of two bytes. The first byte specifies the starting register address: 010r rrrr, where
r rrrr is the starting register address The second command byte is the number of registers to write (minus 1):
000n nnnn, where n nnnn is the number of registers to write minus 1. The following byte (or bytes) is the register
data, most significant bit first. If the byte count exceeds the last register address, the ADC ignores the data. After
the register write command has started, further commands are blocked until one of the following conditions are
met:
• The write operation is completed
• The write operation is terminated by taking CS high
• The write operation is terminated by a serial interface timeout
• The ADC is reset by toggling the RESET pin
Figure 75 shows a two-register write operation example. As shown, the required commands to write data to two
registers starting at register REF (address = 05h) are: command byte 1 = 45h and command byte 2 = 01h.
(1)
CS
1
9
17
25
SCLK
DOUT/DRDY
'21¶7 &$5(
DIN
0100 0101
(1)
'21¶7 &$5(
0000 0001
'21¶7 &$5(
REG DATA 1
'21¶7 &$5(
REG DATA 2
CS can be set high or kept low between commands. If kept low, the command must be completed.
Figure 75. Write Register Sequence
52
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Writing new data to certain configuration registers resets the digital filter and starts a new conversion if a
conversion is in progress. Writing to the following registers triggers a new conversion:
• Channel configuration register (02h)
• Gain setting register (03h)
• Data rate register (04h)
• Reference control register (05h), bits [5:0]
• Excitation current register 1 (06h), bits [3:0]
• Excitation current register 2 (07h)
• System control register (09h), bits [7:5]
When the device is configured with WREG, the first data ready indication occurs after the new conversion
completes with the configuration settings. The previous conversion data are cleared at restart; therefore read the
previous data before the register write operation. Again, a WREG to these registers only starts a new conversion
if a conversion is in progress. If the device is in standby mode, the device sets the configuration according to the
WREG data, but does not start a conversion until the START/SYNC pin is taken high or a START command is
issued.
9.5.4 Interfacing with Multiple Devices
When connecting multiple devices to a single SPI bus, SCLK, DIN, and DOUT/DRDY can be safely shared by
using a dedicated chip-select (CS) line for each SPI-enabled device. When CS transitions high for the respective
device, DOUT/DRDY enters a tri-state mode. Therefore, DOUT/DRDY cannot be used to indicate when new data
are available if CS is high. Only the dedicated DRDY pin indicates that new data are available because the
DRDY pin is actively driven even when CS is high.
In some cases, the DRDY pin cannot be interfaced to the microcontroller. This scenario can occur if there are
insufficient GPIO channels available on the microcontroller or if the serial interface must be galvanically isolated
and thus the amount of channels must be limited. In order to evaluate when a new conversion of one of the
devices is ready, the microcontroller can periodically drop CS to the respective device and poll the state of the
DOUT/DRDY pin.
When CS goes low, the DOUT/DRDY pin immediately drives either high or low. If the DOUT/DRDY line drives
low, new data are available. If the DOUT/DRDY line drives high, no new data are available. This procedure
requires that DOUT/DRDY is forced high after reading each conversion result and before taking CS high. To
make sure DOUT/DRDY is taken high, send a RREG command to read a register where the least significant bit
is 1.
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9.6 Register Map
9.6.1 Configuration Registers
The ADS114S0xB register map consists of 18, 8-bit registers. These registers are used to configure and control
the device to the desired mode of operation. Access the registers through the serial interface by using the RREG
and WREG register commands. After power-on or reset, the registers default to the initial settings, as shown in
the Default column of Table 14.
Data can be written as a block to multiple registers using a single WREG command. If data are written as a
block, the data of certain registers take effect immediately when data are shifted in. Writing new data to certain
registers results in a restart of conversions that are in progress. The registers that result in a conversion restart
are discussed in the WREG section.
Table 14. Configuration Register Map
ADDR
REGISTER
DEFAULT
00h
ID
xxh
BIT 7
BIT 6
BIT 5
BIT 4
BIT 3
01h
STATUS
80h
02h
INPMUX
01h
03h
PGA
00h
0
0
0
04h
DATARATE
14h
0
CLK
MODE
1
05h
REF
10h
0
0
REFP_BUF
REFN_BUF
06h
IDACMAG
00h
0
0
0
0
07h
IDACMUX
FFh
08h
VBIAS
00h
0
VB_AINC
VB_AIN5
VB_AIN4
09h
SYS
10h
0Ah
RESERVED
00h
FL_POR
RDY
0
BIT 1
BIT 0
DEV_ID[2:0]
0
0
0
MUXP[3:0]
0
0
MUXN[3:0]
PGA_EN[1:0]
GAIN[2:0]
DR[3:0]
REFSEL[1:0]
REFCON[1:0]
IMAG[3:0]
I2MUX[3:0]
I1MUX[3:0]
SYS_MON[2:0]
VB_AIN3
CAL_SAMP[1:0]
VB_AIN2
VB_AIN1
VB_AIN0
TIMEOUT
0
0
RESERVED
0Bh
OFCAL0
00h
OFC[7:0]
0Ch
OFCAL1
00h
OFC[15:8]
0Dh
RESERVED
00h
RESERVED
0Eh
FSCAL0
00h
FSC[7:0]
0Fh
FSCAL1
40h
FSC[15:8]
10h
GPIODAT
00h
11h
GPIOCON
00h
54
BIT 2
RESERVED
DIR[3:0]
0
0
DAT[3:0]
0
0
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9.6.2 Register Descriptions
Table 15 lists the access codes for the ADS114S0xB registers.
Table 15. ADS114S0xB Access Type Codes
Access Type
Code
Description
R
R
Read
R-W
R/W
Read or Write
W
W
Write
-n
Value after reset or the default value
9.6.2.1 Device ID Register (address = 00h) [reset = xxh]
Figure 76. Device ID (ID) Register
7
6
5
RESERVED
R-xxh
4
3
2
1
DEV_ID[2:0]
R-xh
0
1
0
R-0h
0
0
R-0h
Table 16. Device ID (ID) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:3
RESERVED
R
xxh
Reserved
2:0
DEV_ID[2:0]
R
xh
Values are subject to change without notice
Device identifier.
Identifies the model of the device.
000 : Reserved
001 : Reserved
010 : Reserved
011 : Reserved
100 : ADS114S08B (12 channels, 16 bits)
101 : ADS114S06B (6 channels, 16 bits)
110 : Reserved
111 : Reserved
9.6.2.2 Device Status Register (address = 01h) [reset = 80h]
Figure 77. Device Status (STATUS) Register
7
FL_POR
R/W-1h
6
RDY
R-0h
5
0
R-0h
4
0
R-0h
3
0
R-0h
2
0
R-0h
Table 17. Device Status (STATUS) Register Field Descriptions
Bit
7
Field
Type
Reset
Description
FL_POR
R/W
1h
POR flag.
Indicates a power-on reset (POR) event has occurred.
0 : Register has been cleared and no POR event has occurred
1 : POR event occurred and has not been cleared. Flag must be cleared by
user register write (default).
6
RDY
R
0h
Device ready flag.
Indicates the device has started up and is ready for communication.
0 : ADC ready for communication (default)
1 : ADC not ready
5:0
RESERVED
R
00h
Reserved
Always write 00h
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9.6.2.3 Input Multiplexer Register (address = 02h) [reset = 01h]
Figure 78. Input Multiplexer (INPMUX) Register
7
6
5
4
3
MUXP[3:0]
R/W-0h
2
1
0
MUXN[3:0]
R/W-1h
Table 18. Input Multiplexer (INPMUX) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:4
MUXP[3:0]
R/W
0h
Positive ADC input selection.
Selects the ADC positive input channel.
0000 : AIN0 (default)
0001 : AIN1
0010 : AIN2
0011 : AIN3
0100 : AIN4
0101 : AIN5
0110 : AIN6 (ADS114S08B only)
0111 : AIN7 (ADS114S08B only)
1000 : AIN8 (ADS114S08B only)
1001 : AIN9 (ADS114S08B only)
1010 : AIN10 (ADS114S08B only)
1011 : AIN11 (ADS114S08B only)
1100 : AINCOM
1101 : Reserved
1110 : Reserved
1111 : Reserved
3:0
MUXN[3:0]
R/W
1h
Negative ADC input selection.
Selects the ADC negative input channel.
0000 : AIN0
0001 : AIN1 (default)
0010 : AIN2
0011 : AIN3
0100 : AIN4
0101 : AIN5
0110 : AIN6 (ADS114S08B only)
0111 : AIN7 (ADS114S08B only)
1000 : AIN8 (ADS114S08B only)
1001 : AIN9 (ADS114S08B only)
1010 : AIN10 (ADS114S08B only)
1011 : AIN11 (ADS114S08B only)
1100 : AINCOM
1101 : Reserved
1110 : Reserved
1111 : Reserved
56
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9.6.2.4 Gain Setting Register (address = 03h) [reset = 00h]
Figure 79. Gain Setting (PGA) Register
7
0
R/W-0h
6
0
R/W-0h
5
0
R/W-0h
4
3
2
1
GAIN[2:0]
R/W-0h
PGA_EN[1:0]
R/W-0h
0
Table 19. Gain Setting (PGA) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:5
RESERVED
R/W
0h
Reserved
4:3
PGA_EN[1:0]
R/W
0h
Always write 0h
PGA enable.
Enables or bypasses the PGA.
00 : PGA is powered down and bypassed. Enables single-ended
measurements with unipolar supply (Set gain = 1 (1)) (default)
01 : PGA enabled (gain = 1 to 128)
10 : Reserved
11 : Reserved
2:0
GAIN[2:0]
R/W
0h
PGA gain selection.
Configures the PGA gain.
000 : 1 (default)
001 : 2
010 : 4
011 : 8
100 : 16
101 : 32
110 : 64
111 : 128
(1)
When bypassing the PGA, the user must also set GAIN[2:0] to 000.
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9.6.2.5 Data Rate Register (address = 04h) [reset = 14h]
Figure 80. Data Rate (DATARATE) Register
7
0
R/W-0h
6
CLK
R/W-0h
5
MODE
R/W-0h
4
1
R/W-1h
3
2
1
0
DR[3:0]
R/W-4h
Table 20. Data Rate (DATARATE) Register Field Descriptions
Bit
Field
Type
Reset
Description
7
RESERVED
R/W
0h
Reserved
6
CLK
R/W
0h
Always write 0h
Clock source selection.
Configures the clock source to use either the internal oscillator or an
external clock.
0 : Internal 4.096-MHz oscillator (default)
1 : External clock
5
MODE
R/W
0h
Conversion mode selection.
Configures the ADC for either continuous conversion or single-shot
conversion mode.
0 : Continuous conversion mode (default)
1 : Single-shot conversion mode
4
RESERVED
R/W
1h
DR[3:0]
R/W
4h
Reserved
Always write 1h
3:0
Data rate selection.
Configures the output data rate (1).
0000 : 2.5 SPS
0001 : 5 SPS
0010 : 10 SPS
0011 : 16.6 SPS
0100 : 20 SPS (default)
0101 : 50SPS
0110 : 60 SPS
0111 : 100 SPS
1000 : 200 SPS
1001 : 400 SPS
1010 : 800 SPS
1011 : 1000 SPS
1100 : 2000 SPS
1101 : 4000 SPS
1110 : 4000 SPS
1111 : Reserved
(1)
58
Data rates of 60 Hz or less can offer line-cycle rejection; see the 50-Hz and 60-Hz Line Cycle Rejection section for more information.
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9.6.2.6 Reference Control Register (address = 05h) [reset = 10h]
Figure 81. Reference Control (REF) Register
7
0
R/W-0h
6
0
R/W-0h
5
REFP_BUF
R/W-0h
4
REFN_BUF
R/W-1h
3
2
1
REFSEL[1:0]
R/W-0h
0
REFCON[1:0]
R/W-0h
Table 21. Reference Control (REF) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:6
RESERVED
R/W
0h
Reserved
5
REFP_BUF
R/W
0h
Always write 0h
Positive reference buffer bypass.
Disables the positive reference buffer. Recommended when V(REFPx) is
close to AVDD.
0 : Enabled (default)
1 : Disabled
4
REFN_BUF
R/W
1h
Negative reference buffer bypass.
Disables the negative reference buffer. Recommended when V(REFNx) is
close to AVSS.
0 : Enabled
1 : Disabled (default)
3:2
REFSEL[1:0]
R/W
0h
Reference input selection.
Selects the reference input source for the ADC.
00 : REFP0, REFN0 (default)
01 : REFP1, REFN1
10 : Internal 2.5-V reference (1)
11 : Reserved
1:0
REFCON[1:0]
R/W
0h
Internal voltage reference configuration (2).
Configures the behavior of the internal voltage reference.
00 : Internal reference off (default)
01 : Internal reference on, but powers down in power-down mode
10 : Internal reference is always on, even in power-down mode
11 : Reserved
(1)
(2)
Disable the reference buffers when the internal reference is selected for measurements.
The internal voltage reference must be turned on to use the IDACs.
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9.6.2.7 Excitation Current Register 1 (address = 06h) [reset = 00h]
Figure 82. Excitation Current Register 1 (IDACMAG)
7
0
R/W-0h
6
0
R/W-0h
5
0
R/W-0h
4
0
R/W-0h
3
2
1
0
IMAG[3:0]
R/W-0h
Table 22. Excitation Current Register 1 (IDACMAG) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:4
RESERVED
R/W
0h
Reserved
3:0
IMAG[3:0]
R/W
0h
Always write 0h
IDAC magnitude selection.
Selects the value of the excitation current sources. Sets IDAC1 and IDAC2
to the same value.
0000 : Off (default)
0001 : 10 µA
0010 : 50 µA
0011 : 100 µA
0100 : 250 µA
0101 : 500 µA
0110 : 750 µA
0111 : 1000 µA
1000 : 1500 µA
1001 : 2000 µA
1010–1111 : Off
60
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9.6.2.8 Excitation Current Register 2 (address = 07h) [reset = FFh]
Figure 83. Excitation Current Register 2 (IDACMUX)
7
6
5
4
3
2
I2MUX[3:0]
R/W-Fh
1
0
I1MUX[3:0]
R/W-Fh
Table 23. Excitation Current Register 2 (IDACMUX) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:4
I2MUX[3:0]
R/W
Fh
IDAC2 output channel selection.
Selects the output channel for IDAC2.
0000 : AIN0
0001 : AIN1
0010 : AIN2
0011 : AIN3
0100 : AIN4
0101 : AIN5
0110 : AIN6 (ADS114S08B), REFP1 (ADS114S06B)
0111 : AIN7 (ADS114S08B), REFN1 (ADS114S06B)
1000 : AIN8 (ADS114S08B only)
1001 : AIN9 (ADS114S08B only)
1010 : AIN10 (ADS114S08B only)
1011 : AIN11 (ADS114S08B only)
1100 : AINCOM
1101–1111 : Disconnected (default)
3:0
I1MUX[3:0]
R/W
Fh
IDAC1 output channel selection.
Selects the output channel for IDAC1.
0000 : AIN0
0001 : AIN1
0010 : AIN2
0011 : AIN3
0100 : AIN4
0101 : AIN5
0110 : AIN6 (ADS114S08B only), REFP1 (ADS114S06B)
0111 : AIN7 (ADS114S08B only), REFN1 (ADS114S06B)
1000 : AIN8 (ADS114S08B only)
1001 : AIN9 (ADS114S08B only)
1010 : AIN10 (ADS114S08B only)
1011 : AIN11 (ADS114S08B only)
1100 : AINCOM
1101–1111 : Disconnected (default)
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9.6.2.9 Sensor Biasing Register (address = 08h) [reset = 00h]
Figure 84. Sensor Biasing (VBIAS) Register
7
0
R/W-0h
6
VB_AINC
R/W-0h
5
VB_AIN5
R/W-0h
4
VB_AIN4
R/W-0h
3
VB_AIN3
R/W-0h
2
VB_AIN2
R/W-0h
1
VB_AIN1
R/W-0h
0
VB_AIN0
R/W-0h
Table 24. Sensor Biasing (VBIAS) Register Field Descriptions
Bit
Field
Type
Reset
Description
7
RESERVED
R/W
0h
Reserved
6
VB_AINC
R/W
0h
Always write 0h
AINCOM VBIAS selection (1).
Enables VBIAS on the AINCOM pin.
0 : VBIAS disconnected from AINCOM (default)
1 : VBIAS connected to AINCOM
5
VB_AIN5
R/W
0h
AIN5 VBIAS selection (1).
Enables VBIAS on the AIN5 pin.
0 : VBIAS disconnected from AIN5 (default)
1 : VBIAS connected to AIN5
4
VB_AIN4
R/W
0h
AIN4 VBIAS selection (1).
Enables VBIAS on the AIN4 pin.
0 : VBIAS disconnected from AIN4 (default)
1 : VBIAS connected to AIN4
3
VB_AIN3
R/W
0h
AIN3 VBIAS selection (1).
Enables VBIAS on the AIN3 pin.
0 : VBIAS disconnected from AIN3 (default)
1 : VBIAS connected to AIN3
2
VB_AIN2
R/W
0h
AIN2 VBIAS selection (1).
Enables VBIAS on the AIN2 pin.
0 : VBIAS disconnected from AIN2 (default)
1 : VBIAS connected to AIN2
1
VB_AIN1
R/W
0h
AIN1 VBIAS selection (1).
Enables VBIAS on the AIN1 pin.
0 : VBIAS disconnected from AIN1 (default)
1 : VBIAS connected to AIN1
0
VB_AIN0
R/W
0h
AIN0 VBIAS selection (1).
Enables VBIAS on the AIN0 pin.
0 : VBIAS disconnected from AIN0 (default)
1 : VBIAS connected to AIN0
(1)
62
The bias voltage can be selected for multiple analog inputs at the same time.
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9.6.2.10 System Control Register (address = 09h) [reset = 10h]
Figure 85. System Control (SYS) Register
7
6
SYS_MON[2:0]
R/W-0h
5
4
3
CAL_SAMP[1:0]
R/W-2h
2
TIMEOUT
R/W-0h
1
0
R/W-0h
0
0
R/W-0h
Table 25. System Control (SYS) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:5
SYS_MON[2:0]
R/W
0h
System monitor configuration (1).
Enables a set of system monitor measurements using the ADC.
000 : Disabled (default)
001 : PGA inputs shorted to (AVDD + AVSS) / 2 and disconnected from
AINx and the multiplexer; gain set by user
010 : Internal temperature sensor measurement; PGA must be enabled
(PGA_EN[1:0] = 01); gain set by user (2)
011 : (AVDD – AVSS) / 4 measurement; gain set to 1 (3)
100 : DVDD / 4 measurement; gain set to 1 (3)
101 : Burn-out current sources enabled, 0.2-µA setting
110 : Burn-out current sources enabled, 1-µA setting
111 : Burn-out current sources enabled, 10-µA setting
4:3
CAL_SAMP[1:0]
R/W
2h
Calibration sample size selection.
Configures the number of samples averaged for self and system offset and
system gain calibration.
00 : 1 sample
01 : 4 samples
10 : 8 samples (default)
11 : 16 samples
2
TIMEOUT
R/W
0h
SPI timeout enable.
Enables the SPI timeout function.
0 : Disabled (default)
1 : Enabled
1:0
RESERVED
R/W
0h
Reserved
Always write 0h
(1)
(2)
(3)
With system monitor functions enabled, the AINx multiplexer switches are open for the (AVDD + AVSS) / 2 measurement, the
temperature sensor, and the supply monitors.
When using the internal temperature sensor, gain must be 4 or less to keep the measurement within the PGA input voltage range.
The PGA gain is automatically set to 1 when the supply monitors are enabled, regardless of the setting in GAIN[2:0].
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9.6.2.11 Reserved Register (address = 0Ah) [reset = 00h]
Figure 86. Reserved Register
7
6
5
4
3
2
1
0
1
0
RESERVED
R-00h
Table 26. Reserved Register Field Descriptions
Bit
Field
Type
Reset
7:0
RESERVED
R
00h
Description
Reserved
Always write 00h
9.6.2.12 Offset Calibration Register 1 (address = 0Bh) [reset = 00h]
Figure 87. Offset Calibration Register 1 (OFCAL0)
7
6
5
4
3
2
OFC[7:0]
R/W-00h
Table 27. Offset Calibration Register 1 (OFCAL0) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:0
OFC[7:0]
R/W
00h
Bits [7:0] of the offset calibration value.
9.6.2.13 Offset Calibration Register 2 (address = 0Ch) [reset = 00h]
Figure 88. Offset Calibration Register 2 (OFCAL1)
7
6
5
4
3
2
1
0
OFC[15:8]
R/W-00h
Table 28. Offset Calibration Register 2 (OFCAL1) Register Field Descriptions
64
Bit
Field
Type
Reset
Description
7:0
OFC[15:8]
R/W
00h
Bits [15:8] of the offset calibration value.
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9.6.2.14 Reserved Register (address = 0Dh) [reset = 00h]
Figure 89. Reserved Register
7
6
5
4
3
2
1
0
1
0
RESERVED
R-00h
Table 29. Reserved Register Field Descriptions
Bit
Field
Type
Reset
7:0
RESERVED
R
00h
Description
Reserved
Always write 00h
9.6.2.15 Gain Calibration Register 1 (address = 0Eh) [reset = 00h]
Figure 90. Gain Calibration Register 1 (FSCAL0)
7
6
5
4
3
2
FSC[7:0]
R/W-00h
Table 30. Gain Calibration Register 1 (FSCAL0) Field Descriptions
Bit
Field
Type
Reset
Description
7:0
FSC[7:0]
R/W
00h
Bits [7:0] of the gain calibration value.
9.6.2.16 Gain Calibration Register 2 (address = 0Fh) [reset = 40h]
Figure 91. Gain Calibration Register 2 (FSCAL1)
7
6
5
4
3
2
1
0
FSC[15:8]
R/W-40h
Table 31. Gain Calibration Register 2 (FSCAL1) Field Descriptions
Bit
Field
Type
Reset
Description
7:0
FSC[15:8]
R/W
40h
Bits [15:8] of the gain calibration value.
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9.6.2.17 GPIO Data Register (address = 10h) [reset = 00h]
Figure 92. GPIO Data (GPIODAT) Register
7
6
5
4
3
2
DIR[3:0]
R/W-0h
1
0
DAT[3:0]
R/W-0h
Table 32. GPIO Data (GPIODAT) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:4
DIR[3:0]
R/W
0h
GPIO direction.
Configures the selected GPIO as an input or output.
0 : GPIO[x] configured as output (default)
1 : GPIO[x] configured as input
3:0
DAT[3:0]
R/W
0h
GPIO data.
Contains the data of the GPIO inputs or outputs.
0 : GPIO[x] is low (default)
1 : GPIO[x] is high
9.6.2.18 GPIO Configuration Register (address = 11h) [reset = 00h]
Figure 93. GPIO Configuration Register
7
0
R-0h
6
0
R-0h
5
0
R-0h
4
0
R-0h
3
2
1
0
CON[3:0]
R/W-0h
Table 33. GPIO Configuration (GPIOCON) Register Field Descriptions
Bit
Field
Type
Reset
Description
7:4
RESERVED
R
0h
Reserved
3:0
CON[3:0]
R/W
0h
Always write 0h
GPIO pin configuration.
Configures the GPIO[x] pin as an analog input or GPIO. CON[x]
corresponds to the GPIO[x] pin.
0 : GPIO[x] configured as analog input (default) (1)
1 : GPIO[x] configured as GPIO
(1)
66
On the ADS114S06B, the GPIO pins default as disabled. Set the CON[3:0] bits to enable the respective GPIO pins.
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10 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
10.1 Application Information
The ADS114S06B and ADS114S08B are precision, 16-bit, ΔΣ ADCs that offer many integrated features to
simplify the measurement of the most common sensor types (including various types of temperature, flow, and
bridge sensors). Primary considerations when designing an application with the ADS114S0xB include analog
input filtering, establishing an appropriate reference, and setting the absolute input voltage for the internal PGA.
Connecting and configuring the serial interface appropriately is another concern. These considerations are
discussed in the following sections.
10.1.1 Serial Interface Connections
The principle serial interface connections for the ADS114S0xB are shown in Figure 94.
21
20
19
18
CLK
GPIO3/AIN11
22
RESET
GPIO2/AIN10
23
GPIO1/AIN9
24
GPIO0/AIN8
REFOUT
REFCOM
1 PF
17
3.3 V
47
NC
5V
25
DVDD
GPIO
16
47
26
AVDD
IOVDD
15
GPIO/IRQ
0.1 PF
330 nF
47
27
DGND
AVSS
14
MISO
47
28
DRDY
AVSS
13
SCLK
47
REFN0
29
DOUT/DRDY
12
REFP0
30
SCLK
11
MOSI
Microcontroller
with SPI
47
GPIO
47
REFN1/AIN7
REFP1/AIN6
31
DIN
32
CS
10
9
GPIO
3.3 V
DVDD
5
AIN5
AIN4
AIN3
AIN2
6
7
8
0.1 PF
START/SYNC
4
AIN0
3
AIN1
2
AINCOM
DVSS
1
Figure 94. Serial Interface Connections
Most microcontroller SPI peripherals can interface with the ADS114S0xB. The interface operates in SPI mode 1
where CPOL = 0 and CPHA = 1. In SPI mode 1, SCLK idles low and data are launched or changed only on
SCLK rising edges; data are latched or read by the master and slave on SCLK falling edges. Details of the SPI
communication protocol employed by the devices are found in the Serial Interface section.
Place 47-Ω resistors in series with all digital input and output pins (CS, SCLK, DIN, DOUT/DRDY, and DRDY).
This resistance smooths sharp transitions, suppresses overshoot, and offers some overvoltage protection. Care
must be taken to meet all SPI timing requirements because the additional resistors interact with the bus
capacitances present on the digital signal lines.
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Application Information (continued)
10.1.2 Analog Input Filtering
Analog input filtering serves two purposes: first, to limit the effect of aliasing during the sampling process and
second, to reduce external noise from being a part of the measurement.
As with any sampled system, aliasing can occur if proper antialias filtering is not in place. Aliasing occurs when
frequency components are present in the input signal that are higher than half the sampling frequency of the
ADC (also known as the Nyquist frequency). These frequency components are folded back and show up in the
actual frequency band of interest below half the sampling frequency. Inside a ΔΣ ADC, the input signal is
oversampled at the modulator frequency, fMOD and not at the output data rate. The filter response of the digital
filter repeats at multiples of fMOD, as shown in Figure 95. Signals or noise up to a frequency where the filter
response repeats are attenuated to a certain amount by the digital filter depending on the filter architecture. Any
frequency components present in the input signal around the modulator frequency or multiples thereof are not
attenuated and alias back into the band of interest, unless attenuated by an external analog filter.
Magnitude
Sensor
Signal
Unwanted
Signals
Unwanted
Signals
Output
Data Rate
fMOD/2
fMOD
Frequency
fMOD
Frequency
fMOD
Frequency
Magnitude
Digital Filter
Aliasing of
Unwanted Signals
Output
Data Rate
fMOD/2
Magnitude
External
Antialiasing Filter
Roll-Off
Output
Data Rate
fMOD/2
Figure 95. Effect of Aliasing
68
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Application Information (continued)
Many sensor signals are inherently band limited; for example, the output of a thermocouple has a limited rate of
change. In this case, the sensor signal does not alias back into the pass band when using a ΔΣ ADC. However,
any noise pick-up along the sensor wiring or the application circuitry can potentially alias into the pass band.
Power line-cycle frequency and harmonics are one common noise source. External noise can also be generated
from electromagnetic interference (EMI) or radio frequency interference (RFI) sources, such as nearby motors
and cellular phones. Another noise source typically exists on the printed circuit board (PCB) itself in the form of
clocks and other digital signals. Analog input filtering helps remove unwanted signals from affecting the
measurement result.
A first-order resistor-capacitor (RC) filter is (in most cases) sufficient to either eliminate aliasing, or to reduce the
effect of aliasing to a level below the noise floor of the sensor. Ideally, any signal beyond fMOD / 2 is attenuated to
a level below the noise floor of the ADC. The digital filter of the ADS114S0xB attenuates signals to a certain
degree, as illustrated in the filter response plots in the Digital Filter section. In addition, noise components are
usually smaller in magnitude than the actual sensor signal. Therefore, using a first-order RC filter with a cutoff
frequency set at the output data rate or 10 times higher is generally a good starting point for a system design.
Internal to the device, prior to the PGA inputs, is an EMI filter; see Figure 43. The cutoff frequency of this filter is
approximately 40 MHz and helps reject high-frequency interference.
10.1.3 External Reference and Ratiometric Measurements
The full-scale range of the ADS114S0xB is defined by the reference voltage and the PGA gain
(FSR = ±VREF / Gain). An external reference can be used instead of the integrated 2.5-V reference to adapt the
FSR to the specific system needs. An external reference must be used if VIN > 2.5 V. For example, an external
5-V reference and an AVDD = 5 V are required in order to measure a single-ended signal that can swing
between 0 V and 5 V.
The reference inputs of the device also allow the implementation of ratiometric measurements. In a ratiometric
measurement, the same excitation source that is used to excite the sensor is also used to establish the reference
for the ADC. As an example, a simple form of a ratiometric measurement uses the same current source to excite
both the resistive sensor element (such as an RTD) and another resistive reference element that is in series with
the element being measured. The voltage that develops across the reference element is used as the reference
source for the ADC. Because current noise and drift are common to both the sensor measurement and the
reference, these components cancel out in the ADC transfer function. The output code is only a ratio of the
sensor element and the value of the reference resistor. The value of the excitation current source itself is not part
of the ADC transfer function.
The example in the Typical Application section describes a system that uses a ratiometric measurement. One
excitation current source is used to drive a reference resistor and an RTD. The ADC measurement represents a
ratiometric measurement between the RTD value and a known reference resistor value.
10.1.4 Establishing a Proper Input Voltage
The ADS114S0xB can be used to measure various types of input signal configurations: single-ended, pseudodifferential, and fully-differential signals (which can be either unipolar or bipolar). However, configuring the device
properly for the respective signal type is important.
Signals where the negative analog input is fixed and referenced to analog ground (VAINN = 0 V) are commonly
called single-ended signals. The input voltage of a single-ended signal consequently varies between 0 V and VIN.
If the PGA is disabled and bypassed, the input voltage of the ADS114S0xB can be as low as 50 mV below AVSS
and as large as 50 mV above AVDD. Therefore, set the PGA_EN bits to 10 in the gain setting register (03h) to
measure single-ended signals when a unipolar analog supply is used (AVSS = 0 V). Only a gain of 1 is possible
in this configuration. Measuring a 0-mA to 20-mA or 4-mA to 20-mA signal across a load resistor of 100 Ω
referenced to GND is a typical example. The ADS114S0xB can directly measure the signal across the load
resistor using a unipolar supply, the internal 2.5-V reference, and gain = 1 when the PGA is bypassed.
If gain is needed to measure a single-ended signal, the PGA must be enabled. In this case, a bipolar supply is
required for the ADS114S0xB to meet the input voltage requirement of the PGA. Signals where the negative
analog input (AINN) is fixed at a voltage other the 0 V are referred to as pseudo-differential signals. The input
voltage of a pseudo-differential signal varies between VAINN and VAINN + VIN.
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Application Information (continued)
Fully-differential signals in contrast are defined as signals having a constant common-mode voltage where the
positive and negative analog inputs swing 180° out-of-phase but have the same amplitude.
The ADS114S0xB can measure pseudo-differential and fully-differential signals both with the PGA enabled or
bypassed. However, the PGA must be enabled in order to measure any input with a gain greater than 1. The
input voltage must meet the input and output voltage restrictions of the PGA, as explained in the PGA InputVoltage Requirements section when the PGA is enabled. Setting the input voltage at or near (AVSS + AVDD) / 2
in most cases satisfies the PGA input voltage requirements.
Signals where both the positive and negative inputs are always ≥ 0 V are called unipolar signals. These signals
can in general be measured with the ADS114S0xB using a unipolar analog supply (AVSS = 0 V). As mentioned
previously, the PGA must be bypassed in order to measure single-ended, unipolar signals when using a unipolar
supply.
A signal is called bipolar when either the positive or negative input can swing below 0 V. A bipolar analog supply
(such as AVDD = 2.5 V, AVSS = –2.5 V) is required in order to measure bipolar signals with the ADS114S0xB. A
typical application task is measuring a single-ended, bipolar, ±10-V signal where AINN is fixed at 0 V and AINP
swings between –10 V and 10 V. The ADS114S0xB cannot directly measure this signal because the 10-V signal
exceeds the analog power-supply limits. However, one possible solution is to use a bipolar analog supply (AVDD
= 2.5 V, AVSS = –2.5 V), gain = 1, and a resistor divider in front of the ADS114S0xB. The resistor divider must
divide the voltage down to ≤ ±2.5 V to be able to measure the voltage using the internal 2.5-V reference.
10.1.5 Unused Inputs and Outputs
To minimize leakage currents on the analog inputs, leave unused analog and reference inputs floating, or
connect the inputs to mid-supply or to AVDD. Connecting unused analog or reference inputs to AVSS is possible
as well, but can yield higher leakage currents than the previously mentioned options. REFN0 is an exception;
leave the REFN0 pin floating when not in use or tie the pin to AVSS.
GPIO pins operate on levels based on the analog supply. Do not float GPIO pins that are configured as digital
inputs. Tie unused GPIO pins that are configured as digital inputs to the appropriate levels, AVDD or AVSS,
including when in power-down mode. Tie unused GPIO output pins to AVSS through a pulldown resistor and set
the output to 0 in the GPIO data register. For unused GPIO pins on the ADS114S06B, leave the GPIOCON
register set to the default register values and connect these GPIO pins in the same manner as for an unused
analog input.
Do not float unused digital inputs; excessive power-supply leakage current can result. Tie all unused digital
inputs to the appropriate levels, IOVDD or DGND, even when in power-down mode. Connections for unused
digital inputs are listed below.
• Tie the CS pin to DGND if CS is not used
• Tie the CLK pin to DGND if the internal oscillator is used
• Tie the START/SYNC pin to DGND to control conversions by commands
• Tie the RESET pin to IOVDD if the RESET pin is not used
• If the DRDY output is not used, leave the DRDY pin unconnected or tie the DRDY pin to IOVDD using a weak
pullup resistor
70
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Application Information (continued)
10.1.6 Pseudo Code Example
The following list shows a pseudo code sequence with the required steps to set up the device and the
microcontroller that interfaces to the ADC in order to take subsequent readings from the ADS114S0xB in
continuous conversion mode. The dedicated DRDY pin is used to indicate availability of new conversion data.
Power-up so that all supplies reach minimum operating levels;
Delay for a minimum of 2.2 ms to allow power supplies to settle and power-up reset to complete;
Configure the SPI interface of the microcontroller to SPI mode 1 (CPOL = 0, CPHA =1);
If the CS pin is not tied low permanently, configure the microcontroller GPIO connected to CS as an
output;
Configure the microcontroller GPIO connected to the DRDY pin as a falling edge triggered interrupt
input;
Set CS to the device low;
Delay for a minimum of td(CSSC);
Send the RESET command (06h) to make sure the device is properly reset after power-up; //Optional
Delay for a minimum of 4096 · tCLK;
Read the status register using the RREG command to check that the RDY bit is 0; //Optional
Clear the FL_POR flag by writing 00h to the status register; //Optional
Write the respective register configuration with the WREG command;
For verification, read back all configuration registers with the RREG command;
Send the START command (08h) to start converting in continuous conversion mode;
Delay for a minimum of td(SCCS);
Clear CS to high (resets the serial interface);
Loop
{
Wait for DRDY to transition low;
Take CS low;
Delay for a minimum of td(CSSC);
Send the RDATA command;
Send 16 SCLK rising edges to read out conversion data on DOUT/DRDY;
Delay for a minimum of td(SCCS);
Clear CS to high;
}
Take CS low;
Delay for a minimum of td(CSSC);
Send the STOP command (0Ah) to stop conversions and put the device in standby mode;
Delay for a minimum of td(SCCS);
Clear CS to high;
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10.2 Typical Application
Figure 96 shows a fault-protected, filtered, 3-wire RTD application circuit with hardware-based, lead-wire
compensation. Two IDAC current sources provide the lead-wire compensation. One IDAC current source
(IDAC1) provides excitation to the RTD element. The ADC reference voltage (pins AIN6 and AIN7) is derived
from the voltage across resistor RREF sourcing the same IDAC1 current, providing ratiometric cancellation of
current-source drift. The other current source (IDAC2) has the same current setting, providing cancellation of
lead-wire resistance by generating a voltage drop across lead-wire resistance RLEAD2 equal to the voltage drop of
RLEAD1. Because the RRTD voltage is measured differentially at ADC pins AIN1 and AIN2, the voltages across the
lead wire resistance cancel. Resistor RBIAS level-shifts the RTD signal to within the ADC specified input range.
The current sources are provided by two additional pins (AIN5 and AIN3) that connect to the RTD through
blocking diodes. The additional pins are used to route the RTD excitation currents around the input filter
resistors, avoiding the voltage drop otherwise caused by the filter resistors RF1 and RF4. The diodes protect the
ADC inputs in the event of a miswired connection. The input filter resistors limit the input fault currents flowing
into the ADC.
3.3 V
5V
0.1 PF
330 nF
IIDAC1
RF4
AVDD
IDAC1
AIN5
(IDAC1)
AVDD
RF3
IOVDD
ADS114S08B
500 A
CCM4
AIN6
REFOUT
(REFP1)
RREF
DVDD
2.5-V
Reference
Reference
Mux
CDIF2
(REFN1)
CCM3
REFCOM
Reference
Buffers
3-Wire RTD
RLEAD1
RF2
CCM2
AIN1
(AINP)
CDIF1
RRTD
RLEAD2
RF1
Input
Mux
16-Bit
û ADC
PGA
AIN2
Digital
Filter
(AINN)
Serial
Interface
and
Control
CCM1
IIDAC2
AIN3
(IDAC2)
IDAC2
START/SYNC
RESET
CS
DIN
DOUT/DRDY
SCLK
DRDY
AVDD
4.096-MHz
Oscillator
500 A
AVSS
RLEAD3
1 PF
AIN7
DGND
CLK
IIDAC1 + IIDAC2
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RBIAS
Figure 96. 3-Wire RTD Application
10.2.1 Design Requirements
Table 34 shows the design requirements of the 3-wire RTD application.
Table 34. Design Requirements
(1)
72
DESIGN PARAMETER
VALUE
ADC supply voltage
4.75 V (minimum)
RTD sensor type
3-wire Pt100
RTD resistance range
20 Ω to 400 Ω
RTD lead resistance range
0 Ω to 10 Ω
RTD self heating
1 mW
Accuracy (1)
±0.1 Ω
TA = 25°C. After offset and full-scale calibration.
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10.2.2 Detailed Design Procedure
The key considerations in the design of a 3-wire RTD circuit are the accuracy, the lead wire compensation, and
the sensor self-heating. As the design values of Table 35 show, several values of excitation currents are
available. The resolution is expressed in units of noise-free resolution (NFR). Noise-free resolution is resolution
with no code flicker. The selection of excitation currents trades off resolution against sensor self-heating. In
general, measurement resolution improves with increasing excitation current. Increasing the excitation current
beyond 1000 µA results in no further improvement in resolution for this example circuit. The design procedure is
based on a 500-µA excitation current, because this level of current results in very low sensor self-heating
(0.4 mW).
Table 35. RTD Circuit Design Parameters
IIDAC
(µA)
NFR
(bits)
PRTD
(mW)
VRTD
(V)
Gain
(V/V)
VREFMIN (1)
(V)
VREF (2)
(V)
RREF
(kΩ)
VAINNLIM (3)
(V)
VAINPLIM (4)
(V)
RBIAS
(kΩ)
VRTDN (5)
(V)
VRTDP (6)
(V)
VIDAC1 (7)
(V)
50
16.8
0.001
0.02
32
0.64
0.70
18
0.6
4.1
7.10
0.7
0.7
1.9
100
17.8
0.004
0.04
32
1.28
1.41
14.1
0.9
3.8
5.10
1.0
1.1
2.8
250
18.8
0.025
0.10
16
1.60
1.76
7.04
1.1
3.7
2.30
1.2
1.3
3.3
500
19.1
0.100
0.20
8
1.60
1.76
3.52
1.0
3.8
1.10
1.1
1.3
3.4
750
18.9
0.225
0.30
4
1.20
1.32
1.76
0.8
4.0
0.57
0.9
1.2
2.8
1000
19.3
0.400
0.40
4
1.60
1.76
1.76
0.9
3.9
0.50
1.0
1.4
3.5
1500
19.1
0.900
0.60
2
1.20
1.32
0.88
0.6
4.2
0.23
0.7
1.3
3.0
2000
18.3
1.600
0.80
1
0.80
0.90
0.45
0.3
4.5
0.10
0.4
1.2
2.4
(1)
(2)
(3)
(4)
(5)
(6)
(7)
VREFMIN is the minimum reference voltage required by the design.
VREF is the design target reference voltage allowing for 10% overrange.
VAINNLIM is the absolute minimum input voltage required by the ADC.
VAINPLIM is the absolute maximum input voltage required by the ADC.
VRTDN is the design target negative input voltage.
VRTDP is the design target positive input voltage.
VIDAC1 is the design target IDAC1 loop voltage.
Initially, RLEAD1 and RLEAD2 are considered to be 0 Ω. Route the IDAC1 current through the external reference
resistor, RREF. IDAC1 generates the ADC reference voltage, VREF, across the reference resistor. This voltage is
defined by Equation 12:
VREF = IIDAC1 · RREF
(12)
Route the second current (IDAC2) to the second RTD lead.
Program the IDAC value by using the IDACMAG register; however, only the IDAC1 current flows through the
reference resistor and RTD. The IDAC1 current excites the RTD to produce a voltage proportional to the RTD
resistance. The RTD voltage is defined by Equation 13:
VRTD = RRTD · IIDAC1
(13)
The ADC amplifies the RTD signal voltage (VRTD) and measures the resulting voltage against the reference
voltage to produce a proportional digital output code, as shown in Equation 14 through Equation 16.
Code ∝ VRTD · Gain / VREF
Code ∝ (RRTD · IIDAC1) · Gain / (IIDAC1 · RREF)
Code ∝ (RRTD · Gain) / RREF
(14)
(15)
(16)
As shown in Equation 16, the RTD measurement depends on the value of the RTD, the PGA gain, and the
reference resistor RREF, but not on the IDAC1 value. Therefore, the absolute accuracy and temperature drift of
the excitation current does not matter.
The second excitation current (IDAC2) provides a second voltage drop across the second RTD lead resistance,
RLEAD2. The second voltage drop compensates the voltage drop caused by IDAC1 and RLEAD1. The leads of a 3wire RTD typically have the same length; therefore, the lead resistance is typically identical. Taking the lead
resistance into account (RLEADx ≠ 0), the differential voltage (VIN) across ADC inputs AIN8 and AIN9 is shown in
Equation 17:
VIN = IIDAC1 · (RRTD + RLEAD1) – IIDAC2 · RLEAD2
(17)
If RLEAD1 = RLEAD2 and IIDAC1 = IIDAC2, the expression for VIN reduces to Equation 18:
VIN = IIDAC1 · RRTD
(18)
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In other words, the measurement error resulting from the voltage drop across the RTD lead resistance is
compensated as long as the lead resistance values and the IDAC values are matched.
Using Equation 13, the value of RTD resistance (400 Ω, maximum) and the excitation current (500 µA) yields an
RTD voltage of VRTD = 500 µA · 400 Ω = 0.2 V. Use the maximum gain of 8 in order to limit the corresponding
loop voltage of IDAC1. Gain = 8 requires a minimum reference voltage VREFMIN = 0.2 V · 8 = 1.6 V. To provide
margin for the ADC operating range, increase the target reference voltage by 10% (VREF = 1.6 V · 1.1 = 1.76 V).
Calculate the value of the reference resistor, as shown in Equation 19:
RREF = VREF / IIDAC1 = 1.76 V / 500 µA = 3.52 kΩ
(19)
For this example application, 3.5 kΩ is chosen for RREF. For best results, use a precision reference resistor RREF
with a low temperature drift (< 10 ppm/°C). Any change in RREF is reflected in the measurement as a gain error.
The next step in the design is determining the value of the RBIAS resistor, in order to level shift the RTD voltage to
meet the ADC absolute input-voltage specification. The required level-shift voltage is determined by calculating
the minimum absolute voltage (VAINNLIM) as shown in Equation 20:
AVSS + 0.15 + VRTDMAX · (Gain – 1) / 2 ≤ VAINNLIM
where
•
•
•
VRTDMAX = maximum differential RTD voltage = 0.2 V
Gain = 8
AVSS = 0 V
(20)
The result of the equation requires a minimum absolute input voltage (VRTDN) > 0.85 V. Therefore, the RTD
voltage must be level shifted by a minimum of 0.85 V. To meet this requirement, a target level-shift value of 1 V
is chosen to provide extra margin. Calculate the value of RBIAS as shown in Equation 21:
RBIAS= VAINN / (IIDAC1+ IIDAC2) = 1 V / ( 2 · 500 µA) = 1 kΩ
(21)
After the level-shift voltage is determined, verify that the positive RTD voltage (VRTDP) is less than the maximum
absolute input voltage (VAINPLIM), as shown in Equation 22:
VAINPLIM ≤ AVDD – 0.15 – VRTDMAX · (Gain – 1) / 2
where
•
•
•
VRTDMAX = maximum differential RTD voltage = 0.2 V
Gain = 8
AVDD = 4.75 V (minimum)
(22)
Solving Equation 22 results in a required VRTDP of less than 3.9 V. Calculate the VRTDP input voltage by
Equation 23:
VAINP = VRTDN + IIDAC1 · (RRTD + RLEAD1) = 1 V + 500 µA · (400 Ω + 10 Ω) = 1.2 V
(23)
Because 1.2 V is less than the 3.9-V maximum input voltage limit, the absolute positive and negative RTD
voltages are within the ADC specified input range.
The next step in the design is to verify that the IDACs have enough voltage headroom (compliance voltage) to
operate. The loop voltage of the excitation current must be less than the supply voltage minus the specified IDAC
compliance voltage. Calculate the voltage drop developed across each IDAC current path to AVSS. In this circuit,
IDAC1 has the largest voltage drop developed across its current path. The IDAC1 calculation is sufficient to
satisfy IDAC2 because the IDAC2 voltage drop is always less than IDAC1 voltage drop. The sum of voltages in
the IDAC1 loop is shown in Equation 24:
VIDAC1 = [(IIDAC1 + IIDAC2) · (RLEAD3 + RBIAS)] + [IIDAC1 · (RRTD + RLEAD1 + RREF)] + VD
where
•
VD = external blocking diode voltage
(24)
The equation results in a loop voltage of VIDAC1 = 3.0 V. The worst-case current source compliance voltage is:
(AVDD – 0.4 V) = (4.75 V – 0.4 V) = 4.35 V. The VIDAC1 loop voltage is less than the specified current source
compliance voltage (3.0 V < 4.35 V).
74
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Many applications benefit from using an analog filter at the inputs to remove noise and interference from the
signal. Filter components are placed on the ADC inputs (RF1, RF2, CDIF1, CCM1, and CCM2), as well as on the
reference inputs (RF3, RF4, CDIF2, CCM3, and CCM4). The filters remove both differential and common-mode noise.
The application shows a differential input noise filter formed by RF1, RF2 and CDIF1, with additional differential
mode capacitance provided by the common-mode filter capacitors, CCM1 and CCM2. Calculate the differential
–3-dB cutoff frequency as shown in Equation 25:
fDIF = 1 / [2π · (RF1 + RF2) · (CDIF1 + CCM1|| CCM2)]
(25)
The common-mode noise filter is formed by components RF1, RF2, CCM1, and CCM2. Calculate the common-mode
signal –3-dB cutoff frequency, as shown in Equation 26:
fCM = 1 / (2π · RF1 · CCM1) = 1 / (2π · RF2 · CCM2)
(26)
Mismatches in the common-mode filter components convert common-mode noise into differential noise. To
reduce the effect of mismatch, use a differential mode filter with a corner frequency that is at least 10 times lower
than the common-mode filter corner frequency. The low-frequency differential filter removes the common-mode
converted noise. The filter resistors (RFx) also serve as current-limiting resistors. These resistors limit the current
into the analog inputs (AINx) of the device to safe levels when an overvoltage occurs on the inputs.
Filter resistors lead to an offset voltage error due to the dc input current leakage flowing into and out of the
device. Remove this voltage error by system offset calibration. Resistor values that are too large generate
excess thermal noise and degrade the overall noise performance. The recommended range of the filter resistor
values is 100 Ω to 10 kΩ. The properties of the capacitors are important because the capacitors are connected to
the signal; use high-quality C0G ceramics or film-type capacitors.
For consistent noise performance across the full range of RTD measurements, match the corner frequencies of
the input and reference filter. See the RTD Ratiometric Measurements and Filtering Using the ADS1148 and
ADS1248 Application Report for detailed information on matching the input and reference filter.
10.2.2.1 Register Settings
The register settings for this design are shown in Table 36.
Table 36. Register Settings
REGISTER
NAME
SETTING
02h
INPMUX
12h
Select AINP = AIN1 and AINN = AIN2
03h
PGA
0Bh
PGA enabled, PGA Gain = 8
04h
DATARATE
14h
Continuous conversion mode, 20-SPS data rate
05h
REF
06h
Positive and negative reference buffers enabled, REFP1 and
REFN1 reference inputs selected, internal reference always on
06h
IDACMAG
05h
IDAC magnitude set to 500 µA
07h
IDACMUX
35h
IDAC2 set to AIN3, IDAC1 set to AIN5
08h
VBIAS
00h
09h
SYS
0Ah
10h
(1)
xxh
0Bh
OFCAL1
xxh
0Ch
OFCAL2
xxh
0Dh
(1)
OFCAL0
FSCAL0
DESCRIPTION
(1)
xxh
0Eh
FSCAL1
0Fh
FSCAL2
xxh
xxh
10h
GPIODAT
00h
11h
GPIOCON
00h
A two-point offset and gain calibration removes errors from the RREF tolerance. The results are used for the OFC and FSC registers.
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10.2.3 Application Curves
To test the accuracy of the acquisition circuit, a series of calibrated high-precision discrete resistors are used as
an input to the system. Measurements are taken at TA = 25°C. Figure 97 displays the resistance measurement
over an input span from 20 Ω to 400 Ω. Any offset error is generally attributed to the offset of the ADC, and the
gain error can be attributed to the accuracy of the RREF resistor and the ADC. The RREF value is also calibrated
to reduce the gain error contribution.
Precision temperature measurement applications are typically calibrated to remove the effects of gain and offset
errors that generally dominate the total system error. The simplest calibration method is a linear, or two-point
calibration that applies an equal and opposite gain and offset term to cancel the measured system gain and
offset error. In this particular tested application, the gain and offset error was very small, and did not require
additional calibration other than the self offset and gain calibration provided by the device. The resulting
measured resistance error is shown in Figure 98.
The results in Figure 98 are converted to temperature accuracy by dividing the results by the RTD sensitivity (α)
at the measured resistance. Over the full resistance input range, the maximum total measured error is
±0.0190 Ω. Equation 27 uses the measured resistance error and the RTD sensitivity at 0°C to calculate the
measured temperature accuracy.
Error (°C) = Error (Ω) / α@0°C = ±0.0190 Ω / 0.39083 Ω / °C = ±0.049°C
(27)
35000
0.03
30000
0.02
Resistance Error (:)
ADC Output (Code)
Figure 99 displays the calculated temperature accuracy of the circuit assuming a linear RTD resistance to
temperature response. This figure does not include any linearity compensation of the RTD, but Figure 99 does
remove offset and gain error, which can be calibrated with the OFC and FSC registers.
25000
20000
15000
10000
0.01
0
-0.01
-0.02
5000
-0.03
0
0
50
100
150
200 250 300
Resistance (:)
350
400
450
Figure 97. ADC Output Code vs Equivalent RTD
Resistance
0
50
100
150
200 250 300
Resistance (:)
350
400
450
Figure 98. Measured Resistance Error vs Equivalent RTD
Resistance
0.06
Temperature Error (qC)
0.04
0.02
0
-0.02
-0.04
-0.06
0
50
100
150
200 250 300
Resistance (:)
350
400
450
Figure 99. Equivalent Temperature Error vs Equivalent RTD Resistance
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10.3 Do's and Don'ts
•
•
•
•
•
•
•
•
•
•
Do partition the analog, digital, and power-supply circuitry into separate sections on the PCB.
Do use a single ground plane for analog and digital grounds.
Do place the analog components close to the ADC pins using short, direct connections.
Do keep the SCLK pin free of glitches and noise.
Do verify that the analog input voltages are within the specified PGA input voltage range under all input
conditions.
Do float unused analog input pins to minimize input leakage current on all other analog inputs. Connecting
unused pins to AVDD is the next best option.
Do provide current limiting to the analog inputs in case overvoltage faults occur.
Do use a low-dropout linear regulator (LDO) to reduce ripple voltage generated by switch-mode power
supplies. Reducing ripple is especially important for AVDD where the supply noise can affect the
performance.
Don't cross analog and digital signals.
Don't allow the analog and digital power supply voltages to exceed 5.5 V under any condition, including
during power-up and power-down.
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Do's and Don'ts (continued)
Figure 100 shows the do's and don'ts of the ADC circuit connections.
INCORRECT
CORRECT
5V
5V
AVDD
AVDD
Device
Device
AINP
AINP
16-bit
û ADC
PGA
AINN
AVSS
0V
16-bit
û ADC
PGA
AINN
AVSS
0V
0V
0V
Single-ended input, PGA enabled
Single-ended input, PGA bypassed
CORRECT
CORRECT
2.5 V
5V
AVDD
AVDD
Device
Device
AINP
AINP
16-bit
û ADC
PGA
2.5 V
AINN
PGA
AVSS
16-bit
û ADC
AVSS
0V
-2.5 V
0V
Single-ended input, PGA enabled
Single-ended input, PGA enabled
INCORRECT
5V
PGA enabled
AINN
AVDD
3.3 V
5V
INCORRECT
3.3 V
DVDD
AVDD
PGA
16-bit
û ADC
PGA
16-bit
û ADC
AVSS
DGND
AVSS
DGND
Device
CORRECT
AVDD
3.3 V
DVDD
AGND/DGND isolation
Inductive supply or ground connections
5V
Device
2.5 V
CORRECT
3.3 V
DVDD
AVDD
PGA
16-bit
û ADC
PGA
16-bit
û ADC
AVSS
DGND
AVSS
DGND
Device
Device
DVDD
-2.5 V
Low impedance AGND/DGND connection
Low impedance AGND/DGND connection
Figure 100. Do's and Don'ts Circuit Connections
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11 Power Supply Recommendations
11.1 Power Supplies
The ADS114S0xB require three power supplies: analog (AVDD, AVSS), digital core (DVDD, DGND), and digital
I/O (IOVDD, DGND). The analog power supply can be bipolar (for example, AVDD = 2.5 V, AVSS = –2.5 V) or
unipolar (for example, AVDD = 3.3 V, AVSS = 0 V) and is independent of the digital power supplies. DVDD is
used to power the digital circuits of the devices. IOVDD sets the digital I/O levels (with the exception of the GPIO
levels that are set by the analog supply of AVDD and AVSS). IOVDD must be equal to or larger than DVDD.
11.2 Power-Supply Sequencing
AVDD and DVDD may be powered up in any order. However, IOVDD is recommended to be powered up before
or at the same time as DVDD. If DVDD comes up before IOVDD, a reset of the device using the RESET pin or
the RESET command may be required.
11.3 Power-On Reset
An internal POR is released after all three supplies exceed approximately 1.65 V. Each supply has an individual
POR circuit. A brownout condition on any of the three supplies triggers a reset of the complete device.
11.4 Power-Supply Decoupling
Good power-supply decoupling is important to achieve best performance. AVDD must be decoupled with at least
a 330-nF capacitor to AVSS. DVDD and IOVDD (when not connected to DVDD) must be decoupled with at least
a 0.1-μF capacitor to DGND. Figure 101 and Figure 102 show typical power-supply decoupling examples for
unipolar and bipolar analog supplies, respectively. Place the bypass capacitors as close to the power-supply pins
of the device as possible using low-impedance connections. Use multi-layer ceramic chip capacitors (MLCCs)
that offer low equivalent series resistance (ESR) and inductance (ESL) characteristics for power-supply
decoupling purposes. To reduce inductance on the supply pins, avoid the use of vias for connecting the
capacitors to the supply pins. The use of multiple vias in parallel lowers the overall inductance and is beneficial
for connections to ground planes. Connect analog and digital grounds together as close to the device as
possible.
23
RESET
24
CLK
17
GPIO3/AIN11
18
GPIO2/AIN10
19
GPIO1/AIN9
20
GPIO0/AIN8
RESET
CLK
21
REFOUT
GPIO3/AIN11
22
±2.5 V
REFCOM
GPIO2/AIN10
23
GPIO1/AIN9
24
GPIO0/AIN8
1 PF
REFOUT
REFCOM
1 PF
22
21
20
19
18
17
3.3 V
NC
5V
25
26
AVDD
DVDD
16
IOVDD
15
+2.5 V
0.1 PF
330 nF
REFN0
27
AVSS
28
AVSS
DGND
29
3.3 V
NC
330 nF
14
13
DRDY
12
DOUT/DRDY
±2.5 V
25
DVDD
16
26
AVDD
IOVDD
15
27
AVSS
DGND
14
28
AVSS
0.1 PF
13
DRDY
REFN0
29
12
DOUT/DRDY
REFP0
30
11
SCLK
REFP1/AIN6
32
9
CS
1
2
3
4
5
6
7
8
Figure 101. Unipolar Analog Power Supply
1
2
3
4
5
6
7
8
START/SYNC
CS
AIN0
9
AIN1
32
AIN2
REFP1/AIN6
AIN3
DIN
AIN4
10
AIN5
31
AINCOM
REFN1/AIN7
AIN0
DIN
START/SYNC
10
AIN1
31
AIN2
REFN1/AIN7
AIN3
SCLK
AIN4
11
AIN5
30
AINCOM
REFP0
Figure 102. Bipolar Analog Power Supply
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12 Layout
12.1 Layout Guidelines
Employing best design practices is recommended when laying out a printed-circuit board (PCB) for both analog
and digital components. This recommendation generally means that the layout separates analog components
[such as ADCs, amplifiers, references, digital-to-analog converters (DACs), and analog MUXs] from digital
components [such as microcontrollers, complex programmable logic devices (CPLDs), field-programmable gate
arrays (FPGAs), radio frequency (RF) transceivers, universal serial bus (USB) transceivers, and switching
regulators]. An example of good component placement is shown in Figure 103. Although Figure 103 provides a
good example of component placement, the best placement for each application is unique to the geometries,
components, and PCB fabrication capabilities employed. That is, there is no single layout that is perfect for every
design and careful consideration must always be used when designing with any analog component.
Ground Fill or
Ground Plane
Supply
Generation
Microcontroller
Device
Optional: Split
Ground Cut
Signal
Conditioning
(RC Filters
and
Amplifiers)
Ground Fill or
Ground Plane
Optional: Split
Ground Cut
Ground Fill or
Ground Plane
Interface
Transceiver
Connector
or Antenna
Ground Fill or
Ground Plane
Figure 103. System Component Placement
The following basic recommendations for layout of the ADS114S0xB help achieve the best possible performance
of the ADC. A good design can be ruined with a bad circuit layout.
• Separate analog and digital signals. To start, partition the board into analog and digital sections where the
layout permits. Route digital lines away from analog lines. This prevents digital noise from coupling back into
analog signals.
• The ground plane can be split into an analog plane (AGND) and digital plane (DGND), but this (splitting) is
not necessary. Place digital signals over the digital plane, and analog signals over the analog plane. As a
final step in the layout, the split between the analog and digital grounds must be connected to together at the
ADC.
• Fill void areas on signal layers with ground fill.
• Provide good ground return paths. Signal return currents will flow on the path of least impedance. If the
ground plane is cut or has other traces that block the current from flowing right next to the signal trace,
another path must be found to return to the source and complete the circuit. If forced into a larger path, the
chance that the signal radiates increases. Sensitive signals are more susceptible to EMI interference.
• Use bypass capacitors on supplies to reduce high-frequency noise. Do not place vias between bypass
capacitors and the active device. Placing the bypass capacitors on the same layer as close to the active
device yields the best results.
• Consider the resistance and inductance of the routing. Often, traces for the inputs have resistances that react
with the input bias current and cause an added error voltage. Reducing the loop area enclosed by the source
signal and the return current reduces the inductance in the path. Reducing the inductance reduces the EMI
pickup and reduces the high-frequency impedance at the input of the device.
• Watch for parasitic thermocouples in the layout. Dissimilar metals going from each analog input to the sensor
can create a parasitic themocouple that can add an offset to the measurement. Differential inputs must be
matched for both the inputs going to the measurement source.
• Analog inputs with differential connections must have a capacitor placed differentially across the inputs. Best
input combinations for differential measurements use adjacent analog input lines (such as AIN0, AIN1 and
AIN2, AIN3). The differential capacitors must be of high quality. The best ceramic chip capacitors are C0G
(NPO) that have stable properties and low noise characteristics.
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Reference
input
REFN0
REFP0
Differential
input or
Reference
input
AIN7
AIN6
12.2 Layout Example
Internal plane connected to GND
(DGND = AVSS)
Via connection to power plane
AIN5
Differential
input
Differential
input
25: NC
27: AVSS
26: AVDD
29: REFN0
28: AVSS
30: REFP0
32: REFP1/
AIN6
31: REFN1/
AIN7
AIN8
AIN4
1: AINCOM
24: REFCOM
2: AIN5
23: REFOUT
3: AIN4
22: GPIO0/
AIN8
4: AIN3
21: GPIO1/
AIN9
5: AIN2
20: GPIO2/
AIN10
6: AIN1
19: GPIO3/
AIN11
7: AIN0
18: RESET
8: START
17: CLK
AIN9
AIN3
Differential
input
AIN10
Differential
input
16: DVDD
15: IOVDD
14: DGND
13: DRDY
11: SCLK
12: DOUT/
DRDY
10: DIN
9: CS
AIN2
AIN11
AIN1
Differential
input
AIN0
RESET
CLK
DRDY
DOUT/DRDY
SCLK
DIN
CS
START
Digital
connections
Figure 104. ADS114S0xB Layout Example
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13 Device and Documentation Support
13.1 Device Support
13.1.1 Development Support
ADS1x4S0x Design Calculator
13.2 Documentation Support
13.2.1 Related Documentation
For related documentation see the following:
• ADS114S0x Low-Power, Low-Noise, Highly Integrated, 6- and 12-Channel, 4-kSPS, 16-Bit, Delta-Sigma ADC
with PGA and Voltage Reference
• ADS124S0x Low-Power, Low-Noise, Highly Integrated, 6- and 12-Channel, 4-kSPS, 24-Bit, Delta-Sigma ADC
with PGA and Voltage Reference
• REF50xx Low-Noise, Very Low Drift, Precision Voltage Reference
• RTD Ratiometric Measurements and Filtering Using the ADS1148 and ADS1248 Application Report
• 3-Wire RTD Measurement System Reference Design, -200°C to 850°C
13.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to order now.
Table 37. Related Links
PARTS
PRODUCT FOLDER
ORDER NOW
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
ADS114S06B
Click here
Click here
Click here
Click here
Click here
ADS114S08B
Click here
Click here
Click here
Click here
Click here
13.4 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
13.5 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
13.6 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
13.7 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
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13.8 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
14 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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8-Sep-2017
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
ADS114S06BIPBS
ACTIVE
TQFP
PBS
32
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
114S06B
ADS114S06BIPBSR
ACTIVE
TQFP
PBS
32
1000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
114S06B
ADS114S06BIRHBR
ACTIVE
VQFN
RHB
32
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
ADS
114S06B
ADS114S06BIRHBT
ACTIVE
VQFN
RHB
32
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
ADS
114S06B
ADS114S08BIPBS
ACTIVE
TQFP
PBS
32
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
114S08B
ADS114S08BIPBSR
ACTIVE
TQFP
PBS
32
1000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
114S08B
ADS114S08BIRHBR
ACTIVE
VQFN
RHB
32
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
ADS
114S08B
ADS114S08BIRHBT
ACTIVE
VQFN
RHB
32
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 125
ADS
114S08B
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
8-Sep-2017
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
15-Aug-2019
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
ADS114S06BIPBSR
TQFP
PBS
32
1000
330.0
16.4
7.2
7.2
1.5
12.0
16.0
Q2
ADS114S06BIRHBR
VQFN
RHB
32
3000
330.0
12.4
5.3
5.3
1.1
8.0
12.0
Q2
ADS114S06BIRHBT
VQFN
RHB
32
250
180.0
12.4
5.3
5.3
1.1
8.0
12.0
Q2
ADS114S08BIPBSR
TQFP
PBS
32
1000
330.0
16.4
7.2
7.2
1.5
12.0
16.0
Q2
ADS114S08BIRHBR
VQFN
RHB
32
3000
330.0
12.4
5.3
5.3
1.1
8.0
12.0
Q2
ADS114S08BIRHBT
VQFN
RHB
32
250
180.0
12.4
5.3
5.3
1.1
8.0
12.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
15-Aug-2019
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
ADS114S06BIPBSR
TQFP
PBS
32
1000
350.0
350.0
43.0
ADS114S06BIRHBR
VQFN
RHB
32
3000
367.0
367.0
35.0
ADS114S06BIRHBT
VQFN
RHB
32
250
210.0
185.0
35.0
ADS114S08BIPBSR
TQFP
PBS
32
1000
350.0
350.0
43.0
ADS114S08BIRHBR
VQFN
RHB
32
3000
367.0
367.0
35.0
ADS114S08BIRHBT
VQFN
RHB
32
250
210.0
185.0
35.0
Pack Materials-Page 2
GENERIC PACKAGE VIEW
RHB 32
VQFN - 1 mm max height
PLASTIC QUAD FLATPACK - NO LEAD
5 x 5, 0.5 mm pitch
Images above are just a representation of the package family, actual package may vary.
Refer to the product data sheet for package details.
4224745/A
www.ti.com
PACKAGE OUTLINE
VQFN - 1 mm max height
RHB0032M
PLASTIC QUAD FLATPACK-NO LEAD
A
5.1
4.9
B
5.1
4.9
PIN 1 INDEX AREA
C
1 MAX
SEATING PLANE
0.08 C
0.05
0.00
2X 3.5
(0.2) TYP
9
16
28X 0.5
8
17
SYMM
33
2X
3.5
1
PIN 1 ID
(OPTIONAL)
24
32
25
SYMM
32X 0.5
0.3
32X 0.3
0.2
0.1
0.05
C A B
C
4223725/A 08/2017
NOTES:
1.
2.
3.
All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
This drawing is subject to change without notice.
The package thermal pad must be soldered to the printed circuit board for optimal thermal and mechanical performance.
www.ti.com
EXAMPLE BOARD LAYOUT
VQFN - 1 mm max height
RHB0032M
PLASTIC QUAD FLATPACK-NO LEAD
(4.8)
(
2.1)
32
32X (0.6)
25
32X (0.25)
1
24
28X (0.5)
33
SYMM
(4.8)
2X
(0.8)
VIA TYP
8
17
(R0.05) TYP
9
16
2X (0.8)
SYMM
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE: 15X
0.07 MAX
ALL AROUND
0.07 MIN
ALL AROUND
METAL
EXPOSED
METAL
SOLDER MASK
OPENING
SOLDER MASK
OPENING
EXPOSED
METAL
NON SOLDER MASK
DEFINED
(PREFERRED)
METAL UNDER
SOLDER MASK
SOLDER MASK
DEFINED
SOLDER MASK DETAILS
4223725/A 08/2017
NOTES: (continued)
4.
5.
This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
number SLUA271 (www.ti.com/lit/slua271).
Vias are optional depending on application, refer to device data sheet. If any vias are implemented, refer to their locations shown
on this view. It is recommended that vias under paste be filled, plugged or tented.
www.ti.com
EXAMPLE STENCIL DESIGN
VQFN - 1 mm max height
RHB0032M
PLASTIC QUAD FLATPACK-NO LEAD
(4.8)
4X (
0.94)
32
32X (0.6)
25
32X (0.25)
1
24
28X (0.5)
33
SYMM
(4.8)
2X
(0.57)
METAL
TYP
8
17
(R0.05) TYP
9
16
2X (0.57)
SYMM
SOLDER PASTE EXAMPLE
BASED ON 0.125 mm THICK STENCIL
EXPOSED PAD
80% PRINTED COVERAGE BY AREA
SCALE: 15X
4223725/A 08/2017
NOTES: (continued)
6.
Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
www.ti.com
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TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATASHEETS), DESIGN RESOURCES (INCLUDING REFERENCE
DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS”
AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY
IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
PARTY INTELLECTUAL PROPERTY RIGHTS.
These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate
TI products for your application, (2) designing, validating and testing your application, and (3) ensuring your application meets applicable
standards, and any other safety, security, or other requirements. These resources are subject to change without notice. TI grants you
permission to use these resources only for development of an application that uses the TI products described in the resource. Other
reproduction and display of these resources is prohibited. No license is granted to any other TI intellectual property right or to any third
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warranties or warranty disclaimers for TI products.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2019, Texas Instruments Incorporated
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