Texas Instruments | ADC14L020 14-Bit, 20 MSPS, 150 mW A/D Converter (Rev. D) | Datasheet | Texas Instruments ADC14L020 14-Bit, 20 MSPS, 150 mW A/D Converter (Rev. D) Datasheet

Texas Instruments ADC14L020 14-Bit, 20 MSPS, 150 mW A/D Converter (Rev. D) Datasheet
ADC14L020
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ADC14L020 14-Bit, 20 MSPS, 150 mW A/D Converter
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FEATURES
DESCRIPTION
•
•
•
•
•
•
The ADC14L020 is a low power monolithic CMOS
analog-to-digital converter capable of converting
analog input signals into 14-bit digital words at 20
Megasamples per second (MSPS). This converter
uses a differential, pipeline architecture with digital
error correction and an on-chip sample-and-hold
circuit to minimize power consumption while providing
excellent dynamic performance and a 150 MHz Full
Power Bandwidth. Operating on a single +3.3V power
supply, the ADC14L020 achieves 12.0 effective bits
at nyquist and consumes just 150 mW at 20 MSPS .
The Power Down feature reduces power consumption
to 15 mW.
1
2
Single +3.3V Supply Operation
Internal Sample-and-Hold
Internal Reference
Outputs 2.4V to 3.6V Compatible
Duty Cycle Stabilizer
Power Down Mode
KEY SPECIFICATIONS
•
•
•
•
•
•
Resolution 14 Bits
DNL ±0.5 LSB (typ)
SNR (fIN = 10 MHz) 74 dB (typ)
SFDR (fIN = 10 MHz) 93 dB (typ)
Data Latency 7 Clock Cycles
Power Consumption
– Operating 150 mW (typ)
– Power Down Mode 15 mW (typ)
APPLICATIONS
•
•
•
•
Medical Imaging
Instrumentation
Communications
Digital Video
The differential inputs provide a full scale differential
input swing equal to 2 times VREF with the possibility
of a single-ended input. Full use of the differential
input is recommended for optimum performance.
Duty cycle stabilization and output data format are
selectable using a quad state function pin. The output
data can be set for offset binary or two's complement.
To ease interfacing to lower voltage systems, the
digital output driver power pins of the ADC14L020
can be connected to a separate supply voltage in the
range of 2.4V to the analog supply voltage.
This device is available in the 32-lead LQFP package
and will operate over the industrial temperature range
of −40°C to +85°C. An evaluation board is available
to ease the evaluation process.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005–2013, Texas Instruments Incorporated
ADC14L020
SNAS325D – JUNE 2005 – REVISED APRIL 2013
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25
AGND
D13 (MSB)
26
27
VA
D12
D11
21
ADC14L020
(Top View)
20
19
D10
D9
D8
VDR
DRGND
D7
D6
16
D5
D4
15
D3
D2
D1
CLK
DGND
14
17
13
8
12
18
9
7
10
PD
28
22
6
AGND
VRN
3
5
VD
29
VRP
23
4
AGND
VA
30
24
2
(LSB) D0
VIN-
1
11
VIN+
DF/DCS
VREF
31
32
VRM
Connection Diagram
Block Diagram
VIN+
S/H
Stage 1
Stage 2
Stage 3
Stage 9
Stage 10
Stage 11
VIN3
3
3
2
2
3
3
3
2
2
3
3
Timing
Control
11-Stage Pipeline Converter
24
Digital
Correction
14
Output
Buffers
14
D0-D13
3
CLK
Duty Cycle
Stabilizer
VRP
VRM
VRN
VREF
2
Reference
Select
Internal
Reference
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Pin Descriptions and Equivalent Circuits
Pin No.
Symbol
Equivalent Circuit
Description
ANALOG I/O
2
3
VIN+
VA
Differential analog input pins. With a 1.0V reference voltage the
differential full-scale input signal level is 2.0 VP-P with each input pin
voltage centered on a common mode voltage, VCM. The negative
input pins may be connected to VCM for single-ended operation, but
a differential input signal is required for best performance.
VIN−
AGND
VA
1
VREF
31
VRP
32
VRM
This pin is the reference select pin and the external reference input.
If (VA - 0.3V) < VREF < VA, the internal 1.0V reference is selected.
If AGND < VREF < (AGND + 0.3V), the internal 0.5V reference is
selected.
If a voltage in the range of 0.4V to (VA - 0.4V) is applied to this pin,
that voltage is used as the reference.
The full scale differential voltage range is 2 * VREF. VREF should be
bypassed to AGND with a 0.1 µF capacitor when an external
reference is used.
AGND
VA
VA
VA
30
VRN
VA
These pins should each be bypassed to AGND with a low ESL
(equivalent series inductance) 0.1 µF capacitor. A 10 µF capacitor
should be placed between the VRP and VRN.
VRM may be loaded to 1mA for use as a temperature stable 1.5V
reference. The remaining pins should not be loaded.
VRM may be used to provide the common mode voltage, VCM, for the
differential inputs.
AGND
AGND
VA
11
VFloat
This is a four-state pin.
DF/DCS = VA, output data format is offset binary with duty cycle
stabilization applied to the input clock
DF/DCS = AGND, output data format is 2's complement, with duty
cycle stabilization applied to the input clock.
DF/DCS = VRM , output data is 2's complement without duty cycle
stabilization applied to the input clock
DF/DCS = "float", output data is offset binary without duty cycle
stabilization applied to the input clock.
DF/DCS
AGND
DIGITAL I/O
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Pin No.
Symbol
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Equivalent Circuit
VD
10
CLK
8
PD
VA
Description
Digital clock input. The range of frequencies for this input is as
specified in the electrical tables with specified performance at 20
MHz. The input is sampled on the rising edge.
PD is the Power Down input pin. When high, this input puts the
converter into the power down mode. When this pin is low, the
converter is in the active mode.
AGND
DGND
VDR VA
12-19
22-27
Digital data output pins that make up the 14-bit conversion result. D0
(pin 12) is the LSB, while D13 (pin 27) is the MSB of the output
word. Output levels are TTL/CMOS compatible. Optimum loading is
< 10pF.
D0–D13
AGND
DR GND
ANALOG POWER
5, 29
VA
4, 7, 28
AGND
Positive analog supply pins. These pins should be connected to a
quiet +3.3V source and bypassed to AGND with 0.1 µF capacitors
located close to these power pins, and with a 10 µF capacitor.
The ground return for the analog supply.
DIGITAL POWER
6
VD
9
DGND
21
20
4
Positive digital supply pin. This pin should be connected to the same
quiet +3.3V source as is VA and be bypassed to DGND with a 0.1 µF
capacitor located close to the power pin and with a 10 µF capacitor.
The ground return for the digital supply.
VDR
Positive driver supply pin for the ADC14L020's output drivers. This
pin should be connected to a voltage source of +2.4V to VD and be
bypassed to DR GND with a 0.1 µF capacitor. If the supply for this
pin is different from the supply used for VA and VD, it should also be
bypassed with a 10 µF capacitor. VDR should never exceed the
voltage on VD. All 0.1 µF bypass capacitors should be located close
to the supply pin.
DR GND
The ground return for the digital supply for the ADC's output drivers.
These pins should be connected to the system digital ground, but
not be connected in close proximity to the ADC's DGND or AGND
pins. See LAYOUT AND GROUNDING for more details.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2) (3)
VA, VD, VDR
4.2V
≤ 100 mV
|VA–VD|
−0.3V to (VA or VD +0.3V)
Voltage on Any Input or Output Pin
Input Current at Any Pin
(4)
±25 mA
Package Input Current (4)
±50 mA
Package Dissipation at TA = 25°C
ESD Susceptibility
See
Human Body Model
(6)
(5)
2500V
Machine Model (6)
250V
−65°C to +150°C
Storage Temperature
Soldering process must comply with TI's Reflow Temperature Profile specifications. Refer to www.ti.com/packaging. (7)
(1)
(2)
(3)
(4)
(5)
(6)
(7)
All voltages are measured with respect to GND = AGND = DGND = 0V, unless otherwise specified.
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the
Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance characteristics may
degrade when the device is not operated under the listed test conditions.
If Military/Aerospace specified devices are required, please contact the TI Sales Office/ Distributors for availability and specifications.
When the input voltage at any pin exceeds the power supplies (that is, VIN < AGND, or VIN > VA), the current at that pin should be
limited to 25 mA. The 50 mA maximum package input current rating limits the number of pins that can safely exceed the power supplies
with an input current of 25 mA to two.
The absolute maximum junction temperature (TJmax) for this device is 150°C. The maximum allowable power dissipation is dictated by
TJmax, the junction-to-ambient thermal resistance (θJA), and the ambient temperature, (TA), and can be calculated using the formula
PDMAX = (TJmax - TA )/θJA. The values for maximum power dissipation listed above will be reached only when the device is operated in
a severe fault condition (e.g. when input or output pins are driven beyond the power supply voltages, or the power supply polarity is
reversed). Obviously, such conditions should always be avoided.
Human body model is 100 pF capacitor discharged through a 1.5 kΩ resistor. Machine model is 220 pF discharged through 0Ω.
Reflow temperature profiles are different for lead-free and non-lead-free packages.
Operating Ratings (1) (2)
Operating Temperature
−40°C ≤ TA ≤ +85°C
Supply Voltage (VA, VD)
+3.0V to +3.6V
Output Driver Supply (VDR)
+2.4V to VD
−0.05V to (VD + 0.05V)
CLK, PD
Clock Duty Cycle (DCS On)
20% to 80%
Clock Duty Cycle (DCS Off)
40% to 60%
Analog Input Pins
0V to 2.6V
VCM
0.5V to 2.0V
≤100mV
|AGND–DGND|
(1)
(2)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the
Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance characteristics may
degrade when the device is not operated under the listed test conditions.
All voltages are measured with respect to GND = AGND = DGND = 0V, unless otherwise specified.
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Converter Electrical Characteristics (1) (2) (3)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V,
VDR = +2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz at -0.5dBFS, tr = tf = 2 ns, CL = 15 pF/pin, Duty
Cycle Stabilizer On. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
Symbol
Parameter
Conditions
Typical
Limits
(4)
(4)
Units
(Limits)
STATIC CONVERTER CHARACTERISTICS
14
Bits (min)
INL
Resolution with No Missing Codes
Integral Non Linearity (5)
±1.4
±3.8
LSB (max)
DNL
Differential Non Linearity
±0.5
±1.0
LSB (max)
PGE
Positive Gain Error
0.3
±3.3
%FS (max)
NGE
Negative Gain Error
0.3
±3.3
%FS (max)
TC GE
Gain Error Tempco
VOFF
Offset Error (VIN+ = VIN−)
±0.85
%FS (max)
TC VOFF
Offset Error Tempco
−40°C ≤ TA ≤ +85°C
2.5
-0.06
−40°C ≤ TA ≤ +85°C
ppm/°C
1.5
ppm/°C
Under Range Output Code
0
Over Range Output Code
16383
REFERENCE AND ANALOG INPUT CHARACTERISTICS
VCM
Common Mode Input Voltage
VRM
Reference Output Voltage
Output load = 1 mA
CIN
VIN Input Capacitance (each pin to GND)
VIN = 1.5 Vdc ± 0.5 V
VREF
1.5
0.5
V (min)
2.0
V (max)
1.5
V
(CLK LOW)
11
pF
(CLK HIGH)
4.5
External Reference Voltage (6)
1.00
Reference Input Resistance
pF
0.8
1.2
V (min)
V (max)
1
MΩ (min)
150
MHz
DYNAMIC CONVERTER CHARACTERISTICS
FPBW
SNR
SINAD
ENOB
(1)
(2)
Full Power Bandwidth
Signal-to-Noise Ratio
Signal-to-Noise Ratio and Distortion
Effective Number of Bits
0 dBFS Input, Output at −3 dB
fIN = 1 MHz
74
fIN =10 MHz
74
fIN = 1 MHz
74
fIN = 10 MHz
74
fIN = 1 MHz
12
fIN = 10 MHz
12
dBc
72.3
dBc
dBc
72.2
dBc
Bits
11.7
Bits
With the test condition for VREF = +1.0V (2VP-P differential input), the 14-bit LSB is 122.1 µV.
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 under Absolute Maximum Ratings. However, errors in the A/D conversion can occur if the input goes above
VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure accurate
conversions.
VA
I/O
To Internal Circuitry
AGND
(3)
(4)
(5)
(6)
6
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are specified to TI's AOQL (Average Outgoing
Quality Level).
Integral Non Linearity is defined as the deviation of the analog value, expressed in LSBs, from the straight line that passes through
positive and negative full-scale.
Optimum performance will be obtained by keeping the reference input in the 0.8V to 1.2V range. The LM4051CIM3-ADJ (SOT-23
package) is recommended for external reference applications.
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Converter Electrical Characteristics (continued) (1) (2) (3) (4)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V,
VDR = +2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz at -0.5dBFS, tr = tf = 2 ns, CL = 15 pF/pin, Duty
Cycle Stabilizer On. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
Symbol
THD
H2
Parameter
Total Harmonic Disortion
Second Harmonic Distortion
H3
Third Harmonic Distortion
SFDR
Spurious Free Dynamic Range
IMD
Intermodulation Distortion
(1)
Conditions
Typical
(5)
fIN = 1 MHz
-90
fIN = 10 MHz
-90
fIN = 1 MHz
-97
fIN = 10 MHz
-97
fIN = 1 MHz
-96
fIN = 10 MHz
-96
fIN = 1 MHz
93
fIN = 10 MHz
93
fIN = 4.8 MHz and 5.2 MHz, each = −6.5
dBFS
−76
Limits
(5)
Units
(Limits)
dBc
-80
dBc
dBc
-81
dBc
dBc
-81
dBc
dBc
81
dBc
dBFS
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 under Absolute Maximum Ratings. However, errors in the A/D conversion can occur if the input goes above
VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure accurate
conversions.
VA
I/O
To Internal Circuitry
AGND
(2)
(3)
(4)
(5)
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
With the test condition for VREF = +1.0V (2VP-P differential input), the 14-bit LSB is 122.1 µV.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are specified to TI's AOQL (Average Outgoing
Quality Level).
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are specified to TI's AOQL (Average Outgoing
Quality Level).
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DC and Logic Electrical Characteristics (1) (2) (3)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
Symbol
Parameter
Typical
Conditions
(4)
Limits
(4)
Units
(Limits)
CLK, PD DIGITAL INPUT CHARACTERISTICS
VIN(1)
Logical “1” Input Voltage
VD = 3.6V
2.0
V (min)
VIN(0)
Logical “0” Input Voltage
VD = 3.0V
1.0
V (max)
IIN(1)
Logical “1” Input Current
VIN = 3.3V
10
µA
IIN(0)
Logical “0” Input Current
VIN = 0V
−10
µA
CIN
Digital Input Capacitance
5
pF
D0–D13 DIGITAL OUTPUT CHARACTERISTICS
VDR = 2.5V
2.3
V (min)
VDR = 3V
2.7
V (min)
0.4
V (max)
VOUT(1)
Logical “1” Output Voltage
IOUT = −0.5 mA
VOUT(0)
Logical “0” Output Voltage
IOUT = 1.6 mA, VDR = 3V
+ISC
Output Short Circuit Source Current
VOUT = 0V
−10
mA
−ISC
Output Short Circuit Sink Current
VOUT = VDR
10
mA
COUT
Digital Output Capacitance
5
pF
POWER SUPPLY CHARACTERISTICS
IA
Analog Supply Current
PD Pin = DGND, VREF = VA
PD Pin = VD
41
4.5
57
mA (max)
mA
ID
Digital Supply Current
PD Pin = DGND
PD Pin = VD , fCLK = 0
4.5
0
8
mA (max)
mA
IDR
Digital Output Supply Current
PD Pin = DGND, CL = 5 pF (5)
PD Pin = VD, fCLK = 0
2.5
0
Total Power Consumption
PD Pin = DGND, CL = 5 pF (6)
150
Power Down Power Consumption
PD Pin = VD, clock on
15
mW
Power Supply Rejection Ratio
Rejection of Full-Scale Error with
VA =3.0V vs. 3.6V
72
dB
PSRR
(1)
mA
mA
215
mW (max)
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 under Absolute Maximum Ratings. However, errors in the A/D conversion can occur if the input goes above
VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure accurate
conversions.
VA
I/O
To Internal Circuitry
AGND
(2)
(3)
(4)
(5)
(6)
8
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
With the test condition for VREF = +1.0V (2VP-P differential input), the 14-bit LSB is 122.1 µV.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are specified to TI's AOQL (Average Outgoing
Quality Level).
IDR is the current consumed by the switching of the output drivers and is primarily determined by load capacitance on the output pins,
the supply voltage, VDR, and the rate at which the outputs are switching (which is signal dependent). IDR=VDR(C0 x f0 + C1 x f1 +....C11 x
f11) where VDR is the output driver power supply voltage, Cn is total capacitance on the output pin, and fn is the average frequency at
which that pin is toggling.
Excludes IDR. See Note 5.
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AC Electrical Characteristics (1) (2) (3) (4)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
Symbol
Parameter
Conditions
Typical
(5)
Limits
(5)
Units
(Limits)
20
MHz (min)
fCLK1
Maximum Clock Frequency
fCLK
Minimum Clock Frequency
tCH
Clock High Time
Duty Cycle Stabilizer On
25
10
ns (min)
tCL
Clock Low Time
Duty Cycle Stabilizer On
25
10
ns (min)
tCH
Clock High Time
Duty Cycle Stabilizer Off
25
20
ns (min)
tCL
Clock Low Time
Duty Cycle Stabilizer Off
25
20
ns (min)
tCONV
Conversion Latency
7
Clock Cycles
tOD
Data Output Delay after Rising Clock
Edge
9.6
ns (max)
tAD
Aperture Delay
2
ns
tAJ
Aperture Jitter
0.7
ps rms
tPD
Power Down Mode Exit Cycle
280
µs
2
(1)
5
6
0.1 µF on pins 30, 31, 32; 10 µF between
pins 30, 31
MHz
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 under Absolute Maximum Ratings. However, errors in the A/D conversion can occur if the input goes above
VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure accurate
conversions.
VA
I/O
To Internal Circuitry
AGND
(2)
(3)
(4)
(5)
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
With the test condition for VREF = +1.0V (2VP-P differential input), the 14-bit LSB is 122.1 µV.
Timing specifications are tested at TTL logic levels, VIL = 0.4V for a falling edge and VIH = 2.4V for a rising edge.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are specified to TI's AOQL (Average Outgoing
Quality Level).
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Specification Definitions
APERTURE DELAY is the time after the rising edge of the clock to when the input signal is acquired or held for
conversion.
APERTURE JITTER (APERTURE UNCERTAINTY) is the variation in aperture delay from sample to sample.
Aperture jitter manifests itself as noise in the output.
CLOCK DUTY CYCLE is the ratio of the time during one cycle that a repetitive digital waveform is high to the
total time of one period. The specification here refers to the ADC clock input signal.
COMMON MODE VOLTAGE (VCM) is the common d.c. voltage applied to both input terminals of the ADC.
CONVERSION LATENCY is the number of clock cycles between initiation of conversion and when that data is
presented to the output driver stage. Data for any given sample is available at the output pins the Pipeline Delay
plus the Output Delay after the sample is taken. New data is available at every clock cycle, but the data lags the
conversion by the pipeline delay.
DIFFERENTIAL NON-LINEARITY (DNL) is the measure of the maximum deviation from the ideal step size of 1
LSB.
EFFECTIVE NUMBER OF BITS (ENOB, or EFFECTIVE BITS) is another method of specifying Signal-to-Noise
and Distortion or SINAD. ENOB is defined as (SINAD - 1.76) / 6.02 and says that the converter is equivalent to a
perfect ADC of this (ENOB) number of bits.
FULL POWER BANDWIDTH is a measure of the frequency at which the reconstructed output fundamental
drops 3 dB below its low frequency value for a full scale input.
GAIN ERROR is the deviation from the ideal slope of the transfer function. It can be calculated as:
Gain Error = Positive Full Scale Error − Negative Full Scale Error
(1)
It can also be expressed as Positive Gain Error and Negative Gain Error, which are calculated as:
PGE = Positive Full Scale Error - Offset Error NGE = Offset Error - Negative Full Scale Error
(2)
INTEGRAL NON LINEARITY (INL) is a measure of the deviation of each individual code from a line drawn from
negative full scale (½ LSB below the first code transition) through positive full scale (½ LSB above the last code
transition). The deviation of any given code from this straight line is measured from the center of that code value.
INTERMODULATION DISTORTION (IMD) is the creation of additional spectral components as a result of two
sinusoidal frequencies being applied to the ADC input at the same time. It is defined as the ratio of the power in
the intermodulation products to the total power in the original frequencies. IMD is usually expressed in dBFS.
LSB (LEAST SIGNIFICANT BIT) is the bit that has the smallest value or weight of all bits. This value is VFS/2n,
where “VFS” is the full scale input voltage and “n” is the ADC resolution in bits.
MISSING CODES are those output codes that will never appear at the ADC outputs. The ADC14L020 is ensured
not to have any missing codes.
MSB (MOST SIGNIFICANT BIT) is the bit that has the largest value or weight. Its value is one half of full scale.
NEGATIVE FULL SCALE ERROR is the difference between the actual first code transition and its ideal value of
½ LSB above negative full scale.
OFFSET ERROR is the difference between the two input voltages [(VIN+) – (VIN-)] required to cause a transition
from code 8191 to 8192.
OUTPUT DELAY is the time delay after the rising edge of the clock before the data update is presented at the
output pins.
PIPELINE DELAY (LATENCY) See CONVERSION LATENCY.
POSITIVE FULL SCALE ERROR is the difference between the actual last code transition and its ideal value of
1½ LSB below positive full scale.
POWER SUPPLY REJECTION RATIO (PSRR) is a measure of how well the ADC rejects a change in the power
supply voltage. PSRR is the ratio of the change in Full-Scale Error that results from a change in the d.c. power
supply voltage, expressed in dB.
10
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SIGNAL TO NOISE RATIO (SNR) is the ratio, expressed in dB, of the rms value of the input signal to the rms
value of the sum of all other spectral components below one-half the sampling frequency, not including
harmonics or d.c.
SIGNAL TO NOISE PLUS DISTORTION (S/N+D or SINAD) Is the ratio, expressed in dB, of the rms value of the
input signal to the rms value of all of the other spectral components below half the clock frequency, including
harmonics but excluding d.c.
SPURIOUS FREE DYNAMIC RANGE (SFDR) is the difference, expressed in dB, between the rms values of the
input signal and the peak spurious signal, where a spurious signal is any signal present in the output spectrum
that is not present at the input.
TOTAL HARMONIC DISTORTION (THD) is the ratio, expressed in dB, of the rms total of the first nine harmonic
levels at the output to the level of the fundamental at the output. THD is calculated as
(3)
where f1 is the RMS power of the fundamental (output) frequency and f2 through f10 are the RMS power of the
first 9 harmonic frequencies in the output spectrum.
SECOND HARMONIC DISTORTION (2ND HARM) is the difference expressed in dB, between the RMS power in
the input frequency at the output and the power in its 2nd harmonic level at the output.
THIRD HARMONIC DISTORTION (3RD HARM) is the difference, expressed in dB, between the RMS power in
the input frequency at the output and the power in its 3rd harmonic level at the output.
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Timing Diagram
Sample N + 7
Sample N + 8
Sample N + 9
|
Sample N + 6
Sample N
Sample N + 10
VIN
tAD
Clock N
Clock N + 7
1
fCLK
90%
90%
CLK
|
10%
tCH
10%
tCL
tf
tr
tOD
| |
D0 - D13
Data N - 1
Data N
Data N + 1
Data N + 2
Latency
Figure 1. Output Timing
Transfer Characteristic
Figure 2. Transfer Characteristic
12
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Typical Performance Characteristics, DNL, INL
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 0 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
DNL
INL
Figure 3.
Figure 4.
DNL vs. fCLK
INL vs. fCLK
Figure 5.
Figure 6.
DNL vs. Clock Duty Cycle
INL vs. Clock Duty Cycle
Figure 7.
Figure 8.
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Typical Performance Characteristics, DNL, INL (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 0 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
14
DNL vs. Temperature
INL vs. Temperature
Figure 9.
Figure 10.
DNL vs. VDR, VA = VD = 3.6V
INL vs. VDR, VA = VD = 3.6V
Figure 11.
Figure 12.
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Typical Performance Characteristics
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
SNR,SINAD,SFDR vs. VA
Distortion vs. VA
Figure 13.
Figure 14.
SNR,SINAD,SFDR vs. VDR, VA = VD = 3.6V
Distortion vs. VDR, VA = VD = 3.6V
Figure 15.
Figure 16.
SNR,SINAD,SFDR vs. VCM
Distortion vs. VCM
Figure 17.
Figure 18.
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Typical Performance Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
16
SNR,SINAD,SFDR vs. fCLK
Distortion vs. fCLK
Figure 19.
Figure 20.
SNR,SINAD,SFDR vs. Clock Duty Cycle
Distortion vs. Clock Duty Cycle
Figure 21.
Figure 22.
SNR,SINAD,SFDR vs. VREF
Distortion vs. VREF
Figure 23.
Figure 24.
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Typical Performance Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
SNR,SINAD,SFDR vs. fIN
Distortion vs. fIN
Figure 25.
Figure 26.
SNR,SINAD,SFDR vs. Temperature
Distortion vs. Temperature
Figure 27.
Figure 28.
tOD vs. VDR, VA = VD = 3.6V
Spectral Response @ 2.4 MHz Input
Figure 29.
Figure 30.
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Typical Performance Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 20 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On.
Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
18
Spectral Response @ 4.4 MHz Input
Spectral Response @ 10 MHz Input
Figure 31.
Figure 32.
Intermodulation Distortion, fIN1= 4.8 MHz, fIN2 = 5.2 MHz
Histogram with input grounded
Figure 33.
Figure 34.
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FUNCTIONAL DESCRIPTION
Operating on a single +3.3V supply, the ADC14L020 uses a pipeline architecture and has error correction
circuitry to help ensure maximum performance. The differential analog input signal is digitized to 14 bits. The
user has the choice of using an internal 1.0 Volt or 0.5 Volt stable reference, or using an external reference. Any
external reference is buffered on-chip to ease the task of driving that pin.
The output word rate is the same as the clock frequency. For the ADC14L020 the clock frequency can be
between 5 MSPS and 20 MSPS (typical) with fully specified performance at 20 MSPS. The analog input is
acquired at the rising edge of the clock and the digital data for a given sample is delayed by the pipeline for 7
clock cycles. Duty cycle stablization and output data format are selectable using the quad state function DF/DCS
pin. The output data can be set for offset binary or two's complement.
A logic high on the power down (PD) pin reduces the converter power consumption to 15 mW.
APPLICATIONS INFORMATION
OPERATING CONDITIONS
We recommend that the following conditions be observed for operation of the ADC14L020:
3.0V ≤ VA ≤ 3.6V
VD = VA
2.4V ≤ VDR ≤ VA
5 MHz ≤ fCLK ≤ 20 MHz
0.8V ≤ VREF ≤ 1.2V (for an external reference)
0.5V ≤ VCM ≤ 2.0V
Analog Inputs
There is one reference input pin, VREF, which is used to select an internal reference, or to supply an external
reference. The ADC14L020 has one analog signal input pairs, VIN + and VIN - . This pair of pins forms a
differential input pair.
Reference Pins
The ADC14L020 is designed to operate with an internal 1.0V or 0.5V reference, or an external 1.0V reference,
but performs well with external reference voltages in the range of 0.8V to 1.2V. Lower reference voltages will
decrease the signal-to-noise ratio (SNR) of the ADC14L020. Increasing the reference voltage (and the input
signal swing) beyond 1.2V may degrade THD for a full-scale input, especially at higher input frequencies.
It is important that all grounds associated with the reference voltage and the analog input signal make connection
to the ground plane at a single, quiet point to minimize the effects of noise currents in the ground path.
The Reference Bypass Pins (VRP, VRM, and VRN) are made available for bypass purposes. All these pins should
each be bypassed to ground with a 0.1 µF capacitor. A 10 µF capacitor should be placed between the VRP and
VRN pins, as shown in Figure 37. This configuration is necessary to avoid reference oscillation, which could result
in reduced SFDR and/or SNR. VRM may be loaded to 1mA for use as a temperature stable 1.5V reference. The
remaining pins should not be loaded.
Smaller capacitor values than those specified will allow faster recovery from the power down mode, but may
result in degraded noise performance. Loading any of these pins other than VRM may result in performance
degradation.
The nominal voltages for the reference bypass pins are as follows:
VRM = 1.5 V
VRP = VRM + VREF / 2
VRN = VRM − VREF / 2
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User choice of an on-chip or external reference voltage is provided. The internal 1.0 Volt reference is in use
when the the VREF pin is connected to VA. When the VREF pin is connected to AGND, the internal 0.5 Volt
reference is in use. If a voltage in the range of 0.8V to 1.2V is applied to the VREF pin, that is used for the voltage
reference. When an external reference is used, the VREF pin should be bypassed to ground with a 0.1 µF
capacitor close to the reference input pin. There is no need to bypass the VREF pin when the internal reference is
used.
Signal Inputs
The signal inputs are VIN + and VIN− . The input signal, VIN, is defined as
VIN = (VIN+) – (VIN−)
(4)
Figure 35 shows the expected input signal range. Note that the common mode input voltage, VCM, should be in
the range of 0.5V to 2.0V.
The peaks of the individual input signals should each never exceed 2.6V.
The ADC14L020 performs best with a differential input signal with each input centered around a common mode
voltage, VCM. The peak-to-peak voltage swing at each analog input pin should not exceed the value of the
reference voltage or the output data will be clipped.
The two input signals should be exactly 180° out of phase from each other and of the same amplitude. For single
frequency inputs, angular errors result in a reduction of the effective full scale input. For complex waveforms,
however, angular errors will result in distortion.
Figure 35. Expected Input Signal Range
For single frequency sine waves the full scale error in LSB can be described as approximately
EFS = 16384 ( 1 - sin (90° + dev))
(5)
Where dev is the angular difference in degrees between the two signals having a 180° relative phase relationship
to each other (see Figure 36). Drive the analog inputs with a source impedance less than 100Ω.
Figure 36. Angular Errors Between the Two Input Signals Will Reduce the Output Level or Cause
Distortion
For differential operation, each analog input pin of the differential pair should have a peak-to-peak voltage equal
to the reference voltage, VREF, be 180 degrees out of phase with each other and be centered around VCM.
Single-Ended Operation
Performance with a differential input signal is better than with a single-ended signal. For this reason, singleended operation is not recommended. However, if single ended-operation is required and the resulting
performance degradation is acceptable, one of the analog inputs should be connected to the d.c. mid point
voltage of the driven input. The peak-to-peak differential input signal at the driven input pin should be twice the
reference voltage to maximize SNR and SINAD performance (Figure 35b). For example, set VREF to 1.0V, bias
VIN− to 1.5V and drive VIN+ with a signal range of 0.5V to 2.5V.
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Because very large input signal swings can degrade distortion performance, better performance with a singleended input can be obtained by reducing the reference voltage when maintaining a full-range output. Table 1 and
Table 2 indicate the input to output relationship of the ADC14L020.
Table 1. Input to Output Relationship – Differential Input
VIN+
VIN−
Binary Output
2’s Complement Output
VCM − VREF/2
VCM + VREF/2
00 0000 0000 0000
10 0000 0000 0000
VCM − VREF/4
VCM + VREF/4
01 0000 0000 0000
11 0000 0000 0000
VCM
VCM
10 0000 0000 0000
00 0000 0000 0000
VCM + VREF/4
VCM − VREF/4
11 0000 0000 0000
01 0000 0000 0000
VCM + VREF/2
VCM − VREF/2
11 1111 1111 1111
01 1111 1111 1111
Table 2. Input to Output Relationship – Single-Ended Input
VIN
VIN−
Binary Output
2’s Complement Output
VCM − VREF
VCM
00 0000 0000 0000
10 0000 0000 0000
VCM − VREF/2
VCM
01 0000 0000 0000
11 0000 0000 0000
VCM
VCM
10 0000 0000 0000
00 0000 0000 0000
VCM + VREF/2
VCM
11 0000 0000 0000
01 0000 0000 0000
VCM + VREF
VCM
11 1111 1111 1111
01 1111 1111 1111
+
Driving the Analog Inputs
The VIN+ and the VIN− inputs of the ADC14L020 consist of an analog switch followed by a switched-capacitor
amplifier. The capacitance seen at the analog input pins changes with the clock level, appearing as 11 pF when
the clock is low, and 4.5 pF when the clock is high.
As the internal sampling switch opens and closes, current pulses occur at the analog input pins, resulting in
voltage spikes at the signal input pins. As a driving amplifier attempts to counteract these voltage spikes, a
damped oscillation may appear at the ADC analog input. Do not attempt to filter out these pulses. Rather, use
amplifiers to drive the ADC14L020 input pins that are able to react to these pulses and settle before the switch
opens and another sample is taken. The LMH6702 LMH6628, LMH6622 and the LMH6655 are good amplifiers
for driving the ADC14L020.
To help isolate the pulses at the ADC input from the amplifier output, use RCs at the inputs, as can be seen in
Figure 37 . These components should be placed close to the ADC inputs because the input pins of the ADC is
the most sensitive part of the system and this is the last opportunity to filter that input.
For Nyquist applications the RC pole should be at the ADC sample rate. The ADC input capacitance in the
sample mode should be considered when setting the RC pole. For wideband undersampling applications, the RC
pole should be set at about 1.5 to 2 times the maximum input frequency to maintain a linear delay response.
A single-ended to differential conversion circuit is shown in Figure 38. Table 3 gives resistor values for that circuit
to provide input signals in a range of 1.0V ±0.5V at each of the differential input pins of the ADC14L020.
Table 3. Resistor Values for Circuit of Figure 38
SIGNAL RANGE
R1
R2
R3
R4
R5, R6
0 - 0.25V
open
0Ω
124Ω
1500Ω
1000Ω
0 - 0.5V
0Ω
openΩ
499Ω
1500Ω
499Ω
±0.25V
100Ω
698Ω
100Ω
698Ω
499Ω
Input Common Mode Voltage
The input common mode voltage, VCM, should be in the range of 0.5V to 2.0V and be a value such that the peak
excursions of the analog signal does not go more negative than ground or more positive than 2.6V. See
Reference Pins.
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DIGITAL INPUTS
Digital TTL/CMOS compatible inputs consist of CLK, PD, and DF/DCS.
CLK
The CLK signal controls the timing of the sampling process. Drive the clock input with a stable, low jitter clock
signal in the range indicated in the Electrical Table with rise and fall times of 2 ns or less. The trace carrying the
clock signal should be as short as possible and should not cross any other signal line, analog or digital, not even
at 90°.
The CLK signal also drives an internal state machine. If the CLK is interrupted, or its frequency too low, the
charge on internal capacitors can dissipate to the point where the accuracy of the output data will degrade. This
is what limits the minimum sample rate.
The clock line should be terminated at its source in the characteristic impedance of that line. Take care to
maintain a constant clock line impedance throughout the length of the line. Refer to Application Note AN-905
(SNLA035) for information on setting characteristic impedance.
It is highly desirable that the the source driving the ADC CLK pin only drive that pin. However, if that source is
used to drive other things, each driven pin should be a.c. terminated with a series RC to ground, as shown in
Figure 37, such that the resistor value is equal to the characteristic impedance of the clock line and the capacitor
value is
(6)
where tPD is the signal propagation rate down the clock line, "L" is the line length and ZO is the characteristic
impedance of the clock line. This termination should be as close as possible to the ADC clock pin but beyond it
as seen from the clock source. Typical tPD is about 150 ps/inch (60 ps/cm) on FR-4 board material. The units of
"L" and tPD should be the same (inches or centimeters).
The duty cycle of the clock signal can affect the performance of the A/D Converter. Because achieving a precise
duty cycle is difficult, the ADC14L020 has a Duty Cycle Stabilizer which can be enabled using the DF/DCS pin. It
is designed to maintain performance over a clock duty cycle range of 20% to 80%.
PD
The PD pin, when high, holds the ADC14L020 in a power-down mode to conserve power when the converter is
not being used. The power consumption in this state is 15 mW. The output data pins are undefined and the data
in the pipeline is corrupted while in the power down mode.
The Power Down Mode Exit Cycle time is determined by the value of the components on pins 30, 31 and 32 and
is about 280 µs with the recommended components on the VRP, VRM and VRN reference bypass pins. These
capacitors loose their charge in the Power Down mode and must be recharged by on-chip circuitry before
conversions can be accurate. Smaller capacitor values allow slightly faster recovery from the power down mode,
but can result in a reduction in SNR, SINAD and ENOB performance.
DF/DCS
Duty cycle stablization and output data format are selectable using this quad state function pin. When enabled,
duty cycle stabilization can compensate for clock inputs with duty cycles ranging from 20% to 80% and generate
a stable internal clock, improving the performance of the part.
With DF/DCS = VA the output data format is offset binary and duty cycle stabilization is applied to the clock. With
DF/DCS = 0 the output data format is 2's complement and duty cycle stabilization is applied to the clock. With
DF/DCS = VRM the output data format is 2's complement and duty cycle stabilization is not used. If DF/DCS is
floating, the output data format is offset binary and duty cycle stabilization is not used. While the sense of this pin
may be changed "on the fly," doing this is not recommended as the output data could be erroneous for a few
clock cycles after this change is made.
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OUTPUTS
The ADC14L020 has 14 TTL/CMOS compatible Data Output pins. Valid data is present at these outputs while
the PD pins is low. Data should be captured with the CLK signal. Depending on the setup and hold time
requirements of the receiving circuit (ASIC), either the rising edge or the falling edge of the CLK signal can be
used to latch the data. Generally, rising-edge capture would maximize setup time with minimal hold time; while
falling-edge-capture would maximize hold time with minimal setup time. However, actual timing for the fallingedge case depends greatly on the CLK frequency and both cases also depend on the delays inside the ASIC.
Refer to the tOD spec in the AC Electrical Characteristics table.
Be very careful when driving a high capacitance bus. The more capacitance the output drivers must charge for
each conversion, the more instantaneous digital current flows through VDR and DR GND. These large charging
current spikes can cause on-chip ground noise and couple into the analog circuitry, degrading dynamic
performance. Adequate bypassing, limiting output capacitance and careful attention to the ground plane will
reduce this problem. Additionally, bus capacitance beyond the specified 15 pF/pin will cause tOD to increase,
making it difficult to properly latch the ADC output data. The result could be an apparent reduction in dynamic
performance.
To minimize noise due to output switching, minimize the load currents at the digital outputs. This can be done by
connecting buffers (74ACQ541, for example) between the ADC outputs and any other circuitry. Only one driven
input should be connected to each output pin. Additionally, inserting series resistors of about 33Ω at the digital
outputs, close to the ADC pins, will isolate the outputs from trace and other circuit capacitances and limit the
output currents, which could otherwise result in performance degradation. See Figure 37.
1.8 to VD Volts
+3.3V
CHOKE
2 x 0.1 PF
0.1 PF
+
10 PF
10 PF
1
32
31
0.1 PF
330
10 PF 30
0.1 PF
V DR
6
VD
VA
VA
1k
21
10 PF
5
29
0.1 PF
VREF
V RM
VRP
V RN
0.1 PF
ADC14L020
D7
D6
D5
**
33
2
T1
1
33
PD
DF/DCS
100 pF
T1-6T
** may be replaced by Ckt in Fig. 5
See
Text
8
PD
2
DF/DCS
2 CLK
74LVTH162374
DRGND
2
2
6
D4
D3
D2
D1
(LSB) D0
VIN+
3 V IN
100 pF
0.1 PF
27
26
25
24
23
22
19
18
17
16
15
14
13
12
Output
Word
CLK
20
3
4
DGND
0.1 PF
9
1
4
7 AGND
AGND
29 AGND
VIN
(MSB) D13
D12
D11
D10
D9
D8
49
49
Clock In
Figure 37. Application Circuit using Transformer Drive Circuit
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2V
R2, 1%
R1, 1%
5k, 1%
SIGNAL
INPUT
+
U2B
U1A
-
51
+
*
2.4k
*
33
to
V IN+
100 pF
5k, 1%
R5, 1%
5k, 1%
5k, 1%
*
R4, 1%
R3, 1%
2.4k
*
5k, 1%
+
U1B
U2A
-
-
100 pF
+
33
to
V IN-
5k, 1%
*
R6, 1%
5k, 1%
The ground
*
connections
indicated with an "*" should
be connected to a common
point in the analog ground
plane.
Amplifiers:
two LMH6622s or
LMH6655s
Figure 38. Differential Drive Circuit of Figure 37
POWER SUPPLY CONSIDERATIONS
The power supply pins should be bypassed with a 10 µF capacitor and with a 0.1 µF ceramic chip capacitor
close to each power pin. Leadless chip capacitors are preferred because they have low series inductance.
As is the case with all high-speed converters, the ADC14L020 is sensitive to power supply noise. Accordingly,
the noise on the analog supply pin should be kept below 100 mVP-P.
No pin should ever have a voltage on it that is in excess of the supply voltages, not even on a transient basis. Be
especially careful of this during power turn on and turn off.
The VDR pin provides power for the output drivers and may be operated from a supply in the range of 2.4V to VD.
This can simplify interfacing to lower voltage devices and systems. Note, however, that tOD increases with
reduced VDR. DO NOT operate the VDR pin at a voltage higher than VD.
LAYOUT AND GROUNDING
Proper grounding and proper routing of all signals are essential to ensure accurate conversion. Maintaining
separate analog and digital areas of the board, with the ADC14L020 between these areas, is required to achieve
specified performance.
The ground return for the data outputs (DR GND) carries the ground current for the output drivers. The output
current can exhibit high transients that could add noise to the conversion process. To prevent this from
happening, the DR GND pins should NOT be connected to system ground in close proximity to any of the
ADC14L020's other ground pins.
Capacitive coupling between the typically noisy digital circuitry and the sensitive analog circuitry can lead to poor
performance. The solution is to keep the analog circuitry separated from the digital circuitry, and to keep the
clock line as short as possible.
Digital circuits create substantial supply and ground current transients. The logic noise thus generated could
have significant impact upon system noise performance. The best logic family to use in systems with A/D
converters is one which employs non-saturating transistor designs, or has low noise characteristics, such as the
74LS, 74HC(T) and 74AC(T)Q families. The worst noise generators are logic families that draw the largest
supply current transients during clock or signal edges, like the 74F and the 74AC(T) families.
The effects of the noise generated from the ADC output switching can be minimized through the use of 33Ω
resistors in series with each data output line. Locate these resistors as close to the ADC output pins as possible.
24
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Since digital switching transients are composed largely of high frequency components, total ground plane copper
weight will have little effect upon the logic-generated noise. This is because of the skin effect. Total surface area
is more important than is total ground plane area.
Generally, analog and digital lines should cross each other at 90° to avoid crosstalk. To maximize accuracy in
high speed, high resolution systems, however, avoid crossing analog and digital lines altogether. It is important to
keep clock lines as short as possible and isolated from ALL other lines, including other digital lines. Even the
generally accepted 90° crossing should be avoided with the clock line as even a little coupling can cause
problems at high frequencies. This is because other lines can introduce jitter into the clock line, which can lead to
degradation of SNR. Also, the high speed clock can introduce noise into the analog chain.
Best performance at high frequencies and at high resolution is obtained with a straight signal path. That is, the
signal path through all components should form a straight line wherever possible.
Be especially careful with the layout of inductors. Mutual inductance can change the characteristics of the circuit
in which they are used. Inductors should not be placed side by side, even with just a small part of their bodies
beside each other.
The analog input should be isolated from noisy signal traces to avoid coupling of spurious signals into the input.
Any external component (e.g., a filter capacitor) connected between the converter's input pins and ground or to
the reference input pin and ground should be connected to a very clean point in the ground plane.
All analog circuitry (input amplifiers, filters, reference components, etc.) should be placed in the analog area of
the board. All digital circuitry and I/O lines should be placed in the digital area of the board. The ADC14L020
should be between these two areas. Furthermore, all components in the reference circuitry and the input signal
chain that are connected to ground should be connected together with short traces and enter the ground plane at
a single, quiet point. All ground connections should have a low inductance path to ground.
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DYNAMIC PERFORMANCE
To achieve the best dynamic performance, the clock source driving the CLK input must be free of jitter. Isolate
the ADC clock from any digital circuitry with buffers, as with the clock tree shown in Figure 39. The gates used in
the clock tree must be capable of operating at frequencies much higher than those used if added jitter is to be
prevented.
Best performance will be obtained with a differential input drive, compared with a single-ended drive, as
discussed in Single-Ended Operation and Driving the Analog Inputs.
As mentioned in LAYOUT AND GROUNDING, it is good practice to keep the ADC clock line as short as possible
and to keep it well away from any other signals. Other signals can introduce jitter into the clock signal, which can
lead to reduced SNR performance, and the clock can introduce noise into other lines. Even lines with 90°
crossings have capacitive coupling, so try to avoid even these 90° crossings of the clock line.
Figure 39. Isolating the ADC Clock from other Circuitry with a Clock Tree
COMMON APPLICATION PITFALLS
Driving the inputs (analog or digital) beyond the power supply rails. For proper operation, all inputs should
not go more than 100 mV beyond the supply rails (more than 100 mV below the ground pins or 100 mV above
the supply pins). Exceeding these limits on even a transient basis may cause faulty or erratic operation. It is not
uncommon for high speed digital components (e.g., 74F and 74AC devices) to exhibit overshoot or undershoot
that goes above the power supply or below ground. A resistor of about 47Ω to 100Ω in series with any offending
digital input, close to the signal source, will eliminate the problem.
Do not allow input voltages to exceed the supply voltage, even on a transient basis. Not even during power up or
power down.
Be careful not to overdrive the inputs of the ADC14L020 with a device that is powered from supplies outside the
range of the ADC14L020 supply. Such practice may lead to conversion inaccuracies and even to device
damage.
Attempting to drive a high capacitance digital data bus. The more capacitance the output drivers must
charge for each conversion, the more instantaneous digital current flows through VDR and DR GND. These large
charging current spikes can couple into the analog circuitry, degrading dynamic performance. Adequate
bypassing and maintaining separate analog and digital areas on the pc board will reduce this problem.
Additionally, bus capacitance beyond the specified 15 pF/pin will cause tOD to increase, making it difficult to
properly latch the ADC output data. The result could, again, be an apparent reduction in dynamic performance.
The digital data outputs should be buffered (with 74ACQ541, for example). Dynamic performance can also be
improved by adding series resistors at each digital output, close to the ADC14L020, which reduces the energy
coupled back into the converter output pins by limiting the output current. A reasonable value for these resistors
is 33Ω.
Using an inadequate amplifier to drive the analog input. As explained in Signal Inputs, the capacitance seen
at the input alternates between 11 pF and 4.5 pF, depending upon the phase of the clock. This dynamic load is
more difficult to drive than is a fixed capacitance.
26
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If the amplifier exhibits overshoot, ringing, or any evidence of instability, even at a very low level, it will degrade
performance. A small series resistor at each amplifier output and a capacitor at the analog inputs (as shown in
Figure 38) will improve performance. The LMH6702 and the LMH6628 have been successfully used to drive the
analog inputs of the ADC14L020.
Also, it is important that the signals at the two inputs have exactly the same amplitude and be exactly 180º out of
phase with each other. Board layout, especially equality of the length of the two traces to the input pins, will
affect the effective phase between these two signals. Remember that an operational amplifier operated in the
non-inverting configuration will exhibit more time delay than will the same device operating in the inverting
configuration.
Operating with the reference pins outside of the specified range. As mentioned in Reference Pins, when
using an external reference, VREF should be in the range of
0.8V ≤ VREF ≤ 1.2V
(7)
Operating outside of these limits could lead to performance degradation.
Inadequate network on Reference Bypass pins (VRP, VRN, and VRM). As mentioned in Reference Pins, these
pins should be bypassed with 0.1 µF capacitors to ground, and 10 µF capacitor should be connected between
pins VRP and VRN.
Using a clock source with excessive jitter, using excessively long clock signal trace, or having other
signals coupled to the clock signal trace. This will cause the sampling interval to vary, causing excessive
output noise and a reduction in SNR and SINAD performance .
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REVISION HISTORY
Changes from Revision C (April 2013) to Revision D
•
28
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 27
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PACKAGE OPTION ADDENDUM
www.ti.com
26-Sep-2015
PACKAGING INFORMATION
Orderable Device
Status
(1)
ADC14L020CIVY/NOPB
ACTIVE
Package Type Package Pins Package
Drawing
Qty
LQFP
NEY
32
250
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
Op Temp (°C)
Device Marking
(4/5)
-40 to 85
ADC14L0
20CIVY
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
26-Sep-2015
Addendum-Page 2
PACKAGE OUTLINE
NEY0032A
LQFP - 1.6 mm max height
SCALE 1.800
PLASTIC QUAD FLATPACK
7.1
6.9
25
32
PIN 1 ID
B
24
1
7.1
6.9
9.4
TYP
8.6
17
8
A
9
16
0.27
32X
0.17
0.2
OPTIONAL:
SHARP CORNERS EXCEPT
PIN 1 ID CORNER
28X 0.8
4X 5.6
C A B
SEE DETAIL A
1.6 MAX
C
SEATING PLANE
0.09-0.20
TYP
0.25
GAGE PLANE
(1.4)
0.1
0.15
0.05
0.75
0.45
0 -7
DETAIL A
DETAIL A
SCALE: 12
TYPICAL
4219901/A 10/2016
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. Reference JEDEC registration MS-026.
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EXAMPLE BOARD LAYOUT
NEY0032A
LQFP - 1.6 mm max height
PLASTIC QUAD FLATPACK
SYMM
25
32
32X (1.6)
1
24
32X (0.4)
SYMM
28X (0.8)
(8.5)
17
8
(R0.05) TYP
16
9
(8.5)
LAND PATTERN EXAMPLE
SCALE:8X
0.07 MAX
ALL AROUND
METAL
SOLDER MASK
OPENING
0.07 MIN
ALL AROUND
SOLDER MASK
METAL UNDER
SOLDER MASK
SOLDER MASK
DEFINED
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
4219901/A 10/2016
NOTES: (continued)
4. Publication IPC-7351 may have alternate designs.
5. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
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EXAMPLE STENCIL DESIGN
NEY0032A
LQFP - 1.6 mm max height
PLASTIC QUAD FLATPACK
SYMM
25
32
32X (1.6)
1
24
32X (0.4)
SYMM
28X (0.8)
(8.5)
8
17
(R0.05) TYP
16
9
(8.5)
SOLDER PASTE EXAMPLE
SCALE 8X
4219901/A 10/2016
NOTES: (continued)
6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
7. Board assembly site may have different recommendations for stencil design.
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