Texas Instruments | ADC16V130 16-Bit, 130 MSPS A/D Converter with LVDS Outputs (Rev. E) | Datasheet | Texas Instruments ADC16V130 16-Bit, 130 MSPS A/D Converter with LVDS Outputs (Rev. E) Datasheet

Texas Instruments ADC16V130 16-Bit, 130 MSPS A/D Converter with LVDS Outputs (Rev. E) Datasheet
ADC16V130
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SNAS458E – NOVEMBER 2008 – REVISED MARCH 2013
ADC16V130 16-Bit, 130 MSPS A/D Converter With LVDS Outputs
Check for Samples: ADC16V130
FEATURES
APPLICATIONS
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1
2
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Dual Supplies: 1.8V and 3.0V Operation
On Chip Automatic Calibration During PowerUp
Low Power Consumption
Multi-Level Multi-Function Pins for CLK/DF and
PD
Power-Down and Sleep Modes
On Chip Precision Reference and Sample-andHold Circuit
On Chip Low Jitter Duty-Cycle Stabilizer
Offset Binary or 2's Complement Data Format
Full Data Rate LVDS Output Port
64-pin WQFN Package (9x9x0.8, 0.5mm PinPitch)
KEY SPECIFICATIONS
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Resolution: 16 Bits
Conversion Rate: 130 MSPS
SNR
– (fIN = 10MHz): 78.5 dBFS (Typ)
– (fIN = 70MHz): 77.8 dBFS (Typ)
– (fIN = 160MHz): 76.7 dBFS (Typ)
SFDR
– (fIN = 10MHz): 95.5 dBFS (Typ)
– (fIN = 70MHz): 92.0 dBFS (Typ)
– (fIN = 160MHz): 90.6 dBFS (Typ)
Full Power Bandwidth: 1.4 GHz (Typ)
Power Consumption
– Core: 650 mW (Typ)
– LVDS Driver: 105 mW (Typ)
– Total: 755 mW (Typ)
Operating Temperature Range: -40°C ~ 85°C
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High IF Sampling Receivers
Multi-carrier Base Station Receivers
– GSM/EDGE, CDMA2000, UMTS, LTE, and
WiMax
Test and Measurement Equipment
Communications Instrumentation
Data Acquisition
Portable Instrumentation
DESCRIPTION
The ADC16V130 is a monolithic high performance
CMOS analog-to-digital converter capable of
converting analog input signals into 16-bit digital
words at rates up to 130 Mega Samples Per Second
(MSPS). This converter uses a differential, pipelined
architecture with digital error correction and an onchip sample-and-hold circuit to minimize power
consumption and external component count while
providing excellent dynamic performance. Automatic
power-up calibration enables excellent dynamic
performance and reduces part-to-part variation, and
the ADC16V130 could be re-calibrated at any time by
asserting and then de-asserting power-down. An
integrated low noise and stable voltage reference and
differential reference buffer amplifier easies board
level design. On-chip duty cycle stabilizer with low
additive jitter allows wide duty cycle range of input
clock without compromising its dynamic performance.
A unique sample-and-hold stage yields a full-power
bandwidth of 1.4 GHz. The digital data is provided via
full data rate LVDS outputs – making possible the 64pin, 9mm x 9mm WQFN package. The ADC16V130
operates on dual power supplies +1.8V and +3.0V
with a power-down feature to reduce the power
consumption to very low levels while allowing fast
recovery to full operation.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2013, Texas Instruments Incorporated
ADC16V130
SNAS458E – NOVEMBER 2008 – REVISED MARCH 2013
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Block Diagram
CLK+
DUTY CYCLE
STABILIZER
CLK-
34
VIN+
16BIT HIGH SPEED
PIPELINE ADC
SHA
VIN-
ERROR CORRECTION
LOGIC
SDR LVDS
BUFFER
2
DO+/-, OR+/OUTCLK+/-
VRN
VRM
VRP
INTERNAL
REFERENCE
MULTI-LEVEL
FUNCTION
PD
CLK/DF
VREF
CALIBRATION ENGINE
CLK_SEL/DF
2
VA1.8
AGND
VIN-
VIN+
AGND
VA3.0
VRM
VREF
AGND
VA3.0
OR+
OR-
D15+
D15-
D14+
D14-
64
63
62
61
60
59
58
57
56
55
54
53
52
51
50
49
Connection Diagram
1
EXPOSED PADDLE ON BOTTOM
OF PACKAGE, PIN 0
48
D13+
47
D13-
VA3.0
2
AGND
3
46
D12+
VRN
4
45
D12-
VRN
5
44
VDR
VRP
6
43
DRGND
VRP
7
42
D11+
AGND
8
41
D11-
VA1.8
9
40
D10+
CLK+
ADC16V130
(Top View)
28
29
30
31
32
D6-
D6+
D7-
D7+
OUTCLK-
D5+
33
27
16
26
DO+
D5-
OUTCLK+
25
34
D4-
15
D4+
DO-
24
D8-
VDR
35
23
14
22
D8+
D3+
36
DRGND
13
PD
21
VAD1.8
D3-
D9-
20
37
19
12
D2-
AGND
D2+
D9+
18
11
D1+
D10-
38
17
39
D1-
10
CLK-
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PIN DESCRIPTIONS
Pin No.
Symbol
Equivalent Circuit
Function and Connection
ANALOG I/O
61
VIN+
VA3.0
62
Differential analog input pins. The differential full-scale input signal
level is 2.4Vpp as default. Each input pin signal centered on a
common mode voltage, VCM.
VIN-
AGND
6,7
Upper reference voltage.
This pin should not be used to source or sink current. The decoupling
capacitor to AGND (low ESL 0.1μF) should be placed very close to
the pin to minimize stray inductance. VRP needs to be connected to
VRN through a low ESL 0.1μF and a low ESR 10μF capacitors in
parallel.
VRP
VA3.0
VRM
VA3.0
4,5
VRN
VRN
VREF
VA3.0
58
VRP
VRM
AGND
IDC
VREF
AGND
CLK+
VA3.0
CLK−
10 k:
11
VA3.0
VA1.8
10 k:
10
Common mode voltage
The decoupling capacitor to AGND (low ESL 0.1μF) should be placed
as close to the pin as possible to minimize stray inductance. It is
recommended to use VRM to provide the common mode voltage for
the differential analog inputs.
Internal reference voltage output / External reference voltage input.
By default, this pin is the output for the internal 1.2V voltage
reference. This pin should not be used to sink or source current and
should be decoupled to AGND with a 0.1μF, low ESL capacitor. The
decoupling capacitors should be placed as close to the pins as
possible to minimize inductance and optimize ADC performance. The
size of decoupling capacitor should not be larger than 0.1μF,
otherwise dynamic performance after power-up calibration can drop
due to the long VREF settling.
This pin can also be used as the input for a low noise external
reference voltage. The output impedance for the internal reference at
this pin is 9 kΩ and this can be overdriven provided the impedance of
the external source is <<9 kΩ. Careful decoupling is just as essential
when an external reference is used. The 0.1µF low ESL decoupling
capacitor should be placed as close to this pin as possible.
The Input differential voltage swing is equal to 2 * VREF.
VA3.0
57
Lower reference voltage.
This pin should not be used to source or sink current. The decoupling
capacitor to AGND (low ESL 0.1μF) should be placed very close to
the pin to minimize stray inductance. VRN needs to be connected to
VRP through a low ESL 0.1μF and a low ESR 10μF capacitors in
parallel.
Differential clock input pins. DC biasing is provided internally. For
single-ended clock mode, drive CLK+ through AC coupling while
decoupling CLK- pin to AGND.
AGND
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PIN DESCRIPTIONS (continued)
Pin No.
Symbol
Equivalent Circuit
Function and Connection
DIGITAL I/O
15 –
25 –
35 –
45 –
22
32
42
52
D0+/- to D3+/D4+/- to D7+/D8+/- to D11+/D12+/- to
D15+/-
LVDS Data Output. The 16-bit digital output of the data converter is
provided on these ports in a full data rate manner. A 100 Ω
termination resistor must be placed between each pair of differential
signals at the far end of the transmission line.
VA3.0
53, 54
OR+/-
D+
AGND
33, 34
VA3.0
VDR
D
DB
DB
D
DRGND
D-
AGND
OUTCLK+/-
14
VA3.0
This is a four-state pin controlling two parameters: input clock
selection and output data format.
CLK_SEL/DF = VA3.0, then CLK+ and CLK− are configured as a
differential clock input and the output data format is 2's complement.
CLK_SEL/DF = VA3.0 * (2/3), then CLK+ and CLK− are configured as
a differential clock input and the output data format is offset binary.
CLK_SEL/DF = VA3.0 * (1/3), then CLK+ is configured as a singleended clock input and CLK− should be tied to AGND. The output data
format is 2's complement.
CLK_SEL/DF = AGND, then CLK+ is configured as a single-ended
clock input and CLK− should be tied to AGND. The output data format
is offset binary.
AGND
CLK_SEL/DF
Output Clock. This pin is used to clock the output data. It has the
same frequency as the sampling clock. One word of data is output in
each cycle of this signal. A 100 Ω termination resistor must be placed
between the differential clock signals at the far end of the transmission
line. The rising edge of this signal should be used to capture the
output data. See the detail Section on Timing Diagrams .
This is a three-state pin.
PD = VA3.0, then Power Down is enabled. In the Power Down state,
only the reference voltage circuitry remains active and power
dissipation is reduced.
PD =VA3.0 * (2/3), then Sleep mode is enabled. In Sleep mode is
similar to Power Down mode - it consumes more power but has a
faster recovery time.
PD = AGND, then Normal operation mode is turned on.
PD
1
Over-Range Indicator. Active High.
This output is set High when analog input signal exceeds full scale of
16 bit conversion range (<0,> 65535). This signal is asserted
coincidently with the over-range data word. A 100 Ω termination
resistor must be placed between the differential signals at the far end
of the transmission.
POWER SUPPLIES
2, 55, 59
VA3.0
Analog Power
3.0V Analog Power Supply. These pins should be connected to a
quiet source and should be decoupled to AGND with 0.1μF capacitors
located close to the power pins.
9, 64
VA1.8
Analog Power
1.8V Analog Power Supply. These pins should be connected to a
quiet source and should be decoupled to AGND with 0.1μF capacitors
located close to the power pins.
13
VAD1.8
Analog/Digital Power
1.8V Analog/Digital Power Supply. These pins should be connected to
a quiet source and should be decoupled to AGND with 0.1μF
capacitors located close to the power pins.
0, 3, 8, 12,
56, 60, 63
AGND
Analog Ground
24, 44
VDR
Power
Output Driver Power Supply. This pin should be connected to a quiet
voltage source and be decoupled to DRGND with a 0.1μF capacitor
close to the power pins.
23, 43
DRGND
Ground
Output Driver Ground Return.
Analog Ground Return. The exposed pad (Pin 0) on back of the
package must be soldered to ground plane to ensure rated
performance.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
4
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Absolute Maximum Ratings (1) (2) (3)
−0.3V to 4.2V
Supply Voltage (VA3.0)
−0.3V to 2.35V
Supply Voltage (VA1.8, VAD1.8, VDR)
Voltage at any Pin except D0-D15, OVR, OUTCLK, CLK, VIN
−0.3V to (VA3.0 +0.3V)
(Not to exceed 4.2V)
Voltage at CLK, VIN Pins
-0.3V to (VDR +0.3V)
(Not to exceed 2.35V)
Voltage at D0-D15, OR, OUTCLK Pins
0.3V to (VDR + 0.3V)
(Not to exceed 2.35V)
Input Current at any pin (4)
5 mA
Storage Temperature Range
-65°C to +150°C
Maximum Junction Temp (TJ)
+150°C
Thermal Resistance (θJA)
20.4°C/W
Thermal Resistance (θJC)
ESD Rating (5)
1.4°C/W
Machine Model
200 V
Human Body Model
2000 V
Soldering process must comply with Reflow Temperature Profile specifications. Refer to www.ti.com/packaging. (6)
(1)
(2)
(3)
(4)
(5)
(6)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is ensured to be functional, but do not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance
characteristics may degrade when the device is not operated under the listed test conditions. Operation of the device beyond the
maximum Operating Ratings is not recommended.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
All voltages are measured with respect to GND = AGND = DRGND = 0V, unless otherwise specified.
When the input voltage at any pin exceeds the power supplies (that is, VIN < AGND, or VIN > VA), the current at that pin should be
limited to ±5 mA. The ±50 mA maximum package input current rating limits the number of pins that can safely exceed the power
supplies with an input current of ±5 mA to 10.
Human Body Model is 100 pF discharged through a 1.5 kΩ resistor. Machine Model is 220 pF discharged through 0 Ω.
Reflow temperature profiles are different for lead-free and non-lead-free packages.
Operating Ratings (1) (2)
Specified Temperature Range:
-40°C to +85°C
3.0V Analog Supply Voltage Range: (VA3.0)
+2.7V to +3.6V
1.8V Supply Voltage Range: VA1.8, VAD1.8, VDR
+1.7V to +1.9V
Clock Duty Cycle
(1)
(2)
30/70 %
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is ensured to be functional, but do not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance
characteristics may degrade when the device is not operated under the listed test conditions. Operation of the device beyond the
maximum Operating Ratings is not recommended.
All voltages are measured with respect to GND = AGND = DRGND = 0V, unless otherwise specified.
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Converter Electrical Characteristics
Unless otherwise specified, the following specifications apply: VA3.0 = +3.0V, VA1.8 = VAD1.8 = VDR = +1.8V, fCLK = 130 MSPS,
AIN = -1dBFS, LVDS Rterm = 100 Ω, CL = 5 pF. Typical values are for TA = 25°C. Boldface limits apply for TMIN ≤ TA ≤ TMAX.
All other limits apply for TA = 25°C, unless otherwise noted. (1)
Symbol
Parameter
Typical
Conditions
(2)
Limits
Units
(Limits)
16
Bits (min)
STATIC CONVERTER CHARACTERISTICS
Resolution with No Missing Codes
INL
Integral Non Linearity
±1.5
DNL
Differential Non Linearity
±0.45
LSB
LSB
PGE
Positive Gain Error
-4.2
%FS
NGE
Negative Gain Error
3.7
%FS
VOFF
Offset Error (VIN+ = VIN−)
0.12
%FS
Under Range Output Code
0
0
Over Range Output Code
65535
65535
REFERENCE AND ANALOG INPUT CHARACTERISTICS (3)
VCM
Common Mode Input Voltage
VRM
VREF
VRM±0.05
V
Reference Ladder Midpoint Output
Voltage
1.15
V
Internal Reference Voltage
1.20
V
2.4
VPP
Differential Analog Input Range
(1)
(2)
(3)
6
VRM is the common mode reference
voltage
Internal Reference
The inputs are protected as shown below. Input voltage magnitudes above VA3.0 or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings table. However, errors in the A/D conversion can occur if the input goes
above 2.6V or below GND as described in the Operating Ratings section see Figure 1.
Typical figures are at TA = 25°C and represent most likely parametric norms at the time of product characterization. The typical
specifications are not ensured.
The input capacitance is the sum of the package/pin capacitance and the sample and hold circuit capacitance.
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Dynamic Converter Electrical Characteristics
Unless otherwise specified, the following specifications apply: VA3.0 = +3.0V, VA1.8 = VAD1.8 = VDR = +1.8V, fCLK = 130 MSPS,
AIN = -1dBFS, LVDS Rterm = 100 Ω, CL = 5 pF. Typical values are for TA = 25°C. Boldface limits apply for TMIN ≤ TA ≤ TMAX.
All other limits apply for TA = 25°C, unless otherwise noted. (1)
Symbol
Parameter
Conditions
Typ
Resolution with no missing codes
DR
SNR
SFDR
THD
H2
H3
Spur-H2/3
SINAD
ENOB
(1)
(2)
Dynamic Range
Single-tone Spurious Free Dynamic
Range (2)
Total Harmonic Distortion (2)
Second-order Harmonic
(2)
Third-order Harmonic (2)
Worst Harmonic or Spurious Tone
excluding H2 and H3 (2)
Signal-to-Noise and Distortion Ratio (2)
Effective Number of Bits
Units
16
0V analog input is applied
Signal-to-Noise Ratio (2)
Limits
Bits
79
dBFS
Fin = 10 MHz
78.5
dBFS
Fin = 40 MHz
78.2
dBFS
Fin = 70 MHz
77.8
dBFS
Fin = 160 MHz
76.7
Fin = 240 MHz
75.6
dBFS
Fin = 10 MHz
95.5
dBFS
Fin = 40 MHz
91
dBFS
75.5
dBFS
Fin = 70 MHz
92
Fin = 160 MHz
90.6
dBFS
Fin = 240 MHz
85.3
dBFS
Fin = 10 MHz
-91.5
dBFS
Fin = 40 MHz
-88.4
dBFS
Fin = 70 MHz
-89.4
dBFS
Fin = 160 MHz
-87.1
Fin = 240 MHz
-82.8
dBFS
Fin = 10 MHz
-95.5
dBFS
Fin = 40 MHz
-104.1
dBFS
Fin = 70 MHz
-95.6
Fin = 160 MHz
-91.5
Fin = 240 MHz
-85.3
dBFS
Fin = 10 MHz
-98.3
dBFS
Fin = 40 MHz
-89.4
dBFS
Fin = 70 MHz
-92
dBFS
Fin = 160 MHz
-90.6
Fin = 240 MHz
-87.8
dBFS
Fin = 10 MHz
106
dBFS
Fin = 40 MHz
103.2
dBFS
Fin = 70 MHz
104.1
Fin = 160 MHz
101.5
Fin = 240 MHz
98.4
dBFS
Fin = 10 MHz
78.3
dBFS
Fin = 40 MHz
77.8
dBFS
Fin = 70 MHz
77.5
dBFS
Fin = 160 MHz
76.3
dBFS
Fin = 240 MHz
74.8
dBFS
Fin = 10 MHz
12.7
Bits
Fin = 40 MHz
12.6
Bits
Fin = 70 MHz
12.6
Bits
Fin = 160 MHz
12.4
Bits
Fin = 240 MHz
12.1
Bits
87
-81
dBFS
dBFS
dBFS
-88
-87
dBFS
dBFS
dBFS
94
dBFS
The inputs are protected as shown below. Input voltage magnitudes above VA3.0 or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings table. However, errors in the A/D conversion can occur if the input goes
above 2.6V or below GND as described in the Operating Ratings section see Figure 1.
This parameter is specified in units of dBFS – indicating the equivalent value that would be attained with a full-scale input signal.
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Dynamic Converter Electrical Characteristics (continued)
Unless otherwise specified, the following specifications apply: VA3.0 = +3.0V, VA1.8 = VAD1.8 = VDR = +1.8V, fCLK = 130 MSPS,
AIN = -1dBFS, LVDS Rterm = 100 Ω, CL = 5 pF. Typical values are for TA = 25°C. Boldface limits apply for TMIN ≤ TA ≤ TMAX.
All other limits apply for TA = 25°C, unless otherwise noted.(1)
Symbol
Parameter
Conditions
Full Power Bandwidth
Typ
-3dB Compression Point
Limits
1.4
Units
GHz
Power Supply Electrical Characteristics
Unless otherwise specified, the following specifications apply: VA3.0 = +3.0V, VA1.8 = VAD1.8 = VDR = +1.8V, fCLK = 130 MSPS,
AIN = -1dBFS, LVDS Rterm = 100 Ω, CL = 5 pF. Typical values are for TA = 25°C. Boldface limits apply for TMIN ≤ TA ≤ TMAX.
All other limits apply for TA = 25°C, unless otherwise noted. (1)
Symbol
Parameter
Conditions
Typical
Limits
Units
(Limits)
IA3.0
Analog 3.0V Supply Current
Full Operation (2)
174.5
208
mA (max)
IA1.8
Analog 1.8V Supply Current
Full Operation (2)
36
42
mA (max)
IAD1.8R
Digital 1.8V Supply Current
Full Operation
34
41
mA (max)
IDR
Output Driver Supply Current
Full Operation
58.3
Core Power Consumption
Excludes IDR (2)
650
773
mW (max)
Driver Power Consumption
Current drawn from the VDR supply; Fin =
10 MHz Rterm = 100Ω
105
mW
Normal operation; Fin = 10 MHz
Total Power Consumption
(1)
(2)
(2)
mA
755
mW
Power down state, with external clock
3
mW
Sleep state, with external clock
30
mW
The inputs are protected as shown below. Input voltage magnitudes above VA3.0 or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings table. However, errors in the A/D conversion can occur if the input goes
above 2.6V or below GND as described in the Operating Ratings section see Figure 1.
This parameter is ensured only at 25°C. For power dissipation over temperature range, refer to Core Power vs. Temperature plot in
Typical Performance Characteristics, Dynamic Performance
LVDS Electrical Characteristics
Unless otherwise specified, the following specifications apply: VA3.0 = +3.0V, VA1.8 = VAD1.8 = VDR = +1.8V, fCLK = 130 MSPS,
AIN = -1dBFS, LVDS Rterm = 100 Ω, CL = 5 pF. Typical values are for TA = 25°C. Boldface limits apply for TMIN ≤ TA ≤ TMAX.
All other limits apply for TA = 25°C, unless otherwise noted. (1)
Symbol
Parameter
Conditions
Min
Typ
Max
Units
mV
LVDS DC SPECIFICATIONS (apply to pins DO to D15, OR)
(1)
8
VOD
Output Differential Voltage
100 Ω Differential Load
175
250
325
VOS
Output Offset Voltage
100 Ω Differential Load
1.15
1.2
1.25
IOS
Output Short Circuit Current
0 Ω Differential Load
IOZ
Output Open Circuit Current
Termination is open
2.5
-20
±1
V
mA
20
µA
The inputs are protected as shown below. Input voltage magnitudes above VA3.0 or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings table. However, errors in the A/D conversion can occur if the input goes
above 2.6V or below GND as described in the Operating Ratings section see Figure 1.
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Timing Specifications
Unless otherwise specified, the following specifications apply: VIN = -1dBFS, AGND = DRGND = 0V, VA3.0= +3.0V, VA1.8 =
VAD1.8 = VDR = +1.8V, Internal VREF = +1.2V, fCLK = 130 MHz, VCM = VRM, CL = 5 pF, Single-Ended Clock Mode, Offset Binary
Format. Typical values are for TA = 25°C. Timing measurements are taken at 50% of the signal amplitude. Boldface limits
apply for TMIN ≤ TA ≤ TMAX. All other limits apply for TA = 25°C, unless otherwise noted. (1)
Parameter
Conditions
Typ
Input Clock Frequency
Minimum Clock Frequency
Units
130
MHz (max)
1
Data Output Setup Time (Tsu) (2)
Data Output Hold Time (Th)
Limits
(2)
MHz (min)
Measured @ Vdr/2; Fclk = 130 MHz.
3.3
2.5
nS (min)
Measured @ Vdr/2; Fclk = 130 MHz.
3.3
2.5
nS (min)
Pipeline Latency (3)
Clock
Cycles
11
Aperture Jitter
80
fS rms
Power-Up Time
From assertion of Power to specified level of
performance.
0.5+ 10 *(2 +2 )/FCLK
mS
Power-Down Recovery Time
From de-assertion of power down mode to
output data available.
0.1+ 103*(219+216)/FCLK
mS
Sleep Recovery Time
From de-assertion of sleep mode to output
data available.
100
μS
(1)
(2)
(3)
3
22
16
The inputs are protected as shown below. Input voltage magnitudes above VA3.0 or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings table. However, errors in the A/D conversion can occur if the input goes
above 2.6V or below GND as described in the Operating Ratings section see Figure 1.
This parameter is a function of the CLK frequency - increasing directly as the frequency is lowered. At frequencies less than 130 MHz,
use the following formulae to calculate the setup and hold times:
For Data and OR+/- Outputs: Tsu = ½*Tp – 0.5 ns (typical)
For Data and OR+/- Outputs: Th = ½*Tp – 0.5 ns (typical)
where Tp = CLK input period = OUTCLK period
Input signal is sampled with the falling edge of the CLK input.
VA3.0
To Internal
Circuitry
I/O
AGND
Figure 1.
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Specification Definitions
APERTURE DELAY is the time after the falling edge of the clock to when the input signal is acquired or held for
conversion.
APERTURE JITTER (APERTURE UNCERTAINTY) is the variation in aperture delay from sample to sample.
Aperture jitter manifests itself as noise in the output.
CLOCK DUTY CYCLE is the ratio of the time during one cycle that a repetitive digital waveform is high to the
total time of one period. The specification here refers to the ADC clock input signal.
COMMON MODE VOLTAGE (VCM) is the common DC voltage applied to both input terminals of the ADC.
CONVERSION LATENCY is the number of clock cycles between initiation of conversion and the time when data
is presented to the output driver stage. Data for any given sample is available at the output pins the Pipeline
Delay plus the Output Delay after the sample is taken. New data is available at every clock cycle, but the data
lags the conversion by the pipeline delay.
DIFFERENTIAL NON-LINEARITY (DNL) is the measure of the maximum deviation from the ideal step size of 1
LSB.
FULL POWER BANDWIDTH is a measure of the frequency at which the reconstructed output fundamental
drops 3 dB below its low frequency value for a full scale input.
GAIN ERROR is the deviation from the ideal slope of the transfer function. It can be calculated as:
Gain Error = Positive Full Scale Error − Negative Full Scale Error
(1)
It can also be expressed as Positive Gain Error and Negative Gain Error, which are calculated as:
PGE = Positive Full Scale Error - Offset Error NGE = Offset Error - Negative Full Scale Error
(2)
INTEGRAL NON LINEARITY (INL) is a measure of the deviation of each individual code from a best fit straight
line. The deviation of any given code from this straight line is measured from the center of that code value.
INTERMODULATION DISTORTION (IMD) is the creation of additional spectral components as a result of two
sinusoidal frequencies being applied to the ADC input at the same time. It is defined as the ratio of the power in
the intermodulation products to the total power in the original frequencies. IMD is usually expressed in dBFS.
LSB (LEAST SIGNIFICANT BIT) is the bit that has the smallest value or weight of all bits. This value is VFS/2n,
where “VFS” is the full scale input voltage and “n” is the ADC resolution in bits.
MISSING CODES are those output codes that will never appear at the ADC outputs. The ADC16V130 is ensured
not to have any missing codes.
MSB (MOST SIGNIFICANT BIT) is the bit that has the largest value or weight. Its value is one half of full scale.
NEGATIVE FULL SCALE ERROR is the difference between the actual first code transition and its ideal value of
½ LSB above negative full scale.
OFFSET ERROR is the difference between the two input voltages (VIN+ – VIN-) required to cause a transition from
code 32767LSB and 32768LSB with offset binary data format.
PIPELINE DELAY (LATENCY) See CONVERSION LATENCY.
POSITIVE FULL SCALE ERROR is the difference between the actual last code transition and its ideal value of
1½ LSB below positive full scale.
SIGNAL TO NOISE RATIO (SNR) is the ratio, expressed in dB, of the power of input signal to the total power of
all other spectral components below one-half the sampling frequency, not including harmonics and DC.
SIGNAL TO NOISE AND DISTORTION (SINAD) Is the ratio, expressed in dB, of the power of the input signal to
the total power of all of the other spectral components below half the clock frequency, including harmonics but
excluding DC.
SPURIOUS FREE DYNAMIC RANGE (SFDR) is the difference, expressed in dB, between the power of input
signal and the peak spurious signal power, where a spurious signal is any signal present in the output spectrum
that is not present at the input.
TOTAL HARMONIC DISTORTION (THD) is the ratio, expressed in dB, of the total power of the first seven
harmonic to the input signal power. THD is calculated as:
10
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THD = 20 log10
f 22 + f 32 + . . . + f 82
f 12
where
•
f12 is the power of the fundamental frequency and f22 through f82 are the powers of the first seven harmonics in
the output spectrum.
(3)
SECOND HARMONIC DISTORTION (2ND HARM or H2) is the difference expressed in dB, from the power of
its 2nd harmonic level to the power of the input signal.
THIRD HARMONIC DISTORTION (3RD HARM or H3) is the difference expressed in dB, from the power of
the 3nd harmonic level to the power of the input signal.
HIGHEST SPURIOUS EXCEPT H2 and H3 (Spur-H2/3) is the difference, expressed in dB, between the
power of input signal and the peak spurious signal power except H2 and H3, where a spurious signal is any
signal present in the output spectrum that is not present at the input.
Timing Diagrams
Sample N+12
Sample N+11
Vin
Sample N
tAD
TP
CLK+
CLKLatency
TP
OUTCLK+
OUTCLKTSU
Word N-1
Dx+/-, OR+/-
Th
Word N
Word N+1
Figure 2. Digital Output Timing
Transfer Characteristic
Figure 3. Transfer Characteristic (Offset Binary Format)
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Typical Performance Characteristics, DNL, INL
Unless otherwise noted, these specifications apply: VA3.0= +3.0V, VA1.8, VAD1.8, VDR = 1.8V, fCLK = 130 MSPS. Differential
Clock Mode, Offset Binary Format. LVDS Rterm = 100 Ω. CL = 5 pF. Typical values are at TA = +25°C. Fin = 10MHz with
–1dBFS.
12
DNL
INL
Figure 4.
Figure 5.
DNL vs.VA3.0
INL vs .VA3.0
Figure 6.
Figure 7.
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Typical Performance Characteristics, Dynamic Performance
Unless otherwise noted, these specifications apply: VA3.0= +3.0V, VA1.8, VAD1.8, VDR = 1.8V, fCLK = 130 MSPS. Differential
Clock Mode, Offset Binary Format. LVDS Rterm = 100 Ω. CL = 5 pF. Typical values are at TA = +25°C. Fin = 160MHz with
–1dBFS..
SNR, SINAD, SFDR vs. fIN
DISTORTION vs. fIN
Figure 8.
Figure 9.
SNR, SINAD, SFDR vs. VA3.0
DISTORTION vs. VA3.0
Figure 10.
Figure 11.
SNR, SINAD, SFDR vs. VAD1.8
DISTORTION vs. VAD1.8
Figure 12.
Figure 13.
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Typical Performance Characteristics, Dynamic Performance (continued)
Unless otherwise noted, these specifications apply: VA3.0= +3.0V, VA1.8, VAD1.8, VDR = 1.8V, fCLK = 130 MSPS. Differential
Clock Mode, Offset Binary Format. LVDS Rterm = 100 Ω. CL = 5 pF. Typical values are at TA = +25°C. Fin = 160MHz with
–1dBFS..
14
Spectral Response @ 10.11 MHz
Spectral Response @ 160.11 MHz
Figure 14.
Figure 15.
Spectral Response @ 40.11 MHz
Spectral Response @ 240.11 MHz
Figure 16.
Figure 17.
Spectral Response @ 70.11 MHz
Core Power vs. Temperature (Excludes IDR)
Figure 18.
Figure 19.
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FUNCTIONAL DESCRIPTION
Operating on dual +1.8 and +3.0V supplies, the ADC16V130 digitizes a differential analog input signals to 16
bits, using a differential pipelined architecture with error correction circuitry and an on-chip sample-and-hold
circuit to ensure maximum performance. The user has the choice of using an internal 1.2V stable reference, or
using an external 1.2V reference. Internal 1.2V reference has high output impedance of > 9 kΩ and can be easily
over-driven by external reference. Two multi-level multi-function pins can program data format, clock mode,
power down and sleep mode.
ADC Architecture
The ADC16V130 architecture consists of a highly linear and wide bandwidth sample-and-hold circuit, followed by
a switched capacitor pipeline ADC. Each stage of the pipeline ADC consists of low resolution flash sub-ADC and
an inter-stage multiplying digital-to-analog converter (MDAC), which is a switched capacitor amplifier with a fixed
stage signal gain and DC level shifting circuits. The amount of DC level shifting is dependent on sub-ADC digital
output code. 16bit final digital output is the result of the digital error correction logic, which receives digital output
of each stage including redundant bits to correct offset error of each sub-ADC.
APPLICATIONS INFORMATION
OPERATING CONDITIONS
We recommend that the following conditions be observed for operation of the ADC16V130:
2.7V ≤ VA3.0 ≤ 3.6V
1.7V ≤ VA1.8 ≤ 1.9V
1.7V ≤ VAD1.8 ≤ 1.9V
1.7V ≤ VDR ≤ 1.9V
5 MSPS ≤ FCLK ≤ 130 MSPS
VREF ≤ 1.2V
VCM = 1.15V (from VRM)
ANALOG INPUTS
Analog input circuit of the ADC16V130 is a differential switched capacitor sample-and-hold circuit (see Figure 20)
that provides optimum dynamic performance wide input frequency range with minimum power consumption. The
clock signal alternates sample mode (QS) and hold mode (QH). An integrated low jitter duty cycle stabilizer
ensures constant optimal sample and hold time over wide range of input clock duty cycle. The duty cycle
stabilizer is always turned on during normal operation.
During sample mode, analog signals (VIN+, VIN-) are sampled across two sampling capacitor (CS) while the
amplifier in the sample-and-hold circuit is idle. The dynamic performance of the ADC16V130 is likely determined
during sampling mode. The sampled analog inputs (VIN+, VIN-) are held during hold mode by connecting input
side of the sampling capacitors to output of the amplifier in the sample-and-hold circuit while driving pipeline ADC
core.
The signal source, which drives the ADC16V130, is recommended to have source impedance less than 100 Ω
over wide frequency range for optimal dynamic performance.
A shunt capacitor can be placed across the inputs to provide high frequency dynamic charging current during
sample mode and also absorb any switching charge coming from the ADC16V130. A shunt capacitor can be
placed across each input to GND for similar purpose. Smaller physical size and low ESR and ESL shunt
capacitor is recommended.
The value of shunt capacitor should be carefully chosen to optimize the dynamic performance at certain input
frequency range. Larger value shunt capacitors can be used for low input frequency range, but the value has to
be reduced at high input frequency range.
Balancing impedance at positive and negative input pin over entire signal path must be ensured for optimal
dynamic performance.
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QH
CS
VIN+
-
+
+
-
QS
QS
CS
VIN -
QH
Figure 20. Simplified Switched-Capacitor Sample-and-hold Circuit
Input Common Mode
The analog inputs of the ADC16V130 are not internally dc biased and the range of input common mode is very
narrow. Hence it is highly recommended to use the common mode voltage (VRM, typically 1.15V) as input
common mode for optimal dynamic performance regardless of DC and AC coupling applications. Input common
mode signal must be decoupled with low ESL 0.1μF at the far end of load point to minimize noise performance
degradation due to any coupling or switching noise between the ADC16V130 and input driving circuit.
Driving Analog Inputs
For low frequency applications, either a flux or balun transformer can convert single-ended input signal into
differential and drive the ADC16V130 without additive noise. An example is shown in Figure 21. VRM pin is used
to bias the input common mode by connecting the center tap of the transformer’s secondary ports. Flux
transformer is used for this example, but AC coupling capacitors should be added once balun type transformer is
used.
VIN +
R
C
R
ADC16V130
VIN -
VRM
0.1 PF
Figure 21. Transformer Drive Circuit for Low Input Frequency
Transformer has a characteristic of band pass filtering. It sets lower band limit by being saturated at frequencies
below a few MHz and sets upper frequency limit due to its parasitic resistance and capacitance. The transformer
core will be saturated with excessive signal power and it causes distortion as equivalent load termination
becomes heavier at high input frequencies. This is a reason to reduce shunt capacitors for high IF sampling
application to balance the amount of distortion caused by transformer and charge kick-back noise from the
device.
As input frequency goes higher with the input network in Figure 22, amplitude and phase unbalance increase
between positive and negative inputs (VIN+ and VIN-) due to the inherent impedance mismatch between the two
primary ports of the transformer while one is connected to the signal source and the other is connected to GND.
Distortion increases as the result.
16
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Cascaded transmission line transformers can be used for high frequency applications like high IF sampling base
station receiver channel. Transmission line transformer has less stray capacitance between primary and
secondary ports and so the amount of impedance at secondary ports is effectively less even with the given
inherent impedance mismatch on the primary ports. Cascading two transmission line transformers further
reduces the effective stray capacitance from the secondary of ports of the secondary transformer to primary ports
of first transformer, where impedance is mismatched. A transmission line transformer, for instance MABACT0040
from M/A-COM, with center tap on secondary port could further reduce amplitude and phase mismatch.
0.1 PF
R
VIN +
C1
C2
ADC16V130
0.1 PF
R
C2
VIN -
VRM
0.1 PF
Figure 22. Transformer Drive Circuit for High Input Frequency
Equivalent Input Circuit and Its S11
Input circuit of the ADC16V130 during sample mode is a differential switched capacitor as shown in Figure 23.
Bottom plate sampling switch is bootstrapped in order to reduce its turn on impedance and its variation across
input signal amplitude. Bottom plate sampling switches and top plate sampling switch are all turned off during
hold mode. The sampled analog input signal is processed throughout the following pipeline ADC core. Equivalent
impedance changes drastically between sample and hold mode while significant amount of charge injection
occurs during the transition between the two operating modes.
Distortion and SNR heavily rely on the signal integrity, impedance matching during sample mode and charge
injection while switching sampling switches.
VIN+
VIN-
Figure 23. Input Equivalent Circuit
A measured S11 of the input circuit of the ADC16V130 is shown in Figure 24 (Currently the figure is a simulated
one. It is subject to be changed later. Note that the simulated S11 closely matches with the measured S11). Up
to 500 MHz, it is predominantly capacitive loading with small stray resistance and inductance as shown in
Figure 24. An appropriate resistive termination at a given input frequency band has to be added to improve
signal integrity. Any shunt capacitor on analog input pin deteriorates signal integrity but it provides high frequency
charge to absorb the charge inject generated while sampling switches are toggling. A optimal shunt capacitor is
dependent on input signal frequency as well as impedance characteristic of analog input signal path including
components like transformer, termination resistor, DC coupling capacitors.
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Figure 24. S11 Curve of Input Circuit
CLOCK INPUT CONSIDERATIONS
Clock Input Modes
The ADC16V130 provides a low additive jitter differential clock receiver for optimal dynamic performance at wide
input frequency range. Input common mode of the clock receiver is internally biased at VA1.8/2 through a 10 kΩ
each to be driven by DC coupled clock input as shown in Figure 25. However while DC coupled clock input
drives CLK+ and CLK-, it is recommend the common mode (average voltage of CLK+ and CLK-) not to be higher
than VA1.8/2 in order to prevent substantial tail current reduction, which might cause lowered jitter performance.
Meanwhile, CLK+ and CLK- should not become lower than AGND. A high speed back-to-back diode connected
between CLK+ and CLK- could limit the maximum swing, but this could cause signal integrity concerns when the
diode turns on and reduce load impedance instantaneously.
A preferred differential clocking through a transformer coupled is shown in Figure 26. A 0.1μF decoupling
capacitor on the center tap of the secondary ports of a flux type transformer stabilizes clock input common mode.
Differential clocking increases the maximum amplitude of the clock input at the pins twice as large as that with
singled-ended mode as shown in Figure 27. Clock amplitude is recommended to be as large as possible while
CLK+ and CLK- both never exceed supply rails of VA1.8 and AGND. With a given equivalent input noise of the
differential clock receiver shown in Figure 25, larger clock amplitude at CLK+ and CLK- pins increases its slope
around zero-crossing point so that higher signal-to-noise could be obtained by reducing the noise contributed by
clock signal path.
VA1.8
CLK +
CLK 10k
10k
VA1.8
2
Figure 25. Equivalent Clock Receiver
The differential receiver of the ADC16V130 has excellent low noise floor but its bandwidth is wide as multiple
times of clock rate. The wide band noise folds back to nyquist frequency band in frequency domain at ADC
output. Increased slope of the input clock lowers the equivalent noise contributed by the differential receiver.
A band-pass filter (BPF) with narrow pass band and low insertion loss could be added on the clock input signal
path when wide band noise of clock source is noticeably large compared to the input equivalent noise of the
differential clock receiver.
18
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Load termination could be a combination of R and C instead of a pure R. This RC termination could improve
noise performance of clock signal path by filtering out high frequency noise through a low pass filter. The size of
R and C is dependent on the clock rate and slope of the clock input.
A LVPECL and/or LVDS driver could also drive the ADC16V130. However the full dynamic performance of the
ADC16V130 might not be achieved due to the high noise floor of the driving circuit itself especially in high IF
sampling application.
CLOCK
INPUT
CLK +
R
C
ADC16V130
CLK -
0.1 PF
Figure 26. Differential Clocking, Transformer Coupled
Singled-ended clock can drive CLK+ pin through a 0.1μF AC coupling capacitor while CLK- is decoupled to
AGND through a 0.1μF capacitor as shown in Figure 27.
0.1 P F
CLOCK
INPUT
CLK +
ADC 16 V 130
R
C
CLK 0.1 P F
Figure 27. Singled-Ended 1.8V Clocking, Capacitive AC Coupled
Duty Cycle Stabilizer
Highest operating speed with optimal performance could be only achieved with 50% of clock duty cycle because
the switched-capacitor circuit of the ADC16V130 is designed to have equal amount of settling time between each
stage. The maximum operating frequency could be reduced accordingly while clock duty cycle departs from 50%.
The ADC16V130 contains a duty cycle stabilizer that adjusts non-sampling (rising) clock edge to make the duty
cycle of the internal clock over 30 to 70% of input clock duty cycle. The duty cycle stabilizer is always on
because the noise and distortion performance are not affected at all. It is not recommended to use the
ADC16V130 at the clock frequencies less than 5 MSPS, at which the feedback loop in the duty cycle stabilizer
becomes unstable.
Clock Jitter vs. Dynamic Performance
High speed and high resolution ADCs require low noise clock input to ensure its full dynamic performance over
wide input frequency range. SNR (SNRFin) at a given input frequency (Fin) can be calculated by:
2
SNR Fin = 10 log10
A /2
VN2 +
2
(2SFin x Tj) / 2
with a given total noise power (VN2) of an ADC, total rms jitter (Tj), and input amplitude (A) in dBFS.
Clock signal path must be treated as an analog signal whenever aperture jitter affects the dynamic performance
of the ADC16V130. Power supplies for the clock drivers has to be separated from the ADC output drive supplies
to prevent modulated clock signal with the ADC digital output signals. Higher noise floor and/or increased
distortion/spur might result from any coupling noise from ADC digital output signals to analog input and clock
signals.
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In IF sampling applications, the signal-to-noise ratio is particularly affected by clock jitter as shown in Figure 28.
Tj is the integrated noise power of the clock signal divided by the slope of clock signal around tripping point.
Upper limit of the noise integration is independent of applications and set by the bandwidth of the clock signal
path. However lower limit of the noise integration highly relies on the applications. In base station receiver
channel applications, the lower limit is determined by channel bandwidth and space from an adjacent channel.
85
80
75
50fs
75fs
100fs
SNR (dBFS)
70
65
60
200fs
55
400fs
50
800fs
45
1.5ps
40
35
1
10
100
1000
INPUT FREQUENCY (MHz)
Figure 28. SNR with given Jitter vs. Input Frequency
CALIBRATION
Automatic calibration engine contained within the ADC16V130 improves dynamic performance and reduces its
part-to-part variation. Digital output signals including output clock (OUTCLK+/-) are all logic low while calibrating.
The ADC16V130 is automatically calibrated when the device is powered up. Optimal dynamic performance might
not be obtained if power-up time is longer than internal delay time (~32mS @ 130 MSPS clock rate). In this case,
the ADC16V130 could be re-calibrated by asserting and then de-asserting power down mode. Re-calibration is
recommended whenever operating clock rate changes.
VOLTAGE REFERENCE
A stable and low noise voltage reference and its buffer amplifier are built into the ADC16V130. The input full
scale is two times of VREF, which is same as VBG (On-chip bandgap output having 9 kΩ output impedance) as
well as VRP - VRN as shown in Figure 29. The input range can be adjusted by changing VREF either internally or
externally. An external reference with low output impedance can easily over-drive VREF pin. Default VREF is 1.2V.
Input common mode voltage (VRM) is a fixed voltage level of 1.15V. Maximum SNR can be achieved at maximum
input range of 1.2V VREF. Although the ADC16V130 dynamic and static performance is optimized at VREF of 1.2V,
reducing VREF can improve SFDR performance with sacrificing SNR of the ADC16V130.
Reference Decoupling
It is highly recommended to place external decoupling capacitors connected to VRP, VRN, VRM and VREF pins as
close to pins as possible. The external decoupling capacitor should have minimal ESL and ESR. During normal
operation, inappropriate external decoupling with large ESL and/or ESR capacitors increase settling time of ADC
core and results in lower SFDR and SNR performance. VRM pin may be loaded up to 1mA for setting input
common mode. The remaining pins should not be loaded. Smaller capacitor values might result in degraded
noise performance. Decoupling capacitor on VREF pin must not exceed 0.1μF, heavier decoupling on this pin will
cause improper calibration during power-up. All reference pins except VREF have very low output impedance.
Driving these pins via low output impedance external circuit for long time period might damage the device.
20
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ADC16V130
9k
1.15V
VRP
VRN
VRM
VREF
0.1 F
0.1 F
10 F
0.1 F
10 F
0.1 F
0.1 F
Figure 29. Internal References and their Decoupling
While VRM pin is used to set input common mode level via transformer, a smaller serial resistor could be placed
on the signal path to isolate any switching noise interfering between ADC core and input signal. The serial
resistor introduces voltage error between VRM and VCM due to charge injection while sampling switches toggling.
The serial resistance should not be larger than 50 Ω.
All grounds associated with each reference and analog input pins should be connected to a solid and quite
ground on PC board. Coupling noise from digital outputs and their supplies to the reference pins and their ground
can cause degraded SNR and SFDR performance.
LAYOUT AND GROUNDING
Proper grounding and proper routing of all signals are essential to ensure accurate conversion. Maintaining
separate analog and digital areas of the board, with the ADC16V130 between these areas, is required to achieve
specified performance.
Even though LVDS output reduces ground bounding during its transition, the positive and negative signal path
has to be well matched and their trace should be kept as short as possible. It is recommend to place LVDS
repeater between the ADC16V130 and digital data receiver block to isolate coupling noise from receiving block
while the length of the traces are long or the noise level of the receiving block is high.
Capacitive coupling between the typically noisy digital circuitry and the sensitive analog circuitry can lead to poor
performance. The solution is to keep the analog circuitry separated from the digital circuitry, and to keep the
clock line as short as possible.
Since digital switching transients are composed largely of high frequency components, total ground plane copper
weight will have little effect upon the logic-generated noise. This is because of the skin effect. Total surface area
is more important than its total ground plane area.
Generally, analog and digital lines should not be crossing each. However whenever it is inevitable, make sure
that these lines are crossing each other at 90° to minimize cross talk. Digital output and output clock signals must
be separated from analog input, references and clock signals unconditionally to ensure the maximum
performance from ADC16V130. Any coupling might result degraded SNR and SFDR performance especially at
high IF applications.
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Be especially careful with the layout of inductors and transformers. Mutual inductance can change the
characteristics of the circuit in which they are used. Inductors and transformers should not be placed side by
side, even with just a small part of their bodies beside each other. For instance, place transformers for the analog
input and the clock input at 90° to one another to avoid magnetic coupling. It is recommended to place the
transformers of input signal path on the top plate, but the transformer of clock signal path on the bottom plate.
Every critical analog signal path like analog inputs and clock inputs must be treated as a transmission line and
should have a solid ground return path with a small loop.
The analog input should be isolated from noisy signal traces to avoid coupling of spurious signals into the input.
Any external component (e.g., a filter capacitor) connected between the converter’s input pins and ground or to
the reference pins and ground should be connected to a very clean point in the ground plane. All analog circuitry
(input amplifiers, filters, reference components, etc.) should be placed in the analog area of the board. All digital
circuitry and dynamic I/O lines should be placed in the digital area of the board. The ADC16V130 should be
between these two areas. Furthermore, all components in the reference circuitry and the input signal chain that
are connected to ground should be connected together with short traces and enter the ground plane at a single,
quiet point. All ground connections should have a low inductance path to ground.
Ground return current path can be well managed when supply current path is precisely controlled and ground
layer is continuous and placed next to the supply layer. This is because of the proximity effect. Ground return
current path with a large loop will cause electro-magnetic coupling and results in poor noise performance. Not
that even if there is a large plane for a current path, high frequency current path is not spread evenly over the
large plane, but only takes a path with lowest impedance. Instead of large plane, using thick trace for supplies
makes it easy to control return current path. It is recommended to place supply next to GND layer with thin
dielectric for smaller ground return loop. Proper location and size of decoupling capacitors provide short and
clean return current path.
SUPPLIES AND THEIR SEQUENCE
There are four supplies for the ADC16V130; one 3.0V supply VA3.0 and three 1.8V supplies VA1.8, VAD1.8 and VDR.
It is recommended to separate VDR from VA1.8 supplies, any coupling from VDR to rest of supplies and analog
signals could cause lower SFDR and noise performance. When VA1.8 and VDR are both from same supply source,
coupling noise can be mitigated by adding ferrite-bead on VDR supply path.
The user can use different decoupling capacitors to provide current over wide frequency range. The decoupling
capacitors should be located close to the point of entry and close to the supply pins with minimal trace length. A
single ground plane is recommended because separating ground under the ADC16V130 could cause
unexpected long return current path.
VA3.0 supply must turn on before VA1.8 and/or VDR reaches single diode turn-on voltage level. If this supply
sequence is reversed, excessive amount of current will flow through VA3.0 supply. Ramp rate of VA3.0 supply must
be kept less than 60V/mS (i.e., 60μS for 3.0V supply) in order to prevent excessive surge current through ESD
protection devices.
The exposed pad (Pin #0) on the bottom of the package should be soldered to AGND in order to get optimal
noise performance. The exposed pad is a solid ground for the device and also is heat sinking path.
22
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Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: ADC16V130
ADC16V130
www.ti.com
SNAS458E – NOVEMBER 2008 – REVISED MARCH 2013
REVISION HISTORY
Changes from Revision D (March 2013) to Revision E
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 22
Submit Documentation Feedback
Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: ADC16V130
23
PACKAGE OPTION ADDENDUM
www.ti.com
13-Sep-2014
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
ADC16V130CISQ/NOPB
ACTIVE
WQFN
NKD
64
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 85
ADC16V130
ADC16V130CISQE/NOPB
ACTIVE
WQFN
NKD
64
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 85
ADC16V130
ADC16V130CISQX/NOPB
ACTIVE
WQFN
NKD
64
2000
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 85
ADC16V130
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
13-Sep-2014
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
16-Apr-2017
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
ADC16V130CISQ/NOPB
Package Package Pins
Type Drawing
WQFN
NKD
64
ADC16V130CISQE/NOPB WQFN
NKD
ADC16V130CISQX/NOPB WQFN
NKD
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
250
178.0
16.4
9.3
9.3
1.3
12.0
16.0
Q1
64
250
178.0
16.4
9.3
9.3
1.3
12.0
16.0
Q1
64
2000
330.0
16.4
9.3
9.3
1.3
12.0
16.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
16-Apr-2017
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
ADC16V130CISQ/NOPB
WQFN
NKD
64
250
210.0
185.0
35.0
ADC16V130CISQE/NOPB
WQFN
NKD
64
250
210.0
185.0
35.0
ADC16V130CISQX/NOPB
WQFN
NKD
64
2000
367.0
367.0
38.0
Pack Materials-Page 2
PACKAGE OUTLINE
NKD0064A
WQFN - 0.8 mm max height
SCALE 1.600
WQFN
9.1
8.9
B
A
PIN 1 INDEX AREA
0.5
0.3
9.1
8.9
0.3
0.2
DETAIL
OPTIONAL TERMINAL
TYPICAL
0.8 MAX
C
SEATING PLANE
(0.1)
TYP
7.2 0.1
SEE TERMINAL
DETAIL
32
17
60X 0.5
33
16
4X
7.5
1
PIN 1 ID
(OPTIONAL)
48
64
49
64X
0.5
0.3
64X
0.3
0.2
0.1
0.05
C A
C
B
4214996/A 08/2013
NOTES:
1. All linear dimensions are in millimeters. Dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. The package thermal pad must be soldered to the printed circuit board for thermal and mechanical performance.
www.ti.com
EXAMPLE BOARD LAYOUT
NKD0064A
WQFN - 0.8 mm max height
WQFN
(
7.2)
SYMM
64X (0.6)
64X (0.25)
64
SEE DETAILS
49
1
48
60X (0.5)
SYMM
(8.8)
(1.36)
TYP
( 0.2) VIA
TYP
8X (1.31)
16
33
32
17
(1.36) TYP
8X (1.31)
(8.8)
LAND PATTERN EXAMPLE
SCALE:8X
0.07 MIN
ALL AROUND
0.07 MAX
ALL AROUND
METAL
SOLDER MASK
OPENING
SOLDER MASK
OPENING
NON SOLDER MASK
DEFINED
(PREFERRED)
METAL
SOLDER MASK
DEFINED
SOLDER MASK DETAILS
4214996/A 08/2013
NOTES: (continued)
4. This package is designed to be soldered to a thermal pad on the board. For more information, refer to QFN/SON PCB application note
in literature No. SLUA271 (www.ti.com/lit/slua271).
www.ti.com
EXAMPLE STENCIL DESIGN
NKD0064A
WQFN - 0.8 mm max height
WQFN
SYMM
64X (0.6)
64X (0.25)
(1.36) TYP
64
49
1
48
(1.36)
TYP
60X (0.5)
SYMM
(8.8)
METAL
TYP
33
16
32
17
25X
(1.16)
(8.8)
SOLDERPASTE EXAMPLE
BASED ON 0.125mm THICK STENCIL
EXPOSED PAD
65% PRINTED SOLDER COVERAGE BY AREA
SCALE:10X
4214996/A 08/2013
NOTES: (continued)
5. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
www.ti.com
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