Texas Instruments | 8-Channel Variable-Gain Amplifier (VGA) With Octal High-Speed ADC (Rev. D) | Datasheet | Texas Instruments 8-Channel Variable-Gain Amplifier (VGA) With Octal High-Speed ADC (Rev. D) Datasheet

Texas Instruments 8-Channel Variable-Gain Amplifier (VGA) With Octal High-Speed ADC (Rev. D) Datasheet
AFE5801
www.ti.com
SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
8-Channel Variable-Gain Amplifier (VGA) With Octal High-Speed ADC
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FEATURES
RELATED DEVICES
•
•
1
•
•
•
•
•
•
•
Eight Variable-Gain Amplifiers (VGA)
– Eight Differential Buffered Inputs With 2Vpp
Maximum Swing
– 5.5nV/√Hz VCA Input Noise (31dB Gain)
– Variable Gain, –5dB to 31dB With 0.125dB
or 1dB Steps
– Digital Gain Control
Third-Order Antialiasing Filter With
Programmable Cutoff Frequency (7.5, 10, or
14MHz)
Clamping
Analog-to-Digital Converter (ADC)
– Octal Channel, 12Bit, 65MSPS
– Internal and External Reference Support
– No External Decoupling Required for
References
– Serial LVDS Outputs
1.8V and 3.3V Supplies
50mW Total Power per Channel at 30MSPS
58mW Total Power per Channel at 50MSPS
64QFN Package (9mm × 9mm)
APPLICATIONS
•
Imaging: Ultrasound, PET
AFE5851: 16-Channel VGA + ADC,
32.5MSPS/Channel
DESCRIPTION
The AFE5801 is an analog front end, targeting
applications where the power and level of integration
are critical. The device contains eight variable-gain
amplifiers (VGA), each followed by a high-speed (up
to 65MSPS) ADC, for a total of eight ADCs per
device.
Each of the eight differential inputs is buffered,
accepts up to 2Vpp maximum input swing, and is
followed by a VGA with a gain range from –5dB to
31dB. The VGA gain is digitally controlled, and the
gain curves versus time can be stored in memory
integrated within the device using the serial interface.
A selectable clamping and antialias low-pass filter
(with 3dB attenuation at 7.5, 10, or 14MHz) is also
integrated between VGA and ADC, for every channel.
The VGA/antialias filter outputs are differential
(limited to 2Vpp) and drive the onboard 12bit,
65MSPS ADC. The ADC also scales down its power
consumption should a lower sampling rate be
selected. The ADC outputs are serialized in LVDS
streams, which further minimizes power and board
area.
The AFE5801 is available in a 64-pin QFN package
(9mm × 9mm) and is specified over the full industrial
temperature range (–40°C to 85°C).
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2010, Texas Instruments Incorporated
AFE5801
SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
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This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
AVDD3
SDOUT
SCLK
SDATA
SEN
RESET
PDN
VREF_IN
TGC SYNC
BLOCK DIAGRAM
DVDD18
Time Gain Block
AVDD18
Control
VCM
Serial
Interface
Memory
LVDS
AAF
IN1
ADC 1
Serializer
D1P
D1M
ADC 2
Serializer
D2P
D2M
ADC 3
Serializer
D3P
D3M
AAF
IN2
AAF
IN3
·
·
·
·
·
·
·
·
·
·
·
·
AAF
IN8
ADC 8
Serializer
D8P
D8M
Frame Clock
FCLKP
FCLKM
Bit Clock
DCLKP
DCLKM
fCLKIN
CLKINP
CLKINM
PLL
AVSS
2
DVSS
6 ´ fCLKIN
B0328-01
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
PINOUT
D1P
D1M
DVSS
DVDD18
SDOUT
SDATA
SEN
RESET
SCLK
PDN
SYNC
AVSS
NC
AVSS
AVDD18
VCM
RGC PACKAGE
(TOP VIEW)
IN1P
64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49
48
1
D2P
IN1M
2
47
D2M
IN2P
3
46
D3P
IN2M
4
45
D3M
IN3P
5
44
D4P
IN3M
6
43
D4M
42
DCLKP
41
DCLKM
40
FCLKP
FCLKM
IN4P
7
IN4M
8
IN5P
9
IN5M
10
39
IN6P
11
38
D5P
IN6M
12
37
D5M
AFE5801
QFN-64
9mm × 9mm
D8P
D8M
DVDD18
DVSS
DVDD18
NC
NC
AVDD18
D7P
D7M
VREF_IN
34
33
16
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
AVSS
15
CLKINM
IN8P
IN8M
AVSS
D6M
CLKINP
D6P
35
AVDD18
36
14
VCM
13
AVDD3
IN7P
IN7M
P0056-13
TERMINAL FUNCTIONS
Name
NUMBER
AVSS
20, 23, 61, 63
DESCRIPTION
Analog ground
AVDD18
19, 24, 62
1.8V analog supply voltage
AVDD3
18
3.3V analog supply voltage
CLKINP, CLKINM
D1P/M–D8P/M
21, 22
50–43, 38–31
Differential clock input pins. Single-ended clock is also supported. See the Clock Inputs section.
LVDS outputs for channels 1 to 8
DCLKM, DCLKP
41, 42
LVDS bit clock output
DVSS
29, 52
Digital ground
DVDD18
(1)
1.8V LVDS buffer supply voltage
FCLKM, FCLKP
39, 40
LVDS frame clock output
IN1P/M–IN8P/M
1–16
Differential analog input pins for channels 1 to 8
NC
28
, 30, 51
26, 27, 60
Do not connect
PDN
59
Global power-down control input (active-high). 100kΩ pulldown resistor
RESET
57
Hardware reset pin (active-high). 100kΩ pulldown resistor
SCLK
56
Serial interface clock input. 100kΩ pulldown resistor
SDATA
55
Serial interface data input. 100kΩ pulldown resistor
SDOUT
53
Serial interface data readout
SEN
54
Serial interface enable. 100kΩ pullup resistor
SYNC
58
TGC/VGA synchronization signal input. 100kΩ pulldown resistor
VCM
17, 64
VREF_IN
Thermal pad
(1)
25
Bottom of package
Common-mode output pins for possible bias of the analog input signals
Reference input in the external reference mode
Connect to AVSS
Pin 28 can be connected to the 1.8V or 3.3V supply, whichever is easier for user. It does not affect the performance.
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AFE5801
SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
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PACKAGING/ORDERING INFORMATION (1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA,
QUANTITY
AFE5801
QFN-64 (2)
RGC
–40°C to 85°C
AFE5801
AFE5801IRGCT
Tape/reel, 250
AFE5801
(2)
RGC
–40°C to 85°C
AFE5801
AFE5801IRGCR
Tape/reel, 2000
(1)
(2)
QFN-64
For the most-current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com.
For the thermal pad size on the package, see the mechanical drawings at the end of this document.
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
VALUE
UNIT
AVDD3 to AVSS
–0.3 to 3.8
V
AVDD18 to AVSS
–0.3 to 2.2
V
DVDD18 to DVSS
–0.3 to 2.2
V
Voltage between AVSS and DVSS
–0.3 to 0.3
V
Analog input pins (INiP, INiM) to AVSS
–0.3V to MIN (3.6V, AVDD3 + 0.3V)
V
VREF_IN to AVSS
–0.3 to 2.2
V
VCLKP, VCLKM to AVSS
–0.3 to 2.2
V
Digital control pins to DVSS: PDN, RESET, SCLK, SDATA, SEN, SYNC
–0.3 to 3.6
V
125
°C
–60 to 150
°C
TJ
Maximum operating junction temperature
Tstg
Storage temperature range
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
THERMAL CHARACTERISTICS
PARAMETER
TYP
UNIT
qJA
0 LFM air flow
TEST CONDITIONS
23.17
°C/W
qJC
2-oz. (0.071-mm thick) copper trace and pad soldered directly to a JEDEC-standard four-layer
3-in. × 3-in. (7.62-cm ×7.62-cm) PCB.
22.1
°C/W
RECOMMENDED OPERATING CONDITIONS
MIN
TA
Ambient temperature
TYP
–40
MAX
UNIT
85
°C
SUPPLIES
AVDD3
Analog supply voltage (VGA)
3
3.3
3.6
V
AVDD18
Analog supply voltage (ADC)
1.7
1.8
1.9
V
DVDD18
Digital supply voltage (ADC, LVDS)
1.7
1.8
1.9
V
VCM + 0.5
V
ANALOG INPUTS
INiP, INiM
Input voltage range to AVSS
VCM – 0.5
VREF_IN in external reference mode
1.35
VCM load
4
1.4
3
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1.45
V
mA
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
RECOMMENDED OPERATING CONDITIONS (continued)
MIN
TYP
MAX
UNIT
CLOCK INPUT
fCLKIN
Input clock frequency
5
Input clock duty cycle
40%
Sine wave, ac-coupled
VCLKP-CLKM
65 MSPS
50%
60%
0.5
LVPECL, ac-coupled
LVDS, ac-coupled
0.25
VCLKP
LVCMOS, single-ended, VCLKM to AVSS
VIH
High-level input voltage
VIL
Low-level input voltage
Vpp
1.6
Vpp
0.7
Vpp
1.8
Vpp
0.75 × AVDD18
V
0.25 × AVDD18
V
DIGITAL OUTPUT
CLOAD
External load capacitance from each output pin to DVSS
RLOAD
Differential load resistance (external) between the LVDS output pairs
5
pF
100
Ω
ELECTRICAL CHARACTERISTICS
Typical values are at 25°C, MIN and MAX values are across the full temperature range of Tmin = –40°C to Tmax = 85°C,
AVDD3 = 3.3V, AVDD18 = 1.8V, DVDD18 = 1.8V, –1dBFS analog input ac-coupled with 0.1mF, internal reference mode,
maximum-rated sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle, antialiasing filter set at
14MHz (3dB corner), output clamp disabled (1) and analog high-pass filter enabled, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
VARIABLE GAIN AMPLIFIER (VGA)
Maximum input voltage swing
VCM
Linear operation, from INP to INM
2
Common-mode voltage
Gain range
Vpp
1.6
Maximum gain – minimum gain
Maximum gain
36
29.5
Gain resolution
V
31
dB
32.5
dB
0.125
or 1
dB
Input resistance
From each input to dc bias level
5
kΩ
Input capacitance
Differential between the inputs
2
pF
ANTIALIAS FILTER (AAF)
7.5MHz filter selected
AAF cutoff frequency
10MHz filter selected
14
7.5MHz filter selected
10
–6dB
20
7.5MHz filter selected
18
–12dB
24
14MHz filter selected
30
7.5MHz filter selected
1.2
10MHz filter selected
At 3.2MHz
14MHz filter selected
(1)
14
14MHz filter selected
10MHz filter selected
In-band attenuation
10
14MHz filter selected
10MHz filter selected
AAF stop-band attenuation
7.5
–3dB
0.5
MHz
MHz
MHz
dB
0.2
Enabling clamping increases distortion values at high swings by about 2dB.
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ELECTRICAL CHARACTERISTICS (continued)
Typical values are at 25°C, MIN and MAX values are across the full temperature range of Tmin = –40°C to Tmax = 85°C,
AVDD3 = 3.3V, AVDD18 = 1.8V, DVDD18 = 1.8V, –1dBFS analog input ac-coupled with 0.1mF, internal reference mode,
maximum-rated sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle, antialiasing filter set at
14MHz (3dB corner), output clamp disabled (1) and analog high-pass filter enabled, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
FULL-CHANNEL CHARACTERISTICS
Gain matching
Gain error
Offset error
Input-referred noise voltage
SNR
Signal-to-noise ratio
HD2
Second-harmonic distortion
Across channels and parts
0.1
0.6
–5dB to 28dB gain
–1.2
±0.3
1.2
Gain > 28dB
–1.8
±0.5
1.8
31dB gain
–50
5MHz, 31dB VGA gain, low-noise mode
50
dB
dB
LSB
5
6.5 nV/√Hz
5MHz, 31dB VGA gain, default noise mode
5.5
nV/√Hz
–1dBFS ADC input, 6dB gain
66
dBc
–1dBFS ADC input, 17dB gain, fin = 2MHz
–75
–60
–1dBFS ADC input, 31dB gain, fin = 2MHz
–75
–60
–1dBFS ADC input, 17dB gain, fin = 2MHz
–65
–55
–1dBFS ADC input, 31dB gain, fin = 2MHz
–60
–52
dBc
HD3
Third-harmonic distortion
dBc
SFDR
Spurious-free dynamic range
–1dBFS ADC input, 17dB gain, fin = 2MHz
65
THD
Total harmonic distortion
–1dBFS ADC input, 17dB gain, fin = 2MHz
64
dBc
IMD
Intermodulation distortion
fin1 = 1MHz, fin2 = 2MHz, Ain1, in2 = –7dBFS, 30dB
VGA gain
–70
dBFS
Group delay variation
f = 100kHz to 14MHz, across gain settings and
channels
±3.5
ns
f = 100kHz to 14MHz, across channels
±1.5
dBc
Input overload recovery
≤ 6dB overload to within 1%
1
Clock
cycle
Clamp level
After amplification. Clamp enabled by default
3
dB
ADC number of bits
Crosstalk
12
Aggressor: fin = 3MHz, 1dB below ADC full-scale
Victims (channel next to aggressor channel): 50Ω
differential (between INiP and INiM)
92
dB
POWER
Total power dissipation
IAVDD3
AVDD3 current consumption
IAVDD18
AVDD18 current consumption
IDVDD18
DVDD18 current consumption
Power down
AC PSRR
(2)
6
Default noise mode
522
600
Low-noise mode
561
636
6
9
Default noise mode
198
222
Low-noise mode
220
244
81
100
See
(2)
Standby mode
Full power-down mode
Power-supply rejection ratio
64
8
30
mW
mA
mA
mA
mW
25
mW
dBc
Using digital modes like averaging, digital gain, digital HPF, etc., (see the Application Information section) might increase the DVDD18
current by about 60mA.
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DIGITAL CHARACTERISTICS (1)
The dc specifications refer to the condition where the digital outputs are not switching, but permanently at a valid logic level 0
or 1. Typical values are at 25°C, min and max values are across the full temperature range of Tmin = –40°C to Tmax = 85°C,
AVDD3 = 3.3V, AVDD18 = 1.8V, DVDD18 = 1.8V, external differential load resistance between the LVDS output pair RLOAD =
100Ω.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
DIGITAL INPUTS (PDN, RESET, SCLK, SDATA, SEN, SYNC)
High-level input voltage
1.4
3.6
Low-level input voltage
0.8
V
V
High-level input current
10
mA
Low-level input current
10
mA
4
pF
High-level output voltage
1375
mV
Low-level output voltage
1025
mV
Input capacitance
DIGITAL OUTPUTS (DiP, DiM)
Output differential voltage |VOD|
Output offset voltage VOS
Common-mode voltage of DiP and DiM
Output capacitance
Output capacitance inside the device, from either output to
DVSS.
270
380
490
mV
0.9
1.15
1.5
V
2
pF
1.8
V
DIGITAL OUTPUTS (SDOUT)
High-level output voltage
1.6
Low-level output voltage
0
Output impedance
(1)
0.2
50 ±25%
V
Ω
All LVDS specifications have been characterized but not tested at production. For clock input levels, see the Recommended Operating
Conditions table.
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OUTPUT INTERFACE TIMING
Typical values are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD = 1.8V, LVCMOS (single ended) clock, CLOAD = 5pF, RLOAD =
100Ω, IO = 3.5mA, unless otherwise noted. Minimum and maximum values are across the full temperature range TMIN =
–40°C to TMAX = 85°C.
PARAMETER
ta
tj
TEST CONDITIONS
Aperture delay
The delay in time between the rising edge of the input sampling
clock and the actual time at which the sampling occurs
Aperture delay matching
Across channels within the same device
MIN
TYP
0.7
3
Aperture jitter
MAX
ns
±150
ps
450
fs rms
Time to valid data after coming out of STANDBY mode
10
50
Time to valid data after coming out of PDN GLOBAL mode
50
200
Time to valid data after stopping and restarting the input clock
30
200
ADC latency
Default, after reset
11
tdelay
Data and frame clock delay
Input clock rising edge (zero cross) to frame clock rising edge (zero
cross) minus half the input clock period (T).
Δtdelay
Delay variation
At fixed supply and 20°C T difference
–1
tRISE
tFALL
Data rise time
Data fall time
Rise time measured from –100mV to 100mV
Fall time measured from 100mV to –100mV
10MHz < fCLKIN < 65MHz
0.1
tFCLKRISE
tFCLKFALL
Frame clock rise time
Frame clock fall time
Rise time measured from –100mV to 100mV
Fall time measured from 100mV to –100mV
10MHz < fCLKIN < 65MHz
Frame clock duty cycle
Zero crossing of the rising edge to zero crossing of the falling edge
Bit clock rise time
Bit clock fall time
Rise time measured from –100mV to 100mV
Fall time measured from 100mV to –100mV
10MHz < fCLKIN < 65MHz
Bit clock duty cycle
Zero crossing of the rising edge to zero crossing of the falling edge
10MHz < fCLKIN < 65MHz
Wake-up time
tDCLKRISE
tDCLKFALL
3
UNIT
4.7
ms
Input
clock
cycles
6.4
ns
1
ns
0.25
0.4
ns
0.1
0.25
0.4
ns
48%
50%
52%
0.1
0.2
0.35
44%
50%
56%
ns
Output Interface Timing (1)
fCLKIN, Input Clock Frequency
Period (T)
MHz
ns
65
15
50
40
30
(1)
8
Setup Time (tsu),
ns
Hold Time (th),
ns
tpdi = 0.5 × T + tdelay, ns
Zero-Cross Data to Zero-Cross
Clock (both edges)
Zero-Cross Clock to Zero-Cross
Data (both edges)
Input Clock Zero-Cross (rising
edge) to Frame Clock Zero-Cross
(rising edge)
MIN
TYP
0.35
0.6
20
0.5
25
0.75
33
20
10
MAX
MIN
TYP
0.3
0.6
MAX
MIN
TYP
12.3
0.8
0.5
0.8
14.6
1
0.75
1
17
1
1.4
1
1.4
21.2
50
1.7
2.1
1.7
2.1
29.5
100
3.8
4.2
3.8
4.2
54.7
MAX
FCLK timing is the same as for the output data lines. It has the same relation to DCLK as the data pins. Setup and hold are the same
for the data and the frame clock.
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Output Data
ChnOUT
Data rate = 12 x fCLKIN
Bit Clock
DCLK
Freq = 6 x fCLKIN
Frame Clock
FCLK
Freq = fCLKIN
Input Clock
CLKIN
Freq = fCLKIN
Input Signal
D1
(D10)
D13
(D2)
D0
(D11)
D11
(D0)
D9
(D2)
D8
(D3)
D7
(D4)
Data Bit in LSB First Mode
Data Bit in MSB First Mode
D10
(D1)
D5
(D6)
D4
(D7)
D2
(D9)
D1
(D10)
Bit Clock
D3
(D8)
D11
(D0)
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CHi out
tsu
DCLKM
DCLKP
D0
(D11)
11 Clock Cycles Latency
D10
(D1)
th
D9
(D2)
Dn
D8
(D3)
D7
(D4)
D5
(D6)
SAMPLE N – 1
D6
(D5)
Sample
N + 10
D2
(D9)
tsu
D3
(D8)
Dn + 1
D4
(D7)
tpdi
th
D1
(D10)
D0
(D11)
D11
(D0)
D10
(D1)
D9
(D2)
D8
(D3)
D7
(D4)
D5
(D6)
SAMPLE N
D6
(D5)
T
Sample
N + 11
D4
(D7)
D3
(D8)
D2
(D9)
D1
(D10)
D0
(D11)
D10
(D1)
T0434-01
SAMPLE
N+1
D11
(D0)
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Output Data Pair
SAMPLE N – 11
D6
(D5)
ta
Sample N
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
Figure 1. Timing Diagram
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TYPICAL CHARACTERISTICS
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
10
10
Ain = −1dBFS
Gain = 12dB
HD2 = −80.1dBc
HD3 = −64.2dBc
SFDR = 63.9dBc
SNR = 65.9dBFS
THD = 63.9dBc
−30
−10
Amplitude − dB
Amplitude − dB
−10
−50
−70
−90
−30
−50
−70
−90
−110
−110
0
5
10
15
20
25
30
35
f − Frequency − MHz
0
20
25
30
35
G002
Figure 3. FFT for 2MHz Input Signal and 30dB Gain
85°C
28
Coarse Gain = 31dB
24
Coarse Gain = 19dB
−40°C
20
16
(Ideal+1dB) Line
12
25°C
8
4
0
Coarse Gain = 7dB
(Ideal−1dB) Line
−4
−8
0.125
0.250
0.375
0.500
0.625
0.750
0
0.875
4
8
12
16
20
24
28
32
36
Gain Code
G021
Figure 4. Fine Gain vs Gain Code
G025
Figure 5. Gain vs Gain Code and Temperature
1.0
185
Output-Referred Noise − nV/√Hz
0.8
0.6
Gain Error − dB
15
32
FINE_GAIN Register Setting
25°C
0.4
−40°C
0.2
0.0
−0.2
−0.4
−0.6
85°C
−0.8
−1.0
165
Low-Noise Enabled
145
125
Low-Noise Disabled
105
85
65
45
−5
0
5
10
15
G − Gain − dB
20
25
30
−5
G026
Figure 6. Gain Error vs Gain Code and Temperature
10
10
G001
G − Gain − dB
32
30
28
26
24
22
20
18
16
14
12
10
8
6
0.000
5
f − Frequency − MHz
Figure 2. FFT for 2MHz Input Signal and 12dB Gain
G − Gain − dB
Ain = −1dBFS
Gain = 30dB
HD2 = −69.8dBc
HD3 = −59.5dBc
SFDR = 59.5dBc
SNR = 58.8dBFS
THD = 58.9dBc
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0
5
10
15
20
25
G − Gain − dB
30
G022
Figure 7. Output-Referred Noise vs Gain
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
TYPICAL CHARACTERISTICS (continued)
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
11
88
Input-Referred Noise − nV/√Hz
Input-Referred Noise − nV/√Hz
98
78
68
Low-Power Mode Enabled
58
48
38
28
18
Low-Power Mode Disabled
8
10
9
Low-Power Mode Disabled
8
7
6
Low-Power Mode Enabled
5
4
−6
−4
−2
0
2
4
6
8
10
12
14
16
G − Gain − dB
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31
18
G − Gain − dB
G023
Figure 8. Input-Referred Noise for Low Gains
−60
−70
Ain = −1dBFS
−74
fIN = 10MHz
HD2 − dBc
HD2 − dBc
fIN = 10MHz
−76
−70
−75
−80
−78
−80
−82
−84
fIN = 5MHz
−86
−85
fIN = 1MHz
−90
−10
0
10
fIN = 5MHz
−88
20
G − Gain − dB
30
40
fIN = 1MHz
−90
−10
0
10
20
30
G − Gain − dB
G003
Figure 10. HD2 Across Coarse Gain and Three fIN
(–1dBFS) (1)
(2)
Ain = −6dBFS
−72
−65
(1)
G024
Figure 9. Input-Referred Noise for High Gains
40
G004
Figure 11. HD2 Across Coarse Gain and Three fIN
(–6dBFS) (2)
For gains ≥5dB, the input amplitude is adjusted to give –1dBFS. At 5dB gain, input amplitude is 4dBm (corresponding to –1dBFS).
For gains less than 5dB, the input is kept constant at 4dBm.
For gains ≥0dB, the input amplitude is adjusted to give –6dBFS. At 0dB gain, input amplitude is 4dBm (corresponding to –6dBFS).
For gains less than 0dB, the input is kept constant at 4dBm.
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TYPICAL CHARACTERISTICS (continued)
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
−40
−30
Ain = −1dBFS
−35
Ain = −6dBFS
−50
fIN = 5MHz
fIN = 10MHz
−45
fIN = 1MHz
−50
fIN = 1MHz
−60
fIN = 10MHz
HD3 − dBc
HD3 − dBc
−40
fIN = 5MHz
−55
−70
−80
−60
−90
−65
−70
−10
0
10
20
30
−100
−10
40
G − Gain − dB
10
20
30
40
G − Gain − dB
G005
Figure 12. HD3 Across Coarse Gain and Three fIN
(–1dBFS)(1)
G006
Figure 13. HD3 Across Coarse Gain and Three fIN
(–6dBFS)(2)
−30
−20
fIN = 2MHz
−30
Gain = 30dB
−50
HD3 − dB
−50
−60
Gain = 6dB
−70
fIN = 2MHz
−40
Gain = 30dB
−40
HD2 − dB
0
−60
−70
−80
−80
Gain = 18dB
Gain = 18dB
−90
−90
Gain = 6dB
−100
−70
−60
−50
−40
−30
−20
−10
Output Amplitude − dBFS
−100
−70
0
G007
−70
2080
Output Offset − Code
Amplitude − dBc
2090
3MHz − Far
−90
−100
−110
−20
−10
0
G008
Analog HPF Enabled
2070
2060
2050
2040
10MHz − Adjacent
Digital HPF Enabled
−120
2030
0
2
4
6
Channel
8
10
−5
0
G009
Figure 16. Crosstalk (3)
12
−30
Analog HPF Disabled
10MHz − Far
(3)
−40
Figure 15. HD3 vs Output Amplitude
−60
3MHz − Adjacent
−50
Output Amplitude − dBFS
Figure 14. HD2 vs Output Amplitude
−80
−60
5
10
15
20
25
G − Gain − dB
30
G027
Figure 17. Output Offset vs Gain
–1dB signal applied on one channel at a time and output is observed on:
1. Adjacent channel - channel next to the aggressor channel, but not a shared channel
2. Far channel - all other channels (neither shared or adjacent)
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TYPICAL CHARACTERISTICS (continued)
1
0
−1
−2
−3
−4
−5
−6
−7
−8
−9
−10
−11
−12
20
K = 7dB
0
7.5MHz
−10
K = 4dB
−20
−30
K = 3dB
K = 6dB
K = 8dB
K = 9dB
K = 10dB
−60
2
4
6
8
10
12
14
16
18
fIN − Input Frequency − MHz
−70
0.0
20
0.5
1.0
1.5
2.0
2.5
3.0
f − Frequency − MHz
G010
Figure 18. Antialiasing Filter Frequency Response
G011
Figure 19. High-Pass Filter Options
450
0.60
fin = 2MHz
AVDD Power − Low-Noise Mode
400
0.55
350
P − Total Power − mW
AVDD3 Power − mW
K = 2dB
K = 5dB
−40
−50
10MHz
0
300
AVDD Power − Default
250
200
150
100
Total Power − Low-Noise Mode
0.50
0.45
0.40
Total Power − Default
0.35
50
fIN = 2MHz
0
0.30
0
10
20
30
40
50
f − Clock Frequency − MSPS
60
70
5
10
15
20
25
30
35
40
45
50
55
60
f − Clock Frequency − MSPS
G012
Figure 20. Analog Power vs Input-Clock Frequency
65
G013
Figure 21. Total Power vs Input-Clock Frequency
7
Count (Number of Channels) − %
4.0
6
Ocurrences − %
Analog Filter
10
14MHz
G − Gain − dB
Normalized Amplitude − dB
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
5
4
3
2
1
0
0.02 0.07 0.12 0.17 0.22 0.27 0.32 0.37 0.42 0.47 0.52
GAIN_MATCHING − dB
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
2005 2016 2027 2038 2049 2060 2071 2072 2083
Output Code − Offset
G030
Figure 22. Gain Matching Measured at a Single Gain (30
dB) as Peak-to-Peak Variation of Gain Across Channels
on Every Device and Measured at Three Temperatures.
Every Device at Each Temperature Counted as One Event
G031
Figure 23. Offset (Average Code) With Signal. Every
Channel Counted as One Event
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TYPICAL CHARACTERISTICS (continued)
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
2800
fIN = 2MHz
2600
Output Code
2400
2200
2000
1800
1600
1400
1200
1000
0
1000
2000
3000
4000
5000
6000
7000
8000
Sample
G014
Figure 24. TGC Sweep With Interpolation Disabled and High-Pass Filter Enabled
2800
fIN = 2MHz
2600
Output Code
2400
2200
2000
1800
1600
1400
1200
1000
0
1000
2000
3000
4000
5000
6000
7000
8000
Sample
G015
Figure 25. TGC Sweep With Interpolation Disabled and High-Pass Filter Disabled
2800
2600
fIN = 2MHz
Output Code
2400
2200
2000
1800
1600
1400
1200
1000
2500
3500
4500
5500
6500
7500
8500
9500
Sample
G016
Figure 26. TGC Sweep With Interpolation Enabled and High-Pass Filter Disabled
14
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
TYPICAL CHARACTERISTICS (continued)
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
0
Gain = 30dB
fIN1 = 2.92MHz at −7dBFS
fIN2 = 3.12MHz at −7dBFS
IMD3 = −70dBFS
Amplitude − dB
−20
−40
−60
−80
−100
−120
0
5
10
15
20
25
30
35
f − Frequency − MHz
G017
Figure 27. Intermodulation Distortion
67
Gain = 30dB
19
17
66
Analog HPF Disabled
65
15
13
SNR − dBFS
Input-Referred Noise − nV/√Hz
21
Default
11
High-Pass Digital Filter K = 4
9
61
60
59
3
0.0
58
−10
1.5
2.0
2.5
3.0
f − Frequency − MHz
20
30
40
G019
1
Normalized Amplitude − dB
65
SNR − dBFS
10
Figure 29. SNR vs Gain, Three fIN (–1dBFS)
66
fIN = 1MHz
fIN = 5MHz
63
fIN = 10MHz
62
61
60
58
−10
0
G018
67
59
Ain = −1dBFS
G − Gain − dB
Figure 28. Input-Referred Noise vs Frequency
64
fIN = 10MHz
62
5
1.0
fIN = 5MHz
63
7
0.5
fIN = 1MHz
64
fS = 65MSPS
Low-Pass Filter = 14MHz
Low-Noise Mode Enabled
0
−1
−2
Gain = −6dB
−3
Gain = 6dB
−4
Gain = 0dB
Gain = 24dB
−5
Gain = 18dB
Ain = −6dBFS
0
10
20
30
G − Gain − dB
40
−6
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
Normalized Frequency − f/fc
G020
Figure 30. SNR vs Gain Three fIN (–6dBFS)
Gain = 12dB
Gain = 30dB
G028
Figure 31. LPF Response Across Coarse Gain
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TYPICAL CHARACTERISTICS (continued)
All graphs are at 25°C, AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, –1dBFS analog input AC coupled with 0.1mF, internal
reference mode, maximum rated channel sampling frequency (65MSPS), LVCMOS (single-ended) clock, 50% duty cycle,
fIN = 2MHz, anti-aliasing filter set at 14MHz (3dB corner), output clamp disable and analog high-pass filter enabled. spacer
0
Normalized Amplitude − dB
1
fS = 65MSPS
Low-Pass Filter = 14MHz
Low-Noise Mode Disabled
0
−1
Gain = 6dB
−2
Gain = −6dB
DCLK
Gain = 12dB
−3
Gain = 30dB
Gain = 0dB
−4
Gain = 24dB
−5
−6
0.0
Gain = 18dB
0.2
0.4
0.6
0.8
1.0
Data
1.2
1.4
1.6
1.8
2.0
Normalized Frequency − f/fc
G029
Figure 32. LPF Response Across Coarse Gain
16
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Figure 33. LVDS Eye Pattern
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APPLICATION INFORMATION
THEORY OF OPERATION
The AFE5801 is a very low-power CMOS monolithic analog front end which includes an eight-channel
variable-gain amplifier (VGA) followed by an eight-channel, 12bit, high-speed pipeline analog-to-digital convereter
(ADC) based on switched-capacitor architecture.
Each of the eight VGA differential inputs is buffered and accepts a maximum swing of 2Vpp centered at a dc
level (VCM) of about 1.6V.
Each VGA has a gain range from –5dB to 31dB, and the gain is digitally controlled, with a resolution of 0.125dB.
Using the serial interface, the gain curves (common to all VGAs) versus time can be stored in the memory
integrated within the device.
A hardware sync input pin is available (SYNC). When a pulse is applied to this pin, all the VGAs in the device
start stepping through the selected time-gain curve at the same clock cycle. This sync can also be initiated by
software using the serial interface.
A selectable anti-alias low-pass filter (AAF) with 3dB attenuation at 7.5MHz, 10MHz, or 14MHz is also integrated,
together with clamping (which can be disabled).
The VGA/AAF can output 2Vpp differential swing without degradation in the specified linearity and can drive an
onboard 12bit ADC. After the input signals are captured by the sample-and-hold circuit, at the rising edge of the
clock, the samples are sequentially converted by a series of low-resolution stages. The outputs of the stages are
combined in a digital correction logic block to form the final 12bit word with a latency of 11 clock cycles, without
taking into account the delays introduced by the optional digital signal-processing functions. These functions are,
in this order, offset correction, channel averaging, digital gain, and high-pass filtering (see General-Purpose
Register Map for more details). The 12bit words of each channel are serialized and output as LVDS levels. For
slower operation speeds, the AFE5801 offers the possibility of multiplexing up to two input channels into one
LVDS output stream, reducing even further the power consumption and routing area. In addition to the data
streams, a bit clock and frame clock are also output. The frame clock is aligned with the 12bit word boundary.
Notice that for the correct operation of the device (see the Serial Interface section), a positive pulse must be
applied to the RESET pin. This sets the internal control registers to zero. There is, nevertheless, no need for any
type of power-up sequencing.
INPUT CONFIGURATION
The analog input for the AFE5801, Figure 34, consists of a differential analog buffer which has inputs biased to
1.6V (usually refered as common-mode voltage, VCM). The biasing is done with two internal resistors of 5kΩ. For
proper operation, the input signal should be either ac-coupled or have a commn-mode value equal to VCM. In the
case of ac coupling, the external input capacitors form a high-pass filter with the internal bias resistors (5kΩ), so,
the value of the capacitors should allow the lowest frequency of interest to pass with minimum attenuation. For
the typical frequencies used in ultrasound (>1MHz) a value of 10nF or bigger is recommended. If dc coupling is
preferred, the user can tap the VCM output pins to set the common-mode level of the input signal. The VCM output
should be connected to high-input-impedance circuits, as its driving capability is limited. Regardless of the
choosen input configuration, a capacitor of 100nF should be connected on each VCM input to AVSS.
For proper operation, the input signal should be in the recommended input range. The maximum input swing is
limited to 2Vpp before saturation/distortion fo the input stage occurs. As the input common mode (VCM) is about
1.6V, each input of the buffer should stay between 1.1V and 2.1V.
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INP
5 kW
Vcm
Channel
Input
Clamp
AC
Coupling
5 kW
INM
CM Buffer
Internal
Voltage
Reference
NOTE: Dashed area denotes one of eight channels.
Figure 34. Common-Mode Biasing of Input Pins
The maximum input swing is limited to 2Vpp before distortion/saturation of the input stage occurs. As the input dc
level (VCM) is about 1.6V, this means that each input of the VGA typically stays between 1.1V and 2.1V.
To drive the AFE5801 with a single-ended signal, the signal should be connected to one of the inputs (INiP, in
principle, to keep the polarity) and the other connected to AVSS through a 100nF capacitor. In fact, all the
AFE5801 inputs that must be connected to ground can be shorted together and connected to ground through a
single 100nF capacitor. The input is limited to 1Vpp (from 1.1V to 2.1V) and the performance is similar to that of
the AFE5851. Every channel on the AFE5851 is similar to a channel on the AFE5801 where the other input has
been shorted internally to VCM. See the AFE5851 16 Channel Variable Gain Amplifier (VGA) With Octal High
Speed ADC data sheet (SLOS574) for more information.
SERIAL INTERFACE
Register Initialization
After power up, the internal registers must be initialized to the default value (zero). Initialization can be done in
one of two ways:
• Through a hardware reset, by applying a positive pulse in the RESET pin
• Through a software reset, using the serial interface, by setting the SOFTWARE RESET bit to high. Setting
this bit initializes the internal registers to the respective default values (all zeros) and then self-resets the
SOFTWARE RESET bit to low. In this case, the RESET pin can stay low (inactive).
It is important to notice that after power up and before the device is reset to its default state, the power
consumption could be up to 2× the specified maximum, due to the unknown setting of the internal registers. In
order to prevent the initial increased power consumption, a potential solution is to connect the RESET pin to
either 1.8V or 3.3V supply with a 10kΩ resistor, so that the device is reset while powering up. For the reset to
take effect, the power must be up on DVDD18. Notice that there is no damage to the part by applying voltage
to the RESET pin while the device power is off.
Notice also that the reset only affects the digital registers, putting the part in its default state. It does not act
as a power down and as such, everything internal just keeps running. As the internal registers change their
values, the effects on the data propagate through the pipe. At the same time, there may be some glitches on
the output due to the transition of the registers values if any of the output controlling modes, for instance,
change. As the reset is level-sensitive and asynchronous with the input clock, it should be inactive in order to
write into the internal registers. Although it takes only 10ns after the reset rising edge to change the registers,
the output data may take, in the worst case, up to 20 clock cycles to be considered stable. Notice that the
output clocks (data and frame) are independent of the RESET and tight to the input clock, avoiding any loss
of synchronization.
18
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Reset Timing
Typical values at 25°C, min and max values across the full temperature range TMIN = –40°C to TMAX = 85°C,
AVDD3 = 3.3V, AVDD18 = DVDD18 = 1.8V, unless otherwise noted.
PARAMATER
CONDITIONS
MIN
t1
Power-on delay time
Delay from power-up of AVDD and LVDD to RESET pulse
active
t2
Reset pulse width
Pulse width of active RESET signal
t3
Register write delay time Delay from RESET disable to SEN active
tPO
Power-up delay time
TYP
MAX
UNIT
5
ms
10
ns
25
Delay from power-up of AVDD and LVDD to output stable
ns
6.5
ms
Power Supply
AVDD, LVDD
t1
RESET
t2
t3
SEN
T0108-04
Figure 35. Reset Timing
Programming of different modes can be done through the serial interface formed by pins SEN (serial interface
enable), SCLK (serial interface clock), SDATA (serial interface data) and RESET. SCLK and SDATA have a
100kΩ pulldown resistor to ground, and SEN has a 100kΩ pullup resistor to DVDD18. Serial shift of bits into the
device is enabled when SEN is low. Serial data SDATA is latched at every rising edge of SCLK when SEN is
active (low). The serial data is loaded into the register at every 24th SCLK rising edge when SEN is low. In case
the word length exceeds a multiple of 24 bits, the excess bits are ignored. Data can be loaded in multiples of
24bit words within a single active SEN pulse (there is an internal counter that counts groups of 24 clocks after
the falling edge of SEN). The interface can work with SCLK frequency from 20MHz down to very low speeds (a
few Hertz) and even with a non-50% duty-cycle SCLK.
The data is divided in two main portions: a register address (8 bits) and the data itself, to load on the addressed
register (16 bits). When writing to a register with unused bits, these should be set to 0. Also, when writing, the
SDOUT signal outputs zeros. The following timing diagram illustrates this process.
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Start Sequence
End Sequence
SEN
t6
t7
t1
t2
Data Latched On Rising Edge of SCLK
SCLK
t3
A7
SDATA
A5
A6
A4
A3
A2
A1
A0 D15 D14 D13 D12 D11 D10 D9
D8
D7
D6
D5
D4
D3
D2
D1
D0
t4
t5
Start Sequence
End Sequence
RESET
T0384-01
Figure 36. Serial Interface Register Write
Minimum values across full temperature range TMIN = –40°C to TMAX = 85°C, AVDD3 = 3.3V, AVDD18 =
DVDD18 = 1.8V.
PARAMETER
DESCRIPTION
MIN
TYP
MAX
UNIT
t1
SCLK period
50
ns
t2
SCLK high time
20
ns
t3
SCLK low time
20
ns
t4
Data setup time
5
ns
t5
Data hold time
5
ns
t6
SEN fall to SCLK rise
8
ns
t7
Time between last SCLK rising edge to SEN rising edge
8
ns
GENERAL-PURPOSE REGISTER MAP
The internal registers can be divided in two groups, a group of registers to control all the general functions and
settings of the device, and a bank of registers to control the TGC/gain curves operation. Those two sets of
registers overlap in all the address space, except for the address 0, which holds the control of the register bank.
One of the bits of this register, TGC_REG_WREN (see following table) is used to access one set of registers or
the other. Its default value is zero and gives access to the general-purpose registers (which are also by default
zero). The TGC control registers (described after the general-purpose registers) can be accessed by writing 1 to
TGC_REG_WREN.
The following table describes the function of the registers when TGC_REGISTER_WREN = 0 (default). The
address format is address[bit of the register].
ADDRESS
FUNCTION
DESCRIPTION
0[2]
TGC_REGISTER_WREN
0: Access to general-purpose registers. 1: Access to TGC registers
0[1]
REGISTER_READOUT_ENABLE
1: Enables readout of the registers
0[0]
SOFTWARE_RESET
1: Resets the device and self-resets the bit to zero
1[14]
OUTPUT_RATE_2X
0: 1× rate (one ADC per LVDS stream). 1: 2× rate (2 ADCs per LVDS stream)
1[13]
EXTERNAL_REFERENCE
0: Internal reference. 1: External reference
1[11]
LOW_FREQUENCY_NOISE_SUPRESSION 0: No supression. 1: Supresses noise at low frequencies and pushes it to fS/2
20
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ADDRESS
FUNCTION
DESCRIPTION
1[10]
STDBY
0: Power up. 1: Standby (fast power-up mode)
1[9:2]
PDN CHANNEL<7:0>
PDN for each individual channel (VCA+ADC). LVDS outputs logic 0.
1[1]
OUTPUT_DISABLE
0: Ouptut enabled. 1: Output disabled
1[0]
GLOBAL_PDN
0: Power up. 1: Global power down (slow power-up mode)
2[15:13]
PATTERN_MODES
Pattern modes for serial LVDS. 000: No pattern. 001: Sync. 010: Deskew.
011: Custom reg. 100: All 1s. 101: toggle. 110: All 0s. 111: Ramp
2[11]
AVERAGING_ENABLE
0: Default (no averaging). 1: Average two channels to increase SNR.
2[10:3]
PDN_LVDS
Power down the eight data-output LVDS pairs.
3[14:13]
SERIALIZED_DATA_RATE
Serialization factor. 00: 12×. 01: 10×. 10: 16×. 11: 14×
3[12]
DIGITAL_GAIN_ENABLE
0: Default (no gain). 1: Apply digital gain set by the following registers.
3[8]
REGISTER_OFFSET_SUBTRACTION_
ENABLE
0: Default (no subtraction). 1: Subtract offset value set in the corresponding
registers.
4[3]
DFS
Data format select. 0: 2s complement. 1: Offset binary
5[13:0]
CUSTOM_PATTERN
Custom pattern data for LVDS (PATTERN_MODES = 011)
7[10]
VCA_LOW_NOISE_MODE_(INCREASE_
POWER)
0: Low power. 1: Low noise, at the expense of increased power (5mW/channel)
7[8:7]
SELF_TEST
00 or 10: No self-test. 01: Self-test enabled. 100 mV dc applied to the input of
the channels. 11: Self-test enabled. 150 mV dc applied to the input of the
channels. NOTE: DC applied before the input buffer. Test does not work if input
is dc-shorted to a different potential. Also notice that the
INTERNAL_AC_COUPLING bit is automatically set to 1 when entering this
mode, and reset back to whatever value it had before, when leaving this mode.
7[3:2]
FILTER_BW
00: 14MHz. 01: 10MHz. 10: 7.5MHz. 11: Not used.
7[1]
INTERNAL_AC_COUPLING
VGA coupling. 0: AC-coupled. 1: DC-coupled
13[15:11]
DIG_GAIN1
0dB to 6dB in steps of 0.2dB
13[9:2]
OFFSET_CH1
Value to be subtracted from channel 1
15[15:11]
DIG_GAIN2
0dB to 6dB in steps of 0.2dB
15[9:2]
OFFSET_CH2
Value to be subtracted from channel 2
17[15:11]
DIG_GAIN3
0dB to 6dB in steps of 0.2dB
17[9:2]
OFFSET_CH3
Value to be subtracted from channel 3
19[15:11]
DIG GAIN4
0dB to 6dB in steps of 0.2dB
19[9:2]
OFFSET_CH4
Value to be subtracted from channel 4
21[4:1]
DIGITAL_HIGH_PASS_FILTER_CORNER_
FREQ_FOR_CHANNELS-1–4
Sets k for the high-pass filter as desribed in General-Purpose Register
Description (k from 2 to 10).
21[0]
DIGITAL_HIGH_PASS_FILTER_ENABLE_
FOR_CHANNELS_1–4
0: No high-pass filter. 1: High-pass filter enabled
25[15:11]
DIG_GAIN8
0dB to 6dB in steps of 0.2dB
25[9:2]
OFFSET_CH8
Value to be subtracted from channel 5
27[15:11]
DIG_GAIN7
0dB to 6dB in steps of 0.2dB
27[9:2]
OFFSET_CH7
Value to be subtracted from channel 6
29[15:11]
DIG_GAIN6
0dB to 6dB in steps of 0.2dB
29[9:2]
OFFSET_CH6
Value to be subtracted from channel 7
31[15:11]
DIG_GAIN5
0dB to 6dB in steps of 0.2dB
31[9:2]
OFFSET_CH5
Value to be subtracted from channel 8
33[4:1]
DIGITAL_HIGH_PASS_FILTER_CORNER_
FREQ_FOR_CHANNELS_5–8
Sets k for the high-pass filter as desribed in General-Purpose Register
Description (k from 2 to 10).
33[0]
DIGITAL_HIGH_PASS_FILTER_ENABLE_
FOR_CHANNELS_5–8
0: No high-pass filter. 1: High-pass filter enabled
70[14]
CLAMP_DISABLE
0: Enabled. 1: Disabled
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GENERAL-PURPOSE REGISTER DESCRIPTION
AVERAGING_ENABLE
Address: 2[11]
When set to 1, two samples, corresponding to two consecutive channels, are averaged (channel 1 with 2, 3
with 4, 5 with 6, and 7 with 8). If both channels receive the same input, the net effect is an improvement in
SNR. The averaging is performed as:
• Channel 1 + channel 2 comes out on channel 3.
• Channel 3 + channel 4 comes out on channel 4.
• Channel 5 + channel 6 comes out on channel 5.
• Channel 7 + channel 8 comes out on channel 6.
DFS
Address: 4[3]
DFS stands for data format select. The ADC output, by default, is in 2s-complement mode. Programming the
DFS bit to 1 inverts the MSB, and the output becomes straight-offset binary mode.
DIGITAL_GAIN_ENABLE
Address: 3[12]
Setting this bit to 1 applies to each channel i the corresponding gain given by DIG_GAINi<15:11>. The gain
is given as 0dB + 0.2dB × DIG_GAINi<15:11>. For instance, if DIG_GAIN5<15:11> = 3, channel 5 is
increased by 0.6dB gain. DIG_GAINi<15:11> = 31 produces the same effect as DIG_GAINi<15:11> = 30,
setting the gain of channel i to 6dB.
DIGITAL_HIGH_PASS_FILTER_ENABLE and DIGITAL_HIGH_PASS_FILTER_CORNER_FREQ
DIGITAL_HIGH_PASS_FILTER_ENABLE (channels 1–4): Address: 21[0]
DIGITAL_HIGH_PASS_FILTER_ENABLE (channels 5–8): Address: 33[0]
DIGITAL_HIGH_PASS_FILTER_CORNER_FREQ (channels 1–4): Address: 21[4:1]
DIGITAL_HIGH_PASS_FILTER_CORNER_FREQ (channels 5–8): Address: 33[4:1]
This group of four registers controls the characteristics of a digital high-pass transfer function applied to the
output data, following the formula: y(n) = 2k/(2k + 1) [x(n) – x(n – 1) + y(n – 1)]. The
DIGITAL_HIGH_PASS_FILTER_CORNER_FREQ registers (one for the first four channels and one for the
second group of four channels) describe the setting of k.
EXTERNAL_REFERENCE
Address: 1[13]
The internal reference mode (default) uses approximately 3mW more power on AVDD (which is already
included in all the specification tables). The AFE5801 can operate in external reference mode by
programming EXTERNAL_REFERENCE to 1. In this mode, the VREF_IN pin should be driven with 1.4V.
Due to the high input impedace of this pin, no special drive capabilities are required. For the same reason,
no decoupling on VREF_IN is needed, although depending on the noise on the 1.4V signal, some filtering
may be required. Nevertheless, when using the internal reference, there is no need to decouple VREF_IN.
The advantage of using the external reference mode is that multiple AFE5801 units can be made to operate
with the same external reference, thereby improving parameters such as gain matching across devices.
FILTER_BW
Address: 7[3:2]
This bit sets the 3dB attenuation frequency for the antialiasing filter (AAF).
GLOBAL_PDN
Address: 1[0]
The global PDN bit is ORed with the signal in the external PDN pin (59). Therefore, a 1 on this bit shuts the
device down completely.
22
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INTERNAL_AC_COUPLING
Address: 7[1]
This bit controls an internal high-pass filter (Figure 34), set between the input buffer and the VCA. This filter
removes the input offset to avoid its amplification by the TGC. An alternative method is to remove the offset
effect on the digital domain, either on the device following the ADC or at the ADC output, by using the
DIGITAL HIGH PASS FILTER registers described previously.
LOW_FREQUENCY_NOISE_SUPPRESSION
Address: 1[11]
The low-frequency noise-suppression mode is specifically useful in applications where good noise
performance is desired in the frequency band of 0MHz to 1MHz (around dc). Setting this mode shifts the
low-frequency noise of the ADC in the AFE5801 to approximately fS/2, thereby reducing the noise floor
around dc to a much lower value.
OUTPUT_DISABLE
Address: 1[1]
A 1 on this bit sets all the LVDS outputs into the high-impedance state.
OUTPUT_RATE_2X
Address: 1[14]
The output data always uses a DDR format, with valid/different bits on the positive as well as the negative
edges of the LVDS bit clock, DCLK. The output rate is set by default to 1× (OUTPUT_RATE_2X = 0), where
each ADC has one LVDS stream associated with it. If the sampling rate is low enough, two ADCs can share
one LVDS stream, in this way lowering the power consumption devoted to the interface. The unused outputs
will output zero. To avoid consumption from those outputs, no termination should be connected to them. The
distribution on the used output pairs is done in the following way:
• Channel 1 and channel 2 come out on channel 3. Channel 1 comes out first.
• Channel 3 and channel 4 come out on channel 4. Channel 3 comes out first.
• Channel 5 and channel 6 come out on channel 5. Channel 5 comes out first.
• Channel 7 and channel 8 come out on channel 6. Channel 7 comes out first.
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PATTERN_MODES and CUSTOM_PATTERN
PATTERN_MODES: Address: 2[15:13]
CUSTOM_PATTERN: Address: 5[13:0]
The AFE5801 can output a variety of test patterns on the LVDS outputs. These test patterns replace the
normal ADC data output and help on debugging and synchronization, with the device reading the output of
the ADC.
• PATTERN_MODE equal to 000 is the default and disables this test mode, i.e., the output data is the
same as the ADC data.
• PATTERN_MODE equal to 001 (SYNC mode) replaces the normal ADC word by a fixed 1111 1100 0000
word.
• PATTERN_MODE equal to 010 sets the DESKEW mode, where the 12-bit ADC output D<11:0> is
replaced with the 0101 0101 0101 word, which creates a continuous stream of 1s and 0s in the data line.
• PATTERN_MODE equal to 011 outputs a constant code set by the bits in CUSTOM_PATTERN<13:0>.
Depending on the value of SERIALIZED_DATA_RATE (see following) the output bits conform to the
following rules:
– In the default case, where SERIALIZED_DATA_RATE is 00, for 12-bit ADC data at the output, the
CUSTOM_PATTERN<13:2> bits are used, replacing the sampled data. These bits are still controlled
by the LSB-first and MSB-first modes in the same way as normal ADC data are.
– For SERIALIZED_DATA_RATE = 01, 10-bit output mode is selected, and the
CUSTOM_PATTERN<13:4> bits are used.
– For SERIALIZED_DATA_RATE = 10, 16-bit output mode is selected. In this case, the
CUSTOM_PATTERN<13:0> bits are used for the first 14 most-significant bits, and two 0s take the
place of the LSBs.
– For SERIALIZED_DATA_RATE = 11, 14-bit mode is selected, and the CUSTOM_PATTERN<13:0>
bits take the place of the output word.
• PATTERN_MODE equal to 100 makes the output always 1, whereas setting it to 110 makes the output
always 0.
• PATTERN_MODE equal to 101 makes the output of the device toggle between two consecutive codes.
On the nth sample clock, the data is 0000 0000 0000, and on the following one (nth + 1), it is
1111 1111 1111.
• PATTERN_MODE equal to 111 causes all the channels to output a repeating full-scale ramp pattern. The
ramp increments from zero code to full-scale code in steps of 1 LSB every clock cycle. After hitting the
full-scale code, it returns back to the zero code and ramps again.
PDN_CHANNEL<7:0>
Address: 1[9:2]
Each bit controls the power down of a channel (buffer, VCA, and ADC). For example, PDN_CHANNEL<0>
powers down channel 1 and the corresponding LVDS pair becomes high-impedance. DCLK and FCLK are
not powered down. They become active if terminated with 100Ω.
PDN_LVDS
Address: 2[10:3]
PDN_LVDS<7:0> selects which LVDS pairs become inactive (zero output current, i.e., high-impedance
state). The frame and clock LVDS streams are powered down only when OUTPUT_DISABLE and/or
GLOBAL_PDN is set.
REGISTER_OFFSET_SUBTRACTION_ENABLE
Address: 3[8]
Setting this bit to 1 enables the substraction of the value on the corresponding OFFSET_CHi<9:2> (offset for
channel i) from the ADC output. The number is specified in 2s-complement format. For example,
24
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OFFSET_CHi<9:2> = 0100 0000 means substract –128. For OFFSET_CHi<9:2> = 0111 1111 the effect is to
substract 127. In effect, both addition and subtraction can be performed. Note that the the offset is applied
before the digital gain (see DIGITAL_GAIN_ENABLE). The whole data path is 2s-complement throughout
internally. Only when DFS = 1 (straight binary output format) is the 2s-complement word translated into offset
binary at the end.
REGISTER_READOUT_ENABLE
Address: 0[1]
The device includes an option where the contents of the internal registers can be read back. This may be
useful as a diagnostic to verify the serial interface communication between the external controller and the
AFE. First, the <REGISTER READOUT ENABLE> bit must be set to 1. Then user should initiate a serial
interface cycle specifying the address of the register (A7–A0) whose content is to be read. The data bits are
don't care. The device outputs the contents (D15–D0) of the selected register on the SDOUT pin. The
external controller can latch the contents at the rising edge of SCLK. To enable serial register writes, set the
<REGISTER READOUT ENABLE> bit back to 0. The following timing diagram shows this operation (the time
specifications follow the same information provided on the table for a serial interface register write):
Start Sequence
End Sequence
SEN
t6
t7
t1
t2
SCLK
t3
A7
SDATA
A6
A5
A4
A3
A2
A1
A0
x
x
x
x
x
x
x
x
x
x
x
x
x
x
x
x
D2
D1
D0
t4
SDOUT to be Latched Externally On the Rising Edge
t5
D15 D14 D13 D12 D11 D10 D9
SDOUT
D8
D7
D6
D5
D4
D3
T0385-01
Figure 37. Serial Interface Register Read
Register readback is incorrect for address 0x97. See the INTERP_ENABLE section for details.
SERIALIZED_DATA_RATE
Address: 3[14:13]
These two bits control the length of the data word, i.e., the number of DCLKs per FCLK period. It is possible,
for instance, to output a 16bit data stream even with a 12bit ADC. In this case, the 4 LSBs are padded with
0s. The pass from higher resolution to lower serialization is not supported, however. I.e, it is not possible to
select a 10bit stream with a 12bit ADC.
TGC_REGISTER_WREN
Address: 0[2]
Set this bit to 1 to access the TGC table and 0 (default after reset) to access the general-purpose register
table. The same address may point, this way, to one bank of registers (general purpose) or to the other
(TGC control). Nevertheless, observe that register 0 of the general-purpose registers is always accessible,
regardless of the value of TGC_REGISTER_WREN. The TGC table starts at address 1.
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VCA_LOW_NOISE_MODE
Address: 7[10]
Setting this bit to 1 reduces the equivalent input noise of the VCA to 5nv/√Hz (for a 51dB gain) at the
expense of an increased power consumption (5mW increase per channel).
TGC CONTROL REGISTER MAP
The TGC operation is described in the VGA/TGC Operation section that follows. This section describes the TGC
control registers which can be accessed by writing 1 to TGC_REG_WREN bit. The following table describes the
register map for all the registers involved in the TGC operation.
ADDRESS
D[15:7]
D[8]
D[7]
D[6]
0x01...0x94
D[5]
D[4]
D[3]
0x95
START_INDEX
0x96
STOP_INDEX
INTERP_
ENABLE
0x97
0x98
D[2]
D[1]
D[0]
REG_VALUES
0
START_GAIN
HOLD_ GAIN _TIME
NOT USED
0x99
0
0
0x9A
0
0
0x9B
SOFT_
SYNC
UNIFORM_
GAIN_
MODE
STATIC_
PGA
FINE_GAIN
COARSE_GAIN
UNIFORM_GAIN_SLOPE
REG_VALUE
Address: 0x01[8:0] to 0x94[8:0]
Each of these 9 bit registers (148 of them) stores the time to stay at a given gain setting, during the gain
ramp. The most significant bit of each register (REG_VALUE<8>) denotes either increment or decrement
gain from current gain value. The other 8 bits (REG_VALUE<7:0>) denote the time (a multiple of 8 × Tclk;
Tclk being the channel sampling clock, i.e., double the period of the device input clock) for the change of the
gain from the CURRENT_GAIN to CURRENT_GAIN ±1dB (depending on the REG_VALUE<8>). The fastest
ramp (shortest time) for this 1dB gain change is set by REG_VALUE<7:0> equal to 0x00 and it is 8 × Tclk.
The slowest ramp (longest time) for this 1dB gain change is set by REG_VALUE<7:0> equal to 0xFF, and it
is 255 × 8 × Tclk (see VGA operation – described later).
START_INDEX
Address: 0x95[7:0]
This 8 bit register specifies/points to the first REG_VALUE register of the TGC curve (i.e., where the curve
starts) and can have values ranging from 1 to 148 (in decimal).
STOP_INDEX
Address: 0x96[7:0]
This 8 bit register specifies/points to the last REG_VALUE register of the TGC curve (i.e., where the curve
finishes) and can have values ranging from 1 to 148 (in decimal).
26
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START_GAIN
Address: 0x97[5:0]
This 6 bit register specifies the start gain value from –5dB to 31dB.
START_GAIN = [–5 + REG_VALUE ] dB
REG_VALUE
GAIN
0x0
0x1
0x24
–5 dB
–4 dB
31 dB
STOP_GAIN (Not a programmable register; it is an internally computed value.)
Case 1:
INTERP_ENABLE = 1,
STOP_GAIN = START_GAIN + (STOP_INDEX – START_INDEX) – ( 2 × Number of
decrements) + 0.875dB.
Case 2:
INTERP_ENABLE = 0,
STOP_GAIN = START_GAIN + (STOP_INDEX – START_INDEX) – ( 2 × Number of
decrements).
HOLD_GAIN_TIME
Address: 0x98[7:0]
This 8 bit register specifies the time for holding of the STOP_GAIN, after reaching either the STOP_GAIN
value as computed in the STOP_GAIN section or the maximum/minimum gain. After this time, the TGC starts
stepping down to the START_GAIN value in 1dB steps every Tclk. The STOP_GAIN value is held for the
following number of clocks:
HOLD_GAIN_TIME = [33 × REG_VALUE] Tclks
where Tclk is the channel sampling clock.
REG_VALUE
0x0
0x1
0xFF
HOLD_GAIN_TIME
0 Tclks
33 Tclks
8415 Tclks
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INTERP_ENABLE
Address: 0x97[7]
This 8 bit register sets the ramp rate. When INTERP_ENABLE = 1, the ramp rate is 0.125dB for every
number of clocks stored in REG_VALUE:
REG_VALUE
0x0
0x1
0x2
0xFF
SLOPE
0.125dB
0.125dB
0.125dB
0.125dB
per
per
per
per
Tclk
Tclk
2 × Tclk
255 × Tclk
When INTERP_ENABLE = 0 the ramp rate is 1dB for every 8 times the number of clocks stored in REG_VALUE:
REG_VALUE
0x0
0x1
0x2
0xFF
SLOPE
1dB per
1dB per
1dB per
1dB per
8 × Tclk
8 × Tclk
16 × Tclk
255× 8 × Tclk
NOTE
Reading back the address 0x97 (INTERP_ENABLE) to verify its value shows the opposite
value of what it actually is. For instance, after setting INTERP_ENABLE to 1 in addres
0x97 to enable interpolation, the bit shows as 0 when reading it back from the same
address (0x97). After setting INTERP_ENABLE to 0 in addres 0x97 to disable
interpolation, the bit shows as 1 when reading it back.
SOFT_SYNC
Address 0x99[5]
Setting SOFT_SYNC bit to 1 enables the TGC engine to run periodically following a given TGC curve,
without the need for a high pulse signal in the SYNC pin (see more details in the Soft Synchronization
section).
UNIFORM_GAIN_MODE
Address 0x99[4]
Setting this bit to 0 (default) directs the TGC engine to follow an arbitrary gain-versus-time curve. If this bit is
set to 1, the gain is ramped up with a slope set by the UNIFORM_GAIN_SLOPE register. (See more details
in the Uniform Gain Increment Mode section.)
UNIFORM_GAIN_SLOPE
Address 0x9B[7:0]
See the Uniform Gain Increment Mode section.
STATIC_PGA
Address 0x99[3]
Setting this bit to 1 disables the TGC engine. COARSE_GAIN and FINE_GAIN control the gain value, which
is independent of time.
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COARSE_GAIN
Address 0x9A[5:0]
This 6 bit register specifies the coarse gain from –5 to 31dB, in 1dB steps. Observe that only values from
0x00 to 0x24, both included, are valid. Setting a value bigger than 0x24 on the COARSE_GAIN register is
the same as setting 0x24. COARSE_GAIN = [–5 + REG_VALUE ] dB
REG_VALUE
0x0
0x1
0x24
GAIN
–5dB
–4dB
31dB
FINE_GAIN
Address 0x99[2:0]
This 3 bit register specifies the fine gain in steps of 0.125dB resolution, from 0dB to 0.875dB. FINE_GAIN =
[0.125 × REG_VALUE ] dB
REG_VALUE
0x0
0x1
0x7
GAIN
0dB
0.125dB
0.875dB
VGA/TGC OPERATION
The gain variation of the variable gain amplifier (VGA) versus time is called the TGC function and on the
AFE5801 is controlled digitally. The gain is implemented by a switched network where the switches controlling
the gain are synchronized with the ADC sampling instant to minimize glitches on the output data. The gain
setting depends on the mode of operation selected by the user. There are 3 possible modes of operation:
non-uniform gain, uniform gain, and static mode. The following sections describe each in detail.
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Sync Period
GAIN Profile
SYNC
Signal Input
to AFE5801
External
System
Signal
Wait time
at start gain,
tw
Ramp up time from
start gain to stop gain,
tru
Hold time at
Wait time
stop gain,
at start gain
th
Ramp down from
stop gain to start
gain, trd
Sync Period = tru + th + trd + tw
Figure 38. SYNC Period
Non-Uniform Gain Increment Mode
In the non-uniform gain increment mode, the user sets an arbitrary shape for the gain versus time curve. For a
given time/sampling instant, the digital gain setting is obtained from an internal memory of 148 positions/registers
(named REG_VALUEs), each 9 bits wide, loaded by the user through the serial port (see Serial Interface
section). Addresses 1 to 148 can be used to access these registers, while TGC_REGISTER_WREN = 1.
As explained above, the most significant bit of each register (REG_VALUE<8>) denotes either increment or
decrement gain from current gain value. The other 8 bits (REG_VALUE<7:0>) denote the time (a multiple of 8 ×
Tclk, being Tclk the sampling clock) for the change of the gain from the CURRENT_GAIN to CURRENT_GAIN
±1dB (depending on the REG_VALUE<8>). The fastest ramp (shortest time) for this 1dB gain change is set by
REG_VALUE<7:0> equal to 0x00 and it is 8 × Tclk. The slowest ramp (longest time) for this 1dB gain change is
set by REG_VALUE<7:0> equal to 0xFF and it is 255 × 8 × Tclk.
INTERP_ENABLE sets the way the gain is increased/decreased. By default the gain ramp is implemented in
steps of 1dB (INTERP_ENABLE equal to 0). If INTERP_ENABLE is equal to 1, the actual 1dB gain step is
implemented in 8 steps of 0.125dB.
The 148 REG_VALUE registers can be used to store either a single curve or multiple TGC curves. The
START_INDEX register points to the REG_VALUE register where the TGC curve starts and the STOP_INDEX
register points to the REG_VALUE register where the TGC curve stops. Using the START_INDEX and
STOP_INDEX registers the desired TGC curves can be chosen.
30
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As shown in Figure 38, a pulse high signal on the SYNC pin will set the starting gain value of the TGC curve to
the START_GAIN register value, and it will initiate the progression through the different REG_VALUEs, starting
at START_INDEX. Observe that there is no option to delay the start of gain stepping after the SYNC pulse is
received. Then, the progression continues until either the STOP_INDEX is reached or maximum/minimum gain is
exceeded. After that, the last valid value of gain is held for an extra given number of clocks set by the register
HOLD_GAIN_TIME.
After the elapsing of clocks mentioned by the HOLD_GAIN_TIME register, the TGC starts to step down (or up) to
the START_GAIN in steps of 1dB every Tclk (channel sampling clock) in preparation for the next TGC profile.
The TGC will start updating/following the REG_VALUEs again after a new high pulse on the SYNC pin is given.
The SYNC signal is latched by the rising edge of the channel sampling clock. In other words, the gain increments
at the rising edge of the channel sampling clock. Setup time with rising edge is 7ns, and hold time 4ns.
Soft Synchronization
The TGC can run periodically following a given TGC curve but without the need for a high pulse signal in the
SYNC pin. This is done by setting SOFT_SYNC bit to 1. Once this bit is set, the sequence of events is the same
as with the hardwired SYNC pulse. The TGC curve updates from START_INDEX to STOP_INDEX. After
reaching STOP_INDEX or the maximum/minimum gain, the STOP_GAIN value is held for HOLD_VALUE_TIME
and then the gain ramps up or down to START_GAIN. After this the TGC update starts again automatically and
repeats all these steps periodically till the SOFT_SYNC bit becomes zero.
The SYNC process through register write occurs at the serial clock edge where the register is written. If serial
clock and sample clock (channel sampling clock) are synchronous then the described relation in the hardwired
SYNC section will hold and the SYNC bit is latched by the rising edge of the channel sampling clock, respecting
a setup time with rising edge of 7ns and hold time of 4ns. If sample clock and serial clock are not synchronous
then this relationship does not apply and a clock uncertainty of ±1 sample will apply in respect to the nearest
sample clock rising edge.
Example 1: In the following example of non-uniform gain mode, all the 148 registers are loaded. Nevertheless,
the start address for the TGC is set in START_INDEX to 2 and the stop address (STOP_INDEX) to 7. The
START_GAIN is set to 6 and HOLD_GAIN_TIME is 4.
With a high pulse on the SYNC pin the gain starts from 1dB (START_GAIN = 0x06). 1dB to 2dB ramp is done in
120Tclks, using eight 0.125dB steps (as INTERP_ENABLE is set to 1), each 15Tclks long. The ramp from 2dB to
3dB is done in 64Tclks, also in 0.125dB steps. The ramp from 3dB to 4dB is done in 40 Tclks. Decrement from
4dB to 3dB in 64Tclks. Gain increment from 3dB to 4dB in 56 Tclks and from 4dB to 4.875dB in 80 Tclks.
Observe that in the case where INTERP_ENABLE = 1, STOP_GAIN = START_GAIN + (STOP_INDEX –
START_INDEX) – ( 2 × Number of decrements) + 0.875dB. In the case where INTERP_ENABLE = 0,
STOP_GAIN = START_GAIN + (STOP_INDEX – START_INDEX) – ( 2 × Number of Decrements). This is due to
the fact that the interpolation engine keeps the gain increasing or decreasing when INTERP_ENABLE = 1, while
the gain is frozen when INTERP_ENABLE = 0.
TGC REG INDEX
REG_VALUE[8:0]
Number of Tclks
Direction of Gain Change
1
0x004
4 × 8 = 32
Increment
2
0x00F
15 × 8 = 120
Increment
3
0x008
8 × 8 = 64
Increment
4
0x005
5 × 8 40
Increment
5
0x108
8 × 8 = 64
Decrement
6
0x007
7 × 8 = 56
Increment
Increment
7
0x00A
10 × 8 = 80
...
...
...
...
147
0x00F
15 × 8 = 120
Increment
148
0x00F
15 × 8 = 120
Increment
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NAME
VALUE
START_INDEX
0x02
STOP_INDEX
0x07
START_GAIN
0x06
HOLD_GAIN_TIME
0x04
INTERP_ENABLE
1
UNIFORM_GAIN_MODE
0
Uniform Gain Increment Mode
By setting UNIFORM_GAIN_MODE to 1, the TGC engine can also be configured for a uniform increment gain
ramp mode where the gain is ramped up from the START_GAIN value to the STOP_GAIN with a slope set by
the UNIFORM_GAIN_SLOPE register. Note: STOP_GAIN is not a programmable register, but just an internally
computed value from START_GAIN, UNIFORM_GAIN_SLOPE, START_INDEX and STOP_INDEX.
If INTERP_ENABLE = 1, UNIFORM_GAIN_SLOPE sets the number of Tclk (channel sampling clock) at a given
gain before incrementing or decrementing 0.125dB. If INTERP_ENABLE = 0, this register sets the number of 8 ×
Tclk (eight sampling periods) at a given gain before incrementing or decrementing 1dB. Observe that in both
cases the time it takes to step by 1dB is the same. In INTERP_ENABLE = 0 the gain is stationary at the same
setting for the given time, whereas in the other case the gain increments in fine gain steps of 0.125dB to cover
that 1dB step
When
INTERP_ENABLE
is
zero,
the
STOP_GAIN
is
computed
as
START_GAIN
+
(STOP_INDEX-START_INDEX). Nevertheless, when INTERP_ENABLE = 1, the STOP_GAIN is equal to
START_GAIN + (STOP_INDEX - START_INDEX) + 0.875dB. This is basically due to the fact that the
interpolation engine keeps the gain increasing on the second case, while, as explained above, is frozen on the
first case. Observe that START_INDEX and STOP_INDEX are not used in this case as pointers to the
REG_VALUEs table. Instead, only the difference between the two is important to compute STOP_GAIN. As
such, START_INDEX can be set to zero and STOP_INDEX will store STOP_GAIN – START_GAIN. Observe
that only positive slope ramps are possible.
Example 1: setting START_GAIN = 0x2 (–3dB), START_INDEX = 0x00, STOP_INDEX = 0x06,
INTERP_ENABLE = 0 and UNIFORM_GAIN_SLOPE = 0x8, will set the gain at –3dB for 8 × 8 × Tclk, then to
–2dB for another 64 Tclk, and so on, through –1, 0, 1, 2 and 3. After spending 64 × Tclk in 3dB, the gain will stay
at that gain setting for HOLD_GAIN_TIME and start stepping down back to START_GAIN, with 1dB per Tclk.
Example 2: for the same settings, START_GAIN = 0x2 (–3dB), START_INDEX = 0x00, STOP_INDEX = 0x06,
and UNIFORM_GAIN_SLOPE = 0x8, if we set INTERP_ENABLE = 1, the gain will start at –3dB for 8Tclk, then
–2.875dB for another 8Tclk, then –2.750dB and so on, till 3dB. At this point, while in example 1, with
INTERP_ENABLE = 0 the gain would be frozen for another 64 Tclk, in this example, the gain will continue to
increase with 0.125dB steps every 8Tclk till 3.875dB is reached. There will stay for another 8Tclk before starting
to wait for HOLD_GAIN_TIME and start stepping down.
Example 3: for START_GAIN = 0x2 (–3dB) , START_INDEX = 0x00, STOP_INDEX = 0x00, INTERP_ENABLE =
1 and UNIFORM_GAIN_SLOPE = 0x1, the gain will step through –3dB, –2.875, –2.75, –2.625, –2.5, –2.375,
–2.25 and –2.125, staying at each of these 8 values 1 clock cycle (8 total). Then it will wait for
HOLD_GAIN_TIME in –2.125dB and then it will start stepping down back to –3dB.
Example 4: same settings as example 3, but with INTERP_ENABLE = 0, would simply set the VGA gain to –3dB
for 8 clock cycles and then the logic would wait for HOLD_GAIN_TIME.
Static PGA Mode
The 3rd mode of operation is actually a mode where the TGC engine is disabled by writing 1 into the
STATIC_PGA bit. This enables the use of a fixed gain mode where the gain is obtained by the sum of a coarse
and a fine gain. Coarse gain can be set from –5 to 31dB, in 1dB steps, by the register COARSE_GAIN (6 bit
word from 0x00 to 0x24). Setting a value bigger than 0x24 on the COARSE_GAIN register is the same as setting
0x24. The fine gain can be set in steps of 0.125dB resolution, from 0dB to 0.875dB by the FINE_GAIN register (3
bit word with range from 0x00 to 0x07). Observe that the maximum gain, when both registers are set to their
maximum gains, is actually 31.875dB.
32
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ANTI-ALIAS FILTER (AAF)
The AFE5801 integrates a selectable third-order low-pass filter for each of the eight channels. The cutoff
frequency can be set for all the channels simultaneously through the serial interface (see FILTER_BW register, in
the General Purpose Register table) between 3 possible settings: 7.5, 10 and 14MHz. Figure 18 shows the
frequency response for each of these settings. The filter characteristics are set by passive components which are
subject to variations over process and temperature. A typical variation of ±5% on the frequency characteristics is
expected.
CLAMPING CIRCUIT AND OVERLOAD RECOVERY
The AFE5801 is designed in particular for ultrasound applications where the front-end device is required to
recover very quickly from an overload condition. Such overload can either be the result of a transmit pulse
feed-through or a strong echo, which can cause overload of the VGA and ADC.
Enabled by default, the AFE5801 includes a clamping circuit to further optimize the overload recovery behavior of
the complete channel (see Figure 34). The circuit can be disabled by writing a 1 in the bit 14 of the address 70
(decimal) of the General Purpose Register Map. The clamp is set to limit the signal at 3dB above the full scale of
the ADC (2Vpp).
CLOCK INPUTS
The eight channels on the device operate from a single clock input. To ensure that the aperture delay and jitter
are the same for all channels, the AFE5801 uses a clock tree network to generate individual sampling clocks to
each channel. The clock lines for all channels are matched from the source point to the sampling circuit of each
of the eight internal ADCs. The variation on this delay is described in the Aperture Delay parameter of the Output
Interface Timing. Its variation over time is described in the Aperture Jitter number of the same table.
The AFE5801 clock input can be driven differentially (sine wave, LVPECL or LVDS) or single-ended (LVCMOS).
The clock input of the device has an internal buffer/clock amplifier (see Figure 39) which is enabled or disabled
automatically depending on the type of clock provided (autodetect feature). When enabled, the device will
consume 6mW more power from the AVDD18 supply rail, but it will also accept differential or single ended inputs
of smaller swing.
AVDD18
VCM
VCM
5 kW
5 kW
CLKP
CLKM
Figure 39. Internal Clock Buffer for Differential Clock Mode
If the preferred clocking scheme for the device is single-ended, CLKINM pin should be connected to ground, i.e.,
shorted directly to AVSS (see Figure 41). In this case, the autodetect feature will shut down the internal clock
buffer and the device will go into single-ended clock input automatically. The user should connect the
single-ended clock source directly (no decoupling) to CLKINP pin, which would be the only device clock input. In
that case, it is recommended the use of low jitter square signals (LVCMOS levels, 1.8V amplitude) to drive the
ADC (see SLYT075 for further details on the theory).
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For single ended sinusoidal clocks or for differential clocks (differential sinewave, LVPECL, LVDS…), the clock
amplifier should be enabled. For that, the connection scheme of Figure 40 should be used. The common-mode
voltage of the clock source should match one of the clock inputs of the AFE5801 (VCM) which is set internally
using 5kΩ resistors, as shown in Figure 39. The easiest way to ensure this is to AC couple the inputs as shown
in Figure 40. The same scheme applies to the case where the clock is single ended but its amplitude is small or
its edges are not sharp (for instance, with a sinusoidal single-ended clock). In this case, the input clock signal
can be connected with a capacitor to CLKINP (as in Figure 40) and the CLKINM should be connected to ground
also through a capacitor, i.e., AC coupled to AVSS.
0.1 mF
CLKP
Differential Sine-Wave
or PECL or LVDS Clock Input
0.1 mF
CLKM
AFE5801
AFE5851
Figure 40. Differential Clock Driving Circuit
If a transformer is used with the secondary floating (for instance, to pass from single-ended to differential) , it can
then obviously be connected directly to the clock inputs, without the need of the 100nF series capacitors.
CMOS Clock Input
CLKP
CLKM
AFE5801
AFE5851
Figure 41. Single-Ended Clock Driving Circuit
Finally, on the differential clock configurations, Figure 42 shows the use of the CDCM7005 to generate the
AFE5801 clock signals.
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VCC
Reference Clock
REF_IN
Y0
CLKP
Y0B
CLKM
VCC
CDCM7005
AFE5801
OUTP
VCXO_INP
OUTM
VCXO_INM
CTRL
CP_OUT
VCXO
Figure 42. PECL Clock Drive Using CDCM7005
DIGITAL OUTPUTS
The conversion results from all eight ADCs are serialized and output using one LVDS data pair per ADC, at 12
times the device input clock rate. Besides that, two more LVDS pairs are used to facilitate the interface to the
circuit reading the ADC output. For one side, a reference frame LVDS signal running at the input clock rate
indicates the beginning and end of the sample word. On top of that, the device outputs a reference clock running
at 6 times the input clock rate, with rise and fall times aligned with the individual bits. See the Output Interface
Timing section for a description of the timing diagram as well as details on the timing margins.
Figure 43 represents the device LVDS output circuit. Observe that for an LVDS output high (OUTP = 1.375V,
OUTM = 1.025V) the high switches would be closed and the low switches would be open. For LVDS output low
(OUTP = 1.025V, OUTM = 1.375V) the low switches would be closed and the high left open. As the high and low
switches have a nominal RON of 50Ω ±10%, notice that the output impedance will be nominally 100Ω in any of
those two configurations (high or low switches closed).
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AFE5801
+0.35 V
Low
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High
SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
OUTP
-0.35 V
High
1.2 V
Low
External
100 W Load
ROUT
OUTM
Switch impedance is
Nominally 50 W ( ±10%)
Figure 43. LVDS Output Circuit
EXTERNAL/INTERNAL REFERENCE
See EXTERNAL_REFERENCE register description in the General Purpose Register Description Section.
POWER SUPPLIES
The use of low noise power supplies with adequate decoupling is recommended, being the linear supplies the
first choice vs switched ones, which tend to generate more noise components that can be coupled to the
AFE5801.
There is no need of any type of power-up sequencing, although a positive pulse must be applied to the Reset pin
once the power supplies are considered stable (see Serial Interface Section)
There are several types of powerdown modes. On the standby mode all circuits but the reference generator are
powered-down. This enables for a fast recovery from power down to full operation. On the full power down mode,
all the blocks are powered down (except some digital circuits). The power savings are bigger but the power-up
will also be slower (see specification tables for more details). The device includes also the possibility of powering
down pairs of channels (corresponding to the same ADC) through the use of PDN_Channel<7:0> and powering
down the LVDS outputs by using PDN_LVDS.
Finally, notice that the metallic heat sink under the package is also connected to analog ground.
LAYOUT INFORMATION
The evaluation board represents a good guideline of how to layout the board to obtain the maximum
performance out of the AFE5801. General design rules as the use of multilayer boards, single ground plane for
both, analog and digital ADC ground connections, and local decoupling ceramic chip capacitors should be
applied. The input traces should be isolated from any external source of interference or noise, including the
digital outputs as well as the clock traces. Clock should also be isolated from other signals although the low
frequencies of the input signal relaxes the jitter requirements.
In order to maintain proper LVDS timing, all LVDS traces should follow a controlled impedance design (for
example, 100Ω differential). In addition, all LVDS trace lengths should be equal and symmetrical. It is
recommended to keep trace length variations less than 150mil (0.150in or 3.81mm).
It is necessary to solder the exposed pad at the bottom of the package to a ground plane for best thermal
performance. For detailed information, see application notes QFN Layout Guidelines (SLOA122A) and QFN/SON
PCB Attachment (SLUA271A).
36
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
DEFINITION OF SPECIFICATIONS
Analog Bandwidth – The analog input frequency at which the power of the fundamental is reduced by 3 dB with
respect to the low frequency value.
Aperture Delay –The delay in time between the rising or the falling edge of the input sampling clock (depending
on the channel) and the actual time at which the sampling occurs. This delay will be different across channels.
The maximum variation is specified as aperture delay variation (channel-channel).
Aperture Uncertainty (Jitter) – The sample-to-sample variation in aperture delay.
Clock Pulse Width/Duty Cycle – The duty cycle of a clock signal is the ratio of the time the clock signal remains
at a logic high (clock pulse width) to the period of the clock signal. Duty cycle is typically expressed as a
percentage. A perfect differential sine-wave clock results in a 50% duty cycle.
Maximum Conversion Rate – The maximum sampling rate at which certified operation is given. All parametric
testing is performed at this sampling rate unless otherwise noted.
Minimum Conversion Rate – The minimum sampling rate at which the ADC functions.
Differential Nonlinearity (DNL) – An ideal ADC exhibits code transitions at analog input values spaced exactly
1 LSB apart. The DNL is the deviation of any single step from this ideal value, measured in units of LSBs.
Integral Nonlinearity (INL) – The INL is the deviation of the ADC's transfer function from a best fit line
determined by a least squares curve fit of that transfer function, measured in units of LSBs.
Gain Error – The difference between the actual gain of a channel & its ideal (theoretical) gain, i.e., the error in
the absolute gain of the channel.
Gain Matching – The gain difference between two channels with same theoretical gain setting. For perfect
matching, the difference should be zero. On the context of this device, the gain matching is obtained in two
different ways:
1. The values on the specification table represent the expected gain matching between any two channels on
the system. The gain is measured on every channel of every device, for a given gain setting, at any
temperature. The difference between the maximum recorded gain and the minimum recorded gain
represents the gain matching at that given gain setting. The same is done for every gain setting and the
maximum difference for any gain setting is presented on the table.
2. The gain matching histogram represents the channel to channel matching inside the same device, i.e., the
maximum expected gain difference between any two channels of the same device, or in other words, the
peak-to-peak variation of absolute gains across all channels in the device. At a given gain setting for all the
channels of a given device (at one temperature assumed common to the whole device), the difference
between the channel with maximum gain and the channel with minimum gain represents one count. The
same thing is done for all the devices and for 3 temperatures (–40C, 25C and 85C). Every measurement of a
device at one given temperature represents one count.
Offset Error – The offset error is the difference, given in mV, between the ADC's actual average idle-channel
output code and the ideal average idle-channel output code.
Temperature Drift – The temperature drift coefficient (with respect to gain error and offset error) specifies the
change per degree Celsius of the parameter from TMIN to TMAX. It is calculated by dividing the maximum deviation
of the parameter across the TMIN to TMAX range by the difference TMAX – TMIN.
Signal-to-Noise Ratio – SNR is the ratio of the power of the fundamental (PS) to the noise floor power (PN),
excluding the power at DC and the first nine harmonics.
P
SNR + 10Log10 S
PN
(1)
SNR is either given in units of dBc (dB to carrier) when the absolute power of the fundamental is used as the
reference, or dBFS (dB to full scale) when the power of the fundamental is extrapolated to the full-scale
converter range.
Signal-to-Noise and Distortion (SINAD) – SINAD is the ratio of the power of the fundamental (PS) to the power
of all the other spectral components including noise (PN) and distortion (PD), but excluding dc.
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SINAD = 10 log 10
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PS
PN + PD
(2)
SINAD is either given in units of dBc (dB to carrier) when the absolute power of the fundamental is used as the
reference, or dBFS (dB to full scale) when the power of the fundamental is extrapolated to the full-scale
converter range.
Effective Number of Bits (ENOB) – The ENOB is a measure the performance of a converter as compared to
the theoretical limit based on quantization noise.
ENOB + SINAD * 1.76
6.02
(3)
Spurious-Free Dynamic Range (SFDR) – SFDR is the ratio of the power of the fundamental (PS) to the highest
FFT bin, harmonic or not, excluding DC. SFDR is typically given in units of dBc (dB to carrier).
Second Harmonic Distortion (HD2) – HD2 is the ratio of the power of the fundamental (PS) to the second
harmonic, typically given in units of dBc (dB to carrier).
Third Harmonic Distortion (HD3) –HD3 is the ratio of the power of the fundamental (PS) to the third harmonic,
typically given in units of dBc (dB to carrier).
Total Harmonic Distortion (THD) – THD is the ratio of the power of the fundamental (PS) to the power of the
first nine harmonics (PD).
THD = 10 log 10
PS
PD
(4)
THD is typically given in units of dBc (dB to carrier).
AC Power Supply Rejection Ratio (AC PSRR) – A measure of the device immunity to variations in its supply
voltage. In this datasheet, if ΔVSUP represents the change in supply voltage and ΔVOUT is the resultant change
of the ADC output code (referred to the input), then:
æ DVout ö
PSRR = 20 log ç
÷
è DVsup ø
38
(5)
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SLOS591D – SEPTEMBER 2008 – REVISED MAY 2010
REVISION HISTORY
Changes from Revision C (January 2010) to Revision D
Page
•
Deleted INVERT_CHANNEL and MSB_FIRST rows from register map table ................................................................... 21
•
Deleted INVERT_CHANNEL register description ............................................................................................................... 23
•
Deleted MSB_FIRSTL register description ......................................................................................................................... 23
Changes from Revision B (April 2009) to Revision C
Page
•
Added pullup/pulldown resistors to descriptions of PDN, RESET, SCLK, SDATA, SEN, and SYNC terminals .................. 3
•
Added a note to the TERMINAL FUNCTIONS table, referenced from pin 28 within the table ............................................ 3
•
Listed names of digital control pins and changed maximum voltage rating for them ........................................................... 4
•
Added a minimum value for LVDS ac-coupled clock input in RECOMMENDED OPERATING CONDITIONS ................... 5
•
Added rows for VIH and VIL to CLOCK INPUT section of RECOMMENDED OPERATING CONDITIONS table ................ 5
•
Added note to ELECTRICAL CHARACTERISTICS table regarding the effects of enabling clamping ................................ 5
•
Added note to ELECTRICAL CHARACTERISTICS table regarding the effects of enabling clamping ................................ 6
•
Added note to ELECTRICAL CHARACTERISTICS table for IDVDD18 row in POWER section .............................................. 6
•
Listed names of digital-input and digital-output pins in Digital Characteristics table ............................................................ 7
•
Added section for SDOUT to Digital Characteristics table ................................................................................................... 7
•
Added a reference to Recommended Operating Conditions table to the note following the Digital Characteristics
table ...................................................................................................................................................................................... 7
•
Added note for FCLK timing to Output Interface Timing table .............................................................................................. 8
•
Modified next-to-last paragraph in Theory of Operation section ......................................................................................... 17
•
Added a new paragraph to the end of the Input Configuration section .............................................................................. 18
•
Added two new paragraphs to the end of the Register Initialization section ...................................................................... 18
•
Added a sentence to the last paragraph of the Reset Timing section ................................................................................ 19
•
Changed "Serial Interface Register Write" from a section heading to a figure caption ...................................................... 20
•
Added a parenthetical expression to a sentence in the General-Purpose Register Map section ...................................... 20
•
Deleted text from ADDRESS 3[8] DESCRIPTION ............................................................................................................. 21
•
Changed DESCRIPTION for ADDRESS 7[8:7] in register map. ........................................................................................ 21
•
Added two sentences to EXTERNAL_REFERENCE register description .......................................................................... 22
•
Changed address of LOW_FREQUENCY_NOISE_SUPPRESSION to 1[11] ................................................................... 23
•
Added reference to INTERP_ENABLE section .................................................................................................................. 25
•
Added a NOTE explaining readback from address 0x97 ................................................................................................... 28
•
Arranged TGC control register-description sections in alphabetical order by register name ............................................. 29
•
Changed "16 channels" to "eight channels" in the ANTI-ALIAS FILTER (AAF) section .................................................... 33
•
Changed "16 channels" to "eight channels" in the CLOCK INPUTS section ..................................................................... 33
•
Changed "clock channels" to "clock lines" in the Clock Inputs section .............................................................................. 33
•
Deleted two sentences from first paragraph of CLOCK INPUTS section ........................................................................... 33
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PACKAGE OPTION ADDENDUM
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11-Apr-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
(3)
(4)
AFE5801IRGCR
ACTIVE
VQFN
RGC
64
2000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
AFE5801
AFE5801IRGCT
ACTIVE
VQFN
RGC
64
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
AFE5801
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
Samples
PACKAGE MATERIALS INFORMATION
www.ti.com
12-Feb-2019
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
AFE5801IRGCR
VQFN
RGC
64
2000
330.0
16.4
9.3
9.3
1.5
12.0
16.0
Q2
AFE5801IRGCT
VQFN
RGC
64
250
180.0
16.4
9.3
9.3
1.5
12.0
16.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
12-Feb-2019
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
AFE5801IRGCR
VQFN
RGC
64
2000
350.0
350.0
43.0
AFE5801IRGCT
VQFN
RGC
64
250
213.0
191.0
55.0
Pack Materials-Page 2
GENERIC PACKAGE VIEW
RGC 64
VQFN - 1 mm max height
PLASTIC QUAD FLATPACK - NO LEAD
9 x 9, 0.5 mm pitch
Images above are just a representation of the package family, actual package may vary.
Refer to the product data sheet for package details.
4224597/A
www.ti.com
PACKAGE OUTLINE
RGC0064H
VQFN - 1 mm max height
SCALE 1.500
PLASTIC QUAD FLATPACK - NO LEAD
A
9.15
8.85
B
PIN 1 INDEX AREA
9.15
8.85
1.0
0.8
C
SEATING PLANE
0.05
0.00
0.08 C
2X 7.5
EXPOSED
THERMAL PAD
SYMM
(0.2) TYP
17
32
16
33
65
SYMM
2X 7.5
7.4 0.1
60X
0.5
1
48
49
64
PIN 1 ID
64X
0.5
0.3
64X
0.30
0.18
0.1
0.05
C A B
4219011/A 05/2018
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. The package thermal pad must be soldered to the printed circuit board for thermal and mechanical performance.
www.ti.com
EXAMPLE BOARD LAYOUT
RGC0064H
VQFN - 1 mm max height
PLASTIC QUAD FLATPACK - NO LEAD
( 7.4)
SEE SOLDER MASK
DETAIL
SYMM
64X (0.6)
49
64
64X (0.24)
1
48
60X (0.5)
(3.45) TYP
(R0.05) TYP
(1.16) TYP
65
SYMM
(8.8)
( 0.2) TYP
VIA
33
16
32
17
(1.16) TYP
(3.45) TYP
(8.8)
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE: 10X
0.07 MIN
ALL AROUND
0.07 MAX
ALL AROUND
METAL UNDER
SOLDER MASK
METAL EDGE
EXPOSED METAL
SOLDER MASK
OPENING
EXPOSED
METAL
NON SOLDER MASK
DEFINED
(PREFERRED)
SOLDER MASK
OPENING
SOLDER MASK DEFINED
SOLDER MASK DETAILS
4219011/A 05/2018
NOTES: (continued)
4. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
number SLUA271 (www.ti.com/lit/slua271).
5. Vias are optional depending on application, refer to device data sheet. If any vias are implemented, refer to their locations shown
on this view. It is recommended that vias under paste be filled, plugged or tented.
www.ti.com
EXAMPLE STENCIL DESIGN
RGC0064H
VQFN - 1 mm max height
PLASTIC QUAD FLATPACK - NO LEAD
SYMM
64X (0.6)
64X (0.24)
64
49
1
48
60X (0.5)
(R0.05) TYP
(1.16) TYP
65
SYMM
(8.8)
(0.58)
36X ( 0.96)
33
16
17
32
(0.58)
(1.16)
TYP
(8.8)
SOLDER PASTE EXAMPLE
BASED ON 0.125 MM THICK STENCIL
SCALE: 10X
EXPOSED PAD 65
61% PRINTED SOLDER COVERAGE BY AREA UNDER PACKAGE
4219011/A 05/2018
NOTES: (continued)
6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
www.ti.com
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IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
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