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Texas Instruments Continuous-Time Sigma-Delta ADCs Application notes
Continuous-Time Sigma-Delta ADCs
Literature Number: SNAA098
Continuous-Time Sigma-Delta ADCs
Scott D. Kulchycki, National Semiconductor
January 2008
Furthermore, the ability of CT∑Δ ADCs to scale with technology will allow such designs to take full advantage of future
CMOS processes.
ontinuous-time sigma-delta (CT∑Δ) analog-to-digital
(A/D) conversion technology shatters the conventional
wisdom that pipeline analog-to-digital converters
(ADCs) are the only conversion technique available for high
dynamic performance, sub-100 MSPS (mega-samples per
second) applications. Besides providing more power-efficient
operation, CT∑Δ technology also offers unique features that
greatly reduce the challenges of deploying such ADCs in
high-speed, high-performance systems. In short, CT∑Δ
technology means:
• An inherently power-efficient architecture that eliminates
the high-speed gain stages required for sampled-input
ADCs, such as pipeline or traditional discrete-time (DT) ∑Δ
(DT∑Δ) ADCs.
• An essentially alias-free Nyquist band that is made available by exploiting inherent over-sampling, an internal
lowpass CT loop filter, and on-chip digital filtering.
• A purely resistive input with no switching, which is easier
to drive and couples less noise to the overall system than
the switching input capacitors of a sampled-input ADC,
such as a pipeline or DT∑Δ ADC.
• An on-chip clock conditioning circuit to provide the
over-sampling clock to the internal modulator. This circuit
increases the frequency and the quality of the input clock,
yielding a low-jitter sampling edge and achieving
high-resolution performance without an expensive,
high-performance input clock.
• Easier migration to future CMOS process technologies.
In a CT∑Δ ADC, the impact of noise and nonlinearity
associated with the sampling process is significantly
reduced, allowing for the reduced supply voltages
required for future CMOS processes.
National’s CT∑Δ technology supports high-resolution ADCs up
to 16 bits and beyond with data output rates up to 100 MHz.
This paper will first review the ADC landscape and explain
how CT∑Δ technology is positioned within that space.
Next, the details and benefits of CT∑Δ technology as applied
to ADCs will be presented, focusing on the advantages and
benefits of National’s new ADC12EU050 versus competing
ADC technologies for high-resolution, sub-100 MSPS applications. Finally, the paper concludes with a summary of the
potential of CT∑Δ ADCs.
Data Conversion Fundamentals
ADCs perform two basic, fundamental operations: discretization
in time and discretization in amplitude. The two functions are
shown conceptually in Figure 1, though the actual ADC may not
be structured as such.
x(t) fs
xD[k] 11
4T 5T 6T
4T 5T 6T
T 2T 3T
T 2T 3T
Figure 1: Analog-to-digital conversion
The first operation of the ADC is to discretize in time, or
sample, the continually time-varying input analog signal. The
input signal is typically sampled at uniformly spaced times at
a frequency of fS, and the samples are thus separated by a
period TS = 1/fS. Once the input signal is sampled, the resultant
exists only as impulses at the sampling interval, kTS. However,
this sampled signal is still able to assume an infinite range of
values, and therefore cannot be represented precisely in a
digital form.
Taken together, the inherent benefits of CT∑Δ technology and
the opportunity to implement an on-chip clock conditioner
greatly simplify the signal path design by:
• Reducing power requirements.
• Eliminating the need (or reducing the requirements) for an
external anti-aliasing filter.
• Reducing the input driver requirements.
• Mitigating the need for high-quality clock sources without
sacrificing performance.
The second function of the ADC is to discretize the sampled signal in amplitude. That is, the ADC approximates the amplitude
of each sample with one of a finite number of possible values.
Because the output of the ADC can take on only a finite number of possible values, the amplitude of each sample can be
represented by a digital code whose bit length determines the
total number of possible converter outputs. The finite number
of output values in a converter introduces error into the digital
representation of the analog input. This so-called quantization
error limits the resolution of the converter.
adjacent comparators to distinguish between inputs that differ
by at least one LSB. The outputs of all the comparators form a
thermometer code, which is typically converted into a binaryweighted digital output.
For N-bit resolution, 2N-1 comparators are required for a flash
ADC, which limits their applicability to low-resolution applications since every additional bit of resolution doubles the
required number and hence, power and area, of comparators [ 1 ]. Increased bits also increase the required accuracy
of each comparator. For these reasons, flash converters are
typically limited to 8-bit resolution. Most design effort in flash
ADCs seeks to reduce their power in high-speed conversion
by minimizing the number of comparators applied, for example
by interpolating and folding designs. National has employed
such strategies to realize its industry-leading very low power,
gigahertz-rate sampling 8-bit ADCs.
ADC Architectures
In general, ADCs are divided into two broad categories:
Nyquist-rate converters and over-sampling converters.
These different converter classes typically offer different compromises between ADC resolution and output sampling rate.
Nyquist-Rate Converters
Pipeline ADCs
Nyquist-rate converters are those that operate at the minimum
sampling frequency necessary to capture all the information
about the entire input bandwidth, and therefore the output data
rate of a Nyquist-rate converter can be very high. Three of the
most popular Nyquist-rate converters are SAR (successive
approximation register), flash, and pipeline ADCs.
Pipeline ADCs have become the standard in data conversion
applications at 8-bit and higher resolutions for sampling rates
from 5 MHz to 100 MHz or more. Indeed, National offers
high-speed 8-, 10-, 12-, and 14-bit ADCs based on pipeline
architectures that achieve sampling rates up to 200 MSPS
and offer very large input sampling bandwidths.
A successive approximation register (SAR) ADC essentially
performs a binary search on the input signal using only a single
comparator [ 1 ]. That is, the ADC first determines whether the
input is greater or less than the midpoint between the reference
voltages and the result of that determination is the most significant bit (MSB) in the digital output. The half of the possible
values in which the input was not found are then discarded and
the ADC next determines whether the input is greater or less
than the midpoint of the remaining possible values; the result of
that operation is the next bit in the digital word.
Rather than providing enough comparators to check the input
against every possible input value as in a flash ADC, the
pipeline architecture employs multiple low-resolution flash
conversion stages cascaded in series to form the pipeline.
At each stage in the pipeline, the previous stage’s quantized
output is subtracted from the original input signal and the
remainder is sent to the next stage for ever finer quantization
[ 1 ]. This process is repeated several times as the signal progresses through the pipeline until the LSBs are determined and
the output of all the stages in the pipeline are then combined to
form the overall digital approximation to the input sample value.
This operation continues, approximating the input value with finer
resolution each successive cycle, reusing the same comparator
each cycle until the least significant bit (LSB) is determined and
the digital word is complete. Because the SAR requires N cycles
to produce an N-bit resolution output, the speed of state-of-theart SARs is often limited to several MSPS; however, the accuracy
can be high at low power since a single high-resolution (possibly
calibrated) comparator can be reused each cycle. National’s
low-power ADCs employ SAR architectures and achieve up to
14-bit resolution and 1-MSPS operation.
Because the pipeline is able to operate concurrently on many
samples, the ADC outputs a complete digital word every clock
cycle. This parallel processing allows the pipeline to offer
high resolution at the full Nyquist rate of the converter. But the
tradeoff for the high output rate is a delay from when the input is
first sampled to when its digital approximation is made available.
This delay is known as the latency of the pipe, which is typically
on the order of ten sample clock cycles. Fortunately for many
applications, the latency of a pipeline ADC is acceptable.
Challenges of the Pipeline ADC
Flash ADCs
As National’s high-speed ADC products demonstrate, the
pipeline ADC is clearly capable of providing high dynamic
performance at sampling rates up to 200 MSPS. Although the
pipeline architecture can achieve very high frequency opera-
A flash ADC features a cascade of parallel comparators connected to a resistor-ladder driven by the most positive and most
negative ADC reference voltages [ 1 ]. Each resistor-ladder tap
is designed to be one LSB away from its neighbors, allowing
tion at moderate-to-high resolution, it compromises in other
design parameters.
The sampling clock provided to the ADC is another important
determinant of the overall dynamic performance of a sampledinput ADC, especially for high-resolution, high-input frequency
applications [ 2 ]. The phase noise of a clock source will
appear as increased noise at the ADC output and therefore,
care must be taken by the system designer to ensure the
overall system resolution is not limited by their clock source.
Clock quality is especially important for high-speed highresolution ADCs because the demands on the purity of the
clock increase with increasing input frequency and increasing
ADC resolution.
High-Speed Circuits
Because each stage in the pipeline must process the previous
stage’s output, a constant input during the conversion process
is provided in each stage by a sample-and-hold (SHA) circuit
[ 1 ]. The first-stage SHA must maintain the accuracy of the
overall ADC at the full sampling rate, requiring the switchedcapacitor circuit to settle within a single clock period [ 1 ].
Similarly, the first-stage adder and DAC must be able to settle
their outputs within a single period. These speed requirements for the first stage (and to a lesser extent for subsequent
stages) typically require large-bandwidth amplifiers and other
circuits, which can lead to high power dissipation.
It is apparent from the preceding discussion that although
they are excellent candidates for high-speed, high-performance applications, pipeline and other sampled-input ADCs
do present design challenges for both the ADC designer
and the system designer using the ADC. In contrast to these
sampled-input ADCs, CT∑Δ ADCs do not require fast-settling
circuits or switched capacitors at their inputs and thus avoid
the increased ADC power and need for high-performance
drivers in high-resolution applications. CT∑Δ ADCs also
include significant anti-aliasing filtering, reducing or eliminating the need for an external AAF and preventing the need to
waste ADC bandwidth. Finally, CT∑Δ technology is well-suited
for migration to future CMOS processes. The advantages of
CT∑Δ technology for high-resolution, sub-100 MSPS applications, where both CT∑Δ and pipeline architectures are applicable, is further developed starting on page 5.
Thermal Noise
The maximum dynamic range of the pipeline ADC is
determined at least partly by the thermal noise at the input of
the converter, including the kT/C noise of the input sampling
capacitor. To reduce kT/C noise, a larger capacitor can be
used, but at the cost of increased switching noise at the input
and a more difficult-to-drive input, requiring a higher-performance, higher-power ADC driver.
Migration to Future CMOS Processes
As for all sampled-input ADCs, pipeline ADCs are also challenging to migrate to future CMOS processes. This challenge
arises from the switched-capacitor input, as a boosted CMOS
switch is often used to sample the input signal onto the sampling capacitor. As CMOS processes and their supply voltages
shrink, the overdrive voltage available for the CMOS switches
also shrinks, greatly reducing the range of input voltages that
can be sampled with high resolution. Furthermore, designing
switches with reduced threshold voltages that work well in
deep sub-micron processes can be difficult.
Over-Sampling ADCs
Whereas Nyquist-rate converters are generally well suited
to achieving moderate resolution at high input bandwidths,
over-sampling converters traditionally provided the opposite
tradeoff. Over-sampling converters are those for which the
sampling frequency is greater than the Nyquist-rate of the input
signal bandwidth and therefore, for a given converter sampling
rate, the output rate of an over-sampling converter will be lower
than that for a Nyquist-rate converter. However, in exchange
for Nyquist bandwidth, over-sampling converters can achieve
(without calibration) higher resolution than Nyquist-rate
converters, regardless of the inherent resolution of the CMOS
circuits composing the converter. Two types of such ADCs are
over-sampling ADCs and ∑Δ ADCs.
Input Filtering and Sampling Clock Requirements
A final challenge in using any sampled-input ADC, including
pipelines, concerns the external circuitry necessary to drive
the converter [ 2 ], specifically the input filtering network
and the sampling clock. With any sampled-input converter,
signals that can be aliased into the band of interest by the
sampling operation must be eliminated using an anti-aliasing
filter (AAF). Steep filter attenuation characteristics are hard to
achieve, leading designers to over-sample the signal of interest. Although over-sampling reduces the range of frequencies
that can alias down in-band and hence, lowers demands on
the AAF roll-off, over-sampling the ADC wastes the Nyquist
bandwidth, which increases system power. In addition, oversampling increases the processing demands on subsequent
digital circuitry.
Over-Sampling A/D Converters
Over-sampling an ADC can perhaps best be understood
beginning with a review of an N-bit flash ADC whose positive
reference voltage is +VREF/2 and whose negative reference
voltage is –VREF/2. The full input range of [-VREF/2, +VREF/2] is
subdivided into 2N smaller regions of width one LSB, or
Figure 3 shows the transfer function, known as the noise
transfer function (NTF), from the quantization noise, ei, to the
modulator output for various loop orders, L [ 3 ].
Because the output of the flash ADC assigns only one of a finite
set of outputs to an infinite range of inputs, the output digital
representation of an input is the sum of the original amplitude
plus an error signal due to the digital approximation; this error
signal is referred to as the quantization error. Typically, it is assumed that the quantization error power has a white frequency
spectrum, distributed equally between all frequencies from 0 to
the sampling frequency, fS. The noise power in the ADC output
is found by integrating this constant quantization noise density
from 0 to fS /2, the Nyquist bandwidth. It can be shown that the
SNR in dB of a flash ADC is SNR = 01.76 + 6N, where N is the
number of bits in the output [ 3 ].
From this figure, it is apparent that the modulator emphasizes
the quantization noise at higher frequencies, while suppressing
the lower-frequency inband noise. In effect, the quantization
noise is shifted to higher frequencies where it can be filtered
out later, significantly reducing the total inband quantization
noise power in the modulator output. Notice that for higher order modulators, more quantization noise is shaped out of band,
leaving less quantization noise inband. However, the loop filter
order can not be increased without bound due to the increased
difficulty of stabilizing higher-order loops [ 5 ].
It can be shown that for a ∑Δ modulator, the achievable SNR in dB is
SNR = 1.76 + 6N + (2L + 1) 10 log10 (M)
[ 3 ].
+ 10 log10 (2L + 1) - (2L)10 log10 (π)
Compared to the SNR of a simple over-sampling ADC then,
the SNR for a ∑Δ modulator is greater if M>π, which is usually
the case. As the over-sampling ratio increases, ∑Δ modulators yield increasingly higher resolution than simple oversampling. This equation shows that the SNR increase due to
the over-sampling ratio is multiplied by (2L+1), and therefore
the bandwidth/resolution tradeoff is much more efficient for
∑Δ modulators than for over-sampling alone, especially as the
modulator order increases. ∑Δ modulators attain this improved
resolution because of the quantization error noise shaping
provided by feedback in the ∑Δ loop.
The basic principle of operation for a ∑Δ modulator is to enclose
a simple quantizer in a feedback loop to shape the quantization noise such that most of the noise is shifted out of the band
of interest, where it can later be suppressed by filtering. An
example of a simple ∑Δ modulator is shown in Figure 2, where
the quantizer has been modeled by the additive white noise
source, ei.
The signal at the output of the quantizer in the ∑Δ modulator contains the input signal and other noise and distortion components
in addition to the shaped quantization noise; furthermore, the loop
output data rate is M times higher than desired [ 4 ]. The final step
of the ∑Δ A/D conversion process is to remove the out-of-band
quantization noise and downsample the output to the desired data
rate: this function is performed by the decimation filter.
Figure 3: Quantization noise shaping in a ∑Δ modulator
The efficiency of the bandwidth/resolution tradeoff in oversampling can be extended by shaping the spectrum of either
the input signal or the quantization noise. The former is typically accomplished using a delta modulator, while the latter is
accomplished with a ∑Δ modulator [ 4 ]. Because ∑Δ modulators
are much more robust to circuit non-idealities than delta modulators, they are usually the preferred architecture [ 4 ].
Frequency (normalized to fs)
Sigma-Delta Modulator ADCs
The preceding discussion concerning the white noise nature of
quantization error distributed between DC and fS /2 suggests a
simple way to reduce noise in the ADC output signal. Because
the finite-power quantization noise is distributed equally across
all frequencies, by restricting the allowable bandwidth of a
converter, the total integrated noise in the output can be reduced and hence, the SNR of signals in that bandwidth will be
increased. That is, if the input bandwidth is limited to fS/2M, the
total integrated noise will be reduced by a factor of M, known
as the over-sampling ratio. Therefore, the achievable SNR in dB
of an over-sampling ADC is
SNR = 1.76 + 6N + 10log10 (M) [ 3 ].
With over-sampling, the SNR increases by one bit (6 dB) for
every fourfold increase in M.
Figure 2: ∑Δ Modulator
Decimation Filter
Challenges of the CT∑Δ ADC
The digital filter at the output of the ∑Δ modulator must lowpass
filter the out-of-band quantization noise and resample the digital
data from the loop sample rate, MfS, to fS, the desired ADC output
rate [ 4 ]. To reduce implementation complexity, the decimation
filter is usually designed in multiple stages [ 4 ].
Of course, as pipeline ADCs offer high speed operation while
compromising other design parameters, the benefits of CT∑Δ
operation also come at the cost of some design challenges for
both the ADC designer and the system architect. A sampledinput, SC ADC can often operate over a wide range of sampling
frequencies from near-zero to its maximum rate. However, the
dynamics of the CT∑Δ are set by the RC or C/gm product of its
component integrators; therefore, the integrator time constants
must be tunable to allow for process variation [ 2 ]. In addition,
the loop dynamics will not scale with sampling frequency, limiting the allowable sampling rate operating range.
One simple implementation uses a simple accumulate-and-dump
or sinc filter as a first stage that is generally limited to a loworder decimation to prevent significant in-band droop; the sinc
transfer function prevents signals at multiples of the resampled
rate from aliasing down in-band. Such a configuration is then
often followed by a lowpass filter that decimates the signal from
the intermediate output rate of the sinc-filter to the desired rate
of fS/M. This lowpass filter can also be designed to compensate
for the remaining in-band droop of the sinc filter [ 6 ]. The decimation filtering in a ∑Δ ADC typically results in longer latencies than
those found in a pipeline ADC but many applications can tolerate
this increased latency.
The input bandwidth of a ∑Δ is also limited to the ADC’s first
Nyquist band. In a Nyquist-rate ADC wherein full-rate sampling
occurs at the system input, the input bandwidth can be many
times larger than the Nyquist rate of the converter, allowing for
IF-sampling. Conversely in a ∑Δ ADC, because of the lowpass
decimation filtering, signals outside the first Nyquist zone will
be removed from the output spectrum. In addition, although a
DT∑Δ would allow signals around its loop sampling rate, MfS, to
fold down inband, the inherent anti-aliasing filtering in a CT∑Δ
ADC precludes this from happening. Therefore, input signals
must be mixed down into the first Nyquist zone to be digitized
by a CT∑Δ ADC.
Continuous-Time ∑Δ Modulators
The first recognizable ∑Δ modulator, introduced in 1962, was
actually implemented as a CT circuit [ 7 ]. Indeed, CT implementations of ∑Δ modulators have appeared regularly since then,
but when switched-capacitor (SC) circuits were introduced,
most ∑Δ modulators were implemented with DT loop filters. SC
circuits remain popular because of their insensitivity to signal
waveform characteristics. In addition, the time constants of SC
integrators scale with sampling frequency, allowing for greater
system flexibility [ 8 ]. However, interest in CT∑Δ modulators
has been renewed because of some of the benefits, such as
employing lower-power integrator amplifiers and including
inherent anti-aliasing filtering versus sampled-input ADCs [ 7 ].
Finally, because of its over-sampling operation, the output rates
of CT∑Δ ADCs are currently limited to less than 100 MSPS while
pipeline ADCs are capable of operation up to 500 MSPS and
beyond. Indeed, for a given technology, Nyquist-rate converters
will always be able to operate faster than ∑Δ ADCs because of
the over-sampling necessary in a ∑Δ design.
Fortunately, the benefits of CT∑Δ technology outweigh the
drawbacks for high-resolution applications at sampling rates
below 100 MSPS. The next section focuses on National’s CT∑Δ
ADCs, highlighting the performance enhancement achieved
versus sampled-input ADCs, such as pipeline and DT∑Δ ADCs.
CT∑Δ ADCs differ from sampled-input ADCs, such as pipeline
and DT∑Δ ADCs, in two important ways:
• A CT∑Δ modulator uses CT integrators rather than DT
integrators or circuits. That is, rather than SC circuits, the
CT∑Δ modulator employs continuous-time circuits, often RC
or C/gm integrators.
• The sampling operation in a CT∑Δ modulator occurs at the
output of the forward loop filter, before the quantizer. In
contrast, for a sampled-input ADC, the sampling occurs at
the input of the ADC.
Advantages of National’s CT∑Δ ADCs
National’s new ADC12EU050 is the industry’s first productionready CT∑Δ ADC. The product offers performance improvement both because of the inherent advantages of CT∑Δ
technology versus sampled-input ADCs and also because of
additional circuitry that National has integrated on-chip.
These differences between CT∑Δ ADCs and sampled-input
ADCs give rise to significant performance differentiation
between the two. Specifically, CT∑Δ ADCs operate at lower
power, include significant anti-aliasing filtering, and present a
much quieter input stage. All of these benefits of CT∑Δ
technology are realized in National’s new ADC12EU050, as
explained in the next section.
Lower Power
The most significant benefit of the CT∑Δ architecture is its low
power consumption relative to comparable sampled-input
ADCs in the high-resolution, sub-100 MSPS application space.
A common way to compare the performance of ADCs is an energy figure of merit (FOM) which typically measures the ratio of
the overall power of the ADC to the resolution and bandwidth of
sampling, which attenuates signals around the modulator loop
sampling rate (MfS) that can alias down in-band. Furthermore,
because these aliased signals are then injected at the input of
the internal quantizer, they are noise-shaped by the loop in the
same manner quantization noise is shaped. These two effects
enable the CT∑Δ to offer significant anti-aliasing filtering versus
the DT∑Δ in addition to the benefits of over-sampling and digital
filtering versus a pipeline design. Figure 4 summarizes the antialiasing performance of CT∑Δ ADCs versus pipeline ADCs.
its output. Aided by the inherent efficiency advantage of CT∑Δ
technology, the ADC12EU050 provides excellent performance at
extremely low power, yielding an outstanding FOM.
The power advantage of a CT∑Δ implementation arises because
of the internal circuitry. In any sampled-input SC circuit – including both pipeline and traditional DT∑Δ ADCs – the internal
amplifiers must settle within some target resolution each
period. This places significant speed constraints on the internal
amplifiers, increasing the power consumption [ 7 ] and limiting
the maximum achievable sampling rate.
Interferer Aliasing
f IN
Requires External
Anti-Aliasing Filter
Noise Aliasing
In a CT∑Δ with CT feedback, because the amplifier output never
attempts to switch its output voltage instantaneously, there
is no output settling required and, hence the amplifier speed
constraints are relaxed [ 7 ]. Although an exact comparison is
difficult, the SC nature of a sampled-input ADC does necessitate higher-speed amplifiers than does a CT∑Δ implementation,
leading to higher power dissipation for a pipeline or DT∑Δ ADC
in general. The lack of a need for high-speed settling in CT∑Δ
ADCs is also the reason they are able to achieve higher sampling rates than traditional DT∑Δ ADCs in a given technology.
f s /2
Aliasing increases in-band noise and interferers
Brickwall Filter Eliminates Aliasing
Anti-Aliasing Filter
f IN
f s /2
Figure 4: Anti-aliasing performance of CT∑Δ and pipeline ADCs.
This significant inherent anti-aliasing filtering greatly reduces the
requirements on or may reduce the need for an external AAF.
Low power, high energy efficiency operation is obviously important for any system but portable devices especially benefit
since reduced power extends battery life and reduces heat dissipation, which is particularly important for handheld applications including handheld ultrasound systems. The 1.2-V power
supply of the ADC12EU050 also positions the converter well for
single battery-powered operation.
The benefit of the anti-aliasing performance of the CT∑Δ
cannot be over-stated; the anti-aliasing requirements are
dependent upon the application and can be a significant
system challenge both in terms of design complexity and
overall system form factor and cost. As previously discussed,
the anti-aliasing requirements can be relaxed in a pipeline or
other Nyquist-rate ADC system by increasing the sampling
rate beyond twice the desired input bandwidth but this wastes
bandwidth and drives down the overall power efficiency of
the system. The design of an analog anti-aliasing filter possessing a steep cut-off characteristic and a very flat passband is a challenging task demanding high-order, potentially
high insertion-loss filter networks, which may necessitate
additional gain in the signal path to compensate for that loss.
Anti-Aliasing Filtering
The CT∑Δ ADC architecture eliminates the need for stringent
input filtering because of its inherent anti-aliasing filtering. In
the ADC12EU050, many of the performance characteristics of
the anti-aliasing filter are set in the digital domain, allowing for
a very high level of passband flatness and steep roll-off (high
effective order).
The anti-aliasing performance of the CT∑Δ is a result of its implementation as both a ∑Δ modulator and a CT circuit. As is the case
for any ∑Δ ADC (CT or DT), the over-sampling and subsequent
decimation filtering of the modulator output results in a very
sharp roll-off lowpass filter with a cutoff frequency at half the
ADC output rate. In contrast, a Nyquist-rate ADC without oversampling must employ a high-order external lowpass filter before
the ADC to prevent signals around multiples of the output sampling rate from aliasing down in-band, as discussed for pipelines
in Input Filtering and Sampling Clock Requirements on page 3.
By eliminating the need for additional over-sampling as with
a sampled-input ADC, the CT∑Δ allows the system designer
to use almost the entire Nyquist bandwidth of the converter,
greatly improving power efficiency. Furthermore, by eliminating the need for expensive, lossy external anti-aliasing filters,
the ADC12EU050 also reduces the demands on the ADC driver,
further reducing system design complexity, cost, and power.
Quiet, Easy-to-Drive Input
However, in addition to the inherent benefit of ∑Δ architectures, the CT offers an additional benefit even over DT∑Δ ADCs.
Because the CT∑Δ ADC samples at the output of the forward
loop filter, the signal is first lowpass filtered by the loop before
The CT∑Δ ADC also offers a much quieter input than a sampledinput ADC because of the CT nature of the internal circuitry.
In a sampled-input ADC, such as a pipeline or traditional DT∑Δ
ADC, the input stage consists of a switched capacitor that is
Instant Overload Recovery
usually large to reduce the overall thermal noise of the ADC.
Driving this large switching capacitor is difficult, especially in
a DT∑Δ whose internal modulator is sampling at several times
the output data rate. In addition, the large switching noise
from such inputs can couple to a system, reducing the overall
system performance. The input voltages that can be applied
to a switched-capacitor input are also limited because of the
gate-source voltage of the input sampling switches.
As opposed to a SC sampled input, CT∑Δ technology instead
presents a constant, resistive input, as illustrated in Figure 5.
Because the ∑Δ modulator is a feedback loop, it is susceptible
to overloading in the presence of large input signals. In a typical
∑Δ modulator, such overloading may require the loop to be reset,
losing data previously stored in the loop and causing a large
glitch in the ADC output. Instead of resetting the loop, the modulator can be allowed to continue operation, allowing the overload
condition to simply work its way out of the loop—but waiting to
clear the overload condition can require several clock cycles,
during which time the ADC output data is corrupted.
The ADC12EU050 includes circuitry that recovers immediately
from an overload condition. When this instant overload recovery (IOR) circuitry is enabled, the ADC maintains signal integrity in the event of an input overload condition, allowing it to
recover faster than even a pipeline ADC.
Technology Scaling
Finally, CT∑Δ technology is capable of scaling well with future
technologies, ensuring a lengthy presence in the ADC marketplace. As discussed above, because the sampling operation in
a CT∑Δ occurs at the output of the loop filter, the performance
impact due to errors in the sampling operation will be greatly
reduced. In sampled-input ADCs such as pipeline or DT∑Δ
ADCs, the sampling occurs at the ADC input and therefore, any
sampling errors are significant. It is for this reason that CT∑Δ
ADCs are more amenable to future, scaled CMOS processes.
Non-idealities in the sampling circuit caused by reduced overdrive, leakage, or other effects in future processes will impact
pipeline, DT∑Δ, and other sampling-input ADCs much more than
Figure 5: Model of the CT∑Δ ADC input
Because the input of a CT∑Δ is not sampled, there are no
switching capacitors and it is easier to drive the input, allowing
for the use of cheaper, lower-power driving circuits. In addition,
the lack of input switching noise will reduce noise coupling to
the system, improving its overall performance. Finally, without
any switches at the input to restrict the input voltage swing,
the allowable input voltage range can be higher than for a SC
sampled-input ADC; indeed, the input voltage can even exceed
the supply rails.
Low-Jitter PLL Provides an Accurate
Sample Clock
National’s New CT∑Δ ADC
The ADC12EU050 12-bit, ultra-low-power, octal CT∑Δ ADC offers
an alias-free sample bandwidth of 20 to 25 MHz and a conversion rate of 40 MSPS to 50 MSPS. The device features 68 dB
of signal-to-noise and distortion (SINAD) and a signal-to-noise
ratio (SNR) of 70 dB full scale (dBFS). Operating from a 1.2V
supply, it consumes 44 mW per channel at 50 MSPS for a total
power consumption of only 350 mW, 30% lower than currently
available competitive pipeline products (see Figure 6).
ADC Power (mW/channel)
A low-jitter sampling clock is crucial in all high-speed, highresolution data conversion systems to realize the full resolution
of an ADC. The modulator over-sampling clock in National’s ADC12EU050 drives the quantizer of the internal ∑Δ loop. This clock
is provided by an on-chip clock conditioner, comprising a PLL
and VCO. The high-performance PLL uses an on-chip LC-tuned
circuit to create a high-Q resonator. This on-chip clock circuit
multiplies up the frequency and provides low-jitter sampling
edges to the modulator loop, allowing for the benefits of CT
∑Δ ADCs to be realized without requiring a high-performance,
high-cost external clock source. The system designer needs
simply to provide a moderate-quality, low-cost crystal at the desired output sampling rate (40-50 MSPS), and the ADC12EU050’s
on-chip clock circuitry takes care of the rest.
A further advantage of the on-chip precision clock is that it can
be routed to external circuits and used as a system reference
clock for other time-critical parts of the system, potentially
eliminating the extra cost of a low-jitter source and saving both
design effort and board area.
Lowest Power
Figure 6: Power Consumption Comparison for ADC12EU050
The ADC12EU050 reduces interconnection complexity by using
programmable serialized outputs, which offer industry-standard
low-voltage differential signaling (LVDS) and scalable lowvoltage signaling (SLVS) modes. The ADC12EU050 operates
over the -40 degrees C to 85 degrees C temperature range and
is supplied in a 68-pin LLP® package.
The author thanks David Barkin, Heribert Geib, Bumha Lee, Stephan
Mechnig, and everyone else at National Semiconductor who helped
with the development and editing of this paper.
[ 1 ] B. Razavi, Data Conversion System Design. Piscataway, NJ: IEEE Press, 1995.
[ 2 ] G. Mitteregger, C. Ebner, S. Mechnig, T. Blon, C. Holuigue, and E. Romani, “A
20-mW 640-MHz CMOS Continuous-Time ∑Δ ADC With 20 MHz Signal Bandwidth, 80-dB Dynamic Range and 12-bit ENOB,” IEEE Journal of Solid-State
Circuits, vol. 41, no. 12, pp. 2641-2649, December 2006.
National’s advanced ADC12EU050 ADC solution finally realizes
the leap in performance that CT∑Δ ADCs have promised for
more than 40 years, successfully migrating the technology from
the research lab to the production line. The power dissipation
is 30% lower than for any of the competitive pipeline products
and it offers 12-bit resolution at an output rate up to 5 times the
fastest currently available DT∑Δ ADC.
[ 3 ] S. D. Kulchycki, “Continuous-Time ∑Δ Modulation for High-Resolution, Broadband A/D Conversion,” Ph.D. dissertation, Stanford University, March 2007.
[ 4 ] J. C. Candy, “An Overview of Basic Concepts,” in Delta-Sigma Data Converters, S. R. Norsworthy, R. Schreier, and G. C. Temes, Eds. Piscataway, NJ: IEEE
Press, 1997, pp. 1-43.
[ 5 ] B. Brandt, D. Wingard, and B. Wooley, “Second-order sigma-delta modulation
for digital-audio signal acquisition,” IEEE Journal of Solid-State Circuits, vol.
26, no. 4, pp. 618-627, April 1991.
The CT∑Δ technology on which the ADC12EU050 is based also
offers significant inherent anti-aliasing filtering and provides
a low-noise, easy-to-drive input stage. To fully exploit these
considerable benefits of CT∑Δ technology, the ADC12EU050
also includes an on-chip clock conditioner that eliminates the
need for a high-performance, expensive clock. Finally, the
ADC12EU050 avoids the hazards of input overload present in ∑Δ
ADCs by offering a means for recovering immediately from an
input overload event.
[ 6 ] S. R. Norsworthy and R. E. Crochiere, “Decimation and Interpolation for Δ∑
Conversion,” in Delta-Sigma Data Converters, S. R. Norsworthy, R. Schreier,
and G. C. Temes, Eds. Piscataway, NJ: IEEE Press, 1997, pp. 406-446.
[ 7 ] E. van der Zwan and E. C. Djikmans, “A 0.2 mW CMOS ∑−Δ modulator for
speech coding with 80 dB dynamic range,” IEEE Journal of Solid-State
Circuits, vol. 31, no. 12, pp. 1873-1880, December 1996.
[ 8 ] V. Dias, G. Palmisano, and F. Maloberti, “Noise in mixed continuous-time
switched-capacitor sigma-delta modulators,” IEE Proceedings G Circuits,
Devices, and Systems, vol. 139, no. 6, pp. 680-684, December 1992.
Beyond the ADC12EU050, National is developing additional CT∑Δ
ADCs for high-resolution applications at sampling rates below
100 MSPS. Their many benefits and ability to scale well with
technology ensure these types of ADCs will find increasing adoption in future systems. National’s expertise in CT∑Δ ADCs ensures
we will continue to be at the forefront of industry adoption. ■
For more information on CT∑Δ ADCs, visit: national.com/adc
National Semiconductor
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