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Texas Instruments AN-1148 Linear Regulators: Theory of Operation and Compensation (Rev. B) Application notes
Application Report
SNVA020B – May 2000 – Revised May 2013
AN-1148 Linear Regulators: Theory of Operation and
Compensation
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ABSTRACT
This application note discusses the operation and compensation of linear regulators.
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Contents
Introduction .................................................................................................................. 3
The NPN Regulator ......................................................................................................... 3
The LDO Regulator ......................................................................................................... 3
The Quasi-LDO Regulator ................................................................................................. 3
Regulator Operation ........................................................................................................ 4
Performance Comparison .................................................................................................. 4
Feedback and Loop Stability .............................................................................................. 4
7.1
Bode Plots .......................................................................................................... 5
7.2
Loop Gain ........................................................................................................... 5
7.3
Feedback ............................................................................................................ 6
7.4
Phase Shift .......................................................................................................... 6
7.5
Phase Margin ....................................................................................................... 7
7.6
Poles ................................................................................................................. 7
7.7
Zeros ................................................................................................................. 7
7.8
Bode Plot Analysis ................................................................................................. 8
NPN Regulator Compensation ............................................................................................ 9
LDO Regulator Compensation ............................................................................................ 9
LDO Compensation Using ESR ......................................................................................... 10
ESR and Stability .......................................................................................................... 11
11.1 High ESR .......................................................................................................... 11
11.2 Low ESR ........................................................................................................... 12
Output Capacitor Selection ............................................................................................... 12
Quasi-LDO Compensation ............................................................................................... 13
Low-ESR Specific LDOs .................................................................................................. 13
The FET Advantage ....................................................................................................... 13
List of Figures
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NPN Regulator .............................................................................................................. 3
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PNP LDO Regulator ........................................................................................................ 3
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Quasi-LDO Regulator
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Voltage Regulator ........................................................................................................... 4
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Typical Bode Plot
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Loop Gain Example ......................................................................................................... 5
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Feedback Signals ........................................................................................................... 6
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Phase Shift Map............................................................................................................. 6
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Pole Gain/Phase Plot ....................................................................................................... 7
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Zero Gain/Phase Plot....................................................................................................... 7
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Bode Plot with Phase Info ................................................................................................. 8
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Bode Plot for NPN Regulator .............................................................................................. 9
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LDO Gain Plot Without Compensation ................................................................................. 10
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Capacitor Showing ESR .................................................................................................. 10
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ESR Zero Stabilizes LDO
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ESR Range for Typical LDO ............................................................................................. 11
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High ESR Causes Unstable Loop ....................................................................................... 12
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Low ESR Causes Unstable Loop
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P-FET LDO Regulator .................................................................................................... 13
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Introduction
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1
Introduction
The explosive proliferation of battery powered equipment in the past decade has created unique
requirements for a voltage regulator that cannot be met by the industry standards like the LM340 or the
LM317. These regulators use an NPN Darlington pass transistor (Figure 1), and will be referred to in this
document as NPN regulators. The demand for higher performance is being met by the newer lowdropout (LDO) regulators and quasi-LDO regulators.
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The NPN Regulator
The NPN Darlington pass transistor with PNP driver used in an NPN regulator requires that at least 1.5V
to 2.5V be maintained from input-to-output for the device to stay in regulation. This minimum voltage
"headroom" (called the dropout voltage) is:
VDROP = 2VBE + VSAT (NPN REG)
(1)
Figure 1. NPN Regulator
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The LDO Regulator
In the low-dropout (LDO) regulator, the pass transistor is a single PNP transistor (Figure 2). The big
advantage of the LDO is that the PNP pass transistor can maintain output regulation with very little voltage
drop across it:
VDROP = VSAT (LDO REGULATOR)
(2)
Full-load dropout voltages < 500 mV are typical. At light loads, dropout voltages can fall as low as 10 to 20
mV.
Figure 2. PNP LDO Regulator
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The Quasi-LDO Regulator
Another regulator configuration that is becoming very popular in certain applications (like 5 - 3.3V
conversion) is the quasi-LDO regulator (Figure 3). The quasi-LDO is so named because it is "half way"
between the NPN Darlington and the true LDO. The pass transistor is made up of a single NPN transistor
being driven by a PNP. As a result, the dropout voltage is less than the NPN Darlington regulator, but
more than an LDO:
VDROP = VBE + VSAT
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Regulator Operation
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Figure 3. Quasi-LDO Regulator
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Regulator Operation
All three of these regulator types regulate the output voltage to a fixed (constant) value using the same
technique (Figure 4).
The output voltage is sampled (measured) through a resistive divider which is fed into the inverting input
of the error amplifier. The non-inverting input is tied to a reference voltage, which is derived from an
internal bandgap reference. The error amplifier will always try to force the voltages at it's input to be equal.
To do this, it sources current as required to provide sufficient load current to maintain the output voltage at
the regulated value which is given by:
VOUT = VREF (1 + R1/R2)
(4)
Figure 4. Voltage Regulator
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Performance Comparison
The primary differences in performance between the NPN, LDO, and quasi-LDO are in the parameters of
dropout voltage (previously defined) and ground pin current. For this analysis, we will define ground pin
current (IGND) as shown in Figure 4, neglecting the small IC bias currents which also flow to ground. It can
be seen that the value of IGND is the load current IL divided by the gain of the pass transistor.
The high gain of the Darlington in an NPN regulator means it requires very little drive to source IL, so it's
ground pin current is very low (typically a few mA). The quasi-LDO also has very good performance, with
products like Texas Instrument's LM1085 being able to source more than 3A with less than 10 mA of
ground pin current.
The ground pin current of an LDO is typically much higher. At full load current, PNP beta values of 15 - 20
are not unusual, which means the LDO ground pin current can be as high as 7% of the load current.
A big advantage of NPN regulators is that they are unconditionally stable (most require no external
capacitors). An LDO does require at least one external capacitor on the output to reduce the loop
bandwidth and provide some positive phase shift. Quasi-LDOs typically require some output capacitance,
but much less than an LDO and with less restrictive limits on its performance characteristics.
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Feedback and Loop Stability
All voltage regulators use a feedback loop to hold the output voltage constant. The feedback signal
experiences changes in both gain and phase as it goes through the loop, and the amount of phase shift
which has occurred at the unity gain (0 dB) frequency determines stability.
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7.1
Bode Plots
Understanding stability requires the use of Bode Plots, which show the loop gain (in dB) plotted as a
function of frequency (Figure 5). Loop gain and associated terms are defined in the next sections.
Loop gain can be measured on a network analyzer, which injects a low-level sine wave into the feedback
path and then measures the gain response while the frequency of the signal is swept from DC up to the
frequency where the gain drops well below 0 dB.
Bode plots are convenient tools because they contain all the information necessary to determine if a
closed-loop system is stable. However, decoding the information contained in a Bode plot requires
understanding the key elements: loop gain, phase margin, poles and zeros.
Figure 5. Typical Bode Plot
7.2
Loop Gain
Every closed-loop system has a characteristic called loop gain. In this analysis of voltage regulators, loop
gain will be defined as the magnitude of the voltage gain that the feedback signal experiences as it travels
through the loop. The block diagram of the LDO in Figure 2 will be redrawn to illustrate this concept
(Figure 6).
Figure 6. Loop Gain Example
A transformer is used to inject an AC signal into the feedback path between points "A" and "B". Using this
transformer, a small-signal sine wave is used to "modulate" the feedback signal. The AC voltages at "A"
and "B" are measured and used to calculate loop gain.
The loop gain is defined as the ratio of the two voltages:
Loop Gain = VA/VB
(5)
it is important to note that the signal starting at the VB point has a phase shift introduced into it as it travels
through the loop (eventually arriving at the VA point). The amount of phase shift is critical in determining
stability.
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Feedback and Loop Stability
7.3
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Feedback
Feedback is used in all voltage regulators to hold the output voltage constant. The output voltage is
sampled through a resistive divider (Figure 6), and that signal is fed back to one input of the error
amplifier. Since the other input of the error amplifier is tied to a reference voltage, the error amplifier will
supply current as required to the pass transistor to keep the regulated output at the correct DC voltage.
It is important to note that for a stable loop, negative feedback must be used. Negative feedback
(sometimes called degenerative feedback) is opposite in polarity to the source signal (see Figure 7).
Figure 7. Feedback Signals
Because it is opposite in polarity with the source, negative feedback will always cause a response by the
loop which opposes any change at the output. This means that if the output voltage tries to rise (or fall),
the loop will respond to force it back to the nominal value.
Positive Feedback occurs when the feedback signal has the same polarity as the source signal. In this
case, the loop responds in the same direction as any change which occurs at the output. This is clearly
unstable, since it does not cancel out changes in output voltage, but amplifies them.
It should be obvious that no one would intentionally design positive feedback into the loop of a linear
regulator, but negative feedback becomes positive feedback if it experiences a phase shift of 180°.
7.4
Phase Shift
Phase shift is defined as the total amount of phase change (referred to the starting point) that is
introduced into the feedback signal as it goes around the loop. Phase shift (expressed in degrees) is most
often measured using a network analyzer.
Ideal negative feedback is 180° out of phase wit the source (Figure 8), so its "starting point" is at −180°.
This "180°" offset can also be seen in Figure 7, as the negative feedback waveforms are exactly one half
cycle shifted with respect to each other.
Figure 8. Phase Shift Map
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It can be seen that starting at −180°, an additional phase shift of 180° (positive or negative) brings the
signal back to zero, which is in phase with the source signal and would cause the loop to be
unstable.
7.5
Phase Margin
Phase margin is defined as the difference (in degrees) between the total phase shift of the feedback
signal and −180° at the frequency where the loop gain is equal to 0 dB (unity gain). A stable loop
typically needs at least 20° of phase margin.
Phase shift and phase margin can be calculated using the poles and zeros present in the Bode plot.
7.6
Poles
A pole (Figure 9) is defined as a point where the slope of the gain curve changes by −20 dB/decade (with
reference to the slope of the curve prior to the pole). Note that the effect is additive: each additional pole
will increase the negative slope by the factor "n" x ( −20 dB/decade), where "n" is the number of
additional poles.
Figure 9. Pole Gain/Phase Plot
The phase shift introduced by a single pole is frequency dependent, varying from 0 to −90° (with a phase
shift added by a pole (frequency). The most important point is that nearly all of the phase shift added by a
pole (or zero) occurs within the frequency range one decade above and one decade below the pole (or
zero) frequency.
NOTE: a single pole can add only −90° of total phase shift, so at least two poles are needed to reach
−180° (which is where instability can occur).
7.7
Zeros
A zero (Figure 10) is defined as a point where the gain changes by +20 dB/decade (with respect to the
slope prior to the zero). As before, the change in slope is additive with additional zeros.
Figure 10. Zero Gain/Phase Plot
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Feedback and Loop Stability
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The phase shift introduced by a zero varies from 0 to +90°, with a +45° shift occurring at the frequency of
the zero.
The most important thing to observe about a zero is that it is an "anti-pole", which is to say its effects on
gain and phase are exactly the opposite of a pole.
This is why zeros are intentionally added to the feedback loops of LDO regulators: they can cancel out of
the effect of one of the poles that would cause instability if left uncompensated.
7.8
Bode Plot Analysis
A bode plot which contains three poles and one zero (Figure 11) will be analyzed for gain and phase
margin.
The DC gain is assumed to be 80 dB, with the first pole occurring at 100 Hz. At that frequency, the slope
of the gain curve changes to −20 dB/decade.
Figure 11. Bode Plot with Phase Info
The zero at 1 kHz changes the slope back to 0 dB/decade until the second pole at 10 kHz, where the gain
curve slope returns to −20 dB/decade.
The third and final pole at 100 kHz changes the gain slope to the final value of −40 dB/decade.
It can also be seen that the unity-gain (0 dB) crossover frequency is 1 MHz. The 0 dB frequency is
sometimes referred to as the loop bandwidth.
The plot of the phase shift shows how the various poles and zeros contribute their effect on the feedback
signal. To produce this plot, the phase shift at each frequency point was calculated based upon summing
the contributions of every pole and zero at that frequency. The phase shift at any frequency "f" which is
caused by a pole frequency located at frequency "fp" can be calculated from:
Pole Phase Shift = − arctan (f/fp)
(6)
The phase shift resulting from a zero located at frequency "fz" can be found using:
Zero Phase Shift = arctan (f/fz)
(7)
Is this loop stable? To answer that question, we need only the phase shift at 0 dB (which is 1 MHz in this
case). Finding this does not require complex calculations:
As stated in the previous sections, a pole or zero contributes nearly its full phase shift in the frequency
range one decade above and below the center frequency of the pole (or zero). Therefore, the first two
poles and the first zero contribute their full phase shifts of −180° to +90°, respectively, resulting in a net
phase shift of −90°.
The final pole is exactly one decade below the 0 dB frequency. Using the formula for Pole Phase Shift,
this pole will contribute −84° of phase shift @ 1 MHz. Added to the −90° from the two previous poles and
the zero, the total phase shift is −174° (which means the phase margin is 6°). This loop would either
oscillate or ring severely.
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NPN Regulator Compensation
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8
NPN Regulator Compensation
The pass transistor of the NPN regulator (see Figure 1) is connected in a circuit configuration known as
common collector. An important characteristic of all common collector circuits is low output impedance,
which means the pole from the power stage that it places in the loop gain occurs at a very high frequency.
The NPN regulator uses a technique called dominant pole compensation because it has no inherent
low-frequency poles. In this case, a capacitor is built into the IC which places a pole in the loop gain at a
low frequency (Figure 12).
Figure 12. Bode Plot for NPN Regulator
This dominant pole (shown as P1) for a typical NPN regulator is set at about 100 Hz. The 100 Hz pole
causes the gain to decrease at a rate of −20 dB/decade until the second pole (P2) which is located at 3
MHz. At that point, the slope of the gain plot changes to −20 dB/decade.
The frequency of P2 is primarily due to the NPN power transistor and associated drive circuitry, so it is
sometimes referred to as the power pole. Since P2 occurs at a frequency where the loop gain is −10 dB,
its contribution to phase shift at the 0 dB frequency (1 MHz) will be small.
To determine stability, it only requires that the phase margin at the 0 dB frequency be calculated:
The first pole (P1) will contribute −90° of phase shift, but the second pole (P2) will add only −18° of
negative phase shift @ 1 MHz (0 dB). This means the total phase shift @ 0 dB is −108°, which yields a
phase margin of 72° (which is very stable).
It should also be noted that simple observation would clearly show this loop is stable, since reaching
−180° of phase shift (the point of instability) would require the full contribution of −90° (each) from both
poles, and P2 is too high in frequency to contribute significant phase shift at the 0 dB frequency (1 MHz).
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LDO Regulator Compensation
The PNP transistor in an LDO regulator (Figure 2) is connected in a configuration called common
emitter, which has a higher output impedance than the common collector configuration in the NPN
regulator. This adds an additional low-frequency pole whose frequency is dependent both on load
resistance and output capacitance. The frequency of this pole (which will be designated PL for load
pole) is found from:
f (PL) = 1/ (2π x RLOAD x COUT)
(8)
The presence of the frequency-variable load pole PL means that the simple dominant pole compensation
method used in the NPN regulator will not work in an LDO unless additional compensation is added. To
illustrate why this is true, the loop gain of a 5V/50 mA LDO regulator will be illustrated using these
assumptions:
At maximum load current, the load pole (PL) occurs at a frequency given by:
PL = 1/ (2π x RLOAD x COUT) = 1/ (2π x 100 x 10−5) = 160 Hz
(9)
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The internal compensation will be assumed to add a fixed pole (P1) at 1 kHz.
A 500 kHz power pole (which will be designated PPWR) is present due to the PNP power transistor and
driver.
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LDO Compensation Using ESR
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The DC gain is assumed to be 80 dB.
RL = 100Ω (which is the value at maximum load current)
COUT = 10 µF.
using the conditions stated above, a Bode plot (Figure 13) is drawn. It is immediately obvious that this
loop is not stable: the two poles PL and P1 will each contribute −90° of phase shift to reach −180° at the 0
dB frequency (which is about 40 kHz in this example).
To reduce the negative phase shift (and prevent oscillations), a zero must be added to the loop. A zero
can contribute as much as +90° of positive phase shift, which will cancel out the effects of one of the two
low frequency poles.
Figure 13. LDO Gain Plot Without Compensation
Nearly all monolithic LDO regulators require that this zero be added to the loop, and they derive it from a
characteristic that is inherent in the output capacitor: equivalent series resistance (usually referred to
ESR).
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LDO Compensation Using ESR
Equivalent series resistance (ESR) is a characteristic that is present in every capacitor. It can be modeled
electrically as a resistance that is placed in series with the capacitor (Figure 14).
Figure 14. Capacitor Showing ESR
The ESR of the output capacitor puts a zero in the loop gain which can be used to reduce excess
negative phase shift.
The frequency where the zero occurs is directly related to the value of the ESR and amount of output
capacitance:
FZERO = 1/(2π x COUT x ESR)
(10)
Using the example in the previous section (Bode plot shown in Figure 13), we will assume that the value
of COUT = 10 µF and the output capacitor ESR = 1Ω, which means a zero will occur at 16 kHz.
Figure 15 shows how this added zero will change the unstable plot into a stable one:
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Figure 15. ESR Zero Stabilizes LDO
The bandwidth of the loop is increased so that the 0 dB crossover frequency moves form 30 kHz to 100
kHz.
The zero adds a total of +81° positive phase shift at 100 kHz (the 0 dB frequency). This will reduce the
negative phase shift caused by the poles PL and P1.
Since the pole PPWR is located at 500 kHz, it adds only −11° of phase shift at 100 kHz.
Summing all poles and zeros, the total phase shift at 0 dB is now −110°. This corresponds to a phase
margin of +70°, which is extremely stable.
This illustrates how an output capacitor with the correct value of ESR can generate a zero that stabilizes
an LDO.
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ESR and Stability
Virtually all LDO regulators require that the ESR of the output capacitor be within a set range to assure
regulator stability.
The LDO manufacturer provides a set of curves which define the boundaries of the stable region, plotted
as a function of load current (Figure 16).
Figure 16. ESR Range for Typical LDO
To explain why these boundaries exist, the effects of low and high ESR on phase margin will be illustrated
using the example previously developed.
11.1 High ESR
Using the examples developed in the previous sections, we will change the conditions and assume the
ESR of the 10 µF output capacitor is increased to 20Ω. This will decrease the frequency of the zero to 800
Hz (Figure 17). Reducing the frequency of the zero causes the loop bandwidth to increase, moving the 0
dB crossover frequency from 100 kHz to 2 MHz.
This increase bandwidth means that the pole PPWR occurs at a gain value of +20 dB (compared to −10 dB
in Figure 14).
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Output Capacitor Selection
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Analyzing the plot (Figure 17) for phase margin, it can be assumed that the zero cancels out either P1 or
PL. This means the loop has a two-pole response with the low frequency pole contributing −90° of phase
shift and the high frequency pole PPWR contributing about −76° of phase shift.
Figure 17. High ESR Causes Unstable Loop
Although this appears to leave a phase margin of 14° (which might be stable), bench test data shows that
ESR values > 10 Ω usually cause instability because of phase shifts contributed by other high-frequency
poles which are not shown in this simplified model.
11.2 Low ESR
An output capacitor with a very low ESR value can cause oscillations for a different reason.
Continuing the example developed in the previous section, we will now reduce the ESR of the 10 µF
output capacitor to 50 mΩ, increasing the frequency of the zero to 320 kHz (Figure 18).
Figure 18. Low ESR Causes Unstable Loop
When the plot is analyzed for phase margin, no calculations are required to see that it is unstable.
The −90° phase shift from each of the two poles P1 and PL will produce a total phase shift of −180° at the
0 dB frequency.
For this system to be stable, a zero is needed that would provide positive phase shift before the 0 dB
point. However, since the zero is at 320 kHz, it's too far out to do any good (and is cancelled out by PPWR).
12
Output Capacitor Selection
Since the output capacitor is the user's tool for compensating a monolithic LDO regulator, it must be
selected very carefully. Most cases of oscillations in LDO applications are caused by the ESR of the
output capacitor being too high or too low.
When selecting an output capacitor for an LDO, a solid tantalum capacitor is usually the best choice
(except for parts specifically designed for ceramic capacitors like the LP2985). Tests performed on an
AVX 4.7 µF Tantalum showed an ESR of 1.3Ω @ 25°C, a value that is almost perfectly centered in the
stable region (Figure 16).
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Also very important, the ESR of the AVX capacitor varied less than 2:1 over the temperature range of
−40°C to +125°C. Aluminium electrolytic capacitors are notorious for exhibiting an exponential increase in
ESR at cold temperatures, and are not suitable for use as an LDO output capacitor.
It must be noted that large (≥ 1 µF) ceramic capacitors typically have very low ESR values (< 20 mΩ), and
will cause most LDO regulators to oscillate if connected directly to the output (except the LP2985). A
ceramic capacitor can be used if some external resistance is added in series with it to increase the
effective ESR. Large value ceramics also have a poor tempco (typically Z5U) which means the
capacitance will drop in half as the temperature is increased or decreased to the operating limits.
13
Quasi-LDO Compensation
When evaluating the quasi-LDO regulator (Figure 3) for stability and compensation, it has some of the
electrical characteristics of both the LDO and NPN regulator. Since the quasi-LDO uses an NPN pass
device, it is in the common-collector configuration which means it's output device node (emitter) looks like
a relatively low impedance.
However, because of the base of the NPN is being driven from a high-impedance PNP current source, the
regulator output impedance of a quasi-LDO is not as low as the NPN regulator with an NPN Darlington
pass device (but is much lower than a true LDO which drives the regulator output off the collector of a
PNP).
This means that the troublesome power pole of a quasi-LDO is at a lower frequency than the NPN
regulator, so some compensation (output capacitance) is required to make a quasi-LDO stable. Of course,
the pole is at a much higher frequency than the LDO, so the quasi-LDO requires less capacitance and the
ESR is not as critical.
For example, the LM1085 quasi-LDO rated for 3A of load current requires only 10 µF of Tantalum output
capacitance to assure complete stability over all line and load conditions. No ESR graphs are given, since
the value of ESR is not critical as it is in an LDO.
14
Low-ESR Specific LDOs
Texas Instruments does have LDO regulators like the LP2985 and LP2989 which are specifically designed
to work with extremely low ESR capacitors like surface-mount ceramics. This type of capacitor can have
ESR values as low as 5-10 mΩ, which will cause most typical LDO regulators to oscillate (as
demonstrated in Figure 18.
To make the LP2985 stable with such low ESR values, an internal zero is built in which takes the place of
the ESR zero previously provided by the Tantalum output capacitor. The effect of this is to shift the stable
ESR range downward. A typical LDO with no added internal zero might be stable ESR range from about
100 mΩ to 5 Ω (well suited for Tantalums but not ceramics). The stable range for the LP2985 extends
down to 3 mΩ, and has an upper limit of about 500 mΩ so it can be used with ceramics.
The reason the upper limit is moved down can be understood by referring to Figure 15. As previously
stated, the zero is now built into the LDO, so the ESR zero resulting from the output capacitor must stay at
a high enough frequency that it does not cause the bandwidth to get too wide where high frequency poles
would add enough phase shift to produce oscillations.
15
The FET Advantage
An LDO regulator can be built using a P-FET as the pass transistor (see Figure 19).
Figure 19. P-FET LDO Regulator
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The FET Advantage
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To see why using a P-FET LDO would be advantageous, it should be noted that all of the base current
required by the power transistor in a PNP LDO (Figure 2) flows out of the ground pin and back to the
negative input voltage return. Therefore, this base drive current is drawn from the input supply but does
not drive the load, so it generates wasted power that must be dissipated within the LDO regulator:
PWR (Base Drive) = VIN X IBASE
(11)
The amount of base current required to drive the PNP is equal to the load current divided by the beta
(gain) of the PNP, and beta may be as low as 15 - 20 (at rated load current) in some PNP LDO regulators.
The wasted power generated by this base drive current is very undesirable (especially in battery-powered
applications). Using a P-FET solves this problem, since the Gate drive is very small.
Another advantage of the P-FET LDO is that the dropout voltage can be made very small by adjusting the
ON-resistance of the FET. For monolithic regulators, FET power transistors typically will give a lower ONresistance per unit area than bipolar ONP devices. This allows making higher current regulators in smaller
packages.
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issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and
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supplied at the time of order acknowledgment.
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to support this warranty. Except where mandated by applicable law, testing of all parameters of each component is not necessarily
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Products
Applications
Audio
www.ti.com/audio
Automotive and Transportation
www.ti.com/automotive
Amplifiers
amplifier.ti.com
Communications and Telecom
www.ti.com/communications
Data Converters
dataconverter.ti.com
Computers and Peripherals
www.ti.com/computers
DLP® Products
www.dlp.com
Consumer Electronics
www.ti.com/consumer-apps
DSP
dsp.ti.com
Energy and Lighting
www.ti.com/energy
Clocks and Timers
www.ti.com/clocks
Industrial
www.ti.com/industrial
Interface
interface.ti.com
Medical
www.ti.com/medical
Logic
logic.ti.com
Security
www.ti.com/security
Power Mgmt
power.ti.com
Space, Avionics and Defense
www.ti.com/space-avionics-defense
Microcontrollers
microcontroller.ti.com
Video and Imaging
www.ti.com/video
RFID
www.ti-rfid.com
OMAP Applications Processors
www.ti.com/omap
TI E2E Community
e2e.ti.com
Wireless Connectivity
www.ti.com/wirelessconnectivity
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Copyright © 2013, Texas Instruments Incorporated
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