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Texas Instruments AN-991 Line Driving and System Design Application notes
AN-991 Line Driving and System Design
Literature Number: SNLA043
National Semiconductor
Application Note 991
April 1995
Successful high-speed system design is dependent on careful system timing design and good board layout. The pitfalls
are many and varied, and this section addresses some of
those problem areas and simplifies the design requirements. All systems must interconnect signals either by short
lines on printed circuit board, long lines on a backplane,
twisted pair cables, or coaxial cables, etc. At high frequency, all of these mediums must be treated as transmission
lines. Two properties of transmission lines, characteristic
impedance (ZO) and propagation delay (tPD), are of concern. Transmission lines store energy, the magnitude of
which is dependent on line length, impedance, applied voltage and source impedance. This stored energy must be dissipated by the terminating device and may couple to other
circuits by crosstalk. The effects of termination on line reflection and crosstalk are discussed, as well as good board
layout practices.
e ZO e
where LO e inductance per unit length, and CO e capacitance per unit length. ZO is in ohms, LO in Henries, and CO
in Farads.
The interactions between wiring and circuitry in high-speed
systems are more easily determined by treating the interconnections as transmission lines. A brief review of basic
concepts is presented and simplified methods of analysis
are used to examine situations commonly encountered in
digital systems. Since the principles and methods apply to
any type of logic circuit, normalized pulse amplitudes are
used in sample waveforms and calculations.
For the great majority of interconnections in digital systems,
resistance of the conductors is much less than the input and
output resistance of the circuits. Similarly, the insulating materials have very good dielectric properties. These circumstances allow such factors as attenuation, phase distortion
and bandwidth limitations to be ignored. With these simplifications, interconnections can be dealt with in terms of characteristic impedance and propagation delay.
The two conductors that interconnect a pair of circuits have
distributed series inductance and distributed capacitance
between them, and thus constitute a transmission line. For
any length in which these distributed parameters are constant, the pair of conductors have a characteristic impedance ZO. Whereas quiescent conditions on the line are determined by the circuits and terminations, ZO is the ratio of
transient voltage to transient current passing by a point on
the line when a signal change or other electrical disturbance
occurs. The relationship between transient voltage, transient current, characteristic impedance, and the distributed
parameters is expressed as follows:
Propagation velocity (n) and its reciprocal, delay per unit
length (e), can also be expressed in terms of LO and CO. A
consistent set of units is nanoseconds, microHenries and
picoFarads, with a common unit of length.
e e 0LOCO
Equations 9-1 and 9-2 provide a convenient means of determining the LO and CO of a line when delay, length and impedance are known. For a length I and delay T, e is the ratio
T/I. To determine LO and CO, combine Equations 9-1 and
LO e eZO
CO e
More formal treatments of transmission line characteristics,
including loss effects, are available from many sources.
A transmission line with a terminating resistor is shown in
Figure 9-1 . As indicated, a positive step function voltage
travels from left to right. To keep track of reflection polarities, it is convenient to consider the lower conductor as the
voltage reference and to think in terms of current flow in the
top conductor only. The generator is assumed to have zero
internal impedance. The initial current I1 is determined by V1
and ZO.
TL/F/12419 – 1
If the terminating resistor matches the line impedance, the
ratio of voltage to current traveling along the line is matched
by the ratio of voltage to current which must, by Ohm’s law,
FASTÉ and TRI-STATEÉ are registered trademarks of National Semiconductor Corporation.
C1995 National Semiconductor Corporation
Line Driving and System Design
Line Driving and System
RRD-B30M105/Printed in U. S. A.
The foregoing has the same form as a simple voltage divider involving a generator V1 with internal impedance ZO driving a load RT, except that the amplitude of VT is doubled.
always prevail at RT. From the viewpoint of the voltage step
generator, no adjustment of output current is ever required;
the situation is as though the transmission line never existed
and RT had been connected directly across the terminals of
the generator.
From the RT viewpoint, the only thing the line did was delay
the arrival of the voltage step by the amount of time T.
When RT is not equal to ZO, the initial current starting down
the line is still determined by V1 and ZO but the final steady
state current, after all reflections have died out, is determined by V1 and RT (ohmic resistance of the line is assumed to be negligible). The ratio of voltage to current in the
initial wave is not equal to the ratio of voltage to current
demanded by RT. Therefore, at the instant the initial wave
arrives at RT, another voltage and current wave must be
generated so that Ohm’s law is satisfied at the line-load
interface. This reflected wave, indicated by Vr and Ir in Figure 9-1 , starts to return toward the generator. Applying Kirchoff’s laws to the end of the line at the instant the initial
wave arrives results in the following:
I1 a Ir e IT e Current into RT
The arrow indicating the direction of Vr, in Figure 9-1 correctly indicates the Vr direction of travel, but the direction of
Ir flow depends on the Vr polarity. If Vr is positive, Ir flows
toward the generator, opposing I1. This relationship between the polarity of Vr and the direction of Ir can be deduced by noting in Equation 9-7 that if Vr is positive it is
because RT is greater than ZO. In turn, this means that the
initial current Ir is larger than the final quiescent current,
dictated by V1 and RT. Hence Ir must oppose I1 to reduce
the line current to the final quiescent value. Similar reasoning shows that if Vr is negative, Ir flows in the same direction
as I1.
It is sometimes easier to determine the effect of Vr on line
conditions by thinking of it as an independent voltage generator in series with RT. With this concept, the direction of Ir is
immediately apparent; its magnitude, however, is the ratio of
Vr to ZO i.e., RT is already accounted for in the magnitude of
Vr. The relationships between incident and reflected signals
are represented in Figure 9-2 for both cases of mismatch
between RT and ZO.
Since only one voltage can exist at the end of the line at this
instant of time, the following is true:
V1 a Vr e VT
V a Vr
e 1
IT e
I1 e
and Ir e b
with the minus sign indicating that Vr is moving toward the
Combining the foregoing relationships algebraically and
solving for Vr yields a simplified expression in terms of V1,
ZO and RT.
V a Vr
b r e 1
e 1a r
Vr e V1
e Vr
TL/F/12419 – 2
a. Incident Wave
TL/F/12419 – 3
b. Reflected Wave for RT l ZO
e rL V1
The term in parentheses is called the coefficient of reflection (rL). With RT ranging between zero (shorted line) and
infinity (open line), the coefficient ranges between b1 and
a 1 respectively. The subscript L indicates that rL refers to
the coefficient at the load end of the line.
Equation 9-7 expresses the amount of voltage sent back
down the line, and since
VT e V1 a Vr
TL/F/12419 – 4
c. Reflected Wave for RT k ZO
FIGURE 9-2. Reflections for RT
then VT e V1 (1 a rL)
VT can also be determined from an expression which does
not require the preliminary step of calculating rL. Manipulating (1 a rL) results in
1 a rL e 1 a
Substituting in Equation 9-8 gives
VT e 2
The incident wave is shown in Figure 9-2a , before it has
reached the end of the line. In Figure 9-2b , a positive Vr is
returning to the generator. To the left of Vr the current is still
I1, flowing to the right, while to the right of Vr the net current
in the line is the difference between I1 and Ir. In Figure 9-2c ,
the reflection coefficient is negative, producing a negative
Vr. This, in turn, causes an increase in the amount of current
flowing to the right behind the Vr wave.
If the source impedance matches the line impedance, a reflected voltage arriving at the source is not reflected back
toward the load end. Voltage and current on the line are
stable with the following values.
VT e V1 a Vr and IT e I1 b Ir
If neither source impedance nor terminating impedance
matches ZO, multiple reflections occur; the voltage at each
end of the line comes closer to the final steady state value
with each succeeding reflection. An example of a line mismatched on both ends is shown in Figure 9-3 . The source is
a step function of VCC e 5.0V amplitude occurring at time
t0. The initial value of V1 starting down the line is 2.4V due
to the voltage divider action of ZO and RS. The time scale in
the photograph shows that the line delay is approximately
6 ns. Since neither end of the line is terminated in its characteristic impedance, multiple reflections occur.
When a reflected voltage arrives back at the source (generator), the reflection coefficient at the source determines the
response to Vr. The coefficient of reflection at the source is
governed by ZO and the source resistance RS.
R S b ZO
rS e
TL/F/12419 – 5
VO e VCC b 2 # VBE e 3.6V
(25 b 50)X
e b 0.33
rS e
(25 a 50)X
Initially: V1 e
# VO
(3.6V) e 2.4
V1 e
(50 a 25)X
(300 b 50)X
e 0.71
rL e
(300 a 50)X
TL/F/12419 – 6
H e 20 ns/div
V e 1.0 V/div
FIGURE 9-3. Multiple Reflections Due to Mismatch at Load and Source
The amplitude and persistence of the ringing shown in Figure 9-3 become greater with increasing mismatch between
the line impedance and source and load impedances. Reducing RS (Figure 9-3) to 13X increases rS to b0.75, and
the effects are illustrated in Figure 9-4 . The initial value of
VT is 1.8V with a reflection of 0.9V from the open end. When
this reflection reaches the source, a reflection of (0.9) c
(b0.75V) starts back toward the open end. Thus, the second increment of voltage arriving at the open end is negative-going. In turn, a negative-going reflection of (0.9) c
(b0.75V) starts back toward the source. This negative increment is again multiplied by b0.75V at the source and
returned toward the open end. It can be deduced that the
difference in amplitude between the first two positive peaks
observed at the open end is:
VT b VÊ T e (1 a rL) V1 b (1 a rL) V1 r2Lr2S
e (1 a rL) V1 (1 b r2Lr2S).
The open-ended line in Figure 9-3 has a reflection coefficient of 0.71 and the successive reflections tend toward the
steady state conditions of zero line current and a line voltage equal to the source voltage. In contrast, a shorted line
has a reflection coefficient of b1 and successive reflections
must cause the line conditions to approach the steady state
conditions of zero voltage and a line current determined by
the source voltage and resistance.
Shorted line conditions are shown in Figure 9-6a with the
reflection coefficient at the source end of the line also negative. A negative coefficient at both ends of the line means
that any voltage approaching either end of the line is reflected in the opposite polarity. Figure 9-6b shows the response
to an input step-function with a duration much longer than
the line delay. The initial voltage starting down the line is
about a 2.4V, which is inverted at the shorted end and returned toward the source as b2.4V. Arriving back at the
source end of the line, this voltage is multiplied by (1 a rS),
causing a b1.61V net change in V1. Concurrently, a reflected voltage of a 0.80V (b2.4V times rS of b0.33) starts
back toward the shorted end of the line. The voltage at V1 is
reduced by 50% with each successive round trip of reflections, thus leading to the final condition of 0V on the line.
When the duration of the input pulse is less than the delay
of the line, the reflections observed at the source end of the
line constitute a train of negative pulses, as shown in
Figure 9-6c . The amplitude decreases by 50% with each
successive occurrence as it did in Figure 9-6b .
The factor (1 b r2Lr2S) is similar to the damping factor
associated with lumped constant circuitry. It expresses the
attenuation of successive positive or negative peaks of ringing.
In the presence of multiple reflections, keeping track of the
incremental waves on the line and the net voltage at the
ends becomes a bookkeeping chore. A convenient and systematic method of indicating the conditions which combines
magnitude, polarity and time utilizes a graphic construction
called a lattice diagram. A lattice diagram for the line conditions of Figure 9-3 is shown in Figure 9-5 .
FIGURE 9-4. Extended Ringing when RS
of Figure 9-3 is Reduced to 13X
The vertical lines symbolize discontinuity points, in this case
the ends of the line. A time scale is marked off on each line
in increments of 2T, starting at t0 for V1 and T for VT. The
diagonal lines indicate the incremental voltages traveling
between the ends of the line; solid lines are used for positive voltages and dashed lines for negative. It is helpful to
write the reflection and transmission multipliers r and (1 a
r) at each vertical line, and to tabulate the incremental and
net voltages in columns alongside the vertical lines. Both
the lattice diagram and the waveform photograph show that
V1 and VT asymptotically approach 3.4V, as they must with
a 3.4V source driving an open-ended line.
TL/F/12419 – 8
FIGURE 9-5. Lattice Diagram
for the Circuit of Figure 9-3
no further changes occur and the line voltage is equal to the
generator voltage. Because the initial signal on the line is
only half the normal signal swing, the loads must be connected at or near the end of line to avoid receiving a 2-step
input signal.
A TTL output driving a series-terminated line is limited in
fanout due to the IR drop associated with the collective IIL
drops of the inputs being driven. For most TTL families other than FASTÉ, series termination should not be considered
since either the input currents are so high (TTL, S, H) or the
input threshold is very low (LS). In either case the noise
margins are severely degraded to the point where the circuit
becomes unusable. In FAST, however, the IIL of 0.6 mA,
through a 25X resistor used as a series terminator, will reduce the low level noise margin by 15.0 mV for each standard FAST input driven.
Driving an open-ended line through a source resistance
equal to the line impedance is called series termination. It is
particularly useful when transmitting signals which originate
on a PC board and travel through the backplane to another
board with the attendant discontinuities, since reflections
coming back to the source are absorbed and ringing thereby
controlled. Figure 9-7 shows a 93X line driven from a 1V
generator through a source impedance of 93X. The photograph illustrates that the amplitude of the initial signal sent
down the line is only half of the generator voltage, while the
voltage at the open end of the line is doubled to full amplitude (1 a rL e 2). The reflected voltage arriving back at
the source raises V1 to the full amplitude of the generator
signal. Since the reflection coefficient at the source is zero,
TL/F/12419 – 9
a. Reflection Coefficients for Shorted Line
TL/F/12419 – 10
b. Input Pulse Duration n Line Delay
TL/F/12419 – 11
c. Input Pulse Duration k Line Delay
FIGURE 9-6. Reflections of Long and Short Pulses on a Shorted Line
TL/F/12419 – 15
a. Series-Terminated Line with Load Capacitance
TL/F/12419 – 16
b. Output Rise Time Increase with
Increasing Load Capacitance
FIGURE 9-7. Series-Terminated Line Waveforms
FIGURE 9-8. TTL Element Driving
a Series-Terminated Line
TL/F/12419 – 17
c. Extra Delay DT Due to Rise Time Increase
FIGURE 9-9. Extra Delay with Termination Capacitance
The increase in propagation delay can be calculated by using a ramp approximation for the incident voltage and characterizing the circuit as a fixed impedance in series with the
load capacitance, as shown in Figure 9-10 . One general
solution serves both series and parallel termination cases
by using an impedance ZÊ and a time constant, u, defined in
Figure 9-10a and 9-10b .
Calculated and observed increases in delay time to the 50%
point show close agreement when u is less than half the
ramp time. At large ratios of u/a (where a e ramp time),
Designers should consider the effect of the load capacitance at the end of the line when using series termination.
Figure 9-9 shows how the output waveform changes with
increasing load capacitance. Figure 9-9b shows the effect
of load capacitances of 0, 12, 24, 48 pF. With no load, the
delay between the 50% points of the input and output is just
the line delay T. A capacitive load at the end of the line
causes an extra delay DT due to the increase in rise time of
the output signal. The midpoint of the signal swing is a good
approximation of the FAST threshold since VOL e 0.5V and
VOH e 2.7V and the actual input switching threshold of
FAST is 1.5V at 25§ C.
measured delays exceed calculated values by approximately 7%. Figure 9-11 , based on measured values, shows the
increase in delay to the 50% point as a function of the ZÊ C
time constant, both normalized to the 10% to 90% rise time
of the input signal. As an example of using the graph, consider a 100X series-terminated line with 30 pF load capacitance at the end of the line. The 3 ns rise time assumed is
typical of FAST in an actual line driving application. From
Figure 9-10a , ZÊ is equal to 100X; the ratio
is 1. From
the graph, the ratio DT/tr is 0.8. Thus the increase in the
delay to the 50% point of the output waveform is 0.8 tr, or
2.4 ns, which is then added to the no-load line delay T to
determine the total delay.
Had the 100X line in the foregoing example been parallel
rather than series terminated at the end of the line, ZÊ would
be 50X. The added delay would be only 1.35 ns with the
same 30 pF loading at the end. The added delay would be
only 0.75 ns if the line were 50X and parallel-terminated.
The various trade-offs involving type of termination, line impedance, and loading are important considerations for critical delay paths.
TL/F/12419 – 20
FIGURE 9-11. Increase in 50% Point Delay
Due to Capacitive Loading at the
End of the Line, Normalized to tr
When capacitive loads such as TTL inputs are connected
along a transmission line, each one causes a reflection with
a polarity opposite to that of the incident wave. Reflections
from two adjacent loads tend to overlap if the time required
for the incident wave to travel from one load to the next is
equal to or less than the signal rise time. Figure 9-12a illustrates an arrangement for observing the effects of capacitive loading, while Figure 9-12b shows an incident wave followed by reflections from two capacitive loads. The two capacitors causing the reflections are separated by a distance
requiring a travel time of 1 ns. The two reflections return to
the source 2 ns a part, since it takes 1 ns longer for the
incident wave to reach the second capacitor and an additional 1 ns for the second reflection to travel back to the
source. In the upper trace of Figure 9-12b , the input signal
rise time is 1 ns and there are two distinct reflections, although the trailing edge of the first overlaps the leading
edge of the second. The input rise time is longer in the
middle trace, causing a greater overlap. In the lower trace,
the 2 ns input rise time causes the two reflections to merge
and appear as a single reflection which is relatively constant
(at & b 10%) for half its duration. This is about the same
reflection that would occur if the 93X line had a middle section with an impedance reduced to 75X.
TL/F/12419 – 18
a.Thevenin Equivalent for Series-Terminated Case
TL/F/12419 – 19
b. Thevenin Equivalent for
Parallel-Terminated Case
FIGURE 9-10. Determining the Effect
of End-of-Line Capacitance
waveform of Figure 9-12c . Note that the final (steady state)
value of the line voltage is reduced by about the same
amount as that caused by the capacitive reflections. In the
lower trace of Figure 9-12c the source resistance RS is reduced from 93X to 75X, restoring both the initial and final
line voltage values to the same amplitude as the final value
in the upper trace. From the standpoint of providing a desired signal voltage on the line and impedance matching at
either end, the effect of distributed capacitive loading can
be treated as a reduction in line impedance.
With a number of capacitors distributed all along the line of
Figure 9-12a , the combined reflections modify the observed
input waveform as shown in the top trace of Figure 9-12c .
The reflections persist for a time equal to the 2-way line
delay (15 ns), after which the line voltage attains its final
value. The waveform suggests a line terminated with a resistance greater than its characteristic impedance (RT l
ZO). This analogy is strengthened by observing the effect of
reducing RT from 93X to 75X which leads to the middle
TL/F/12419 – 21
a. Arrangement for Observing Capacitive Loading Effects
TL/F/12419 – 22
b. Capacitive Reflections Merging as Rise Time Increases
TL/F/12419 – 23
c. Matching the Altered Impedance of a Capacitively Loaded Line
FIGURE 9-12. Capacitive Reflections and Effects on Line Characteristics
The reduced line impedance can be calculated by considering the load capacitance CL as an increase in the intrinsic
line capacitance CO along that portion of the line where the
loads are connected. Denoting this length of line as I, the
distributed value CD of the load capacitance is as follows:
CD e
CD is then added to CO in Equation 9-1 to determine the
reduced line impedance ZO.
ZÊ O e
O a CD
ZÊ O a
CO 1 a
TL/F/12419 – 24
In the example of Figure 9-12c , the total load capacitance
(ICO) is 60 pF. Note that the ratio CD/CO is the same as
CL/ICO. The calculated value of the reduced impedance is
ZÊ O e
FIGURE 9-13. Capacitive Loading Effects
on Line Delay and Impedance
Worst case reflections from a capacitively loaded section of
transmission line can be accurately predicted by using the
modified impedance of Equation 9-9. When a signal originates on an unloaded section of line, the effective reflection
coefficient is as follows:
e 75X
This correlates with the results observed in Figure 9-12c
when RT and RS are reduced to 75X.
The distributed load capacitance also increases the line delay, which can be calculated from Equation 9-2.
Reflections occur not only from mismatched load and
source impedances but also from changes in line impedance. These changes could be caused by bends in coaxial
cable, unshielded twisted-pair in contact with metal, or mismatch between PC board traces and backplane wiring. With
the coax or twisted-pair, line impedance changes run about
5% to 10% and reflections are usually no problem since the
percent reflection is roughly half the percent change in impedance. However, between PC board and backplane wiring, the mismatch can be 2 or 3 to 1. This is illustrated in
Figure 9-14 and analyzed in the lattice diagram of Figure 915 . Line 1 is driven in the series-terminated mode so that
reflections coming back to the source are absorbed.
The reflection and transmission at the point where impedances differ are determined by treating the downstream line
as though it were a terminating resistor. For the example of
Figure 9-14 , the reflection coefficient at the intersection of
lines 1 and 2 for a signal traveling to the right is as follows:
eÊ e 0LO (CO a CD) e 0LOCO
The line used in the example of Figure 9-12c has an intrinsic
delay of 6 ns and a loaded delay of 7.5 ns which checks
with Equation 9-15.
IeÊ e Ie 01.55 e 601.55 e 7.5 ns
Equation 9-15 can be used to predict the delay for a given
line and load. The ratio CD/CO (hence the loading effect)
can be minimized for a given loading by using a line with a
high intrinsic capacitance CO.
A plot of ZÊ and eÊ for a 50X line as a function of CD is
shown in Figure 9-13 . This figure illustrates that relatively
modest amounts of load capacitance will add appreciably to
the propagation delay of a line. In addition, the characteristic impedance is reduced significantly.
r12 e
Z2 b Z1
93 b 50
e a 0.3
Z2 a Z1
TL/F/12419 – 25
TL/F/12419 – 26
FIGURE 9-14. Reflections from Mismatched Lines
Thus the signal reflected back toward the source and the
signal continuing along line 2 are, respectively, as follows:
V1r e r12 V1 e a 0.3 V1
Thus, Equation 9-22 is the general expression for the initial
step of output voltage for three lines when the input is series-terminated and the output is open-ended. Note that the
reflection coefficients at the intersections of lines 1 and 2
and lines 2 and 3 in Figure 9-15 have reversed signs for
signals traveling to the left. Thus the voltage reflected from
the open output and the signal reflecting back and forth on
line 2 both contribute additional increments of output voltage in the same polarity as VO. Lines 2 and 3 have the
same delay time; therefore, the two aforementioned increments arrive at the output simultaneously at time 5T on the
lattice diagram (Figure 9-15) .
In the general case of series lines with different delay times,
the vertical lines on the lattice diagram should be spaced
apart in the ratio of the respective delays. Figure 9-16
shows this for a hypothetical case with delay ratios 1:2:3.
For a sequence of transmission lines with the highest impedance line in the middle, at least three output voltage
increments with the same polarity as VO occur before one
can occur of opposite polarity. On the other hand, if the
middle line has the lowest impedance, the polarity of the
second increment of output voltage is the opposite of VO.
The third increment of output voltage has the opposite polarity, for the time delay ratios of Figure 9-16 .
When transmitting logic signals, it is important that the initial
step of line output voltage passes through the threshold
region of the receiving circuit, and that the next two increments of output voltage augment the initial step. Thus in a
series-terminated sequence of three mismatched lines, the
middle line should have the highest impedance.
V2 e (1 a r12) V1 e a 1.3 V1
At the intersection of lines 2 and 3, the reflection coefficient
for signals traveling to the right is determined by treating Z3
as a terminating resistor.
Z b Z2
39 b 93
e b 0.41
r23 e 3
Z3 a Z2
When V2 arrives at this point, the reflected and transmitted
signals are as follows:
V2r e r23 V2 e b0.41 V2
e ( b 0.41) (1.3) V1
e b 0.53 V1
V3 e (1 a r23) V2 e 0.59 V2
e (0.59) (1.3) V1
e 0.77 V1
Voltage V3 is doubled in magnitude when it arrives at the
open-ended output, since rL is a 1. This effectively cancels
the voltage divider action between RS and Z1.
V4 e (1 a rL) (1 a r23) V2
e (1 a rL) (1 a r23) (1 a r12) V1
e (1 a rL) (1 a r23) (1 a r12) O
V4 e (1 a r23) (1 a r12) VO
TL/F/12419 – 27
FIGURE 9-15. Lattice Diagram for the Circuit of Figure 9-14
VT rises to 0.5V, i.e., 1 a rL e 0.5. The negative reflection
arrives back at the source at time 2T, causing a net change
of b0.4V, i.e., (1 a rS) (b0.5) e b0.4.
The negative coefficient at the source changes the polarity
of the other 0.1V of the reflection and returns it to the end of
the line, causing VT to go positive by another 50 mV at time
3T. The remaining 50 mV is inverted and reflected back to
the source, where its effect is barely distinguishable as a
small negative change at time 4T.
In Figure 9-17c , the input rise time (0% to 100%) is increased to such an extent that the input ramp ends just as
the negative reflection arrives back at the source end. Thus
the input rise time is equal to 2T.
The input rise time is increased to 4T in Figure 9-17d , with
the negative reflection causing a noticeable change in input
slope at about its midpoint. This change in slope is more
visible in the double exposure photo of Figure 9-17e , which
shows V1 (tr still set for 4T) with and without the negative
reflection. The reflection was eliminated by terminating the
line in its characteristic impedance.
The net input voltage at any particular time is determined by
adding the reflection to the otherwise unaffected input. It
must be remembered that the reflection arriving back at the
input at a given time is proportional to the input voltage at a
time 2T earlier. The value of V1 in Figure 9-17d can be
calculated by starting with the 1V input ramp.
TL/F/12419 – 28
FIGURE 9-16. Lattice Diagram for
Three Lines with Delay Ratios 1:2:3
When the 2-way line delay is less than the rise time of the
input wave, any reflections generated at the end of the line
are returned to the source before the input transition is completed. Assuming that the generator has a finite source resistance, the reflected wave adds algebraically to the input
wave while it is still in transition, thereby changing the shape
of the input. This effect is illustrated in Figure 9-17 , which
shows input and output voltages for several comparative
values of rise time and line delay.
In Figure 9-17b where the rise time is much shorter than the
line delay, V1 rises to an initial value of 1V. At time T later,
# t for 0 s t s 4T
e 1V for t t 4T
V1 e
a. Test Arrangement for Rise Time Analysis
TL/F/12419 – 30
b. Line Voltages for tr k k T
TL/F/12419 – 32
d. Line Voltages for tr e 4T
c. Line Voltages for tr e 2T
TL/F/12419 – 33
e. Input Voltage With and Without Reflection
FIGURE 9-17. Line Voltages for Various Ratios of Rise Time to Line Delay
The voltage at the output end of the line is expressed in a
similar manner.
The reflection from the end of the line is
rL (t b 2T)
Vr e
for 0 k t k T
the portion of the reflection that appears at the input is
(1 a rS) rL (t b 2T)
VÊ r e
the net value of the input voltage is the sum.
(1 a rS) rL (t b 2T)
VÊ 1 e a
The peak value of the input voltage in Figure 9-17d is determined by substituting values and letting t equal 4T.
e (1 a rL) V1(t b T)
e (1 a rL) V1(t b T) a
(1 a rL) rSrL V1(t b 3T)
e (1 a rL) V1(t b T) a
(1 a rL) rSrL V1(t b 3T) a
(1 a rL) rS2rL2 V1(t b 5T)
for T k t k 3T
for 3T k t k 5T
(0.8) (b0.5) (4T b 2T)
e 1 b 0.04 (0.5) e 0.8V
After this peak point, the input ramp is no longer increasing
but the reflection is still arriving. Hence the net value of the
input voltage decreases. In this example, the later reflections are too small to be detected and the input voltage is
thus stable after time 6T. For the general case of repeated
reflections, the net voltage V1(t) seen at the driven end of
the line can be expressed as follows, where the signal
caused by the generator is V1(t):
VÊ 1(t) e V1(t)
for 0 k t k 2T
VÊ 1 e
VÊ 1(t) e V1(t) a (1 a rS) rL V1(t
for 2T k t k 4T
e 0
for 5T k t k 7T, etc.
Multiple reflections occur on a transmission line when neither the signal source impedance nor the termination (load)
impedance matches the line impedance. When the source
reflection coefficient rS and the load reflection coefficient
rL are of opposite polarity, the reflections alternate in polarity. This causes the signal voltage to oscillate about the final
steady state value, commonly recognized as ringing.
When the signal rise time is long compared to the line delay,
the signal shape is distorted because the individual reflections overlap in time. The basic relationships among rise
time, line delay, overshoot and undershoot are shown in a
simplified diagram, Figure 9-18 . The incident wave is a ramp
of amplitude B and rise duration A. The reflection coefficient
at the open-ended line output is a 1 and the source reflection coefficient is assumed to be b0.8, i.e., RO e ZO/9.
Figure 9-18b shows the individual reflections treated separately. Rise time A is assumed to be three times the line
delay T. The time scale reference is the line output and the
first increment of output voltage VO rises to 2B in the time
interval A. Simultaneously, a positive reflection (not shown)
of amplitude B is generated and travels to the source,
whereupon it is multiplied by b0.8 and returns toward the
end of the line. This negative-going ramp starts at time 2T
(twice the line delay) and doubles to b1.6B at time 2T a A.
b 2T)
VÊ 1(t) e V1(t) a (1 a rS) rL V1(t b 2T) a
(1 a rS) rSrL2 (V1(t b 4T)
for 4T k t k 6T
VÊ 1(t) e V1(t) a (1 a rS) rL V1(t b 2T) a
(1 a rS) rSrL2 V1(t b 4T) a
(1 a rS) rS2rL3 V1(t b 6T)
for 6T k t k 8T, etc.
TL/F/12419 – 34
a. Ramp Generator Driving Open-Ended Line
TL/F/12419 – 35
b. Increments of Output Voltage Treated Individually
TL/F/12419 – 36
c. Net Output Signal Determined by Superposition
FIGURE 9-18. Basic Relationships Involved in Ringing
limits load placement to the end of the line. TTL input capacitance increases the rise time at the end of the line, thus
increasing the rise time at the end of the line, thus increasing the effective delay. With parallel termiantion, i.e., at the
end of the line, loads can be distributed along the line. TTL
input capacitance modifies the line characteristics and
should be taken into account when determining line delay.
The negative-going increment also generates a reflection of
amplitude b0.8B which makes the round trip to the source
and back, appearing at time 4T as a positive ramp rising to
a 1.28B at time 4T a A. The process of reflection and rereflection continues, and each successive increment changes in polarity and has an amplitude of 80% of the preceding
In Figure 9-18c , the output increments are added algebraically by superposition. The starting point of each increment
is shifted upward to a voltage value equal to the algebraic
sum of the quiescent levels of all the preceding increments
(i.e., 0, 2B, 0.4B, 1.68B, etc.). For time intervals when two
ramps occur simultaneously, the two linear functions add to
produce a third ramp that prevails during the overlap time of
the two increments.
It is apparent from the geometric relationships, that if the
ramp time A is less than twice the line delay, the first output
increment has time to rise to the full 2B amplitude and the
second increment reduces the net output voltage to 0.4B.
Conversely, if the line delay is very short compared to the
ramp time, the excursions about the final value VG are
All interconnects, such as coaxial cable, defined impedance
transmission lines and feeders, can be considered as transmission lines, whereas printed circuit traces and hook-up
wire tend to be ignored as transmission lines. With any highspeed logic family, all interconnects should be considered
as transmission lines, and evaluated as such to see if termination is required. Of the many properties of transmission
lines, two are of major interest to us: ZO (the effective equivalent resistive value that causes zero reflection) and tPD
(propagation delay down the transmission line). Both of
these parameters are geometry dependent. Here are some
common configurations:
h e dielectric thickness
Figure 9-18c shows that the peak of each excursion is
reached when the earlier of the two constituent’s ramps
reaches its maximum value, with the result that the first
peak occurs at time A. This is because the earlier ramp has
a greater slope (absolute value) than the one that follows.
Actual waveforms such as produced by TTL do not have a
constant slope and do not start and stop as abruptly as the
ramp used in the example of Figure 9-18 . Predicting the
time at which the peaks of overshoot and undershoot occur
is not as simple as with ramp excitation. A more rigorous
treatment is required, including an expression for the driving
waveform which closely simulates its actual shape. In the
general case, a peak occurs when the sum of the slopes of
the individual signal increment is zero.
fr e
trace thickness
trace length
dielectric thickness between ground planes
trace width
dielectric constant
TL/F/12419 – 37
FIGURE 9-19. Micro Stripline
ZO e
The foregoing discussions are by no means an exhaustive
treatment of transmission line characteristics. Rather, they
are intended to focus attention on the general methods
used to determine the interactions between high-speed logic circuits and their interconnections. Considering an interconnection in terms of distributed rather than lumped inductance and capacitance leads to the line impedance concept,
i.e., mismatch between this characteristic impedance and
the terminations causes reflections and ringing.
Series termination provides a means of absorbing reflections when it is likely that discontinuities and/or line impedance changes will be encountered. A disadvantage is that
the incident wave is only one-half the signal swing, which
limits load placement to the end of the line. TTL input capacitance increases the rise time at the end of the line, thus
increasing the effective delay. With parallel termination, i.e.,
at the end of the line, loads can be distributed along the line.
TTL input capacitance modifies the line characteristics and
should be taken into account when determining line delay.
Series termination provides a means of absorbing reflections when it is likely that discontinuities and/or line impedance changes will be encountered. A disadvantage is that
the incident wave is only one-half the signal swing, which
0fr a 1.41
# 0.8 b c J X
tPD e 1.017 00.475 fr a 0.67 ns/ft.
TL/F/12419 – 38
FIGURE 9-20. Stripline
ZO e f ln
0 r
0.67 qb
0.8 a
tPD e 1.017 0fr ns/ft.
FIGURE 9-21. Side by Side
ZO e 120
#b cJ X
tPD e 1.017 00.475 fr a 0.67 ns/ft.
TL/F/12419 – 43
# J
ZO e f ln
0 r
FIGURE 9-24. Twisted Pair or Ribbon Cable
All of the above rely on the complex relationship
FIGURE 9-22. Flat Parallel Conductors
for b n h and h n c
ZO e 377 ln
RO a j 0LO
ZO e
GO a j 0 CO
and can be simplified to
ZO e
if we assume
GO j RO j 0
Note that ZO is real, not complex, appears resistive and is
not a function of length.
#bJ X
tPD e 1.017 00.475fr a 0.67 ns/ft.
a. Wire Over Ground Plane
ZO e f ln
0 r
# J
The inductance of PC trace can be determined by the formula
LO e 0.0051 Ln
# a J(
where a e
b. Coaxial Cable
ZO e f ln
0 r
FIGURE 9-23. Wiring
a 0.00127 mH/inch
For power and ground planes in a multilayer board, the capacitance of the plane can be calculated by the formula for
parallel plates separated by a dielectric:
# J
C e 0.2212
where A e surface area of one plate.
The above formula (E9-40) cannot be used to calculate PC
trace capacitance. This must either be measured or an appropriate value may be taken from the following curves.
The impedance of striplines and microstriplines can be
found quickly from the following curves. For characteristics
of cables, refer to manufacturers’ data.
TL/F/12419 – 47
FIGURE 9-28. Impedance of Striplines
TL/F/12419 – 44
TABLE 9-1. Relative Dielectric
Constants of Various Materials
FIGURE 9-25. Capacitance of Microstriplines
Polyethylene Foam
Cellular Polyethylene
Silicon Rubber
Polyvinylchloride (PVC)
Epoxy Resin
Epoxy Glass
All the above information on impedance and propagation
delays are for the circuit interconnect only. The actual impedance and propagation delays will differ from this by the
loading effects of gate input and output capacitances, and
by any connectors that may be in line. The effective impedance and propagation delay can be determined from the
following formula:
tPD e 0LO CO . . tPDÊ e tPD 1 a
where CL is the total of all additional loading.
The results of these formulas will frequently give effective
impedances of less than half ZO, and interconnect propagation delays greater than the driving device propagation delays, thus becoming the predominant delay.
TL/F/12419 – 45
FIGURE 9-26. Capacitance of Striplines
0 # J
0 # J
TL/F/12419 – 46
FIGURE 9-27. Impedance of Microstriplines
This consumes no DC current with outputs in either state. If
this is used on a TRI-STATE bus, then the quiescent voltage
on the line can be established at VCC or GND by a high
value pull up (down) resistor to the appropriate supply rail.
1. Unterminated
The maximum length for an unterminated line can be determined by
TL/F/12419 – 50
A. RT to VCC
fimax e
(For FAST, tr e 3 ns)
. . fimax e 10 inches for trace on G10 epoxy glass P.C.
The voltage wave propagated down the transmission line (V
step) is the full output drive of the deivce into ZOÊ . Reflections will not be a problem if fi s fimax. Lines longer than
fimax will be subject to ringing and reflections and will drive
the inputs and outputs below ground.
TL/F/12419 – 51
B. RT to GND
2. Series-Terminated
TL/F/12419 – 52
Series termination has limited use in TTL interconnect
schemes due to the voltage drop across RTS in the LOW
state, reducing noise margins at the receiver. Series termination is the ideal termination for highly capacitive memory
arrays whose DC loadings are minimal. RTS values of 10X
to 50X are normally found in these applications.
C. Thevenin Termination
RT e 2 ZOÊ
3. Parallel-Terminated
Four possibilities for parallel termination exist:
A. ZÊ O to VCC. This will consume current from VCC when
output is LOW;
B. ZÊ O to GND. This will consume current from VCC when
output is HIGH;
C. Thevenin equivalent termination. This will consume half
the current of A and B from the output stage, but will
have reduced noise margins, and consume current from
VCC with outputs HIGH or LOW. If used on a TRISTATEÉ bus, this will set the quiescent line voltage to
D. AC Termination. An RC termination to GND, R a XC e
ZOÊ , XC to be less than 2% of ZOÊ at
TL/F/12419 – 53
D. AC Termination to GND
Choose capacitor, CT, such
that: TRC l 3TD (Line Delay)
The buffer output effectively sees two 100X lines in parallel
and thus a 50X load. For this value of load impedance, the
buffer output will force an initial LOW-to-HIGH transition
from 0.2V to 2.7V in about 3 ns. This net charge of 2.5V into
a 50X load causes an output-HIGH current change of
50 mA.
If all eight outputs of an octal buffer switch simultaneously,
in this application the current demand on VCC would be
0.4A. Clearly, a nearby VCC bypass capacitor is needed to
accommodate this demand.
Figure 9-32 shows several schemes for power and ground
distribution on logic boards. Figure 9-32 is a cross-section,
with a, b, and c showing a 0.1 inch wide VCC bus and
ground on the opposite side. Figure 9-32d shows side-byside VCC and ground strips, each 0.04 inch wide. Figure
9-32e shows a four layer board with embedded power and
ground planes.
In Figure 9-32a , the dynamic impedance of VCC with respect
to ground is 50X, even though the VCC trace width is generous and there is a complete ground plane. In Figure 9-32b ,
the ground plane stops just below the edge of the VCC bus
and the dynamic impedance doubles to 100X. In Figure
9-32c , the ground bus is also 0.1 inch wide and runs along
under the VCC bus and exhibits a dynamic impedance of
about 68X. In Figure 9-32d , the trace widths and spacing
are such that the traces can run under a DIP, between two
rows of pins. The impedance of the power and ground
planes in Figure 9-32e is typically less than 2X.
These typical dynamic impedances point out why a sudden
current demand due to an IC output switching can cause a
momentary reduction in VCC, unless a bypass capacitor is
located near the IC.
TL/F/12419 – 55
Buffer Output Sees Net 50X Load.
50X Load Line on IOH b VOH Characteristic
Shows LOW-to-HIGH Step of Approx. 2.5V.
Figure 9-33 illustrates the sudden demand for current from
VCC when a buffer output forces a LOW-to-HIGH transition
into the midpoint of a data bus. The sketch shows a wireover-ground transmission line, but it could also be twisted
pair, flat cable or PC interconnect.
Worst-Case Octal Drain e 8 c 50 mA e 0.4A
TL/F/12419 – 54
A VCC bus with bypass capacitors connected periodically
along its length is shown in Figure 9-34 . Also shown is a
current source representing the current demand of the buffer in the preceding application.
The equations illustrate an approximation method of estimating the size of a bypass capacitor based on the current
demand, the drop in VCC that can be tolerated and the
length of time that the capacitor must supply the charge.
While the current demand is known, the other two parameters must be chosen. A VCC droop of 0.1V will not cause
any appreciable change in performance, while a time duration of 3 ns is long enough for other nearby bypass capacitors to help supply charge. If the current demand continues
over a long period of time, charge must be supplied by a
very large capacitor on the board. This is the reason for the
recommendation that a large capacitor be located where
VCC comes onto a board. If the buffers are also located
near the connector end of the board, the large capacitor
helps supply charge sooner.
With the advent of Fairchild Advanced Schottky Technology
(FAST) with considerably faster edge rates and switching
times, proper grounding practice has become of primary
concern in printed circuit layout. Poor circuit grounding layout techniques may result in crosstalk and slowed switching
rates. This reduces overall circuit performance and may necessitate costly redesign. Also when FAST chips are substituted for standard TTL-designed printed circuit boards, faster edge rates can cause noise problems. The source of
these problems can be sorted into three categories:
1. VCC droop due to faster load capacitance charging;
2. Coupling via ground paths adjacent to both signal sources and loads; and
3. Crosstalk caused by parallel signal paths.
TL/F/12419 – 56
Q e CV
I e CDV/Dt
C e IDt/DV
Dt e 3 c 10b9
Specify VCC Droop e 0.1V Max.
0.4 c 3 c 10b9
12 c 10b9 e 0.012 mF
Select CB t 0.02 mF
Place one bypass capacitor near each buffer package. Distribute other bypass capacitors evenly throughout the logic, one capacitor per two packages.
VCC droop can be remedied with better or more bypassing
to ground. The rule here is to place 0.01 mF capacitors from
VCC to ground for every two FAST circuits used, as near the
IC as possible. The other two problems are not as easily
corrected, because PC boards may already be manufactured and utilized. In this case, simply replacing TTL circuits
with FAST compatible circuits is not always as easy as it
may seem, especially on two-sided boards. In this situation
IC placement is critical at high speeds. Also when designing
high density circuit layout, a ground-plane layer is imperative
to provide both a sufficiently low inductance current return
path and to provide electromagnetic and electrostatic
shielding thus preventing noise problem 2 and reducing, by
a large degree, noise problem 3.
When considering the two-sided PC board, more than one
ground trace is often found in a parallel or non-parallel configuration. For this illustration parallel traces tied together at
one end are shown. This arrangement is referred to as a
ground comb. The ground comb is placed on one side of the
PC board while the signal traces are on the other side, thus
the two-sided circuit board (Figure 9-35) .
TL/F/12419 – 57
TL/F/12419 – 58
Figure 9-36 illustrates how noise is generated even though
there is no apparent means of crosstalk between the circuits. If package A has an output which drives package D
input and package B output drives package C input, there is
no apparent path for crosstalk since mutual signal traces
are remotely located. What is significant, and must be emphasized here, is that circuit packages A and B accept their
ground link from the same trace. Hence, circuit A may well
couple noise to circuit B via the common or shared portion
of the trace. This is especially true at high switching speeds.
System faults occur if the sum of VOL quiescent level plus
current spike amplitude reaches the threshold region of
gate C. From this it can be seen that erroneous switching
may be transmitted throughout the system. In the illustration
the glitch at gate B’s output is given by the following formula:
L1 (di/dt)
Ground trace noise coupling is illustrated by a model circuit
in Figure 9-37 . With the ground comb configuration, the
ground strips may be shown to contain distributed inductance, as is indeed the case. Referring to the above illustration we can see that if we switch gate A from HIGH to LOW,
the current for the transition is drawn from ground strip number two. Current flows in the direction indicated by the arrow
to the common tie point. It can be seen that gate B shares
ground strip number two with gate A from the point where
gate B is grounded back to the common tie point. This
length is represented by L1. When A switches states there is
a current transient which occurs on the ground strip in the
positive direction. This current spike is caused by the
ground strip inductance and it is ‘‘felt’’ by gate B. If gate B is
in a LOW state (VOL) the spike will appear on the output
since gate B’s VOL level is with reference to ground. Thus if
gate B’s ground reference rises momentarily VOL will also
rise. Consequently, if gate B is output to another gate (C in
the illustration) problems may arise.
TL/F/12419 – 60
TL/F/12419 – 59
However, when designing with FAST products, consider
that sockets increase total lead inductance and interlead
capacitance, thus circuit performance may be adversely affected. If boards without ground and power grids must be
used or if non-standard pin connections must be accommodated, the use of copper strips or braid is recommended.
Copper strip is readily available as shim stock while a brand
of solder wick can be used for braid material. Please note
here that jumper wires must be avoided because of wire
inductance. Wire inductance, like stripline inductance, will
slow rise and fall times. Jumpers also promote crosstalk
The following sketch (Figure 9-39) shows one method of
effectively reducing ground path length when using the
ground comb layout. By using topside traces to tie the underside ground comb together we can reduce ground strip
distributed inductances. Therefore, current transients are
significantly smaller or nonexistent in amplitude. In the application of these topside strips, care need not be exercised in
their spacing or arrangement. Parallelism is not of paramount importance either. Another advantage is evident: if
one or more of these strips is placed between topside signal
traces, crosstalk can be eliminated between those traces.
As stated earlier under ‘‘GroundÐAn Essential Link’’, it was
noted that the common rule of thumb is to decouple every
other FAST package with 0.01 mF capacitors. This is fine for
most gates in the majority of applications. However, with
buffer/driver packages, decoupling should occur at each
package since the possibility of all outputs switching coincidentally exists and can cause large loads on VCC.
An alternative to standard decoupled power traces on the
two-sided P.C. board is a product called Q/PAC*. Q/PAC is
a low impedance, high capacitance power distribution system which uses wide VCC and ground conductors in close
proximity with ceramic insulators to effectively represent integral decoupling capacitors. Packages are available in
varying lengths and pinout spacings.
TL/F/12419 – 61
An area which warrants special consideration is bus driver/buffer package placement. Here we refer to products
such as the ’F240, ’F241, ’F244, ’F540, ’F827 and ’F828.
These units have a minimum of eight outputs. A problem
may arise if all eight outputs happen to switch from HIGH to
LOW or vice versa at the same time. In this case the chance
for a large current transient on the ground circuit is apparent. This is possible even on short runs of ground strip (1 to
2 inches). Here, extra care is advised and it is suggested
that buffer/driver groups driving backplanes be segregated
to one area in the circuit. This area should have its own
ground reference. Ideally it should be a ground plane configuration or contain minimal or negligible length ground trace
Crosstalk is an interference effect of an active signal line on
an inactive signal line in close proximity to the active line.
There are two forms of crosstalk that are of concern in system design: forward and reverse crosstalk. The causes of
both kinds are similar, but the effects are significantly different. Four possible crosstalk conditions can exist at the inactive receiver: (1) a positive pulse on a LOW, (2) a negative
pulse on a LOW, (3) a positive pulse on a HIGH, (4) a negative pulse on a HIGH. Of the four previously mentioned conditions (1) and (4) are of major concern in logic systems,
and (2) and (3) are less problematic. Crosstalk is caused by
a number of interrelated factors which fall into two groups:
mutual impedance and velocity difference.
Mutual impedance is caused by the mutual inductance and
mutual capacitance distributed along two signal lines in
close proximity. The electrical effects are akin to transformer action with well defined polarities. The induced crosstalk
voltage pulse is of opposite polarity to the inducing pulse.
Figure 9-40 shows the schematic representation.
It is important when breadboarding, creating prototype circuits for evaluation or making special function generators,
to use optimum techniques for connecting VCC and ground.
Breadboard-type selection is of certain consequence here
and should be attended to wisely.
The best choice, when designing with high-speed logic, is
board material which has power and ground already connected to circuit trace grids. Boards may offer the designer
the option of using IC sockets although these are not recommended for high-speed applications. Socket layout is
convenient and may be necessary when special or one-ofa-kind circuits are utilized in initial circuit arrangements.
TL/F/12419 – 62
*Rogers Corporation
Q/PAC Division, 5750 East McKellips Road
Mesa, AZ 85205 Telephone: 602-830-3370
Here Z1 and Z2 represent the adjacent signal line impedances, and ZC is the mutual impedance coupling the two
signal lines. An equivalent circuit is shown in Figure 9-41 .
Crosstalk on printed circuit traces exhibits both velocity difference and mutual impedance effects. This can be seen
clearly in Figure 9-42 . The jig, two 50X parallel traces, 34
inches long and 0.100 inches apart, was characterized using a 5V 3 ns rise time signal from a 50X source and all
traces terminated in 50X. Figures 9-43 through 9-50 show
the effects of forward and reverse crosstalk on terminated
and unterminated cases using the jig of Figure 9-42 . All of
the cases show no approach to the logic threshold on this
test jig; other circuit configurations and impedances may not
act in a similar fashion and crosstalk avoidance procedures
may have to be taken.
RS is the effective source resistance; for VOH, RS e 33X
and VOL, RS e 3X. These are the typical FAST gate sink
and source resistances. VC is the crosstalk voltage and
should be adjusted for polarity. The crosstalk voltage can be
calculated with the following simplified formula:
VC e
RS a ZC a Z1/2 a Z2/2
Velocity differences are caused when a signal propagates
along a conductive medium that is in contact with substances of different dielectric constants, i.e., epoxy glass
and air in printed circuit board applications. The different
dielectric constants of the materials cause the wave propagating at the epoxy glass interface to be traveling slower
than the wave at the air interface. This has the effect of
generating a pulse that will couple electrostatically into the
adjacent signal line and add to the pulse caused by mutual
impedance coupling. The velocity difference pulse will have
the same rise time as the signal on the active line and its
duration will be twice the difference between the arrival of
the wave front in air and the wave front in epoxy glass.
TL/F/12419 – 64
a. Velocity Difference
Forward Crosstalk
Forward crosstalk is the effect when the active driver and
the driver on the non-active line are at the same end: the
wave front propagates toward the active and non-active receiver simultaneously. Forward crosstalk is classically attributed almost entirely to velocity differences, but in practice it
is a mixture of both velocity difference and mutual impedance effects.
TL/F/12419 – 65
Reverse Crosstalk
Reverse or backward crosstalk is the effect when the active
driver and the non-active receiver are at the same end of
the signal lines: the wave front propagates toward the active
receiver and the non-active driver simultaneously. Reverse
crosstalk is due entirely to mutual impedance effects. Forward and reverse crosstalk tests have been performed on
both parallel circuit board traces and ribbon cable.
b. Mutual Impedance
FIGURE 9-42. 34-Inch, 50X Crosstalk Jig
TL/F/12419 – 66
TL/F/12419 – 67
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – 68
TL/F/12419 – 69
c. Active Receiver
d. Active Driver
FIGURE 9-43. Reverse PC Board Crosstalk
through 34-Inch, 0.100 Trace Unterminated
TL/F/12419 – 71
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – 73
c. Active Receiver
d. Active Driver
FIGURE 9-44. Forward PC Board Crosstalk through 34-Inch Trace Unterminated
TL/F/12419 – 74
TL/F/12419 – 75
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – 76
TL/F/12419 – 77
c. Active Receiver
d. Active Driver
FIGURE 9-45. PC Board Crosstalk through 36-Inch,
0.100 Trace, Forward with No Termination
TL/F/12419 – 79
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – 81
c. Active Receiver
d. Active Driver
FIGURE 9-46. Reverse PC Board Crosstalk through 34-Inch, 0.100 Trace Unterminated
TL/F/12419 – 82
TL/F/12419 – 83
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – 84
TL/F/12419 – 85
c. Active Receiver
d. Active Driver
FIGURE 9-47. Reverse PC Board Crosstalk through 34-Inch, 0.100 Trace Terminated
TL/F/12419 – 87
b. Adjacent Driver
a. Adjacent Receiver
TL/F/12419 – 89
c. Active Receiver
d. Active Driver
FIGURE 9-48. Forward PC Board Crosstalk through 34-Inch Trace Terminated
TL/F/12419 – 90
a. Adjacent Receiver
TL/F/12419 – 91
b. Adjacent Driver
TL/F/12419 – 92
TL/F/12419 – 93
c. Active Receiver
d. Active Driver
FIGURE 9-49. Forward PC Board Crosstalk through 34-Inch Trace with Termination
TL/F/12419 – 95
b. Adjacent Driver
a. Adjacent Receiver
TL/F/12419 – 97
c. Active Receiver
d. Active Driver
FIGURE 9-50. Reverse PC Board Crosstalk through 34-Inch, 0.100 Trace Terminated
hibiting only leakage currents. This high impedance state
causes the current that has been induced into the line to
reflect from both ends and induce crosstalk back into the
active line. This action will continue until damped by circuit
resistance and leakages. The charge pump effect will leave
the adjacent line at around 7V. If this line is then switched
low, twice the normal energy is required to switch the line,
thus almost doubling the crosstalk generated in the previous case. The terminated lines show the true magnitude of
the crosstalk. Note that when the adjacent line is in the
LOW state, the crosstalk will cause the driver output to turn
off until clamped by the diode in the output structure.
Crosstalk on Ribbon Cable
Crosstalk on ribbon cable shows no velocity difference effectsÐbecause the cable insulation is a homogeneous medium, all effects are due to mutual impedance. The results
of tests on three foot sections of 160X ribbon cable are
shown in Figures 9-51 through 9-58 . From these it can be
seen that the unterminated lines exhibit large amounts of
ringing due to unterminated energy being transferred between lines. Note also that when the adjacent line is in a
HIGH state a charge pump effect occurs, forcing the HIGH
output above the VCC supply and into a high impedance
state with the output structure turned off and the input ex-
TL/F/12419 – 98
a. Adjacent Receiver
TL/F/12419 – A0
c. Active Driver
TL/F/12419 – 99
b. Adjacent Driver
FIGURE 9-51. Forward Crosstalk Using FAST and 3-Foot Ribbion Cable, Unterminated
TL/F/12419 – A2
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – A3
c. Active Driver
FIGURE 9-52. Forward Crosstalk Using FAST and 3-Foot, 2-Conductor Ribbon Cable, Unterminated
TL/F/12419 – A4
a. Adjacent Receiver
TL/F/12419 – A6
c. Active Driver
TL/F/12419 – A5
b. Adjacent Driver
FIGURE 9-53. Reverse Crosstalk Using FAST and 3-Foot, 2-Conductor Ribbon Cable, Unterminated
TL/F/12419 – A8
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – A9
c. Active Driver
FIGURE 9-54. Reverse Crosstalk Using FAST and 3-Foot, 2-Conductor Ribbon Cable, Unterminated
TL/F/12419 – B0
a. Adjacent Receiver
TL/F/12419 – B3
a. Adjacent Receiver
TL/F/12419 – B1
b. Adjacent Driver
TL/F/12419 – B4
b. Adjacent Driver
TL/F/12419 – B2
c. Active Driver
FIGURE 9-55. Forward Crosstalk Using FAST and
3-Foot, 2-Conductor Ribbon Cable, Terminated
TL/F/12419 – B5
c. Active Driver
FIGURE 9-56. Reverse Crosstalk Using FAST and
3-Foot, 2-Conductor Ribbon Cable, Terminated
a. Adjacent Receiver
TL/F/12419 – B9
a. Adjacent Receiver
b. Adjacent Driver
TL/F/12419 – C0
b. Adjacent Driver
c. Active Driver
FIGURE 9-57. Reverse Crosstalk Using FAST and
3-Foot, 2-Conductor Ribbon Cable, Terminated
TL/F/12419 – C1
c. Active Driver
FIGURE 9-58. Forward Crosstalk Using FAST and
3-Foot, 2-Conductor Ribbon Cable, Terminated
In order to minimize crosstalk it is necessary to consider the
causes during the design of systems. Some preventative
measures are as follows:
1. Always use maximum allowable spacing between signal
2. Minimize spacing between signal lines and ground lines;
3. Run ground strips alongside either the cross-talker or the
cross-listener and between the two when possible;
4. In backplane and wire-wrap applications use twisted pair
for sensitive functions such as clocks, asynchronous set
or clear, asynchronous parallel load (especially leading to
LS inputs); and
5. For ribbon or flat cabling make every other conductor a
ground line.
In the case where systems or boards are already built and
problems are encountered, some temporary or quick fixes
may be utilized. They are:
1. With printed circuit boards, glue a source of ground, either a wire or a copper strip, alongside the cross-talker or
cross-listenerÐpreferably between them;
2. For the backplane or wire-wrap situation, spiral a ground
wire around the talker to confine its electromagnetic field
or around the listener in order to shield it, or do both;
3. Try the split-resistor termination on the offending line
(Figure 9-41) ;
TL/F/12419 – C3
TL/F/12419 – C4
Trace proximity and coupled trace length are the two main
factors which affect the amount of reverse crosstalk that
occurs. Therefore, if coupled length is long, noise will be at
a maximum. For short lengths, noise may appear only as a
short spike which can cause difficulties and even system
TL/F/12419 – C2
where R1/R2 e Equivalent Thevenin Resistance of Termination
4. Cut the offending crosstalk trace from the PC board and
replace it with a wire. In this method reverse and forward
crosstalk can be reduced. The line in this case may be
lengthened, thereby increasing propagation delays, but a
rerouting of the generating signal line may eliminate the
Termination can be used to reduce the effects of crosstalk.
It can be seen here that a little termination is better than no
Energy which can be stored in a capacitor is given by the
When two lines do not run between the same points but are
in proximity over part of their length, signal propagation time
(line delay) along this coupled length is T. If T is long compared to the rise of the signal on the active line, the crosstalk pulse has time to develop its full amplitude. The trailing
edge of the noise pulse is caused by the reflection from the
driven end of the passive line. When T is half the rise time,
the reflection from the driven end of the passive line arrives
and beings to pull the noise pulse down just as it reaches
full amplitude. Any value of T less than half the rise time of
the active signal will cause a reflection to arrive and oppose
the noise pulse voltage before it can reach full amplitude.
The noise will therefore be lower in amplitude.
E e (/2 CV2
E e Energy in joules (watts-sec)
V e Applied voltage
C e Capacitance in farads
A capacitor is a reactive component which reacts against a
change in potential across it. This is shown by the equation
for the linear charge of a capacitor:
I e Current
C e Capacitance
dV/dt e Slope of voltage transition across capacitor
Thus an infinite current would be required to instantly
change the potential across a capacitor, and the amount of
current a capacitor can ‘‘sink’’ is given by the above equation.
A capacitor, as a practical device, exhibits not only capacitance but also resistance and inductance. A simplified schematic for the equivalent circuit is:
A capacitor is a component which is capable of storing electrical energy. It consists of conductive plates (electrodes)
separated by insulating material which is called the dielectric. A typical formula for determining capacitance is:
Iideal e C
0.224 KA
Capacitance (farads)
Dielectric constant (Vacuum e 1)
Area in square inches
Separation between plates in inches (thickness
of dielectric)
0.224 e Conversion constant (0.0884 for metric system
in cm)
CapacitanceÐThe standard unit of capacitance is the farad.
A capacitor has a capacitance of 1 farad when 1 coulomb
charges it to 1V. One farad is a very large unit and most
capacitors have values in the micro (10b6), nano (10b9), or
pico farad (10b12) level.
Dielectric ConstantÐIn the formula for capacitance given
above, the dielectric constant of a vacuum is arbitrarily chosen as the number 1. Dielectric constants of other materials
are then compared to the dielectric constant of a vacuum.
Dielectric constants of some typical materials are as follows:
Ruby Mica
Ceramic (Class 1)
Ceramic (Class 2)
Aluminum Oxide
Tantalum Oxide
Dielectric ThicknessÐCapacitance is indirectly proportional
to the separation between electrodes. Lower voltage requirements mean thinner dielectrics and greater capacitance per volume.
AreaÐCapacitance is directly proportional to the area of the
electrodes. Since the other variables in the equation are
usually set by the performance desired, area is the easiest
parameter to modify to obtain a specific capacitance within
a material group.
TL/F/12419 – C5
C e Capacitance
RS e Series Resistance
L e Inductance
Rp e Parallel Resistance
All the factors shown above are important in the application
of capacitors. The inductance determines the usefulness of
the capacitor at high frequency, the parallel resistance affects performance in timing and coupling circuits (normally
expressed as Insulation Resistance) and the series resistance is a measure of the loss in the capacitor and is a major
factor in Power Factor and/or Dissipation Factor.
Since the insulation resistance (Rp) is normally very high,
the total impedance of a capacitor is:
Z e 0RS2 a (XC b XL)2
Z e Total Impedance
RS e Series Resistance
XC e Capacitive Reactance e (/2qfc
XL e Inductive Reactance e 2qfL
The variation of a capacitor’s impedance with frequency determines its effectiveness in many applications.
Power Factor and Dissipation Factor are often confused
since they are both measures of the loss in a capacitor under AC application and are often almost identical in value. In
a ‘‘perfect’’ capacitor the current in the capacitor will lead
the voltage by 90§ .
discharge instantaneously upon application of a short circuit, but drains gradually after the capacitance proper has
been discharged. It is common practice to measure the dielectric absorption by determining the ‘‘reappearing voltage’’ which develops across a capacitor at some point in
time after it has been fully discharged under short circuit
Corona is the ionization of air or other vapors which causes
them to conduct current. It is especially prevalent in high
voltage units but can occur with low voltages as well where
high voltage gradients occur. The energy discharged degrades the performance of the capacitor and can in time
cause catastrophic failures.
The usual characteristics that are specified for a capacitor
include Capacitance, Dissipation Factor or ESR, Insulation
Resistance or Leakage Current and Dielectric Strength. The
electrical and environmental parameters that are of most
interest with respect to these four basic measurements are
temperature, voltage and test frequency. The reference
temperature for most capacitor measurements is 25§ C. Voltage is dependent on the rating applied by the manufacturer
and the test frequency typically depends on the class of
As the ambient temperature changes, the dielectric constant and hence the capacitance of many capacitors changes. In general, when the dielectric constant is lower, materials tend to change capacitance less with temperature or
with relatively predictable changes that are linear with temperature. High dielectric constant meterials tend to have capacitance changes that are non-linear and expressed as
percent capacitance change over a temperature range. Increasing temperature usually reduces Insulation Resistance, increases Leakage Current and Power Factor/Dissipation Factor and reduces the voltage rating of the part.
Some ceramic capacitors actually exhibit a decrease in DF
with increasing temperature. Conversely, reducing temperature normally improves most characteristics.
The effects of applied voltage on capacitors are a prime
consideration in use. Capacitance and other parameter
changes occur under both AC and DC applied voltages.
Even those cases where voltage application does not
change the parametric characteristics of a capacitor, the
level of voltage applied will determine the life expectancy of
the capacitor.
Frequency is the third factor which is of great concern in the
application of capacitors. This is an area that is often overlooked by designers. Earlier an equivalent circuit was given
for a capacitor. Inductance which is caused by the leads
and the electrodes was depicted. As the frequency applied
to the capacitor increases it eventually passes through selfresonance and becomes inductive with gradually increasing
impedance. Even though a capacitor is beyond the self-resonant point it still blocks DC and has a low impedance and
thus is useful in bypass, coupling and many other applications. Care should be taken in feedback, tuning, phase shift
and such applications.
TL/F/12419 – C6
In practice the current leads the voltage by some other
phase angle due to the series resistance RS. The complement of this angle is called the loss angle and:
Power Factor (PF) e cos w or sine e
Dissipation Factor (DF) e tan e
For small values of e the tan and sine are essentially equal,
which has led to the common interchangeability of the two
terms in the industry.
The term ESR or Equivalent Series Resistance combines all
losses, both series and parallel, in a capacitor at a given
frequency so that the equivalent circuit is reduced to a simple R-C series connection.
TL/F/12419 – C7
e (2q fc) (ESR)
Dissipation Factor e
The DF/PF of a capacitor tells what percent of the apparent
power input will turn to heat in the capacitor. The watts loss
Watts Loss e (2q fcE2) (DF)
Very low values of dissipation factor are expressed as their
reciprocal for convenience. These are called the ‘‘Q’’, or
Quality factor of capacitors.
Insulation Resistance is the resistance measured across the
terminals of a capacitor and consists principally of the parallel resistance Rp shown in the equivalent circuit. As capacitance values and hence the area of dielectric increases, the
IR decreases. The product (C c IR or RC) is often specified
in ohm farads or commonly megohm microfarads.
Dielectric Strength is an expression of the ability of a material to withstand an electrical stress. Although dielectric
strength is ordinarily expressed in volts, it is actually dependent on the thickness of the dielectric and thus is also more
generically a function of volts/mil.
Other specialized factors which may be of interest to the
user, especially in high voltage applications, are corona and
dielectric absorption.
The phenomenon of Dielectric Absorption is exhibited in the
following manner: charging current from a steady unidirectional source continues to flow at a gradually decreasing
rate into a capacitor of negligible series resistance for some
time after the almost instantaneous charge is completed. A
steady value proportional to the capacitor parallel resistance is finally reached. The additional charge apparently is
absorbed by the dielectric. Conversely, a capacitor does not
Ceramic capacitors are the most widely used capacitors.
They come in an extremely wide range of mechanical configurations and electrical characteristics. The common mechanical variations are discs, tubulars, feed throughs and
monolithics. A 0.01 mF disc is about (/2 inch in diameter
while the 0.01 mF monolithic chip capacitor is only 0.050× x
0.075× x 0.030× . Electrically, ceramic capacitors are broken
From a mechanical point of view, ceramic capacitors are
manufactured by two basic techniques. One method involves pressing or extruding the ceramic material, firing (sintering) the ceramic and subsequently applying electrodes
(typically with silver materials) which are fired onto the ceramic at lower temperatures after the maturation of the ceramic. This is the method employed in the fabrication of
single layer devices. The most common form of single layer
capacitors is disc capacitors with radial leads or tubular capacitors which are available with axial leads, radial leads or
in feed-through form with both bolt and eyelet types being
common. There are specialized versions of pressed ceramic capacitors, such as high voltage cartwheels and double
cup high voltage units. These and other types may be considered as jumbo size disc pressed units.
The second method of fabricating ceramic capacitors
evolved in recent years as a result of the demand for lower
voltages and smaller sizes consistent with the advent of
semiconductor usage. The miniaturization in the ceramic capacitor area was made possible through the manufacture of
monolithic types of ceramic capacitors. These capacitors
are manufactured by mixing the ceramic powder in an organic binder (slurry) and casting it by one technique or another into thin layers typically ranging from about 3 mils in
thickness down to 1 mil or thinner.
Metal electrodes are deposited onto the green ceramic layers which are then stacked to form a laminated structure.
The metal electrodes are arranged so that their terminations
alternate from one edge of the capacitor to another. Upon
sintering at high temperature the part becomes a monolithic
block which can provide extremely high capacitance values
in small mechanical volumes. Figure 9-65 shows a pictorial
view of a monolithic ceramic capacitor.
into two classes. Class 1 ceramic dielectrics are also called
temperature compensating ceramics and feature zero TC
and other predictable and relatively linear TC bodies. The
insulation resistance is high, the losses are low and the
parts are essentially unaffected by voltage or frequency and
are usually used for tuned circuits, timing applications and
other precision circuits.
Where Class 1 ceramics are completely predictable, Class 2
general purpose ceramics are full of surprises for the unsuspecting engineer. Not only does capacitance change with
temperature but the ‘‘high K’’ units which are so enticingly
small in size may lose 90% of their room temperature capacitance at b55§ C. Further care must be exercised when
voltage is applied, particularly with monolithic capacitors
with their thin dielectrics. AC voltage caused the capacitance to increase and DC voltage causes a capacitance
loss. A change in frequency also changes capacitance and
The fact that more ceramic Class 2 capacitors are used
than all other types combined proves that the variability of
characteristics not only can be overcome by wise selection
but can in many cases be an advantage. Considerably more
detailed information is given below and a number of articles
and booklets are also available on this subject.
The ceramic capacitor is defined as a capacitor manufactured from metallic oxides, sintered at a high temperature.
As a general rule, the electrical ceramics used in capacitors
are based on complex titanate compounds, principally barium titanate, rare earth titanates, calcium titanates, sodium
titanate, etc. Occasionally other materials, such as lead niobiate, may be used.
TL/F/12419 – C8
aging characteristics. Thus they operate in a manner similar
to mica capacitors except for the TC which is controllable.
Normally the TCs of Class 1 capacitors are deemed to run
between P100 and N750. Class 1 extended temperature
compensating capacitors are also manufactured in TCs
from negative 1400 through negative 5600, however, these
may start developing a slight aging characteristic and voltage susceptibility.
Most TC formulations are available in pressed and extruded
construction while only NPO (zero TC) is provided by most
manufacturers in monolithic construction. NPO ceramics in
monolithic capacitors are available in high enough values to
cover most applications requiring extreme stability. With the
exception of some NPO capacitors almost all temperature
compensating capacitors have a TC curve which is a true
curve and not a straight line. The TC tends to become more
negative at the cold end than it is from the 25§ C reference to
a 85§ C. Both EIA specification RS-198 and military specification MIL-C-20 contain information about curvature. This
informatin is contained in Table II. These charts are based
on industry accepted standard TC values.
While pressed and extruded ceramic capacitors are in general low cost and provide limited capacitance values, monolithic units are typically smaller in size, feature excellent high
frequency characteristics because of the small size and provide considerably higher capacitance values with low voltage ratings.
Ceramic capacitors are available in a tremendous variety of
characteristics. Electronic Industries Association (EIA) and
the military have established categories to help divide the
basic characteristics into more easily specified classes. The
basic industry specifications for ceramic capacitors is EIA
specification RS-198 and as noted in the general section it
specifies temperature compensating capacitors as Class 1
capacitors. These are specified by the military under specification MIL-C-20. General purpose capacitors with non-linear temperature coefficients are called Class 2 capacitors
by EIA and are specified by the military under MIL-C-11015
and MIL-C-39014. EIA specifications further include a Class
3 category which is defined as reduced titanates.
Class 1 or temperature compensating capacitors are usually
made from mixtures of titanates where barium titanate is
normally not a major part of the mix. They have predictable
temperature coefficients and in general do not have any
TABLE 9-2. TC Tolerances (Note 1)
in pF
a 30
b 80
a 30
b 90
a 30
b 105
a 30
b 120
a 60
b 180
a 60
b 210
a 120
b 340
a 250
b 670
a 500
b 1100
b 55§ C to a 25§ C in PPM/§ C
10 and Over
a 30
b 75
a 25§ C to a 85§ C in PPM/§ C
10 and over
g 30
g 30
g 30
g 30
g 30
g 60
g 60
g 120
g 250
g 500
EIA Desig.
Note 1: Table II indicates the tolerance available on specific temperature characteristics. It may be noted that limits are established on the basis of measurements
at a 25§ C and a 85§ C and that TC becomes more negative at low temperature. Wider tolerances are required on low capacitance values because of the effects of
stray capacitance.
now use higher dielectric constants than 1200 so the term is
now taken to mean only X7R and is commonly called semistable material.
General purpose ceramic capacitors are called Class 2 capacitors and have become extremely popular because of
the high capacitance values available in very small size.
Class 2 capacitors are ‘‘ferro electric’’ and vary in capacitance value under the influence of the environmental and
electrical operating conditions. Class 2 capacitors are affected by temperature, voltage (both AC and DC), frequency
and time. Temperature effects for Class 2 are exhibited as
non-linear capacitance changes with temperature.
In specifying capacitance change with temperature, EIA expresses capacitance change over an operating temperature
range by a 3-symbol code. The first symbol represents the
cold temperature end of the range, the second represents
the upper limit of the operating range and a third symbol
represents the capacitance change allowed over the operating temperature range. Table 9-3 provides a detailed explanation of the EIA system. As an example, a capacitor
with a characteristic X7R would change g 15% over the
temperature range b55§ C to a 125§ C and is often identical
to military characteristics BX. Parts with characteristics are
also sometimes called ‘‘K1200’’ but most manufacturers
A Z5U temperature characteristic is also extremely popular.
It allows a capacitance change of a 22% to b56% over the
temperature range of a 10§ C to a 85§ C, and is usually made
with materials with a dielectric constant in the range of 5000
to 10,000.
Whereas variations in temperature affect all of the parameters of ceramic capacitors, voltage basically affects only the
capacitance and dissipation factor. The application of DC
voltage reduces both the capacitance and dissipation factor
while the application of an AC voltage within a reasonable
range tends to increase both capacitance and dissipation
factor readings. If a high enough AC voltage is applied,
eventually it will reduce capacitance just as a DC voltage
will. However, the application of this high an AC voltage is
normally not encountered.
EIA Code
Percent Capacity Change Over Temperature Range
MIL-C-11015D Code
Temperature Range
Temperature Range
b 55§ C to a 85§ C
b 55§ C to a 125§ C
b 55§ C to a 150§ C
b 55§ C to a 125§ C
b 55§ C to a 85§ C
b 30§ C to a 85§ C
a 10§ C to a 85§ C
Per Cent Capacity Change
g 3.3%
Cap. Change
Zero Volts
Cap. Change
Rated Volts
a 15%, b 15%
a 15%, b 40%
a 22%, b 56%
a 22%, b 66%
a 15%, b 15%
a 15%, b 25%
a 30%, b 70%
a 30%, b 80%
a 20%, b 20%
a 20%, b 30%
Temperature characteristic is specified by combining range
and change symbols, for example BR or AW. Specification
slash sheets indicate the characteristic applicable to a given
style of capacitor.
g 4.7%
g 7.5%
g 10%
g 15%
g 22%
a 22%, b 33%
a 22%, b 56%
a 22%, b 82%
ExampleÐA capacitor is desired with the capacitance value
at 25§ C to increase no more than 7.5% or decrease no more
than 7.5% from b30§ C to a 85§ C. EIA Code will be Y5F.
Since the magnitude of the effect is dependent on the thickness of the dielectric versus the voltage applied (volts per
mil) the curve is based on percent of rated voltage in order
to give a basic idea of the order of magnitude of the changes in question. Figure 9-66 shows the effects of AC voltage.
These are of major significance in some applications but are
perhaps of even more significance when it comes to measuring the capcitance value and dissipation factor of the
capacitors. Capacitor specifications specify the AC voltage
at which to measure (normally 1 VAC) and application of the
wrong voltage can cause spurious readings. Figure 9-67
gives the voltage coefficient of dissipation factor for AC
based on 1000 cycles reading a 1 kHz readings. Applications of different frequencies will affect the percentage
changes versus voltages.
TL/F/12419 – D1
TL/F/12419 – D2
TL/F/12419 – C9
Frequency affects capacitance and dissipation factor as is
the case with voltage. Curves of capacitance change and
dissipation factor change with normal type ceramics are
shown in Figure 9-70 and 9-71 .
TL/F/12419 – D0
The effect of the application of DC voltage is once again
dependent on the thickness of the dielectric (volts per mil)
and is shown in a similar manner in Figure 9-68 . As will be
noted in general, the voltage coefficient is more pronounced
for higher K dielectrics. These figures are shown for room
temperature conditions. Of considerable interest to the user
is a combination characteristic known as voltage temperature limit which shows the effects of rated voltage over the
operating temperature range. Figure 9-69 shows a capacitor
of military specification type BX.
TL/F/12419 – D3
Variation of impedance with frequency is an important consideration for decoupling capacitor applications. Lead
length, lead configuration and body size all affect the impedance level as well as the ceramic formulation variations.
Special ceramic materials are also made for use at extremely high frequencies.
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