Texas Instruments | TPS54360B 60-V Input, 3.5-A, Step-Down DC/DC Converter With Eco-Mode™ | Datasheet | Texas Instruments TPS54360B 60-V Input, 3.5-A, Step-Down DC/DC Converter With Eco-Mode™ Datasheet

Texas Instruments TPS54360B 60-V Input, 3.5-A, Step-Down DC/DC Converter With Eco-Mode™ Datasheet
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TPS54360B
SNVSB93 – DECEMBER 2018
TPS54360B 60-V Input, 3.5-A, Step-Down DC/DC Converter With Eco-Mode™
1 Features
3 Description
•
•
The TPS54360B is a 60-V, 3.5-A, step-down
regulator with an integrated high side MOSFET.
Current mode control provides simple external
compensation and flexible component selection. A
low ripple pulse skip mode reduces the no-load
supply current to 146 μA. Shutdown supply current is
reduced to 2 μA when the enable pin is pulled low.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
4.5-V to 60-V (65-V Abs Max) Input Range
3.5-A Continuous Current, 4.5-A Minimum Peak
Inductor Current Limit
Current Mode Control DC/DC Converter
92-mΩ High-Side MOSFET
High Efficiency at Light Loads with Pulse Skipping
Eco-mode™
Low Dropout at Light Loads with Integrated BOOT
Recharge FET
146-μA Operating Quiescent Current
2-μA Shutdown Current
100-kHz to 2.5-MHz Fixed Switching Frequency
Synchronizes to External Clock
Adjustable UVLO Voltage and Hysteresis
Internal Soft Start
Accurate Cycle-by-Cycle Current Limit
Thermal, Overvoltage, and Frequency Foldback
Protection
0.8 V 1% Internal Voltage Reference
8-Pin HSOIC with PowerPAD™ Package
–40°C to 150°C TJ Operating Range
Create a Custom Design using the TPS54360B
with the WEBENCH® Power Designer
2 Applications
Undervoltage lockout is internally set at 4.3 V but can
be increased using the enable pin. The output voltage
start-up ramp is internally controlled to provide a
controlled start-up and eliminate overshoot.
A wide switching-frequency range allows either
efficiency or external component size to be optimized.
Frequency foldback and thermal shutdown protect
internal and external components during an overload
condition.
The TPS54360B is available in an 8-pin thermally
enhanced HSOIC PowerPAD™ package.
Device Information
PART NUMBER
TPS54360B
PACKAGE
BODY SIZE
HSOIC (8)
4.89 mm × 3.90 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
spacer
spacer
spacer
12-V,
24-V
and
48-V
Industrial
Communications Power Systems
and
spacer
spacer
Simplified Schematic
Efficiency vs Load Current
100
VIN
90
VIN
80
TPS54360B
EN
VOUT
SW
RT/CLK
GND
COMP
FB
Efficiency - %
70
BOOT
5V
3.3 V
60
50
40
VIN = 12 V
30
20
VOUT = 5 V, fsw = 600 kHz
10
VOUT = 3.3 V, fsw = 300 kHz
0
GND
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
IO - Output Current - A
Copyright © 2018, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54360B
SNVSB93 – DECEMBER 2018
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
4
5
6
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Typical Characteristics ..............................................
7.4 Device Functional Modes........................................ 21
8
Application and Implementation ........................ 22
8.1 Application Information............................................ 22
8.2 Typical Application .................................................. 22
8.3 Other Applications................................................... 34
9 Power Supply Recommendations...................... 35
10 Layout................................................................... 36
10.1 Layout Guidelines ................................................. 36
10.2 Layout Example .................................................... 36
11 Device and Documentation Support ................. 37
11.1
11.2
11.3
11.4
11.5
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 11
7.3 Feature Description................................................. 11
Device Support......................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
37
37
37
37
37
12 Mechanical, Packaging, and Orderable
Information ........................................................... 38
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
2
DATE
REVISION
NOTES
December 2018
*
Initial release
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5 Pin Configuration and Functions
DDA Package
8-Pin HSOIC
Top View
BOOT
1
VIN
2
8
SW
7
GND
PowerPAD
9
EN
3
6
COMP
RT/CLK
4
5
FB
Pin Functions
PIN
NAME
NO.
I/O
DESCRIPTION
BOOT
1
O
A bootstrap capacitor is required between BOOT and SW. If the voltage on this capacitor is below the
minimum required to operate the high-side MOSFET, the output is switched off until the capacitor is
refreshed.
VIN
2
I
Input supply voltage with 4.5 V to 60 V operating range.
EN
3
I
Enable pin, with internal pullup current source. Pull below 1.2 V to disable. Float to enable. Adjust the input
undervoltage lockout with two resistors. See the Enable and Adjusting Undervoltage Lockout section.
RT/CLK
4
I
Resistor Timing and External Clock. An internal amplifier holds this pin at a fixed voltage when using an
external resistor to ground to set the switching frequency. If the pin is pulled above the PLL upper threshold,
a mode change occurs and the pin becomes a synchronization input. The internal amplifier is disabled and
the pin is a high impedance clock input to the internal PLL. If clocking edges stop, the internal amplifier is reenabled and the operating mode returns to resistor frequency programming.
FB
5
I
Inverting input of the transconductance (gm) error amplifier.
COMP
6
O
Error amplifier output and input to the output switch current (PWM) comparator. Connect frequency
compensation components to this pin.
GND
7
–
Ground
SW
8
I
The source of the internal high-side power MOSFET and switching node of the converter.
Thermal Pad
9
–
GND pin must be electrically connected to the exposed pad on the printed circuit board for proper operation.
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SNVSB93 – DECEMBER 2018
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6 Specifications
6.1 Absolute Maximum Ratings (1)
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
VIN
–0.3
65
EN
–0.3
8.4
BOOT
Input voltage
73
FB
–0.3
COMP
–0.3
3
RT/CLK
–0.3
3.6
V
3
BOOT-SW
Output voltage
UNIT
8
SW
–0.6
65
–2
65
Operating junction temperature
–40
150
°C
Storage temperature, Tstg
–65
150
°C
SW, 10-ns transient
(1)
V
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
MAX
VESD
(1)
(2)
(3)
(1)
Human body model (HBM) esd stress voltage
(2)
Charged device model (HBM) ESD stress voltage
UNIT
±2000
(3)
V
±500
Electrostatic discharge (ESD) to measure device sensitivity and immunity to damage caused by assembly line electrostatic discharges
into the device.
Level listed above is the passing level per ANSI/ESDA/JEDEC JS-001. JEDEC document JEP155 states that 500V HBM allows safe
manufacturing with a standard ESD control process. pins listed as 1000 V may actually have higher performance.
Level listed above is the passing level per EIA-JEDEC JESD22-C101. JEDEC document JEP157 states that 250V CDM allows safe
manufacturing with a standard ESD control process. pins listed as 250 V may actually have higher performance.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
VO + VDO
60
V
Output voltage
0.8
58.8
V
IO
Output current
0
3.5
A
TJ
Junction temperature
–40
150
°C
VIN
Supply input voltage (1)
VO
(1)
UNIT
See Equation 1
6.4 Thermal Information
TPS54360B
THERMAL METRIC (1)
DDA (HSOIC)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
42
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
45.8
°C/W
RθJB
Junction-to-board thermal resistance
23.4
°C/W
ψJT
Junction-to-top characterization parameter
5.9
°C/W
ψJB
Junction-to-board characterization parameter
23.4
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.6
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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6.5 Electrical Characteristics
TJ = –40°C to +150°C, VIN = 4.5 to 60V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
60
V
4.3
4.48
V
SUPPLY VOLTAGE (VIN PINS)
Operating input voltage
Internal undervoltage lockout threshold
4.5
Rising
4.1
Internal undervoltage lockout threshold
hysteresis
325
mV
Shutdown supply current
EN = 0 V, 25°C, 4.5 V ≤ VIN ≤ 60 V
2.25
4.5
Operating: nonswitching supply current
FB = 0.9 V, TA = 25°C
146
175
1.2
1.3
μA
ENABLE AND UVLO (EN pinS)
Enable threshold voltage
Input current
No voltage hysteresis, rising and falling
1.1
Enable threshold +50 mV
Enable threshold –50 mV
Hysteresis current
–4.6
V
μA
–0.58
–1.2
-1.8
–2.2
–3.4
-4.5
μA
0.792
0.8
0.808
V
92
190
VOLTAGE REFERENCE
Voltage reference
HIGH-SIDE MOSFET
On-resistance
VIN = 12 V, BOOT-SW = 6 V
mΩ
ERROR AMPLIFIER
Input current
Error amplifier transconductance (gM)
–2 μA < ICOMP < 2 μA, VCOMP = 1 V
Error amplifier transconductance (gM) during
–2 μA < ICOMP < 2 μA, VCOMP = 1 V, VFB = 0.4 V
soft-start
Error amplifier DC gain
VFB = 0.8 V
Min unity gain bandwidth
Error amplifier source/sink
V(COMP) = 1 V, 100-mV overdrive
COMP to SW current transconductance
50
nA
350
μS
77
μS
10,000
V/V
2500
kHz
±30
μA
12
A/V
CURRENT LIMIT
Current limit threshold
All VIN and temperatures, Open Loop (1)
4.5
5.5
6.8
All temperatures, VIN = 12 V, Open Loop (1)
4.5
5.5
6.25
VIN = 12 V, TA = 25°C, Open Loop (1)
5.2
5.5
5.85
A
THERMAL SHUTDOWN
Thermal shutdown
Thermal shutdown hysteresis
176
°C
12
°C
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK pinS)
Switching frequency range using RT mode
fSW
Switching frequency
100
RT = 200 kΩ
Switching frequency range using CLK mode
450
160
RT/CLK high threshold
2500
kHz
550
kHz
2300
kHz
1.55
RT/CLK low threshold
(1)
500
0.5
2
1.2
V
V
Open Loop current limit measured directly at the SW pin and is independent of the inductor value and slope compensation.
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6.6 Timing Requirements
MIN
NOM
MAX
UNIT
INTERNAL SOFT-START TIME
Soft-start time
fSW = 500 kHz, 10% to 90%
2.1
ms
Soft-start time
fSW = 2.5 MHz, 10% to 90%
0.42
ms
VIN = 12 V, TA = 25°C
135
ns
60
ns
15
ns
HIGH-SIDE MOSFET
Minimum controllable on time
CURRENT LIMIT
Current limit threshold delay
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PINS)
Minimum CLK input pulse width
RT/CLK falling edge to SW rising edge
delay
Measured at 500 kHz with RT resistor in series
55
ns
PLL lock-in time
Measured at 500 kHz
78
μs
6.7 Typical Characteristics
0.25
0.814
VIN = 12 V
VFB − Voltage Reference (V)
RDS(ON) − On-State Resistance (Ω)
BOOT-SW = 3 V
BOOT-SW = 6 V
0.2
0.15
0.1
0.05
0
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
0.809
0.804
0.799
0.794
0.789
0.784
−50
Figure 1. On-Resistance vs Junction Temperature
150
G002
TJ = −40°C
TJ = 25°C
TJ = 150°C
6.3
High-Side Switch Current (A)
High-Side Switch Current (A)
125
6.5
VIN = 12 V
6.3
6.1
5.9
5.7
5.5
5.3
5.1
4.9
4.7
6.1
5.9
5.7
5.5
5.3
5.1
4.9
4.7
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
4.5
0
G003
Figure 3. Switch Current Limit vs Junction Temperature
6
0
25
50
75
100
TJ − Junction Temperature (°C)
Figure 2. Voltage Reference vs Junction Temperature
6.5
4.5
−50
−25
G001
10
20
30
40
VIN − Input Voltage (V)
50
60
G004
Figure 4. Switch Current Limit vs Input Voltage
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Typical Characteristics (continued)
500
RT = 200 kΩ, VIN = 12 V
540
ƒSW − Switching Frequency (kHz)
ƒSW − Switching Frequency (kHz)
550
530
520
510
500
490
480
470
460
450
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
350
300
250
200
150
100
50
0
200
300
G005
400
500
600
700
800
RT/CLK − Resistance (kΩ)
900
1000
G006
Figure 6. Switching Frequency vs RT/CLK Resistance
Low Frequency Range
500
2500
VIN = 12 V
450
2000
400
1500
gm (uS)
ƒSW − Switching Frequency (kHz)
400
150
Figure 5. Switching Frequency vs Junction Temperature
1000
350
300
500
0
250
0
50
100
150
RT/CLK − Resistance (kΩ)
200
−50
200
VIN = 12 V
EN − Threshold (V)
100
90
80
70
60
50
40
30
20
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
1.3
1.29
1.28
1.27
1.26
1.25
1.24
1.23
1.22
1.21
1.2
1.19
1.18
1.17
1.16
1.15
−50
G009
Figure 9. EA Transconductance During Soft Start vs
Junction Temperature
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
G008
Figure 8. EA Transconductance vs Junction Temperature
120
110
−25
G007
Figure 7. Switching Frequency vs RT/CLK Resistance
High Frequency Range
gm (uS)
ƒSW (kHz) = 92417 × RT (kΩ)−0.991
RT (kΩ) = 101756 × fSW (kHz)−1.008
450
VIN = 12 V
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
G010
Figure 10. EN Pin Voltage vs Junction Temperature
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Typical Characteristics (continued)
−4
−0.5
VIN = 5 V, I EN = Threshold+50mV
−0.9
−4.2
−1.1
−4.3
−1.3
−4.4
−1.5
−1.7
−4.5
−4.6
−1.9
−4.7
−2.1
−4.8
−2.3
−4.9
−2.5
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
−5
−50
150
0
25
50
75
100
Tj − Junction Temperature (°C)
% of Nominal Switching Frequency
−2.9
−3.1
−3.3
−3.5
−3.7
−3.9
−4.1
−4.3
VIN = 12 V
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
75
50
25
0
150
0
0.1
G112
0.2
0.3
0.4
VFB (V)
0.5
0.6
0.7
0.8
G013
Figure 14. Switching Frequency vs FB
3
3
VIN = 12 V
TJ = 25°C
Shutdown Supply Current (µA)
Shutdown Supply Current (µA)
G012
VFB Falling
VFB Rising
Figure 13. EN Pin Current Hysteresis vs Junction
Temperature
2.5
2
1.5
1
0.5
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
Figure 15. Shutdown Supply Current vs Junction
Temperature
8
150
100
−2.7
0
−50
125
Figure 12. EN pin Current vs Junction Temperature
−2.5
EN PIN Current Hysteresis (µA)
−25
G011
Figure 11. EN pin Current vs Junction Temperature
−4.5
−50
VIN = 12 V, I EN = Threshold+50mV
−4.1
IEN (µA)
IEN (µA)
−0.7
2.5
2
1.5
1
0.5
0
0
G014
10
20
30
40
VIN − Input Voltage (V)
50
60
G015
Figure 16. Shutdown Supply Current vs Input Voltage (VIN)
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Typical Characteristics (continued)
210
210
VIN = 12 V
TJ = 25°C
190
VIN − Supply Current (µA)
VIN − Supply Current (uA)
190
170
150
130
110
90
70
−50
170
150
130
110
90
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
70
150
Figure 17. VIN Supply Current vs Junction Temperature
50
60
G017
4.4
4.3
Input Voltage (V)
BOOT-SW UVLO (V)
20
30
40
VIN − Input Voltage (V)
4.5
BOOT-SW UVLO Falling
BOOT-SW UVLO Rising
2.4
2.3
2.2
2.1
4.2
4.1
4
2
3.9
1.9
3.8
1.8
−50
10
Figure 18. VIN Supply Current vs Input Voltage
2.6
2.5
0
G016
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
UVLO Start Switching
UVLO Stop Switching
3.7
−50
−25
G018
Figure 19. BOOT-SW UVLO vs Junction Temperature
0
25
50
75
100
Tj − Junction Temperature (°C)
125
150
G019
Figure 20. Input Voltage UVLO vs Junction Temperature
10
VIN = 12V,
o
TJ = 25 C
9
Soft-Start Time (ms)
8
7
6
5
4
3
2
1
0
100 300 500 700 900 11001300 1500 17001900 2100 2300 2500
Switching Frequency (kHz)
G021
Figure 21. Soft-Start Time vs Switching Frequency
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7 Detailed Description
7.1 Overview
The TPS54360B is a 60-V, 3.5-A, step-down (buck) regulator with an integrated high side n-channel MOSFET.
The device implements constant frequency, current mode control which reduces output capacitance and
simplifies external frequency compensation. The wide switching frequency range of 100 kHz to 2500 kHz allows
either efficiency or size optimization when selecting the output filter components. The switching frequency is
adjusted using a resistor to ground connected to the RT/CLK pin. The device has an internal phase-locked loop
(PLL) connected to the RT/CLK pin that synchronizes the power switch turnon to a falling edge of an external
clock signal.
The TPS54360B has a default input start-up voltage of approximately 4.3 V. The EN pin can be used to adjust
the input voltage undervoltage lockout (UVLO) threshold with two external resistors. An internal pull up current
source enables operation when the EN pin is floating. The operating current is 146 μA under no load condition
(not switching). When the device is disabled, the supply current is 2 μA.
The integrated 92-mΩ high side MOSFET supports high efficiency power supply designs capable of delivering
3.5 Amperes of continuous current to a load. The gate drive bias voltage for the integrated high side MOSFET is
supplied by a bootstrap capacitor connected from the BOOT to SW pins. The TPS54360B reduces the external
component count by integrating the bootstrap recharge diode. The BOOT pin capacitor voltage is monitored by a
UVLO circuit which turns off the high side MOSFET when the BOOT to SW voltage falls below a preset
threshold. An automatic BOOT capacitor recharge circuit allows the TPS54360B to operate at high duty cycles
approaching 100%. Therefore, the maximum output voltage is near the minimum input supply voltage of the
application. The minimum output voltage is the internal 0.8-V feedback reference.
Output overvoltage transients are minimized by an overvoltage transient protection (OVP) comparator. When the
OVP comparator is activated, the high side MOSFET is turned off and remains off until the output voltage is less
than 106% of the desired output voltage.
The TPS54360B includes an internal soft-start circuit that slows the output rise time during start-up to reduce inrush current and output voltage overshoot. Output overload conditions reset the soft-start timer. When the
overload condition is removed, the soft-start circuit controls the recovery from the fault output level to the nominal
regulation voltage. A frequency foldback circuit reduces the switching frequency during start-up and overcurrent
fault conditions to help maintain control of the inductor current.
10
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7.2 Functional Block Diagram
EN
VIN
Thermal
Shutdown
UVLO
Enable
Comparator
OV
Shutdown
Shutdown
Logic
Enable
Threshold
Boot
Charge
Voltage
Reference
Boot
UVLO
Minimum
Clamp
Pulse
Skip
Error
Amplifier
Current
Sense
PWM
Comparator
FB
BOOT
Logic
Shutdown
6
Slope
Compensation
SW
COMP
Frequency
Foldback
Reference
DAC for
Soft- Start
Maximum
Clamp
Oscillator
with PLL
8/8/ 2012 A 0192789
GND
POWERPAD
RT/ CLK
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7.3 Feature Description
7.3.1 Fixed Frequency PWM Control
The TPS54360B uses fixed-frequency, peak-current-mode control with adjustable switching frequency. The
output voltage is compared through external resistors connected to the FB pin to an internal voltage reference by
an error amplifier. An internal oscillator initiates the turnon of the high side power switch. The error amplifier
output at the COMP pin controls the high side power switch current. When the high side MOSFET switch current
reaches the threshold level set by the COMP voltage, the power switch is turned off. The COMP pin voltage
increases and decreases as the output current increases and decreases. The device implements current limiting
by clamping the COMP pin voltage to a maximum level. The pulse skipping Eco-mode is implemented with a
minimum voltage clamp on the COMP pin.
7.3.2 Slope Compensation Output Current
The TPS54360B adds a compensating ramp to the MOSFET switch current sense signal. This slope
compensation prevents sub-harmonic oscillations at duty cycles greater than 50%. The peak current limit of the
high-side switch is not affected by the slope compensation and remains constant over the full duty cycle range.
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Feature Description (continued)
7.3.3 Pulse Skip Eco-mode
The TPS54360B operates in a pulse-skipping Eco-mode at light load currents to improve efficiency by reducing
switching and gate drive losses. If the output voltage is within regulation and the peak switch current at the end
of any switching cycle is below the pulse-skipping-current threshold, the device enters Eco-mode. The pulseskipping-current threshold is the peak switch current level corresponding to a nominal COMP voltage of 600 mV.
When in Eco-mode, the COMP pin voltage is clamped at 600 mV and the high side MOSFET is inhibited. Since
the device is not switching, the output voltage begins to decay. The voltage control loop responds to the falling
output voltage by increasing the COMP pin voltage. The high side MOSFET is enabled and switching resumes
when the error amplifier lifts COMP above the pulse skipping threshold. The output voltage recovers to the
regulated value, and COMP eventually falls below the Eco-mode pulse skipping threshold at which time the
device again enters Eco-mode. The internal PLL remains operational when in Eco-mode. When operating at light
load currents in Eco-mode, the switching transitions occur synchronously with the external clock signal.
During Eco-mode operation, the TPS54360B senses and controls peak switch current, not the average load
current. Therefore the load current at which the device enters Eco-mode is dependent on the output inductor
value. The circuit in enters Eco-mode at about 24-mA output current. As the load current approaches zero, the
device enters a pulse-skip mode during which it draws only 146-μA input quiescent current.
7.3.4 Low Dropout Operation and Bootstrap Voltage (BOOT)
The TPS54360B provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT and
SW pins provides the gate-drive voltage for the high side MOSFET. The BOOT capacitor is refreshed when the
high side MOSFET is off and the external low side diode conducts. The recommended value of the BOOT
capacitor is 0.1 μF. TI recommends a ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating
of 10 V or higher for stable performance over temperature and voltage.
When operating with a low voltage difference from input to output, the high side MOSFET of the TPS54360B
operate at 100% duty cycle as long as the BOOT to SW pin voltage is greater than 2.1 V. When the voltage from
BOOT to SW drops below 2.1 V, the high side MOSFET is turned off and an integrated low-side MOSFET pulls
SW low to recharge the BOOT capacitor. To reduce the losses of the small low-side MOSFET at high output
voltages, it is disabled at 24-V output and re-enabled when the output reaches 21.5 V.
Because the gate drive current sourced from the BOOT capacitor is small, the high-side MOSFET can remain on
for many switching cycles before the MOSFET is turned off to refresh the capacitor. Thus the effective duty cycle
of the switching regulator can be high, approaching 100%. The effective duty cycle of the converter during
dropout is mainly influenced by the voltage drops across the power MOSFET, the inductor resistance, the lowside diode voltage and the printed-circuit-board resistance.
Equation 1 calculates the minimum input voltage required to regulate the output voltage and ensure normal
operation of the device. This calculation must include tolerance of the component specifications and the variation
of these specifications at their maximum operating temperature in the application.
VIN min
VOUT
VF Rdc u IOUT
0.99
RDS on u IOUT
VF
where
•
•
•
12
VF = Schottky diode forward voltage
Rdc = DC resistance of inductor and PCB
RDS(on) = High-side MOSFET RDS(on)
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Feature Description (continued)
At heavy loads, the minimum input voltage must be increased to ensure a monotonic start-up. Use Equation 2 to
calculate the minimum input voltage for this condition.
V OUT(max) = D (max) x (V IN(min) - I OUT(max) x R DS(on) + V F ) - V F + I OUT(max) x R dc
where
•
•
•
•
•
•
D(max) ≥ 0.9
IB2SW = 100 µA
tSW = 1 / fSW(MHz)
VB2SW = VBOOT + VF
VBOOT = (1.41 × VIN – 0.554 – VF / tSW – 1.847 × 103 × IB2SW) / (1.41 + 1 / tSW)*
RDS(on) = 1 / (–0.3 × VB2SW2 + 3.577 × VB2SW – 4.246)
*VBOOT is clamped by the IC. If VBOOT calculates to greater than 6 V, set VBOOT = 6 V
(2)
7.3.5 Error Amplifier
The TPS54360B voltage regulation loop is controlled by a transconductance error amplifier. The error amplifier
compares the FB pin voltage to the lower of the internal soft-start voltage or the internal 0.8-V voltage reference.
The transconductance (gm) of the error amplifier is 350 μA/V during normal operation. During soft-start operation,
the transconductance is reduced to 78 μA/V, and the error amplifier is referenced to the internal soft-start
voltage.
The frequency compensation components (capacitor, series resistor, and capacitor) are connected between the
error-amplifier-output COMP pin and GND pin.
7.3.6 Adjusting the Output Voltage
The internal voltage reference produces a precise 0.8 V ±1% voltage reference over the operating temperature
and voltage range by scaling the output of a bandgap reference circuit. The output voltage is set by a resistor
divider from the output node to the FB pin. It is recommended to use 1% tolerance or better divider resistors.
Select the low side resistor RLS for the desired divider current and use Equation 3 to calculate RHS. To improve
efficiency at light loads consider using larger value resistors. However, if the values are too high, the regulator is
more susceptible to noise and voltage errors from the FB input current may become noticeable.
æ Vout - 0.8V ö
RHS = RLS ´ ç
÷
0.8 V
è
ø
(3)
7.3.7 Enable and Adjusting Undervoltage Lockout
The TPS54360B is enabled when the VIN pin voltage rises above 4.3 V, and the EN pin voltage exceeds the
enable threshold of 1.2 V. The TPS54360B is disabled when the VIN pin voltage falls below 4 V or when the EN
pin voltage is below 1.2 V. The EN pin has an internal pullup current source, i1, of 1.2 μA that enables operation
of the TPS54360B when the EN pin floats.
If an application requires a higher undervoltage lockout (UVLO) threshold, use the circuit shown in Figure 22 to
adjust the input voltage UVLO with two external resistors. When the EN pin voltage exceeds 1.2 V, an additional
3.4 μA of hysteresis current, Ihys, is sourced out of the EN pin. When the EN pin is pulled below 1.2 V, the 3.4
μA Ihys current is removed. This addional current facilitates adjustable input voltage UVLO hysteresis. Use
Equation 4 to calculate RUVLO1 for the desired UVLO hysteresis voltage. Use Equation 5 to calculate RUVLO2 for
the desired VIN start voltage.
In applications designed to start at relatively low input voltages (for example, from 4.5 V to 9 V) and withstand
high input voltages (for example, from 40 V to 60 V), the EN pin may experience a voltage greater than the
absolute maximum voltage of 8.4 V during the high input voltage condition. It is recommended to use a zener
diode to clamp the pin voltage below the absolute maximum rating.
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Feature Description (continued)
VIN
TPS54360
i1
ihys
RUVLO1
EN
Optional
VEN
RUVLO2
Copyright © 2017, Texas Instruments Incorporated
Figure 22. Adjustable Undervoltage Lockout (UVLO)
- VSTOP
V
RUVLO1 = START
IHYS
(4)
VENA
RUVLO2 =
VSTART - VENA
+ I1
RUVLO1
(5)
7.3.8 Internal Soft Start
The TPS54360B has an internal digital soft start that ramps the reference voltage from zero volts to its final value
in 1024 switching cycles. The internal soft-start time (10% to 90%) is calculated using Equation 6.
1024
tSS (ms) =
fSW (kHz)
(6)
If the EN pin is pulled below the stop threshold of 1.2 V, switching stops and the internal soft-start resets. The
soft start also resets in thermal shutdown.
7.3.9 Constant Switching Frequency and Timing Resistor (RT/CLK) pin)
The switching frequency of the TPS54360B is adjustable over a wide range from 100 kHz to 2500 kHz by placing
a resistor between the RT/CLK pin and GND pin. The RT/CLK pin voltage is typically 0.5 V and must have a
resistor to ground to set the switching frequency. To determine the timing resistance for a given switching
frequency, use Equation 7 or Equation 8 or the curves in Figure 6 and Figure 7. To reduce the solution size one
would typically set the switching frequency as high as possible, but tradeoffs of the conversion efficiency,
maximum input voltage and minimum controllable on time should be considered. The minimum controllable ontime is typically 135 ns, which limits the maximum operating frequency in applications with high input to output
step down ratios. The maximum switching frequency is also limited by the frequency foldback circuit. A more
detailed discussion of the maximum switching frequency is provided in Accurate Current Limit Operation and
Maximum Switching Frequency.
101756
RT (kW) =
f sw (kHz)1.008
(7)
f sw (kHz) =
14
92417
RT (kW)0.991
(8)
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Feature Description (continued)
7.3.10 Accurate Current Limit Operation and Maximum Switching Frequency
The TPS543060B implements peak-current-mode control in which the COMP pin voltage controls the peak
current of the high side MOSFET. A signal proportional to the high-side switch current and the COMP pin voltage
are compared each cycle. When the peak switch current intersects the COMP control voltage, the high side
switch is turned off. During overcurrent conditions that pull the output voltage low, the error amplifier increases
switch current by driving the COMP pin high. The error amplifier output is clamped internally at a level which sets
the peak switch current limit. The TPS54360B provides an accurate current limit threshold with a typical current
limit delay of 60 ns. With smaller inductor values, the delay results in a higher peak inductor current. The
relationship between the inductor value and the peak inductor current is shown in Figure 23.
Peak Inductor Current
Inductor Current (A)
ΔCLPeak
Open Loop Current Limit
ΔCLPeak = VIN/L x tCLdelay
tCLdelay
tON
Figure 23. Current Limit Delay
To protect the converter in overload conditions at higher switching frequencies and input voltages, the
TPS54360B implements a frequency foldback. The oscillator frequency is divided by 1, 2, 4, and 8 as the FB pin
voltage falls from 0.8 V to 0 V. The TPS54360B uses a digital frequency foldback to enable synchronization to an
external clock during normal start-up and fault conditions. During short-circuit events, the inductor current can
exceed the peak current limit because of the high input voltage and the minimum controllable on time. When the
output voltage is forced low by the shorted load, the inductor current decreases slowly during the switch off time.
The frequency foldback effectively increases the off time by increasing the period of the switching cycle providing
more time for the inductor current to ramp down.
With a maximum frequency foldback ratio of 8, there is a maximum frequency at which the inductor current can
be controlled by frequency foldback protection. Equation 9 calculates the maximum switching frequency at which
the inductor current remains under control when VOUT is forced to VOUT(SC). The selected operating frequency
should not exceed the calculated value.
Equation 10 calculates the maximum switching frequency limitation set by the minimum controllable on time and
the input to output step down ratio. Setting the switching frequency above this value causes the regulator to skip
switching pulses to achieve the low duty cycle required at maximum input voltage.
æ I ´R + V
dc
OUT + Vd
´ç O
ç VIN - IO ´ RDS(on ) + Vd
è
ö
÷
÷
ø
(9)
fDIV æç ICL ´ Rdc + VOUT(sc ) + Vd
´
tON ç VIN - ICL ´ RDS(on ) + Vd
è
ö
÷
÷
ø
(10)
fSW (max skip ) =
fSW(shift) =
1
tON
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Feature Description (continued)
Where:
IO
Output current
ICL
Current limit
Rdc
inductor resistance
VIN
maximum input voltage
VOUT
output voltage
VOUTSC
output voltage during short
Vd
diode voltage drop
RDS(on)
switch on resistance
tON
controllable on time
ƒDIV
frequency divide equals (1, 2, 4, or 8)
7.3.11 Synchronization to RT/CLK pin
The RT/CLK pin can receive a frequency synchronization signal from an external system clock. To implement
this synchronization feature connect a square wave to the RT/CLK pin through either circuit network shown in
Figure 24. The square wave applied to the RT/CLK pin must switch lower than 0.5 V and higher than 1.7 V and
have a pulsewidth greater than 15 ns. The synchronization frequency range is 160 kHz to 2300 kHz. The rising
edge of the SW is synchronized to the falling edge of RT/CLK pin signal. Design the external synchronization
circuit so that the default frequency set resistor is connected from the RT/CLK pin to ground when the
synchronization signal is off. When using a low-impedance-signal source, the frequency set resistor is connected
in parallel with an AC-coupling capacitor to a termination resistor (for example, 50 Ω) as shown in Figure 24. The
two resistors in series provide the default frequency setting resistance when the signal source is turned off. The
sum of the resistance must set the switching frequency close to the external CLK frequency. TI recommends AC
coupling the synchronization signal through a 10-pF ceramic capacitor to RT/CLK pin.
The first time the RT/CLK is pulled above the PLL threshold the TPS54360B switches from the RT resistor freerunning frequency mode to the PLL synchronized mode. The internal 0.5-V voltage source is removed, and the
RT/CLK pin becomes high impedance as the PLL starts to lock onto the external signal. The switching frequency
can be higher or lower than the frequency set with the RT/CLK resistor. The device transitions from the resistor
mode to the PLL mode and locks onto the external clock frequency within 78 microseconds. During the transition
from the PLL mode to the resistor programmed mode, the switching frequency falls to 150 kHz and then
increases or decreases to the resistor programmed frequency when the 0.5-V bias voltage is reapplied to the
RT/CLK resistor.
The switching frequency is divided by 8, 4, 2, and 1 as the FB pin voltage ramps from 0 to 0.8 volts. The device
implements a digital frequency foldback to enable synchronizing to an external clock during normal start-up and
fault conditions. Figure 25, Figure 26, and Figure 27 show the device synchronized to an external system clock in
continuous conduction mode (CCM), discontinuous conduction (DCM), and pulse-skip mode (Eco-Mode).
SPACER
RT/CLK
TPS54360B
TPS54360B
RT/CLK
PLL
RT
Clock
Source
PLL
Hi-Z
Clock
Source
RT
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Figure 24. Synchronizing to a System Clock
16
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Feature Description (continued)
SW
SW
EXT
EXT
IL
IL
Figure 25. Plot of Synchronizing in CCM
Figure 26. Plot of Synchronizing in DCM
SW
EXT
IL
Figure 27. Plot of Synchronizing in Eco-Mode
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Feature Description (continued)
7.3.12 Overvoltage Protection
The TPS54360B incorporates an output overvoltage protection (OVP) circuit to minimize voltage overshoot when
recovering from output fault conditions or strong unload transients in designs with low output capacitance. For
example, when the power supply output is overloaded the error amplifier compares the actual output voltage to
the internal reference voltage. If the FB pin voltage is lower than the internal reference voltage for a considerable
time, the output of the error amplifier increases to a maximum voltage corresponding to the peak current limit
threshold. When the overload condition is removed, the regulator output rises and the error amplifier output
transitions to the normal operating level. In some applications, the power supply output voltage can increase
faster than the response of the error amplifier output resulting in an output overshoot.
The OVP feature minimizes output overshoot when using a low value output capacitor by comparing the FB pin
voltage to the rising OVP threshold, which is nominally 109% of the internal voltage reference. If the FB pin
voltage is greater than the rising OVP threshold, the high side MOSFET is immediately disabled to minimize
output overshoot. When the FB voltage drops below the falling OVP threshold which is nominally 106% of the
internal voltage reference, the high-side MOSFET resumes normal operation.
7.3.13 Thermal Shutdown
The TPS54360B provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 176°C. The high side MOSFET stops switching when the junction temperature exceeds the thermal trip
threshold. Once the die temperature falls below 164°C, the device reinitiates the power up sequence controlled
by the internal soft-start circuitry.
7.3.14 Small Signal Model for Loop Response
Figure 28 shows an equivalent model for the TPS54360B control loop which can be simulated to check the
frequency response and dynamic load response. The error amplifier is a transconductance amplifier with a gmEA
of 350 μA/V. The error amplifier can be modeled using an ideal voltage controlled current source. The resistor Ro
and capacitor Co model the open loop gain and frequency response of the amplifier. The 1-mV AC voltage
source between the nodes a and b effectively breaks the control loop for the frequency response measurements.
Plotting c/a provides the small signal response of the frequency compensation. Plotting a/b provides the small
signal response of the overall loop. The dynamic loop response can be evaluated by replacing RL with a current
source with the appropriate load step amplitude and step rate in a time domain analysis. This equivalent model is
only valid for continuous conduction mode (CCM) operation.
SW
VO
Power Stage
gmps 12 A/V
a
b
RESR
R1
RL
COMP
c
0.8 V
CO
R3
C2
RO
FB
COUT
gmea
R2
350 mA/V
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 28. Small Signal Model for Loop Response
18
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Feature Description (continued)
7.3.15 Simple Small Signal Model for Peak-Current-Mode Control
Figure 29 describes a simple small signal model that can be used to design the frequency compensation. The
TPS54360B power stage can be approximated by a voltage-controlled current source (duty cycle modulator)
supplying current to the output capacitor and load resistor. The control to output transfer function is shown in
Equation 11 and consists of a DC gain, one dominant pole, and one ESR zero. The quotient of the change in
switch current and the change in COMP pin voltage (node c in Figure 28) is the power stage transconductance,
gmPS. The gmPS for the TPS54360B is 12 A/V. The low-frequency gain of the power stage is the product of the
transconductance and the load resistance as shown in Equation 12.
As the load current increases and decreases, the low-frequency gain decreases and increases, respectively. This
variation with the load may seem problematic at first glance, but fortunately the dominant pole moves with the
load current (see Equation 13). The combined effect is highlighted by the dashed line in the right half of
Figure 29. As the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB
crossover frequency the same with varying load conditions. The type of output capacitor chosen determines
whether the ESR zero has a profound effect on the frequency compensation design. Using high ESR aluminum
electrolytic capacitors may reduce the number frequency compensation components needed to stabilize the
overall loop because the phase margin is increased by the ESR zero of the output capacitor (see Equation 14).
VO
Adc
VC
RESR
fp
RL
gmps
COUT
fz
Copyright © 2017, Texas Instruments Incorporated
Figure 29. Simple Small Signal Model and Frequency Response for Peak-Current-Mode Control
æ
s ö
ç1 +
÷
2p ´ fZ ø
VOUT
è
= Adc ´
VC
æ
s ö
ç1 +
÷
2
p
´ fP ø
è
Adc = gmps ´ RL
(11)
(12)
1
fP =
COUT ´ RL ´ 2p
(13)
1
fZ =
COUT ´ RESR ´ 2p
(14)
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Feature Description (continued)
7.3.16 Small Signal Model for Frequency Compensation
The TPS54360B uses a transconductance amplifier for the error amplifier and supports three of the commonlyused frequency compensation circuits. Compensation circuits Type 2A, Type 2B, and Type 1 are shown in
Figure 30. Type 2 circuits are typically implemented in high bandwidth power-supply designs using low ESR
output capacitors. The Type 1 circuit is used with power-supply designs with high-ESR aluminum electrolytic or
tantalum capacitors. Equation 15 and Equation 16 relate the frequency response of the amplifier to the small
signal model in Figure 30. The open-loop gain and bandwidth are modeled using the RO and CO shown in
Figure 30. See Application and Implementation for a design example using a Type 2A network with a low ESR
output capacitor.
Equation 15 through Equation 24 are provided as a reference. An alternative is to use WEBENCH software tools
to create a design based on the power supply requirements.
VO
R1
FB
gmea
Type 2A
COMP
Type 2B
Type 1
Vref
R2
RO
R3
CO
C2
R3
C2
C1
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 30. Types of Frequency Compensation
Aol
A0
P1
Z1
P2
A1
BW
Figure 31. Frequency Response of the Type 2A and Type 2B Frequency Compensation
Aol(V/V)
gmea
gmea
=
2p ´ BW (Hz)
Ro =
CO
20
(15)
(16)
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Feature Description (continued)
æ
ö
s
ç1 +
÷
2p ´ fZ1 ø
è
EA = A0 ´
æ
ö æ
ö
s
s
ç1 +
÷ ´ ç1 +
÷
2
2
p
´
p
´
f
f
P1 ø è
P2 ø
è
(17)
R2
R1 + R2
R2
´ Ro| | R3 ´
R1 + R2
A0 = gmea ´ Ro ´
A1 = gmea
P1 =
Z1 =
P2 =
(18)
(19)
1
2p ´ Ro ´ C1
(20)
1
2p ´ R3 ´ C1
(21)
1
2p ´ R3 | | RO ´ (C2 + CO )
type 2a
(22)
1
P2 =
type 2b
2p ´ R3 | | RO ´ CO
P2 =
2p ´ R O
(23)
1
type 1
´ (C2 + C O )
(24)
7.4 Device Functional Modes
7.4.1 Operation with VIN ≤ 4.5 V (Minimum VIN)
TI recommends operating the device with input voltages above 4.5 V. The typical VIN UVLO threshold is 4.3 V,
and the device may operate at input voltages down to the UVLO voltage. At input voltages below the actual
UVLO voltage, the device does not switch. If EN is externally pulled up to VIN or left floating, when VIN passes the
UVLO threshold the device become actives. Switching is enabled, and the soft start sequence is initiated. The
TPS54360B starts at the soft-start time determined by the internal soft-start time.
7.4.2 Operation with EN Control
The enable threshold voltage is 1.2 V typical. With EN held below that voltage the device is disabled and
switching is inhibited even if VIN is above its UVLO threshold. The IC quiescent current is reduced in this state. If
the EN voltage is increased above the threshold while VIN is above its UVLO threshold, the device becomes
active. Switching is enabled, and the soft start sequence is initiated. The TPS54360B starts at the soft-start time
determined by the internal soft start time.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS54360B is a 60-V, 3.5-A, step-down regulator with an integrated high-side MOSFET. Ideal applications
are: 12 V, 24 and 48 V industrial and communications power systems.
8.2 Typical Application
L1
8.2 µH
C4
5.0 V, 3.5 A
0.1 …F
U1
TPS54360BDDA
VIN
8.5 V to 60 V
C1
2.2 …F
C2
2
3
R1
523k
4
2.2 …F
R2
84.5
BOOT
SW
VIN
GND
COMP
EN
RT/CLK
PWRPD
1
D1
9
R3
162k
FB
8
B560C
C6
C7
47 …F
47 …F
7
R5
53.6k
6
5
FB
GND
C8
R4
13.0k
FB
39 pF
C5
R6
10.2k
6800 pF
GND
VOUT
GND
GND
GND
GND
Copyright © 2018, Texas Instruments Incorporated
Figure 32. 5 V Output TPS54360B Design Example
8.2.1 Design Requirements
This guide illustrates the design of a high frequency switching regulator using ceramic output capacitors. A few
parameters must be known in order to start the design process. These requirements are typically determined at
the system level. For this example, start with the following known parameters:
Table 1. Design Parameters
22
PARAMETER
VALUE
Output voltage
5V
Transient response 0.875-A to 2.625-A load step
ΔVOUT = 4%
Maximum output current
3.5 A
Input voltage
12 V nom. 8.5 V to 60 V
Output voltage ripple
0.5% of VOUT
Start input voltage (rising VIN)
8V
Stop input voltage (falling VIN)
6.25 V
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8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design with WEBENCH® Tools
Click here to create a custom design using the TPS54360B device with the WEBENCH® Power Designer.
1. Start by entering your VIN, VOUT, and IOUT requirements.
2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and
compare this design with other possible solutions from Texas Instruments.
3. The WEBENCH Power Designer provides you with a customized schematic along with a list of materials with
real time pricing and component availability.
4. In most cases, you will also be able to:
– Run electrical simulations to see important waveforms and circuit performance
– Run thermal simulations to understand the thermal performance of your board
– Export your customized schematic and layout into popular CAD formats
– Print PDF reports for the design, and share your design with colleagues
5. Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Selecting the Switching Frequency
The first step is to choose a switching frequency for the regulator. Typically, the designer uses the highest
switching frequency possible since this produces the smallest solution size. High switching frequency allows for
lower value inductors and smaller output capacitors compared to a power supply that switches at a lower
frequency. The switching frequency that can be selected is limited by the minimum on-time of the internal power
switch, the input voltage, the output voltage and the frequency foldback protection.
Equation 9 and Equation 10 should be used to calculate the upper limit of the switching frequency for the
regulator. Choose the lower value result from the two equations. Switching frequencies higher than these values
results in pulse skipping or the lack of overcurrent protection during a short circuit.
The typical minimum on time, tonmin, is 135 ns for the TPS54360B. For this example, the output voltage is 5 V
and the maximum input voltage is 60 V, which allows for a maximum switch frequency up to 710 kHz to avoid
pulse skipping from Equation 9. To ensure overcurrent runaway is not a concern during short circuits use
Equation 10 to determine the maximum switching frequency for frequency foldback protection. With a maximum
input voltage of 60 V, assuming a diode voltage of 0.7 V, inductor resistance of 25 mΩ, switch resistance of 92
mΩ, a current limit value of 4.7 A and short circuit output voltage of 0.1 V, the maximum switching frequency is
902 kHz.
For this design, a lower switching frequency of 600 kHz is chosen to operate comfortably below the calculated
maximums. To determine the timing resistance for a given switching frequency, use Equation 7 or the curve in
Figure 6. The switching frequency is set by resistor R3 shown in . For 600 kHz operation, the closest standard
value resistor is 162 kΩ.
1
æ 3.5 A x 25 mW + 5 V + 0.7 V ö
fSW(max skip) =
´ ç
÷ = 710 kHz
135ns
è 60 V - 3.5 A x 92 mW + 0.7 V ø
(25)
8
æ 4.7 A x 25 mW + 0.1 V + 0.7 V ö
´ ç
÷ = 902 kHz
135 ns
è 60 V - 4.7 A x 92 mW + 0.7 V ø
101756
RT (kW) =
= 161 kW
600 (kHz)1.008
fSW(shift) =
(26)
(27)
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8.2.2.3 Output Inductor Selection (LO)
To calculate the minimum value of the output inductor, use .
KIND is a ratio that represents the amount of inductor ripple current relative to the maximum output current. The
inductor ripple current is filtered by the output capacitor. Therefore, choosing high inductor ripple currents
impacts the selection of the output capacitor since the output capacitor must have a ripple current rating equal to
or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion of the designer,
however, the following guidelines may be used.
For designs using low ESR output capacitors such as ceramics, a value as high as KIND = 0.3 may be desirable.
When using higher ESR output capacitors, KIND = 0.2 yields better results. Since the inductor ripple current is
part of the current mode PWM control system, the inductor ripple current should always be greater than 150 mA
for stable PWM operation. In a wide input voltage regulator, it is best to choose relatively large inductor ripple
current. This provides sufficienct ripple current with the input voltage at the minimum.
For this design example, KIND = 0.3 and the minimum inductor value is calculated to be 7.3 μH. The nearest
standard value is 8.2 μH. It is important that the RMS current and saturation current ratings of the inductor not be
exceeded. The RMS and peak inductor current can be found from Equation 30 and Equation 31. For this design,
the RMS inductor current is 3.5 A and the peak inductor current is 3.97 A. The chosen inductor is a WE
7447797820, which has a saturation current rating of 5.8 A and an RMS current rating of 5.05 A.
As the equation set demonstrates, lower ripple currents reduces the output voltage ripple of the regulator but it
requires a larger value of inductance. Selecting higher ripple currents increases the output voltage ripple of the
regulator but allow for a lower inductance value.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults, or transient load conditions, the inductor current can increase above the peak inductor current level
calculated above. In transient conditions, the inductor current can increase up to the switch current limit of the
device. For this reason, the most conservative design approach is to choose an inductor with a saturation current
rating equal to or greater than the switch current limit of the TPS54360B, which is nominally 5.5 A.
LO(min ) =
VIN(max ) - VOUT
IOUT ´ KIND
´
VOUT
60 V - 5 V
5V
=
´
= 7.3 mH
VIN(max ) ´ fSW
3.5 A x 0.3
60 V ´ 600 kHz
(28)
spacer
IRIPPLE =
VOUT ´ (VIN(max ) - VOUT )
VIN(max ) ´ LO ´ fSW
=
5 V x (60 V - 5 V)
= 0.932 A
60 V x 8.2 mH x 600 kHz
(29)
spacer
IL(rms ) =
(IOUT )
2
(
æ
1 ç VOUT ´ VIN(max ) - VOUT
+
´
12 çç
VIN(max ) ´ LO ´ fSW
è
)ö÷
2
÷ =
÷
ø
2
(3.5 A )
2
æ
5 V ´ (60 V - 5 V ) ö
1
+
´ ç
÷ = 3.5 A
ç 60 V ´ 8.2 mH ´ 600 kHz ÷
12
è
ø
(30)
spacer
IL(peak ) = IOUT +
IRIPPLE
0.932 A
= 3.5 A +
= 3.97 A
2
2
(31)
8.2.2.4 Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the most stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the increased load current until the regulator responds to the load step. The regulator does not respond
immediately to a large, fast increase in the load current such as transitioning from no load to a full load. The
regulator usually needs two or more clock cycles for the control loop to sense the change in output voltage and
adjust the peak switch current in response to the higher load. The output capacitance must be large enough to
24
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supply the difference in current for 2 clock cycles to maintain the output voltage within the specified range.
Equation 32 shows the minimum output capacitance necessary, where ΔIOUT is the change in output current, ƒsw
is the regulators switching frequency and ΔVOUT is the allowable change in the output voltage. For this example,
the transient load response is specified as a 4% change in VOUT for a load step from 0.875 A to 2.625 A.
Therefore, ΔIOUT is 2.625 A – 0.875 A = 1.75 A and ΔVOUT = 0.04 × 5 = 0.2 V. Using these numbers gives a
minimum capacitance of 29.2 μF. This value does not take the ESR of the output capacitor into account in the
output voltage change. For ceramic capacitors, the ESR is usually small enough to be ignored. Aluminum
electrolytic and tantalum capacitors have higher ESR that must be included in load step calculations.
The output capacitor must also be sized to absorb energy stored in the inductor when transitioning from a high to
low load current. The catch diode of the regulator can not sink current so energy stored in the inductor can
produce an output voltage overshoot when the load current rapidly decreases. A typical load step response is
shown in Figure 33. The excess energy absorbed in the output capacitor increases the voltage on the capacitor.
The capacitor must be sized to maintain the desired output voltage during these transient periods. Equation 33
calculates the minimum capacitance required to keep the output voltage overshoot to a desired value, where LO
is the value of the inductor, IOH is the output current under heavy load, IOL is the output under light load, Vf is the
peak output voltage, and Vi is the initial voltage. For this example, the worst case load step is from 2.625 A to
0.875 A. The output voltage increases during this load transition and the stated maximum in our specification is
4 % of the output voltage. This makes Vf = 1.04 × 5 = 5.2. Vi is the initial capacitor voltage which is the nominal
output voltage of 5 V. Using these numbers in Equation 33 yields a minimum capacitance of
24.6 μF.
Equation 34 calculates the minimum output capacitance needed to meet the output voltage ripple specification,
where ƒsw is the switching frequency, VORIPPLE is the maximum allowable output voltage ripple, and IRIPPLE is the
inductor ripple current. Equation 34 yields 7.8 μF.
Equation 35 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 35 indicates the ESR should be less than 27 mΩ.
The most stringent criteria for the output capacitor is 29.2 μF required to maintain the output voltage within
regulation tolerance during a load transient.
Capacitance de-ratings for aging, temperature and DC bias increases this minimum value. For this example, 2 ×
47-μF, 10-V ceramic capacitors with 5 mΩ of ESR is used. The derated capacitance is 58.3 µF, well above the
minimum required capacitance of 29.2 µF.
Capacitors are generally rated for a maximum ripple current that can be filtered without degrading capacitor
reliability. Some capacitor data sheets specify the root mean square (RMS) value of the maximum ripple current.
Equation 36 can be used to calculate the RMS ripple current that the output capacitor must support. For this
example, Equation 36 yields 269 mA.
2 ´ DIOUT
2 ´ 1.75 A
=
= 29.2 mF
COUT >
fSW ´ DVOUT 600 kHz x 0.2 V
(32)
((I ) - (I ) ) = 8.2 mH x (2.625 A - 0.875 A ) = 24.6 mF
x
(5.2 V - 5 V )
((V ) - (V ) )
2
OH
COUT > LO
2
2
2
OL
2
f
2
2
2
I
1
1
1
1
´
=
= 7.8 mF
x
8 ´ fSW æ VORIPPLE ö 8 x 600 kHz
æ 25 mV ö
ç 0.932 A ÷
ç
÷
è
ø
è IRIPPLE ø
V
25 mV
= 27 mW
RESR < ORIPPLE =
IRIPPLE
0.932 A
(33)
COUT >
ICOUT(rms) =
(
VOUT ´ VIN(max ) - VOUT
)=
12 ´ VIN(max ) ´ LO ´ fSW
5V ´
(60 V
(34)
(35)
- 5 V)
12 ´ 60 V ´ 8.2 mH ´ 600 kHz
= 269 mA
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8.2.2.5 Catch Diode
The TPS54360B requires an external catch diode between the SW pin and GND. The selected diode must have
a reverse voltage rating equal to or greater than VIN(max). The peak current rating of the diode must be greater
than the maximum inductor current. Schottky diodes are typically a good choice for the catch diode due to their
low forward voltage. The lower the forward voltage of the diode, the higher the efficiency of the regulator.
Typically, diodes with higher voltage and current ratings have higher forward voltages. A diode with a minimum of
60-V reverse voltage is preferred to allow input voltage transients up to the rated voltage of the TPS54360B.
For the example design, the B560C-13-F Schottky diode is selected for its lower forward voltage and good
thermal characteristics compared to smaller devices. The typical forward voltage of the B560C-13-F is 0.7 volts
at 5 A.
The diode must also be selected with an appropriate power rating. The diode conducts the output current during
the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input
voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by
the forward voltage of the diode to calculate the instantaneous conduction losses of the diode. At higher
switching frequencies, the AC losses of the diode need to be taken into account. The AC losses of the diode are
due to the charging and discharging of the junction capacitance and reverse recovery charge. Equation 37 is
used to calculate the total power dissipation, including conduction losses and AC losses of the diode.
The B560C-13-F diode has a junction capacitance of 300 pF. Using Equation 37, the total loss in the diode is
2.58 Watts.
If the power supply spends a significant amount of time at light load currents or in sleep mode, consider using a
diode which has a low leakage current and slightly higher forward voltage drop.
PD =
(V
IN(max ) - VOUT
)´ I
OUT
+
VIN(max )
(60 V
2
´ Vf d
- 5 V ) ´ 3.5 A x 0.7 V
60 V
+
C j ´ fSW ´ (VIN + Vf d)
=
2
300 pF x 600 kHz x (60 V + 0.7 V)2
= 2.58 W
2
(37)
8.2.2.6 Input Capacitor
The TPS54360B requires a high quality ceramic type X5R or X7R input decoupling capacitor with at least 3 μF of
effective capacitance. Some applications benefit from additional bulk capacitance. The effective capacitance
includes any loss of capacitance due to DC bias effects. The voltage rating of the input capacitor must be greater
than the maximum input voltage. The capacitor must also have a ripple current rating greater than the maximum
input current ripple of the TPS54360B. The input ripple current can be calculated using Equation 38.
The value of a ceramic capacitor varies significantly with temperature and the DC bias applied to the capacitor.
The capacitance variations due to temperature can be minimized by selecting a dielectric material that is more
stable over temperature. X5R and X7R ceramic dielectrics are usually selected for switching regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The input capacitor
must also be selected with consideration for the DC bias. The effective value of a capacitor decreases as the DC
bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 60-V voltage rating is required to support the
maximum input voltage. Common standard ceramic capacitor voltage ratings include 4 V, 6.3 V, 10 V, 16 V, 25
V, 50 V, or 100 V. For this example, two 2.2-μF, 100-V capacitors in parallel are used. Table 2 shows several
choices of high voltage capacitors.
The input capacitance value determines the input ripple voltage of the regulator. The input voltage ripple can be
calculated using Equation 39. Using the design example values, IOUT = 3.5 A, CIN = 4.4 μF, ƒsw = 600 kHz, yields
an input voltage ripple of 331 mV and an RMS input ripple current of 1.72 A.
ICI(rms ) = IOUT x
26
VOUT
x
VIN(min )
(V
IN(min ) - VOUT
VIN(min )
) = 3.5 A
5V
´
8.5 V
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(8.5 V
- 5 V)
8.5 V
= 1.72 A
(38)
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´ 0.25
I
3.5 A ´ 0.25
DVIN = OUT
=
= 331 mV
CIN ´ fSW
4.4 mF ´ 600 kHz
(39)
Table 2. Capacitor Types
VALUE (μF)
1 to 2.2
1 to 4.7
1
1 to 2.2
1 to 1.8
1 to 1.2
1 to 3.9
1 to 1.8
1 to 2.2
1.5 to 6.8
1 to 2.2
1 to 3.3
1 to 4.7
1
1 to 4.7
1 to 2.2
EIA SIZE
VOLTAGE
DIALECTRIC
100 V
1210
GRM32 series
50 V
100 V
1206
COMMENTS
GRM31 series
50 V
50 V
2220
100 V
VJ X7R series
50 V
2225
100 V
100 V
1812
X7R
50 V
100 V
1210
50 V
C series C4532
C series C3225
50 V
1210
100 V
50 V
1812
X7R dielectric series
100 V
8.2.2.7 Bootstrap Capacitor Selection
A 0.1-μF ceramic capacitor must be connected between the BOOT and SW pins for proper operation. A ceramic
capacitor with X5R or better grade dielectric is recommended. The capacitor must have a 10 V or higher voltage
rating.
8.2.2.8 Undervoltage Lockout Set Point
The undervoltage lockout (UVLO) can be adjusted using an external voltage divider on the EN pin of the
TPS54360B. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power
down or brown outs when the input voltage is falling. For the example design, the supply should turn on and start
switching once the input voltage increases above 8 V (UVLO start). After the regulator starts switching, it should
continue to do so until the input voltage falls below 6.25 V (UVLO stop).
Programmable UVLO threshold voltages are set using the resistor divider of RUVLO1 and RUVLO2 between VIN and
ground connected to the EN pin. Equation 4 and Equation 5 calculate the resistance values necessary. For the
example application, a 523 kΩ between VIN and EN (RUVLO1) and a 84.5 kΩ between EN and ground (RUVLO2)
are required to produce the 8-V and 6.25-V start and stop voltages.
V
- VSTOP
8 V - 6.25 V
=
= 515 kW
RUVLO1 = START
IHYS
3.4 mA
(40)
RUVLO2 =
VENA
1.2 V
=
= 84.5 kW
VSTART - VENA
8 V - 1.2 V
+ 1.2 mA
+ I1
523 kW
RUVLO1
(41)
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8.2.2.9 Output Voltage and Feedback Resistors Selection
The voltage divider of R5 and R6 sets the output voltage. For the example design, 10.2 kΩ was selected for R6.
Using Equation 3, R5 is calculated as 53.5 kΩ. The nearest standard 1% resistor is 53.6 kΩ. Due to the input
current of the FB pin, the current flowing through the feedback network should be greater than 1 μA to maintain
the output voltage accuracy. This requirement is satisfied if the value of R6 is less than 800 kΩ. Choosing higher
resistor values decreases quiescent current and improves efficiency at low output currents but may also
introduce noise immunity problems.
V
- 0.8 V
æ 5 V - 0.8 V ö
= 10.2 kW x ç
RHS = RLS x OUT
÷ = 53.5 kW
0.8 V
0.8 V
è
ø
(42)
8.2.2.10 Minimum VIN
To ensure proper operation of the device and to keep the output voltage in regulation, the input voltage at the
device must be above the value calculated with Equation 43. Using the typical values for the RHS, RDC and VF in
this application example, the minimum input voltage is 5.56 V. The BOOT-SW = 3 V curve in Figure 1 was used
for RDS(on) = 0.12 Ω because the device operates with low drop out. When operating with low dropout, the BOOTSW voltage is regulated at a lower voltage because the BOOT-SW capacitor is not refreshed every switching
cycle. In the final application, the values of RDS(on), Rdc and VF used in this equation must include tolerance of the
component specifications and the variation of these specifications at their maximum operating temperature in the
application.
VIN min
VIN min
VOUT
VF Rdc u IOUT
RDS on u IOUT VF
0.99
5V 0.5V 0.0253: u 3.5A
0.12: u 3.5A 0.5V
0.99
5.56V
(43)
8.2.2.11 Compensation
There are several methods to design compensation for DC/DC regulators. The method presented here is easy to
calculate and ignores the effects of the slope compensation that is internal to the device. Because the slope
compensation is ignored, the actual crossover frequency is lower than the crossover frequency used in the
calculations. This method assumes the crossover frequency is between the modulator pole and the ESR zero
and the ESR zero is at least 10 times greater the modulator pole.
To get started, the modulator pole, ƒp(mod), and the ESR zero, ƒz1 must be calculated using Equation 44 and
Equation 45. For COUT, use a derated value of 58.3 μF. Use equations Equation 46 and Equation 47 to estimate
a starting point for the crossover frequency, ƒco. For the example design, ƒp(mod) is 1912 Hz and ƒz(mod) is 1092
kHz. Equation 45 is the geometric mean of the modulator pole and the ESR zero and Equation 47 is the mean of
modulator pole and the switching frequency. Equation 46 yields 45.7 kHz and Equation 47 gives 23.9 kHz. Use
the lower value of Equation 46 or Equation 47 for an initial crossover frequency. For this example, the target ƒco
is 23.9 kHz.
Next, the compensation components are calculated. A resistor in series with a capacitor is used to create a
compensating zero. A capacitor in parallel to these two components forms the compensating pole.
IOUT(max )
3.5 A
fP(mod) =
=
= 1912 Hz
2 ´ p ´ VOUT ´ COUT 2 ´ p ´ 5 V ´ 58.3 mF
(44)
f Z(mod) =
1
1
=
= 1092 kHz
2 ´ p ´ RESR ´ COUT
2 ´ p ´ 2.5 mW ´ 58.3 mF
fco =
fp(mod) x f z(mod) =
fco =
fp(mod) x
fSW
2
=
1912 Hz x 1092 kHz
1912 Hz x
600 kHz
2
= 45.7 kHz
= 23.9 kHz
(45)
(46)
(47)
To determine the compensation resistor, R4, use Equation 48. Assume the power stage transconductance,
gmps, is 12 A/V. The output voltage, VO, reference voltage, VREF, and amplifier transconductance, gmea, are 5
V, 0.8 V and 350 μA/V, respectively. R4 is calculated to be 13 kΩ which is a standard value. Use Equation 49 to
set the compensation zero to the modulator pole frequency. Equation 49 yields 6404 pF for compensating
capacitor C5. 6800 pF is used for this design.
28
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ö
VOUT
æ 2 ´ p ´ fco ´ COUT ö æ
ö
5V
æ 2 ´ p ´ 23.9 kHz ´ 58.3 mF ö æ
R4 = ç
÷ = ç
÷ x ç
÷ x ç 0.8 V x 350 mA / V ÷ = 13 kW
gmps
V
x
gmea
12
A
/
V
è
ø è
ø
è
ø è REF
ø
(48)
1
1
=
= 6404 pF
C5 =
2 ´ p ´ R4 x fp(mod)
2 ´ p ´ 13 kW x 1912 Hz
(49)
A compensation pole can be implemented if desired by adding capacitor C8 in parallel with the series
combination of R4 and C5. Use the larger value calculated from Equation 50 and Equation 51 for C8 to set the
compensation pole. The selected value of C8 is 39 pF for this design example.
C
x RESR
58.3 mF x 2.5 mW
=
= 11.2 pF
C8 = OUT
R4
13 kW
(50)
1
1
=
= 40.8 pF
C8 =
R4 x f sw x p
13 kW x 600 kHz x p
(51)
8.2.2.12 Discontinuous Conduction Mode and Eco-mode Boundary
With an input voltage of 12 V, the power supply enters discontinuous conduction mode when the output current
is less than 300 mA. The power supply enters Eco-mode when the output current is lower than 24 mA. The input
current draw is 270 μA with no load.
8.2.2.13 Power Dissipation Estimate
The following formulas show how to estimate the TPS54360B power dissipation under continuous conduction
mode (CCM) operation. Do not use these equations if the device is operating in discontinuous conduction mode
(DCM).
The power dissipation of the IC includes conduction loss (PCOND), switching loss (PSW), gate drive loss (PGD) and
supply current (PQ). Example calculations are shown with the 12-V typical input voltage of the design example.
PCOND =
(IOUT )2
æV
ö
5V
´ RDS(on ) ´ ç OUT ÷ = 3.5 A 2 ´ 92 mW ´
= 0.47 W
V
12
V
è IN ø
(52)
spacer
PSW = VIN ´ fSW ´ IOUT ´ trise = 12 V ´ 600 kHz ´ 3.5 A ´ 4.9 ns = 0.123 W
(53)
spacer
PGD = VIN ´ QG ´ fSW = 12 V ´ 3nC ´ 600 kHz = 0.022 W
(54)
spacer
PQ = VIN ´ IQ = 12 V ´ 146 mA = 0.0018 W
(55)
Where:
IOUT
is the output current (A).
RDS(on)
is the on-resistance of the high-side MOSFET (Ω).
VOUT
is the output voltage (V).
VIN
is the input voltage (V).
ƒsw
is the switching frequency (Hz).
trise
is the SW pin voltage rise time and can be estimated by trise = VIN x 0.16ns/V + 3.0ns.
QG
is the total gate charge of the internal MOSFET.
IQ
is the operating nonswitching supply current.
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Therefore,
PTOT = PCOND + PSW + PGD + PQ = 0.47 W + 0.123 W + 0.022 W + 0.0018 W = 0.616 W
(56)
For given TA,
TJ = TA + RTH ´ PTOT
(57)
For given TJMAX = 150°C
TA (max ) = TJ(max ) - RTH ´ PTOT
(58)
Where:
Ptot
is the total device power dissipation (W).
TA
is the ambient temperature (°C).
TJ
is the junction temperature (°C).
RTH
is the thermal resistance of the package (°C/W).
TJMAX
is maximum junction temperature (°C).
TAMAX
is maximum ambient temperature (°C).
There are additional power losses in the regulator circuit due to the inductor AC and DC losses, the catch diode
and PCB trace resistance impacting the overall efficiency of the regulator.
30
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10 V/div
1 A/div
8.2.3 Application Curves
C4: IOUT
VIN
20 mV/div
C3: VOUT ac coupled
VOUT -5 V offset
Time = 100 ms/div
Time = 5 ms/div
Figure 33. Load Transient
Figure 34. Line Transient (8 V to 40 V)
5 V/div
C3
5 V/div
100 mV/div
C4
C1: VIN
C1: VIN
2 V/div
1 V/div
C1
C2: EN
C2
C1
C2: EN
2 V/div
C3: VOUT
C3
Time = 2 ms/div
Time = 2 ms/div
Figure 35. Start-up With VIN
Figure 36. Start-up With EN
10 V/div
C4: IL
IOUT = 3.5 A
C3: VOUT ac coupled
C3
C4
C1: SW
C1
C4: IL
500 mA/div
C1
1 A/div
10 V/div
C1: SW
20 mV/div
C3: VOUT
C3
C4
20 mV/div
2 V/div
C2
C3
IOUT = 100 mA
C3: VOUT ac coupled
Time = 2 ms/div
Time = 2 ms/div
Figure 37. Output Ripple CCM
Figure 38. Output Ripple DCM
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10 V/div
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C1: SW
C1
C1
1 A/div
C4
IOUT = 3.5 A
C3: VOUT ac coupled
C3: VIN ac coupled
200 mV/div
C3
No Load
C3
C4
Time = 2 ms/div
Time = 2 ms/div
Figure 39. Output Ripple PSM
Figure 40. Input Ripple CCM
C1: SW
2 V/div
C1: SW
C1
200 mA/div
C4: IL
C4
IOUT = 100 mA
20 mV/div
C3: VIN ac coupled
C3
C4
C4: IL
C3
C3: VOUT ac coupled
VIN = 5.5 V
VOUT = 5 V
No Load
EN Floating
Time = 2 ms/div
Time = 20 ms/div
Figure 41. Input Ripple DCM
Figure 42. Low Dropout Operation
100
100
90
90
80
80
70
70
60
50
40
VOUT = 5V, fsw = 600 kHz
30
20
VOUT = 5V, fsw = 600 kHz
60
50
40
30
20
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
0
0.001
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
32
C1: SW
C4: IL
C4: IL
Efficiency - %
Efficiency - %
20 mV/div
500 mA/div
10 V/div
20 mV/div
200 mA/div
10 V/div
SNVSB93 – DECEMBER 2018
0.01
0.1
IO - Output Current - A
IO - Output Current - A
Figure 43. Efficiency vs Load Current
Figure 44. Light Load Efficiency
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100
100
90
90
80
80
70
70
Efficiency - %
Efficiency - %
www.ti.com
60
50
40
VOUT = 3.3V, fsw = 300 kHz
30
60
50
40
30
20
20
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
0.5
1.0
1.5
2.5
2.0
3.0
3.5
IO - Output Current - A
Figure 45. Efficiency vs Load Current
Figure 46. Light Load Efficiency
180
Output Voltage Deviation - %
60
Gain
0
0
-60
-20
VIN = 12V,
VOUT = 5V,
IOUT = 3.5A
10
100
Phase - degree
120
20
-120
VIN = 12V, VOUT = 5V,
fsw = 600 kHz
0.6
0.4
0.2
0
-0.2
0.4
-0.6
-0.8
-180
1000
10000
100000
1
1
0.8
Phase
40
-60
0.1
0.01
IO - Output Current - A
60
Gain - dB
0
0.001
4.0
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
-40
VOUT = 3.3V, fsw = 300 kHz
-1
0
1000000
0.5
1.0
1.5
2.0
2.5
3.0
3.5
IO - Output Current - A
Frequency - Hz
Figure 47. Overall Loop Frequency Response
Figure 48. Regulation vs Load Current
0.5
VOUT = 5V,
fsw = 600 kHz, IOUT = 3.5A
Output Voltage Deviation - %
0.4
0.3
0.2
0.1
0
-0.1
0.2
-0.3
-0.4
-0.5
0
5
10
15
20
25
30
35
40
45
50
55
60
VIN - Input Voltage - V
Figure 49. Regulation vs Input Voltage
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TPS54360B
SNVSB93 – DECEMBER 2018
www.ti.com
8.3 Other Applications
8.3.1 Inverting Power
The TPS54360B can be used to convert a positive input voltage to a negative output voltage. Idea applications
are amplifiers requiring a negative power supply. For a more detailed example, see Create an Inverting Power
Supply from a Step-Down Regulator.
VIN
+
Cin
Cboot
Lo
BOOT
VIN
Cd
GND
SW
TPS54360B
R1
GND
+
Co
FB
R2
VOUT
EN
COMP
RT/CLK
Rcomp
RT
Czero
Cpole
Copyright © 2018, Texas Instruments Incorporated
Figure 50. TPS54360B Inverting Power Supply
8.3.2 Split-Rail Power Supply
The TPS54360 can be used to convert a positive input voltage to a split rail positive and negative output voltage
by using a coupled inductor. Idea applications are amplifiers requiring a split rail positive and negative voltage
power supply. For a more detailed example see Create a Split-Rail Power Supply with a Wide Input Voltage
Buck Regulator.
34
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TPS54360B
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SNVSB93 – DECEMBER 2018
Other Applications (continued)
VIN
+
VOPOS
Cin
Cboot
+
GND
Copos
BOOT
VIN
Cd
GND
SW
TPS54360B
R1
GND
+
Coneg
FB
R2
VONEG
EN
COMP
RT/CLK
Rcomp
RT
Czero
Cpole
Copyright © 2018, Texas Instruments Incorporated
Figure 51. TPS54360B Split Rail Power Supply
9 Power Supply Recommendations
The device is designed to operate from an input voltage supply range between 4.5 V and 60 V. This input supply
must be well regulated. If the input supply is located more than a few inches from the TPS54360B converter
additional bulk capacitance may be required in addition to the ceramic bypass capacitors. A 100-μF electrolytic
capacitor is a typical choice
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www.ti.com
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. There are several signal paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade performance.
• To reduce parasitic effects, bypass the VIN pin to ground with a low ESR ceramic bypass capacitor with X5R
or X7R dielectric.
• Take care to minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the anode
of the catch diode.
• Tie the GND pin directly to the power pad under the IC and the PowerPAD.
• Connect the PowerPAD to internal PCB ground planes using multiple vias directly under the device. Route
the SW pin to the cathode of the catch diode and to the output inductor.
• Because the SW connection is the switching node, the catch diode and output inductor must be located close
to the SW pins, and the area of the PCB conductor minimized to prevent excessive capacitive coupling.
• For operation at full rated load, the top-side ground area must provide adequate heat dissipating area.
• The RT/CLK pin is sensitive to noise so place the RT resistor as close as possible to the IC and routed with
minimal lengths of trace.
• The additional external components can be placed approximately as shown.
• Acceptable performance can be attained with alternate PCB layouts; however, this layout has been shown to
produce good results and is meant as a guideline.
10.2 Layout Example
Vout
Output
Capacitor
Topside
Ground
Area
Input
Bypass
Capacitor
Vin
UVLO
Adjust
Resistors
Output
Inductor
Route Boot Capacitor
Trace on another layer to
provide wide path for
topside ground
BOOT
Catch
Diode
SW
VIN
GND
EN
COMP
RT/CLK
FB
Frequency
Set Resistor
Compensation
Network
Resistor
Divider
Thermal VIA
Signal VIA
Figure 52. PCB Layout Example
10.2.1 Estimated Circuit Area
Boxing in the components in the design of Figure 32 the estimated printed circuit board area is 1.025 in2 (661
mm2). This area does not include test points or connectors.
36
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SNVSB93 – DECEMBER 2018
11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Custom Design with WEBENCH® Tools
Click here to create a custom design using the TPS54360B device with the WEBENCH® Power Designer.
1. Start by entering your VIN, VOUT, and IOUT requirements.
2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and
compare this design with other possible solutions from Texas Instruments.
3. The WEBENCH Power Designer provides you with a customized schematic along with a list of materials with
real time pricing and component availability.
4. In most cases, you will also be able to:
– Run electrical simulations to see important waveforms and circuit performance
– Run thermal simulations to understand the thermal performance of your board
– Export your customized schematic and layout into popular CAD formats
– Print PDF reports for the design, and share your design with colleagues
5. Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
Eco-mode, PowerPAD, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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TPS54360B
SNVSB93 – DECEMBER 2018
www.ti.com
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
38
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Product Folder Links: TPS54360B
PACKAGE OPTION ADDENDUM
www.ti.com
15-Dec-2018
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
TPS54360BDDA
ACTIVE SO PowerPAD
DDA
8
75
Green (RoHS
& no Sb/Br)
CU NIPDAUAG
Level-2-260C-1 YEAR
-40 to 150
54360C
TPS54360BDDAR
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU NIPDAUAG
Level-2-260C-1 YEAR
-40 to 150
54360C
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
15-Dec-2018
OTHER QUALIFIED VERSIONS OF TPS54360B :
• Automotive: TPS54360B-Q1
NOTE: Qualified Version Definitions:
• Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
13-Dec-2018
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
TPS54360BDDAR
Package Package Pins
Type Drawing
SO
Power
PAD
DDA
8
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
2500
330.0
12.8
Pack Materials-Page 1
6.4
B0
(mm)
K0
(mm)
P1
(mm)
5.2
2.1
8.0
W
Pin1
(mm) Quadrant
12.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
13-Dec-2018
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS54360BDDAR
SO PowerPAD
DDA
8
2500
366.0
364.0
50.0
Pack Materials-Page 2
IMPORTANT NOTICE AND DISCLAIMER
TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATASHEETS), DESIGN RESOURCES (INCLUDING REFERENCE
DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS”
AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY
IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
PARTY INTELLECTUAL PROPERTY RIGHTS.
These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate
TI products for your application, (2) designing, validating and testing your application, and (3) ensuring your application meets applicable
standards, and any other safety, security, or other requirements. These resources are subject to change without notice. TI grants you
permission to use these resources only for development of an application that uses the TI products described in the resource. Other
reproduction and display of these resources is prohibited. No license is granted to any other TI intellectual property right or to any third
party intellectual property right. TI disclaims responsibility for, and you will fully indemnify TI and its representatives against, any claims,
damages, costs, losses, and liabilities arising out of your use of these resources.
TI’s products are provided subject to TI’s Terms of Sale (www.ti.com/legal/termsofsale.html) or other applicable terms available either on
ti.com or provided in conjunction with such TI products. TI’s provision of these resources does not expand or otherwise alter TI’s applicable
warranties or warranty disclaimers for TI products.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2019, Texas Instruments Incorporated
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