Texas Instruments | TPS54622 4.5-V to 17-V Input, 6-A Synchronous Step-Down SWIFT™ Converter With Hiccup Protection (Rev. F) | Datasheet | Texas Instruments TPS54622 4.5-V to 17-V Input, 6-A Synchronous Step-Down SWIFT™ Converter With Hiccup Protection (Rev. F) Datasheet

Texas Instruments TPS54622 4.5-V to 17-V Input, 6-A Synchronous Step-Down SWIFT™ Converter With Hiccup Protection (Rev. F) Datasheet
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TPS54622
SLVSA70F – MARCH 2011 – REVISED OCTOBER 2017
TPS54622 4.5-V to 17-V Input, 6-A Synchronous Step-Down SWIFT™ Converter With
Hiccup Protection
1 Features
3 Description
•
•
•
•
•
•
•
•
The TPS54622 device in thermally enhanced 3.5-mm
× 3.5-mm VQFN package is a full-featured 17-V, 6-A
synchronous step-down converter optimized for small
designs through high efficiency and integrating the
high-side and low-side MOSFETs. Further space
savings are achieved through current mode control,
which reduces component count, and by selecting a
high switching frequency, reducing the footprint of the
inductor.
1
•
•
•
•
•
Integrated 26-mΩ and 19-mΩ MOSFETs
Split Power Rail: 1.6 V to 17 V on PVIN
200-kHz to 1.6-MHz Switching Frequency
Synchronizes to External Clock
0.6V ±1% Voltage Reference Overtemperature
Hiccup Current Limit
Monotonic Start-Up Into Prebiased Outputs
–40°C to 150°C Operating Junction Temperature
Range
Adjustable Slow Start and Power Sequencing
Power Good Output Monitor for Undervoltage and
Overvoltage
Adjustable Input Undervoltage Lockout
For SWIFT™ Documentation, visit
http://www.ti.com/swift
Create a Custom Design Using the TPS54622
With the WEBENCH® Power Designer
2 Applications
•
•
•
High-Density Distributed Power Systems
High-Performance Point-of-Load Regulation
Broadband, Networking, and Optical
Communications Infrastructure
The output voltage start-up ramp is controlled by the
SS/TR pin, which allows operation as either a standalone power supply or in tracking situations. Power
sequencing is also possible by correctly configuring
the enable and the open-drain power good pins.
Cycle-by-cycle current limiting on the high-side FET
protects the device in overload situations and is
enhanced by a low-side sourcing current limit that
prevents current runaway. There is also a low-side
sinking current limit that turns off the low-side
MOSFET to prevent excessive reverse current.
Hiccup protection is triggered if the overcurrent
condition has persisted for longer than the preset
time. Thermal hiccup protection disables the device
when the die temperature exceeds the thermal
shutdown temperature and enables the part again
after the built-in thermal shutdown hiccup time.
Device Information(1)
PART NUMBER
PACKAGE
TPS54622
BODY SIZE (NOM)
VQFN (14)
3.50 mm × 3.50 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
TPS54622
VIN
VIN
BOOT
Efficiency vs Load Current
100
CBOOT
95
LO
EN
PH
90
VOUT
85
SS
RT/CLK
COMP
R3
R1
VSENSE
VIN = 12 V
75
VIN = 17 V
70
65
60
PowerPad
C1
80
R2
GND
CSS RRT
Efficiency - %
VIN = 8 V
PWRGD
55
50
0
C2
1
2
3
Output Current - A
4
5
6
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54622
SLVSA70F – MARCH 2011 – REVISED OCTOBER 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configurations and Functions .......................
Specifications.........................................................
1
1
1
2
4
5
6.1
6.2
6.3
6.4
6.5
6.6
5
5
5
6
6
8
Absolute Maximum Ratings .....................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 11
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
11
12
12
20
8
Application and Implementation ........................ 23
8.1 Application Information............................................ 23
8.2 Typical Application ................................................. 23
9 Power Supply Recommendations...................... 32
10 Layout................................................................... 32
10.1 Layout Guidelines ................................................. 32
10.2 Layout Examples................................................... 33
10.3 Estimated Circuit Area .......................................... 34
11 Device and Documentation Support ................. 35
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Device Support......................................................
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
35
35
35
35
35
36
36
12 Mechanical, Packaging, and Orderable
Information ........................................................... 36
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision E (December 2016) to Revision F
Page
•
Added top nav icon for TI Design .......................................................................................................................................... 1
•
Added links for WEBENCH on page 1 and in Application and Implementation and Device and Documentation
Support sections .................................................................................................................................................................... 1
•
Changed RθJA value from "47.2" to "40.1" ............................................................................................................................. 6
•
Changed RθJCtop value from "64.8" to "34.4" .......................................................................................................................... 6
•
Changed RθJB value from "14.4" to "11.4" ............................................................................................................................. 6
•
Changed ψJB value from "14.7" to "11.4" ............................................................................................................................... 6
•
Changed RθJCbot value from "3.2" to "1.8" .............................................................................................................................. 6
•
Added new paragraph to end of Sequencing (SS/TR)......................................................................................................... 22
Changes from Revision D (August 2016) to Revision E
Page
•
Changed Error amplifier dc gain Test Condition From: VSENSE = 0.8 V To: VSENSE = 0.6 V in the Electrical
Characteristics table ............................................................................................................................................................... 6
•
Changed From: (Vref) is 0.8 V To: (Vref) is 0.6 V in section Slow Start (SS/TR)................................................................ 15
•
Changed text From: "voltage reference of 0.8 V. Above 0.8 V,.." To: "voltage reference of 0.6 V. Above 0.6 V,.." in
the Minimum Output Voltage section.................................................................................................................................... 27
Changes from Revision C (August 2015) to Revision D
Page
•
Changed text string from "should be 0.1 μF" to "should be between 0.1 μF and 1.0 μF in section Bootstrap Voltage
(BOOT) and Low Dropout Operation." ................................................................................................................................. 21
•
Changed text string from "A 0.1 μF ceramic capacitor" to "A 0.1 μF to 1 μF ceramic capacitor" in section Bootstrap
Capacitor Selection............................................................................................................................................................... 26
2
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Changes from Revision B (January 2014) to Revision C
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section .................................................................................................. 1
•
Changed From separate RHL and RGY packages To a combined RHL and RGY package ................................................ 4
Changes from Revision A (March 2013) to Revision B
•
Page
Changed FEATURE From: Low 2 µA Shutdown Quiescent Current To: Hiccup Current Limit ............................................. 1
Changes from Original (March 2010) to Revision A
•
Page
Added PH 5ns Transient to the ABSOLUTE MAXIMUM RATINGS table.............................................................................. 5
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SLVSA70F – MARCH 2011 – REVISED OCTOBER 2017
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5 Pin Configurations and Functions
RHL Package
14-Pin VQFN With Exposed Thermal Pad
Top View
RT/CLK
1
PWRGD
14
GND 2
13 BOOT
GND 3
PVIN 4
PVIN 5
12 PH
Exposed
Thermal Pad
(15)
11 PH
10 EN
VIN 6
9 SS/TR
7
VSENSE
8
COMP
Pin Functions
PIN
I/O (1)
DESCRIPTION
NAME
NO.
BOOT
13
I
A bootstrap cap is required between BOOT and PH. The voltage on this cap carries the gate drive
voltage for the high-side MOSFET.
COMP
8
O
Error amplifier output, and input to the output switch current comparator. Connect frequency
compensation to this pin.
EN
10
I
Enable pin. Float to enable. Adjust the input undervoltage lockout with two resistors.
GND
2, 3
G
Return for control circuitry and low-side power MOSFET.
11, 12
O
The switch node.
PVIN
4, 5
P
Power input. Supplies the power switches of the power converter.
PWRGD
14
G
Power Good fault pin. Asserts low if output voltage is low due to thermal shutdown, dropout, overvoltage, EN shutdown or during slow start.
RT/CLK
1
I
Automatically selects between RT mode and CLK mode. An external timing resistor adjusts the
switching frequency of the device; In CLK mode, the device synchronizes to an external clock.
SS/TR
9
O
Slow start and tracking. An external capacitor connected to this pin sets the internal voltage reference
rise time. The voltage on this pin overrides the internal reference. It can be used for tracking and
sequencing.
VIN
6
P
Supplies the control circuitry of the power converter.
VSENSE
7
I
Inverting input of the gm error amplifier.
Exposed
Thermal
PAD
15
G
Thermal pad of the package and signal ground and it must be soldered down for proper operation.
PH
(1)
4
I = input, O = output, G = GND, P = Power
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6 Specifications
6.1 Absolute Maximum Ratings (1)
Input voltage
Output voltage
MIN
MAX
VIN
–0.3
20
PVIN
–0.3
20
EN
–0.3
6
BOOT
–0.3
27
VSENSE
–0.3
3
COMP
–0.3
3
PWRGD
–0.3
6
SS/TR
–0.3
3
RT/CLK
–0.3
6
BOOT-PH
0
7.5
PH
–1
20
PH 10-ns transient
–3
20
PH 5-ns transient
–4
20
Vdiff (GND to exposed thermal pad)
Source current
Sink current
–0.2
UNIT
V
V
0.2
V
±100
µA
PH
Current Limit
A
PH
Current Limit
A
PVIN
Current Limit
A
±200
µA
RT/CLK
COMP
–0.1
5
mA
Operating junction temperature
PWRGD
–40
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged device model (CDM), per JEDEC specification JESD22-C101 (2)
±500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
NOM
MAX
UNIT
Input voltage
VIN
4.5
17
V
Power stage input voltage
PVIN
1.6
17
V
0
6
A
–40
150
°C
Output current
Operating junction temperature, TJ
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6.4 Thermal Information
TPS54622
THERMAL METRIC (1) (2)
RHL (VQFN)
UNIT
14 PINS
RθJA
Junction-to-ambient thermal resistance
(3)
40.1
°C/W
RθJA
Junction-to-ambient thermal resistance
32
°C/W
RθJCtop
Junction-to-case (top) thermal resistance
34.4
°C/W
RθJB
Junction-to-board thermal resistance
11.4
°C/W
ψJT
Junction-to-top characterization parameter
0.5
°C/W
ψJB
Junction-to-board characterization parameter
11.4
°C/W
RθJCbot
Junction-to-case (bottom) thermal resistance
1.8
°C/W
(1)
(2)
(3)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
Power rating at a specific ambient temperature TA should be determined with a junction temperature of 150°C. This is the point where
distortion starts to substantially increase. Thermal management of the PCB should strive to keep the junction temperature at or below
150°C for best performance and long-term reliability. See the power dissipation estimate in the application section of this datasheet for
more information.
Test Board Conditions:
(a) 2.5 inches × 2.5 inches, 4 layers, thickness: 0.062 inch
(b) 2 oz. copper traces located on the top of the PCB
(c) 2 oz. copper ground planes on the 2 internal layers of and the bottom layer
(d) 4 0.010 inch thermal vias located under the device package
6.5 Electrical Characteristics
TJ = –40°C to 150°C, VIN = 4.5 V to 17 V, PVIN = 1.6 V to 17 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN AND PVIN PINS)
PVIN operating input voltage
1.6
17
V
VIN operating input voltage
4.5
17
V
VIN internal UVLO threshold
VIN rising
4
VIN internal UVLO hysteresis
4.5
150
VIN shutdown supply Current
EN = 0 V
VIN operating – non switching supply current
V
mV
2
5
μA
VSENSE = 810 mV
600
800
μA
Enable threshold
Rising
1.21
1.26
V
Enable threshold
Falling
Input current
EN = 1.1 V
1.15
μA
Hysteresis current
EN = 1.3 V
3.3
μA
ENABLE AND UVLO (EN PIN)
1.1
1.17
VOLTAGE REFERENCE
0 A ≤ IOUT ≤ 6 A
Voltage reference
0.594
0.6
0.606
V
BOOT-PH = 3 V
32
60
mΩ
BOOT-PH = 6 V
26
40
mΩ
VIN = 12 V
19
30
mΩ
MOSFET
High-side switch resistance
High-side switch resistance
(1)
Low-side switch resistance (1)
ERROR AMPLIFIER
Error amplifier transconductance (gm)
–2 μA < ICOMP < 2 μA, V(COMP) = 1 V
Error amplifier dc gain
VSENSE = 0.6 V
Error amplifier source/sink
V(COMP) = 1 V, 100 mV input
overdrive
Start switching threshold
6
1300
μMhos
3100
V/V
±110
μA
0.25
COMP to Iswitch gm
(1)
1000
16
V
A/V
Measured at pins.
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Electrical Characteristics (continued)
TJ = –40°C to 150°C, VIN = 4.5 V to 17 V, PVIN = 1.6 V to 17 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
8
11
14
A
6.5
10
15
A
3
4
CURRENT LIMIT
High-side switch current limit threshold
Low-side switch sourcing current limit
Low-side switch sinking current limit
2
Hiccup wait time
Hiccup time before re-start
A
512
Cycles
16384
Cycles
THERMAL SHUTDOWN
Thermal shutdown
160
Thermal shutdown hysteresis
175
°C
10
Thermal shutdown hiccup time
°C
16384
Cycles
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
Minimum switching frequency
Rrt = 240 kΩ (1%)
160
200
240
kHz
Switching frequency
Rrt = 100 kΩ (1%)
400
480
560
kHz
Maximum switching frequency
Rrt = 29 kΩ (1%)
1440
1600
1760
kHz
Minimum pulse width
20
RT/CLK high threshold
RT/CLK low threshold
RT/CLK falling edge to PH rising edge delay
ns
2
V
0.8
Measure at 500 kHz with RT resistor
in series
Switching frequency range (RT mode set point
and PLL mode)
V
66
200
ns
1600
kHz
145
ns
PH (PH PIN)
Minimum on-time
Measured at 90% to 90% of VIN,
25°C, IPH = 2A
Minimum off-time
BOOT-PH ≥ 3 V
94
0
ns
BOOT (BOOT PIN)
BOOT-PH UVLO
2.1
3
V
60
mV
SLOW START AND TRACKING (SS/TR PIN)
SS charge current
SS/TR to VSENSE matching
2.3
V(SS/TR) = 0.4 V
20
VSENSE falling (Fault)
μA
POWER GOOD (PWRGD PIN)
VSENSE threshold
92
% Vref
VSENSE rising (Good)
94
% Vref
VSENSE rising (Fault)
106
% Vref
VSENSE falling (Good)
104
Output high leakage
VSENSE = Vref, V(PWRGD) = 5.5 V
Output low
I(PWRGD) = 2 mA
Minimum VIN for valid output
V(PWRGD) < 0.5 V at 100 μA
Minimum SS/TR voltage for PWRGD
30
0.6
% Vref
100
nA
0.3
V
1
V
1.4
V
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6.6 Typical Characteristics
30
40
VIN = 12 V
RDS(on) − On Resistance − mW
RDS(on) − On Resistance − mW
VIN = 12 V
35
30
25
20
−50
−25
0
25
50
75
100
125
27
24
21
18
15
−50
150
−25
Figure 1. High-Side RDS(on) vs Temperature
50
75
100
125
150
Figure 2. Low-Side RDS(on) vs Temperature
485
fO − Oscillator Frequency − kHz
0.606
Vref − Voltage Reference − V
25
TJ − Junction Temperature - ° C
TJ − Junction Temperature − °C
0.604
0.602
0.600
0.598
0.596
0.594
−50
−25
0
25
50
75
100
125
480
475
470
RT = 100 kΩ
465
−50
150
−25
0
25
50
75
100
125
150
TJ − Junction Temperature − °C
Figure 3. Voltage Reference vs Temperature
Figure 4. Oscillator Frequency vs Temperature
N
μ
Isd – Shutdown Quiescent Current – mA
TJ − Junction Temperature − °C
Figure 5. Shutdown Quiescent Current vs Input Voltage
8
0
Figure 6. EN Pin Hysteresis Current vs Temperature
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Typical Characteristics (continued)
1.220
μ
En Pin UVLO Threshold − V
VIN = 12 V
1.215
1.210
1.205
1.200
−50
−25
°C
25
50
75
100
125
150
TJ − Junction Temperature − °C
Figure 8. Pin UVLO Threshold vs Temperature
Figure 7. Pin Pullup Current vs Temperature
2.5
ISS − Slow Start Charge Current − mA
Non-Switching Operating Quiescent Current − mA
0
800
TJ = −40°C
700
TJ = −25°C
TJ = 150°C
600
500
2.4
2.3
2.2
2.1
−50
400
3
6
9
12
−25
15
0
25
50
75
100
125
150
TJ − Junction Temperature − °C
VI − Input Voltage − V
Figure 10. Slow Start Charge Current vs Temperature
Figure 9. Non-Switching Operating Quiescent Current (VIN)
vs Input Voltage
120
ISS − SS Charge Current − μA
Voff − SS/TR to Vsense Offset − V
0.040
0.030
0.020
0.010
−50
−25
0
25
50
75
100
125
150
Vin = 12 V
110
VSENSE Rising
VSENSE Falling
100
VSENSE Rising
90
VSENSE Falling
80
−50
−25
0
25
50
75
100
125
150
TJ − Junction Temperature − °C
TJ − Junction Temperature − °C
Figure 11. (SS/TR - VSENSE) Offset vs Temperature
Figure 12. PWRGD Threshold vs Temperature
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Typical Characteristics (continued)
140
Minimum Controllable On Time − ns
IcI − Current Limit Threshold − A
13
12
11
10
TJ = −40°C
9
TJ = 25°C
TJ = 150°C
8
7
6
5
1
5
9
13
130
120
110
100
90
VIN = 12 V
80
−50
17
VI − Input Voltage − V
−25
0
25
50
75
100
125
150
TJ − Junction Temperature − °C
Figure 13. High-Side Current Limit Threshold vs Input
Voltage
Figure 14. Minimum Controllable On-Time vs Temperature
6.0
5.0
RT = 100 kΩ
VIN = 12 V
BOOT-PH UVLO Threshold – V
7.0
4.0
°C
Figure 15. Minimum Controllable Duty Ratio vs Junction
Temperature
10
Figure 16. BOOT-PH UVLO Threshold vs Temperature
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7 Detailed Description
7.1 Overview
The TPS54622 device is a 17-V, 6-A, synchronous step-down (buck) converter with two integrated N-channel
MOSFETs. To improve performance during line and load transients the device implements a constant frequency,
peak current mode control which also simplifies external frequency compensation. The wide switching frequency
of 200 kHz to 1600 kHz allows for efficiency and size optimization when selecting the output filter components.
The switching frequency is adjusted using a resistor to ground on the RT/CLK pin. The device also has an
internal phase lock loop (PLL) controlled by the RT/CLK pin that can be used to synchronize the switching cycle
to the falling edge of an external system clock.
The device has been designed for safe monotonic start-up into prebiased loads. The default start-up is when VIN
is typically 4 V. The EN pin has an internal pullup current source that can be used to adjust the input voltage
undervoltage lockout (UVLO) with two external resistors. In addition, the EN pin can be floating for the device to
operate with the internal pullup current. The total operating current for the device is approximately 600 μA when
not switching and under no load. When the device is disabled, the supply current is typically less than 2 μA.
The integrated MOSFETs allow for high-efficiency power supply designs with continuous output currents up to 6
amperes. The MOSFETs have been sized to optimize efficiency for lower duty cycle applications.
The device reduces the external component count by integrating the boot recharge circuit. The bias voltage for
the integrated high-side MOSFET is supplied by a capacitor between the BOOT and PH pins. The boot capacitor
voltage is monitored by a BOOT to PH UVLO (BOOT-PH UVLO) circuit allowing PH pin to be pulled low to
recharge the boot capacitor. The device can operate at 100% duty cycle as long as the boot capacitor voltage is
higher than the preset BOOT-PH UVLO threshold which is typically 2.1 V. The output voltage can be stepped
down to as low as the 0.6-V voltage reference (Vref).
The device has a power good comparator (PWRGD) with hysteresis which monitors the output voltage through
the VSENSE pin. The PWRGD pin is an open-drain MOSFET which is pulled low when the VSENSE pin voltage
is less than 92% or greater than 106% of the reference voltage Vref and asserts high when the VSENSE pin
voltage is 94% to 104% of the Vref.
The SS/TR (slow start/tracking) pin is used to minimize inrush currents or provide power supply sequencing
during power up. A small value capacitor or resistor divider should be coupled to the pin for slow start or critical
power supply sequencing requirements.
The device is protected from output overvoltage, overload, and thermal fault conditions. The device minimizes
excessive output overvoltage transients by taking advantage of the overvoltage circuit power good comparator.
When the overvoltage comparator is activated, the high-side MOSFET is turned off and prevented from turning
on until the VSENSE pin voltage is lower than 104% of the Vref. The device implements both high-side MOSFET
overload protection and bidirectional low-side MOSFET overload protections which help control the inductor
current and avoid current runaway. The device also shuts down if the junction temperature is higher than thermal
shutdown trip point. The device is restarted under control of the slow start circuit automatically after the built-in
thermal shutdown hiccup time.
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7.2 Functional Block Diagram
PWRGD EN
VIN
UV
i1
Logic
iHYS
Thermal
Shutdown
UVLO
Enable
Comparator
OV
Shutdown
Logic
Voltage
Reference
VSENSE
PVIN
Enable
Threshold
Hiccup
Shutdown
Boot
Charge
Boot
UVLO
+
+
SS
Shutdown
BOOT
OV
Minimum
COMP Clamp
Logic
HS FET
Current
Comparator
COMP
Dead Time
Logic and
PWM Latch
PH
Slope
Compensation
Overload
Recovery
Maximum
Clamp
OSC with
PLL
Hiccup
Shutdown
RT/CLK
VIN
LS FET
Current
Limit
Regulator
GND
Copyright © 2016, Texas Instruments Incorporated
7.3 Feature Description
7.3.1 Fixed-Frequency PWM Control
The device uses a adjustable fixed-frequency, peak current mode control. The output voltage is compared
through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives
the COMP pin. An internal oscillator initiates the turnon of the high-side power switch. The error amplifier output
is converted into a current reference which compares to the high-side power switch current. When the power
switch current reaches current reference generated by the COMP voltage level the high-side power switch is
turned off and the low-side power switch is turned on.
7.3.2 Continuous Current Mode Operation (CCM)
As a synchronous buck converter, the device normally works in continuous conduction mode (CCM) under all
load conditions.
7.3.3 VIN and Power VIN Pins (VIN and PVIN)
The device allows for a variety of applications by using the VIN and PVIN pins together or separately. The VIN
pin voltage supplies the internal control circuits of the device. The PVIN pin voltage provides the input voltage to
the power converter system.
If tied together, the input voltage for VIN and PVIN can range from 4.5 V to 17 V. If using the VIN separately from
PVIN, the VIN pin must be from 4.5 V to 17 V, and the PVIN pin can range from as low as 1.6 V to 17 V. A
voltage divider connected to the EN pin can adjust the either input voltage UVLO appropriately. Adjusting the
input voltage UVLO on the PVIN pin helps to provide consistent power-up behavior.
12
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Feature Description (continued)
7.3.4 Voltage Reference
The voltage reference system produces a precise ±1% voltage reference overtemperature by scaling the output
of a temperature stable bandgap circuit.
7.3.5 Adjusting the Output Voltage
The output voltage is set with a resistor-divider from the output (VOUT) to the VSENSE pin. TI recommends
using 1% tolerance or better divider resistors. Referring to the application schematic of Figure 29, start with a 10
kΩ for R6 and use Equation 1 to calculate R5. To improve efficiency at light loads, consider using larger value
resistors. If the values are too high, the regulator is more susceptible to noise and voltage errors from the
VSENSE input current and are noticeable.
Vo - Vref
R5 =
R6
Vref
where
•
Vref = 0.6 V
(1)
The minimum output voltage and maximum output voltage can be limited by the minimum on-time of the highside MOSFET and bootstrap voltage (BOOT-PH voltage) respectively. More discussions are located in Minimum
Output Voltage and Bootstrap Voltage (BOOT) and Low Dropout Operation.
7.3.6 Safe Start-Up Into Prebiased Outputs
The device has been designed to prevent the low-side MOSFET from discharging a prebiased output. During
monotonic prebiased startup, the low-side MOSFET is not allowed to sink current until the SS/TR pin voltage is
higher than 1.4 V.
7.3.7 Error Amplifier
The device uses a transconductance error amplifier. The error amplifier compares the VSENSE pin voltage to the
lower of the SS/TR pin voltage or the internal 0.6-V voltage reference. The transconductance of the error
amplifier is 1300 μA/V during normal operation. The frequency compensation network is connected between the
COMP pin and ground.
7.3.8 Slope Compensation
The device adds a compensating ramp to the switch current signal. This slope compensation prevents subharmonic oscillations. The available peak inductor current remains constant over the full duty cycle range.
7.3.9 Enable and Adjusting Undervoltage Lockout
The EN pin provides electrical on/off control of the device. Once the EN pin voltage exceeds the threshold
voltage, the device starts operation. If the EN pin voltage is pulled below the threshold voltage, the regulator
stops switching and enters low Iq state.
The EN pin has an internal pullup current source, allowing the user to float the EN pin for enabling the device. If
an application requires controlling the EN pin, use open-drain or open collector output logic to interface with the
pin.
The device implements internal UVLO circuitry on the VIN pin. The device is disabled when the VIN pin voltage
falls below the internal VIN UVLO threshold. The internal VIN UVLO threshold has a hysteresis of 150 mV.
If an application requires either a higher UVLO threshold on the VIN pin or a secondary UVLO on the PVIN, in
split rail applications, then the EN pin can be configured as shown in Figure 17, Figure 18, and Figure 19. When
using the external UVLO function, TI recommends setting the hysteresis to be greater than 500 mV.
The EN pin has a small pullup current Ip which sets the default state of the pin to enable when no external
components are connected. The pullup current is also used to control the voltage hysteresis for the UVLO
function since it increases by Ih once the EN pin crosses the enable threshold. The UVLO thresholds can be
calculated using Equation 2 and Equation 3.
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Feature Description (continued)
VIN
IP
IH
R1
EN
+
R2
Copyright © 2016, Texas Instruments Incorporated
Figure 17. Adjustable VIN Undervoltage Lockout
PVIN
IP
IH
R1
EN
+
R2
Copyright © 2016, Texas Instruments Incorporated
Figure 18. Adjustable PVIN Undervoltage Lockout, VIN ≥ 4.5 V
14
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Feature Description (continued)
VIN
IP
IH
R1
EN
+
R2
Copyright © 2016, Texas Instruments Incorporated
Figure 19. Adjustable VIN and PVIN Undervoltage Lockout
æV
ö
VSTART ç ENFALLING ÷ - VSTOP
è VENRISING ø
R1 =
æ V
ö
Ip ç1 - ENFALLING ÷ + Ih
V
ENRISING ø
è
R2 =
VSTOP
(2)
R1´ VENFALLING
- VENFALLING + R1(Ip + Ih )
where
•
•
•
•
Ih = 3.4 μA
Ip = 1.15 μA
VENRISING = 1.21 V
VENFALLING = 1.17 V
(3)
7.3.10 Adjustable Switching Frequency and Synchronization (RT/CLK)
The RT/CLK pin can be used to set the switching frequency of the device in two modes.
In RT mode, a resistor (RT resistor) is connected between the RT/CLK pin and GND. The switching frequency of
the device is adjustable from 200 kHz to 1600 kHz by placing a maximum of 240 kΩ and minimum of 29 kΩ
respectively. In CLK mode, an external clock is connected directly to the RT/CLK pin. The device is synchronized
to the external clock frequency with PLL.
The CLK mode overrides the RT mode. The device is able to detect the proper mode automatically and switch
from the RT mode to CLK mode.
7.3.11 Slow Start (SS/TR)
The device uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as the reference
voltage and regulates the output accordingly. A capacitor on the SS/TR pin to ground implements a slow start
time. The device has an internal pullup current source of 2.3 μA that charges the external slow-start capacitor.
The calculations for the slow start time (tSS, 10% to 90%) and slow-start capacitor (Css) are shown in Equation 4.
The voltage reference (Vref) is 0.6 V and the slow start charge current (Iss) is 2.3 μA.
Css (nF) ´ Vref (V)
t SS (ms) =
Iss (µA)
(4)
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Feature Description (continued)
When the input UVLO is triggered, the EN pin is pulled below 1.21 V, or a thermal shutdown event occurs the
device stops switching and enters low current operation. At the subsequent power up, when the shutdown
condition is removed, the device does not start switching until it has discharged its SS/TR pin to ground ensuring
proper soft start behavior.
7.3.12 Power Good (PWRGD)
The PWRGD pin is an open-drain output. Once the VSENSE pin is between 94% and 104% of the internal
voltage reference the PWRGD pin pulldown is deasserted and the pin floats. TI recommends using a pullup
resistor from the values of 10 kΩ to 100 kΩ to a voltage source that is 5.5 V or less. The PWRGD is in a defined
state once the VIN input voltage is greater than 1 V but with reduced current sinking capability. The PWRGD
achieves full current sinking capability once the VIN input voltage is above 4.5 V.
The PWRGD pin is pulled low when VSENSE is lower than 92% or greater than 106% of the nominal internal
reference voltage. Also, the PWRGD is pulled low, if the input UVLO or thermal shutdown are asserted, the EN
pin is pulled low or the SS/TR pin is below 1.4 V.
7.3.13 Output Overvoltage Protection (OVP)
The device incorporates an output overvoltage protection (OVP) circuit to minimize output voltage overshoot. For
example, when the power supply output is overloaded the error amplifier compares the actual output voltage to
the internal reference voltage. If the VSENSE pin voltage is lower than the internal reference voltage for a
considerable time, the output of the error amplifier demands maximum output current. Once the condition is
removed, the regulator output rises and the error amplifier output transitions to the steady state voltage. In some
applications with small output capacitance, the power supply output voltage can respond faster than the error
amplifier. This leads to the possibility of an output overshoot. The OVP feature minimizes the overshoot by
comparing the VSENSE pin voltage to the OVP threshold. If the VSENSE pin voltage is greater than the OVP
threshold the high-side MOSFET is turned off preventing current from flowing to the output and minimizing output
overshoot. When the VSENSE voltage drops lower than the OVP threshold, the high-side MOSFET is allowed to
turn on at the next clock cycle.
7.3.14 Overcurrent Protection
The device is protected from overcurrent conditions by cycle-by-cycle current limiting on both the high-side
MOSFET and the low-side MOSFET.
7.3.14.1 High-Side MOSFET Overcurrent Protection
The device implements current mode control which uses the COMP pin voltage to control the turnoff of the highside MOSFET and the turnon of the low-side MOSFET on a cycle-by-cycle basis. Each cycle the switch current
and the current reference generated by the COMP pin voltage are compared, when the peak switch current
intersects the current reference the high-side switch is turned off.
7.3.14.2 Low-Side MOSFET Overcurrent Protection
While the low-side MOSFET is turned on its conduction current is monitored by the internal circuitry. During
normal operation the low-side MOSFET sources current to the load. At the end of every clock cycle, the low-side
MOSFET sourcing current is compared to the internally set low-side sourcing current limit. If the low-side
sourcing current is exceeded the high-side MOSFET is not turned on and the low-side MOSFET stays on for the
next cycle. The high-side MOSFET is turned on again when the low-side current is below the low-side sourcing
current limit at the start of a cycle.
The low-side MOSFET may also sink current from the load. If the low-side sinking current limit is exceeded the
low-side MOSFET is turned off immediately for the rest of that clock cycle. In this scenario both MOSFETs are
off until the start of the next cycle.
Furthermore, if an output overload condition (as measured by the COMP pin voltage) has lasted for more than
the hiccup wait time which is programmed for 512 switching cycles, the device will shut down itself and restart
after the hiccup time of 16384 cycles. The hiccup mode helps to reduce the device power dissipation under
severe overcurrent conditions.
16
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Feature Description (continued)
7.3.15 Thermal Shutdown
The internal thermal shutdown circuitry forces the device to stop switching if the junction temperature exceeds
175°C typically. Once the junction temperature drops below 165°C typically, the internal thermal hiccup timer will
start to count. The device reinitiates the power-up sequence after the built-in thermal shutdown hiccup time
(16384 cycles) is over.
7.3.16 Small Signal Model for Loop Response
Figure 20 shows an equivalent model for the device control loop which can be modeled in a circuit simulation
program to check frequency response and transient responses. The error amplifier is a transconductance
amplifier with a gm of 1300 μA/V. The error amplifier can be modeled using an ideal voltage controlled current
source. The resistor ROUT(ea) (2.38 MΩ) and capacitor COUT(ea) (20.7 pF) model the open loop gain and frequency
response of the error amplifier. The 1-mV AC voltage source between the nodes a and b effectively breaks the
control loop for the frequency response measurements. Plotting a/c and c/b show the small signal responses of
the power stage and frequency compensation respectively. Plotting a/b shows the small signal response of the
overall loop. The dynamic loop response can be checked by replacing the RL with a current source with the
appropriate load step amplitude and step rate in a time domain analysis.
PH
VOUT
Power Stage
16 A/V
a
RESR
b
R1
VSENSE
COMP
RLOAD
COUT
c
+
C2
R3
COUT(ea)
ROUT(ea)
0.6 V
gM
1300 µA/V
R2
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 20. Small Signal Model for Loop Response
7.3.17 Simple Small Signal Model for Peak Current Mode Control
Figure 21 is a simple small signal model that can be used to understand how to design the frequency
compensation. The device power stage can be approximated to a voltage controlled current source (duty cycle
modulator) supplying current to the output capacitor and load resistor. The control to output transfer function is
shown in Equation 5 and consists of a DC gain, one dominant pole and one ESR zero. The quotient of the
change in switch current and the change in COMP pin voltage (node c in Figure 20) is the power stage
transconductance (gmps) which is 16 A/V for the device. The DC gain of the power stage is the product of gmps
and the load resistance ®L) as shown in Equation 6 with resistive loads. As the load current increases, the DC
gain decreases. This variation with load may seem problematic at first glance, but fortunately the dominant pole
moves with load current (see Equation 7). The combined effect is highlighted by the dashed line in Figure 22. As
the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB crossover
frequency the same for the varying load conditions which makes it easier to design the frequency compensation.
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Feature Description (continued)
VC
RESR
RLOAD
gm(ps)
COUT
Copyright © 2016, Texas Instruments Incorporated
Figure 21. Simplified Small Signal Model for Peak Current Mode Control
Gain
Adc
fZ
fP
Frequency
Figure 22. Simplified Frequency Response for Peak Current Mode Control
æ
ç1+
2p
VOUT
= Adc ´ è
VC
æ
ç1+
è 2p
ö
s
÷
´ ¦z ø
ö
s
÷
´ ¦p ø
(5)
Adc = gmps ´ RL
(6)
1
¦p =
C O ´ R L ´ 2p
(7)
¦z =
1
CO ´ RESR ´ 2p
(8)
where
gmps is the power stage gain (16 A/V).
RL is the load resistance.
CO is the output capacitance.
RESR is the equivalent series resistance of the output capacitor.
7.3.18 Small Signal Model for Frequency Compensation
The device uses a transconductance amplifier for the error amplifier and readily supports two of the commonly
used Type II compensation circuits and a Type III frequency compensation circuit, as shown in Figure 23. In
Type 2A, one additional high-frequency pole, C6, is added to attenuate high frequency noise. In Type III, one
additional capacitor, C11, is added to provide a phase boost at the crossover frequency. See Designing Type III
Compensation for Current Mode Step-Down Converters for a complete explanation of Type III compensation.
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Feature Description (continued)
The design guidelines below are provided for advanced users who prefer to compensate using the general
method. The following equations only apply to designs whose ESR zero is above the bandwidth of the control
loop. This is usually true with ceramic output capacitors. See Application and Implementation for a step-by-step
design procedure using higher ESR output capacitors with lower ESR zero frequencies.
VOUT
VSENSE
COMP
R8
C11
gM(ea)
Type III
+
R4
C6
C4
Type IIB
VREF
R9
R4
C4
COUT(ea)
ROUT(ea)
Type IIA
Copyright © 2016, Texas Instruments Incorporated
Figure 23. Types of Frequency Compensation
The general design guidelines for device loop compensation are as follows:
1. Determine the crossover frequency, fc. A good starting point is 1/10th of the switching frequency, fsw.
2. R4 can be determined by:
2p ´ ¦ c ´ VOUT ´ Co
R4 =
gmea ´ Vref ´ gmps
where
•
•
•
gmea is the GM amplifier gain (1300 μA/V).
gmps is the power stage gain (16 A/V).
Vref is the reference voltage (0.6 V).
(9)
æ
ö
1
ç ¦p =
÷
CO ´ RL ´ 2p ø
3. Place a compensation zero at the dominant pole: è
C4 can be determined by:
R ´ Co
C4 = L
R4
(10)
4. C6 is optional. It can be used to cancel the zero from the equivalent series resistance (ESR) of the output
capacitor CO.
´ Co
R
C6 = ESR
R4
(11)
5. Type III compensation can be implemented with the addition of one capacitor, C11. This allows for slightly
higher loop bandwidths and higher phase margins. If used, C11 is calculated from Equation 12.
1
C11 =
(2 × p × R8 × fc )
(12)
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7.4 Device Functional Modes
7.4.1 Adjustable Switching Frequency (RT Mode)
To determine the RT resistance for a given switching frequency, use Equation 13 or the curve in Figure 24. To
reduce the solution size one would set the switching frequency as high as possible, but tradeoffs of the supply
efficiency and minimum controllable on-time should be considered.
- 0.997
Rrt(k W ) = 48000 × Fsw (kHz )
-2
(13)
RT − Resistance − kW
250
200
150
100
50
0
200
400
600
800
1000
1200
1400
1600
Fsw − Oscillator Frequency − kHz
Figure 24. RT Set Resistor vs Switching Frequency
7.4.2 Synchronization (CLK Mode)
An internal phase locked loop (PLL) has been implemented to allow synchronization from 200 kHz to 1600 kHz,
and to easily switch from RT mode to CLK mode.
To implement the synchronization feature, connect a square wave clock signal to the RT/CLK pin with a duty
cycle from 20% to 80%. The clock signal amplitude must transition lower than 0.8 V and higher than 2 V. The
start of the switching cycle is synchronized to the falling edge of RT/CLK pin.
In applications where both RT mode and CLK mode are needed, the device can be configured as shown in
Figure 25. Before the external clock is present, the device works in RT mode and the switching frequency is set
by RT resistor. When the external clock is present, the CLK mode overrides the RT mode. The first time the
SYNC pin is pulled above the RT/CLK high threshold (2 V), the device switches from the RT mode to the CLK
mode and the RT/CLK pin becomes high impedance as the PLL starts to lock onto the frequency of the external
clock. TI does not recommended switching from the CLK mode back to the RT mode because the internal
switching frequency drops to 100 kHz first before returning to the switching frequency set by RT resistor.
RT/CLK Mode Select
RT/CLK
RRT
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Figure 25. Works With Both RT Mode and CLK Mode
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Device Functional Modes (continued)
7.4.3 Bootstrap Voltage (BOOT) and Low Dropout Operation
The device has an integrated boot regulator, and requires a small ceramic capacitor between the BOOT and PH
pins to provide the gate drive voltage for the high-side MOSFET. The boot capacitor is charged when the BOOT
pin voltage is less than VIN and BOOT-PH voltage is below regulation. The value of this ceramic capacitor
should be between 0.1 μF and 1 μF. TI recommends a ceramic capacitor with an X7R or X5R grade dielectric
with a voltage rating of 10 V or higher because of the stable characteristics over temperature and voltage.
To improve drop out, the device is designed to operate at 100% duty cycle as long as the BOOT to PH pin
voltage is greater than the BOOT-PH UVLO threshold which is typically 2.1 V. When the voltage between BOOT
and PH drops below the BOOT-PH UVLO threshold the high-side MOSFET is turned off and the low-side
MOSFET is turned on allowing the boot capacitor to be recharged. In applications with split input voltage rails
100% duty cycle operation can be achieved as long as (VIN – PVIN) > 4 V.
7.4.4 Sequencing (SS/TR)
Many of the common power supply sequencing methods can be implemented using the SS/TR, EN and PWRGD
pins.
The sequential method is illustrated in Figure 26 using two TPS54622 devices. The power good of the first
device is coupled to the EN pin of the second device which enables the second power supply once the primary
supply reaches regulation.
TPS54622
TPS54622
PWRGD
EN
EN
SS/TR
SS/TR
PWRGD
Copyright © 2016, Texas Instruments Incorporated
Figure 26. Sequential Start-Up Sequence
Figure 27 shows the method implementing ratiometric sequencing by connecting the SS/TR pins of two devices
together. The regulator outputs ramp up and reach regulation at the same time. When calculating the slow start
time the pullup current source must be doubled in Equation 4.
TPS54622
EN
CSS
TPS54622
EN
SS
SS
PWRGD
PWRGD
Copyright © 2016, Texas Instruments Incorporated
Figure 27. Ratiometric Start-Up Sequence
Ratiometric and simultaneous power supply sequencing can be implemented by connecting the resistor network
of R1 and R2 shown in Figure 28 to the output of the power supply that needs to be tracked or another voltage
reference source. Using Equation 14 and Equation 15, the tracking resistors can be calculated to initiate the
Vout2 slightly before, after or at the same time as Vout1. Equation 16 is the voltage difference between Vout1
and Vout2.
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Device Functional Modes (continued)
To design a ratiometric start-up in which the Vout2 voltage is slightly greater than the Vout1 voltage when Vout2
reaches regulation, use a negative number in Equation 14 and Equation 15 for deltaV. Equation 16 results in a
positive number for applications where the Vout2 is slightly lower than Vout1 when Vout2 regulation is
achieved. .
The deltaV variable is zero volt for simultaneous sequencing. To minimize the effect of the inherent SS/TR to
VSENSE offset (Vssoffset, 29 mV) in the slow start circuit and the offset created by the pullup current source
(Iss, 2.3 μA) and tracking resistors, the Vssoffset and Iss are included as variables in the equations.
To ensure proper operation of the device, the calculated R1 value from Equation 14 must be greater than the
value calculated in Equation 17.
R1 =
Vout2 + D V
Vssoffset
´
Vref
Iss
(14)
Vref ´ R1
R2 =
Vout2 + DV - Vref
DV = Vout1 - Vout2
R1 > 2800 ´ Vout1- 180 ´ DV
(15)
(16)
(17)
TPS54622
VOUT(1)
EN
SS
CSS
PWRGD
TPS54622
VOUT(2)
EN
R1
R3
SS
R2
R4
PWRGD
Copyright © 2016, Texas Instruments Incorporated
Figure 28. Ratiometric and Simultaneous Start-Up Sequence
There are two final considerations when using a resistor divider to the SS/TR pin for simultaneous start-up. First,
as described in Power Good (PWRGD), for the PWRGD output to be active the SS/TR voltage must be above
1.4 V. The external divider may prevent the SS/TR voltage from charging above the threshold. For the SS/TR pin
to charge above the threshold, an external MOSFET may be needed to disconnect the resistor divider or modify
the resistor divider ratio after start-up is complete. The PWRGD pin of the VOUT(1) converter could be used to turn
on or turn off the external MOSFET. Second, a pre-bias on VOUT(1) may prevent VOUT(2) from turning on. When
the TPS54622 is enabled, an internal 700-Ω MOSFET at the SS/TR pin turns on to discharge the SS/TR voltage
as described in Slow Start (SS/TR). The SS/TR pin voltage must discharge below 20 mV before the TPS54622
starts up. If the upper resistor at the SS/TR pin is too small, the SS/TR pin does not discharge below the
threshold, and VOUT(2) does not ramp up. The upper resistor in the SS/TR divider may need to be increased to
allow the SS/TR pin to discharge below the threshold.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS54622 device is a highly integrated synchronous step-down DC-DC converter. This device is used to
convert a higher DC input voltage to a lower DC output voltage, with a maximum output current of 6 A.
8.2 Typical Application
The application schematic of Figure 29 was developed to meet the requirements above. This circuit is available
as the TPS54622EVM-012 evaluation module. The design procedure is given in this section. For more
information about Type II and Type III frequency compensation circuits, see Designing Type III Compensation for
Current Mode Step-Down Converters and design calculator (SLVC219).
U1
TPS54622RHL
1
2
3
VIN = 8-17V
4
5
6
C1
10 PF
VSNS
R1
35.7k
7
RT/CLK
GND
GND
PVIN
PVIN
VIN
VSNS
PWPD
R3
100k
15
EN
C3
0.1 PF
PWRGD 14
13
BOOT
12
PH
11
PH
10
EN
9
SS/TR
8
COMP
L1
3.3 H
VOUT
EN
R5
10k
C7
100 PF
C8
Optional
VSNS
R3
3.74k
R7
2.21k
C2
4.7 PF
R2
8.06k
VOUT = 3.3V, 6A
C5
47pF
C4
0.01 PF
C6
0.022 PF
Copyright © 2016, Texas Instruments Incorporated
Figure 29. Typical Application Circuit
8.2.1 Design Requirements
This example details the design of a high-frequency switching regulator design using ceramic output capacitors.
A few parameters must be known to start the design process. These parameters are typically determined at the
system level. For this example, begin with the known parameters listed in Table 1.
Table 1. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Output voltage
3.3 V
Output current
6A
Transient response 1-A load step
ΔVOUT = 5%
Input voltage
12 V nominal, 8 V to 17 V
Output voltage ripple
33 mV p-p
Start input voltage (rising VIN)
6.528 V
Stop input voltage (falling VIN)
6.190 V
Switching frequency
480 kHz
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8.2.2 Detailed Design Procedures
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS54622 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Operating Frequency
The first step is to decide on a switching frequency for the regulator. There is a trade off between higher and
lower switching frequencies. Higher switching frequencies may produce smaller a solution size using lower
valued inductors and smaller output capacitors compared to a power supply that switches at a lower frequency.
However, the higher switching frequency causes extra switching losses, which hurt the converter’s efficiency and
thermal performance. In this design, a moderate switching frequency of 480 kHz is selected to achieve both a
small solution size and a high-efficiency operation.
8.2.2.3 Output Inductor Selection
To calculate the value of the output inductor, use Equation 18. KIND is a coefficient that represents the amount
of inductor ripple current relative to the maximum output current. The inductor ripple current is filtered by the
output capacitor. Therefore, choosing high inductor ripple currents impact the selection of the output capacitor
since the output capacitor must have a ripple current rating equal to or greater than the inductor ripple current. In
general, the inductor ripple value is at the discretion of the designer; however, KIND is normally from 0.1 to 0.3
for the majority of applications.
L1 =
Vinm ax - Vout
Vout
×
Io × Kind
Vinm ax × f sw
(18)
For this design example, use KIND = 0.3 and the inductor value is calculated to be 3.08 µH. For this design, a
nearest standard value was chosen: 3.3 µH. For the output filter inductor, it is important that the RMS current
and saturation current ratings not be exceeded. The RMS and peak inductor current can be found from
Equation 20 and Equation 21.
Vinmax - Vout
Vout
×
Iripple =
L1
Vinmax × f sw
(19)
1 æ Vo × (Vinmax - Vo ) ö
ILrms = Io + × ç
÷
12 çè Vinmax × L1× f sw ÷ø
2
2
Iripple
ILpeak = Iout +
2
(20)
(21)
For this design, the RMS inductor current is 6.02 A and the peak inductor current is 6.84 A. The chosen inductor
is a Coilcraft MSS1048 series 3.3 µH. It has a saturation current rating of 7.38 A and a RMS current rating of
7.22 A.
24
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The current flowing through the inductor is the inductor ripple current plus the output current. During power-up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
8.2.2.4 Output Capacitor Selection
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the more stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator can not. This situation would occur if there are desired hold-up
times for the regulator where the output capacitor must hold the output voltage above a certain level for a
specified amount of time after the input power is removed. The regulator is also temporarily not able to supply
sufficient output current if there is a large, fast increase in the current needs of the load such as a transition from
no load to full load. The regulator usually needs two or more clock cycles for the control loop to see the change
in load current and output voltage and adjust the duty cycle to react to the change. The output capacitor must be
sized to supply the extra current to the load until the control loop responds to the load change. The output
capacitance must be large enough to supply the difference in current for 2 clock cycles while only allowing a
tolerable amount of droop in the output voltage. Equation 22 shows the minimum output capacitance necessary
to accomplish this.
2 × DIout
Co >
f sw × DVout
where
•
•
•
ΔIout is the change in output current.
fSW is the regulators switching frequency.
ΔVout is the allowable change in the output voltage.
(22)
For this example, the transient load response is specified as a 5% change in Vout for a load step of 1 A. For this
example, ΔIout = 3 A and ΔVout = 0.05 × 3.3 = 0.165 V. Using these numbers gives a minimum capacitance of
75.8 μF. This value does not take the ESR of the output capacitor into account in the output voltage change. For
ceramic capacitors, the ESR is usually small enough to ignore in this calculation.
Equation 23 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
Where fsw is the switching frequency, Vripple is the maximum allowable output voltage ripple, and Iripple is the
inductor ripple current. In this case, the maximum output voltage ripple is 33 mV. Under this requirement,
Equation 23 yields 13.2 µF.
1
1
Co >
×
8 × f sw Voripple
Iripple
(23)
Equation 24 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 24 indicates the ESR should be less than 19.7 mΩ. In this case, the ESR of the ceramic
capacitors is much smaller than 19.7 mΩ.
Voripple
Resr <
Iripple
(24)
Additional capacitance deratings for aging, temperature and DC bias should be factored in which increases this
minimum value. For this example, a 100-μF, 6.3-V X5R ceramic capacitor with 3 mΩ of ESR is be used.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the RMS (Root Mean Square) value of the maximum ripple current. Equation 25 can be used
to calculate the RMS ripple current the output capacitor needs to support. For this application, Equation 25 yields
485 mA.
Vout × (Vinmax - Vout )
Icorms =
12 × Vinmax × L1× f sw
(25)
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8.2.2.5 Input Capacitor Selection
The TPS54622 requires a high-quality ceramic, type X5R or X7R, input decoupling capacitor of at least 4.7 µF of
effective capacitance on the PVIN input voltage pins and 4.7 µF on the Vin input voltage pin. In some
applications, additional bulk capacitance may also be required for the PVIN input. The effective capacitance
includes any DC bias effects. The voltage rating of the input capacitor must be greater than the maximum input
voltage. The capacitor must also have a ripple current rating greater than the maximum input current ripple of the
TPS54622. The input ripple current can be calculated using Equation 26.
Icirms = Iout ×
Vout (Vinmin - Vout )
×
Vinmin
Vinmin
(26)
The value of a ceramic capacitor varies significantly over temperature and the amount of DC bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the DC bias taken into account. The capacitance value of a capacitor
decreases as the DC bias across a capacitor increases. For this example design, a ceramic capacitor with at
least a 25 V voltage rating is required to support the maximum input voltage. For this example, one 10 μF and
one 4.7 µF 25-V capacitors in parallel have been selected as the VIN and PVIN inputs are tied together so the
TPS54622 may operate from a single supply. The input capacitance value determines the input ripple voltage of
the regulator. The input voltage ripple can be calculated using Equation 27. Using the design example values,
Ioutmax = 6 A, Cin = 14.7 μF, Fsw = 480 kHz, yields an input voltage ripple of 213 mV and a RMS input ripple
current of 2.95 A.
Ioutmax × 0.25
DVin =
Cin × f sw
(27)
8.2.2.6 Slow-Start Capacitor Selection
The slow-start capacitor determines the minimum amount of time it takes for the output voltage to reach its
nominal programmed value during power up. This is useful if a load requires a controlled voltage slew rate. This
is also used if the output capacitance is very large and would require large amounts of current to quickly charge
the capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the
TPS54622 reach the current limit or excessive current draw from the input power supply may cause the input
voltage rail to sag. Limiting the output voltage slew rate solves both of these problems. The soft-start capacitor
value can be calculated using Equation 28. For the example circuit, the soft-start time is not too critical since the
output capacitor value is 100 μF which does not require much current to charge to 3.3 V. The example circuit has
the soft-start time set to an arbitrary value of 6 ms which requires a 22-nF capacitor. In TPS54622, Iss is 2.3 uA
and Vref is 0.6 V.
Tss(ms) × Iss( m A )
C6(nF) =
Vref ( V )
(28)
8.2.2.7 Bootstrap Capacitor Selection
A 0.1-µF to 1-μF ceramic capacitor must be connected between the BOOT to PH pin for proper operation. TI
recommends using a ceramic capacitor with X5R or better grade dielectric. The capacitor should have 10-V or
higher voltage rating.
8.2.2.8 Undervoltage Lockout Setpoint
The undervoltage lockout (UVLO) can be adjusted using the external voltage divider network of R3 and R4. R3 is
connected between VIN and the EN pin of the TPS54622 and R4 is connected between EN and GND . The
UVLO has two thresholds, one for power up when the input voltage is rising and one for power down or
brownouts when the input voltage is falling. For the example design, the supply should turn on and start
switching once the input voltage increases above 6.528 V (UVLO start or enable). After the regulator starts
switching, it should continue to do so until the input voltage falls below 6.19 V (UVLO stop or disable). Equation 2
and Equation 3 can be used to calculate the values for the upper and lower resistor values. For the stop voltages
specified, the nearest standard resistor value for R3 is 35.7 kΩ and for R4 is 8.06 kΩ.
26
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8.2.2.9 Output Voltage Feedback Resistor Selection
The resistor-divider network R5 and R6 is used to set the output voltage. For the example design, 10 kΩ was
selected for R5. Using Equation 29, R6 is calculated as 2.22 kΩ. The nearest standard 1% resistor is 2.21 kΩ.
R5 × Vref
R6 =
Vo - Vref
(29)
8.2.2.9.1 Minimum Output Voltage
Due to the internal design of the TPS54622, there is a minimum output voltage limit for any given input voltage.
The output voltage can never be lower than the internal voltage reference of 0.6 V. Above 0.6 V, the output
voltage may be limited by the minimum controllable on-time. The minimum output voltage in this case is given by
Equation 30:
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Voutmin = Ontimemin × Fsmax (Vinmax + Ioutmin (RDS2min - RDS1min ))- Ioutmin (RL + RDS2min )
where
•
•
•
•
•
•
•
•
Voutmin = minimum achievable output voltage
Ontimemin = minimum controllable on-time (135 ns maximum)
Fsmax = maximum switching frequency including tolerance
Vinmax = maximum input voltage
Ioutmin = minimum load current
RDS1min = minimum high-side MOSFET ON-resistance (36-32 mΩ typical)
RDS2min = minimum low-side MOSFET ON-resistance (19 mΩ typical)
RL = series resistance of output inductor
(30)
8.2.2.10 Compensation Component Selection
There are several industry techniques used to compensate DC-DC regulators. The method presented here is
easy to calculate and yields high phase margins. For most conditions, the regulator has a phase margin from 60
to 90 degrees. The method presented here ignores the effects of the slope compensation that is internal to the
TPS54622. Since the slope compensation is ignored, the actual crossover frequency is usually lower than the
crossover frequency used in the calculations.
First, the modulator pole, fpmod, and the ESR zero, fzmod, must be calculated using Equation 31 and
Equation 32. For Cout, use a derated value of 75 µF. Use Equation 33 and Equation 34 to estimate a starting
point for the closed loop crossover frequency, fco. Then the required compensation components may be derived.
For this design example, fpmod is 3.86 kHz and fzmod is 707.4 kHz. Equation 33 is the geometric mean of the
modulator pole and the ESR zero and Equation 34 is the geometric mean of the modulator pole and one half the
switching frequency. Use a frequency near the lower of these two values as the intended crossover frequency,
fco. In this case Equation 33 yields 52.2 kHz and Equation 34 yields 30.4 kHz. The lower value is 30.4 kHz. A
slightly higher frequency of 30 kHz is chosen as the intended crossover frequency.
Iout
f pmod =
2 × p × Vout × Cout
(31)
f zm od =
1
2 × p × RESR × Cout
f co =
f pmod × f zmod
f co =
f pmod ×
(32)
(33)
f sw
2
(34)
Now the compensation components can be calculated. First calculate the value for R2 which sets the gain of the
compensated network at the crossover frequency. Use Equation 35 to determine the value of R2.
2p × f c × Vout × Cout
R4 =
gmea × Vref × gmps
(35)
Next calculate the value of C3. Together with R2, C3 places a compensation zero at the modulator pole
frequency. Equation 36 to determine the value of C3.
Vout × Cout
C4 =
Iout × R4
(36)
Using Equation 35 and Equation 36 the standard values for R4 and C4 are 3.74 kΩ and 0.01 µF.
An additional high-frequency pole can be used if necessary by adding a capacitor in parallel with the series
combination of R4 and C4. The pole frequency can be placed at the ESR zero frequency of the output capacitor
as given by Equation 8. Use Equation 37 to calculate the required capacitor value for C5.
RESR × Cout
C5 =
R4
(37)
28
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8.2.2.11 Fast Transient Considerations
In applications where fast transient responses are very important, Type III frequency compensation can be used
instead of the traditional Type II frequency compensation.
For more information about Type II and Type III frequency compensation circuits, see Designing Type III
Compensation for Current Mode Step-Down Converters and design calculator (SLVC219).
8.2.3 Application Curves
VOUT = 100 mV / div (dc coupled, -3.13 V offset)
VIN = 5 V / div
IOUT = 2 A / div
Load step = 1.5 A to 4.5 A Slew rate = 100 mA / µsec
VOUT = 1 V / div
Time = 200 µsec / div
Time = 2 msec / div
Figure 30. Load Transient
Figure 31. Start-Up With VIN
VIN = 5 V / div
VIN = 10 V / div
VOUT = 2 V / div
EN = 2 V / div
VOUT = 2 V / div
PH = 10 V / div
Time = 2 msec / div
Time = 2 msec / div
Figure 33. Start-Up With PRE-BIAS
Figure 32. Start-Up With EN
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VOUT = 20 mV / div (ac coupled)
VIN = 500 mV / div
PH = 5 V / div
PH = 5 V / div
Time = 1 µsec / div
Time = 1 µsec / div
Figure 34. Output Voltage Ripple With No Load
Figure 35. Input Voltage Ripple With Full Load
60
180
50
150
0.4
0.3
30
90
20
60
10
30
0
0
Gain
-10
-30
-20
-60
-30
-90
-40
-120
-50
-150
Output Voltage Deviation - %
120
Phase - Degrees
Gain - dB
Phase
40
0.2
IOUT = 3 A
0.1
0
-0.1
-0.2
-0.3
-60
100
1000
10000
-0.4
-180
1000000
100000
8
9
10
11
Frequency - Hz
Figure 36. Closed Loop Response
13
12
Input Voltage - V
14
15
16
Figure 37. Line Regulation
0.4
10
10
Vout
0.3
1
1
0.1
0
-0.1
0.1
0.1
Ideal Vsense
Vsense
0.01
0.01
0.001
0.001
0.0001
0.0001
Vsense Voltage - V
VIN = 12 V
0.2
Output Voltage - V
Output Voltage Deviation - %
17
-0.2
-0.3
-0.4
0
1
2
3
Output Current - A
4
5
6
0.00001
0.001
0.1
1
10
Track In Voltage - V
Figure 39. Tracking Performance
Figure 38. Load Regulation
30
0.00001
0.01
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150
TA - Maximum Ambient Temperature - °C
TA - Maximum Ambient Temperature - °C
150
125
125
100
100
75
VIN = 12 V,
VOUT = 3.3 V,
Fsw = 480 kHz,
room temp, no air flow
50
75
50
25
25
0
1
2
3
4
Load Current - A
5
6
Figure 40. Maximum Ambient Temperature vs Load
Current
0
0.5
1
1.5
2 2.5
3
3.5
PD - IC Power Dissipation - W
4
Figure 41. Maximum Ambient Temperature vs IC Power
Dissipation
100
150
TA = room temperature,
no air flow
95
125
90
85
VIN = 8 V
Efficiency - %
TJ - Junction Temperature - °C
Tjmax = 150 °C,
no air flow
100
75
80
VIN = 12 V
75
VIN = 17 V
70
65
50
60
55
25
0
0.5
1
1.5
2
2.5
3
3.5
Pic - IC Power Dissipation - W
50
4
0
Figure 42. Junction Temperature vs IC Power Dissipation
1
3
Output Current - A
2
4
5
6
Figure 43. Efficiency vs Load Current
100
VIN = 12 V
VIN = 17 V
90
80 VIN = 8 V
VOUT = 2 V / div
Efficiency - %
70
60
PH = 10 V / div
50
Inductor Current = 5 A / div
40
30
20
10
0
0.01
Time = 20 msec / div
0.1
10.0
1.0
Output Current - A
Figure 44. Efficiency vs Load Current
Figure 45. Hiccup Mode Current Limit
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9 Power Supply Recommendations
The TPS54622 is designed to operate from an input voltage supply range from 4.5 V to 17 V. This supply voltage
must be well regulated. Power supplies must be well bypassed for proper electrical performance. This includes a
minimum of one 4.7-µF (after derating) ceramic capacitor, type X5R or better from PVIN to GND, and from VIN to
GND. Additional local ceramic bypass capacitance may be required in systems with small input ripple
specifications, in addition to bulk capacitance if the TPS54622 device is located more than a few inches away
from its input power supply. In systems with an auxiliary power rail available, the power stage input, PVIN, and
the analog power input, VIN, may operate from separate input supplies. See Figure 46 for recommended bypass
capacitor placement.
10 Layout
10.1 Layout Guidelines
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
32
Layout is a critical portion of good power supply design. See Figure 46 for a PCB layout example.
The top layer contains the main power traces for VIN, VOUT, and VPHASE. Also on the top layer are
connections for the remaining pins of the TPS54622 and a large top-side area filled with ground.
Connect the top layer ground area to the internal ground layers using vias at the input bypass capacitor, the
output filter capacitor, and directly under the TPS54622 device to provide a thermal path from the exposed
thermal pad land to ground
Tie the GND pin directly to the power pad under the IC and the power pad.
For operation at full rated load, the top side ground area together with the internal ground plane, must provide
adequate heat dissipating area.
There are several signals paths that conduct fast changing currents or voltages that can interact with stray
inductance or parasitic capacitance to generate noise or degrade the power supplies performance.
To help eliminate these problems, the PVIN pin should be bypassed to ground with a low ESR ceramic
bypass capacitor with X5R or X7R dielectric.
Care should be taken to minimize the loop area formed by the bypass capacitor connections, the PVIN pins,
and the ground connections.
The VIN pin must also be bypassed to ground using a low ESR ceramic capacitor with X5R or X7R dielectric.
Make sure to connect this capacitor to the quite analog ground trace rather than the power ground trace of
the PVIn bypass capacitor.
Since the PH connection is the switching node, the output inductor should be located close to the PH pins,
and the area of the PCB conductor minimized to prevent excessive capacitive coupling.
The output filter capacitor ground should use the same power ground trace as the PVIN input bypass
capacitor.
Try to minimize this conductor length while maintaining adequate width.
The small signal components should be grounded to the analog ground path as shown.
The RT/CLK pin is sensitive to noise so the RT resistor must be located as close as possible to the IC and
routed with minimal lengths of trace.
The additional external components can be placed approximately as shown.
It may be possible to obtain acceptable performance with alternate PCB layouts, however, this layout has
been shown to produce good results and is meant as a guideline.
Land pattern and stencil information is provided in the data sheet addendum.
The dimension and outline information is for the standard RHL (S-PVQFN-N14) package.
There may be slight differences between the provided data and actual lead frame used on the TPS54622RHL
package.
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10.2 Layout Examples
TOPSIDE
GROUND
AREA
FREQUENCY SET RESISTOR
PVIN
INPUT
BYPASS
CAPACITOR
RT/CLK
PWRGD
GND
GND
OUTPUT
FILTER
CAPACITOR
BOOT
CAPACITOR
BOOT
EXPOSED THERMAL
PAD AREA
PVIN
PH
PVIN
EN
VIN
SS/TR
VSENSE
PVIN
OUTPUT
INDUCTOR
PH
VOUT
PH
COMP
VIN
SLOW START
CAPACITOR
VIN
INPUT
BYPASS
CAPACITOR
FEEDBACK
RESISTORS
UVLO SET
RESISTORS
COMPENSATION
NETWORK
ANALOG GROUND TRACE
0.010 in. Diameter
Thermal VIA to Ground Plane
VIA to Ground Plane
Etch Under Component
Figure 46. PCB Layout
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Layout Examples (continued)
Figure 47. Ultra-Small PCB Layout Using TPS54622 (PMP4854-2)
10.3 Estimated Circuit Area
The estimated printed-circuit-board area for the components used in the design of Figure 29 is 0.58 in2
(374mm2). This area does not include test points or connectors.
34
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
For the Design Calculator, see SLVC219.
11.1.3 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS54622 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation, see the following:
Designing Type III Compensation for Current Mode Step-Down Converters, SLVA352
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.5 Trademarks
E2E is a trademark of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
Submit Documentation Feedback
Copyright © 2011–2017, Texas Instruments Incorporated
Product Folder Links: TPS54622
35
TPS54622
SLVSA70F – MARCH 2011 – REVISED OCTOBER 2017
www.ti.com
11.5 Trademarks (continued)
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
36
Submit Documentation Feedback
Copyright © 2011–2017, Texas Instruments Incorporated
Product Folder Links: TPS54622
PACKAGE OPTION ADDENDUM
www.ti.com
17-Jan-2018
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
TPS54622RHLR
ACTIVE
VQFN
RHL
14
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 150
54622
TPS54622RHLT
ACTIVE
VQFN
RHL
14
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 150
54622
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
17-Jan-2018
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
24-Jan-2019
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
TPS54622RHLR
VQFN
RHL
14
3000
330.0
12.4
3.75
3.75
1.15
8.0
12.0
Q2
TPS54622RHLT
VQFN
RHL
14
250
180.0
12.4
3.75
3.75
1.15
8.0
12.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
24-Jan-2019
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS54622RHLR
VQFN
RHL
14
3000
367.0
367.0
35.0
TPS54622RHLT
VQFN
RHL
14
250
210.0
185.0
35.0
Pack Materials-Page 2
IMPORTANT NOTICE AND DISCLAIMER
TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATASHEETS), DESIGN RESOURCES (INCLUDING REFERENCE
DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS”
AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY
IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
PARTY INTELLECTUAL PROPERTY RIGHTS.
These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate
TI products for your application, (2) designing, validating and testing your application, and (3) ensuring your application meets applicable
standards, and any other safety, security, or other requirements. These resources are subject to change without notice. TI grants you
permission to use these resources only for development of an application that uses the TI products described in the resource. Other
reproduction and display of these resources is prohibited. No license is granted to any other TI intellectual property right or to any third
party intellectual property right. TI disclaims responsibility for, and you will fully indemnify TI and its representatives against, any claims,
damages, costs, losses, and liabilities arising out of your use of these resources.
TI’s products are provided subject to TI’s Terms of Sale (www.ti.com/legal/termsofsale.html) or other applicable terms available either on
ti.com or provided in conjunction with such TI products. TI’s provision of these resources does not expand or otherwise alter TI’s applicable
warranties or warranty disclaimers for TI products.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2019, Texas Instruments Incorporated
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