Texas Instruments | Multiphase Buck Design from Start to Finish, Part 1 (Rev. A) | Application notes | Texas Instruments Multiphase Buck Design from Start to Finish, Part 1 (Rev. A) Application notes

Texas Instruments Multiphase Buck Design from Start to Finish, Part 1 (Rev. A) Application notes
Application Report
SLVA882A – April 2017 – Revised May 2019
Multiphase Buck Design From Start to Finish (Part 1)
Carmen Parisi
ABSTRACT
This application report covers the basics of multiphase buck regulators. A comparison versus single-phase
regulators is presented before diving into a detailed design example aimed at powering the core rail of a
generic networking ASIC setting up a second application report discussing printed-circuit board (PCB)
layout techniques and performance testing.
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Contents
Introduction ................................................................................................................... 2
Multiphase Buck Regulator Overview ..................................................................................... 2
Advantages of Multiphase Regulators .................................................................................... 4
Multiphase Challenges ...................................................................................................... 9
Multiphase Design Example - Component Selection .................................................................. 10
Conclusion .................................................................................................................. 17
References .................................................................................................................. 18
List of Figures
1
Multiphase Regulator Example ............................................................................................ 2
2
TPS53679 Demo Board With Controller and Power Stage ICs Highlighted ......................................... 3
3
Input Current Waveforms ................................................................................................... 4
4
Normalized Input Capacitance RMS Current
5
Inductor Ripple Current Waveforms ....................................................................................... 6
6
Normalized Output Capacitance Ripple
7
Efficiency vs Phase Number ............................................................................................... 8
8
TPS53661 5-PH Efficiency Curve ......................................................................................... 8
9
Simplified Comparison Between Current Sense Methods
10
Capacitor Derating Curves Courtesy of Murata Left:1210 Case, GRM32ER61C226ME20L, Right: 1206
Case, GRM31CR61C226ME15 .......................................................................................... 13
11
Load Transient Waveforms ............................................................................................... 14
12
Load Transient with DC Load Line ....................................................................................... 15
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..................................................................................
.............................................................
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9
List of Tables
...............................................................................................
1
Multiphase Design Targets
2
Summary of Driver and FET Implementations ......................................................................... 12
3
Power Stage Loss Calculations per Phase ............................................................................. 12
4
Output Capacitor Options ................................................................................................. 16
5
Output Capacitor Solution Comparison
6
Multiphase Design Comparison .......................................................................................... 17
7
Case Study Design Summary ............................................................................................ 17
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1
Introduction
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Introduction
In today’s computing environment CPUs, FPGAs, ASICs, and even peripherals are growing increasingly
complex. In turn so do their power delivery requirements. To handle the higher demands, multiphase
regulators are becoming more common on motherboards in many areas of computing from laptops and
tablets to servers and Ethernet switches. Designing with these regulators is more challenging than using
conventional switchers and linear regulators but the benefits of multiphase outweigh the complexity for
high-performance power applications. This tutorial is designed to provide the necessary equations and
guidance to get a new multiphase design up and running and ready for validation. After an overview of
multiphase benefits, an in-depth design example of a multiphase buck regulator for an ASIC core rail is
presented. Part 1 of this series focuses on the design specifications and component selection. Part 2
covers the PCB layout and basic performance testing.
2
Multiphase Buck Regulator Overview
A multiphase buck regulator is a parallel set of buck power stages as shown in Figure 1 and Figure 2,
each with its own inductor and set of power MOSFETs. Collectively, these components are called a
phase. These phases are connected in parallel and share both input and output capacitors. During steadystate operation, individual phases are active at spaced intervals equal to 360° / n throughout the switching
period where n is the total number of phases. Figure 2 shows a TPS53679 multiphase controller
demonstration board and TI power stages for a six-phase design.
Figure 1. Multiphase Regulator Example
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Figure 2. TPS53679 Demo Board With Controller and Power Stage ICs Highlighted
Today’s controllers most commonly support applications needing two to eight phases. Techniques exist to
extend the phase count to 12 or more, but these are outside the scope of this document. As a general
guideline, the maximum phase current should be kept between 30 to 40 A. Depending on budget,
efficiency targets, and available cooling methods the maximum phase current can be increased but it is
highly recommended to do a thorough study of the ramifications before committing to the design.
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Advantages of Multiphase Regulators
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Advantages of Multiphase Regulators
Compared to single-phase buck regulators, multiphase converters offer several key performance
advantages that make them the default choice for high-power, high-performance applications:
• Reduced input capacitance
• Reduced output capacitance
• Improved thermal performance and efficiency at high load currents
• Improved overshoot and undershoot during load transients
3.1
Input Capacitance Reduction
Adding additional phases to a design decreases the RMS input current flowing through the decoupling
capacitors thereby reducing the ripple on the input voltage, VIN. Fewer capacitors are then needed to keep
VIN ripple within specifications. Self-heating effects within the capacitors themselves due to equivalent
series resistance (ESR) are also reduced.
Figure 3. Input Current Waveforms
Figure 3 shows the input current waveforms for a two-phase buck compared to a single-phase design
(dashed line). Lower RMS and peak currents from the addition of a second phase not only reduces the
input capacitance, CIN, but also provides less stress on the upper MOSFET of each phase.
I CIN
norm
RMS
m · §1 m
§
¨D n ¸ u ¨ n
©
¹ ©
·
D¸
¹
where
•
•
•
D = VOUT / VIN
n = # of phases
m = floor (n × D)
(1)
Calculating the normalized RMS input current of a regulator can be done using the formula in Equation 1.
Plotting this equation as a function of duty cycle and phase number gives the curves in Figure 4. These
graphs show a higher phase count can reduce the amount of current the input capacitors have to handle
by 50% or more depending on the duty cycle.
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Figure 4. Normalized Input Capacitance RMS Current
At several points on the graphs in Figure 4, the input RMS current drops to zero as the individual ripple
currents for each phase cancel one another out. While mathematically it may be possible set the phase
number and duty cycle of a design to operate at a zero current point and eschew input caps altogether, in
reality this is unachievable. Noise, line transients, load transients, and natural variations in the duty cycle
make no input current ripple unrealizable in practice. Spacing between phases can reach several inches
for 4+ phase designs causing PCB inductance to reduce the effects of ripple cancellation and so
capacitors must always be used.
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Advantages of Multiphase Regulators
3.2
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Output Capacitance Reduction
Because all phases of a multiphase design are tied together at the output node, the inductor currents of
each phase are concurrently charging and discharging the output capacitors depending on whether or not
a given phase is active. This charging and discharging produce one overall current, ISUM, the AC portion of
which gets absorbed by the output capacitance, COUT. Compared to the ripple current of an individual
phase ISUM has a lower peak-to-peak value in steady state as shown in Figure 5. Smaller ripple current in
the output capacitors lowers the overall output voltage ripple which in turn lowers the amount of
capacitance needed to keep VOUT within tolerance.
Figure 5. Inductor Ripple Current Waveforms
The normalized ripple current for the output capacitors is calculated using Equation 2 and plotted in
Figure 6 for two-, three-, and four-phase buck converters. Setting n = 1 gives ICOUTnorm = 1 for all duty
cycles making Equation 2 invalid for single-phase calculations. Much like with the input capacitor current,
at various duty cycles the currents of the inductors mathematically cancel out suggesting no output current
ripple. Even when designed to operate at one of these points, a converter always requires some amount
of output capacitance due to noise, transients, and duty cycle variation. However, for fixed output
applications, operating near one of these zero points leads to an optimal design with the fewest number of
output capacitors.
n
m · §1 m
§
·
u ¨D
u
I COUTripple,norm
D¸
Du 1 D
n ¸¹ ¨© n
©
¹
(2)
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Figure 6. Normalized Output Capacitance Ripple
Unlike with input ripple cancellation, output ripple cancellation is less affected by the PCB layout. Usually,
a significant number of output capacitors are tightly packed close to the CPU or point of load reducing the
effects of parasitic inductance between components. Also, the inductor value of each phase dominates
parasitics for all but the highest frequency designs allowing for better cancellation between phases.
3.3
Thermal Performance and Efficiency Improvements
Single-phase converters by definition have all the output power flowing through a single inductor and pair
of FETs. Any power loss is contained solely within those components. For an application with greater than
100 A of output current, sourcing FETs and inductors rated to such large currents becomes difficult and
expensive. Concentrating the entirety of the losses of a design into one small area of a PCB and set of
components comes at an undesirable loss of efficiency.
Multiphase regulators spread power loss evenly across all phases. Since each phase is dealing with only
a portion of the total output current, selecting FETs and inductors becomes easier as less thermal strain is
placed on these components. Regulator efficiency is also able to remain much higher over the entire load
range when compared to an equivalent single-phase design. Performance is further improved by the
reductions in CIN and COUT discussed previously as lower ripple current in the capacitors produces less
self-heating and lower power loss.
Modern DC/DC controllers allow for phases to be added and dropped as needed depending on the load
current as shown in Figure 7. These add and drop points can be tuned to account for various FET and
inductor combinations for optimal efficiency across many applications and conditions.
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Figure 7. Efficiency vs Phase Number
At low currents, fewer phases are used, down to a single phase operating in Discontinuous Conduction
Mode, to minimize the FET switching losses and the current draw associated with the power stage and
gate drivers of each phase. As the load current increases, conduction losses begin to dominate over
switching loss and more phases are activated to keep the efficiency as high as possible. The optimum set
point to turn on a phase occurs at the intersection of two efficiency curves. For example, phase two should
be turned on where the falling single-phase efficiency curve crosses the rising two-phase efficiency curve.
Figure 8 depicts an efficiency curve taken of a five-phase design using the TPS53661 controller and
CSD95372B power stage. The design called for VIN = 12 V, VOUT = 1.8 V, used a switching frequency of
600-kHz and 150-nH inductors. An efficiency > 90% is maintained from 5 A to 200 A, a feat which for all
intents and purposes is impossible to do with only a single-phase buck.
Figure 8. TPS53661 5-PH Efficiency Curve
3.4
Transient Response Improvements
In many high-performance applications the capacitance requirements demanded by load transients far
exceed what is called for to successfully hit DC ripple targets. During load transients multiphase
converters offer the advantage of needing fewer output capacitors to keep VOUT within the specifications of
a given design.
During a transient a multiphase controller overlaps phases during a load step or turns all phases off during
a load release, effectively putting the inductors in parallel with one another. This reduces equivalent
inductance,(LEQ) seen at the output node by a factor of n, where n is the total number of phases. With a
smaller LEQ, charge can quickly be supplied from the supply to the output caps reducing undershoot.
Similarly, overshoot is reduced as less excess charge stored in the inductors is transferred to the output
capacitors when the phases are all shut off.
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Multiphase Challenges
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4
Multiphase Challenges
While multiphase bucks offer many benefits over single-phase converters, they do present some
challenges that must be overcome in order to successfully implement a design. Adding additional phases
to a converter increases bill of materials (BOM) cost and PCB area. The price of more inductors and FETs
must be weighed against the increased cost of sourcing more robust components and needing higher
capacitor counts to implement a single-phase regulator instead. To minimize the greater board area
needed for multiphase solutions, a balance between current capabilities and thermal performance versus
overall phase number must be found.
Perhaps the biggest challenge of multiphase converters is phase management. In order to achieve the
highest possible performance, current must be evenly balanced between active phases to avoid thermally
stressing any one phase and provide optimal ripple cancellation. Additionally, phases must be quickly
added or removed during transients to minimize excursions on the output voltage. Keeping the phases
balanced requires a more sophisticated controller versus a single-phase buck. The sophistication comes
from more sense lines, signal routing, current sense components, and so forth, that must be fed back to
the controller in order to accurately balance phase currents.
Determining the phase current is traditionally done through a current sense resistor in series with each
inductor or by utilizing the parasitic DC resistance (DCR) of the inductor. These methods are sensitive to
component placement and signal routing making implementation difficult. The sense circuitry for each
phase requires additional passive components to provide filtering and in the case of resistor sensing, adds
an additional point of power loss. However, Smart Power Stages, such as the CSD95372B and
CSD95490, have recently hit the market integrating current sense capabilities directly in the DriverMOSFET package. When paired with a compatible controller, these ICs offer increased current sense
accuracy, eliminate a number of passive components, and require fewer differential signals, if any, to be
routed across the PCB as seen in Figure 9.
DCR Sense
Driver
Smart Power Stage
Smart Power Stage
VIN
VIN
BOOT
PWM1
PWM
VIN
VCC
BOOT
VCC
PWM1
PWM
UG
VCC
PHASE
VOUT
VCC
VOUT
PHASE
COUT
LG
CS1P
CS1
IOUT
TSEN
TSEN
DIFF
Pair
CS1N
Smart Power Stage
Driver
VIN
VIN
PWM2
PWM
VCC
PHASE
VCC
VCC
PHASE
CS2P
DIFF
Pair
CS2N
Driver
VIN
Multiphase Controller
Multiphase Controller
CS2
LG
IOUT
PWM
Smart Power Stage
VIN
PWM
PWM3
VCC
PHASE
VCC
BOOT
PHASE
CS3
IOUT
LSET
TSEN
LG
CS3P
VIN
VCC
UG
VCC
LSET
TSEN
BOOT
CS3N
BOOT
UG
PWM
PWM3
VIN
VCC
BOOT
PWM2
COUT
LSET
DIFF
Pair
Figure 9. Simplified Comparison Between Current Sense Methods
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Multiphase Design Example - Component Selection
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Multiphase Design Example - Component Selection
To illustrate the benefits of multiphase buck regulators a design using the specifications in Table 1 is
worked through from initial component selection, to PCB layout, and finally performance testing. Only the
initial design is currently discussed; layout and testing are the subjects of a future application report. While
working through the design process component count, efficiency, and layout complexity are studied to
strike a balance between performance and ease of implementation.
Table 1. Multiphase Design Targets
VIN
12 V
Input Voltage
VOUT
0.9 V
Nominal Output Voltage
ITDC
200 A
Thermal Design Current
IMAX
240 A
Max Current
ISTEP
150 A
Max Load Step
DCLL
0.5 mΩ
DC Load Line
ΔVOUT(DC)
±1%
VOUT DC Ripple
ΔVOUT(AC)
±5%
VOUT Transient Specifications
ΔVIN(DC)
240 mVpp
VIN DC Ripple
ΔVIN(AC)
±360 mV
VIN Overshoot and Undershoot
PMBus with
Telemetry
Yes
Requires PMBus interface with VIN, IIN, VOUT, IOUT, and Temp
readings
The requirements in Table 1 are typical specifications for the core voltage rail of a generic networking
ASIC that may be found on an enterprise motherboard. Most of the specifications are fairly straightforward
to anyone who has done a DC/DC switcher design before with the possible exception of the DC load line
and PMBUS requirements.
With a DC load line, a buck regulator essentially presents itself as a fixed resistance to the output load.
From the example numbers - with a 200-A load being pulled by the ASIC the nominal output voltage of 0.9
V drops by 200 A × 0.5 mΩ, or 100 mV, to 0.8 V. This lowers the power consumption of the processor by
20 W, easing the strain on whatever heatsink or thermal solution is in place. This 20 W difference is not
dissipated by the regulator; it simply is not drawn from the input supply. When the load current drops
below 200 A the output voltage rises accordingly. Load lines also make meeting the transient
specifications much easier by reducing the amount of output caps needed, as discussed in Section 5.5.
Power Management Bus or, PMBus™, is an open, industry standard interface based on I2C that can be
found on many modern single and multiphase regulators. When implemented, the bus allows for easy
adjustment of the output voltage, reporting of load conditions and FET temperature, as well as fault
recording. If a digital or hybrid modulator is used in the controller PMBus can also be used to change the
compensation of a converter during design validation.
5.1
Phase Count
With a 200-A TDC and 240-A maximum current, the design requires six phases to keep the individual
phase currents below 40 A. Four- and five-phase designs result in TDC current levels that make power
loss through the inductors and FETs difficult to manage. Conversely, a six-phase solution only has 33 A
flowing per phase at ITDC and 40 A while at IMAX, providing a more manageable power loss scenario. The
additional phases also provide a significant reduction in the amount of capacitors required to maintain
regulation during load transients which can be seen in Table 6 of the Design Summary section.
5.2
Inductor
To choose an inductor, the switching frequency must first be decided. Frequencies around 300 kHz can
provide low switching loss and high efficiency at the price of slow transient response as larger inductors
are needed and the control loop bandwidth must be set lower than it otherwise would be at higher
frequencies. Similarly, higher switching frequencies around 1 MHz suffer from greater switching loss but
offer faster transient response.
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For this design, a switching frequency of 600 kHz is used to provide a balanced tradeoff between transient
response and efficiency. Using the standard buck design equation for calculating inductance and a ripple
current target of 25%, an inductance of 0.138 µH per phase is calculated using Equation 3. Rounding
towards the closest standard value gives an inductance of 0.15 µH per phase.
§ 0.9 V ·
0.9 V u ¨ 1
VOUT u 1 D
12 V ¸¹
©
L
0.138 µH
240 A ·
fSW u I PP
§
600 kHz u ¨ 0.25 u
6 ¸¹
©
(3)
The inductor for this design was chosen from the popular IHLP line of inductors from Vishay Dale,
specifically the IHLP-5050FD series. The 150-nH choke from this series has a typical DCR of 0.53 mΩ for
low conduction losses as well as minimal AC loss that can be estimated using the Vishay online
calculator. It is also thermally rated out to 55 A, providing margin since only 40 A per phase is expected.
The soft saturation curve of the powdered core on this inductor means the inductance remains relatively
flat out to its saturation current rating before slowly rolling off giving predictable performance over the
range of expected operating conditions. Should a severe over-current event occur above the saturation
current rating, a powdered core makes damage to the FETs and PCB much less likely than with a ferrite
core. With a ferrite core, the inductance drops off quickly at the saturation point and the inductor
essentially becomes a short which can pull a damaging amount of current.
5.3
Driver and Power MOSFETs
When working through a multiphase design there are three options available to a designer when it comes
to deciding how to implement the controller, drivers, and power MOSFETs. Table 2 summarizes the
general pros and cons of each option.
1. Discrete ICs for the controller, MOSFET drivers, and FETs
2. A controller with integrated drivers and discrete FETs
3. A driverless controller with the FETs and IC combined into one IC package
Option 1 offers the most design flexibility, provided common footprints are used, as the FETs and drivers
can be swapped out easily if requirements change. The controller sends a PWM signal out to each driver
IC which then converts the signal into the upper and lower gate drive signals for the MOSFETs. This
option may also prove to be the cheapest since the individual ICs themselves are neither highly integrated
nor sophisticated. However, going with an all discrete solution places the optimization of the driver-FET
combination on the designer which increases the design complexity and may not be an option in a timeconstrained scenario. Performance is also much more affected by the PCB layout as opposed to more
integrated solutions as there are a greater number of high-power nodes, drive signals, and sense lines to
route along with additional parasitic elements.
Option 2 restricts the design freedom an engineer has since the drivers are paired with the controller and
may not be suitable for driving all possible FETs. It also requires that the controller be located relatively
close to the phases because the gate signals cannot be run for long distances without compromising
performance. Layout area and complexity compared to an all discrete solution depends on the phase
count. As the phase count increases, the controller size balloons out as at least four additional pins per
phase (Upper Gate Drive, Lower Gate Drive, Phase Sense, and Boot) are needed. For designs greater
than two or three phases, maintaining a proper layout with this option becomes difficult at best. Finding a
controller that supports a high phase count with integrated drivers may not be possible at all. Stacking
multiple controllers together only further complicates the design.
Option 3 provides the easiest design and layout at the expense of BOM cost because of the high
integration in the ICs. Only PWM signals are sent between the controller and driver-FET IC. No gate drive
signal routing is required. This option also provides the optimal driver FET combination, with the lowest
parasitics, translating into higher efficiency and a lower chance of shoot-through. If telemetry data for
parameters such as input current, output current, and temperature are required, these features can be
easily added into a driver-FET power stage instead of requiring additional discrete circuitry.
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Table 2. Summary of Driver and FET Implementations
Design Parameter
Option 1 –
Discrete Solution
Option 2 –
Controller+Driver with FETs
Option 3 –
Controller with Driver+FET
Flexibility
High
Average
Average
BOM Cost
Low
Phase # Dependent
High
Complexity
High
High
Low
Density
Low
Phase # Dependent
High
Performance
Average
Average
High
For the current design, Option 2 can be eliminated right away. A controller and driver package that can
handle six phases does not exist and stacking multiple controllers adds unneeded complexity when
controllers exist with six PWM outputs. Option 1 looks attractive because of the potential for a cheaper
BOM cost but the PCB area needed to layout a driver, FETs, and associated passives, multiplied by six
phases increases the board area and raises the cost of its production and assembly.
Choosing Option 3 reduces the overall component count and provides for the simplest board layout. It also
eliminates the challenge of selecting an optimal pair of FETs and drivers to use for each phase, a topic
that merits its own application note (Multiphase Buck Regulator Portal). Choosing a Smart Power Stage
provides support for PMBus telemetry by integrating the needed circuitry on the chip.
Two possible options for power stages to consider for this design are the CSD95372AQ5M and the
CSD95490Q5MC. Each stage is rated for a continuous current of 60 A and 75 A respectively, and
supports the input/output voltages required, can switch at 600 kHz, and has a built in temperature monitor
pin. Both parts come in low inductance packages to reduce parasitics that can affect steady-state
switching and transient response. Finally, both are compatible with 3.3-V and 5-V PWM signals allowing
for more flexibility when choosing a controller IC.
Upon closer inspection, the CSD95490Q5MC proves to be a better fit for powering the networking ASIC.
No DCR matching or resistor sense filter circuit is needed, thanks to the integrated bi-directional currentsense capability, removing six differential current sense signals routed back to the controller. An amplified,
single-ended, current sense signal per phase is reported back instead. Because this current sense signal
is amplified at the power stage it is much less susceptible to corruption from noise and other switching
signals simplifying the circuit layout. A single resistor value on the LSET pin is all that is needed to
properly configure this part. Additionally, a small amount of power loss is eliminated because a minimum
sense resistor or DCR value is no longer needed to keep the sense signal SNR high enough to accurately
balance the phase currents.
Most importantly, the CSD95490Q5MC has much lower power loss than the CSD95372AQ5M under
identical conditions. Power loss is calculated at 33 A (TDC) and 40 A (maximum) and shown in Table 3
using the loss curves in both data sheets for the following conditions: VIN = 12 V, VOUT = 0.9 V, fSW = 600
kHz, L = 150 nH, TJ = 100°C. With losses 1.4 W less per phase at TDC and 3 W less at maximum current,
the CSD95490Q5MC is the clear choice.
Table 3. Power Stage Loss Calculations per Phase
5.4
Phase Current
CSD95490Q5MC
CSD95372AQ5M
33 A (TDC)
3.36 W
4.71 W
40 A (MAX)
4.56 W
7.54 W
Input Capacitors
Typically input capacitor requirements are met via a combination of multi-layer ceramic capacitors
(MLCCs) and either aluminum or polymer electrolytic bulk capacitors. The MLCCs are sized to handle the
RMS current and DC ripple in steady-state conditions while the bulk capacitance is used to provide charge
and keep VIN within tolerance during load transients.
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To calculate the number of MLCCs simply multiply the RMS current value calculated from Equation 1 by
the maximum current and divide by RMS current rating for an individual MLCC, rounding up to the nearest
whole number. Capacitor current rating can be obtained from the manufacturer’s website. The RMS input
current for this application is 19.9 Arms. A 22-µF, X5R, 1210, 16-V capacitor is rated at approximately 5 A
of RMS ripple current at 600 kHz with a 20°C rise. Under these conditions, four total capacitors would be
needed to carry the current.
Equation 4 calculates the amount of ceramic capacitance per phase needed to keep the input voltage
ripple within its limits. In order to get a better estimate of the capacitance required the duty cycle can be
divided by the target efficiency, η, at the maximum phase current in order to get an adjusted duty cycle
term, DADJ.
I PHASEmax u DADJ u n 1 DADJ
40 A u 0.0882 u 1 0.0882
C INphase
22.3 µF
fSW u 'VIN DC
600 kHz u 240 mVpp
where
•
DADJ = VOUT / VIN × η
(4)
Assuming a conservative efficiency of 85%, η = 0.85, at 40 A, a minimum of 22 µF is needed to keep VIN
within tolerance. You may initially think only one ceramic capacitor is needed per phase to hit both the
ripple and RMS current requirements but the derating of each capacitor as a function of the DC bias
voltage proves otherwise. From Figure 10, a single 1210, 22-µF capacitor derates to approximately 15 µF
with a 12-V bias. Taking this into account, two 22-µF capacitors per phase are needed to meet the input
ripple requirements. Using identical capacitors from the same vendor but in a smaller 1206 package, a 22µF capacitor derates to about 5 µF at 12 V, requiring four capacitors per phase instead of two.
Figure 10. Capacitor Derating Curves Courtesy of Murata
Left:1210 Case, GRM32ER61C226ME20L, Right: 1206 Case, GRM31CR61C226ME15
Choosing a bulk capacitor to decouple the input voltage is more of an art than a science. Equations can
give an engineer a starting point for a design but ultimately the performance must be verified on the board
during validation. A tradeoff must be made between minimizing the ESR spike caused by the bulk
capacitor while at the same time maintaining a high enough resistance to dampen any oscillations caused
by ceramic capacitor ringing during a transient.
For this design, the process outlined in the How to Select Input Capacitors for a Buck Converter Technical
Brief is used to get a starting bulk capacitance value assuming a 10-kHz bandwidth for the 12-V bus
regulator. After completing the process, 550 µF should be the minimum capacitance with an ESR of less
than 27 mΩ. Two 330-µF, 16-V, 20-mΩ Aluminum polymer capacitors are used as bulk decoupling on VIN.
Additionally, a single 0.33-µF, 0603 ceramic capacitor is placed on each phase to help suppress ringing
on the phase node and reduce the requirements of a snubbing circuit should testing reveal one to be
needed.
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Multiphase Design Example - Component Selection
5.5
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Output Capacitors
Calculating the output capacitance requires taking into account both the DC ripple and AC transient
specifications of an application. As previously discussed, the AC transient requirements are typically more
demanding than the DC ripple specifications and dictate how much total output capacitance is needed.
Just as when choosing input capacitors, a mix of MLCCs and bulk caps are used.
Ceramic capacitors keep the output impedance of the converter low before the control loop can respond
during fast transients, minimizing overshoot and undershoot. Bulk capacitors provide enough of a charge
reservoir for the output voltage to stay within tolerance as the controller ramps the inductor current the
new load current level.
Assuming minimal ESR and ESL in the capacitor network, the amount of output capacitance needed to
handle the DC ripple can be calculated using Equation 5. In this equation, IPP is the ripple current for a
single phase of the converter (calculated using the 150nH inductor value) as there is no inductor current
cancellation in single-phase operation making it the worst-case scenario.
I PP
9.25 APP
COUT,Ripple
214 µF
8 u fsw u 'VOUT DC
8 u 600 kHz u 0.01 u 0.9 V
(5)
Figure 11 and Equation 6 to Equation 9 explain the process behind calculating starting capacitance values
needed to handle load transients. During a load step the inductance, L or LEQ – depending on the total
phase number – takes some amount of time, tUndershoot, to slew to the high current level. In that time, an
amount of charge equal to QUndershoot is pulled from the output capacitors while VOUT dips below its set point.
Upon load release, excess charge in the inductor, QOvershoot, is dumped into the output capacitors during
time tOvershoot, causing VOUT to swing above its regulation point.
Figure 11. Load Transient Waveforms
t UNDERSHOOT
Q UNDERSHOOT
t OVERSHOOT
Q OVERSHOOT
L EQ u I STEP
V IN
V OUT
150 nH
u 150 A
6
12 V 0.9 V
338 ns
(6)
1
1
u t UNDERSHOOT u I STEP
u 438 ns u 150 A
2
2
150 nH
u 150 A
L EQ u ISTEP
6
4.16 µs
VOUT
0.9 V
1
u t OVERSHOOT u I STEP
2
1
u 4.3 µs u 150 A
2
25.35 µC
(7)
(8)
312.5 µC
(9)
After calculating QOvershoot and QUndershoot, finding the output capacitance is simply a matter of dividing the
charge by the allowable swing on VOUT. The current design specifies a DC load line which must be taken
into account as shown in Figure 12. The total capacitance needed to handle the maximum transient of the
application is calculated in Equation 10 for the load step and Equation 11 for the load release. For
applications without a DC load line, simply set DCLL = 0.
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Figure 12. Load Transient with DC Load Line
CUNDERSHOOT
COVERSHOOT
QUNDERSHOOT
'VOUT AC
ISTEP u DCLL
QOVERSHOOT
'VOUT AC ISTEP u DCLL
25.35 µC
0.05 u 0.9V 150 A u 0.5 m
312.5 µC
0.05 u 0.9 V 150 A u 0.5 m
211.1 µF
(10)
2,604 µF
(11)
Comparing the values calculated for CRipple, CUndershoot, and COvershoot, the load release dictates the amount of
capacitance needed to keep VOUT within regulation. COvershoot comes out to be much greater than CUndershoot
because during load release, less energy is required by the processor and so any excess stored in the
inductor gets transferred to the output capacitors causing VOUT to overshoot. During a load step the
processor is pulling energy from the capacitors and the energy stored in the inductor refills them helping
mitigate undershoot.
Table 4 and Table 5 are used to come up with a mix of output capacitors that can satisfy the transient
requirements while balancing component count and BOM cost. Table 4 compares the prices and
specifications of several popular capacitor options while Table 5 looks at combinations of capacitors that
meet the necessary requirements and can be used as a starting point for the design. Depending on bench
results, the amount and type of capacitors may be adjusted. The total capacitance of each option is set
higher than COvershoot to provide margin and account for derating on the MLCCs. Since the DC bias on each
capacitor is lower than on the input side of the regulator, less derating occurs and the capacitors still retain
most of their nominal capacitance.
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Multiphase Design Example - Component Selection
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Table 4. Output Capacitor Options
Capacitor Type
Capacitance
Specifications
Price/1000 Units
Ceramic
22 µF
0805, 6.3 V, X5R
$0.054
Ceramic
47 µF
0805, 6.3 V, X5R
$0.131
Organic Polymer
470 µF
V-Case, 2.5 V, 6 mΩ
$1.357
Organic Polymer
680 µF
D-Case, 2.5 V, 6 mΩ
$2.537
Table 5. Output Capacitor Solution Comparison
Capacitor Mix
Total Capacitance
Component Count
Price
3 × 470 µF + 20 × 47 µF + 25 × 22 µF
2900 µF
48
$8.04
1 × 680 µF + 32 × 47 µF + 35 × 22 µF
2950 µF
68
$8.62
2 × 680 µF + 20 × 47 µF + 20 × 22 µF
2850 µF
47
$9.04
47 × 47 µF + 35 × 22 µF
2980 µF
82
$8.05
From Table 5, a combination of 470-µF bulk capacitors and MLCCs provide the best balance between
component count and price. For applications that may require an all ceramic solution, the component
count increases substantially though not necessarily at the expense of BOM cost.
5.6
Controller
Studying the TPS53679 Dual-Channel Multiphase Controller data sheet (SLUSC47) proves it to be a good
fit for this ASIC core rail. The D-CAP+ modulator is optimized for multiphase control and keeping the
current balanced between phases. Six PWM channels offer a great deal of design flexibility to work with a
variety of power stages, including the chosen CSD95490, while minimizing the size of the controller
package. Support for PMBus communication checks the box to meet the telemetry specification of the
design. The PMBus also enables tuning functionality of the phase add and drop points so that optimal
efficiency can be achieved over the whole load range. For a deeper look into the D-CAP+ modulator see
the Synchronous Buck NexFET Power Stage, CSD95372AQ5M Data Sheet and Enabling Loadline for
Memory and ASIC VR Applications to Save Output Capacitors Application Report.
As an added bonus, the controller also supports full digital compensation through the PMBus making
tuning the design on the board much easier than reworking components on an analog compensation pin.
Finally, the second single-phase buck regulator can be used to power any auxiliary rails that the ASIC
may require saving money and PCB area.
5.7
Design Summary
Table 6 gives a comparison of the current six-phase design compared to alternatives using one, two, or
four phases with the same power stage and inductor. Fewer phases are not a feasible option for this
design when looking at the results. Power loss can be mitigated to some degree by selecting components
rated to the higher currents but between component cost, power loss concentration, plus modifications to
fans and heatsinks, any benefits from these changes are likely be equivalent when compared to a sixphase solution.
The output capacitance to hit the overshoot requirement drops by thousands of micro-Farads as the
phase count increases. Input ceramic capacitor count is also more manageable with a higher phase count.
As an academic exercise, the benefit of a DC load line is shown for each case by recalculating the value
of COvershoot after setting DCLL = 0 from Equation 11. Without a load line, VOUT cannot swing more than 45
mV, 5%, in either direction during a 150-A transient. The ability of the ASIC to handle a 0.5-mΩ load line
on its core voltage rail allows VOUT to swing an additional 75 mV for the same transient for a total of 120
mV, drastically reducing the output capacitance.
16
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Table 6. Multiphase Design Comparison
Phases
1
2
4
6
IIN (Arms)
63.2
42.8
27.5
19.9
RMS Input Current
IMAX,PH (A)
240.0
120.0
60.0
40.0
Max Current per Phase
ITDC,PH (A)
200
100
50
40
PFET,TDC (W)
-
-
6.81
3.36
Thermal Design Current per Phase
FET Loss @ TDC
PIND,TDC (W)
-
7.04
2.07
1.15
Inductor Loss @ TDC
CIN,MLCC (µF)
134.1
57.0
33.5
22.3
Ceramic Input Capacitance per Phase
COvershoot (µF)
15 625
7812
3906
2604
Output Capacitance to Meet Overshoot
COvershoot (µF)
41 666
20 833
10 416
6944
Output Capacitance to Meet Overshoot, no load line
Table 7 displays a brief summary of the major design decisions and components selected for this case
study. These components are used in Part 2 of this multiphase series when the PCB is laid out and tested
in the lab.
Table 7. Case Study Design Summary
VIN
12 V
VOUT
0.9 V
IMAX
240 A
TDC
200 A
Phase Count
6
Inductor
150nH, 0.53 mΩ, 55 A ITEMP
FETs
CSD95490
TDC Power Loss
FETs - 20.1 W
Inductors - 6.87 W
TDC Eff. Estimate
86.9%
CIN
2 × 330 µF, 10 mΩ, 16 V, Al Poly
12 × 22 µF, 1210, X5R, 16 V
COUT
3 × 470 µF, 6 mΩ, 6.3 V
20 × 47 µF, 0805, X5R, 2.5 V
25 × 22 µF, 0805, X5R, 6.3 V
Controller
6
TPS53679
Conclusion
After an introduction to the pros and cons of multiphase regulators, a paper design of a high-performance,
six-phase buck has been completed. During the design tradeoffs between component count, power loss,
ease of design, and BOM cost were made resulting in an optimal solution. Looking forward to the next
portion of the tutorial, a PCB based on this design is completed and tested on the bench against the target
specifications. For more information on TI’s multiphase controllers, both with and without PMBus, visit the
web portal referred to in the D-CAP+TM Control for Multiphase Step-Down Voltage Regulators for
Powering Microprocessors Application Report.
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17
References
7
References
•
•
•
•
•
•
•
•
•
•
•
•
•
18
www.ti.com
Texas Instruments, Benefits of a Multiphase Buck Converter Technical Brief (SLYT449)
Texas Instruments, Choosing the Right Variable Frequency Buck Regulator Control Strategy White
Paper (SLUP319)
Texas Instruments, Synchronous Buck NexFET Power Stage, CSD95372AQ5M Data Sheet
(SLPS416)
Texas Instruments, How to Select Input Capacitors for a Buck Converter Technical Brief (SLYT670)
Texas Instruments, D-CAP+TM Control for Multiphase Step-Down Voltage Regulators for Powering
Microprocessors Application Report (SLVA867)
Texas Instruments, Enabling Loadline for Memory and ASIC VR Applications to Save Output
Capacitors Application Report (SLUA819)
Texas Instruments, CSD95490Q5MC Synchronous Buck NexFET™ Smart Power Stage Data Sheet
(SLPS669)
Texas Instruments, Power Loss Calculation With CSI Consideration for Synchronous Buck Converters
Application Note (SLPA009)
Multiphase Buck Regulator Portal
IHLP Inductor Loss Calculator Tool
Introduction to the PMBus
Under the Hood of a DC/DC Boost Converter Seminar
Vishay Dale, “Low Profile, High Current IHLP Inductors,” data sheet 34123, 2016
Multiphase Buck Design From Start to Finish (Part 1)
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Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (April 2017) to A Revision .......................................................................................................... Page
•
•
•
•
•
Edited application report for clarity. ..................................................................................................... 1
Edited list of performance advantages. ................................................................................................ 4
Changed "out power" to "output power" ................................................................................................ 7
Added "(calculated using the 150 nH inductor value)". ............................................................................. 14
Changed the bottom rows in the Multiphase Design Comparison table. ......................................................... 17
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