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Texas Instruments PSR Flyback DC/DC Converter Transformer Design for mHEV Applications Application notes
PSR Flyback DC/DC Converter Transformer Design for
mHEV Applications
Timothy Hegarty
Magnetic component design is an important aspect
when implementing isolated DC/DC converters. For
the primary-side regulated (PSR) flyback converter in
particular, the transformer plays a critical role as it sets
the flyback operating mode boundaries and has an
outsized impact on the performance of the converter.
This tech note describes a condensed transformer
design procedure for a low-power PSR flyback DC/DC
converter in an automotive application. The steps are
useful whether the transformer design is carried out inhouse or outsourced to a magnetic component vendor.
mHEV Power Solution
Instead, the device senses the reflected isolated
output voltage from the primary-side flyback voltage
waveform, resulting in accurate load and line
regulation performance. This simplifies the design,
enables a smaller solution size and requires only one
component crossing the isolation barrier.
VIN = 4.5 V...42 V
T1
DFLY
VOUT = 12 V
DZ
CIN
EN/UVLO
As an example, consider a mild-hybrid electric vehicle
(mHEV) system with 12 V and 48 V batteries as
shown in Figure 1. The isolated DC/DC power supply
highlighted in red provides a tightly-regulated 12-V
bias rail on the 48-V side.
1:1
DF
VIN
10 F
COUT
22 F
SW
LM25180-Q1
RFB
124 k:
Option for ±12V output
DFLY2 V
OUT2 = ±12 V
FB
GND
RSET
COUT2
22 F
RSET
SS/BIAS
12.1 k:
TC
Copyright © 2019, Texas Instruments Incorporated
Figure 2. PSR Flyback DC/DC Converter Schematic
Designing a Flyback Magnetic Component
The following outlines a 6-step transformer design
procedure tailored for a PSR flyback DC/DC converter.
1. Define Specification
Table 1 gives the relevant design parameters for this
mHEV application example.
Table 1. PSR Flyback Specifications
SYMBOL
Figure 1. mHEV System Block Diagram
Figure 2 shows the LM25180-Q1 PSR flyback DC/DC
converter schematic with a 12-V output up to 200 mA.
Additional outputs are easily configured depending on
the application requirements. The schematic above
includes an optional negative output if bipolar output
rails (±12 V) are required.
Note that the LM25180-Q1 does not need an
optocoupler, voltage reference or transformer auxiliary
winding for output voltage regulation.
Min, nom, max input voltage
VOUT, IOUT
Output voltage and current
VALUE
5.5 V, 13.5 V, 42 V
12 V, 0.2 A
100 kHz (BCM), 220
kHz (BCM), 350 kHz
(DCM)
FSW
Full-load switching frequency
at min/nom/max VIN
IPRI-PK
Peak primary current
1.2 A
VD
Flyback diode voltage drop
0.4 V
Equation 1 gives the duty cycle of a flyback converter
when operating in boundary conduction mode (BCM).
D
SNVA805 – April 2019
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PARAMETER
VIN
VOUT
VIN
VD ˜ NPS
VOUT
VD ˜ NPS
(1)
PSR Flyback DC/DC Converter Transformer Design for mHEV Applications
Copyright © 2019, Texas Instruments Incorporated
Timothy Hegarty
1
www.ti.com
Assuming a maximum duty cycle of 70% at minimum
input voltage, find an initial estimate for the
transformer turns ratio using Equation 2.
NPS
VIN(min)
DMAX
˜
1 DMAX VOUT VD
0.7
5.5 V
˜
|1
1 0.7 12 V + 0.4 V
Find the total copper loss using Equation 5 as 0.2 W.
PCU
PCU-PRI
RPRI ˜ IPRI-RMS
PCU-SEC
1 ª
2
˜ D ˜ RPRI ˜ IPRI-PK
3 ¬«
2
RSEC ˜ ISEC-RMS
1 D ˜ RSEC ˜ NPS ˜ IPRI-PK
2º
»¼
(5)
(2)
2. Core Selection
5. Calculate Flux Density and Core Loss
Select a core based on the required output power.
Refer to section 4 of the Magnetics Design Handbook
for more detail. Choose an EP7 ferrite core for this
application with relevant parameters given in Table 2.
Given the minimum cross-sectional area of the
selected core, calculate the peak flux density in an
overcurrent (OC) condition using Equation 6 and
ensure that it is less than the saturation level of the
core material, typically 250 mT or higher for a ferrite.
The flux cycles from zero to peak in the first quadrant
of the B-H curve of the core.
Table 2. EP7 Ferrite Core Parameters
SYMBOL
PARAMETER
VALUE
Ae
Effective area (mm2)
Amin
Minimum area (mm2)
8.65
Ve
Effective volume (mm3)
165
le
Effective length (mm)
15.5
10.7
BPK
BAC
To ensure adequate time to reset the magnetizing
current to zero, Equation 3 determines the minimum
magnetizing inductance using current and off-time
parameters specific to the LM25180-Q1.
VOUT ˜ NPS ˜ t OFF(min)
IPRI-PK(FFM)
12 V ˜ 1˜ 0.45 V
0.3 A
18 +
A magnetizing inductance of 30 µH provides an
acceptable design margin and enables the converter
to operate in BCM at a lower switching frequency for a
greater portion of the load and line range.
Using a core with 880-µm airgap that sets an
inductance factor AL of 25 nH per turn squared,
calculate the number of primary turns using
Equation 4. Given the unity turns ratio, the number of
secondary turns is also 36.
LMAG
AL
30 +
25nH/Turns
2
| 36
(4)
4. Calculate Copper Loss
Determine the appropriate wire gauge and number of
paralleled strands using the bobbin fit factor and
buildup calculations found in the Practical Magnetic
Design: Inductor and Coupled Inductors Seminar.
As described there, derive the AC/DC wire resistance
ratio due to skin effect at the applicable frequency.
Selecting 34 AWG wire size for both primary and
secondary, calculate the primary and secondary
winding resistances as 180 mΩ based on the mean
length of the bobbin per turn (MLT) of 17.9 mm.
2
30 + ˜ $
NP ˜ Amin
36 ˜ 8.5mm
LMAG ˜ IPRI-PK
2 ˜ NP ˜ A e
(6)
30 + ˜
$
2 ˜ 36 ˜ 10.7mm
47mT
2
(7)
Identify a specific power loss of 40 kW/m using flux
density and frequency as parameters in the applicable
characterization plot of the core vendor. Multiply by the
effective volume to obtain a core loss of 7 mW.
5. Calculate Temperature Rise
Given a total power loss of 207 mW, use the core and
bobbin effective thermal impedance of 40°C/W to
estimate a temperature rise of 8°C.
Iterate these calculations as needed to obtain a
transformer design optimized for the desired input
voltage and ambient temperature ranges.
6. Construct Transformer
Figure 3 shows a transformer winding construction
with single-filar primary and secondary windings. The
split-primary winding sandwiches two secondary layers
to obtain a low leakage inductance of 300 nH.
18T primary,
outer layer
36T secondary
wound in two layers
18T primary,
inner layer
19
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28
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30
31
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36
19
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18
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2
1
1
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18
Two
layers
of tape
Bobbin
CORE
Figure 3. Interleaved Transformer Construction
with Split-primary Winding
As functional grade isolation is typically sufficient for
mHEV applications, two layers of tape are placed
between adjacent primary and secondary layers.
PSR Flyback DC/DC Converter Transformer Design for mHEV Applications
Timothy Hegarty
196mT
2
3
(3)
NP
LMAG ˜ IPRI-PK(OC)
Find the flux density swing for core loss calculations
using Equation 7.
3. Calculate Inductance
LMAG t
2
Copyright © 2019, Texas Instruments Incorporated
SNVA805 – April 2019
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