TPA3112D1 - Texas Instruments

TPA3112D1 - Texas Instruments
TPA3112D1
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SLOS654C – SEPTEMBER 2009 – REVISED JULY 2012
25-W FILTER-FREE MONO CLASS-D AUDIO POWER AMPLIFIER with SPEAKER GUARD™
Check for Samples: TPA3112D1
FEATURES
1
•
2
•
•
•
•
•
•
•
•
•
•
25-W into an 8-Ω Load at < 0.1% THD+N From
a 24-V Supply
20-W into an 4-Ω Load at 10% THD+N From a
12-V Supply
94% Efficient Class-D Operation into 8-Ω Load
Eliminates Need for Heat Sinks
Wide Supply Voltage Range Allows Operation
from 8 to 26 V
Filter-Free Operation
SpeakerGuard™ Speaker Protection Includes
Adjustable Power Limiter plus DC Protection
Flow Through Pin Out Facilitates Easy Board
Layout
Robust Pin-to-Pin Short Circuit Protection and
Thermal Protection with Auto-Recovery Option
Excellent THD+N/ Pop Free Performance
Four Selectable, Fixed Gain Settings
Differential Inputs
APPLICATIONS
•
•
DESCRIPTION
The TPA3112D1 is a 25-W efficient, Class-D audio
power amplifier for driving a bridge tied speaker.
Advanced EMI Suppression Technology enables the
use of inexpensive ferrite bead filters at the outputs
while meeting EMC requirements. SpeakerGuard™
speaker protection system includes an adjustable
power limiter and a DC detection circuit. The
adjustable power limiter allows the user to set a
"virtual" voltage rail lower than the chip supply to limit
the amount of current through the speaker. The DC
detect circuit measures the frequency and amplitude
of the PWM signal and shuts off the output stage if
the input capacitors are damaged or shorts exist on
the inputs.
The TPA3112D1 can drive a mono speaker as low as
4Ω. The high efficiency of the TPA3112D1, > 90%,
eliminates the need for an external heat sink when
playing music.
The outputs are fully protected against shorts to
GND, VCC, and output-to-output. The short-circuit
protection and thermal protection includes an autorecovery feature.
Televisions
Consumer Audio Equipment
1uF
Audio
Source
OUT+
INP
OUT -
INN
TPA3112D1
OUTP
FERRITE
BEAD
FILTER
OUTN
25W
8Ω
GAIN0
GAIN1
PLIMIT
Fault
SD
PVCC
8 to 26V
Figure 1. Simplified Application Diagram
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
SpeakerGuard, PowerPad are trademarks of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009–2012, Texas Instruments Incorporated
TPA3112D1
SLOS654C – SEPTEMBER 2009 – REVISED JULY 2012
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
UNIT
VCC
Supply voltage
AVCC, PVCC
–0.3 V to 30 V
SD, FAULT,GAIN0, GAIN1, AVCC (Pin 14)
VI
Interface pin voltage
PLIMIT
–0.3 V to VCC + 0.3 V
< 10 V/ms
–0.3 V toGVDD + 0.3 V
INN, INP
–0.3 V to 6.3 V
Continuous total power dissipation
See Thermal Inforamtion Table
TA
Operating free-air temperature range
–40°C to 85°C
TJ
Operating junction temperature range (2)
–40°C to 150°C
Tstg
Storage temperature range
–65°C to 150°C
RL
Minimum Load Resistance
BTL
Human body model
Electrostatic discharge
(1)
(2)
(3)
(4)
3.2
(3)
Charged-device model
(all pins)
(4)
±2 kV
(all pins)
±500 V
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operations of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
The TPA3112D1 incorporates an exposed thermal pad on the underside of the chip. This acts as a heatsink, and it must be connected
to a thermally dissipating plane for proper power dissipation. Failure to do so may result in the device going into thermal protection
shutdown. See TI Technical Briefs SCBA017D and SLUA271 for more information about using the QFN thermal pad. See TI Technical
Briefs SLMA002 for more information about using the HTQFP thermal pad.
In accordance with JEDEC Standard 22, Test Method A114-B.
In accordance with JEDEC Standard 22, Test Method C101-A
THERMAL INFORMATION
THERMAL METRIC (1) (2)
TPA3112D1
PWP (28 PINS)
θJA
Junction-to-ambient thermal resistance
30.3
θJCtop
Junction-to-case (top) thermal resistance
33.5
θJB
Junction-to-board thermal resistance
17.5
ψJT
Junction-to-top characterization parameter
0.9
ψJB
Junction-to-board characterization parameter
7.2
θJCbot
Junction-to-case (bottom) thermal resistance
0.9
(1)
(2)
2
UNITS
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
For thermal estimates of this device based on PCB copper area, see the TI PCB Thermal Calculator
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SLOS654C – SEPTEMBER 2009 – REVISED JULY 2012
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
MAX
26
UNIT
VCC
Supply voltage
PVCC, AVCC
8
VIH
High-level input voltage
SD, GAIN0, GAIN1
2
V
VIL
Low-level input voltage
SD, GAIN0, GAIN1
0.8
V
VOL
Low-level output voltage
FAULT, RPULLUP=100kΩ, VCC=26V
0.8
V
IIH
High-level input current
SD, GAIN0, GAIN1, VI = 2, VCC = 18 V
50
µA
IIL
Low-level input current
SD, GAIN0, GAIN1, VI = 0.8V, VCC = 18 V
5
µA
TA
Operating free-air temperature
85
°C
V
–40
DC CHARACTERISTICS
TA = 25°C, VCC = 24 V, RL = 8 Ω (unless otherwise noted)
PARAMETER
TEST CONDITIONS
| VOS |
Class-D output offset voltage (measured
differentially)
VI = 0 V, Gain = 36 dB
ICC
Quiescent supply current
SD = 2 V, no load, PVcc=21V
ICC(SD)
Quiescent supply current in shutdown mode
SD = 0.8 V, no load, PVcc=21V
rDS(on)
IO = 500 mA,
TJ = 25°C
Drain-source on-state resistance
GAIN1 = 0.8 V
G
Gain
GAIN1 = 2 V
tON
Turn-on time
SD = 2 V
tOFF
Turn-off time
SD = 0.8 V
GVDD
Gate Drive Supply
IGVDD = 2mA
MIN
TYP MAX
1.5
15
mV
40
mA
400
µA
High Side
240
Low side
240
mΩ
GAIN0 = 0.8 V
19
20
21
GAIN0 = 2 V
25
26
27
GAIN0 = 0.8 V
31
32
33
GAIN0 = 2 V
35
36
37
6.5
UNIT
dB
dB
10
ms
2
μs
6.9
7.3
V
DC CHARACTERISTICS
TA = 25°C, VCC = 12 V, RL = 8 Ω (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
| VOS |
Class-D output offset voltage (measured
differentially)
VI = 0 V, Gain = 36 dB
1.5
ICC
Quiescent supply current
SD = 2 V, no load, PVcc=12V
20
mA
ICC(SD)
Quiescent supply current in shutdown mode
SD = 0.8 V, no load, PVcc=12V
200
µA
Drain-source on-state resistance
IO = 500 mA,
TJ = 25°C
rDS(on)
GAIN1 = 0.8 V
G
Gain
GAIN1 = 2 V
High Side
240
Low side
240
15
mΩ
GAIN0 = 0.8 V
19
20
21
GAIN0 = 2 V
25
26
27
GAIN0 = 0.8 V
31
32
33
GAIN0 = 2 V
35
36
37
tON
Turn-on time
SD = 2 V
tOFF
Turn-off time
SD = 0.8 V
GVDD
Gate Drive Supply
IGVDD = 2mA
PLIMIT
Output Voltage maximum under PLIMIT
control
VPLIMIT=2.0 V; VI=6.0V differential
10
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dB
dB
ms
μs
2
6.5
6.9
7.3
V
6.75
7.90
8.75
V
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mV
3
TPA3112D1
SLOS654C – SEPTEMBER 2009 – REVISED JULY 2012
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AC CHARACTERISTICS
TA = 25°C, VCC = 24 V, RL = 8 Ω (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
KSVR
Power Supply ripple rejection
200 mVPP ripple from 20 Hz–1 kHz,
Gain = 20 dB, Inputs ac-coupled to AGND
PO
Continuous output power
THD+N ≤ 0.1%, f = 1 kHz, VCC = 24 V
THD+N
Total harmonic distortion + noise
VCC = 24 V, f = 1 kHz, PO = 12 W (half-power)
Vn
Output integrated noise
20 Hz to 22 kHz, A-weighted filter, Gain = 20 dB
Crosstalk
SNR
Signal-to-noise ratio
fOSC
Oscillator frequency
TYP
MAX
UNIT
–70
dB
25
W
<0.05
%
65
µV
–80
dBV
VO = 1 Vrms, Gain = 20 dB, f = 1 kHz
–70
dB
Maximum output at THD+N < 1%, f = 1 kHz,
Gain = 20 dB, A-weighted
102
dB
250
Thermal trip point
Thermal hysteresis
310
350
kHz
150
°C
15
°C
AC CHARACTERISTICS
TA = 25°C, VCC = 12 V, RL = 8 Ω (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
KSVR
Supply ripple rejection
200 mVPP ripple from 20 Hz–1 kHz,
Gain = 20 dB, Inputs ac-coupled to AGND
PO
Continuous output power
PO
Continuous output power
THD+N
Total harmonic distortion + noise
RL = 8 Ω, f = 1 kHz, PO = 5 W (half-power)
Vn
TYP
MAX
UNIT
–70
dB
THD+N ≤ 10%, f = 1 kHz , RL = 8Ω
10
W
THD+N ≤ 10%, f = 1 kHz , RL = 4Ω
20
W
<0.06
%
65
µV
–80
dBV
Output integrated noise
20 Hz to 22 kHz, A-weighted filter, Gain = 20 dB
Crosstalk
Po = 1 W, Gain = 20 dB, f = 1 kHz
–70
dB
SNR
Signal-to-noise ratio
Maximum output at THD+N < 1%, f = 1 kHz,
Gain = 20 dB, A-weighted
102
dB
fOSC
Oscillator frequency
250
Thermal trip point
Thermal hysteresis
310
350
kHz
150
°C
15
°C
PWP (TSSOP) Package
(Top View)
4
SD
FAULT
1
28
2
27
GND
GND
GAIN0
GAIN1
3
26
4
25
5
24
6
23
AVCC
AGND
GVDD
PLIMIT
7
22
8
21
INN
INP
NC
AVCC
9
20
10
19
11
18
12
17
13
16
14
15
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PVCC
PVCC
BSN
OUTN
PGND
OUTN
BSN
BSP
OUTP
PGND
OUTP
BSP
PVCC
PVCC
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SLOS654C – SEPTEMBER 2009 – REVISED JULY 2012
PIN FUNCTIONS
PIN
NAME
SD
Pin #
1
I/O
DESCRIPTION
I
Shutdown logic input for audio amp(LOW = outputs Hi-Z, HIGH = outputs enabled).
TTL logic levels with compliance to AVCC.
O
Open drain output used to display short circuit or dc detect fault status. Voltage
compliant to AVCC. Short circuit faults can be set to auto-recovery by connecting
FAULT pin to SD pin. Otherwise both the short circuit faults and dc detect faults
must be reset by cycling PVCC.
FAULT
2
GND
3
GND
4
GAIN0
5
I
Gain select least significant bit. TTL logic levels with compliance to AVCC.
GAIN1
6
I
Gain select most significant bit. TTL logic levels with compliance to AVCC.
AVCC
7
P
Analog supply.
AGND
8
GVDD
9
O
High-side FET gate drive supply. Nominal voltage is 7V. May also be used as
supply for PLILMIT divider. Add a 1μF cap to ground at this pin.
PLIMIT
10
I
Power limit level adjust. Connect directly to GVDD pin for no power limiting. Add a
1μF cap to ground at this pin.
INN
11
I
Negative audio input. Biased at 3V.
INP
12
I
Positive audio input. Biased at 3V.
NC
13
AVCC
14
P
Connect AVCC supply to this pin
PVCC
15
P
Power supply for H-bridge. PVCC pins are also connected internally.
PVCC
16
P
Power supply for H-bridge. PVCC pins are also connected internally.
BSP
17
I
Bootstrap I/O for positive high-side FET.
OUTP
18
O
Class-D H-bridge positive output.
PGND
19
OUTP
20
O
Class-D H-bridge positive output.
BSP
21
I
Bootstrap I/O for positive high-side FET.
BSN
22
I
Bootstrap I/O for negative high-side FET.
OUTN
23
O
Class-D H-bridge negative output.
PGND
24
OUTN
25
O
Class-D H-bridge negative output.
BSN
26
I
Bootstrap I/O for negative high-side FET.
PVCC
27
P
Power supply for H-bridge. PVCC pins are also connected internally.
PVCC
28
P
Power supply for H-bridge. PVCC pins are also connected internally.
Connect to local ground
Connect to local ground
Analog supply ground. Connect to the thermal pad.
Not connected
Power ground for the H-bridges.
Power ground for the H-bridges.
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FUNCTIONAL BLOCK DIAGRAM
GVDD
PVCC
BSP
PVCC
OUTP FB
OUTP FB
INP
Gain
Control
PWM
Logic
PLIMIT
Gate
Drive
OUTP
INN
OUTN FB
PGND
FAULT
SD
GAIN0
TTL
Buffer
Gain
Control
GAIN1
PLIMIT
Reference
PLIMIT
GVDD
AVDD
AVCC
PVCC
BSN
PVCC
LDO
Regulator
SC Detect
GVDD
DC Detect
GVDD
Ramp
Generator
Biases and
References
Startup Protection
Logic
Thermal
Detect
Gate
Drive
OUTN
OUTN FB
UVLO/OVLO
PGND
AGND
6
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TYPICAL CHARACTERISTICS
(All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3112D2 EVM which is
available at ti.com.)
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
10
Gain = 20 dB
VCC = 12 V
ZL = 8 Ω + 66 µH
THD − Total Harmonic Distortion − %
THD − Total Harmonic Distortion − %
10
1
0.1
PO = 1 W
0.01
PO = 5 W
Gain = 20 dB
VCC = 24 V
ZL = 8 Ω + 66 µH
1
PO = 1 W
0.1
0.01
PO = 10 W
PO = 5 W
PO = 2.5 W
0.001
20
100
1k
10k
0.001
20
20k
100
1k
f − Frequency − Hz
10k
G001
G002
Figure 2.
Figure 3.
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT POWER
10
Gain = 20 dB
VCC = 12 V
ZL = 4 Ω + 33 µH
THD+N − Total Harmonic Distortion + Noise − %
THD − Total Harmonic Distortion − %
10
1
PO = 5 W
PO = 10 W
0.1
0.01
0.001
20
20k
f − Frequency − Hz
PO = 1 W
100
1k
10k
20k
Gain = 20 dB
VCC = 12 V
ZL = 8 Ω + 66 µH
1
f = 1 kHz
f = 20 Hz
0.1
0.01
f = 10 kHz
0.001
0.01
f − Frequency − Hz
G003
Figure 4.
0.1
1
10
PO − Output Power − W
30
G004
Figure 5.
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TYPICAL CHARACTERISTICS (continued)
(All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3112D2 EVM which is
available at ti.com.)
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT POWER
10
Gain = 20 dB
VCC = 24 V
ZL = 8 Ω + 66 µH
THD+N − Total Harmonic Distortion + Noise − %
THD+N − Total Harmonic Distortion + Noise − %
10
1
f = 1 kHz
f = 20 Hz
0.1
0.01
f = 10 kHz
0.001
0.01
0.1
1
10
PO − Output Power − W
Gain = 20 dB
VCC = 12 V
ZL = 4 Ω + 33 µH
1
f = 1 kHz
0.1
0.01
f = 10 kHz
0.001
0.01
30
0.1
1
G005
G006
OUTPUT POWER
vs
PLIMIT VOLTAGE
30
30
Gain = 20 dB
VCC = 24 V
ZL = 8 Ω + 66 µH
25
PO − Output Power − W
PO(Max) − Maximum Output Power − W
30
Figure 7.
MAXIMUM OUTPUT POWER
vs
PLIMIT VOLTAGE
20
15
10
Gain = 20 dB
VCC = 12 V
ZL = 4 Ω + 33 µH
20
15
10
5
5
0
0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
0
0.0
VPLIMIT − PLIMIT Voltage − V
0.5
G007
Note: Dashed line represents thermally limited region.
Figure 8.
8
10
PO − Output Power − W
Note: Dashed lines represent thermally limited region.
Figure 6.
25
f = 20 Hz
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1.0
1.5
2.0
2.5
3.0
VPLIMIT − PLIMIT Voltage − V
3.5
4.0
G008
Figure 9.
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TYPICAL CHARACTERISTICS (continued)
(All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3112D2 EVM which is
available at ti.com.)
GAIN/PHASE
vs
FREQUENCY
EFFICIENCY
vs
OUTPUT POWER
40
100
35
50
100
90
Phase
VCC = 24 V
80
30
0
25
−50
20
−100
15
−150
CI = 1 µF
Gain = 20 dB
Filter = Audio Precision AUX-0025
VCC = 12 V
VI = 0.1 Vrms
ZL = 8 Ω + 66 µH
10
5
0
10
100
1k
Phase − °
Gain
η − Efficiency − %
Gain − dB
70
VCC = 12 V
60
50
40
30
−200
20
−250
Gain = 20 dB
ZL = 8 Ω + 66 µH
10
−300
100k
10k
0
0
f − Frequency − Hz
5
10
G009
20
25
30
G012
Note: Dashed line represents thermally limited region.
Figure 11.
Figure 10.
EFFICIENCY
vs
OUTPUT POWER
SUPPLY CURRENT
vs
TOTAL OUTPUT POWER
1.2
100
Gain = 20 dB
ZL = 8 Ω + 66 µH
90
1.0
VCC = 24 V
70
ICC − Supply Current − A
80
η − Efficiency − %
15
PO − Output Power − W
VCC = 12 V
60
50
40
30
20
VCC = 12 V
0.8
VCC = 24 V
0.6
0.4
0.2
Gain = 20 dB
ZL = 4 Ω + 33 µH
10
0
0.0
0
5
10
15
20
25
PO − Output Power − W
30
0
10
15
20
25
PO(Tot) − Total Output Power − W
G013
Note: Dashed line represents thermally limited region.
Figure 12.
5
30
G014
Note: Dashed line represents thermally limited region.
Figure 13.
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TYPICAL CHARACTERISTICS (continued)
(All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3112D2 EVM which is
available at ti.com.)
SUPPLY CURRENT
vs
TOTAL OUTPUT POWER
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
1.8
1.6
KSVR − Supply Ripple Rejection Ratio − dB
Gain = 20 dB
VCC = 12 V
ZL = 4 Ω + 33 µH
ICC − Supply Current − A
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.0
0
5
10
15
20
25
30
Gain = 20 dB
VCC = 12 V
ZL = 8 Ω + 66 µH
−20
−40
−60
−80
−100
−120
20
100
1k
10k
20k
f − Frequency − Hz
PO(Tot) − Total Output Power − W
G016
G015
Figure 14.
Figure 15.
DEVICE INFORMATION
Gain setting via GAIN0 and GAIN1 inputs
The gain of the TPA3112D1 is set by two input terminals, GAIN0 and GAIN1. The voltage slew rate of these gain
terminals, along with terminals 1 and 14, must be restricted to no more than 10V/ms. For higher slew rates, use
a 100kΩ resistor in series with the terminals.
The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance (ZI) to be dependent on the gain setting. The actual gain settings are controlled by
ratios of resistors, so the gain variation from part-to-part is small. However, the input impedance from part-to-part
at the same gain may shift by ±20% due to shifts in the actual resistance of the input resistors.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 7.2 kΩ, which is the absolute minimum input impedance of the TPA3112D1. At the lower gain
settings, the input impedance could increase as high as 72 kΩ
Table 1. Gain Setting
10
AMPLIFIER GAIN (dB)
INPUT IMPEDANCE
(kΩ)
TYP
TYP
20
60
1
26
30
1
0
32
15
1
1
36
9
GAIN1
GAIN0
0
0
0
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SD OPERATION
The TPA3112D1 employs a shutdown mode of operation designed to reduce supply current (ICC) to the absolute
minimum level during periods of nonuse for power conservation. The SD input terminal should be held high (see
specification table for trip point) during normal operation when the amplifier is in use. Pulling SD low causes the
outputs to mute and the amplifier to enter a low-current state. Never leave SD unconnected, because amplifier
operation would be unpredictable.
For the best power-off pop performance, place the amplifier in the shutdown mode prior to removing the power
supply voltage.
PLIMIT
The voltage at pin 10 can used to limit the power to levels below that which is possible based on the supply rail.
Add a resistor divider from GVDD to ground to set the voltage at the PLIMIT pin. An external reference may also
be used if tighter tolerance is required. Also add a 1μF capacitor from pin 10 to ground.
The PLIMIT circuit sets a limit on the output peak-to-peak voltage. This limit can be thought of as a "virtual"
voltage rail which is lower than the supply connected to PVCC. This "virtual" rail is 4 times the voltage at the
PLIMIT pin. This output voltage can be used to calculate the maximum output power for a given maximum input
voltage and speaker impedance.
TPA3112D1 PLimit Operation
Figure 16. PLIMIT Circuit Operation
The PLIMIT circuits sets a limit on the output peak-to-peak voltage. The limiting is done by limiting the duty cycle
to fixed maximum value. This limit can be thought of as a “virtual” voltage rail which is lower than the supply
connected to PVCC. This “virtual” rail is 4 times the voltage at the PLIMIT pin. This output voltage can be used to
calculate the maximum output power for a given maximum input voltage and speaker impedance.
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POUT
ææ
ö
ö
RL
çç ç
÷ ´ VP ÷÷
è RL + 2 ´ RS ø
ø
=è
2 ´ RL
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2
for unclipped power
(1)
Where:
RS is the total series resistance including RDS(on), and any resistance in the output filter.
RL is the load resistance.
VP is the peak amplitude of the output possible within the supply rail.
VP = 4 × PLIMIT voltage if PLIMIT < 4 × VP
POUT(10%THD) = 1.25 × POUT(unclipped)
Table 2. PLIMIT Typical Operation
Test Conditions ()
PLIMIT Voltage
Output Power (W)
Output Voltage
Amplitude (VP-P)
PVCC=24V, Vin=1Vrms,
RL=4Ω, Gain=20dB
6.97
22.1
26.9
PVCC=24V, Vin=1Vrms,
RL=4Ω, Gain=20dB
1.92
10
15.0
PVCC=24V, Vin=1Vrms,
RL=4Ω, Gain=20dB
1.24
5
10.0
PVCC=12V, Vin=1Vrms,
RL=4Ω, Gain=20dB
6.95
17.2
20.9
PVCC=12V, Vin=1Vrms,
RL=4Ω, Gain=20dB
1.75
10
15.3
PVCC=12V, Vin=1Vrms,
RL=4Ω, Gain=20dB
1.20
5
10.3
GVDD Supply
The GVDD Supply is used to power the gates of the output full bridge transistors. It can also used to supply the
PLIMIT voltage divider circuit. Add a 1μF capacitor to ground at this pin.
DC Detect
TPA3112D1 has circuitry which will protect the speakers from DC current which might occur due to defective
capacitors on the input or shorts on the printed circuit board at the inputs. A DC detect fault will be reported on
the FAULT pin as a low state. The DC Detect fault will also cause the amplifier to shutdown by changing the
state of the outputs to Hi-Z. To clear the DC Detect it is necessary to cycle the PVCC supply. Cycling SD will
NOT clear a DC detect fault.
A DC Detect Fault is issued when the output differential duty-cycle exceeds 14% (eg. +57%, -43%) for more than
420 ms at the same polarity. This feature protects the speaker from large DC currents or AC currents less than 2
Hz. To avoid nuisance faults due to the DC detect circuit, hold the SD pin low at power-up until the signals at the
inputs are stable. Also, take care to match the impedance seen at the positive and negative input to avoid
nuisance DC detect faults.
The minimum differential input voltages required to trigger the DC detect are shown in Table Table 3. The inputs
must remain at or above the voltage listed in the table for more than 420 ms to trigger the DC detect.
Table 3. DC Detect Threshold
12
AV(dB)
Vin (mV, differential)
20
112
26
56
32
28
36
17
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SHORT-CIRCUIT PROTECTION AND AUTOMATIC RECOVERY FEATURE
TPA3112D2 has protection from over-current conditions caused by a short circuit on the output stage. The short
circuit protection fault is reported on the FAULT pin as a low state. The amplifier outputs are switched to a Hi-Z
state when the short circuit protection latch is engaged. The latch can be cleared by cycling the SD pin through
the low state.
If automatic recovery from the short circuit protection latch is desired, connect the FAULT pin directly to the SD
pin. This will allow the FAULT pin function to automatically drive the SD pin low which will clear the short circuit
protection latch.
THERMAL PROTECTION
Thermal protection on the TPA3112D1 prevents damage to the device when the internal die temperature
exceeds 150°C. There is a ±15°C tolerance on this trip point from device to device. Once the die temperature
exceeds the thermal set point, the device enters into the shutdown state and the outputs are disabled. This is not
a latched fault. The thermal fault is cleared once the temperature of the die is reduced by 15°C. The device
begins normal operation at this point with no external system interaction.
Thermal protection faults are NOT reported on the FAULT terminal.
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APPLICATION INFORMATION
PVCC
100 μF
0.1 μF
1000pF
100k Ω
Control
System
1
SD
PVCC
FAULT
PVCC
28
1 kΩ
2
3
4
5
6
AVCC
PVCC
7
10 Ω
1 uF
8
GND
BSN
GND
OUTN
GAIN0
PGND
GAIN1
OUTN
BSN
AVCC
TPA3112D1
AGND
BSP
GVDD
OUTP
PLIMIT
PGND
INN
OUTP
INP
BSP
NC
PVCC
AVCC
PVCC
27
26
0.47 μF
25
24
FB
23
1000 pF
22
21
1000 pF
9
1 uF
10
11
1 uF
Audio
Source
12
1 uF
AVCC
100 kW
13
(1)
20
FB
19
0.47 μF
18
17
16
100 μF
14
15
0.1 μF
1000pF
GND
29
PowerPAD
PVCC
(1)
100 kΩ resistor is needed if the PVCC slew rate is more than 10 V/ms.
Figure 17. Mono Class-D Amplifier with BTL Output
14
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CLASS-D OPERATION
This section focuses on the class-D operation of the TPA3112D1.
TPA3112D1 Modulation Scheme
The TPA3112D1 uses a modulation scheme that allows operation without the classic LC reconstruction filter
when the amp is driving an inductive load. Each output is switching from 0 volts to the supply voltage. The OUTP
and OUTN are in phase with each other with no input so that there is little or no current in the speaker. The duty
cycle of OUTP is greater than 50% and OUTN is less than 50% for positive output voltages. The duty cycle of
OUTP is less than 50% and OUTN is greater than 50% for negative output voltages. The voltage across the load
sits at 0 V throughout most of the switching period, greatly reducing the switching current, which reduces any I2R
losses in the load.
OUTP
OUTN
Differential
Voltage
Across
Load
Output = 0 V
+12 V
0V
-12 V
Current
OUTP
OUTN
Differential
Voltage
Across
Load
Output > 0 V
+12 V
0V
-12 V
Current
Figure 18. The TPA3112D1 Output Voltage and Current Waveforms Into an Inductive Load
Ferrite Bead Filter Considerations
Using the Advanced Emissions Suppression Technology in the TPA3112D1 amplifier it is possible to design a
high efficiency Class-D audio amplifier while minimizing interference to surrounding circuits. it is also possible to
accomplish this with only a low-cost ferrite bead filter. In this case it is necessary to carefully select the ferrite
bead used in the filter.
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One important aspect of the ferrite bead selection is the type of material used in the ferrite bead. Not all ferrite
material is alike, so it is important to select a material that is effective in the 10 to 100 MHz range which is key to
the operation of the Class D amplifier. Many of the specifications regulating consumer electronics have
emissions limits as low as 30 MHz. It is important to use the ferrite bead filter to block radiation in the 30 MHz
and above range from appearing on the speaker wires and the power supply lines which are good antennas for
these signals. The impedance of the ferrite bead can be used along with a small capacitor with a value in the
range of 1000 pF to reduce the frequency spectrum of the signal to an acceptable level. For best performance,
the resonant frequency of the ferrite bead/ capacitor filter should be less than 10 MHz.
Also, it is important that the ferrite bead is large enough to maintain its impedance at the peak currents expected
for the amplifier. Some ferrite bead manufacturers specify the bead impedance at a variety of current levels. In
this case it is possible to make sure the ferrite bead maintains an adequate amount of impedance at the peak
current the amplifier will see. If these specifications are not available, it is also possible to estimate the bead
current handling capability by measuring the resonant frequency of the filter output at very low power and at
maximum power. A change of resonant frequency of less than fifty percent under this condition is desirable.
Examples of ferrite beads which have been tested and work well with the TPA3112D2 include 28L0138-80R-10
and HI1812V101R-10 from Steward and the 742792510 from Wurth Electronics.
A high quality ceramic capacitor is also needed for the ferrite bead filter. A low ESR capacitor with good
temperature and voltage characteristics will work best.
Additional EMC improvements may be obtained by adding snubber networks from each of the class D outputs to
ground. Suggested values for a simple RC series snubber network would be 10 ohms in series with a 330 pF
capacitor although design of the snubber network is specific to every application and must be designed taking
into account the parasitic reactance of the printed circuit board as well as the audio amp. Take care to evaluate
the stress on the component in the snubber network especially if the amp is running at high PVCC. Also, make
sure the layout of the snubber network is tight and returns directly to the PGND or the PowerPad™ beneath the
chip.
Efficiency: LC Filter Required With the Traditional Class-D Modulation Scheme
The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results
in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is
large for the traditional modulation scheme, because the ripple current is proportional to voltage multiplied by the
time at that voltage. The differential voltage swing is 2 x VCC, and the time at each voltage is half the period for
the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half cycle for
the next half cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive,
whereas an LC filter is almost purely reactive.
The TPA3112D1 modulation scheme has little loss in the load without a filter because the pulses are short and
the change in voltage is VCC instead of 2 x VCC. As the output power increases, the pulses widen, making the
ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for most
applications the filter is not needed.
An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow
through the filter instead of the load. The filter has less resistance but higher impedance at the switching
frequency than the speaker, which results in less power dissipation, therefore increasing efficiency.
When to Use an Output Filter for EMI Suppression
The TPA3112D1 has been tested with a simple ferrite bead filter for a variety of applications including long
speaker wires up to 125 cm and high power. The TPA3112D1 EVM passes FCC Class B specifications under
these conditions using twisted speaker wires. The size and type of ferrite bead can be selected to meet
application requirements. Also, the filter capacitor can be increased if necessary with some impact on efficiency.
There may be a few circuit instances where it is necessary to add a complete LC reconstruction filter. These
circumstances might occur if there are circuits near which are sensitive to noise. Therefore, a classic second
order Butterworth filter similar to those shown in Figure 19 through Figure 21 can be used.
16
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33 mH
OUTP
L1
C2
1 mF
33 mH
OUTN
L2
C3
1 mF
Figure 19. Typical LC Output Filter, Cutoff Frequency of 27 kHz, Speaker Impedance = 8 Ω
15 mH
OUTP
L1
C2
2.2 mF
15 mH
OUTN
L2
C3
2.2 mF
Figure 20. Typical LC Output Filter, Cutoff Frequency of 27 kHz, Speaker Impedance = 4 Ω
Ferrite
Chip Bead
OUTP
1 nF
Ferrite
Chip Bead
OUTN
1 nF
Figure 21. Typical Ferrite Chip Bead Filter (Chip Bead Example: Steward HI0805R800R-10)
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INPUT RESISTANCE
Changing the gain setting can vary the input resistance of the amplifier from its smallest value, 9 kΩ ±20%, to the
largest value, 60 kΩ ±20%. As a result, if a single capacitor is used in the input high-pass filter, the -3 dB or
cutoff frequency may change when changing gain steps.
Zf
Ci
IN
Input
Signal
Zi
The -3-dB frequency can be calculated using Equation 2. Use the ZI values given in Table 1.
f =
1
2p Zi Ci
(2)
INPUT CAPACITOR, CI
In the typical application, an input capacitor (CI) is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier (ZI) form a highpass filter with the corner frequency determined in Equation 3.
-3 dB
fc =
1
2p Zi Ci
fc
(3)
The value of CI is important, as it directly affects the bass (low-frequency) performance of the circuit. Consider
the example where ZI is 60 kΩ and the specification calls for a flat bass response down to 20 Hz. Equation 3 is
reconfigured as Equation 4.
Ci =
1
2p Zi fc
(4)
In this example, CI is 0.13 µF; so, one would likely choose a value of 0.15 μF as this value is commonly used. If
the gain is known and is constant, use ZI from Table 1 to calculate CI. A further consideration for this capacitor is
the leakage path from the input source through the input network CI) and the feedback network to the load. This
leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially
in high gain applications. For this reason, a low-leakage tantalum or ceramic capacitor is the best choice. If a
ceramic capacitor is used, use a high quality capacitor with good temperature and voltage coefficient. An X7R
type works well and if possible use a higher voltage rating than required. This will give a better C vs voltage
characteristic. When polarized capacitors are used, the positive side of the capacitor should face the amplifier
input in most applications as the dc level there is held at 3 V, which is likely higher than the source dc level. Note
that it is important to confirm the capacitor polarity in the application. Additionally, lead-free solder can create dc
offset voltages and it is important to ensure that boards are cleaned properly.
POWER SUPPLY DECOUPLING, CS
The TPA3112D1 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker.
18
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Optimum decoupling is achieved by using a network of capacitors of different types that target specific types of
noise on the power supply leads. For higher frequency transients due to parasitic circuit elements such as bond
wire and copper trace inductances as well as lead frame capacitance, a good quality low equivalent-seriesresistance (ESR) ceramic capacitor of value between 220 pF and 1000 pF works well. This capacitor should be
placed as close to the device PVCC pins and system ground (either PGND pins or PowerPad) as possible. For
mid-frequency noise due to filter resonances or PWM switching transients as well as digital hash on the line,
another good quality capacitor typically 0.1 mF to 1 μF placed as close as possible to the device PVCC leads
works best For filtering lower frequency noise signals, a larger aluminum electrolytic capacitor of 220 mF or
greater placed near the audio power amplifier is recommended. The 220 mF capacitor also serves as a local
storage capacitor for supplying current during large signal transients on the amplifier outputs. The PVCC
terminals provide the power to the output transistors, so a 220 μF or larger capacitor should be placed on each
PVCC terminal. A 10 μF capacitor on the AVCC terminal is adequate. Also, a small decoupling resistor between
AVCC and PVCC can be used to keep high frequency class D noise from entering the linear input amplifiers.
BSN and BSP CAPACITORS
The full H-bridge output stage uses only NMOS transistors. Therefore, they require bootstrap capacitors for the
high side of each output to turn on correctly. A 470-nF ceramic capacitor, rated for at least 16 V, must be
connected from each output to its corresponding bootstrap input. Specifically, one 470-nF capacitor must be
connected from OUTP to BSP, and one 470-nF capacitor must be connected from OUTN to BSN. (See the
application circuit diagram in Figure 1.)
The bootstrap capacitors connected between the BSx pins and corresponding output function as a floating power
supply for the high-side N-channel power MOSFET gate drive circuitry. During each high-side switching cycle,
the bootstrap capacitors hold the gate-to-source voltage high enough to keep the high-side MOSFETs turned on.
DIFFERENTIAL INPUTS
The differential input stage of the amplifier cancels any noise that appears on both input lines of the channel. To
use the TPA3112D1 with a differential source, connect the positive lead of the audio source to the INP input and
the negative lead from the audio source to the INN input. To use the TPA3112D1 with a single-ended source, ac
ground the INP or INN input through a capacitor equal in value to the input capacitor on INN or INP and apply
the audio source to either input. In a single-ended input application, the unused input should be ac grounded at
the audio source instead of at the device input for best noise performance. For good transient performance, the
impedance seen at each of the two differential inputs should be the same.
The impedance seen at the inputs should be limited to an RC time constant of 1 ms or less if possible. This is to
allow the input dc blocking capacitors to become completely charged during the 14 msec power-up time. If the
input capacitors are not allowed to completely charge, there will be some additional sensitivity to component
matching which can result in pop if the input components are not well matched.
USING LOW-ESR CAPACITORS
Low-ESR capacitors are recommended throughout this application section. A real (as opposed to ideal) capacitor
can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor
minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance,
the more the real capacitor behaves like an ideal capacitor.
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PRINTED-CIRCUIT BOARD (PCB) LAYOUT
The TPA3112D1 can be used with a small, inexpensive ferrite bead output filter for most applications. However,
since the Class-D switching edges are very fast, it is necessary to take care when planning the layout of the
printed circuit board. The following suggestions will help to meet EMC requirements.
• Decoupling capacitors—The high-frequency decoupling capacitors should be placed as close to the PVCC
and AVCC terminals as possible. Large (220 μF or greater) bulk power supply decoupling capacitors should
be placed near the TPA3112D1 on the PVCC supplies. Local, high-frequency bypass capacitors should be
placed as close to the PVCC pins as possible. These caps can be connected to the thermal pad directly for
an excellent ground connection. Consider adding a small, good quality low ESR ceramic capacitor between
220 pF and 1000 pF and a larger mid-freqency cap of value between 0.1mF and 1mF also of good quality to
the PVCC connections at each end of the chip.
• Keep the current loop from each of the outputs through the ferrite bead and the small filter cap and back to
PGND as small and tight as possible. The size of this current loop determines its effectiveness as an
antenna.
• Output filter—The ferrite EMI filter should be placed as close to the output terminals as possible for the best
EMI performance. The LC filter should be placed close to the outputs. The capacitors used in both the ferrite
and LC filters should be grounded to power ground.
• Thermal Pad—The thermal pad must be soldered to the PCB for proper thermal performance and optimal
reliability. The dimensions of the thermal pad and thermal land should be 6.46 mm by 2.35 mm. Seven rows
of solid vias (three vias per row, 0.33 mm or 13 mils diameter) should be equally spaced underneath the
thermal land. The vias should connect to a solid copper plane, either on an internal layer or on the bottom
layer of the PCB. The vias must be solid vias, not thermal relief or webbed vias. See TI Application Report
SLMA002 for more information about using the TSSOP thermal pad.
For an example layout, see the TPA3112D1 Evaluation Module (TPA3112D1EVM) User Manual. Both the EVM
user manual and the thermal pad application note are available on the TI Web site at http://www.ti.com.
SPACER
REVISION HISTORY
Changes from Original (September 2009) to Revision A
Page
•
Added slew rate adjustment information ............................................................................................................................. 10
•
Added updates for Figure 17, pin 7 .................................................................................................................................... 14
Changes from Revision A (July 2010) to Revision B
•
Page
In the BSN and BSP CAPACITORS section, the 220-nf capacitor rated for at least 25V was changed to a 470-nf
capacitor rated to at least 16V ............................................................................................................................................ 19
Changes from Revision B (August 2010) to Revision C
Page
•
Added < 10 V/ms to VI in the Absolute Maximum Ratings table .......................................................................................... 2
•
Added a 100kΩ resistor to AVCC Pin 14 and Note 1 to Figure 17 .................................................................................... 14
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PACKAGE OPTION ADDENDUM
www.ti.com
17-Mar-2012
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package
Drawing
Pins
Package Qty
Eco Plan
(2)
Lead/
Ball Finish
MSL Peak Temp
(3)
TPA3112D1PWP
ACTIVE
HTSSOP
PWP
28
50
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-3-260C-168 HR
TPA3112D1PWPR
ACTIVE
HTSSOP
PWP
28
2000
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-3-260C-168 HR
Samples
(Requires Login)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
TPA3112D1PWPR
Package Package Pins
Type Drawing
SPQ
HTSSOP
2000
PWP
28
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
330.0
16.4
Pack Materials-Page 1
6.9
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
10.2
1.8
12.0
16.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPA3112D1PWPR
HTSSOP
PWP
28
2000
367.0
367.0
38.0
Pack Materials-Page 2
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voids all express and any implied warranties for the associated TI component or service and is an unfair and deceptive business practice.
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Buyer acknowledges and agrees that it is solely responsible for compliance with all legal, regulatory and safety-related requirements
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anticipate dangerous consequences of failures, monitor failures and their consequences, lessen the likelihood of failures that might cause
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In some cases, TI components may be promoted specifically to facilitate safety-related applications. With such components, TI’s goal is to
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Products
Applications
Audio
www.ti.com/audio
Automotive and Transportation www.ti.com/automotive
Amplifiers
amplifier.ti.com
Communications and Telecom www.ti.com/communications
Data Converters
dataconverter.ti.com
Computers and Peripherals
www.ti.com/computers
DLP® Products
www.dlp.com
Consumer Electronics
www.ti.com/consumer-apps
DSP
dsp.ti.com
Energy and Lighting
www.ti.com/energy
Clocks and Timers
www.ti.com/clocks
Industrial
www.ti.com/industrial
Interface
interface.ti.com
Medical
www.ti.com/medical
Logic
logic.ti.com
Security
www.ti.com/security
Power Mgmt
power.ti.com
Space, Avionics and Defense
www.ti.com/space-avionics-defense
Microcontrollers
microcontroller.ti.com
Video and Imaging
www.ti.com/video
RFID
www.ti-rfid.com
OMAP Mobile Processors
www.ti.com/omap
TI E2E Community
e2e.ti.com
Wireless Connectivity
www.ti.com/wirelessconnectivity
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2012, Texas Instruments Incorporated
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