Linear Technology LTC2400, LTC1599 20-bit DAC, LTC2400 A-to-D converter, RV-722, 720A, VDR-307 Kelvin-Varley divider, LTC1152, LT1010, LTC1150 op amp, LTZ1000A voltage source, HP3458A DVM Application Note

Application Note 86

January 2001

A Standards Lab Grade 20-Bit DAC with 0.1ppm/

°

C Drift

The Dedicated Art of Digitizing One Part Per Million

Jim Williams

J. Brubaker

P. Copley

J. Guerrero

F. Oprescu

INTRODUCTION

Significant progress in high precision, instrumentation grade D-to-A conversion has recently occurred. Ten years ago 12-bit D-to-A converters (DACs) were considered premium devices. Today, 16-bit DACs are available and increasingly common in system design. These are true precision devices with less than 1LSB linearity error and

1ppm/

°

C drift.

1

Nonetheless, there are DAC applications that require even higher performance. Automatic test equipment, instruments, calibration apparatus, laser trimmers, medical electronics and other applications often require DAC accuracy beyond 16 bits. 18-bit DACs have been produced in circuit assembly form, although they are expensive and require frequent calibration. 20 and even

23+ (0.1ppm!) bit DACs are represented by manually switched Kelvin-Varley dividers. These devices, although amazingly accurate, are large, slow and extremely costly.

Their use is normally restricted to standards labs.

2

A useful development would be a practical, 20-bit (1ppm)

DAC that is easily constructed and does not require frequent calibration.

20-Bit DAC Architecture

Figure 1 diagrams the architecture of a 20-bit (1ppm) DAC.

This scheme is based on the availability of a true 1ppm analog-to-digital converter with scale and zero drifts below 0.02ppm/

°

C. This device, the LTC

®

2400, is used as a feedback element in a digitally corrected loop to realize a

20-bit DAC.

3

In practice, the “slave” 20-bit DAC’s output is monitored by the “master” LTC2400 A-to-D, which feeds digital information to a code comparator. The code comparator differences the user input word with the LTC2400 output, presenting a corrected code to the slave DAC. In this fashion, the slave DAC’s drifts and nonlinearity are continuously corrected by the loop to an accuracy determined by the A-to-D converter and V

REF

4

. The sole DAC requirement is that it be monotonic. No other components in the loop need to be stable.

20-BIT

USER

INPUT

CODE

DIGITAL

CODE

COMPARATOR

FEEDBACK

CODE

V

REF

LTC2400

“MASTER”

A-TO-D

V

IN

CORRECTED

CODE

20-BIT

(1PPM)

ANALOG

OUTPUT

20-BIT

“SLAVE”

DAC

V

OUT

AN86 F01

Figure 1. Conceptual Loop-Based 20-Bit DAC. Digital

Comparison Allows A-to-D to Correct DAC Errors. LTC2400

A-to-D’s Low Uncertainty Characteristics Permit 1ppm

Output Accuracy

, LTC and LT are registered trademarks of Linear Technology Corporation.

Note 1: See Appendix A, “A History of High Accuracy Digital-to-Analog

Conversion,” for a review of high accuracy digital-to-analog conversion.

Note 2: Consult Appendix C, “Verifying Data Converter Linearity to

1ppm,” for discussion on Kelvin-Varley dividers. Also, see Appendix A,

“A History of High Accuracy Digital-to-Analog Conversion.”

Note 3: The LTC2400 analog-to-digital converter is profiled in

Appendix␣ B, “The LTC2400—A Monolithic 24-Bit Analog-to-Digital

Converter.”

Note 4: D-to-A converters have been placed in loops to make A-to-D converters for a long time. Here, an A-to-D converter feeds back a loop to form a D-to-A converter. There seems a certain justified symmetry to this development. Turnabout is indeed fair play.

AN86-1

Application Note 86

This loop has a number of desireable attributes. As mentioned, accuracy limitations are set by the A-to-D converter and its reference. No other components need be stable. Additionally, loop behavior averages low order bit indexing and jitter, obviating the loop’s inherent smallsignal instability. Finally, classical remote sensing may be used or digitally based sensing is possible by placing the

A-to-D converter at the load. The A-to-D’s SO-8 package and lack of external components makes this digitally incarnated Kelvin sensing scheme practical.

5

Circuitry Details

Figure 2 is a detailed schematic of the 1ppm DAC. The slave DAC is comprised of two DACs. The upper 16 bits of the code comparator’s output are fed to a 16-bit DAC

(“MSB DAC”), while the lower bits are converted by a separate DAC (“LSB DAC”). Although a total of 32 bits are presented to the two DACs, there are 8 bits of overlap, assuring loop capture under all conditions. The composite

DACs’ resultant 24-bit resolution provides 4 bits of indexing range below the 20th bit, ensuring a stable LSB of

1ppm of scale. A1 and A2 transform the DAC’s output currents into voltages, which are summed at A3. A3’s scaling is arranged so that the correction loop can always capture and correct any combination of zero- and fullscale errors. A3’s output, the circuit output, feeds the

LTC2400 A-to-D. The LT

®

1010 provides buffering to drive loads and cables. The A-to-D’s digital output is differenced against the input word by the code comparator, which produces a corrected code. This corrected code is applied to the MSB and LSB DACs, closing a feedback loop.

6

The loop’s integrity is determined by A-to-D converter and voltage reference errors.

7

The resistor and diodes at the

5V powered A-to-D protect it from inadvertent A3 outputs

(power up, transient, lost supply, etc.). A4 is a reference inverter and A5 provides a clean ground potential to both

DACs.

20-BIT

USER

INPUT

CODE

COMMAND

INPUTS FROM

CODE COMPARATOR

CODE

COMPARATOR

(SEE APPENDIX D

FOR DETAILS)

CORRECTED

CODE OUT—

24-BIT WORD

+

A4

–

LT1001

5V

24-BIT

FEEDBACK

CODE

SERIAL

DIGITAL

OUT

5V

REF

(SEE APPENDIX I

FOR OPTIONS)

LTC2400

A-TO-D

REF

IN

CS

1k

5V

1N5817

1N5817

* = 1% METAL FILM

DATA

INPUTS

5V

REF

100pF

REF R

OS

R

FS

R

LD

WR

ML

BYTE

R

LTC1599

CLVL CLR

FB

I

I

OUT

–

5V

–

+

A5

LT1001

DATA

INPUTS

5V

LD

ML

BYTE

WR

CLVL

R

OS

CLR

REF

LTC1599

V

REF

5V

R

FB

I

–

I

OUT

MSB DAC

100pF

–

+

A1

LT1001

LSB DAC

100pF

–

A2

+

LT1001

2k*

4.12M*

15V

OUTPUT

AMPLIFIER

+

–

A3

LT1001

LT1010

–15V

0.1

µ

F

2k*

4.12M*

Figure 2. Detail of 1ppm DAC. Composite DAC Is Comprised of Two DAC Values Summed at Output Amplifier. LTC2400 A-to-D and Code Comparator Furnish Stabilizing Feedback

Note 5: One wonders what Lord Kelvin’s response would be to the digizatation of his progeny. Such uncertainties are the residue of progress.

Note 6: The code comparator is detailed in Appendix D, “A Processor

Based Code Comparator.”

OUTPUT

AN86 F02

Note 7: Voltage reference options are discussed in Appendix I,

“Voltage References.” For tutorial on the LTC2400, refer to

Appendix␣ B.

AN86-2

Linearity Considerations

A-to-D linearity determines overall DAC linearity. The

A-to-D has about

±

2ppm nonlinearity. In applications where this error is permissible, it may be ignored. If 1ppm linearity is required, it is obtainable by correcting the residual linearity error with software techniques. Details on LTC2400 linearity and this feature are presented in

Appendices D and E.

DC Performance Characteristics

Figure 3 is a plot of linearity vs output code. The data shows linearity is within 1ppm over all codes.

8

Output noise, measured in a 0.1Hz to 10Hz bandpass, is seen in

Figure 4 to be about 0.2LSB.

9

This measurement is somewhat corrupted by equipment limitations, which set a noise floor of about 0.2

µ

V.

Application Note 86

Dynamic Performance

The A-to-D’s conversion rate combines with the loop’s sampled data characteristic and slow amplifiers to dictate relatively slow DAC response. Figure 5’s slew response requires about 150 microseconds.

Figure 6 shows full-scale DAC settling time to within 1ppm

(

±

5

µ

V) requires about 1400 milliseconds. A smaller step

(Figure 7) of 500

µ

V needs only 100 milliseconds to settle within 1ppm.

10

Conclusion

Summarized 1ppm DAC specifications appear in Figure 9.

These specifications should be considered guidelines, as the options and variations noted will affect performance.

Consult the appropriate appendices for design specifics and trade-offs.

1.0

INDICATED MEASUREMENT RESULTS

0.5

0

–0.5

SHADED REGION DELINEATES

MEASUREMENT UNCERTAINTY DUE TO

DAC OUPUT NOISE AND

INSTRUMENTATION LIMITATIONS

–1.0

0 262,144 524,288 786,432 1,048,576

DIGITAL INPUT CODE

AN86 F03

Figure 3. Linearity Plot Shows No

Error Outside 1ppm for All Codes

2s/DIV

AN86 F04.tif

Figure 4. Output Noise Indicates Less Than 1

µ

V,

About 0.2LSB. Measurement Noise Floor, Due to Equipment Limitations, Is 0.2

µ

V

Note 8: Establishing and maintaining confidence in a 1ppm linearity measurement is uncomfortably close to the state of the art. The technique used is shown in Appendix C, “Verifying Data Converter

Linearity to 1ppm.”

Note 9: Noise measurement considerations appear in Appendix H,

“Microvolt Level Noise Measurement.”

Note 10: Measuring DAC settling time to 1ppm is by no means straightforward, even at the relatively slow speed involved here. See

Appendix G, “Measuring DAC Settling Time.”

AN86-3

Application Note 86

50

µ s/DIV

AN86 F05.tif

Figure 5. DAC Output Full-Scale Slew Characteristics

50ms/DIV

AN86 F07.tif

Figure 7. Small Step Settling Time Measures 100

Milliseconds to Within 1ppm (

±

5

µ

V) for a 500

µ

V

Transition

500ms/DIV

AN86 F06.tif

Figure 6. High Resolution Settling Detail After a Full-Scale Step.

Settling Time Is 1400 Milliseconds to Within 1ppm (

±

5

µ

V)

PARAMETER

Resolution

SPECIFICATION

1ppm

Full-Scale Error

Full-Scale Error Drift

Offset Error

Offset Error Drift

Nonlinearity

4ppm of V

REF

(Trimmable to 1ppm by V

REF

Adjustment)

0.04ppm/

°

C Exclusive of Reference

(0.1ppm/

°

C with LTZ1000A Reference

1

)

0.5ppm

0.01ppm/

°

C

±

2ppm, Trimmable to Less Than 1ppm

2

Output Noise

Slew Rate

0.2ppm

(

≈

0.9

µ

V, 0.1Hz to 10Hz BW)

0.033V/

µ s

Settling Time—Full-Scale Step 1400 Milliseconds

Settling Time—500

µ

V Step 100 Milliseconds

Output Voltage Range 0V to 5V. For Other Ranges See Note 3

Note 1: See Appendix I

Note 2: See Appendix E

Note 3: See Appendices E and F

Figure 8. Summarized Specifications for the 20-Bit DAC

Note: This Application Note was derived from a manuscript originally prepared for publication in EDN magazine.

AN86-4

REFERENCES

1. Linear Technology Corporation, “LTC2400 Data Sheet,”

Linear Technology Corporation, January 1999.

2. Linear Technology Corporation, “LTC2410 Data Sheet,”

Linear Technology Corporation, April 2000.

3. Keithley Instruments, “Low Level Measurements,”

Keithley Instruments, 1984.

4. Williams, J., “Testing Linearity of the LTC2400 24-Bit

A/D Converter,” Linear Technology Corporation, Design Solution 11, November 1999.

5. Seebeck, T. Dr., “Magnetische Polarisation der Metalle und Erze durch Temperatur-Differenz,” Abhaandlungen der Preussischen Akademic der Wissenschaften

(1822–1823), pp. 265–373.

6. Williams, J., “Component and Measurement Advances

Ensure 16-Bit DAC Settling Time,” Linear Technology

Corporation, Application Note 74, July 1998.

7. Lee, M., “Understanding and Applying Voltage References,” Linear Technology Corporation, Application

Note 82, November 1999.

8. Williams, J., “Applications Considerations and Circuits for a New Chopper-Stabilized Op Amp,” Linear

Technology Corporation, Application Note 9, August

1987.

9. Huffman, B., “Voltage Reference Circuit Collection,”

Linear Technology Corporation, Application Note 42,

June 1991.

Application Note 86

10. Spreadbury, P. J., “The Ultra-Zener—A Portable Replacement for the Weston Cell?” IEEE Transactions on

Instrumentation and Measurement, Vol. 40, No. 2,

April 1991, pp. 343–346.

11. Williams, J., “Thermocouple Measurement,” Linear

Technology Corporation, Application Note 28, February 1988.

12. Hueckel, J. H., “Input Connection Practices for Differential Amplifiers,” Neff Inst. Corporation, Duarte, California.

13. Gould Inc., “Elimination of Noise in Low Level Circuits,” Gould Inc., Instrument Systems Division, Cleveland, Ohio.

14. Williams, J., “Prevent Low Level Amplifier Problems,”

Electronic Design, February 15, 1975, p. 62.

15. Pascoe, G., “The Choice of Solders for High Gain

Devices,” New Electronics (Great Britain), February 6,

1977.

16. Pascoe, G., “The Thermo-E.M.F. of Tin-Lead Alloys,”

Journal Phys. E, December 1976.

17. Brokaw, A. P., “Designing Sensitive Circuits? Don’t

Take Grounds for Granted,” EDN, October 5, 1975, p.␣ 44.

18. Morrison, R., “Noise and Other Interfering Signals,”

John Wiley and Sons, 1992.

19. Morrison, R., “Grounding and Shielding Techniques in Instrumentation,” Wiley-Interscience, 1986.

20. Vishay Inc., “Vishay Foil Resistors,” Vishay Inc., 1999.

AN86-5

Application Note 86

APPENDIX A

A HISTORY OF HIGH ACCURACY

DIGITAL-TO-ANALOG CONVERSION

People have been converting digital-to-analog quantities for a long time. Probably among the earliest uses was the summing of calibrated weights (Figure A1, left center) in weighing applications. Early electrical digital-to-analog conversion inevitably involved switches and resistors of different values, usually arranged in decades. The application was often the calibrated balancing of a bridge or reading, via null detection, some unknown voltage. The most accurate resistor-based DAC of this type is Lord

Kelvin’s Kelvin-Varley divider (Figure, large box). Based on switched resistor ratios, it can achieve ratio accuracies of 0.1ppm (23+ bits) and is still widely employed in standards laboratories.

1

High speed digital-to-analog conversion resorts to electronically switching the resistor network. Early electronic DACs were built at the board level using discrete precision resistors and germanium transistors (Figure, center foreground, is a 12-bit DAC from a

Minuteman missile D-17B inertial navigation system, circa

1962). The first electronically switched DACs available as standard product were probably those produced by

Pastoriza Electronics in the mid 1960s. Other manufacturers followed and discrete- and monolithically-based modular DACs (Figure, right and left) became popular by the

1970s. The units were often potted (Figure, left) for ruggedness, performance or to (hopefully) preserve proprietary knowledge. Hybrid technology produced smaller package size (Figure, left foreground). The development of

Si-Chrome resistors permitted precision monolithic DACs such as the LTC1595 (Figure, immediate foreground). In keeping with all things monolithic, the cost-performance trade off of modern high resolution IC DACs is a bargain.

Think of it! A 16-bit DAC in an 8-pin IC package. What Lord

Kelvin would have given for a credit card and LTC’s phone number.

Note 1: See Appendix C, “Verifying Data Converter Linearity to 1ppm,” for details on Kelvin-Varley Dividers.

AN86 FA01.tif

Figure A1. Historically Significant Digital-to-Analog Converters Include: Weight Set (Center Left), 23+ Bit Kelvin-Varley Divider

(Large Box), Hybrid, Board and Modular Types, and the LTC1595 IC (Foreground). Where Will It All End?

AN86-6

Application Note 86

APPENDIX B

THE LTC2400—A MONOLITHIC 24-BIT

ANALOG-TO-DIGITAL CONVERTER

The LTC2400 is a micropower 24-bit A-to-D converter with an integrated oscillator, 4ppm nonlinearity and 0.3ppm

RMS noise. It uses delta-sigma technology to provide extremely high stability. The device can be configured for better than 110dB rejection at 50Hz or 60Hz

±

2%, or it can be driven by an external oscillator for a user defined rejection frequency in the range 1Hz to 120Hz.

This ultraprecision A-to-D converter in an SO-8 pin package forms the heart of the 20-bit DAC described in the text.

It is significant that the device is used here as a circuit component rather than in the traditional standalone role accorded precision A-to-D converters. This freedom, in keeping with the IC’s economy and ease of use, is a noteworthy opportunity. Alert designers will recognize this development and capitalize on it. Key specifications for the A-to-D are given in Figure B1.

PARAMETER CONDITIONS

Resolution (No Missing Codes) 0.1V

≤

V

REF

≤

V

CC

Integral Nonlinearity

Offset Error

Offset Error Drift

Full-Scale Error

Full-Scale Error Drift

V

REF

= 2.5V

V

REF

= 5V

2.5V

≤

V

REF

≤

V

CC

2.5V

≤

V

REF

≤

V

CC

2.5V

≤

V

REF

≤

V

CC

2.5V

≤

V

REF

≤

V

CC

Total Unadjusted Error V

REF

= 2.5V

V

REF

= 5V

Output Noise

Normal Mode Rejection

60Hz

±

2%

Normal Mode Rejection

50Hz

±

2

Input Voltage Range

Reference Voltage Range

Supply Voltage

24 Bits

2ppm of V

4ppm of V

0.5ppm of V

0.01ppm of V

0.02ppm of V

REF

4ppm of V

REF

REF

REF

REF

/

°

C

REF

/

°

C

5ppm of V

REF

1ppm of V

REF

1.5

µ

V

RMS

110dB (Min)

110dB (Min)

0.125V • V

REF

to 1.125V • V

REF

0.1V

≤

V

REF

≤

V

CC

2.7V

≤

V

CC

≤

5.5V

Supply Current

Conversion Mode

Sleep Mode

CS = 0V

CS = V

CC

200

20

µ

µ

A

A

Figure B1. Key Specifications for LTC2400 A-to-D Converter.

High Linearity and Extreme Stability Allow Realization of 1ppm DAC

AN86-7

Application Note 86

APPENDIX C

VERIFYING DATA CONVERTER LINEARITY TO 1PPM

Help from the Nineteenth Century

INTRODUCTION

Verifying 1ppm linearity of the DAC and the analog-todigital converter used to construct it requires special considerations. Testing necessitates some form of voltage source that produces equal amplitude output steps for incremental digital inputs. Additionally, for measurement confidence, it is desirable that the source be substantially more linear than the 1ppm requirement. This is, of course, a stringent demand and painfully close to the state of the art.

The most linear “D to A” converter is also one of the oldest.

Lord Kelvin’s Kelvin-Varley divider (KVD), in its most developed form, is linear to 0.1ppm. This manually switched device features ten million individual dial settings arranged in seven decades. It may be thought of as a

3-terminal potentiometer with fixed “end-to-end” resistance and a 7-decade switched wiper position (Figure C1).

INPUT

10k 2k 400

Ω

80

Ω

80

Ω

80

Ω

OUT

R = 100k

SEVEN-DECADE SWITCHED

WIPER POSITION PERMITS

SETTING TO 0.1ppm LINEARITY

AN86 FC01

Figure C1. Conceptual Kelvin-Varley Divider

The actual construction of a 0.1ppm KVD is more artistry and witchcraft than science. The market is relatively small, the number of vendors few and resultant price high. If

$13,000 for a bunch of switches and resistors seems offensive, try building and certifying your own KVD. Figure

C2 shows a detailed schematic.

The KVD shown has a 100k

Ω

input impedance. A consequence of this is that wiper output resistance is high and varies with setting. As such, a very low bias current follower is required to unload the KVD without introducing significant error. Now, our KVD looks like Figure␣ C3. The

LT1010 output buffer allows driving cables and loads and, more subtly, maintains the amplifier’s high open-loop gain.

COMMON

AN86 FC02

Figure C2. A 4-Decade Kelvin-Varley Divider. Additional

Decades Are Implemented By Opening Last Switch, Deleting

Two Associated 80

Ω

Values and Continuing

÷

5 Resistor Chains

E

INPUT

KVD

–

+

LTC1152

0.1

µ

F

10k

LT1010 OUTPUT

KVD = ELECTRO SCIENTIFIC INDUSTRIES RV-722,

FLUKE 720A OR JULIE RESEARCH LABS VDR-307

AN86 FC03

Figure C3. KVD with Buffer Gives Output Drive Capability

AN86-8

Approach and Error Considerations

This schematic is deceptively simple. In practice, construction details are crucial. Parasitic thermocouples

(Seebeck effect), layout, grounding, shielding, guarding, cable choice and other issues affect achievable performance.

1

In fact, as good as the chopper-stabilized LTC1152 is with respect to drift, offset, bias current and CMRR, selection is required if we seek sub-ppm nonlinearity performance. Figure C4, an error budget analysis, details some of the selection criteria.

KVD R

IN

= 100k

≈

30k

WORST-CASE

OUTPUT

RESISTANCE

5V

KVD

5V

–

+

LTC1152

0.1

µ

F

ERROR

SOURCE

E

OS

E

OS

∆

T

I

B

CMRR

FINITE GAIN

WORST-CASE

SPEC

5

µ

V

0.05

µ

V/

°

C

50pA

110dB

140dB

REALISTIC SELECTION

TARGET

0.5

µ

V

0.05

µ

V/

°

C

10pA

140dB

140dB

ERROR IN

PPM

0.1

0.01/

°

C

0.1

0.1

0.1

AN86 FC04

Figure C4. Error Budget Analysis for the KVD Buffer.

Selection Permits

≈

0.4ppm Predicted Linearity Error

–

+

LTC1152

0.1

µ

F

LT1010

10k

LT1010

10k

OUTPUT

OUTPUT

FLOATING, BATTERY-POWERED

µ

V NULL DETECTOR

HP-419A

AN86 FC05

Figure C5. Determining Buffer Error By Measuring Input-Output

Deviation with Floating Microvolt Null Detector. Technique

Permits Evaluation of Fixed and Operating Point Induced Errors

Application Note 86

The buffer is tested with Figure C5’s circuit. As the KVD is run through its entire range, the floating null detector must remain well within 1ppm (5

µ

V), preferably below 0.5ppm.

This test ensures that all error sources, particularly I

B

and

CMRR, whose effects vary with operating point, are accounted for. Measured performance indicates the sum of all errors called out in Figure C4 is well within desired limits.

In Figure C6, we add offset trim, a stable voltage source and a second KVD to drive the main KVD. Additionally, an ensemble of three HP3458A voltmeters monitor the output.

The offset trim bleeds a small current into the main KVD ground return, producing a few microvolts of offset-trim range. This functionally trims out all sources of zero error

(amplifier offsets, parasitic thermocouple mismatches and the like), permitting a true zero volt output when the main KVD is set to all zeros.

The voltmeters, specified for < 0.1ppm nonlinearity on the

10V range, “vote” on the source’s output.

Circuitry Details

Figure C7 is a more detailed schematic. It is similar to

Figure C6 but highlights issues and concerns. The grounding scheme is single point, preventing mixing of return currents and the attendant errors. The shielded cables used for interconnections between the KVDs and voltmeters should be specified for low thermal activity. Keithley type SC-93 and Guildline #SCW are suitable. Crush type copper lugs (as opposed to soldered types) provide lower parasitic thermocouple activity at KVD and DVM connection points. However, they must be kept clean to prevent oxidation, thus avoiding excessive thermal voltages.

2

A copper deoxidant (Caig Labs “Deoxit” D100L) is quite effective for maintaining such cleanliness. Low thermal lugs and jacks, preterminated to cables, are also available

(Hewlett-Packard 11053, 11174A) and convenient.

Thermal baffles enclosing KVD and DVM connections tend

Note 1: See Appendix J, “Cables, Connections, Solder, Layout,

Component Choice, Terror and Arcana,” for relevant tutorial.

Note 2: See above Footnote.

AN86-9

Application Note 86

to thermally equilibrate their associated banana jack terminals, minimizing residual parasitic thermocouple activity. Additionally, restrict the number of connections in the signal path. Necessary connections should be matched in number and material so that differential cancellation occurs. Complying with this guideline may necessitate deliberate introduction of solder-copper junctions (marked “X” on Figure␣ C7) to obtain optimum differential cancellation.

3

This is normally facilitated by simply breaking the appropriate wire or PC trace and soldering it. Ensure that the introduced thermocouples temperature track the junctions they are supposed to cancel. This is usually accomplished by locating all junctions within close physical proximity.

The noise filtering capacitor at the main KVD is a low leakage type, with its metal case driven by the output buffer to guard out surface leakage.

When studying the approach used, it is essential to differentiate between linearity and absolute accuracy. This eliminates concerns with absolute standards, permitting certain freedoms in the measurement scheme. In particular, although single-point grounding was used, remote sensing was not. This is a deliberate choice, made to minimize the number of potential error-causing parasitic thermocouples in the signal path. Similarly, a ratiometric reference connection between the KVD LTZ1000A voltage source and the voltmeters was not utilized for the same reason. In theory, a ratiometric connection affords lower drift. In practice, the resultant introduced parasitic thermocouples obviate the desired advantage. Additionally, the aggregate stability of the LTZ1000A reference and the voltmeter references (also, incidentally, LTZ1000A based) is comfortably inside 0.1ppm for periods of 10 minutes.

4

This is more than enough time for a 10-point linearity measurement.

STABLE

VOLTAGE

SOURCE

(LTZ1000A

BASED)

KVD

ADJUST FOR

5.000000V AT A

0.1

µ

F

–

+

LTC1150

OFFSET

TRIM

+V

–V

10k

LT1010

20k

MAIN

KVD

A

R

WIRE

–

+

LTC1152

0.1

µ

F

10k

LT1010

HP3458A

HP3458A

HP3458A

Figure C6. Simplified Sub-ppm Linearity Voltage Source

OUTPUT

0.000000V

TO

5.000000V

AN86 FC06

Note 3: See Appendix J, “Cables, Connections, Solder, Layout,

Component Choice, Terror and Arcana,” for further discussion.

Note 4: The LTZ1000A reference is detailed in Appendix I, “ Voltage

References.”

AN86-10

Construction

Figures C8 and C9 are photographs of the voltage source and the reference-buffer box internal construction. The figure captions annotate some significant features.

Results

This KVD-based, high linearity voltage source has been in use in our lab for nearly two years. During this period, the total linearity uncertainty defined by the source and its monitoring voltmeters has been just 0.3ppm (see Figure

Application Note 86

C10’s measurement regime). This is more than 3 times better than the desired 1ppm performance, promoting confidence in our measurements.

5

Acknowledgments

The author is indebted to Lord Kelvin and to Warren Little of the C. S. Draper Laboratory (née M. I. T. Instrumentation

Laboratory) standards lab. Warren taught me, with great patience, the wonders of KVDs some thirty years ago and

I am still trading on his efforts.

STABLE

VOLTAGE

SOURCE

(LTZ1000A

BASED)

SEE APPENDIX F

FOR CIRCUIT DETAILS

OF LTZ1000A

ADJUST FOR

5.000000V AT A

KVD

10k

–

+

LTC1150

0.1

µ

F

10k

LT1010

A

–

+

LTC1152

0.1

µ

F

OFFSET

TRIM

2k

+V

–V

20k

MAIN

KVD

10k

2

µ

F

CASE

R

WIRE

= SOLDER-COPPER JUNCTION.

PLACEMENT AND NUMBER AS REQUIRED

2

µ

F = POLYSTYRENE. COMPONENT RSCH. CORP.

USE LOW THERMAL, LOW TRIBOELECTRIC

SHIELDED CABLE FOR KVD AND DVM CONNECTIONS.

SEE TEXT

HIGH QUALITY

GROUND

Figure C7. Complete Sub-ppm Linearity Voltage Source

10k

LT1010

HP3458A

HP3458A

HP3458A

OUTPUT

0.000000V

TO

5.000000V

AN86 FC07

Note 5: The author, wholly unenthralled by web surfing, has spent many delightful hours “surfing the Kelvin.” This activity consists of dialing various Kelvin-Varley divider settings and noting monitoring

A-to-D agreement within 1ppm. This is astonishingly nerdy behavior, but thrills certain types.

AN86-11

Application Note 86

AN86 FC08.tif

Figure C8. The Sub-ppm Linearity Voltage Source. Box Upper Right Is LTZ1000A Based Reference and Buffers. Upper Left Is Offset

Trim. Reference and Main Kelvin-Varley Dividers Are Photo Center—Upper and Center-Middle, Respectively. Three HP3458 DVMs

(Photo Lower) Monitor Output. Computer (Left Foreground) Aids Linearity Calculations

AN86-12

Application Note 86

AN86 FC09.tif

Figure C9. Reference-Buffer Box Construction. LTZ1000A Reference Circuitry Is Photo Lower Left, Buffer Amplifiers Photo Center. Note

Capacitor Case Bootstrap Connection (Photo Center—Right). Single Point Ground Mecca Appears Photo Upper Left. Power Supply

(Photo Top) Mounts Outside Box, Minimizing Magnetic Field Disturbance

1. VERIFY KVD LINEARITY BY INTERCOMPARISON AND INDEPENDENT CAL. LAB.

2. TAKE WORST-CASE VOLTMETER ENSEMBLE DEVIATIONS OVER 0V TO 5V, EVERY 0.5V

3. 100 RUNS (10 PER DAY, ONCE PER HOUR)

4. INDICATED RESULT IS 0.3ppm NONLINEARITY

AN86 FC10

Figure C10. Testing Regime for the High Linearity Voltage Source

AN86-13

Application Note 86

APPENDIX D

A PROCESSOR-BASED CODE COMPARATOR

The code comparator enforces the loop by setting the slave DAC inputs to the code that equalizes the user input and the LTC2400 A-to-D output. This action is more fully described on page one of the text.

Figure D1 is the code comparator’s digital hardware. It is composed of three input data latches and a PIC-16C5X processor. Inputs include user data (e.g., DAC inputs), linearity curvature correction (via DIP switches), convert command (“DA WR”) and a selectable filter time constant.

An output (“DAC RDY”) indicates when the DAC output is settled to the user input value. Additional outputs and an

1ppm

5V

2ppm

5V

4ppm

5V

DAC WR

LSB

5

6

7

8

2

3

4

9

11

1

10

D3

D4

D5

D6

D0

D1

D2

D7

CLK

OE

GND

U2 74HCT574

19

18

17

16

15

14

13

12

V

Q0

Q1

Q2

Q3

Q4

Q5

Q6

Q7

CC

20

0.1

µ

F

FILTER

5V

USER

DATA

INPUT

6

7

8

9

2

3

4

5

11

1

10

D0

D1

D2

D3

D4

D5

D6

D7

U3 74HCT574

Q0

Q1

Q2

Q3

Q4

Q5

19

18

17

16

15

14

Q6

Q7

13

12

CLK

OE

GND

V

CC

20

0.1

µ

F

MSB input control and monitor the analog section (text Figure␣ 2) to effect loop closure. Note that although a total of

32 bits are presented to the two 16-bit slave DACs, there are 8 bits of overlap, allowing a total dynamic range of 24 bits. This provides 4 bits of indexing range below the 20th bit, ensuring a stable LSB of 1ppm of scale. The 8-bit overlap assures the loop will always be able to capture the correct output value.

The processor is driven by software code, authored by

Florin Oprescu, which is described below.

MCLR

5V

10k

10k

10k

10k

10k

10k

10k

10k

10k

D3

D2

D1

D0

D7 MSB

D6

D5

D4

DAC

COMMANDS

TO ANALOG

SECTION

10k

10k

10k

10k

10k

DAC LD

MLBYTE

10k

MSB WR

LSB WR

ADC SDO

ADC SCK

TO LTC2400

IN ANALOG

SECTION

DAC RDY

6

7

8

9

2

3

4

5

11

1

10

D0

D1

D2

D3

D4

D5

D6

D7

U4 74HCT574

Q0

Q1

Q2

Q3

Q4

Q5

19

18

17

16

15

14

Q6

Q7

13

12

CLK

OE

GND V

CC

20

0.1

µ

F

5V

0.1

µ

F

10k

20pF

20pF

4MHz

CRYSTAL

9

10

11

12

13

14

5

6

7

8

1

2

3

4

U1

USER INPUT

V

DD

NC

V

SS

NC

ADC SCK

ADC SDO

NC

NC

LSB OE

MID OE

MSB OE

LSB WR

MSB WR

PIC16C5X

MCLR

CLKIN

CLKIN2

MSB D7

D6

D5

D4

D3

D2

D1

LSB D0

DAC RDY

DAC LD

MLBYTE

28

27

26

25

24

23

22

21

20

19

18

17

16

15

Figure D1. Code Comparator Hardware. User Control Lines Are at Left, Analog Section Connections Appear at Right Side

AN86 FD01

AN86-14

Application Note 86

;20bit DAC code comparator

;

;***************************************************

; *

; Filename:

; Date

; File Version:

; dac20.asm

12/4/2000

1.1

*

*

*

*

; Author:

; Company:

Florin Oprescu *

Linear Technology Corp. *

;

;

*

*

;***************************************************

;

; Variables

;============

; uses 17 bytes of RAM as follows:

;

; {UB2, UB1, UB0} user input word buffer

;———————————————————————————————————————

; 24 bits unsigned integer (3 bytes):

;

; The information is transferred from the external input register

; into {UB2, UB1, UB0} whenever a “user input update” event

; is detected by testing the timer0 content. Following the data

; transfer, the UIU (“user input update”) flag is set and the DAC

; ready flags RDY and RDY2 are cleared. UB0 uses the same physical

; location as U0. The user input double buffering is necessary

; because the loop error corresponding to the current ADC reading

; must be calculated using the previous user input value.

; The old user input value can be replaced by the new user input

; value only after the loop error calculation.

; The worst case minimum response time to an UIU event must be

;

;

;

;

; calculated. The user shall not update the external input register

; at intervals shorter than this response time. For the moment the

; program can not block the user access to the external input

; register during a read operation. In such a situation the result

;

;

; of the read operation can be very wrong.

;

UB0 - least significant byte. Same physical location as U0

UB1 - second byte.

UB2 - most significant byte.

;

; {U2, U1, U0} user input word

;——————————————————————————————

; 24 bits unsigned integer (3 bytes):

;

; The information is transferred from {UB2, UB1, UB0[7:4], [0000]}

; into {U2, U1, U0} whenever the UIU flag is found set within the

; CComp (“code comparator”) procedure. The UIU flag is reset

; following the data transfer.

AN86-15

Application Note 86

;

;

;

;

;

;

;

; U0 -

U1 least significant byte of current DAC input

The 4 least significant bits U0[3:0] are set to zero.

second byte of current DAC input

U2 most significant byte of current DAC input

;

;

; {CON} control byte

;————————————————————

; (1 byte):

;

; The 3 least significant bits CON[2:0] represent the ADC linearity

; correction factor transferred from UB[2:0] when the UIU flag

; is found set within the CComp procedure - at the same time as the

; {U2, U1, U0} variable is updated.

;

; The effect of CON[2:0] is additive and its weight is as follows:

;

;

;

; CON[0] = 1 linearity correction effect is about 1ppm

CON[1] = 1 linearity correction effect is about 2ppm

CON[2] = 1 linearity correction effect is about 4ppm

;

; The LTC2400 has a typical 4ppm INL error therefore the default

; curvature correction value can be set at CON[2:0] = 100

;

; CON[3] is the control loop integration factor transferred from

; UB[3] when the UIU flag is found set within the CComp procedure.

; If CON[3]=0, after the control loop error becomes less than 4ppm

; the error correction gain is reduced from 1 to 1/4

; If CON[3]=1, after the control loop error becomes less than 4ppm

; the error correction gain is reduced from 1 to 1/16

;

; CON[7] is used as the UIU (“user input update”) flag. It is set

; when {UB2, UB1, UB0} is updated and it is reset when {U2, U1, U0}

; and CON[3:0] are updated.

;

; CON[6] is used as the RDY (“DAC ready”) flag. It is set when

; the DAC loop error becomes less than 4ppm and it is reset when

; the UIU flag is set.

;

; CON[5] is used as the RDY2 (“DAC ready twice”) flag. It is set

;

;

; whenever the DAC loop error becomes less than 4ppm and the RDY

; flag has been previously set. It is reset when the UIU flag is set.

; The bit CON[4] is not used and is always set to 0.

;

;

AN86-16

Application Note 86

; {ADC3, ADC2, ADC1, ADC0} formatted ADC conversion result

;—————————————————————————————————————————————————————————

; 32 bits signed integer (4 bytes).

;

; The ADC reading is necessary only for the calculation of the control

; loop error and in order to save RAM space, it can share the same

; physical space as the loop error variable.

;

; The LTC2400 output format is offset binary. It must be converted

; to 2’s complement before any arithmetic operation. A number of

; possible codes are not valid LTC2400 output codes. If such codes

; are detected it can be inferred that a serial transfer error has

; occurred, the data must be discarded and a new conversion must

; be started. For all LTC2400 devices B31=0 and B30=0 always. In

; addition, with the exception of some early samples of the device

; the sequence B[29:28]=00 should not occur in a valid transaction.

;

;

;

; ADC0 - least significant byte

contains ADC output bits B11(MSBIT) to B4 (LSBIT)

;

;

;

;

ADC1 - second byte

contains ADC output bits B19(MSBIT) to B12 (LSBIT)

;

;

;

;

ADC2 - third byte

contains ADC output bits B27(MSBIT) to B20 (LSBIT)

ADC3 - most significant byte

contains ADC output bits ~B29(as 7 MSBITS for

2’s complement sign extension) and B28 (LSBIT) ;

;

;

; {ADCC} ADC curvature correction

;————————————————————————————————

; 8 bits unsigned integer (1 byte)

;

; The LTC2400 transfer characteristic has a typical INL of about

; 4ppm and a parabolic shape symmetric with respect to mid-scale.

; This error can be corrected to a first and second order and

; ADDC contains the magnitude of this correction.

;

;

; {ER3, ER2, ER1, ER0} control loop error value

;——————————————————————————————————————————————

; signed integer (4 bytes)

;

; Contains the value of the current control loop error calculated

; as the difference between the previous user input and the last

; ADC reading. It is used to adjust the Low_DAC setting. Uses the

;

;

; same physical location as {ADC3, ADC2, ADC1, ADC0}:

;

ER0 - least significant byte, same location as ADC0

; ER1 - second byte, same location as ADC1

AN86-17

Application Note 86

;

;

;

;

;

;

;

;

;

;

;

; ER2 - third byte, same location as ADC2

ER3 - most significant byte, same location as ADC3

;

;

; {DL3, DL2, DL1, DL0} Low DAC control value

;———————————————————————————————————————————

;

;

;

;

; signed integer (4 bytes):

;

; Contains the Low_DAC setting in a 2’s complement, 32 bit

; format. Must be initialized to 0!

DL0 - least significant byte - used for Low_DAC

control

DL1 - second byte - used for Low_DAC control after

conversion to offset binary format {DL1, DL0}

DL2 - third byte - may be only 00 or FF

DL3 - most significant byte - may be only 00 or FF

; {INDX} Index variable for various program functions

;————————————————————————————————————————————————————

; 1 byte.

;

;

; {TMPV} Temporary variable for various program functions

;————————————————————————————————————————————————————————

; 1 byte.

;

;

;

; Algorithm

;===========

;

; After each ADC conversion cycle the processor calculates the control

; loop error value as the difference between the desired output and

; the latest conversion result. Than it updates the DACs command

; such as to reduce the error magnitude. A new ADC conversion cycle

; is started following the DACs update operation.

;

; In order to maintain adequate control loop stability it is necessary

; for the DACs and the associated amplifiers to settle to better than

; 20 bits accuracy before the ADC starts sampling the system output. For

; an LTC2400 based system this settling time is 66ms.

;

; Initialization:

; Initializes the PIC controller and the hardware interface

; and starts the Scan procedure.

;

AN86-18

Application Note 86

; 1. Load ADC control port with default values

; SCKAD = 0

; SDOAD = 1

; 2. Set ADC control port I and O pins

;

;

SCKAD = output

SDOAD = input

; 3. Load register control port with default values

; NCSR[2:0] = 111

;

;

;

;

NCSD[1:0] = 11

ADDAC = 1

NLDAC = 1

DACRDY = 0

; 4. Set register control port in output mode

; 5. Set data bus to default value DBUS[7:0]=0x00

; 6. Set data bus in output mode

; 7. Initialize internal registers and variables:

;

;

; OPTION = 0x2F

; Timer0 used as counter is incremented by low-to-high edge

Prescaler works with watch dog timer in div128 mode

CON = 0x80

;

;

; Simulate a UIU “user interface update” event to force

; the update of both Low_DAC and High_DAC

{DL3, DL2, DL1, DL0} = 0

{ U2, U1, U0} = 0

; 8. Update hardware using the initialized variables

; 9. Start new ADC conversion by reading and discarding

; 32 serial bits.

; 10.Start the Scan procedure

;

; Scan:

; Continuously looks for “user input update” events. When

; a “user input update” event is detected updates the

; input buffer {UB2, UB1, UB0}, resets timer, sets UIU flag

; and resets RDY and RDY2 flags.

;

; The active low write signal for the external input register

; (which is the same as the user interface NWR input signal)

; is driven by the user and it is connected to the counter

; input of Timer0. The Timer0 is used in counter mode without a

; prescaler and it increments whenever a low-to-high transition

; is detected at its input. This is the same transition which

; latches in the input register a new user command.

; Because of the PIC controller timing constraints, this write

; signal must be maintained low for at least 2*Tosc + 20ns

; where Tosc is the maximum PIC clock period. When a 4 MHz

; clock is used for the PIC processor, the low time must be

; longer than about 520ns.

;

;

;

; 1. Test for “user input update” events by testing the Timer0

; value.

;

If Timer0>0 an UIU event has occurred. Reset the timer

and answer Yes.

If Timer0=0 answer No.

AN86-19

Application Note 86

;

;

; 1.1 If Yes, read input latch into {UB2, UB1, UB0},

; reset DACRDY output line, set UIU flag and

and reset RDY and RDY2 flags (CON[7:5]=100)

Than continue

; 1.2 If No continue

;

; Continuously looks for the ADC end of conversion event. When

; the “end of conversion” is detected it reads the 28 most

;

;

; significant bits from the ADC and it constructs the ADC

; result {ADC3, ADC2, ADC1, ADC0} in 2’s complement format

If ADC3[1] == 0 => ADC3[7:1]=1111 111

If ADC3[1] == 1 => ADC3[7:1]=0000 000

;

;

; For very early LTC2400 samples only, it is possible

; to obtain as a valid 0 conversion result ADC3[1:0]=00

In this case:

If ADC3[1:0] == 0 => ADC3=0

; It also calculates the first (x1) and second (x2) order ADC

; curvature correction ADCC as follows:

; x1 = {0x00, 0x80} -

; -abs({ADC3, ADC2, ADC1, ADC0}/(2^16)-{0x00, 0x80})

; x2 = {0x00, 0x40} -

; -abs({0x00,{0,ADC2[6:0]},ADC1,ADC0}/(2^16)-{0x00,0x40})

; ADCC = floor((x1 + x2/2) * {00000 CON[2:0]} / 4 )

; The actual implementation uses only the least significant

;

;

; byte of x without any substantial additional error.

; Thus the above relation can be modified as follows:

ADCC = floor((abs(ADC2) + abs({ADC2[6],ADC2[6:0]})/2) *

* {00000 CON[2:0]} / 4 )

; The maximum correction range is about 7ppm INL at mid

; scale for CON[2:0] = 111.

;

; 2. Test for ADC “end of conversion” event by testing the

;

;

;

;

; value of the ADC_SDO signal.

; If ADC_SDO = LOW answer Yes.

; If ADC_SDO = HIGH answer No.

; 2.1 If Yes read 28 most significant bits from the ADC,

update {ADC3, ADC2, ADC1, ADC0} and calculate the

curvature correction byte ADCC. Than start the CComp

procedure.

It should be noticed that while reading the first 28

; most significant bits from the ADC the controller

; generates 27 serial clock pulses. An additional 5 serial

; clock pulses (for a total of 32) are necessary to restart

; the conversion.

; 2.2 If No restart the Scan procedure.

;

;

; CComp:

; Calculates the current control loop error as:

;

; error = current_user_input - ADC_reading +

; + new_user_input_LSB - current_user_input_LSB

;

AN86-20

Application Note 86

; The curvature correction is included in the ADC

; conversion result and is always positive therefore:

;

; ADC_reading = {ADC3, ADC2, ADC1, ADC0} +

; + { 0, 0, 0, ADCC}

;

; The term “new_user_input_LSB - current_user_input_LSB”

; represents the residue of the new user command which

; is added to the Low_DAC.

;

; {ER3, ER2, ER1, ER0} =

; = {0, U2, U1, U0} - {ADC3, ADC2, ADC1, ADC0} -

; - { 0, 0, 0, ADCC} +

; + {0, 0, 0, UB0} - { 0, 0, 0, U0} =

;

; = {0, U2, U1, UB0} - {ADC3, ADC2, ADC1, ADC0} -

; - { 0, 0, 0, ADCC}.

;

; The loop error {ER3, ER2, ER1, ER0} is a 32 bit signed number

; and the weight of the least significant bit is 1/16ppm of

; the ADC reference voltage. A 4ppm error value is represented

; as {0, 0, 0, 0x40}.

;

; The ADC output noise is dominated by thermal noise and has a

; white distribution. The control loop noise can be reduced by

; the square root of N by averaging N successive ADC readings.

; The obvious penalty is a slow settling time. Due to the

; limited amount of RAM available a direct implementation

; of this noise reduction strategy is difficult. In an equivalent

; implementation, when the absolute value of the loop error

; {ER3, ER2, ER1, ER0} decreases below a certain threshold, the

; gain of the error correction loop can be decreased. The default

; threshold is set at a very conservative 4ppm. This value must

; always be larger than the peak noise level of the ADC. A very

; quiet implementation can probably operate with a threshold of

; 2ppm. If CON[3]=0 the gain of the error correction loop is

; decreased from 1 to 1/4. If CON[3]=1 the gain of the error

; correction loop is decreased from 1 to 1/16.

;

; The High_DAC is always controlled by the 16 most significant

; bits of the most recent user command {UB2, UB1}

;

; The Low_DAC is controlled by the {DL3, DL2, DL1, DL0}

; variable which integrates the control loop error. Under

; correct operating condition abs({DL3, DL2, DL1, DL0})<2^15.

; In order to avoid roll-overs during large transients the

; {DL3, DL2, DL1, DL0} must be clamped within the +/- 2^15 range.

; The 16 bit Low_DAC can than be controlled by {DL1, DL0}

; after conversion to offset binary format.

;

AN86-21

Application Note 86

; The DACRDY output line reflects the state of the

; internal RDY2 flag.

;

; After the updates are completed we must start a new ADC

; conversion by completing the serial transfer.

;

; 1. Test if UIU flag is set

; 1.1 If Yes, move UB[3:0] into CON[3:0]

; and {UB0[7:4], 0000} into U0. The last ADC result

; is curvature corrected using the previous CON[3:0] value!.

; 2. Calculate {ER3, ER2, ER1, ER0}.

; 3. Test if UIU flag is set

; 3.1 If Yes, move {UB2, UB1} into {U2, U1} and

; clear UIU, RDY and RDY2 flags (CON[7:5]=000 )

; 3.2 If No, test if abs({ER3, ER2, ER1, ER0}) < 4ppm

; 3.2.1 If Yes, test if CON[6]=1 (RDY flag)

; 3.2.1.1 If Yes, set RDY2 flag (CON[5]=1 )

; 3.2.1.2 If No, set RDY flag (CON[6]=1 )

; and test if CON[3]=0 (filter flag)

; 3.2.1.3 If Yes, {ER3, ER2, ER1, ER0} =

; = {ER3, ER2, ER1, ER0}/4

; 3.2.1.4 If No, {ER3, ER2, ER1, ER0} =

; = {ER3, ER2, ER1, ER0}/16

; 3.2.2 If No, clear UIU, RDY and RDY2

; flags (CON[7:5]=000 )

; 4 {DL3, DL2, DL1, DL0} = {DL3, DL2, DL1, DL0} +

; +{ER3, ER2, ER1, ER0}.

; 5. Update High_DAC, Low_DAC and DACRDY output line

; 6. Read the 4 least significant bits from ADC and start

; a new conversion

; 7. Restart the Scan procedure

;

;

; Hardware resources

;====================

;

; Uses 8 input/output pins, 9 output pins, 1 input pin and 1

; counter input pin

;

; DBUS[7:0] data bus

;———————————————————

; 8 bit bi-directional data bus is used to read the 20 bit input

; command IC[19:0], the one bit filter selection FS[0] and the 3 bit

; curvature correction selection CC[2:0]. It is also used to write

; the 16 bit Low_DAC command LDAC[15:0] and the 16 bit High_DAC

; command HDAC[15:0].

;

; assigned to PIC port C[7:0]

;

AN86-22

Application Note 86

; The data format for the read and write operations is as follows:

;

; DBUS[ 7:0] = IC[19:12] when NCSR[2] = 0

; DBUS[ 7:0] = IC[11: 4] when NCSR[1] = 0

; DBUS[ 7:0] = {IC[3:0], FS[0], CC[2:0]} when NCSR[0] = 0

; LDAC[ 7:0] = DBUS[7:0] when NCSD[0] = 0 and ADDAC = 0

; LDAC[15:8] = DBUS[7:0] when NCSD[0] = 0 and ADDAC = 1

; HDAC[ 7:0] = DBUS[7:0] when NCSD[1] = 0 and ADDAC = 0

; HDAC[15:7] = DBUS[7:0] when NCSD[1] = 0 and ADDAC = 1

;

;

; NCSR[2:0] active low output enable controls for input registers

;————————————————————————————————————————————————————————————————

; 3 output lines used to selectively enable the three 8-bit input

; registers in order to read the user updated DAC command, the 3

; curvature correction bits and the one filter control bit.

;

; NCSR[0] enables the low input byte (LSB) and is assigned to port B[0]

;

; NCSR[1] enables the second input byte and is assigned to port B[1]

;

;

;

; NCSR[2] enables the high input byte (MSB) and is assigned to port B[2]

; NCSD[1:0] active low input enable controls for the DACs

;————————————————————————————————————————————————————————

; 2 output lines used to selectively enable the two DACs

;

; NCSD[0] enables the Low_DAC and is assigned to port B[3]

;

; NCSD[1] enables the High_DAC and is assigned to port B[4]

;

;

;

;

; ADDAC DAC address control

;——————————————————————————

; output line. A low enables a write operation to the low byte of

; Low_DAC or High_DAC. A high enables a write operation to the high

; byte of Low_DAC or High_DAC.

;

; ADDAC is assigned to port B[5]

AN86-23

Application Note 86

; NLDAC active low DAC load control

;——————————————————————————————————

; output line. A high to low transition on this line updates the

; Low_DAC and High_DAC output values

;

;

;

; NLDAC is assigned to port B[6]

; DACRDY active high ready output signal

;———————————————————————————————————————

; output line. Indicates that the control loop error has been

; within a +/- 4ppm range for at least 250 ms

;

;

;

; DACRDY is assigned to port B[7]

; SCKAD external serial clock line for the ADC

;—————————————————————————————————————————————

; output line. ADC external serial clock. An external 10Kohm

; pull-down resistor is necessary on this line for correct

; power-up configuration.

;

; SCKAD is assigned to port A[0]

;

;

; SDOAD serial data line from ADC

;————————————————————————————————

; input line. Used to read ADC serial data.

;

;

;

; SDOAD is assigned to port A[1]

;

;

;

; NWRUI active low user interface write control

;——————————————————————————————————————————————

; input line. The user must bring this line low in order to update

; the DAC input value. A minimum low and high time is required !

;

; NWRUI is assigned to TOCKI counter input pin

;

AN86-24

Application Note 86

; The spare I/O pins A[3:2] are configured as outputs and maintained LOW.

;

;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;; list

#include

__CONFIG p=16c55A

<p16c5x.inc>

; list directive to define processor

; processor specific variable definitions

_CP_OFF & _WDT_ON & _XT_OSC

ER0

ER1

ER2

ER3

DL0

DL1

DL2

DL3

ADC0

ADC1

ADC2

ADC3

ADCC

INDX

TMPV

;VARIABLE DEFINITIONS

;====================

UB0

UB1

UB2

EQU

EQU

EQU

H’0008'

H’0009'

H’000A’

U0

U1

U2

CON

EQU

EQU

EQU

EQU

H’0008'

H’000B’

H’000C’

H’000D’

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

EQU

H’000E’

H’000F’

H’0010'

H’0011'

H’0012'

H’000E’

H’000F’

H’0010'

H’0011'

H’0013'

H’0014'

H’0015'

H’0016'

H’0017'

H’0018'

#define OPRDF 0x2F

#define CONDF 0x80

; user input word buffer LSB

; user input word buffer second byte

; user input word buffer MSB

; user input word LSB

; user input word second byte

; user input word MSB

; control byte

; ADC conversion result LSB

; ADC conversion result second byte

; ADC conversion result third byte

; ADC conversion result MSB

; ADC curvature correction byte

; control loop error LSB

; control loop error second byte

; control loop error third byte

; control loop error MSB

; Low_DAC LSB

; Low_DAC second byte

; Low_DAC third byte

; Low_DAC MSB

; index variable

; temporary variable

; OPTION register default value

; CON register default value

AN86-25

Application Note 86

;HARDWARE ASSIGNMENT DEFINITIONS

;===============================

#define DBUS PORTC ; 8bit I/O data bus

#define REGCN PORTB

#define REGDF 0x7F

#define NCSR0 PORTB,0

#define NCSR1 PORTB,1

#define NCSR2 PORTB,2

#define NCSD0 PORTB,3

#define NCSD1 PORTB,4

#define ADDAC PORTB,5

#define NLDAC PORTB,6

#define DACRDY PORTB,7

#define ADCCN PORTA

#define ADCTR 0x02

#define ADCDF 0x02

#define SCKAD PORTA,0

#define SDOAD PORTA,1

; register control port

; register control port default value

; LSB input register active low output enable

; second byte input register active low output enable

; MSB input register active low output enable

; Low_DAC active low write enable

; High_DAC active low write enable

; address bit for Low_DAC and High_DAC

; active low load control for Low_DAC and High_DAC

; 20bit_DAC ready indicator

; ADC control port

; ADC control port configuration

; SDOAD input, the rest outputs

; ADC control port default value

; ADC external serial clock

; ADC serial data output

;THE CODE

;===============================

RESET ORG 0x1FF goto start

; processor reset vector

ORG 0x000 start movlw movwf movlw tris movlw movwf clrw tris movwf tris movlw option

ADCDF

ADCCN

ADCTR

ADCCN

REGDF

REGCN

REGCN

DBUS

DBUS

OPRDF

;Initialization procedure

;————————————————————————

;write ADC control port default value

;

;set the I and O pin states for the

;ADC control port

;

;write register control port default value

;

;set register control port pins as

;output only

;

;

;

;set DBUS default value of 0

;set DBUS as output

;

;set OPTION register default value

AN86-26

fladc rddmy rduiu

Application Note 86

clrf TMR0 ; btfss STATUS,NOT_TO ;if this is not a power-on reset movwf TMR0 ;load Timer0 with a nonzero value

;to force an initial read of the clrf clrf clrf DL1 clrf DL0 clrf clrf

DL3

DL2

U2

U1 clrf U0

;external input register

;

;initialize {DL3, DL2, DL1, DL0}=0

;

;

;

;initialize {U2, U1, U0}=0

; movlw movwf

CONDF

CON

;

;

;set CON variable default value

; movlw movwf goto

0x20

INDX iupdt

;prepare to trigger a new ADC conversion

;after completing a hardware update

;read and discard 32 serial bits from

;the ADC

;

;go to the hardware update function movlw movwf

0x20

INDX

;ADC output buffer flush function

;————————————————————————————————

;reads and discards 32 serial bits from

;the ADC bsf bcf decfsz INDX,1 goto rddmy btfss SDOAD goto fladc goto

SCKAD

SCKAD scan movlw 0xFF tris DBUS bcf nop bcf

NCSR0 movf DBUS,0 bsf NCSR0

NCSR1 movwf UB0 movf DBUS,0 bsf NCSR1

;ADC dummy serial read function

;——————————————————————————————

;reads and discards the number of serial

;bits indicated by the INDX variable

;low-to-high ADC serial clock edge

;high-to-low ADC serial clock edge

;test if we read enough bits

;if No, read one more bit

;if Yes test that a new conversion has started

;if No, there is an interface problem. Flush the

;ADC output buffer and start a new conversion

;if Yes restart the scan procedure

;external input register read function

;—————————————————————————————————————

;input register read routine

;set data bus in read mode (input)

;output enable for input reg. LSB

;wait for data bus to settle

;read input reg. LSB

;output disable for input reg. LSB

;output enable for input reg. second byte

;store input reg. LSB into input buffer

;read input reg. second byte

;output disable for input reg. second byte

AN86-27

Application Note 86

bcf NCSR2 movwf UB1 movf DBUS,0 movwf UB2 clrw bsf NCSR2 tris DBUS clrf TMR0 bcf DACRDY bsf bcf bcf

CON,7

CON,6

CON,5 scan rdadc rdbit movf TMR0,1 btfss STATUS,Z goto rduiu btfsc SDOAD goto scan movlw 0x1B movwf INDX bsf bcf

SCKAD

SCKAD bcf STATUS,C btfsc SDOAD bsf rlf rlf rlf

STATUS,C

ADC0,1

ADC1,1

ADC2,1 rlf ADC3,1 decfsz INDX,1 goto rdbit

;output enable for input reg. MSB

;store input reg. second byte into input buffer

;read input reg. MSB

;store input reg. MSB into input buffer

;terminate input reg. read operation

;output disable for input reg. MSB

;return data bus to write mode

;clear Timer0 to continue wait for a UIU event

;signal user that input data has been read

;set UIU flag

;clear RDY flag

;clear RDY2 flag

;scan procedure

;——————————————

;monitors UIU and end-of-conversion events

;test if Timer0 = 0

;if Timer0=0 no UIU has occurred, skip next

;a user interface update has occurred

;go and read the new DAC input data

;test ADC end of conversion signal

;conversion not ready, rescan

;ADC serial read function

;————————————————————————

;ADC conversion is done, read first 28 bits

;the first bit must be “0” to get here

;so do not bother with it

;low-to-high ADC serial clock edge

;high-to-low ADC serial clock edge

;move ADC output bit to carry. First clear carry

;read ADC output bit

;if ADC output is “1” set carry

;load carry as msb of ADC result

;and shift left all 4 bytes of the ADC result

;

;

;test if all 28 bits have been read

;if not, continue

;

;we have skipped the first ADC bit (ADC bit31=0)

;which has been tested as =0 when we detected the

;end of conversion.

;we have read 27 additional bits and have generated

;27 clock pulses. To restart the conversion we must

;produce the 5 remaining clock pulses

AN86-28

; btfsc ADC3,2 goto fladc movlw 0x03 andwf ADC3,1 btfsc STATUS,Z goto rdend goto fladc movlw 0x02 btfss ADC3,1 movlw 0xFE xorwf ADC3,1 rdend movf ADC2,0 btfsc ADC2,7 comf ADC2,0 movwf ADCC movf ADC2,0 btfsc ADC2,6 comf ADC2,0 movwf TMPV rrf TMPV,0 andlw 0x1f addwf ADCC,0 movwf TMPV clrf ADCC bcf STATUS,C btfsc CON,2

Application Note 86

;verify validity of ADC serial data and format it

;————————————————————————————————————————————————

;test if the ADC bit30 is “0”

;if not there is an interface problem. Flush the

;ADC output buffer and start a new conversion

;if yes, put the ADC result in 2’s complement form

;first clear the 6 most significant bits in ADC3

;

;tests for the [ADC_B29,ADC_B28]=00 ADC output

;if Yes the formatting is completed.

;in very early LTC2400 samples the ADC output code

;[ADC_B29,ADC_B28]=00 is valid

;for current LTC2400 devices improved error

;detection capability is obtained if the

;previous line is replaced with this line.

;The replacement is not mandatory.

;For current LTC2400 parts the output code

;[ADC_B29,ADC_B28]=00 is not valid thus it may

;be assumed that an ADC interface error has

;occurred. Flush the ADC output buffer and start

;a new conversion

;

;

;if No, convert ADC3 in 2’s complement form

;

;curvature correction calculator

;———————————————————————————————

;first order curvature correction multiplier

;use ADC2[7:0] as a 2’s complement number

;calculate abs(ADC2)

;if ADC2[7]=0 w = ADC2

;else w = !ADC2

;ADCC=w=abs(ADC2)

;second order curvature correction multiplier

;use ADC2[6:0] as a 2’s complement number

;calculate abs(ADC2[6:0])

;if ADC2[6]=0 w = ADC2

;else w = !ADC2

;TMPV=w=abs(ADC2[6:0])

;w=TMPV/2 in order to scale the second order

;curvature correction

;clear 3 MSB of w to complete calculation

;w=abs(ADC2)+abs(ADC2[6:0])/2

;TMPV contains the curvature correction multiplier

;

;

;clear carry for div-by-2 operation

;if CON[2]=1

AN86-29

Application Note 86

movwf ADCC rrf TMPV,1 movf TMPV,0 bcf STATUS,C btfsc CON,1 addwf ADCC,1 rrf TMPV,1 movf TMPV,0 btfsc CON,0 addwf ADCC,1 ccomp ercalc btfss CON,7 goto ercalc movlw 0xF0 andwf CON,1 movlw 0x0F andwf UB0,0 iorwf CON,1 movlw 0xF0 andwf UB0,1 comf ADCC,1 comf ADC0,1 movlw 0x02 clrf TMPV addwf UB0,0 btfsc STATUS,C incf TMPV,1 addwf ADCC,0 btfsc STATUS,C incf TMPV,1 addwf ADC0,1 btfsc STATUS,C incf TMPV,1

;ADCC=ADCC+abs(ADC2)

;TMPV=TMPV/2

;

;clear carry for div-by-2 operation

;if CON[1]=1

;ADCC=ADCC+abs(ADC2)/2

;TMPV=TMPV/2

;

;if CON[0]=1

;ADCC=ADCC+abs(ADC2)/4

;code comparator procedure

;—————————————————————————

;if the UIU flag is clear

;skip CON[3:0] and U0 update

;else update CON[3:0]

;clear CON[3:0]

;extract new CON[3:0]

;from input buffer

;and load it

;move UB[7:4] to U0[7:4]

;UB0 and U0 use the same

;physical location

;calculate control loop error

;————————————————————————————

;ADCC 1’s complement

;ADC0 1’s complement

;add carry-in for ADCC and for ADC0

;2’s complement conversion

;prepare carry-out accumulator

;w=carry-in + UB0

;if there is a carry-out

;accumulate it

;w=carry-in + UB0 - ADCC

;if there is a carry-out

;accumulate it

;ER0=UB0 - ADC0 - ADCC

;has same location as ADC0

;if there is a carry-out

;accumulate it

AN86-30

comf ADC1,1 movlw 0xff addwf TMPV,0 clrf TMPV btfsc STATUS,C incf TMPV,1 addwf U1,0 btfsc STATUS,C incf TMPV,1 addwf ADC1,1 btfsc STATUS,C incf TMPV,1 comf ADC2,1 movlw 0xff addwf TMPV,0 clrf TMPV btfsc STATUS,C incf TMPV,1 addwf U2,0 btfsc STATUS,C incf TMPV,1 addwf ADC2,1 btfsc STATUS,C incf TMPV,1 comf ADC3,1 decf TMPV,1 movf TMPV,0 addwf ADC3,1 btfsc CON,7 goto lduiu

Application Note 86

;ADC1 1’s complement

;w=0xff (1’s complement of ADCC second byte)

;w=0xff + carry-in

;prepare carry-out accumulator

;if there is a carry-out

;accumulate it

;w=0xff + carry-in + UB1

;if there is a carry-out

;accumulate it

;ER1=U1 - ADC1 - 0 + carry-in

;has same location as ADC1

;if there is a carry-out

;accumulate it

;ADC2 1’s complement

;w=0xff (1’s complement of ADCC third byte)

;w=0xff + carry-in

;prepare carry-out accumulator

;if there is a carry-out

;accumulate it

;w=0xff + carry-in + UB2

;if there is a carry-out

;accumulate it

;ER2=U2 - ADC2 - 0 + carry-in

;has same location as ADC2

;if there is a carry-out

;accumulate it

;ADC3 1’s complement

;ADCC 2’s complement term. The next

;carry-in is not useful - discard

;w=carry-in

;ER3= 0 - ADC3 - 0 + carry-in

;has same location as ADC3

;test if the UIU flag is set

;go to U1, U2 update

AN86-31

Application Note 86

movf ER3,0 btfsc ER3,7 comf ER3,0 btfss STATUS,Z goto nrdy movf ER2,0 btfsc ER3,7 comf ER2,0 btfss STATUS,Z goto nrdy movf ER1,0 btfsc ER3,7 comf ER1,0 btfss STATUS,Z goto nrdy movf ER0,0 btfsc ER3,7 comf ER0,0 andlw 0xC0 btfss STATUS,Z goto nrdy btfsc bsf bsf

CON,6

CON,5

CON,6

;error comparator

;————————————————

;calculate absolute value of loop error and

;compare loop error magnitude with the 4ppm

;threshold

;W = ER3

;test if {ER3, ER2, ER1, ER0} < 0

;if yes W = -ER3

;test if W=0

;if not absolute error >= 4ppm

;W = ER2

;test if {ER3, ER2, ER1, ER0} < 0

;if yes W = -ER2

;test if W=0

;if not absolute error >= 4ppm

;W = ER1

;test if {ER3, ER2, ER1, ER0} < 0

;if yes W = -ER1

;test if W=0

;if not absolute error >= 4ppm

;W = ER0

;test if {ER3, ER2, ER1, ER0} < 0

;if yes W = -ER0

;keep only W[7:6] which are bits >= 4ppm

;test if W[7:6]=0

;if not absolute error >= 4ppm

;if we are here the absolute loop error is

;less than 4 ppm. Set the flags and

;scale the loop error.

;test if RDY flag is already set

;if Yes, set RDY2 flag

;set RDY flag in any case

AN86-32

div4 btfsc CON,3 goto div4 rrf rrf

ER0,1

ER0,0 andlw 0x3F btfsc ER3,7 iorlw 0xC0 movwf ER0 rrf rrf

ER0,1

ER0,0 andlw 0x3F btfsc ER3,7 iorlw 0xC0 movwf ER0 goto eracc lduiu nrdy movf UB1,0 movwf U1 movf UB2,0 movwf U2 movlw 0x1F andwf CON,1 eracc movf ER0,0 clrf TMPV addwf DL0,1 btfsc STATUS,C incf TMPV,1 movf TMPV,0 clrf TMPV addwf ER1,0 btfsc STATUS,C incf TMPV,1 addwf DL1,1 btfsc STATUS,C incf TMPV,1 movf TMPV,0 clrf TMPV addwf ER2,0

Application Note 86

;error scaling

;—————————————

;reduce error correction value for loop

;damping and ADC noise reduction

;test if CON[3]=0

;if Yes ER0=ER0/4

;if No ER0=ER0/16

;*1/2

;*1/2

;clear 2 most significant bits

;if {ER3, ER2, ER1, ER0} < 0

;set 2 most significant bits

;ER0=ER0/4

;*1/2

;*1/2

;clear 2 most significant bits

;if {ER3, ER2, ER1, ER0} < 0

;set 2 most significant bits

;ER0=ER0/4

;go to error accumulator

;load latest user input

;——————————————————————

;

;U1=UB1

;

;U2=UB2

;

;clear UIU, RDY and RDY2 flags

;error accumulator

;—————————————————

;adds the current loop error to the

;previous Low_DAC control value

;{DL3, DL2, DL1, DL0}={DL3, DL2, DL1, DL0}+

; +{ER3, ER2, ER1, ER0}

;the carry-in is 0

;clear carry-in accumulator

;DL0=DL0+ER0

;if there is a carryout

;accumulate in carry-in

;load carry-in

;clear carry-in accumulator

;W=ER1+carry-in

;if there is a carryout

;accumulate in carry-in

;DL1=DL1+ER1

;if there is a carryout

;accumulate in carry-in

;load carry-in

;clear carry-in accumulator

;W=ER2+carry-in

AN86-33

Application Note 86

btfsc STATUS,C incf TMPV,1 addwf DL2,1 btfsc STATUS,C incf TMPV,1 movf TMPV,0 addwf ER3,0 addwf DL3,1 ovflow btfss STATUS,Z goto negpot movf DL2,1 btfss STATUS,Z goto ovflow btfsc DL1,7 goto ovflow goto updt clrf DL3 clrf DL2 movlw 0xFF movwf DL0 movwf DL1 bcf DL1,7 goto updt

;if there is a carryout

;accumulate in carry-in

;DL2=DL2+ER2

;if there is a carryout

;accumulate in carry-in

;load carry-in

;W=ER3+carry-in

;DL3=DL3+ER3

;Low_DAC control truncation

;——————————————————————————

;limits the {DL3, DL2, DL1, DL0} range to

; abs({DL3, DL2, DL1, DL0}) < 2^15 by

;truncation

;test if DL3=0

;if No, DL may be negative

;if Yes, DL is positive

;test for overflow (>= 2^15)

;

;test if DL2=0

;if No, DL >= 2^15, must truncate

;if Yes continue testing for overflow

;test if DL1[7]=1

;if No, DL >= 2^15, must truncate

;if Yes we are done with DL range control

;if we arrive here DL >= 2^15. Must

;truncate to DL=2^15-1 => DL3=DL2=0

;and DL1=0xEF, DL0=0xFF

;

;

;

;done with overflow correction

AN86-34

udflow negpot clrf DL1 bsf DL1,7 clrf DL0 movlw 0xFF movwf DL3 movwf DL2 goto updt btfss DL3,7 goto ovflow incf DL3,0 btfss STATUS,Z goto udflow incf DL2,0 btfss STATUS,Z goto udflow btfss DL1,7 goto udflow updt movlw 0x05 movwf INDX

Application Note 86

;if we arrive here DL < -2^15. Must

;truncate to DL=-2^15-1 => DL3=DL2=0xFF

;and DL1=0x80, DL0=0

;

;

;

;done with underflow correction

;DL may be negative. Test if DL3[7]=1

;if No, DL > 2^15, must truncate

;if Yes, DL <0.

;test if DL3=FF

;if No, DL < -2^15, must truncate

;if Yes continue testing for underflow

;test if DL2=FF

;if No, DL < -2^15, must truncate

;if Yes continue testing for underflow

;test if DL1[7]=0

;if No, DL < -2^15, must truncate

;if Yes we are done with DL range control

;Hardware update function

;————————————————————————

;Low_DAC and High_DAC update

;

;This is the hardware update function

;entry point for normal operation.

;

;prepare to generate the last 5 ADC external

;serial clock pulses

;when going to restart the scan procedure

;at the end of the hardware update function

;This will trigger a new ADC conversion.

;

;This is the hardware update function

;entry point for initial operation.

;The INDX variable has been initialized

;before to 0x2F which will generate

;32 serial clock pulses to the ADC thus

;starting a new conversion

;

AN86-35

Application Note 86

iupdt clrw tris DBUS bcf ADDAC movf U1,0 movwf DBUS bcf NCSD1 bsf NCSD1 movf DL0,0 movwf DBUS bcf NCSD0 bsf bsf

NCSD0

ADDAC movf U2,0 movwf DBUS bcf bsf

NCSD1

NCSD1 movlw 0x80 xorwf DL1,0 movwf DBUS bcf NCSD0 bsf bcf bsf

NCSD0

NLDAC

NLDAC btfsc CON,5 bsf DACRDY btfss CON,5 bcf DACRDY clrwdt goto rddmy

END

;set the data bus in write mode

;

;set DAC address for LSB

;load High_DAC LSB

;

;write to High_DAC

;

;load Low_DAC LSB

;

;write to Low_DAC

;

;set DAC address for MSB

;load High_DAC MSB

;

;write to High_DAC

;

;change DL1 to offset binary

;load Low_DAC MSB

;

;write to Low_DAC

;

;load Low_DAC and High_DAC

;

;DACRDY output update

;test if RDY2 flag is set

;if Yes, set DACRDY output

;if No

;and only if No, clear DACRDY output

;

;clear watch dog timer

;

;generate the necessary number

;of ADC serial clock pulses in order

;to start a new conversion

; directive ‘end of program’

AN86-36

Application Note 86

APPENDIX E

LINEARITY AND OUTPUT RANGE OPTIONS

The LTC2400 used as the feedback A-to-D element in the

DAC has a typical

±

2ppm residual nonlinearity. Figure E1’s lower curve shows this, along with the first order correction necessary (upper curve) to get nonlinearity inside

1ppm (center curve). If true 1ppm performance is necessary, the software based correction described in Appendix␣ D can be utilized. The software generates the desired “inverted bowl” correction characteristic. The correction may be set to complement the residual nonlinearity characteristics of any individual LTC2400 via DIP switches at the code comparator.

The LTC2410 offers another approach to improved linearity. This LTC2400 variant has improved linearity but specifies a maximum 2.5V input range. Figure E2 divides the

DAC output with a precision resistor ratio set, allowing

LTC2410 use while maintaining the 5V full-scale output.

The disadvantage of this approach is the ratio set’s additional 0.1ppm/

°

C and 5ppm/year error contribution.

1

Figure E3 is similar, although the ratio set’s new value permits a 10V full-scale output.

0

–2

4

2

10

8

6

–4

–6

–8

–10

0

CORRECTION

CHARACTERISITIC

IDEAL CORRECTED ERROR

INHERENT

ERROR

8,338,608

OUTPUT CODE (DECIMAL)

16,777,215

AN86 FE01

Figure E1. LTC2400 A-to-D Inherent Residual Linearity Error (Lower),

Correction Characteristic (Upper) and Resultant Corrected Linearity

(Center). Correction Ensures <1ppm Nonlinearity

FROM

DAC OUTPUT

AMPLIFIER

*10ppm RATIO SET

VISHAY VHD-200

0.1ppm/

°

C

5k

*

5k

IN

(2.5V

MAX)

LTC2410

(PARTIAL)

OUTPUT

0V TO 5V

DIGITAL OUTPUT

TO CODE COMPARATOR

AN86 E02

Figure E2. Precision Resistor Ratio Set Divides DAC Output,

Permitting Higher Inherent Linearity LTC2410 Utilization.

Disadvantage Is 5ppm/Yr and 0.1ppm/

°

C Additional Drift Terms

FROM

DAC OUTPUT

AMPLIFIER

*10ppm RATIO SET

VISHAY VHD-200

0.1ppm/

°

C

7.5k

*

2.5k

IN

(2.5V

MAX)

LTC2410

(PARTIAL)

OUTPUT

0V TO 10V

DIGITAL OUTPUT

TO CODE COMPARATOR

AN86 E03

Figure E3. Similar to Figure E5, Except 3:1 Ratio Set Permits

10V Output While Accomodating LTC2410’s 2.5V Input

Note 1: The strata is becoming rarified when “error contribution” is delineated in fractional parts-per-million and the yearly drift rate noted.

AN86-37

Application Note 86

APPENDIX F

OUTPUT STAGES

Some applications may require outputs other than the text circuit’s 0V to 5V range. The simplest variation is a bipolar output, shown in Figure F1. The circuit, a summing inverter, subtracts the DAC output from a reference to obtain a bipolar output. Resistor and reference values may be varied to obtain different output excursions. The LT1010 output buffer provides drive capability and the chopper stabilized amplifier maintains 0.05

µ

V/

°

C stability. The resistors introduce a 0.3ppm/

°

C error contribution

1

Figure F2 yields voltage gain by dividing the DAC output prior to its application to the feedback A-to-D. In this case, the 1:1 divider ratio sets a 10V output, assuming an A-to-D reference of 5V. As in Figure F1, the resistors add a slight temperature error, about 0.1ppm/

°

C for the ratio set specified.

2

Figure F3 uses active devices for voltage outputs as high as

±

100V. The discrete high voltage stage is driven in closed-loop fashion by a chopper stabilized amplifier. Q1 and Q2 furnish voltage gain, and feed the Q3-Q4 emitter follower outputs. Q5 and Q6 set current limit at 25mA by diverting output drive when voltages across the 27

Ω shunts become too high. The local 1M-50k feedback pairs set stage gain at 20, allowing LTC1152 drives to cause full

±

120V output swing. The local feedback reduces stage gain-bandwidth, making dynamic control easier. This stage is relatively simple to frequency compensate because only

Q1 and Q2 contribute voltage gain. Additionally, the high voltage transistors have large junctions, resulting in low f t s, and no special high frequency roll-off precautions are needed. Because the stage inverts, feedback is returned to the amplifier’s positive input. Frequency compensation is achieved by rolling off the amplifier with the local 0.005

µ

F-

10k pair.

Heating and voltage coefficient errors are minimized in the feedback term by using four individual resistors. Trimming involves selecting the indicated resistor for exactly

100.0000V output with the DAC at full scale.

Figure F4 increases output current capability with a current gain stage inside the DAC output amplifier’s feedback loop. This stage replaces the LT1010 150mA buffer shown in the text. The figure shows two options, differing in output capacity. It is worth noting that as output current rises, wiring resistance becomes a large potential error term. For example, at only 10mA output, 0.001

Ω

of wiring resistance introduces 10

µ

V drop—a 2ppm error. Because of this, heavy loads should be supplied via short, highly conductive paths and remote sensing employed.

0V TO 5V

FROM DAC

OUTPUT

AMPLIFIER

10k*

20k*

–5V

REFERENCE

–

+

LTC1152

10k*

LT1010

OUTPUT

–2.5V TO 2.5V

*= VISHAY TYPE VHP-100 MATCHED SET

Figure F1. Precision Resistors and Chopper Stabilized

Output Amplifier Allow Bipolar DAC Output. Trade-Off Is

≈

0.3ppm/

°

C Additional Resistor Based Error

AN86 FF01

FROM

DAC OUTPUT

AMPLIFIER

OUTPUT

0V TO 10V

5k

*VISHAY VHD-200

RATIO SET

0.1ppm/

°

C

*

5k

TO LTC2400/LTC2410

A-TO-D INPUT

USE 7.5k-2.5k VALUES

FOR LTC2410

AN86 F02

Figure F2.

×

2 Voltage Gain Obtained By

Feedback Division at A-to-D. Slight Increase in

Overall Temperature Coefficient Results

Note 1: See Note 1 in Appendix E.

Note 2: See above footnote.

AN86-38

Application Note 86

1

µ

F

125V

510

Ω

330

Ω

Q1

2N5415

0.005

µ

F

15V

10k

INPUT FROM DAC

R

SELECT

***

TYPICAL

6

Ω 19.994k**

–

+

LTC1152

– 15V

50k 1M

50k 1M

Q5

2N2222

1N4148

1N4148

Q6

2N2907

1k

Q3

2N3440*

27

Ω

1k

27

Ω

Q3

2N5415*

* HEAT SINK

** VISHAY VHP-100 RESISTOR 0.01%

*** 1% METAL FILM RESISTOR

510

Ω

Q2

2N3440

330

Ω

1

µ

F

– 125V

100k** 100k** 100k** 100k**

Figure F3. High Voltage Output Stage Delivers

±

100V at 25mA. Multiple

Feedback Resistors Minimize Dissipation and Voltage Coefficient Effects

OUTPUT

AN86 FF03

FROM TEXT FIGURE 2

(PARTIAL)

2k

0.1

µ

F

0.1

µ

F

(OPTIONAL—

FOR INCREASED

LOAD STABILITY)

–

FROM

MSB

AND LSB

DACs

C

C

–

+

LT1001

+

LT1206: 250mA OUTPUT

LT1210: 1.1A OUTPUT

AN86 FF04

Figure F4. LT1206/LT1210 Output Stages Supply 250mA and 1.1A Loads,

Respectively. Remote Sensing Is Usually Necessary to Compensate IR Drops

AN86-39

Application Note 86

APPENDIX G

MEASURING DAC SETTLING TIME

Measuring the 20-bit DAC’s output settling time is a challenging task. Although the time scale involved is relatively slow, the 5

µ

V LSB step size presents problems.

The issue reduces to obtaining a great deal of gain without inducing overdrive in the monitoring oscilloscope. Such overdrive will corrupt the measurement, rendering displayed results meaningless.

Figure G1 is a solution. The DAC output is resistively balanced against a precision variable reference supply, adjustable in 0.5

µ

V steps.

1

The circuit’s remainder constitutes a clamped, distributed gain of 2000 amplifier. Diode clamping, placed at each gain stage input, prevents saturation from occurring even with large DAC-reference supply imbalances. The distributed gain allows 10kHz bandwidth while maintaining clamping effectiveness. The monitoring oscilloscope, operating at 5mV or 10mV/DIV

(5

µ

V to 10

µ

V at the DAC output), can readily discern 5

µ

V settling without incurring deleterious overdrive.

Layout and construction of this circuit requires care.

Figure G2 shows construction details. A linear layout minimizes parasitic feedback paths, preventing oscillation. The DAC input step is fully shielded, preventing feedthrough to various sensitive points within the amplifier. Finally, the entire circuit is built into a shielded enclosure to minimize effects of stray RF and pick up.

The circuit is tested by applying a test step that settles much faster than the DAC. Figure G3 uses a mercury wetted reed relay based pulse generator to supply the step.

The unit noted is commercially produced, although similar results are obtainable with standard mercury based reed relays. When the relay opens the circuit’s output settles essentially instantaneously (Figure G4) relative to DAC speed and settling time amplifier bandwidth.

Figure G1’s response is tested by grounding one of its inputs and driving the other with the pulse generator.

Figure G5 shows settling to within 1ppm (

±

5

µ

V) in 2ms.

This is much faster than the DAC settles, lending confidence to text Figures 6 and 7 indicated results.

0V TO –5V

IN 0.5

µ

V STEPS

FROM BUFFERED

KVD BIASED FROM

–5V REFERENCE

5pF

20k*

20k*

+

–

LT1008

5.1k

FROM

DAC

OUTPUT

0V TO 5V

9k

1k

* VISHAY VHD-200 RATIO SET 10ppm MATCHING

1N4148

1N5712

15V

CONNECT OUTPUT DIRECTLY TO

OSCILLOSCOPE. DO NOT USE CABLE

TO SET ZERO, GROUND BOTH INPUTS

AND ADJUST “ZERO” FOR V

OUT

= <1mV

909k

10k

ZERO

10

Ω

–15V

+

–

LT1008

5pF

5.1k

9k

1k

+

–

LT1008

5pF

5.1k

9k

1k

Figure G1. Clamped, Distributed Gain-of-2000 Amplifier Permits

DAC Settling Time Measurement Without Saturation Effects

+

–

LT1008

20pF

5.1k

5.1k

5.1k

OUTPUT

AN86 FG01

Note 1: See Appendix C for details on such a supply.

AN86-40

Application Note 86

AN86 FG02.tif

Figure G2. Settling Time Amplifier Construction. Bandwidth Is Only 10kHz, Although Gain of 2000 Necessitates Layout Care to Avoid

Parasitic Feedback Induced Oscillation. Input (Photo Lower Left) Is Fully Shielded, Preventing Radiative Feedthrough to Amplifier.

Enclosure Shields Circuit from Stray RF and Pickup

AN86-41

Application Note 86

TEKTRONIX

067-0608-00

FROM

PULSE

GENERATOR

PULSE

IN

+V

Hg REED RELAY

50

Ω

25ms

OUTPUT

TO DAC INPUT OF

SETTLE CIRCUIT

30ms

(GROUND SETTLE CIRCUIT

REFERENCE INPUT)

5V

AN86 FG03

Figure G3. Reed Relay Based Pulser Supplies Clean Step to Test Settling Time Circuit

AN86 FG03.tif

Figure G4. Mercury Wetted Reed Relay Opens in 5 Nanoseconds, Settles Quickly to Zero.

500MHz Ring-Off Derives from Source-Termination Impedance Mismatch

AN86-42

10

µ

V/DIV

HORIZ = 2ms/DIV

AN86 FE03.tif

Figure G5. Settling Time Circuit Responds to Test Step with 2ms Settling to

±

1ppm (

±

5

µ

V)

Application Note 86

APPENDIX H

MICROVOLT LEVEL NOISE MEASUREMENT

Verifying DAC output noise requires a quiet, high gain amplifier at the oscilloscope. Figure H1 shows one way to take the measurement. The input preamplifier, operating at a gain of 1000, supplies a high pass cutoff at 0.1Hz. It drives the oscilloscope via a 10Hz discrete low pass filter.

The oscilloscope, set to 1mV/DIV, indicates 1

µ

V/DIV referred to the preamplifier input. Figure H2 indicates DAC output noise well below an LSB, about 0.9

µ

V. Equipment limitations set measurement noise floor at 0.2

µ

V.

Figure H3 shows the noise measurement test setup. Note that the signal levels involved dictate a completely shielded, coaxial path from breadboard to oscilloscope.

Figure H4 lists some applicable high sensitivity amplifiers suitable for the noise measurement. Current generation oscilloscopes rarely have greater than 2mV/DIV sensitivity, although older instruments offer more capability. The figure lists representative preamplifiers and oscilloscope plug-ins suitable for noise measurement. These units feature wideband, low noise performance. It is particularly significant that many of these instruments are no longer produced. This is in keeping with current instrumentation trends, which emphasize digital signal acquisition as opposed to analog measurement capability.

The monitoring oscilloscope should have exceptional trace clarity. In the latter regard high quality analog oscilloscopes are unmatched. The exceptionally small spot size of these instruments is well-suited to low level noise measurement.

1

The digitizing uncertainties and raster scan limitations of DSOs impose display resolution penalties. Many DSO displays will not even register the fine structure of the noise waveform.

FROM

DAC

OUTPUT

PREAMPLIFIER

A = 1000

HIGH PASS

CUTOFF = 0.1Hz

1.6k

(10Hz)

+

10

µ

F

OSCILLOSCOPE

1mv/DIV = 1

µ

V/DIV

AN86 FH01

Figure H1. Microvolt Noise Measurement Necessitates High

Gain Preamplifier for Oscilloscope. Preamplifier and Discrete

Filter Set 0.1Hz to 10Hz Measurement Bandpass

2s/DIV

AN86 F04.tif

Figure H2. Indicated DAC Output Noise in a 0.1Hz to 10Hz

Bandpass Is Below 1

µ

V, About 0.2LSB. Equipment

Limitations Set Measurement Noise Floor at 0.2

µ

V

Note 1: In our work we have found Tektronix types 453, 453A, 454,

454A, 547 and 556 excellent choices. Their pristine trace presentation is ideal for discerning small signals of interest against a noise floor limited background.

AN86-43

Application Note 86

AN86 FH03.tif

Figure H3. Noise Measurement Test Setup Includes Shielded DAC Breadboard (Foreground), Preamplifier

(Left) and Low Pass Filter Attached to Oscilloscope (Center). Measurement Path Is Fully Coaxial

AN86-44

Application Note 86

INSTRUMENT

TYPE

Differential Amplifier

MODEL MAXIMUM

MANUFACTURER NUMBER BANDWIDTH SENSITIVITY/GAIN AVAILABILITY

Tektronix 1A7/1A7A 500kHz/1MHz 10

µ

V/DIV Secondary Market

COMMENTS

Requires 500 Series Mainframe,

Settable Bandstops

Differential Amplifier Tektronix 7A22 1MHz 10

µ

V/DIV Secondary Market Requires 7000 Series Mainframe,

Settable Bandstops

Differential Amplifier Tektronix 5A22 1MHz 10

µ

V/DIV Secondary Market Requires 5000 Series Mainframe,

Settable Bandstops

Differential Amplifier Tektronix ADA-400A 1MHz 10

µ

V/DIV Current Production Standalone with Optional Power

Supply, Settable Bandstops

Differential Amplifier Tektronix AM-502 1MHz 100,000 Secondary Market Standalone with Optional Power

Supply, Settable Bandstops

Differential Amplifier Preamble 1822

Differential Amplifier Stanford Research SR-560

Systems

10MHz

1MHz

Gain = 1000

Gain = 50000

Current Production Standalone, Settable Bandstops

Current Production Standalone, Settable Bandstops,

Battery or Line Operation

Figure H4. Some Applicable High Sensitivity, Low Noise Amplifiers.

Trade-Offs Include Compatibility, Sensitivity and Availability

APPENDIX I

VOLTAGE REFERENCES

Figure I1 lists some voltage reference options for use with the DAC. The self-contained types are convenient and easily applied. The LM199A and the LTZ1000A require external circuitry but offer higher performance. All choices must be trimmed to establish absolute DAC accuracy. The

LTZ1000A offers the highest stability and is discussed below.

Figure I2 shows the LTZ1000A and its support circuitry.

A1 senses LTZ1000A die temperature and accordingly controls the IC heater via the 2N3904. A2 controls reference current. The Zener reference is sensed via Kelvin connections, minimizing voltage drop effects. A single point ground eliminates return current mixing and the attendant errors that would be produced.

TYPE

LTZ1000A

LM199A

LT1021

LT1027

LT1236

Figure I3 offers choices for reference buffering. All employ a chopper stabilized amplifier augmented with a buffer output stage. Buffer error is extremely low, as noted in

Appendix C’s discussion. I3a, a simple unity-gain stage, transmits the input to the output with low error and minimal reference loading. I3b takes moderate gain, allowing a 7V reference input to produce (in this case) 10V at the output. I3c offers two ways to get 5V from the nominal 7V input. A precision divider lightly loads

the reference in one case; the 5V output is taken at the

LT1010. Reference loading is avoided by placing the divider at the output (optional case shown) and driving the

A-to-D reference input from the divider output, which is permissible.

VOLTAGE

7.2V

6.95V

INITIAL

ACCURACY

Minimum 7V

Maximum 7.5V

2%

5V, 7V, 10V 0.05V (7V)

5V 0.02%

5V, 10V 0.05%

TEMPERATURE

DRIFT

0.05ppm/

°

C

0.5ppm/

°

C

2ppm/

°

C (7V)

2ppm/

°

C

5ppm/

°

C

LONG-TERM

STABILITY

4ppm/Yr Typical

COMMENTS

Highest Stability Zener Available. Requires

External Heater Control and Reference Buffer Circuitry

10ppm/Yr Typical Self-Contained, Including Heater Control Circuitry.

Zener Output Is Unbuffered

20ppm/kHr Noncumulative Fully Self-Contained. Trimmable

20ppm/kHr Noncumulative Fully Self-Contained. Trimmable

20ppm/kHr Noncumulative Fully Self-Contained. Trimmable

Figure I1. Reference Choices Compared for Output Voltage, Accuracy and Stability. Highest Stability Types Require External Circuitry

AN86-45

Application Note 86

ZENER + SENSE

V

+

15V

R4

13k

R3

70k

3

R2

70k

1N4148

2N3904

1k

A1

LT1013

–

+

1M

10k

6

8

TEMP

SENSOR

7

4

5

7

+

–

A2

LT1013

R1 THROUGH R5:

VISHAY VHP-100 0.1%

0.1

µ

F ZENER – SENSE

HEATER,

UNLABELED

AND

ZENER = LTZ1000A

PIN NUMBERS APPEAR

FOR INDIVIDUAL

LTZ1000A ELEMENTS

1

HEATER

0.1

µ

F

R5

1k

R1

120

Ω

0.022

µ

F

ZENER – FORCE

1N4148

2

HEATER RETURN

Figure I2. 7V Reference Includes A1 Heater Control Amplifier, A2 Zener Current Regulator and LTZ1000A Zener. Note Zener Kelvin Connections and Single Point Ground

SINGLE

POINT

GROUND

AN86 FI02

TO ZENER

SENSE

10k

–

+

LTC1150

0.1

µ

F

LT1010 OUT = 7.2V NOMINAL

5k**

TO ZENER

SENSE

12.9k**

–

+

LTC1150

0.1

µ

F

LT1010

OUT = 10V NOMINAL

AN86-46

10k

–

+

LTC1150

0.1

µ

F

TO ZENER

SENSE

22k*

50k*

LT1010

OUT = 5V NOMINAL

2.2k**

OPTIONAL, DELETE INPUT DIVIDER.

SEE TEXT

OUT = 5V NOMINAL

5k**

*

**

VISHAY VHP-100 – 1%

VISHAY VHD-200 RATIO SET – 0.1%

Figure I3. Chopper Stabilized Reference Buffer Options Include Unity Gain (a),

10V (b) and 5V (c) Output. Trimming Is Required for Absolute Accuracy

AN86 I03

APPENDIX J

CABLES, CONNECTIONS, SOLDER, COMPONENT

CHOICE, TERROR AND ARCANA

Subtle parasitic effects can have pronounced and seemingly inexplicable effects on low level circuit performance.

Perhaps the most prevalent detractor to microvolt level circuitry is unintended thermocouples. Considerable discussion for dealing with thermocouples appeared in Appendix C and should be considered preliminary to this section’s material.

In 1822, Thomas Seebeck, an Estonian physician, accidentally joined semicircular pieces of bismuth and copper

(Figure J1) while studying thermal effects on galvanic arrangements. A nearby compass indicated a magnetic disturbance. Seebeck experimented repeatedly with different metal combinations at various temperatures, noting relative magnetic field strengths. Curiously, he did not believe that electric current was flowing and preferred to describe the effect as “thermomagnetism.” He published his results in a paper, “Magnetische Polarisation der

Metalle und Erze durch Temperatur-Differenz” (see References).

JUNCTION

COPPER BISMUTH

Application Note 86

In low drift circuits, unwanted thermocouples are probably the primary source of error. Connectors, switches, relay contacts, sockets, wire and even solder are all candidates for thermal EMF generation. It is relatively clear that connectors and sockets can form thermal junctions.

However, it is not at all obvious that junctions of wire from different manufacturers can easily generate 200nV/

°

C— four times a precision amplifier’s drift specification! Figure J2 shows a plot obtained for such a wire junction. Even solder can become an error term at low levels, creating a junction with copper or Kovar wires or PC traces (see

Figure J3). Figure J4 lists thermocouple potentials for some common materials found in electronic assemblies.

1.2

1.0

0.8

0.6

0.4

0.2

0

25

2.2

2.0

1.8

1.6

1.4

3.0

2.8

2.6

2.4

30 35 40

DEGREES CENTIGRADE

45

AN86 FJ02

Figure J2. Thermal EMF Generated by Two “Identical”

Copper Wires Due to Oxidation and Impurities

W

N

E

JUNCTION

S

COMPASS

AN86 FJ01

Figure J1. The Arrangement for Dr. Seebeck’s

Accidental Discovery of “Thermomagnetism”

100

SLOPE

≈

1.5

µ

V/

°

C BELOW 25

°

C

50

0

64% Sn/36% Pb

60% Cd/40% Sn

SLOPE

≈

160nV/

°

C BELOW 25

°

C

–50

Subsequent investigation has shown the “Seebeck Effect” to be fundamentally electrical in nature, repeatable and quite useful. Thermocouples, by far the most common transducer, are Seebeck’s descendants. Unfortunately, unintended and unwanted thermocouples are also

Seebeck’s progeny.

–100

0 10 20 30 40 50

SOLDER-COPPER JUNCTION DIFFERENTIAL TERMPERATURE

SOURCE: NEW ELECTRONICS 2/6/77

AN86 FJ03

Figure J3. Solder-Copper Thermal EMFs. Cd/Sn Has Notably

Lower Activity but Is Toxic, Not Available and Not Recommended

AN86-47

Application Note 86

The unusually energetic response of Cu-CuO necessitated the treatment described in Appendix C (Figure C7 and associated text) for cleaning DVM and Kelvin-Varley divider connections. Readers finding this figure’s information seemingly academic should be awakened by Figure

J5. This chart lists thermoelectric potentials for commonly employed laboratory connectors. Thermocouple activity of some types is more than 20 times greater than others. Be careful!

Minimizing thermal EMF induced errors is possible if judicious attention is given to circuit board layout. In general, it is good practice to limit the number of junctions in the signal path. Avoid connectors, sockets, switches and other potential error sources to the extent possible. In some cases this will not be possible. In these instances, attempt to balance the number and type of junctions in the signal path so that differential cancellation occurs. Doing this may involve deliberately creating and introducing junctions to offset unavoidable junctions. This can be a tricky procedure. Repeated deliberate temperature excursions may be necessary to determine the optimal number and placement of added junctions. Experimentation, tempered by a healthy reserve of patience and abundance of time, is required. This practice, borrowed from standards lab procedures, can be quite effective in reducing thermal

EMF originated drifts. Figure J6 shows a simple example where a nominally unnecessary resistor is included to promote such thermal balancing. For remote signal sources connectors may be unavoidable. In these cases, choose a connector specified for relatively low thermal EMF activity and ensure a similarly balanced approach in routing signals through the connector along the circuit board and to circuitry. If some imbalance is unavoidable, deliberately introduce an intentional counterbalancing junction. In all cases maintain the differencing junctions in close physical proximity, which will keep them at the same temperature.

Avoid drafts and temperature gradients, which can introduce thermal imbalances and cause problems. Figure J7 shows the LTC1150 set up in a test circuit to measure its temperature stability. The lead lengths of the resistors connected to the amplifier’s inputs are identical. The thermal capacity each input sees is also balanced because of the symmetrical connection of the resistors and their identical size. Thus, thermal EMF induced shifts are equal in phase and amplitude and cancellation occurs. Very slight air currents can still affect even this arrangement.

Figure J8 shows a strip chart of output noise with the circuit covered by a small styrofoam cup (HANDI-KUP

TM

Company Model H8-S) and with no cover in “still” air. This data illustrates why it is often prudent to enclose low level circuitry inside some form of thermal baffle.

MATERIALS

Cu-Cu

Cu-Ag

Cu-Au

Cu-Cd/Sn

Cu-Pb/Sn

Cu-Kovar

Cu-Si

Cu-CuO

POTENTIAL (

< 0.2

0.3

0.3

0.3

1 to 3

40

400

1000

µ

V/

Source: Low Level Measurements,

Keithley Instruments, 1984

(see References)

°

C)

Figure J4. Thermoelectric Potentials for Various Materials

Indicates Inadvisability of Mixing Materials in Signal Path.

Cu-Cu Connections (Chart Top) Must Be Kept Clean or

5000:1 Degradation Occurs As They Oxidize (Chart Bottom)

HANDI-KUP is a trademark of WinCup.

AN86-48

Application Note 86

CONNECTION TYPE

BNC-BNC Mate

BNC-Banana Adapter

BNC-BNC “Barrel” Adapter

Male/Female Banana Mate Sample #1

Male/Female Banana Mate Sample #2

Male/Female Banana Mate (Type Specified for Low Thermal Activity) Sample #3

Copper Lug-Copper Banana Binding Post

Copper Lug-Standard Banana Binding Post

Plated Lug-Copper Banana Binding Post

DESCRIPTION

Figure J5. Measured Thermoelectric Potentials for Some Common Laboratory Connectors.

Pronounced Difference Between “Banana” Samples Is Due to Manufacturer’s Materials Choice. Note

That Copper Lug/Copper Banana Post Has 20

×

Lower Activity Than Plated Lug/Copper Banana Post

THERMOELECTRIC

POTENTIAL (

µ

V/

°

C)

0.4

0.35

0.5

1.7

0.4

0.35

1.1

0.07

0.08

AN86-49

Application Note 86

Thermal EMFs are the most likely, but not the only, potential low level error source. Electrostatic and electromagnetic shielding may be required. Power supply transformer fields are notorious sources of errors often mistakenly attributed to amplifier DC drift and noise. A transformer’s magnetic field impinging on a PC trace can easily generate microvolts across that conductor in accordance with well known magnetic theory. The circuit cannot distinguish between this spurious signal and the desired input. Attempts to eliminate the problem by rolling off circuit response may work, but often the filtered version of the undesired pickup masquerades as an unstable DC term. The most direct approach is to use shielded transformers but careful layout may be equally effective and less costly. A circuit that requires the transformer to be close by to achieve a good quality grounding scheme may be disturbed by the transformer’s magnetic field. An RF choke connected across a scope probe can determine the presence and relative intensity of transformer fields, aiding layout experimentation.

Another source of parasitic error is stray leakage current.

Such leakage currents must be prevented from influencing circuit operation. The simplest way to do this is to connect leakage sensitive points via teflon standoffs.

Because the points never contact the PC board, stray leakage currents do not affect them. Although this approach is effective, its implementation may not be acceptable in production. Guarding is another technique for minimizing board leakage effects. The guard is a PC trace completely encircling the leakage sensitive points. This trace is driven at a potential equal to that of the point, preventing leakage to the “guarded” point. On PC boards, the guard should enclose the node(s) to be protected.

Guarding was used to eliminate the effects of capacitor surface leakage in Appendix C’s Figure C7.

NOMINALLY UNNECESSARY

RESISTOR USED TO

THERMALLY BALANCE OTHER

INPUT RESISTOR

DELIBERATE SPLICE MAY BE DESIRABLE

TO BALANCE OTHER JUNCTIONS

LEAD WIRE/SOLDER/COPPER

TRACE JUNCTION

+

OUTPUT

–

RESISTOR LEAD, SOLDER,

COPPER TRACE JUNCTION

AN86-50

AN86 FJ06

Figure J6. Typical Thermal Layout Considerations Emphasize Minimizing and Differencing

Parasitic Thermocouples. Thermal Mass at Amplifier Inputs Should Be Equal, Allowing

Differenced Parasitic Thermocouple Outputs to Arrive Matched In Phase and Amplitude

50k

100

Ω

–

+

LTC1150

E

OS

×

1000

50k

AN86 FJ06

Figure J7. Amplifier Drift Test Circuit. Thermal EMFs and Thermal

Capacity at Each Input Must Be Similar for Cancellation to Occur

Application Note 86

AN86 FJ08.tif

Figure J8. Effect of Thermal Baffle on Low Frequency Amplifier Noise in “Still” Air. Amplifier Is Covered

By Small Cup in Upper Trace, Uncovered in Lower Trace. Instability Worsens If Air Movement Increases

AN86-51

Application Note 86

AN86-52

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

●

FAX: (408) 434-0507

●

www.linear-tech.com

an86f LT/TP 0101 4K • PRINTED IN USA



LINEAR TECHNOLOGY CORPORATION 2001