Keysight Technologies Impedance Measurement Handbook A guide to measurement technology and techniques

Keysight Technologies Impedance Measurement Handbook A guide to measurement technology and techniques
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Keysight Technologies
Impedance Measurement Handbook
A guide to measurement technology and techniques
5th Edition
Application Note
Distributed by:
dataTec ▪ Ferdinand-Lassalle-Str. 52 ▪ 72770 Reutlingen ▪ Tel. 07121 / 51 50 50 ▪ Fax 07121 / 51 50 10 ▪ info@datatec.de ▪ www.datatec.de
Introduction
In this document, not only currently available products but also discontinued and/or obsolete products will be shown as
reference solutions to leverage Keysight's impedance measurement expertise for speciic application requirements. For
whatever application or industry you work in, Keysight offers excellent performance and high reliability to give you conidence when making impedance measurements. The table below shows product status of instruments, accessories, and
ixtures listed in this document. Please note that the status is subject to change without notice.
Product status
Available
Discontinued
(� replacement product)
Obsolete
(� replacement product)
Instrument
4285A
E4980A
E4980AL
E4981A
E4982A
E4990A
E4991B
E5061B-3L5
E5071C
4263B
4287A (� E4982A)
4294A (� E4990A)
4338B
E4991A (� E4991B)
4268A (� E4981A)
4284A (� E4980A/AL)
4288A (� E4981A)
4395A (� E5061B-3L5/005)
Accessories, ixtures
16034E/G/H
16044A
16047A/E
16048A/D/E/G/H
16060A
16065A/C
16089A/B/C/D
16196A/B/C/D
16200B
16334A
16451B
16452A
42941A
42942A
16047D
16089E
16316A
16317A
42841A
42842A/B/C
43961A
Table of Contents
1.0
Impedance Measurement Basics
1.1 Impedance ........................................................................................................................ 1-01
1.2 Measuring impedance ..................................................................................................... 1-03
1.3 Parasitics: There are no pure R, C, and L components .................................................. 1-03
1.4 Ideal, real, and measured values ..................................................................................... 1-04
1.5 Component dependency factors ..................................................................................... 1-05
1.5.1 Frequency .............................................................................................................. 1-05
1.5.2 Test signal level ..................................................................................................... 1-07
1.5.3 DC bias................................................................................................................... 1-07
1.5.4 Temperature........................................................................................................... 1-08
1.5.5 Other dependency factors....................................................................................... 1-08
1.6 Equivalent circuit models of components........................................................................ 1-08
1.7 Measurement circuit modes ............................................................................................. 1-10
1.8 Three-element equivalent circuit and sophisticated component models ...................... 1-13
1.9 Reactance chart................................................................................................................ 1-15
2.0
Impedance Measurement Instruments
2.1 Measurement methods .................................................................................................... 2-01
2.2 Operating theory of practical instruments ...................................................................... 2-04
LF impedance measurement
2.3 Theory of auto balancing bridge method ....................................................................... 2-04
2.3.1 Signal source section ............................................................................................ 2-06
2.3.2 Auto-balancing bridge section ............................................................................. 2-07
2.3.3 Vector ratio detector section................................................................................. 2-08
2.4 Key measurement functions ............................................................................................ 2-09
2.4.1 Oscillator (OSC) level ............................................................................................ 2-09
2.4.2 DC bias .................................................................................................................. 2-10
2.4.3 Ranging function .................................................................................................. 2-11
2.4.4 Level monitor function .......................................................................................... 2-12
2.4.5 Measurement time and averaging ....................................................................... 2-12
2.4.6 Compensation function ........................................................................................ 2-13
2.4.7 Guarding ............................................................................................................... 2-14
2.4.8 Grounded device measurement capability .......................................................... 2-15
RF impedance measurement
2.5 Theory of RF I-V measurement method ......................................................................... 2-16
2.6 Difference between RF I-V and network analysis measurement methods ...................... 2-17
2.7 Key measurement functions ............................................................................................ 2-19
2.7.1 OSC level ............................................................................................................... 2-19
2.7.2 Test port ................................................................................................................ 2-19
2.7.3 Calibration ............................................................................................................ 2-20
2.7.4 Compensation ....................................................................................................... 2-20
2.7.5 Measurement range ............................................................................................. 2-20
2.7.6 DC bias .................................................................................................................. 2-20
3.0
Fixturing and Cabling
LF impedance measurement
3.1 Terminal configuration ..................................................................................................... 3-01
3.1.1 Two-terminal configuration ................................................................................... 3-02
3.1.2 Three-terminal configuration ................................................................................ 3-02
3.1.3 Four-terminal configuration .................................................................................. 3-04
3.1.4 Five-terminal configuration ................................................................................... 3-05
3.1.5 Four-terminal pair configuration ........................................................................... 3-06
3.2 Test fixtures ...................................................................................................................... 3-07
3.2.1 Keysight-supplied test fixtures.............................................................................. 3-07
3.2.2 User-fabricated test fixtures.................................................................................. 3-08
3.2.3 User test fixture example....................................................................................... 3-09
3.3 Test cables ....................................................................................................................... 3-10
3.3.1 Keysight supplied test cables ............................................................................... 3-10
3.3.2 User fabricated test cables .................................................................................. 3-11
3.3.3 Test cable extension ............................................................................................. 3-11
3.4 Practical guarding techniques ........................................................................................ 3-15
3.4.1 Measurement error due to stray capacitances ..................................................... 3-15
3.4.2 Guarding techniques to remove stray capacitances ............................................ 3-16
RF impedance measurement
3.5 Terminal configuration in RF region ................................................................................ 3-16
3.6 RF test fixtures ................................................................................................................. 3-17
3.6.1 Keysight-supplied test fixtures ............................................................................. 3-18
3.7 Test port extension in RF region....................................................................................... 3-19
4.0 Measurement Error and Compensation
Basic concepts and LF impedance measurement
4.1 Measurement error .......................................................................................................... 4-01
4.2 Calibration ........................................................................................................................ 4-01
4.3 Compensation .................................................................................................................. 4-03
4.3.1 Offset compensation ............................................................................................ 4-03
4.3.2 Open and short compensations ........................................................................... 4-04
4.3.3 Open/short/load compensation .......................................................................... 4-06
4.3.4 What should be used as the load? ...................................................................... 4-07
4.3.5 Application limit for open, short, and load compensations ................................ 4-09
4.4 Measurement error caused by contact resistance ......................................................... 4-09
4.5 Measurement error induced by cable extension ............................................................ 4-11
4.5.1 Error induced by four-terminal pair (4TP) cable extension .................................. 4-11
4.5.2 Cable extension without termination .................................................................... 4-13
4.5.3 Cable extension with termination ......................................................................... 4-13
4.5.4 Error induced by shielded 2T or shielded 4T cable extension ............................. 4-13
4.6 Practical compensation examples .................................................................................. 4-14
4.6.1 Keysight test fixture (direct attachment type) ...................................................... 4-14
4.6.2 Keysight test cables and Keysight test fixture ...................................................... 4-14
4.6.3 Keysight test cables and user-fabricated test fixture (or scanner) ...................... 4-14
4.6.4 Non-Keysight test cable and user-fabricated test fixture .................................... 4-14
RF impedance measurement
4.7
Calibration and compensation in RF region .................................................................. 4-16
4.7.1 Calibration ........................................................................................................... 4-16
4.7.2 Error source model .............................................................................................. 4-17
4.7.3 Compensation method ....................................................................................... 4-18
4.7.4 Precautions for open and short measurements in RF region ............................ 4-18
4.7.5 Consideration for short compensation ............................................................... 4-19
4.7.6 Calibrating load device ....................................................................................... 4-20
4.7.7 Electrical length compensation .......................................................................... 4-21
4.7.8 Practical compensation technique ..................................................................... 4-22
4.8
Measurement correlation and repeatability .................................................................. 4-22
4.8.1 Variance in residual parameter value ................................................................. 4-22
4.8.2 A difference in contact condition ........................................................................ 4-23
4.8.3 A difference in open/short compensation conditions ........................................ 4-24
4.8.4 Electromagnetic coupling with a conductor near the DUT ............................... 4-24
4.8.5 Variance in environmental temperature .............................................................. 4-25
5.0 Impedance Measurement Applications and Enhancements
5.1
Capacitor measurement ................................................................................................ 5-01
5.1.1 Parasitics of a capacitor ....................................................................................... 5-02
5.1.2 Measurement techniques for high/low capacitance .......................................... 5-04
5.1.3 Causes of negative D problem ............................................................................. 5-06
5.2
Inductor measurement ................................................................................................... 5-08
5.2.1 Parasitics of an inductor ...................................................................................... 5-08
5.2.2 Causes of measurement discrepancies for inductors ......................................... 5-10
5.3
Transformer measurement ............................................................................................. 5-14
5.3.1 Primary inductance (L1) and secondary inductance (L2) ................................... 5-14
5.3.2 Inter-winding capacitance (C) ............................................................................. 5-15
5.3.3 Mutual inductance (M) ......................................................................................... 5-15
5.3.4 Turns ratio (N)....................................................................................................... 5-16
5.4
Diode measurement ....................................................................................................... 5-18
5.5
MOS FET measurement ................................................................................................. 5-19
5.6
Silicon wafer C-V measurement .................................................................................... 5-20
5.7
High-frequency impedance measurement using the probe ......................................... 5-23
5.8
Resonator measurement ................................................................................................ 5-24
5.9
Cable measurements ..................................................................................................... 5-27
5.9.1 Balanced cable measurement ............................................................................. 5-28
5.10 Balanced device measurement ..................................................................................... 5-29
5.11 Battery measurement ..................................................................................................... 5-31
5.12 Test signal voltage enhancement .................................................................................. 5-32
5.13 DC bias voltage enhancement ...................................................................................... 5-34
5.13.1 External DC voltage bias protection in 4TP configuration ................................ 5-35
5.14 DC bias current enhancement ....................................................................................... 5-36
5.14.1 External current bias circuit in 4TP configuration ............................................. 5-37
5.15 Equivalent circuit analysis function and its application ................................................ 5-38
Appendix A: The Concept of a Test Fixture’s Additional Error .......
A-01
A.1 System configuration for impedance measurement ......................................................
A.2 Measurement system accuracy........................................................................................
A.2.1 Proportional error ..................................................................................................
A.2.2 Short offset error ...................................................................................................
A.2.3 Open offset error...................................................................................................
A.3 New market trends and the additional error for test fixtures ...........................................
A.3.1 New devices ...........................................................................................................
A.3.2 DUT connection configuration ..............................................................................
A.3.3 Test fixture’s adaptability for a particular measurement ......................................
A-01
A-01
A-02
A-02
A-03
A-03
A-03
A-04
A-05
Appendix B: Open and Short Compensation .........................................
Appendix C: Open, Short, and Load Compensation .............................
Appendix D: Electrical Length Compensation .......................................
Appendix E: Q Measurement Accuracy Calculation .............................
B-01
C-01
D-01
E-01
1.0 Impedance Measurement Basics
1.1
Impedance
Impedance is an important parameter used to characterize electronic circuits, components, and the materials used
to make components. Impedance (Z) is generally defined as the total opposition a device or circuit offers to the flow
of an alternating current (AC) at a given frequency, and is represented as a complex quantity which is graphically
shown on a vector plane. An impedance vector consists of a real part (resistance, R) and an imaginary part
(reactance, X) as shown in Figure 1-1. Impedance can be expressed using the rectangular-coordinate form R + jX or
in the polar form as a magnitude and phase angle: |Z|_ θ. Figure 1-1 also shows the mathematical relationship
between R, X, |Z|, and θ. In some cases, using the reciprocal of impedance is mathematically expedient. In which
case 1/Z = 1/(R + jX) = Y = G + jB, where Y represents admittance, G conductance, and B susceptance. The unit of
impedance is the ohm (Ω), and admittance is the siemen (S). Impedance is a commonly used parameter and is
especially useful for representing a series connection of resistance and reactance, because it can be expressed
simply as a sum, R and X. For a parallel connection, it is better to use admittance (see Figure 1-2.)
Figure 1-1. Impedance (Z) consists of a real part (R) and an imaginary part (X)
Figure 1-2. Expression of series and parallel combination of real and imaginary components
Reactance takes two forms: inductive (XL) and capacitive (Xc). By definition, XL = 2πfL and Xc = 1/(2πfC), where f is
the frequency of interest, L is inductance, and C is capacitance. 2πf can be substituted for by the angular frequency
(ω: omega) to represent XL = ωL and Xc =1/(ωC). Refer to Figure 1-3.
Figure 1-3. Reactance in two forms: inductive (XL) and capacitive (Xc)
A similar reciprocal relationship applies to susceptance and admittance. Figure 1-4 shows a typical representation for
a resistance and a reactance connected in series or in parallel.
The quality factor (Q) serves as a measure of a reactance’s purity (how close it is to being a pure reactance, no resistance), and is defined as the ratio of the energy stored in a component to the energy dissipated by the component. Q
is a dimensionless unit and is expressed as Q = X/R = B/G. From Figure 1-4, you can see that Q is the tangent of the
angle θ. Q is commonly applied to inductors; for capacitors the term more often used to express purity is dissipation
factor (D). This quantity is simply the reciprocal of Q, it is the tangent of the complementary angle of θ, the angle δ
shown in Figure 1-4 (d).
Figure 1-4. Relationships between impedance and admittance parameters
1.2
Measuring impedance
To find the impedance, we need to measure at least two values because impedance is a complex quantity. Many
modern impedance measuring instruments measure the real and the imaginary parts of an impedance vector and
then convert them into the desired parameters such as |Z|, θ, |Y|, R, X, G, B, C, and L. It is only necessary to connect
the unknown component, circuit, or material to the instrument. Measurement ranges and accuracy for a variety of
impedance parameters are determined from those specified for impedance measurement.
Automated measurement instruments allow you to make a measurement by merely connecting the unknown
component, circuit, or material to the instrument. However, sometimes the instrument will display an unexpected
result (too high or too low.) One possible cause of this problem is incorrect measurement technique, or the natural
behavior of the unknown device. In this section, we will focus on the traditional passive components and discuss
their natural behavior in the real world as compared to their ideal behavior.
1.3
Parasitics: There are no pure R, C, and L components
The principal attributes of L, C, and R components are generally represented by the nominal values of capacitance,
inductance, or resistance at specified or standardized conditions. However, all circuit components are neither purely
resistive, nor purely reactive. They involve both of these impedance elements. This means that all real-world devices
have parasitics—unwanted inductance in resistors, unwanted resistance in capacitors, unwanted capacitance in
inductors, etc. Different materials and manufacturing technologies produce varying amounts of parasitics. In fact,
many parasitics reside in components, affecting both a component’s usefulness and the accuracy with which you can
determine its resistance, capacitance, or inductance. With the combination of the component’s primary element and
parasitics, a component will be like a complex circuit, if it is represented by an equivalent circuit model as shown in
Figure 1-5.
Figure 1-5. Component (capacitor) with parasitics represented by an electrical equivalent circuit
Since the parasitics affect the characteristics of components, the C, L, R, D, Q, and other inherent impedance
parameter values vary depending on the operating conditions of the components. Typical dependence on the
operating conditions is described in Section 1.5.
1.4
Ideal, real, and measured values
When you determine an impedance parameter value for a circuit component (resistor, inductor, or capacitor), it is
important to thoroughly understand what the value indicates in reality. The parasitics of the component and the
measurement error sources, such as the test fixture’s residual impedance, affect the value of impedance.
Conceptually, there are three sorts of values: ideal, real, and measured. These values are fundamental to
comprehending the impedance value obtained through measurement. In this section, we learn the concepts of ideal,
real, and measured values, as well as their significance to practical component measurements.
—
An ideal value is the value of a circuit component (resistor, inductor, or capacitor) that excludes the effects of its
parasitics. The model of an ideal component assumes a purely resistive or reactive element that has no
frequency dependence. In many cases, the ideal value can be defined by a mathematical relationship involving
the component’s physical composition (Figure 1-6 (a).) In the real world, ideal values are only of academic
interest.
—
The real value takes into consideration the effects of a component’s parasitics (Figure 1-6 (b).) The real value
represents effective impedance, which a real-world component exhibits. The real value is the algebraic sum of
the circuit component’s resistive and reactive vectors, which come from the principal element (deemed as a pure
element) and the parasitics. Since the parasitics yield a different impedance vector for a different frequency, the real
value is frequency dependent.
—
The measured value is the value obtained with, and displayed by, the measurement instrument; it reflects the
instrument’s inherent residuals and inaccuracies (Figure 1-6 (c).) Measured
values always contain errors when compared to real values. They also vary intrinsically from one measurement
to another; their differences depend on a multitude of considerations in regard to measurement uncertainties.
We can judge the quality of measurements by comparing how closely a measured value agrees with the real
value under a defined set of measurement conditions. The measured value is what we want to know, and the
goal of measurement is to have the measured value be as close as possible to the real value.
Figure 1-6. Ideal, real, and measured values
1.5
Component dependency factors
The measured impedance value of a component depends on several measurement conditions, such as test
frequency, and test signal level. Effects of these component dependency factors are different for different types of
materials used in the component, and by the manufacturing process used. The following are typical dependency
factors that affect the impedance values of measured components.
1.5.1
Frequency
Frequency dependency is common to all real-world components because of the existence of parasitics. Not all parasitics affect the measurement, but some prominent parasitics determine the component’s frequency characteristics.
The prominent parasitics will be different when the impedance value of the primary element is not the same. Figures
1-7 through 1-9 show the typical frequency response for real-world capacitors, inductors, and resistors.
Ls C R s
Ls: Lead inductance
Rs: Equivalent series resistance (ESR)
90º
L og | Z|
1
C
| Z|
q
90º
L og | Z |
1
C
| Z|
q
0º
Ls
0º
Ls
Rs
–90º
Rs
SRF
–90º
Log f
Log f
SR F
Frequency
Frequency
(a) General capacitor
(b) Capacitor with large ESR
Figure 1-7. Capacitor frequency response
Cp
Cp
L
L
Cp: Stray capacitance
Rs: Resistance of winding
Rs
90º
Log | Z|
q
1
wCp
| Z|
Rs
Rp: Parallel resistance
equivalent to core loss
Rp
90º
L og | Z|
q
q
0º
wL
1
wCp
Rp
| Z|
q
0º
wL
Rs
–90º
SRF
Frequency
(a) General inductor
Figure 1-8. Inductor frequency response
Log f
Rs
–90º
SRF
Frequency
(b) Inductor with high core loss
Log f
Cp
R
R
Cp: Stray capacitance
Ls
Ls: Lead inductance
90º
Log | Z|
1
w Cp
| Z|
90º
Log | Z |
q
q
0º
q
|Z|
0º
q
wL
–90º
–90º
Log f
Log f
Frequency
Frequency
(a) High value resistor
(b) Low value resistor
Figure 1-9. Resistor frequency response
As for capacitors, parasitic inductance is the prime cause of the frequency response as shown in Figure 1-7. At low
frequencies, the phase angle (θ) of impedance is around –90°, so the reactance is capacitive. The capacitor frequency
response has a minimum impedance point at a self-resonant frequency (SRF), which is determined from the
capacitance and parasitic inductance (Ls) of a series equivalent circuit model for the capacitor. At the self-resonant
frequency, the capacitive and inductive reactance values are equal (1/(ωC) = ωLs.) As a result, the phase angle is 0°
and the device is resistive. After the resonant frequency, the phase angle changes to a positive value around +90°
and, thus, the inductive reactance due to the parasitic inductance is dominant.
Capacitors behave as inductive devices at frequencies above the SRF and, as a result, cannot be used as a capacitor.
Likewise, regarding inductors, parasitic capacitance causes a typical frequency response as shown in Figure 1-8. Due
to the parasitic capacitance (Cp), the inductor has a maximum impedance point at the SRF (where ωL = 1/(ωCp).) In
the low frequency region below the SRF, the reactance is inductive. After the resonant frequency, the capacitive
reactance due to the parasitic capacitance is dominant. The SRF determines the maximum usable frequency of
capacitors and inductors.
1.5.2 Test signal level
The test signal (AC) applied may affect the measurement result for some components. For example, ceramic
capacitors are test-signal-voltage dependent as shown in Figure 1-10 (a). This dependency varies depending on
the dielectric constant (K) of the material used to make the ceramic capacitor.
Cored-inductors are test-signal-current dependent due to the electromagnetic hysteresis of the core material.
Typical AC current characteristics are shown in Figure 1-10 (b).
Figure 1-10. Test signal level (AC) dependencies of ceramic capacitors and cored-inductors
1.5.3 DC bias
DC bias dependency is very common in semiconductor components such as diodes and transistors. Some passive
components are also DC bias dependent. The capacitance of a high-K type dielectric ceramic capacitor will vary
depending on the DC bias voltage applied, as shown in Figure 1-11 (a).
In the case of cored-inductors, the inductance varies according to the DC bias current flowing through the coil. This
is due to the magnetic flux saturation characteristics of the core material. Refer to Figure 1-11 (b).
Figure 1-11. DC bias dependencies of ceramic capacitors and cored-inductors
1.5.4 Temperature
Most types of components are temperature dependent. The temperature coefficient is an important specification for
resistors, inductors, and capacitors. Figure 1-12 shows some typical temperature dependencies that affect ceramic
capacitors with different dielectrics.
1.5.5 Other dependency factors
Other physical and electrical environments, e.g., humidity, magnetic fields, light, atmosphere, vibration, and time,
may change the impedance value. For example, the capacitance of a high-K type dielectric ceramic capacitor
decreases with age as shown in Figure 1-13.
Figure 1-12. Temperature dependency of ceramic capacitors
1.6
Figure 1-13. Aging dependency of ceramic capacitors
Equivalent circuit models of components
Even if an equivalent circuit of a device involving parasitics is complex, it can be lumped as the simplest series or
parallel circuit model, which represents the real and imaginary (resistive and reactive) parts of total equivalent circuit
impedance. For instance, Figure 1-14 (a) shows a complex equivalent circuit of a capacitor. In fact, capacitors have
small amounts of parasitic elements that behave as series resistance (Rs), series inductance (Ls), and parallel
resistance (Rp or 1/G.) In a sufficiently low frequency region, compared with the SRF, parasitic inductance (Ls) can
be ignored. When the capacitor exhibits a high reactance (1/(ωC)), parallel resistance (Rp) is the prime determinative,
relative to series resistance (Rs), for the real part of the capacitor’s impedance. Accordingly, a parallel equivalent
circuit consisting of C and Rp (or G) is a rational approximation to the complex circuit model. When the reactance of
a capacitor is low, Rs is a more significant determinative than Rp. Thus, a series equivalent circuit comes to the
approximate model. As for a complex equivalent circuit of an inductor such as that shown in Figure 1-14 (b), stray
capacitance (Cp) can be ignored in the low frequency region. When the inductor has a low reactance, (ωL), a series
equivalent circuit model consisting of L and Rs can be deemed as a good approximation. The resistance, Rs, of a
series equivalent circuit is usually called equivalent series resistance (ESR).
Cp
Rp (G)
(a) Capacitor
(b) Inductor
L
C
Ls
Rs
Rs
Rp (G)
Parallel (High | Z| )
Series (Low | Z| )
Series (Low |Z| )
Parallel ( High|Z|)
Log | Z|
Log | Z|
Rp
High Z
Rp
High Z
1
C
|Z|
|Z|
L
Low Z
Low Z
Rs
Rs
Log f
Frequency
Frequency
Log f
Rp (G)
Rp (G)
C
C
Rs
L
Ls-Rs
Cs-Rs
Cp-Rp
Rs
L
Lp-Rp
Figure 1-14. Equivalent circuit models of (a) a capacitor and (b) an inductor
Note: Generally, the following criteria can be used to roughly discriminate between low, middle, and high
impedances (Figure 1-15.) The medium Z range may be covered with an extension of either the low Z
or high Z range. These criteria differ somewhat, depending on the frequency and component type.
1k
Low Z
100 k
Medium Z
High Z
Series
Parallel
Figure 1-15. High and low impedance criteria
In the frequency region where the primary capacitance or inductance of a component exhibits almost a flat frequency
response, either a series or parallel equivalent circuit can be applied as a suitable model to express the real
impedance characteristic. Practically, the simplest series and parallel models are effective in most cases when
representing characteristics of general capacitor, inductor, and resistor components.
1.7
Measurement circuit modes
As we learned in Section 1.2, measurement instruments basically measure the real and imaginary parts of impedance
and calculate from them a variety of impedance parameters such as R, X, G, B, C, and L. You can choose from series
and parallel measurement circuit modes to obtain the measured parameter values for the desired equivalent circuit
model (series or parallel) of a component as shown in Table 1-1.
Table 1-1. Measurement circuit modes
Equivalent circuit models of component
Series
R
jX
Measurement circuit modes and impedance parameters
Series mode: Cs, Ls, Rs, Xs
G
Parallel mode: Cp, Lp, Rp, Gp, Bp
Parallel
jB
Though impedance parameters of a component can be expressed by whichever circuit mode (series or parallel) is
used, either mode is suited to characterize the component at your desired frequencies. Selecting an appropriate
measurement circuit mode is often vital for accurate analysis of the relationships between parasitics and the
component’s physical composition or material properties. One of the reasons is that the calculated values of C, L, R,
and other parameters are different depending on the measurement circuit mode as described later. Of course,
defining the series or parallel equivalent circuit model of a component is fundamental to determining which
measurement circuit mode (series or parallel) should be used when measuring C, L, R, and other impedance
parameters of components. The criteria shown in Figure 1-15 can also be used as a guideline for selecting the
measurement circuit mode suitable for a component.
Table 1-2 shows the definitions of impedance measurement parameters for the series and parallel modes. For the
parallel mode, admittance parameters are used to facilitate parameter calculations.
Table 1-2. Definitions of impedance parameters for series and parallel modes
Series mode
Rs ±jXs
Parallel mode
|Z| = √Rs2 + Xs2
θ = tan–1 (Xs/Rs)
Gp
±jBp
Rs: Series resistance
Xs: Series reactance (XL = wLs, XC = –1/(wCs))
Ls: Series inductance (= XL/w)
Cs: Series capacitance (= –1/(wXC))
D: Dissipation factor (= Rs/Xs = Rs/(wLs) or wCsRs)
Q: Quality factor (= Xs/Rs = wLs/Rs or 1/(wCsRs))
|Y| = √Gp2 + Bp2
θ = tan–1 (Bp/Gp)
Gp: Parallel conductance (= 1/Rp)
Bp: Parallel susceptance (BC = wCp, BL = –1/(wLp))
Lp: Parallel inductance (= –1/(wBL))
Cp: Parallel capacitance (= BC/w)
D: Dissipation factor (= Gp/Bp = Gp/(wCp)
= 1/(wCpRp) or wLpGp = wLp/Rp)
Q: Quality factor (= Bp/Gp = wCp/Gp
= wCpRp or 1/(wLpGp) = Rp/(wLp))
Though series and parallel mode impedance values are identical, the reactance (Xs), is not equal to reciprocal of parallel susceptance (Bp), except when Rs = 0 and Gp = 0. Also, the series resistance (Rs), is not equal to parallel resistance (Rp) (or reciprocal of Gp) except when Xs = 0 and Bp = 0. From the definition of Y = 1/Z, the series and parallel
mode parameters, Rs, Gp (1/Rp), Xs, and Bp are related with each other by the following equations:
Z = Rs + jXs = 1/Y = 1/(Gp + jBp) = Gp/(Gp2 + Bp2) – jBp/(Gp2 + Bp2)
Y = Gp + jBp = 1/Z = 1/(Rs + jXs) = Rs/(Rs2 + Xs2) – jXs/(Rs2 + Xs2)
Rs = Gp/(Gp2 + Bp2) ) Rs = RpD2/(1 + D2)
Gp = Rs/(Rs2 + Xs2) ) Rp = Rs(1 + 1/D2)
Xs = –Bp/(Gp2 + Bp2) ) Xs = Xp/(1 + D2)
Bp = –Xs/(Rs2 + Xs2) ) Xp = Xs(1 + D2)
Table 1-3 shows the relationships between the series and parallel mode values for capacitance, inductance, and
resistance, which are derived from the above equations.
Table 1-3. Relationships between series and parallel mode CLR values
Series
Rs ±jXs
Parallel
Dissipation factor
Gp
(Same value for series and parallel)
±jBp
Capacitance
Cs = Cp(1 + D2)
Cp = Cs/(1 + D2)
D = Rs/Xs = wCsRs
D = Gp/Bp = Gp/(wCp) = 1/(wCpRp)
Inductance
Ls = Lp/(1 + D2)
Lp = Ls(1 + D2)
D = Rs/Xs = Rs/(wLs)
D = Gp/Bp = wLpGp = wLp/Rp
Resistance
Rs = RpD2/(1 + D2)
Rp = Rs(1 + 1/D2)
–––––
Cs, Ls, and Rs values of a series equivalent circuit are different from the Cp, Lp, and Rp values of a parallel equivalent
circuit. For this reason, the selection of the measurement circuit mode can become a cause of measurement discrepancies. Fortunately, the series and parallel mode measurement values are interrelated by using simple equations that
are a function of the dissipation factor (D.) In a broad sense, the series mode values can be converted into parallel
mode values and vice versa.
Figure 1-16 shows the Cp/Cs and Cs/Cp ratios calculated for dissipation factors from 0.01 to 1.0. As for inductance,
the Lp/Ls ratio is same as Cs/Cp and the Ls/Lp ratio equals Cp/Cs.
1.01
1.009
Cp
Cs
Cs
Cp
0.999
1.9
1
0.95
Cs
Cp
Cp
Cs
0.998
1.8
0.997
1.7
0.85
1.006
0.996
1.6
0.8
1.005
0.995
1.5
0.75
1.004
0.994
1.4
0.7
1.003
0.993
1.3
0.65
1.002
0.992
1.2
0.6
1.008
1.007
Cs
Cp
2
1
1.001
1
0.01
Cp
Cs
Cs
Cp
1.1
0.991
0.55
0.1
0.1
Cp
Cs
0.5
1
0.99
0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09
0.9
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
Dissipation factor
Dissipation factor
Figure 1-16. Relationships of series and parallel capacitance values
For high D (low Q) devices, either the series or parallel model is a better approximation of the real impedance
equivalent circuit than the other one. Low D (high Q) devices do not yield a significant difference in measured
C or L values due to the measurement circuit mode. Since the relationships between the series and parallel mode
measurement values are a function of D2, when D is below 0.03, the difference between Cs and Cp values (also
between Ls and Lp values) is less than 0.1 percent. D and Q values do not depend on the measurement circuit
modes.
Figure 1-17 shows the relationship between series and parallel mode resistances. For high D (low Q) components,
the measured Rs and Rp values are almost equal because the impedance is nearly pure resistance. Since the
difference between Rs and Rp values increases in proportion to 1/D2, defining the measurement circuit mode is vital
for measurement of capacitive or inductive components with low D (high Q.)
10000
1000
Rp
Rs
100
10
1
0.01
0.1
1
Dissipation factor
Figure 1-17. Relationships of series and parallel resistance values
10
1.8
Three-element equivalent circuit and sophisticated component models
The series and parallel equivalent circuit models cannot serve to accurately depict impedance characteristics of
components over a broad frequency range because various parasitics in the components exercise different influence
on impedance depending on the frequency. For example, capacitors exhibit typical frequency response due to
parasitic inductance, as shown in Figure 1-18. Capacitance rapidly increases as frequency approaches the resonance
point. The capacitance goes down to zero at the SRF because impedance is purely resistive. After the resonant
frequency, the measured capacitance exhibits a negative value, which is calculated from inductive reactance. In the
aspect of the series Cs-Rs equivalent circuit model, the frequency response is attributed to a change in effective
capacitance. The effect of parasitic inductance is unrecognizable unless separated out from the compound
reactance. In this case, introducing series inductance (Ls) into the equivalent circuit model enables the real
impedance characteristic to be properly expressed with three-element (Ls-Cs-Rs) equivalent circuit parameters.
When the measurement frequency is lower than approximately 1/30 resonant frequency, the series Cs-Rs
measurement circuit mode (with no series inductance) can be applied because the parasitic inductance scarcely
affects measurements.
+C
3-element equivalent circuit model
Capacitive
Inductive
Ls
(Negative Cm value)
Cm =
Cm
Cs
1-
2 CsLs
Effective range of
0
Equivalent L =
–C
SRF
Frequency
Figure 1-18. Influence of parasitic inductance on capacitor
Log f
–1
= L s (1 2 Cm
1
2 CsLs
)
Cs
Rs
When both series and parallel resistances have a considerable amount of influence on the impedance of a reactive
device, neither the series nor parallel equivalent circuit models may serve to accurately represent the real C, L, or R
value of the device. In the case of the capacitive device shown in Figure 1-19, both series and parallel mode
capacitance (Cs and Cp) measurement values at 1 MHz are different from the real capacitance of the device. The
correct capacitance value can be determined by deriving three-element (C-Rp-Rs) equivalent circuit parameters
from the measured impedance characteristic. In practice, C-V characteristics measurement for an ultra-thin CMOS
gate capacitance often requires a three-element (C-Rs-Rp) equivalent circuit model to be used for deriving real
capacitance without being affected by Rs and Rp.
10 pF
Rs
700
Cs = C +
Cp =
D=
at 1 MHz
1
2
CRs +
10.3
0.8
+
Cs = 10.11 pF
10.1
2
C 2 Rp 2 Rs 2
1
Rs
(1 +
)
CRp
Rp
0.7
0.6
Cs
10.0
9.9
0.5
Cp
0.4
Cp = 9.89 pF
9.7
CRp 2
2
0.9
9.8
CRp 2
(Rs + Rp)
10.4
10.2
Capacitance (pF)
Xc = 15.9 k
1.0
0.3
0.2
D
9.6
9.5
100 k
Dissipation factor (D)
Rp
150 k
C
10.5
0.1
0.0
1M
10 M
Frequency (Hz)
Figure 1-19. Example of capacitive device affected by both Rs and Rp
By measuring impedance at a frequency you can acquire a set of the equivalent resistance and reactance values, but
it is not enough to determine more than two equivalent circuit elements. In order to derive the values of more than
two equivalent circuit elements for a sophisticated model, a component needs to be measured at least at two
frequencies. The Keysight Technologies, Inc. impedance analyzers have the equivalent circuit analysis function that
automatically calculates the equivalent circuit elements for three- or four-element models from a result of a swept
frequency measurement. The details of selectable three-/four-element equivalent circuit models and the equivalent
circuit analysis function are described in Section 5.15.
1.9
Reactance chart
The reactance chart shows the impedance and admittance values of pure capacitance or inductance at arbitrary
frequencies. Impedance values at desired frequencies can be indicated on the chart without need of calculating
1/(ωC) or ωL values when discussing an equivalent circuit model for a component and also when estimating the
influence of parasitics. To cite an example, impedance (reactance) of a 1 nF capacitor, which is shown with an
oblique bold line in Figure 1-20, exhibits 160 kΩ at 1 kHz and 16 Ω at 10 MHz. Though a parasitic series resistance
of 0.1 Ω can be ignored at 1 kHz, it yields a dissipation factor of 0.0063 (ratio of 0.1 Ω to 16 Ω) at 10 MHz. Likewise,
though a parasitic inductance of 10 nH can be ignored at 1 kHz, its reactive impedance goes up to 0.63 Ω at 10 MHz
and increases measured capacitance by +4 percent (this increment is calculated as 1/(1 – XL/XC) = 1/(1 – 0.63/16).)
At the intersection of 1 nF line (bold line) and the 10 nH line at 50.3 MHz, the parasitic inductance has the same
magnitude (but opposing vector) of reactive impedance as that of primary capacitance and causes a resonance
(SRF). As for an inductor, the influence of parasitics can be estimated in the same way by reading impedance
(reactance) of the inductor and that of a parasitic capacitance or a resistance from the chart.
10
pF
100 M
1
pF
10
10
k H 0 fF
1k
1
H 0f
F
10
H
0 1 fF
10
10
H0a
F
1
10
H aF
10
0
m
H
m
10
10
H
0
pF
10 M
1
m
H
1
H
nF
10
1M
0
10
nF
100 k
H
10
10
0
nF
10 k
H
1
1
1k
F
|Z|
C
10
0
nH
10
F
10
100
nH
10
0
F
10
1
nH
1
m
F
1
10
10
0
pH
m
F
100 m
10
10
pH
0
m
F
10 m
1
1m
100
1k
10 k
100 k
1M
Frequency (Hz)
Figure 1-20. Reactance chart
10 M
100 M
1G
pH
L
Most of the modern impedance measuring instruments basically measure vector impedance (R + jX) or vector
admittance (G + jB) and convert them, by computation, into various parameters, Cs, Cp, Ls, Lp, D, Q, |Z|, |Y|, θ, etc.
Since measurement range and accuracy are specified for the impedance and admittance, both the range and
accuracy for the capacitance and inductance vary depending on frequency. The reactance chart is also useful when
estimating measurement accuracy for capacitance and inductance at your desired frequencies. You can plot the
nominal value of a DUT on the chart and find the measurement accuracy denoted for the zone where the DUT
value is enclosed. Figure 1-21 shows an example of measurement accuracy given in the form of a reactance chart.
The intersection of arrows in the chart indicates that the inductance accuracy for 1 µH at 1 MHz is ±0.3 percent. D
accuracy comes to ±0.003 (= 0.3/100.) Since the reactance is 6.28 Ω, Rs accuracy is calculated as ±(6.28 x 0.003) =
±0.019 Ω. Note that a strict accuracy specification applied to various measurement conditions is given by the
accuracy equation.
Figure 1-21. Example of measurement accuracy indicated on a reactance chart
2.0 Impedance Measurement Instruments
2.1
Measurement methods
There are many measurement methods to choose from when measuring impedance, each of which has advantages
and disadvantages. You must consider your measurement requirements and conditions, and then choose the most
appropriate method, while considering such factors as frequency coverage, measurement range, measurement
accuracy, and ease of operation. Your choice will require you to make tradeoffs as there is not a single measurement
method that includes all measurement capabilities. Figure 2-1 shows six commonly used impedance measurement
methods, from low frequencies up to the microwave region. Table 2-1 lists the advantages and disadvantages of
each measurement method, the Keysight instruments that are suited for making such measurements, the
instruments’ applicable frequency range, and the typical applications for each method. Considering only
measurement accuracy and ease of operation, the auto-balancing bridge method is the best choice for
measurements up to 120 MHz. For measurements from 100 MHz to 3 GHz, the RF I-V method has the best
measurement capability, and from 3 GHz and up the network analysis is the recommended technique.
Bridge method
When no current lows through the detector (D), the value of the
unknown impedance (Zx) can be obtained by the relationship of the
other bridge elements. Various types of bridge circuits, employing
combinations of L, C, and R components as the bridge elements, are
used for various applications.
Resonant method
When a circuit is adjusted to resonance by adjusting a tuning
capacitor (C), the unknown impedance Lx and Rx values are
obtained from the test frequency, C value, and Q value. Q is
measured directly using a voltmeter placed across the tuning
capacitor. Because the loss of the measurement circuit is very
low, Q values as high as 300 can be measured. Other than the
direct connection shown here, series and parallel connections are
available for a wide range of impedance measurements.
Figure 2-1. Impedance measurement method (1 of 3)
I-V method
The unknown impedance (Zx) can be calculated from measured voltage and current values. Current is calculated using
the voltage measurement across an accurately known low
value resistor (R.) In practice, a low loss transformer is used
in place of R to prevent the effects caused by placing a low
value resistor in the circuit. The transformer, however, limits
the low end of the applicable frequency range.
RF I-V method
While the RF I-V measurement method is based on
the same principle as the I-V method, it is conigured
in a different way by using an impedance-matched
measurement circuit (50 Ω) and a precision coaxial test
port for operation at higher frequencies. There are two
types of the voltmeter and current meter arrangements
that are suited to low impedance and high impedance
measurements.
Impedance of DUT is derived from measured voltage
and current values, as illustrated. The current that
lows through the DUT is calculated from the voltage
measurement across a known R. In practice, a low loss
transformer is used in place of the R. The transformer limits
the low end of the applicable frequency range.
Network analysis method
The relection coeficient is obtained by measuring the ratio
of an incident signal to the relected signal. A directional
coupler or bridge is used to detect the relected signal
and a network analyzer is used to supply and measure the
signals. Since this method measures relection at the DUT,
it is usable in the higher frequency range.
Figure 2-1. Impedance measurement method (2 of 3)
Auto-balancing bridge method
Ix
The current Ix balances with the current Ir which lows through the
range resistor (Rr), by operation of the I-V converter. The potential at
the Low point is maintained at zero volts (thus called a virtual ground.)
The impedance of the DUT is calculated using the voltage measured
at the High terminal (Vx) and across Rr (Vr).
Ir
DUT
High
OSC
Zx
Rr
Low
Vr
Vx
Vx
Zx
Vr
= Ix = Ir =
Zx =
Rr
Vx
Ix
= Rr
Vx
Vr
Note: In practice, the coniguration of the auto-balancing bridge
differs for each type of instrument. Generally, an LCR meter, in a low
frequency range typically below 100 kHz, employs a simple operational ampliier for its I-V converter. This type of instrument has a
disadvantage in accuracy at high frequencies because of performance
limits of the ampliier. Wideband LCR meters and impedance analyzers employ the I-V converter consisting of sophisticated null detector,
phase detector, integrator (loop ilter), and vector modulator to ensure
a high accuracy for a broad frequency range over 1 MHz. This type of
instrument can attain to a maximum frequency of 120 MHz.
Figure 2-1. Impedance measurement method (3 of 3)
Table 2-1. Common impedance measurement methods
Advantages
Disadvantages
Applicable
Keysight measurement
frequency range instruments
Common
applications
Bridge
method
– High accuracy (0.1% typ.)
– Wide frequency coverage
by using different types of
bridges
– Low cost
– Needs to be manually
balanced
– Narrow frequency
coverage with a single
instrument
DC to 300 MHz
None
Standard lab
Resonant
method
– Good Q accuracy up to
high Q
– Needs to be tuned to
resonance
– Low impedance
measurement accuracy
10 kHz to 70 MHz
None
High Q device
measurement
I-V
method
– Grounded device
measurement
– Suitable to probe-type test
needs
– Operating frequency range 10 kHz to 100
is limited by transformer
MHz
used in probe
None
Grounded
device
measurement
RF I-V
method
– High accuracy (1% typ.)
and wide impedance range
at high frequencies
– Operating frequency range 1 MHz to 3 GHz
is limited by transformer
used in test head
E4991B, E4991A
4287A
4395A+43961A
RF component
measurement
Network
analysis
method
– Wide frequency coverage
from LF to RF
– Good accuracy when the
unknown impedance is
close to characterisitic
impedence
– Recalibration required
when the measurement
frequency is changed
– Narrow impedance
measurement range
5 Hz and above
E5061B-3L5
E5071C
4395A
RF component
measurement
Autobalancing
bridge
method
– Wide frequency coverage
from LF to HF
– High accuracy over a wide
impedance measurement
range
– Grounded device
measurement
– High frequency range not
available
20 Hz to 120 MHz E4980A/AL
E4981A
E4990A, 4294A
Generic
component
measurement
E4990A/4294A+42941A1 1. Grounded
E4990A/4294A+42942A1 device
measurement
Note: Keysight Technologies currently offers no instruments for the bridge method and the resonant method
shaded in the above table.
2.2
Operating theory of practical instruments
The operating theory and key functions of the auto balancing bridge instrument are discussed in Sections 2.3
through 2.4. A discussion on the RF I-V instrument is described in Sections 2.5 through 2.7.
2.3
Theory of auto-balancing bridge method
The auto-balancing bridge method is commonly used in modern LF impedance measurement instruments. Its
operational frequency range has been extended up to 120 MHz.
Basically, in order to measure the complex impedance of the DUT it is necessary to measure the voltage of the test
signal applied to the DUT and the current that flows through it. Accordingly, the complex impedance of the DUT can
be measured with a measurement circuit consisting of a signal source, a voltmeter, and an ammeter as shown in
Figure 2-2 (a). The voltmeter and ammeter measure the vectors (magnitude and phase angle) of the signal voltage
and current, respectively.
Ix
I
DUT
High
DUT
Low
V
High
A
Z =
Ir
Vx
V
I
(a) The simplest model for
impedance measurement
Figure 2-2. Principle of auto-balancing bridge method
Low
Zx =
Rr
Vr
Vx
Ix
= Rr
Vx
Vr
(b) Impedance measurement
using an operational amplifier
The auto-balancing bridge instruments for low frequency impedance measurement (below 100 kHz) usually employ a
simple I-V converter circuit (an operational amplifier with a negative feedback loop) in place of the ammeter as
shown in Figure 2-2 (b). The bridge section works to measure impedance as follows:
The test signal current (Ix) flows through the DUT and also flows into the I-V converter. The operational amplifier of the
I-V converter makes the same current as Ix flow through the resistor (Rr) on the negative feedback loop. Since the
feedback current (Ir) is equal to the input current (Ix) flows through the Rr and the potential at the Low terminal is
automatically driven to zero volts. Thus, it is called virtual ground. The I-V converter output voltage (Vr) is represented
by the following equation:
Vr = Ir x Rr = Ix x Rr
(2-1)
Ix is determined by the impedance (Zx) of the DUT and the voltage Vx across the DUT as follows:
Ix =
Vx
Zx
(2-2)
From the equations 2-1 and 2-2, the equation for impedance (Zx) of the DUT is derived as follows:
Zx =
Vx
Vx
Rr =
Ix
Vr
(2-3)
The vector voltages Vx and Vr are measured with the vector voltmeters as shown in Figure 2-2 (b). Since the value of
Rr is known, the complex impedance Zx of the DUT can be calculated by using equation 2-3. The Rr is called the
range resistor and is the key circuit element, which determines the impedance measurement range. The Rr value is
selected from several range resistors depending on the Zx of the DUT as described in Section 2.4.3.
In order to avoid tracking errors between the two voltmeters, most of the impedance measuring instruments measure
the Vx and Vr with a single vector voltmeter by alternately selecting them as shown in Figure 2-3. The circuit block,
including the input channel selector and the vector voltmeter, is called the vector ratio detector, whose name comes
from the function of measuring the
vector ratio of Vx and Vr.
Vector ratio detector section
Signal source section
Vx
Rs
DUT
High
Low
Rr
Vr
V
Auto-balancing bridge section
Figure 2-3. Impedance measurement using a single vector voltmeter
Note:
The balancing operation that maintains the low terminal potential at zero volts has the
following advantages in measuring the impedance of a DUT:
(1) The input impedance of ammeter (I-V converter) becomes virtually zero and does not
affect measurements.
(2) Distributed capacitance of the test cables does not affect measurements because there is
no potential difference between the inner and outer shielding conductors of (Lp and Lc)
cables. (At high frequencies, the test cables cause measurement errors as described in
Section 4.5.)
(3) Guarding technique can be used to remove stray capacitance effects as described in
Sections 2.4.7 and 3.4.
Block diagram level discussions for the signal source, auto-balancing bridge, and vector ratio detector are described
in Sections 2.3.1 through 2.3.3.
2.3.1.
Signal source section
The signal source section generates the test signal applied to the unknown device. The frequency of the test signal
(fm) and the output signal level are variable. The generated signal is output at the Hc terminal via a source resistor,
and is applied to the DUT. In addition to generating the test signal that is fed to the DUT, the reference signals used
internally are also generated in this signal source section. Frequency synthesizer and frequency conversion
techniques are employed to generate high-resolution test signals (1 mHz minimum resolution), as well as to expand
the upper frequency limit up to 120 MHz.
2.3.2
Auto-balancing bridge section
The auto-balancing bridge section balances the range resistor current with the DUT current while maintaining a zero
potential at the Low terminal. Figure 2-4 (a) shows a simplified circuit model that expresses the operation of the
auto-balancing bridge. If the range resistor current is not balanced with the DUT current, an unbalance current that
equals Ix – Ir flows into the null detector at the Lp terminal. The unbalance current vector represents how much the
magnitude and phase angle of the range resistor current differ from the DUT current. The null detector detects the
unbalance current and controls both the magnitude and phase angle of the OSC2 output so that the detected
current goes to zero.
Low frequency instruments, below 100 kHz, employ a simple operational amplifier to configure the null detector and
the equivalent of OSC2 as shown in Figure 2-4 (b). This circuit configuration cannot be used at frequencies higher
than 100 kHz because of the performance limits of the operational amplifier. The instruments that cover frequencies
above 100 kHz have an auto balancing bridge circuit consisting of a null detector, 0°/90° phase detectors, and a
vector modulator as shown in Figure 2-4 (c). When an unbalance current is detected with the null detector, the
phase detectors in the next stage separate the current into 0° and 90° vector components. The phase detector
output signals go through loop filters (integrators) and are applied to the vector modulator to drive the 0°/90°
component signals. The 0°/90° component signals are compounded and the resultant signal is fed back through
range resistor (Rr) to cancel the current flowing through the DUT. Even if the balancing control loop has phase errors,
the unbalance current component, due to the phase errors, is also detected and fed back to cancel the error in the
range resistor current. Consequently, the unbalance current converges to exactly zero, ensuring Ix = Ir over a broad
frequency range up to 120 MHz.
If the unbalance current flowing into the null detector exceeds a certain threshold level, the unbalance detector after
the null detector annunciates the unbalance state to the digital control section of the instrument. As a result, an error
message such as “OVERLOAD” or “BRIDGE UNBALANCED” is displayed.
Ix
Ir
VX
DUT
Hc
Vr
Lc
Lp
Rr
Null
detector
OSC1
OSC2
VX
Hp
Lc
DUT
(a) Operation image of the auto-balancing bridge
Hc
Rr
Vr
Lp
Null detector
VX
OSC
Hp
Lc
Vr
DUT
Hc
v
Lp
0°
Phase
detector
90°
Rr
OSC
Vector modulator
–90°
(b) Auto-balancing bridge for frequency below 100 kHz
Figure 2-4. Auto-balancing bridge section block diagram
(c) Auto-balancing bridge for frequency above 100 kHz
2.3.3
Vector ratio detector section
The vector ratio detector (VRD) section measures the ratio of vector voltages across the DUT, Vx, and across the
range resistor (Vr) series circuit, as shown in Figure 2-5 (b). The VRD consists of an input selector switch (S), a phase
detector, and an A-D converter, also shown in this diagram.) The measured vector voltages, Vx and Vr, are used to
calculate the complex impedance (Zx) in accordance with equation 2-3.
Buffer
90º
S
Lc
Rr
0º, 90º
DU T
Hc
0º
a
A/D
To digital
section
Vr
Hp
V r = c + jd
d
ATT
Buffer
V X = a + jb
b
Phase
detector
VX
Lp
c
(b) Block diagram
(a) Vector diagram of Vx and Vr
Figure 2-5. Vector ratio detector section block diagram
In order to measure the Vx and Vr, these vector signals are resolved into real and imaginary components, Vx = a + jb
and Vr = c + jd, as shown in Figure 2-5 (a). The vector voltage ratio of Vx/Vr is represented by using the vector
components a, b, c, and d as follows:
Vx
Vr
=
a + jb
c + jd
=
ac + bd
c2 + d2
+ j
bc - ad
(2-4)
c2 + d2
The VRD circuit is operated as follows. First, the input selector switch (S) is set to the Vx position. The phase detector
is driven with 0° and 90° reference phase signals to extracts the real and imaginary components (a and jb) of the Vx
signal. The A-D converter next to the phase detector outputs digital data for the magnitudes of a and jb. Next, S is
set to the Vr position. The phase detector and the A-D converter perform the same for the Vr signal to extract the
real and imaginary components (c and jd) of the Vr signal.
From the equations 2-3 and 2-4, the equation that represents the complex impedance Zx of the DUT is derived as
follows (equation 2-5):
Zx = Rx + jXx = Rr
[
ac + bd
bc - ad
Vx
+ j 2
= Rr 2
2
Vr
c +d
c + d2
]
(2-5)
The resistance and the reactance of the DUT are thus calculated as:
Rx = Rr
ac + bd
bc - ad
, Xx = Rr 2
c2 + d2
c + d2
(2-6)
Various impedance parameters (Cp, Cs, Lp, Ls, D, Q, etc) are calculated from the measured Rx and Xx values by
using parameter conversion equations which are described in Section 1.
2.4
Key measurement functions
The following discussion describes the key measurement functions for advanced impedance measurement instruments. Thoroughly understanding these measurement functions will eliminate the confusion sometimes caused by
the measurement results obtained.
2.4.1
Oscillator (OSC) level
The oscillator output signal is output through the Hc terminal and can be varied to change the test signal level
applied to the DUT. The specified output signal level, however, is not always applied directly to the DUT. In general,
the specified OSC level is obtained when the High terminal is open. Since source resistor (Rs) is connected in series
with the oscillator output, as shown in Figure 2-6, there is a voltage drop across Rs. So, when the DUT is connected,
the applied voltage (Vx) depends on the value of the source resistor and the DUT’s impedance value. This must be
taken into consideration especially when measuring low values of impedance (low inductance or high capacitance).
The OSC level should be set as high as possible to obtain a good signal-to-noise (S/N) ratio for the vector ratio
detector section. A high S/N ratio improves the accuracy and stability of the measurement. In some cases, however,
the OSC level should be decreased, such as when measuring cored-inductors, and when measuring semiconductor
devices in which the OSC level is critical for the measurement and to the device itself.
Figure 2-6. OSC level divided by source resistor (Rs) and DUT impedance (Zx)
2.4.2
DC bias
In addition to the AC test signal, a DC voltage can be output through the Hc terminal and applied to the DUT. A
simplified output circuit, with a DC bias source, is shown in Figure 2-7. Many of the conventional impedance
measurement instruments have a voltage bias function, which assumes that almost no bias current flows (the DUT
has a high resistance.) If the DUT’s DC resistance is low, a bias current flows through the DUT and into the resistor
(Rr) thereby raising the DC potential of the virtual ground point. Also, the bias voltage is dropped at source resistor
(Rs.) As a result, the specified bias voltage is not applied to the DUT and, in some cases, it may cause measurement
error. This must be taken into consideration when a low-resistivity semiconductor device is measured.
The Keysight E4990A (and some other impedance analyzers) has an advanced DC bias function that can be set to
either voltage source mode or current source mode. Because the bias output is automatically regulated according to
the monitored bias voltage and current, the actual bias voltage or current applied across the DUT is always
maintained at the setting value regardless of the DUT’s DC resistance. The bias voltage or current can be regulated
when the output is within the specified compliance range.
Inductors are conductive at DC. Often a DC current dependency of inductance needs to be measured. Generally the
internal bias output current is not enough to bias the inductor at the required current levels. To apply a high DC bias
current to the DUT, an external current bias unit or adapter can be used with specific instruments. The 42841A and
its bias accessories are available for high current bias measurements using the Keysight E4980A.
Figure 2-7. DC bias applied to DUT referenced to virtual ground
2.4.3
Ranging function
To measure impedance from low to high values, impedance measurement instruments have several measurement
ranges. Generally, seven to ten measurement ranges are available and the instrument can automatically select the
appropriate measurement range according to the DUT’s impedance. Range changes are generally accomplished by
changing the gain multiplier of the vector ratio detector, and by switching the range resistor (Figure 2-8 (a).) This
insures that the maximum signal level is fed into the analog-to-digital (A-D) converter to give the highest S/N ratio
for maximum measurement accuracy.
The range boundary is generally specified at two points to give an overlap between adjacent ranges. Range changes
occur with hysteresis as shown in Figure 2-8 (b), to prevent frequent range changes due to noise.
On any measurement range, the maximum accuracy is obtained when the measured impedance is close to the fullscale value of the range being used. Conversely, if the measured impedance is much lower than the full-scale value
of the range being used, the measurement accuracy will be degraded. This sometimes causes a discontinuity in the
measurement values at the range boundary. When the range change occurs, the impedance curve will skip. To
prevent this, the impedance range should be set manually to the range which measures higher impedance.
Figure 2-8. Ranging function
2.4.4
Level monitor function
Monitoring the test signal voltage or current applied to the DUT is important for maintaining accurate test conditions,
especially when the DUT has a test signal level dependency. The level monitor function measures the actual signal
level across the DUT. As shown in Figure 2-9, the test signal voltage is monitored at the High terminal and the test
signal current is calculated using the value of range resistor (Rr) and the voltage across it.
Instruments equipped with an auto level control (ALC) function can automatically maintain a constant test signal
level. By comparing the monitored signal level with the test signal level setting value, the ALC adjusts the oscillator
output until the monitored level meets the setting value. There are two ALC methods: analog and digital. The analog
type has an advantage in providing a fast ALC response, whereas the digital type has an advantage in performing a
stable ALC response for a wide range of DUT impedance (capacitance and inductance.)
Figure 2-9. Test signal level monitor and ALC function
2.4.5
Measurement time and averaging
Achieving optimum measurement results depends upon measurement time, which may vary according to the control
settings of the instrument (frequency, IF bandwidth, etc.) When selecting the measurement time modes, it is
necessary to take some tradeoffs into consideration. Speeding up measurement normally conflicts with the accuracy,
resolution, and stability of measurement results.The measurement time is mainly determined by operating time
(acquisition time) of the A-D converter in the vector ratio detector. To meet the desired measurement speed, modern
impedance measurement instruments use a high speed sampling A-D converter, in place of the previous technique,
which used a phase detector and a dual-slope A-D converter. Measurement time is proportional to the number of
sampling points taken to convert the analog signal (Edut or Err) into digital data for each measurement cycle.
Selecting a longer measurement time results in taking a greater number of sampling points for more digital data,
thus improving measurement precision. Theoretically, random noise (variance) in a measured value proportionately
decreases inversely to the square root of the A-D converter operating time.
Averaging function calculates the mean value of measured parameters from the desired number of measurements.
Averaging has the same effect on random noise reduction as that by using a long measurement time.
Figure 2-10. Relationship of measurement time and precision
2.4.6
Compensation function
Impedance measurement instruments are calibrated at UNKNOWN terminals and measurement accuracy is specified
at the calibrated reference plane. However, an actual measurement cannot be made directly at the calibration plane
because the UNKNOWN terminals do not geometrically fit to the shapes of components that are to be tested.
Various types of test fixtures and test leads are used to ease connection of the DUT to the measurement terminals.
(The DUT is placed across the test fixture’s terminals, not at the calibration plane.) As a result, a variety of error
sources (such as residual impedance, admittance, electrical length, etc.) are involved in the circuit between the DUT
and the UNKNOWN terminals. The instrument’s compensation function eliminates measurement errors due to these
error sources. Generally, the instruments have the following compensation functions:
- Open/short compensation or open/short/load compensation
- Cable length correction
The open/short compensation function removes the effects of the test fixture’s residuals. The open/short/load
compensation allows complicated errors to be removed where the open/short compensation is not effective. The
cable length correction offsets the error due to the test lead’s transmission characteristics.
The induced errors are dependent upon test frequency, test fixture, test leads, DUT connection configuration, and
surrounding conditions of the DUT. Hence, the procedure to perform compensation with actual measurement setup
is the key to obtaining accurate measurement results. The compensation theory and practice are discussed
comprehensively in Section 4.
2.4.7
Guarding
When in-circuit measurements are being performed or when one parameter of a three-terminal device is to be
measured for the targeted component, as shown in Figure 2-11 (a), the effects of paralleled impedance can be
reduced by using guarding techniques. The guarding techniques can also be utilized to reduce the outcome of stray
capacitance when the measurements are affected by the strays present between the measurement terminals, or
between the DUT terminals and a closely located conductor. (Refer to Section 3.5 for the methods of eliminating the
stray capacitance effects.)
The guard terminal is the circuit common of the auto-balancing bridge and is connected to the shields of the fourterminal pair connectors. The guard terminal is electrically different from the ground terminal, which is connected
directly to the chassis (Figure 2-11 (b).) When the guard is properly connected, as shown in Figure 2-11 (c), it
reduces the test signal's current but does not affect the measurement of the DUT’s impedance (Zx) because Zx is
calculated using DUT current (Ix.)
The details of the guard effects are described as follows. The current (I1) which flows through Z1, does not flow into
the ammeter. As long as I1 does not cause a significant voltage drop of the applied test signal, it scarcely influences
on measurements. The current I2, which is supposed to flow through Z2, is small and negligible compared to Ix,
because the internal resistance of the ammeter (equivalent input impedance of the auto-balancing bridge circuit) is
very low in comparison to Z2. In addition, the potential at the Low terminal of the bridge circuit, in the balanced
condition, is zero (virtual ground.) However, if Z2 is too low, the measurement will become unstable because ammeter
noise increases.
Note: In order to avoid possible bridge unbalance and not cause significant measurement errors, Z2 should not be
lower than certain impedance. Minimum allowable value of Z2 depends on Zx, test cable length, test
frequency, and other measurement conditions.
The actual guard connection is shown in Figure 2-11 (d). The guard lead impedance (Zg) should be as small as
possible. If Zg is not low enough, an error current will flow through the series circuit of Z1 and Z2 and, it is parallel
with Ix.
Note: Using the ground terminal in place of the guard terminal is not recommend because the ground potential is
not the true zero reference potential of the auto-balancing bridge circuit. Basically, the ground terminal is
used to interconnect the ground (chassis) of the instrument and that of a system component, such as an
external bias source or scanner, in order to prevent noise interference that may be caused by mutual ground
potential difference.
Figure 2-11. Guarding techniques
2.4.8
Grounded device measurement capability
Grounded devices such as the input/output of an amplifier can be measured directly using the I-V measurement
method or the reflection coefficient measurement method (Figure 2-12 (a).) However, it is difficult for an auto-balancing bridge to measure low-grounded devices because the measurement signal current bypasses the ammeter
(Figure 2-12 (b).) Measurement is possible only when the chassis ground is isolated from the DUT’s ground. (Note:
The E4990A used with the Keysight 42941A impedance probe kit or the Keysight 42942A terminal adapter will result
in grounded measurements.)
Figure 2-12. Low-grounded device measurement
2.5
Theory of RF I-V measurement method
The RF I-V method featuring Keysight’s RF impedance analyzers and RF LCR meters is an advanced technique to
measure impedance parameters in the high frequency range, beyond the frequency coverage of the auto-balancing
bridge method. It provides better accuracy and a wider impedance range than the network analysis (reflection coefficient measurement) instruments can offer. This section discusses the brief operating theory of the RF I-V method
using a simplified block diagram as shown in Figure 2-13.
Figure 2-13. Simplified block diagram for RF I-V method
The signal source section generates an RF test signal applied to the unknown device and typically has a variable
frequency range from 1 MHz to 3 GHz. Generally, a frequency synthesizer is used to meet frequency accuracy,
resolution, and sweep function needs. The amplitude of signal source output is adjusted for the desired test level by
the output attenuator.
The test head section is configured with a current detection transformer, V/I multiplexer, and test port. The
measurement circuit is matched to the characteristic impedance of 50 Ω to ensure optimum accuracy at high
frequencies. The test port also employs a precision coaxial connector of 50 Ω characteristic impedance. Since the
test current flows through the transformer in series with the DUT connected to the test port, it can be measured from
the voltage across the transformer’s winding. The V channel signal, Edut, represents the voltage across the DUT and
the I channel signal (Etr) represents the current flowing through the DUT. Because the measurement circuit
impedance is fixed at 50 Ω, all measurements are made in reference to 50 Ω without ranging operation.
The vector ratio detector section has similar circuit configurations as the auto-balancing bridge instruments. The V/I
input multiplexer alternately selects the Edut and Etr signals so that the two vector voltages are measured with an
identical vector ratio detector to avoid tracking errors. The measuring ratio of the two voltages derives the
impedance of the unknown device as Zx = 50 × (Edut/Etr.) To make the vector measurement easier, the mixer circuit
down-converts the frequency of the Edut and Etr signals to an IF frequency suitable for the A-D converter’s operating
speed. In practice, double or triple IF conversion is used to obtain spurious-free IF signals. Each vector voltage is
converted into digital data by the A-D converter and is digitally separated into 0° and 90° vector components.
2.6
Difference between RF I-V and network analysis measurement methods
When testing components in the RF region, the RF I-V measurement method is often compared with network analysis. The difference, in principle, is highlighted as the clarifying reason why the RF I-V method has advantages over
the reflection coefficient measurement method, commonly used with network analysis.
The network analysis method measures the reflection coefficient value (Γx) of the unknown device. Γx is correlated
with impedance, by the following equation:
Γx = (Zx - Zo)/(Zx + Zo)
Where, Zo is the characteristic impedance of the measurement circuit (50 Ω) and Zx is the DUT impedance. In
accordance with this equation, measured reflection coefficient varies from –1 to 1 depending on the impedance (Zx.)
The relationship of the reflection coefficient to impedance is graphically shown in Figure 2-14. The reflection coefficient curve in the graph affirms that the DUT is resistive. As Figure 2-14 indicates, the reflection coefficient sharply
varies, with difference in impedance (ratio), when Zx is near Zo (that is, when Γx is near zero). The highest accuracy is
obtained at Zx equal to Zo because the directional bridge for measuring reflection detects the “null” balance point.
The gradient of reflection coefficient curve becomes slower for lower and higher impedance, causing deterioration of
impedance measurement accuracy. In contrast, the principle of the RF I-V method is based on the linear relationship
of the voltage-current ratio to impedance, as given by Ohm’s law. Thus, the theoretical impedance measurement
sensitivity is constant, regardless of measured impedance (Figure 2-15 (a).) The RF I-V method has measurement
sensitivity that is superior to the reflection coefficient measurement except for a very narrow impedance range
around the null balance point (Γ = 0 or Zx = Zo) of the directional bridge.
Figure 2-14. Relationship of reflection coefficient to impedance
Note: Measurement sensitivity is a change in measured signal levels (ΔV/I or ΔV/V) relative to a change
in DUT impedance (ΔZ/Z.) The measurement error approximates to the inverse of the sensitivity.
The reflection coefficient measurement never exhibits such high peak sensitivity for capacitive and inductive DUTs
because the directional bridge does not have the null balance point for reactive impedance. The measurement
sensitivity of the RF I-V method also varies, depending on the DUT’s impedance, because the measurement circuit
involves residuals and the voltmeter and current meter are not ideal (Figure 2-15 (b).) (Voltmeter and current meter
arrangement influences the measurement sensitivity.) Though the measurable impedance range of the RF I-V
method is limited by those error sources, it can cover a wider range than in the network analysis method. The RF I-V
measurement instrument provides a typical impedance range from 0.2 Ω to 20 kΩ at the calibrated test port, while
the network analysis is typically from 2 Ω to 1.5 kΩ (depending upon the required accuracy and measurement
frequency.)
Figure 2-15. Measurement sensitivity of network analysis and RF I-V methods
Note: Typical impedance range implies measurable range within 10 percent accuracy.
Moreover, because the vector ratio measurement is multiplexed to avoid phase tracking error and, because
calibration referenced to a low loss capacitor can be used, accurate and stable measurement of a low dissipation
factor (high Q factor) is enabled. The Q factor accuracy of the network analysis and the RF I-V methods are
compared in Figure 2-16.
Figure 2-16. Comparison of typical Q accuracy
2.7
Key measurement functions
2.7.1
OSC level
The oscillator output signal is output through the coaxial test port (coaxial connector) with a source impedance of
50 Ω. The oscillator output level can be controlled to change the test signal level applied to the DUT. Specified test
signal level is obtained when the connector is terminated with a 50 Ω load (the signal level for open or short
condition is calculated from that for 50 Ω.) When a DUT is connected to the measurement terminals, the current that
flows through the DUT will cause a voltage drop at the 50 Ω source impedance (resistive.) The actual test signal level
applied to the device can be calculated from the source impedance and the DUT’s impedance as shown in Figure
2-6. Those instruments equipped with a level monitor function can display the calculated test signal level and
measurement results.
2.7.2
Test port
The test port of the RF I-V instrument usually employs a precision coaxial connector to ensure optimum accuracy
throughout the high frequency range. The coaxial test port allows RF test fixtures to be attached and the instrument
to be calibrated using traceable coaxial standard terminations. The test port is a two-terminal configuration and does
not have a guard terminal separate from a ground terminal. Therefore, the guarding technique does not apply as well
to the RF I-V measurements as compared to network analysis.
2.7.3
Calibration
Most of the RF vector measurement instruments, such as network analyzers, need to be calibrated each time a
measurement is initiated or a frequency setting is changed. The RF I-V measurement instrument requires calibration
as well. At higher frequencies, a change in the instrument’s operating conditions, such as environmental
temperature, humidity, frequency setting, etc., have a greater effect on measurement accuracy. This nature of RF
vector measurement makes it difficult to sufficiently maintain the calibrated measurement performance over a long
period of time. Thus, users have to periodically perform requisite calibration.
Note: Calibration is necessary each time a measurement setup is changed.
Calibration is executed in reference to three standard terminations: open, short, and load. All three must be
performed. To improve the accuracy of low dissipation factor measurements (high Q factor), calibration with a
low-loss capacitor can be performed. The theory of calibration and appropriate calibration methods are discussed
in Section 4.
2.7.4
Compensation
Two kinds of compensation functions are provided: open/short compensation for eliminating the errors due to test
fixture residuals, and electrical length compensation for minimizing the test port extension induced error. Practical
compensation methods are discussed in Section 4.
2.7.5
Measurement range
The RF I-V measurement method, as well as network analysis, covers the full measurement range from low
impedance to high impedance without ranging operation. All measurements are made at single broad range.
2.7.6
DC bias
The internal DC bias source is connected to the center conductor of the coaxial test port and applies a bias voltage
to the DUT. The internal bias function can be set to either the voltage source mode or the current source mode. The
voltage source mode is adequate to the voltage-biased measurement of the capacitive DUT. The current source
mode is to the current-biased measurement of the inductive DUT. Actual bias voltage and current across the DUT are
monitored and, within specified voltage/current output compliance ranges, automatically regulated at the same level
as the bias setting value regardless of the DUT’s DC resistance, thus allowing accurate DC bias to be applied across
the DUT. Since the internal bias source cannot output bias current large enough for inductor measurements,
generally, current-biased measurement (in excess of maximum output current) requires an external bias method be
used. For biasing up to 5 A and 40 V in a frequency range below 1 GHz, the Keysight 16200B external DC bias
adapter compatible with RF I-V instruments is available.
3.0 Fixturing and Cabling
Connecting a DUT to the measurement terminals of the auto-balancing bridge instrument requires a test fixture or
test cables. The selection of the appropriate test fixtures and cables, as well as the techniques for obtaining the
optimum DUT connection configuration, are important for maximizing the total measurement accuracy. This section
introduces the basic theory and use of each connection configuration, focusing on the auto-balancing bridge
instrument. In RF impedance measurements, the usable connection configuration is the two-terminal (2T)
configuration only. Since the measurement technique for RF impedance is different from that for LF, it is described
separately after the discussion of the auto-balancing bridge instrument.
3.1
Terminal coniguration
An auto-balancing bridge instrument is generally equipped with four BNC connectors, Hcur, Hpot, Lpot, and Lcur, as
measurement terminals (see Figure 3-1.) These terminals are conventionally named "UNKNOWN" terminals. There are
several connection configurations used to interconnect a DUT with the UNKNOWN terminals. Because each method
has advantages and disadvantages, the most suitable method should be selected based on the DUT’s impedance and
required measurement accuracy.
LCR Meter
H cur : High current
H po t : High potential
L pot : Low potential
L cur : Low current
Figure 3-1. Measurement terminals of auto balancing bridge instrument
3.1.1
Two-terminal configuration
The two-terminal (2T) configuration is the simplest method of connecting a DUT but contains many error sources.
Lead inductances (LL), lead resistances (RL), and stray capacitance (Co) between two leads are added to the
measurement result (see Figure 3-2.) Contact resistances (R) between the test fixture’s electrodes and the DUT are
also added to measured impedance. Because of the existence of these error sources, the typical impedance
measurement range (without doing compensation) is limited to 100 Ω to 10 kΩ.
Hc
Rs
V
Hc
Hp
DUT
Null
detector
Lp
Hp
Lp
Lc
Lc
Rr
(b) Connection image
(a) Schematic diagram
Hc
RL
LL
Hp
RC
CO
Lp
Lc
DUT
1 m 10 m 100 m 1
10
100
1 K 10 K 100 K 1 M 10 M ()
RC
RL
LL
(c) Residual parameters
(d) Typical impedance
measurement range
Figure 3-2. Two-terminal (2T) configuration
3.1.2
Three-terminal configuration
The three-terminal (3T) configuration employs coaxial cables to reduce the effects of stray capacitance. The outer
shielding conductors of the coaxial cables are connected to the guard terminal. Measurement accuracy is improved
on the higher impedance measurement range but not on the lower impedance measurement range, because lead
impedances (ωLL and RL) and contact resistances (Rc) still remain (see Figure 3-3.) The typical impedance range will
be extended above 10 kΩ. If the two outer conductors are connected to each other at the ends of the cables as
shown in Figure 3-4, the accuracy for the lower impedance measurement is improved a little. This configuration is
called the shielded 2T configuration.
Hc
Hc
Rs
Hp
V
Hp
DU T
Lp
Null
detector
Lp
Lc
Lc
Rr
(b) Connection image
(a) Schematic diagram
LL
RL
Hc
RC
Hp
DU T
1 m 10 m 100 m 1
Lp
10
100
1 K 10 K 100 K 1 M 10 M ()
RC
Lc
LL
RL
(d) Typical impedance
measurement range
(c) Residual parameters
Figure 3-3. Three-terminal (3T) configuration
Connect here
Connect here
Hc
Hc
Rs
Hp
V
Lp
Null
detector
DUT
Hp
Lp
Lc
Rr
(a) Schematic diagram
Figure 3-4. Shielded two-terminal (2T) configuration
Lc
(b) Connection image
3.1.3
Four-terminal configuration
The four-terminal (4T) configuration can reduce the effects of lead impedances (ωLL and RL) and contact resistances
(Rc) because the signal current path and the voltage sensing leads are independent, as shown in Figures 3-5 (a) and
(b). The voltage sensing leads do not detect the voltage drop caused by the RL, LL, and Rc on the current leads. The
impedances on the voltage sensing leads do not affect measurement because signal current scarcely flows through
these leads. Measurement errors due to the lead impedances and contact resistances are thereby eliminated.
Accuracy for the lower impedance measurement range is thus improved typically down to 10 mΩ. Measurement
accuracy on the higher impedance range is not improved because the stray capacitances between the leads still
remain. The 4T configuration is also called Kelvin connection configuration.
When the DUT’s impedance is below 10 mΩ, large signal current flows through the current leads, generating external
magnetic fields around the leads. The magnetic fields induce error voltages in the adjacent voltage sensing leads.
The effect of mutual coupling (M) between the current and voltage leads is illustrated in Figure 3-5 (e). The induced
error voltages in the voltage sensing leads cause a measurement error in very low impedance measurements.
Hc
Rs
Hc
Hp
V
Hp
DU T
Lp
Lp
Null
detector
Lc
Lc
(b) Connection image
Rr
(a) Schematic diagram
1 m 10 m 100 m 1
RL
LL
10
100
1K
10 K 100 K 1 M 10 M ()
(d) Typical impedance
measurement range
Hc
M
RC
H c or L c cable
Hp
DUT
CO
H p or L p cable
Magnetic flux generated
by test current
Lp
M
Lc
RC
Voltage is induced
(e) Mutual coupling
(c) Residual parameters
Figure 3-5. Four-terminal (4T) configuration
Test current
3.1.4
Five-terminal configuration
The five-terminal (5T) configuration is a combination of the three-terminal (3T) and four-terminal (4T) configurations.
It is equipped with four coaxial cables and all of the outer shielding conductors of the four cables are connected to
the guard terminal (see Figures 3-6 (a) and (b).) This configuration has a wide measurement range from 10 mΩ to 10
MΩ, but the mutual coupling problem still remains. If the outer conductors are connected to each other at the ends of
the cables, as shown in Figure 3-7, the accuracy for the lower impedance measurement is improved a little. This
configuration is called the shielded 4T configuration.
Hc
Rs
Hc
Hp
V
Hp
DUT
Lp
Null
detector
Lp
Lc
Lc
Rr
(b) Connection image
(a) Schematic diagram
Hc
M
Hp
DU T
1 m 10 m 100 m 1
10
100
1K
10 K 100 K 1 M 10 M ()
Lp
M
Lc
RL
(d) Typical impedance
measurement range
LL
(c) Residual parameters
Figure 3-6. Five-terminal (5T) configuration
Hc
Rs
Connect here
Connect here
Hc
Hp
V
Hp
Null
detector
Lp
DUT
Lp
Lc
Lc
Rr
(b) Connection image
(a) Schematic diagram
Figure 3-7. Shielded four-terminal (4T) configuration
3.1.5
Four-terminal pair configuration
The four-terminal pair (4TP) configuration solves the effects of mutual coupling between the leads by employing the
following techniques either 1) and/or 2).
1)
The outer shield conductors work as the return path for the test signal current (they are not grounded). The
magnetic fields produced by the inner and outer currents cancel each other out because of the opposite direc
tions and same amount of current flow. Hence there is little inductive magnetic field, test leads do not contribut
to additional errors due to self or mutual inductance between the individual leads (Fig 3-8. (e))
2)
A vector voltmeter measures the differential voltage between the inner and outer conductors. The differential
measurement method can minimize the influence of the mutual inductance. (Fig 3-8. (a))
As a result, the mutual coupling problem is eliminated. The 4TP configuration can improve the impedance measurement range to below 1 mW. The measurement range achieved by this configuration depends on how well the 4TP
configuration is strictly adhered to up to the connection point of the DUT.
Note: If the shielding conductors of the coaxial test cables are not interconnected properly at the ends of the
cables, the 4TP configuration will not work effectively and, as a result, the measurement range will be limited,
or in some cases, measurements cannot be made.
Do not interconnect the outer
shielding conductors of the cables
at UNKNOWN terminals side
Hc
Rs
Connect here
Hc
Hp
Hp
V
DUT
Lp
Lp
Null
detector
Lc
Lc
Rr
(b) Connection image
Test signal current
Return current
(a) Typical schematic diagram
1 m 10 m 100 m 1
10
100
1K
10 K 100 K 1 M 10 M ()
(d) Typical impedance measurement range
Hc
Magnetic flux generated by
return current
Hp
DUT
Return current
(through outer shield)
Lp
Lc
Magnetic flux generated by
test current
Test current
(through inner cable)
RL
LL
(c) Residual parameters
Figure 3-8. Four-terminal pair (4TP) configuration
(e) Cancellation of magnetic fluxes
3.2
Test ixtures
The test fixture plays an important role in impedance measurement both mechanically and electrically. The quality of
the fixture determines the limit of the total measurement accuracy. This section discusses how to choose or fabricate
a test fixture for use with auto-balancing bridge instruments.
3.2.1
Keysight-supplied test fixtures
Keysight Technologies supplies various types of test fixtures depending on the type of device being tested. To choose
the most suitable test fixture for the DUT, consider not only the physical layout of the contacts but also the usable
frequency range, residual parameters (usable impedance range), and the allowable DC voltage that can be applied.
The contact terminals of the test fixtures (DUT connection) can be either 2T or 4T which are respectively suited to
different applications. The DUT connection configuration and suitable application of Keysight’s test fixtures are
summarized in Table 3-1. The advantages and disadvantages of 2T and 4T test fixtures are detailed in Appendix A.
Note: The meaning of “DUT connection configuration” in this paragraph differs from that of the terminal
configuration in Section 3.1. While the terminal configuration mainly refers to the cabling methods, the DUT
connection configuration describes the particular configuration of test fixture’s contact terminals. The test
fixtures are classified into the groups of 2T and 4T fixtures by the DUT connection configuration as shown in
Table 3-1.
Table 3-1. DUT connection conigurations of Keysight test ixtures and their
characteristics
DUT connection
coniguration
Applicable
device type
Keysight
test ixture
2-terminal
Leaded device
16047D
16047E
16065A
42842A/B/C
SMD
Surface mounted
device)
16034E
16034G
16034H
16334A
Material
16451B
16452A
In-circuit device
42941A
Leaded device
16047A
16089A/B/C/D/E
SMD
(Surface mounted
device)
16044A
4-terminal
Basic characteristics
Suitable applications
– Measurement is
Impedance: Middle and high
susceptible to the
effect of residual
Frequency: High
impedance and contact
resistance
– Usable frequency limit
is high
– Additional error at high
frequencies is smaller
than in 4-terminal
connection
– Measurement is less
Impedance: Low and middle
affected by residual
impedance and contact Frequency: Low
resistance (at relatively
low frequencies)
– Usable frequency limit
is low
– Additional error at high
frequencies is greater
than in 2-terminal
connection
3.2.2
User-fabricated test fixtures
If the DUT is not applicable to Keysight-supplied test fixtures, create an application-specific test fixture. Key points to
consider when fabricating a test fixture are:
(1)
(2)
(3)
Residuals must be minimized. To minimize the residuals, the 4TP configuration should be maintained as close
as possible to the DUT. Also, proper guarding techniques will eliminate the effects of stray capacitance. For
details, refer to “Practical guarding techniques” in Section 3.4.
Contact resistance must be minimized. Contact resistance will cause additional error. In the case of the 2T
configuration, it directly affects the measurement result. The contact electrodes should hold the DUT firmly and
should always be clean. Use a corrosion-free material for the electrodes.
Contacts must be able to be opened and shorted. Open/short compensation can easily reduce the effects of
the test fixture's residuals. To perform an open measurement, the contact electrodes should be located the
same distance apart as when the DUT is connected. For the short measurement, a lossless (low impedance)
conductor should be connected between the electrodes, or the contact electrodes should be directly
interconnected. If the 4T configuration is kept to the electrodes, make the connections of current and potential
terminals, and then make an open or short as shown in Figure 3-9.
Hc
DU T
Lc
Contact
electrodes
Lp
Hp
DUT electrodes
(a) DUT connection
Hc
Lc
Hp
Lp
Hc
Lc
OR
Hp
Lp
Low loss
conductors
(b) OPEN measurement
Hc
Lc
Hp
Lp
Hc
Lc
OR
Hp
Lp
Low loss
conductor
(c) SHORT measurement
Figure 3-9. User-fabricated test fixture open/short methods
3.2.3
User test fixture example
Figure 3-10 shows an example of a user-fabricated test fixture. It is equipped with alligator clips as the contact
electrodes for flexibility in making a connection to DUTs. Also, this test fixture can be connected directly to 4TP
instruments.
Figure 3-10. Example of fixture fabrication
3.3
Test cables
When the DUT is tested apart from the instrument, it is necessary to extend the test ports (UNKNOWN terminals)
using cables. If the cables are extended without regard to their length, it will cause not only a measurement error, but
will also result in bridge unbalance making measurement impossible. This section provides a guideline for choosing
or fabricating test cables.
3.3.1
Keysight-supplied test cables
Keysight Technologies supplies 1, 2, and 4 m cables as listed in Table 3-2. The Keysight 16048A and 16048E test
leads are manufactured using the same cable material. The Keysight 16048G and 16048H test leads employ a highquality cable to insure low-loss transmission characteristics that specifically match the E4990A, and the 4294A. The
cable length and the usable frequency range must be considered when selecting a test cable. Keysight's instruments
can minimize additional measurement errors because the characteristic of Keysight's test cables are known. Though
the cable compensation function is effective for Keysight-supplied test cables, the measurement inaccuracy will
increase according to the cable length and the measurement frequency.
Table 3-2. Keysight-supplied test cable
Test cable
Cable length
Maximum frequency
Connector type
Applicable instruments
16048A
1m
30 MHz
BNC
4263B, 4268A, 4284A, 4285A,
4288A, E4980A/AL, E4981A
SMC
16048-65000
16048D
2m
30 MHz
16048E
4m
2 MHz
4263B, 4284A, E4980A/AL
16048G
1m
120 MHz
110 MHz
E4990A
4294A
16048H
2m
120 MHz
110 MHz
E4990A
4294A
BNC
4263B, 4268A, 4284A, 4285A,
4288A, E4980A/AL, E4981A
3.3.2
User-fabricated test cables
Using cables other than those supplied by Keysight is not recommended. The cable compensation function of the
instrument may not work properly in non-Keysight cables. If there is an unavoidable need to use non-Keysight
cables, then employ the cable equivalent to Keysight test cables. The Keysight part number of the cable used for
frequencies below 30 MHz is 8121-1218 (not applicable to the E4990A.) Electrical specifications for these cables are
provided in Figure 3-11. Do not use test cables other than Keysight-supplied cables for higher frequencies.
To extend the cables using the 4TP configuration, the cable length should be adapted to the instrument’s cable
length correction function (1 m, 2 m, or other selectable cable length.) An error in the cable length will cause
additional measurement error. A detailed discussion on the cable extension is provided in Section 3.3.3 and in
Section 4.
Figure 3-11. Specifications of recommended cable (Keysight part number 8121-1218)
3.3.3
Test cable extension
If the required test cable is longer than 1, 2, or 4 m, it is possible to extend the Keysight-supplied test cable by using
the following techniques.
4TP-4TP extension
As shown in Figure 3-12 (a), all the outer shielding conductors are interconnected at far ends of the extension cables.
Actual connection can be made using four BNC (f) to BNC (f) adapters (Keysight part number 1250-0080 x 4) as
illustrated in Figure 3-12 (b). It is recommended that the BNC adapters be held in place with an insulation plate to
keep the adapters isolated (so as to not break the 4TP configuration.)
Note: If a conductive plate is used to hold the BNC adapters (without inserting insulators between the BNC
adapters and the plate), the 4TP configuration is terminated at the plate and the return current does not flow through
the extension cables.
Although this technique can provide the best accuracy, especially for low impedance measurement, the extension
length is limited by the measurement frequency. This is because the total length of the series cables must be
sufficiently shorter than the wavelength of the measurement signal. The following equation gives a guideline for
determining typical cable length limitation:
F (MHz) x L (m) ≤ 15
F: Measurement frequency (MHz)
L: Cable length (m)
When the cable length is one meter, the maximum frequency limit will be approximately 15 MHz. If the cable length
or frequency exceeds this limit, the auto-balancing bridge may not balance. For higher frequency measurements or
longer extension, the shielded 2T extension technique, which is described next, should be used.
Note: The E4990A helps prevent the cable length limitation by terminating the test ports with the same impedance
as the characteristic impedance of specified test cables at high frequencies. However, the practical cable length limit
due to increase in measurement error still exists.
Note: Additional measurement error and the compensation regarding the 4TP-4TP extension are
described in Section 4.5.
LCR meter
Test cables
(16048A etc.)
Extended cables
Hc
Hp
DUT
Lp
Lc
(a) Schematic diagram
LCR meter
BNC (f ) – BNC (f ) adapters
(Part number: 1250-0080)
Plate for holding the adapters
Connect here
Hc
Test cables
Hp
Lp
Lc
BNC cables
(b) Connection image
Figure 3-12. 4TP-4TP extension
Shielded 2T extension
As shown in Figure 3-13, the 4TP configuration is terminated and the extension cables configure a modified 3T
(shielded 2T). The two outer shielding conductors are connected together at each end of the cable. This decreases
the magnetic field induced by the inner conductors. This technique is used in the higher frequency region, up to 15
MHz. The residual impedance of the cables will be directly added to the measurement result, but can be an
insignificant error source if the DUT’s impedance is greater than the impedance due to the residuals. For the actual
connection, a connector plate (Keysight part number 16032-60071) supplied with Keysight test cables can be used
as shown in Figure 3-13.
LCR meter
Test cables
(16048A etc.)
Extended cables
Hc
Hp
DUT
Lp
Lc
(a) Schematic diagram
LCR meter
Hc
Connector plate
(Part number: 16032-60071)
Test cables
Hp
Lp
Lc
Coaxial cables
(b) Connection image
Figure 3-13. Shielded 2T extension
Shielded 4T extension
The outer shielding conductors of coaxial cables are interconnected at each end of the cables, as shown in Figure
3-14. The shielded 4T extension can be used for accurate low-impedance measurements. However, when applied to
high-frequency measurements (typically above 3 MHz), this extension method produces greater measurement errors
than the shielded 2T extension because the error sources at high frequencies are complicated. The length of the
shielded 4T extension in the high frequency region should be made as short as possible.
LCR meter
Test cables
(16048A etc.)
Extended cables
Hc
Hp
DUT
Lp
Lc
(a) Schematic diagram
LCR meter
Hc
Connector plate
(Part number: 16032-60071)
Test cables
Hp
Lp
Lc
Coaxial cables
(b) Connection image
Figure 3-14. Shielded 4T extension
Table 3-3 summarizes the extension techniques and their applicable impedance/frequency range.
Table 3-3. Summary of cable extension
Typical measurement frequency
Measured impedance
100 kHz and below
Low
(Typically 100 Ω and below)
Medium
(Typically 100 Ω to 100 kΩ)
High
(Typically 100 kΩ and above)
Above 100 kHz
4TP - 4TP
4TP - 4TP
4TP - Shielded 4T
4TP - Shielded 2T
3.4
Practical guarding techniques
3.4.1
Measurement error due to stray capacitances
When the DUT is located near a conductor (for example, a metallic desktop) and a measurement signal is applied to
the DUT, a voltage difference will appear between the DUT and the nearby conductor. This creates stray
capacitances and allows the measurement signal to leak towards the conductor as shown in Figure 3-15 (a).
Unshielded portions of test leads also have stray capacitances to the conductor.
Signal leakage through the stray capacitance on the High side of the DUT will bypass the DUT by flowing through
the conductor and the stray capacitance on the Low side. The ammeter (I-V converter) on the Lc side measures the
sum of the DUT current and the additional leakage current caused by the stray capacitances. Thus, the effect of stray
capacitances results in measurement error. The stray capacitances produce greater measurement error for higher
impedance of DUT and at higher measurement frequencies.
Null
detector
Lc
Leakage
current
Nul l
detector
V
Lp
Hp
DUT
Stray capacitance
(a) Stray capacitance and leakage current
Figure 3-15. Guarding technique (1)
Hc
Lc
Connect to
guard
Leakage
current
V
Lp
Hp
DUT
No stray
capacitance!
Conductor
(e.g. desktop)
Hc
Insert a
shielding plate
Conductor
(e.g. desktop)
(b) Removing the stray capacitance
3.4.2
Guarding technique to remove stray capacitances
By inserting a shielding plate between the DUT and the conductor, and by connecting it to the guard terminal of the
instrument as shown in Figure 3-15 (b), the leakage current flow through the stray capacitances can be eliminated.
Since the Low side of the DUT has a potential of zero volts (virtual ground) equal to the guard potential, the voltage
difference that yields the stray capacitance on the Low side is extinguished. Basically, the guard terminal is the outer
shielding conductor of the test cables.
Note: If the conductor yielding the stray capacitances is isolated from the ground and is free of
noise, it may be directly connected to the guard terminal without using the additional
shielding plate. On the contrary, if the conductor has a noise potential, this method should be
avoided because noise current flows into the outer shielding conductor of test cables and may
measurements.
disturb
When a stray capacitance in parallel with the DUT is present between High and Low terminals, as shown in Figure
3-16 (a), it can be removed by inserting a shielding plate between the High and Low terminals and by connecting the
plate to the guard terminal (as shown in Figure 3-16 (b).)
Null
detector
Lc
Null
detector
V
Lp
Hp
Hc
Leakage
current
Lc
V
Lp
Hp
Hc
Insert a
shielding plate
Connect to
guard
Stray C
DU T
(a) Stray capacitance between
High and Low terminals
DUT
(b) Removing the stray capacitance
Figure 3-16. Guarding technique (2)
3.5
Terminal coniguration in RF region
RF impedance measuring instruments have a precision coaxial test port, which is actually a 2T configuration in
principle. The center conductor of the coaxial test port connector is active (High side) terminal and the outer
conductor is grounded Low side terminal, as shown in Figure 3-17. To measure the DUT, only the simplest 2T
connection configuration can be used. Residual inductance, residual resistance, stray capacitance, and stray
conductance of the test fixture will add to measurement results (before compensation.) Whether using the RF I-V
method or network analysis, RF impedance measurement has lower accuracy as the measured impedance differs
greater from 50 Ω.
Instrument inaccuracy, rather than the error factors in the 2T test fixture, primarily limits the measurement range.
The effect of residuals increases with frequency and narrows the measurable impedance range in very high
frequencies.
Figure 3-17. Coaxial test port circuit configuration
3.6
RF test ixtures
RF test fixtures are designed so that the lead length (electrical path length) between the DUT and the test port is
made as short as possible to minimize residuals. At frequencies typically below 100 MHz, measurement error due to
test fixture residuals is small compared to instrument error and is normally negligible after compensation is made.
But, especially when measuring low or high impedance close to the residual parameter values, variance in the
residuals of the test fixture will cause measurement repeatability problems. For example, when measuring a 1 nH
inductor (a very low inductance), a slight variance of 0.1 nH in residual inductance will produce a 10 percent
difference in the measured value. The variance in the residual, and resultant measurement instability, is dependent on
the accurate positioning of the DUT on the test fixture terminals. For repeatable measurements, RF test fixtures
should be able to precisely position the DUT across measurement terminals.
The test fixture residuals will have greater effects on measurements at higher frequencies (typically above 500 MHz)
and will narrow the practical measurement range. Therefore, the usable frequency range of the test fixture is limited
to the maximum frequency specified for each test fixture.
The measurement inaccuracy for the DUT is given by sum of the instrument’s inaccuracy and the test-fixture induced
errors. Because only the 2T measurement configuration is available, the compensation method is crucial for
optimizing measurement accuracy. The measurement error sources and compensation techniques are discussed in
Section 4.
Each test fixture has unique characteristics and different structures. Since not only the residuals but also the
surrounding conditions of the DUT (such as ground plate, terminal layout, dielectric constant of insulator, etc.)
influence the measured values of the DUTs, the same type of test fixture should be used to achieve good
measurement correlation.
3.6.1
Keysight-supplied RF test fixtures
Keysight Technologies offers various types of RF test fixtures that meet the type of the DUT and required test
frequency range. Consider measurable DUT size, electrode type, frequency, and bias condition to select a suitable
test fixture.
There are two types of RF test fixtures: coaxial and non-coaxial test fixtures, which are different from each other in
geometrical structures and electrical characteristics. As the non-coaxial test fixture has open-air measurement
terminals as shown in Figure 3-18 (a), it features ease of connecting and disconnecting DUTs. The non-coaxial type is
suitable for testing a large number of devices efficiently. Trading off the benefit of measurement efficiency, the
measurement accuracy tends to be sacrificed at high frequencies because discontinuity (miss-match) in electrical
characteristics exists between the coaxial connector part and the measurement terminals. The coaxial test fixture holds
DUTs using a similar configuration to the coaxial terminations, as shown in Figure 3-18 (b). The DUT is connected
across the center electrode and the outer conductor cap electrode of the test fixture. With 50 Ω characteristic
impedance continuously maintained from test port to the DUT, the coaxial test fixture provides the best measurement
accuracy and the best frequency response. As the diameter of its replaceable insulator can be selected to minimize
the gap between the DUT and the insulator, the DUT can be positioned with a good repeatability across the test
fixture’s terminals independently of operator skill. The coaxial test fixture ensures less additional errors and much
better measurement repeatability than the non-coaxial test fixtures.
Figure 3-18. Types of RF impedance test fixtures
3.7
Test port extension in RF region
In RF measurements, connect the DUT closely to the test port to minimize additional measurement
errors. When there is an unavoidable need for extending the test port, such as in-circuit testing of devices
and on-wafer device measurement using a prober, make the length of test port extension as short as
possible. If the instrument has a detachable test head, it is better for accuracy to place the test head
near the DUT in order to minimize the test port extension length, and interconnect the test head and the
instrument using coaxial cables. (Observe the limit of maximum interconnection cable length specified for
the instrument.) Using a long test port extension will involve large residual impedance and admittance of
the extension cable in the measurement results, and significantly deteriorate the accuracy even if
calibration and compensation are completed.
Figure 3-19 shows an equivalent circuit model of the port extension. The inductance (Lo), resistance (Ro),
capacitance (Co), and conductance (Go) represent the equivalent circuit parameter values of the
extension cable. When the DUT’s impedance (Zx) is nearly 50 Ω, the test signal is mostly fed to the DUT
as the cable causes only a phase shift and (relatively small) propagation loss like a transmission line
terminated with its characteristic impedance. However, most likely the DUTs have a different value from
50 Ω. If the impedance of the DUT is greater than that of Co, the test signal current mainly bypasses
through Co, flowing only a little through the DUT. Conversely, if the impedance of the DUT is lower than
that of Lo and Ro, the test signal voltage decreases by a voltage drop across the cable and is applied only
a little to the DUT. As a result, the cable residuals lead to measurement inaccuracy and instability,
particularly, in high-impedance and low-impedance measurements. As illustrated in Figure 3-19, the Lo,
Ro, Co, and Go not only get involved in the measurement results (before compensation), but also affect
measurement sensitivity. Note that the measurable impedance range becomes narrow due to port
extension even though the calibration and compensation have been performed appropriately.
Figure 3-19. Calibration plane extension
In addition, electrical length of the extension cable will vary with environmental temperature, causing phase
measurement instability. Using longer extension makes measurement results more susceptible to the influence of
environmental temperature changes. Bending the cable will also cause variance in measured phase angle,
deteriorating measurement repeatability. Accordingly, in any application the port extension should be minimized.
The RF I-V and network analysis instruments commonly employ an N-type or 7-mm type coaxial connector as the
UNKNOWN terminal. Naturally, test port extension is made using a low-loss, electrically-stable coaxial transmission
line (cable) with 50 Ω characteristic impedance. When choosing the cable, consideration should be given to
temperature coefficients of propagation constants and rigidity to restrain the cable from easily bending. Figure 3-20
shows an example of the test fixture connected at the end of a 7 mm-7 mm connector cable. Calibration should be
performed first at the end of the extension before connecting to the test fixture. Next, the electrical length and open/
short compensations for the test fixture can be performed. (Alternatively, instead of the compensation, the open/
short/load calibration may be performed with working-standards connected at the test fixture's measurement
terminals. This method does not require the calibration at the end of the extension.) A detailed discussion on
measurement error sources, calibration, and compensation is provided in Section 4.
Figure 3-20. Practical calibration and compensation at extended test port
4.0
4.1
Measurement Error and Compensation
Measurement error
For real-world measurements, we have to assume that the measurement result always contains some error. Some
typical error sources are:
- Instrument inaccuracies (including DC bias inaccuracy, test signal level inaccuracy, and impedance
measurement inaccuracy)
- Residuals in the test fixture and cables
- Noise
The DUT’s parasitics are not included in the above list because they are a part of the DUT. The parasitics are the
cause of component dependency factors (described in Section 1.5) and dominate the real characteristics of
components. The objective of component measurement is to accurately determine the real value of a component
including parasitics. In order to know the real values of the DUTs, we need to minimize the measurement errors by
using proper measurement techniques. In the listed error sources, the residuals in the test fixture and test cables can
be compensated for if they are constant and stable.
4.2
Calibration
Calibration verifies instrument accuracy by comparing the instrument with "standard devices." To calibrate an
instrument, standard devices are connected at the calibration plane and the instrument is adjusted (through
computation/data storage) so that it measures within its specified accuracy. The calibration plane indicates the
electrical reference plane at which the standard devices are connected and measured. Accordingly, calibration
defines the calibration plane at which the specified measurement accuracy can be obtained.
The calibration plane of auto-balancing bridge instruments is at the UNKNOWN BNC connectors (see Figure 4-1.)
When the cable length correction is performed, the calibration reference plane moves to the tip of the test cables.
After an auto-balancing bridge instrument is shipped from the factory, calibration is usually required for maintenance
and service purposes. To maintain the instrument within the specified accuracy, calibration should be performed
periodically at the recommended calibration intervals (typically once a year.)
LCR meter
x [m ]
Side view
Side view
Calibration plane
(b) When cable length correction is performed for
Keysight test cables (x = 1, 2, and 4 [m])
Calibration plane
(a) Without cable extension
Figure 4-1. Calibration plane of auto-balancing bridge instruments
RF-IV instruments require calibration every time the instrument is powered on or every time the frequency setting is
changed. This is because ambient temperature, humidity, frequency setting, etc. have a much greater influence on
measurement accuracy than in low frequency impedance measurements. Calibration is performed using open, short,
and load reference terminations (a low loss capacitor termination is also used as necessary) as described in Section
4.7.1. The calibration plane is at the test port or the tip of test port extension where the calibration reference
terminations are connected (see Figure 4-2.)
Note: The calibration of the RF I-V instruments that should be performed prior to measurements eliminates
impedance measurement errors under the desired measurement conditions. The RF I-V instruments also
require periodic calibration at the recommended intervals for maintaining their overall operating performance
within specifications.
Impedance Analyzer
Open
Open
Short
Load
Short
Load
Calibration plane
Calibration plane
(a) Open/short/load (+ LLC) calibration at test port
(b) Open/short/load (+ LLC) calibration at the tip of a port
extension cable
Open
(No device)
Test fixture
Short
Load
Calibration plane
(c) Open/ short/ load calibration at DUT
contact terminals of a test fixture
Figure 4-2. Calibration plane of RF-IV instruments
LLC: Calibration using low loss
capacitor termination
4.3
Compensation
Compensation is also called correction and reduces the effects of the error sources existing between the DUT and
the instrument’s calibration plane. Compensation, however, can not always completely remove the error. Thus, the
measurement accuracy obtained after compensation is not as good as that obtained at the calibration plane.
Compensation is not the same as calibration and can not replace calibration. Compensation data is obtained by
measuring the test fixture residuals. The accuracy of compensation data depends on the calibration accuracy of the
instrument, so compensation must be performed after calibration has been completed.
Compensation improves the effective measurement accuracy when a test fixture, test leads, or an additional
measurement accessory (such as a component scanner) is used with the instrument. The following paragraphs
describe three commonly used compensation techniques:
- Offset compensation
- Open/short compensation
- Open/short/load compensation
Note: The open/short/load compensation for the auto-balancing bridge instrument (described in Section 4.3.3) is
not applied to RF-IV instruments because the compensation theory for the RF-IV method is different from
that for the auto-balancing bridge method.
4.3.1
Offset compensation
When a measurement is affected by only a single component of the residuals, the effective value can be corrected by
simply subtracting the error value from the measured value. For example, in the case of the low value capacitance
measurement shown in Figure 4-3, the stray capacitance (Co), paralleled with the DUT’s capacitance (Cx) is
significant to the measurement and can be removed by subtracting the stray capacitance value from the measured
capacitance value (Cxm). The stray capacitance value is obtained with the measurement terminals left open (Com).
Com
Cxm
LCR meter
LCR meter
Co
Co
DUT
Cx
Co = Com
Cx + Co = Cxm
Cx = Cxm - Com
Figure 4-3. Offset compensation
Cx: Corrected capacitance of the DUT
Cxm: Measured capacitance of the DUT
Co: Stray capacitance
Com: Measured stray capacitance
4.3.2
Open and short compensations
Open and short compensations are the most popular compensation technique used in recent LCR measurement
instruments. This method assumes that the residuals of the test fixture can be represented by the simple L/R/C/G
circuit as shown in Figure 4-4 (a). When the DUT contact terminals of the test fixture are open, as shown in Figure
4-4 (b), stray admittance Go + jωCo is measured as Yo because residual impedance (Zs) is negligible, (1/Yo >> Zs).
When the DUT contact terminals of the test fixture are shorted, as shown in Figure 4-4 (c), the measured impedance
represents residual impedance Zs = Rs + jωLs because Yo is bypassed. As a result, each residual parameter is known
and, the DUT’s impedance (Zdut) can be calculated from the equation given in Figure 4-4 (d).
Note: Keysight’s impedance measurement instruments actually use a slightly different equation. Refer to Appendix
B for more detailed information.
This compensation method can minimize the errors when the actual residual circuit matches the assumed model in
the specific situations listed below:
- Measurement by connecting a Keysight test fixture to the UNKNOWN terminals
- Measurement with a Keysight test fixture connected by a Keysight test cable that is compensated for
electrical length
In other situations, the open/short compensation will not thoroughly correct the measured values. In addition, this
method cannot correlate measurement results from different instruments. To resolve these compensation limitations,
the open/short/load compensation is required. Refer to “Open/short/load compensation” described in Section 4.3.3.
Test fixture residuals
Residual
Stray
impedance (Z s ) admittance (Yo )
Rs
Hc
Ls
Hp
Zm
Lp
Co
Go
Z du t
Lc
(a) Test fixture residuals
Rs
Hc
Ls
Rs
Hc
Hp
Hp
Yo
Lp
Co
Go
Open
Lp
Lc
Zs
Co
Go
Short
Lc
Z s = R s + jwL s
Yo = Go + jwC o
1
(R s + jwL s <<
)
Go + jwCo
SHORT impedance << Rs + jwL s
(b) Open measurement
(c) Short measurement
Zs
Hc
Hp
Lp
Ls
Z xm
Lc
Yo
Z dut
Z dut =
Z xm - Z s
1- (Z xm - Z s )Yo
Z dut : Corrected DUT impedance
Z xm : Measured DUT impedance
Yo : Stray admittance
Z s: Residual impedance
(d) Open/short compensation formula
.
Figure 4-4. Open/short compensation
Precautions for open and short measurements
Open measurement must be performed so that it accurately measures the stray capacitance. To do this, keep the
distance between the test fixture terminals the same as when they are holding the DUT. In addition, set the
integration time, averaging, and test signal level so that the instrument measures with maximum accuracy. If an open
measurement is performed under improper conditions, stray admittance (Yo) is not correctly measured, resulting in
an open compensation error.
Short measurement is performed by connecting the test fixture terminals directly together or by connecting a
shorting device to the terminals. The residual impedance of the shorting device should be much lower than the DUT’s
impedance, otherwise it will directly affect the measurement results. Figure 4-5 shows an example of a shorting
device that is applicable to the Keysight 16047A, and 16047D test fixtures. This shorting bar (Keysight part number
5000-4226) typically has residuals of 20 nH and 1 mΩ. Hence, the shorting bar is not suitable for low impedance
measurement. For very low impedance measurement, you should use a test fixture in which the fixture terminals can
be connected directly together.
Material: Brass (Ni-dipped)
Thickness: 1.0 mm
Residual impedance: 20 nH, 1 mΩ
t
o
g
e
Figure 4-5. Example of shorting device (Keysight part number 5000-4226)
t
h
e
r
.
4.3.3
Open/short/load compensation
There are numerous measurement conditions where complicated residual parameters cannot be modeled as the
simple equivalent circuit in Figure 4-4. Open/short/load compensation is an advanced compensation technique that
can be applied to complicated residual circuits. To carry out the open/short/load compensation, three measurements
are required before measuring the DUT, with the test fixture terminals opened, shorted, and with a reference DUT
(load) connected. These measurement results (data) are used for compensation calculation when the DUT is
undergoing measurement. As shown in Figure 4-6, the open/short/load compensation models the test fixture
residuals as a four-terminal network circuit represented by the ABCD parameters. Each parameter value is derived by
calculation if three conditions are known and if the four-terminal circuit is a linear circuit. The details of the
calculation method for the open/short/load compensation are described in Appendix C.
The open/short/load compensation should be used in the following situations:
(1)
(2)
(3)
(4)
(5)
(6)
An additional passive circuit or component (e.g. external DC bias circuit, balun transformer, attenuator and
filter) is connected.
A component scanner, multiplexer, or matrix switch is used.
Non-standard length test cables are used or 4TP extension cables are connected in series with Keysight test
cables.
An amplifier is used to enhance the test signal.
A component handler is used.
A custom-made test fixture is used.
In the cases listed above, open/short compensation will not work effectively and the measurement result contains
some error. It is not necessary to use the open/short/load compensation for simple measurement, like measuring an
axial leaded component using the Keysight 16047A test fixture. The open/short compensation is adequate for such
measurements.
I1
Measurement
instrument
V1
I2
AB
C D
V2
Unknown four-terminal
circuit
Open/short/load compensation formula
(Z s – Z xm ) (Z sm – Z o )
Z du t =
(Z xm – Z o ) (Z s – Z sm )
Figure 4-6. Open/short/load compensation
Z std
Z du t
DUT
Zdut: Corrected DUT impedance
Zxm: Measured DUT impedance
Zo: Measured open impedance
Zs: Measured short impedance
Zsm: Measured impedance of the load device
Zstd: True value of the load device
4.3.4
What should be used as the load?
The key point in open/short/load compensation is to select a load whose impedance value is accurately known. The
criteria is as follows.
Use a stable resistor or capacitor as the load device.
The load device’s impedance value must be stable under conditions of varying temperature, magnetic flux, and other
component dependency factors. So, avoid using inductors that are relatively sensitive to measurement conditions for
the load.
Use a load of the same size and measure it in the same way as the DUT will be measured.
As shown in Figure 4-7, if the load is measured under different electrode conditions, its measured data will not
effectively compensate for the residuals. It is a good idea to use one of the actual DUTs as a working standard. If the
load is a different type from the DUT (e.g. load is C and the DUT is R), at least keep the same distance between the
electrodes.
Use a load that is close in value to the DUT.
Whatever the load value is, the load compensation is effective over the entire measurement range if the
measurement circuit has a linear characteristic. In practice, the circuit between the UNKNOWN terminals and the
DUT may have a non-linear factor, especially when an additional circuit includes a non-linear component such as an
inductor, active switch, amplifier, etc. As shown in Figure 4-8, additional measurement error will be added when the
measured DUT value is far from the load value used for the compensation. So, the impedance value of the load
should be as close as possible to that of the DUT to be measured. If various impedances are to be measured, select a
load that is nearly the center value of the DUT’s impedance range. In addition, the load value should not be near the
open or short impedance. Otherwise, the load compensation will not be effective and the result of the open/short/
load compensation will be much the same as (or even worse than) that of the open/short compensation.
Use an accurately known load value.
The impedance value of the load must be known before performing the open/short/load compensation. To measure
the load value, it is practical to use the same measurement instrument, but under the best possible measurement
conditions. Set the measurement time, averaging, and test signal level so that the instrument can measure the load
with maximum accuracy. Also, use a test fixture that mounts directly to the instrument. Figure 4-9 shows an example
of such a measurement.
Figure 4-7. Electrode distance in load measurement
Figure 4-8. Load value must be close to the DUT’s value
Step 1: Using a direct-connected test
ixture, measure the load.
Step 2: Measure load compensation data using
ixture to be compensated.
Figure 4-9. Actual open/short load measurement example
4.3.5
Application limit for open, short, and load compensations
When the residuals are too significant compared to the DUT’s impedance value, compensation may not work
properly. For example, if the measured short impedance (Zsm) is about the same as the DUT’s impedance, total
measurement error will be doubled. The following are typical criteria for this limitation:
(1)
(2)
4.4
Measured open impedance (Zom) must be more than 100 times the measured impedance of the DUT.
Measured short impedance (Zsm) should be less than 1/100 of the measured impedance of the DUT.
Measurement error caused by contact resistance
Any contact resistance existing between the DUT electrodes and the contact electrodes of the test fixture or test
station will result in measurement error. The effects of the contact resistance are different for the DUT connection
methods, 2T and 4T. In the case of a 2T connection, the contact resistance is added to the DUT impedance in series
and produces a positive error in the dissipation factor (D) reading (see Figure 4-10 (a).) In the case of a 4T
connection, contact resistances Rhc, Rhp, Rlc, and Rlp exist as shown in Figure 4-10 (b.) The effects of the contact
resistance differ depending on the terminals. Rhc decreases the test signal level applied to the DUT, but it does not
directly produce measurement error. Rlp may cause the auto-balancing bridge to be unstable, but generally its effect
is negligible. Rhp and Chp (distributed capacitance of the coaxial test cable) form a low-pass filter, which causes
attenuation and phase shift of the Hp input signal, producing measurement error. Rlc and Clc also form a low-pass
filter and cause an error in measured DUT current and phase angle. Since the resultant dissipation factor error is
proportional to –ωRhp × Chp and –ωRlc × Clc, the D error is a negative value and increases with frequency. This error
becomes significant when the 4T connection method is used in high frequency measurements. The 4T connection
gives a constant D error (that is determined by the contact resistance and test lead capacitance only) while the error
of the 2T connection varies depending on the DUT’s value (Figure 4-10 (c).) The 4T connection provides minimal error
only when the effects of contact resistance and test lead capacitance are negligible (mainly at low frequencies.)
Low
High
R lc
R hc
C lc
Hc
Hp
RH
RL
Cx
Lc
Lp
Cx
C hp
R hp
R lp
Error caused by R lc with C lc
and R hp with C hp :
Error caused by R hp with C hp :
Magnitude error:
D erro r = wC x ( R L + R H )
1
-1
1 + w2 C hp2 R hp 2
D erro r = – w(C hp R hp + C lc R lc )
Phase error: wC hp R hp [rad]
(b) Contact resistance in four-terminal connection
rm
Tw
ote
D
err o r
in
al
co
nt
ac
t
(a) Contact resistance in twoterminal connection
Four-terminal contact
Chp
If R H = R L = R hp = R lc and C hp = C lc , D errors
of two-terminal and fou r-terminal contacts become the
same when C x = C hp .
This means that the two- terminal connection
is a better choice when the DUT capacitance
is smaller than cable capacitance (C hp or C lc ).
Capacitance value Cx
(c) Error of two-terminal and four-terminal connections
Figure 4-10. Effect of contact resistance
4.5
Measurement error induced by cable extension
4.5.1
Error induced by four-terminal pair (4TP) cable extension
A simplified schematic of test cable extension for the auto-balancing bridge instrument is shown in Figure 4-11.
Extending a 4TP measurement cable from the instrument will cause a magnitude error and phase shift of the
measurement signal according to the extension cable length and measurement frequency.
The following two problems will arise from the cable extension:
(1)
Bridge unbalanced
(2)
Error in the impedance measurement result
Bridge unbalance is caused by the phase shift in the feedback loop that includes the range resistor, (Rr), amplifier,
and the Lp and Lc cables. If the Lp or Lc cable is too long, it causes a significant change in phase angle of range
resistor current (IRr) flowing through the feedback loop. The vector current (IRr) cannot balance with the DUT current
vector because of the phase error and, as a result the unbalance current that flows into the Lp terminal is detected
by the unbalance detector (which annunciates the unbalance state to digital control section.) Some instruments such
as the Keysight E4990A can compensate for the effect of a long extension cable by producing an intentional phase
shift in the feedback loop.
The bridge unbalance is also caused by a standing wave (an effect of reflection) generated when the cable length is
not sufficiently shorter than the test signal wavelength. A guideline for the cable length limitation caused by this
effect is given by the following equation (as described in Section 3.3.3.)
F [MHz] x L [m] ≤ 15
The errors in impedance measurement results are mainly caused by the phase shift, attenuation, and reflection of
test signal on the cables connected to the Hp and Lc terminals. These errors can be corrected by the instrument if
the propagation constants and the length of the cable are known.
These two problems are critical only at high frequencies (typically above 100 kHz), and Keysight’s impedance measurement instruments can compensate for Keysight-supplied test cables. In the lower frequency region, the capacitance of the cable will only degrade the measurement accuracy without affecting the bridge balance. This effect of
the cable extension is shown in Figure 4-12.
Figure 4-11. Cable length correction
Cable
Source resistor
Cable
Hc
Lc
DUT
Hp
Range resistor
Lp
V
(a) Extended 4TP cables
Source resistor
High
Edut
DUT
Low
Range resistor
Rr
Z hg
Z lg
(Chc + Chp )
(C lc + C lp )
Zm = R r
Err
Edut
Err
Z hg : An impedance determined by the capacitance of the Hcur and Hpot
test cables. This impedance does not directly affect the measured
impedance (Zm) because a leakage current flow through Zhg does
not flow into the Low terminal (I-V converter.) However, it degrades
measurement accuracy by decreasing the test signal level.
Z lg : An impedance determined by the capacitance of the Lcur and Lpot
test cables. Unless very long cables are used, this impedance does not
cause significant error in the measured impedance (Zm), because the
potential across Zlg is zero volts when the bridge circuit is balanced.
(b) Effects of cable extension in low frequency region (≤ 100 kHz)
Figure 4-12. Measurement error due to extended cable length
The cable length correction works for test cables whose length and propagation constants are known, such as the
Keysight-supplied test cables of 1, 2, or 4 m. If different types of cable in different lengths are used, it may cause
bridge unbalance in addition to measurement error.
In practice, the measurement error is different for the cable termination types of the instrument. High frequency 4TP
instruments, such as the Keysight 4285A and the E4990A, which internally terminate cables with their characteristic
impedance, differ from low frequency 4TP instruments without cable termination.
4.5.2
Cable extension without termination
Extending test cable from the 4TP instrument without cable termination will produce an impedance measurement
error, which is typically given by the following equation:
Error = k × ΔL × f2 (%)
Where,
k: A coefficient specific to the instrument
ΔL: Cable length difference (m) from standard length (cable length setting)
f: Measurement frequency (MHz)
The k value is a decimal number mostly within the range of –1 to +1 and is different depending on instruments. As
the above equation shows, the error rapidly increases in proportion to the square of measurement frequency. Using
the open/short compensation will not reduce this error. Only the open/short/load compensation can minimize this
error.
4.5.3
Cable extension with termination
Extending the test cables from the instrument with cable termination will not produce a large error for the magnitude
of measured impedance (because the effect of reflections is decreased.) However, it causes a phase error in
proportion to the extension length and measurement frequency. (In practice, an error for the magnitude of
impedance also occurs because the actual cable termination is not ideal.) Performing the open/short/load
compensation at the end of the cable can eliminate this error.
4.5.4
Error induced by shielded 2T or shielded 4T cable extension
When the 4TP test cables and the shielded 2T (or shielded 4T) extension cables are connected in series as shown in
Figures 3-13 and 3-14, the cable length limitation and measurement error (discussed in Sections 4.4.2 and 4.4.3)
apply to the 4TP test cables only. The cable extension portion in the shielded 2T or shielded 4T configuration does
not cause the bridge unbalance, but produces additional impedance measurement error. There are some error
sources specific to the shielded 2T or shielded 4T configuration (as described in Sections 3.1.2 and 3.1.4) in the cable
extension portion. In this case, different compensation methods are applied to the 4TP test cables and the cable
extension portion, respectively.
Keysight-supplied test cables should be used in order to apply the cable length correction to the 4TP test cables. The
cable length correction moves the calibration plane to the tip of the 4TP test cables from the UNKNOWN terminals.
To minimize errors, perform the cable length correction for the Keysight test cables and then the open/short/load
compensation at the tip of extension cables. When the cable extension is sufficiently short and is used in the low
frequency region, the open/short compensation can be used in place of the open/short/load compensation. Note
that the cable length correction must be done to avoid the bridge unbalance caused by the phase shift of the
measurement signal in the 4TP test cables.
4.6
Practical compensation examples
The error sources present in a practical measurement setup are different for the configuration of test fixtures, test
cables, or circuits which may be connected between the instrument and the DUT. Appropriate compensation
methods need to be applied depending on the measurement configuration used. Figure 4-13 shows examples of the
compensation methods that should be used for typical measurement setups.
4.6.1
Keysight test fixture (direct attachment type)
When a Keysight direct attachment type test fixture is used, open/short compensation is enough to minimize the
additional measurement errors. Since the characteristics of Keysight test fixtures can be properly approximated by
the circuit model shown in Figure 4-4, the open/short compensation effectively removes the errors. Open/short/load
compensation is not required as long as the fundamental measurement setup is made as shown in Figure 4-13 (a).
4.6.2
Keysight test cables and Keysight test fixture
When Keysight test cables and a Keysight test fixture are connected in series as shown in Figure 4-13 (b), perform the
cable length correction first. The cable length correction moves the calibration plane to the tip of the test cables.
Then, perform the open/short compensation at the DUT terminals of the test fixture in order to minimize the test
fixture induced errors.
4.6.3
Keysight test cables and user-fabricated test fixture (or scanner)
When Keysight test cables and a user-fabricated test fixture are connected in series as shown in Figure 4-13 (c),
perform the cable length correction first in order to move the calibration plane to the tip of the test cables. The
characteristics of the user-fabricated test fixture are usually unknown. Thus, the open/short/load compensation
should be performed to effectively reduce the errors even if the test fixture has complicated residuals.
4.6.4
Non-Keysight test cable and user-fabricated test fixture
When a non-Keysight test cable and a user-fabricated test fixture is used, the 4TP measurement is basically limited to
the low frequency region. In the higher frequency region, this type of test configuration may produce complicated
measurement errors or, in the worst cases, cause the bridge unbalance which disables measurements. When
measurement setup is made as shown in Figure 4-13 (d), the cable length correction cannot be used because it will
not match the characteristics of the non-Keysight cables. As a result, the calibration reference plane stays at the
instrument’s UNKNOWN terminals (as shown in Figure 4-1 (a).) Initially, verify that the bridge unbalance does not
arise at the desired test frequencies. Next, perform the open/short/load compensation at the DUT terminals of the
test fixture. This method can comprehensively reduce measurement errors due to the test cables and fixture.
Instrument
Instrument
(1)
Keysight test fixture
(e.g. 16047E)
Keysight test cable
(16048X)
(1)
(2)
Keysight test fixture
(e.g. 16047E)
DUT
DUT
(1 ) Cable length correction
(1 ) Open/short compensation
(2 ) Open/short compensation
(a ) Keysight test fixture is used
(b ) Keysight test cable and
Keysight test fixture are used
Instrument
Instrument
Keysight test cable
(16048X )
Non-Keysight test cable
(1)
User-fabricated test
fixture or scanner
User-fabricated test
fixture or scanner
(2)
DUT
(1 ) Cable length correction
(1)
DUT
(1) Open/short/load compensation
(2 ) Open/short/load compensation
(c ) Keysight test cable and userfabricated test fixture are used
Figure 4-13. Compensation examples
(d ) Non-Keysight test cable and userfabricated test fixture are used
4.7 Calibration and compensation in RF region
4.7.1
Calibration
Whether the RF I-V method or network analysis, the open, short, and load calibration minimizes instrument
inaccuracies. To perform calibration, open, short, and load reference terminations are connected to the test port and
each of the terminations is measured. This calibration data is stored in the instrument’s memory and used in the
calculation to remove the instrument errors. Impedance values of these reference terminations are indicated in both
vector impedance coordinates and a Smith chart in Figure 4-14.
Note: A 7-mm coaxial connector has a fringe capacitance of typically 0.082 pF when terminated with Open. This
fringe capacitance value has been memorized in the instrument and is used to calculate accurate open
calibration data.
Note: Open impedance is infinite,
so it is not shown in the graph.
(a) Vector impedance plane
(b) Smith chart
Figure 4-14. Calibration standard values
Though all three terminations are indispensable for calibration, the load termination impedance (50 Ω) is particularly
important for precise calibration and has a large influence on resultant measurement accuracy. The uncertainty of
the load termination impedance is represented by a circle that encloses the error vector (see Figure 4-14 (a).) The
uncertainty of its phase angle increases with frequency and becomes a considerable error factor, especially in
measurements of high Q (low ESR or low D) devices at high frequencies.
To improve accuracy for the high Q (low loss) measurement, the RF I-V measurement instrument can be calibrated
using a low loss capacitor (LLC) termination in addition to the open/short/load terminations. The LLC provides a
reference for calibration with respect to the 90°-phase component of impedance. As a result, the instrument can
measure high Q (low dissipation factor) devices more accurately than when basic open/short/load calibration is
performed. The LLC calibration takes place only in the high frequency range (typically above 300 MHz) because the
phase angle of the load impedance is accurate at relatively low frequencies.
When the test port is extended, calibration should be performed at the end of extension cable, as discussed in
Section 3. Thereby, the calibration plane is moved to the end of cable.
To perform measurements to meet specified accuracy, the instrument should be calibrated before the measurement
is initiated and each time the frequency setting is changed. The calibration defines the calibration reference plane at
which measurement accuracy is optimized.
If a component could be measured directly at the calibration plane, it would be possible to obtain measured values
within the specified accuracy of the instrument. However, the real-world components cannot be connected directly
to the calibrated test port and a suitable test fixture is used for measurements. Calibration is not enough to measure
the DUT accurately. Because measurement is made for the DUT connected at the contact terminals of the test fixture
(different from calibration plane), the residual impedance, stray admittance, and electrical length that exist between
the calibration plane and the DUT will produce additional measurement errors. As a result, compensation is required
to minimize those test fixture induced errors.
4.7.2
Error source model
Regarding ordinary, non-coaxial test fixtures, consider an error source model similar to that in low frequency
measurements. Figure 4-15 (a) illustrates a typical test fixture configuration and a model of error sources. The test
fixture is configured with two electrically different sections: a coaxial connector section and a non-coaxial terminal
section for connecting the DUT. The characteristic of the coaxial section can be modeled using an equivalent
transmission line (distributed constant circuit) and represented by propagation constants. Normally, as the coaxial
section is short enough to neglect the propagation loss, we can assume that only the phase shift (error) expressed as
electrical length exists. The characteristic of the non-coaxial section can be described using the residual impedance
and stray admittance model in a two-terminal measurement configuration as shown in Figure 4-15 (b). We can
assume residual impedance (Zs) is in series with the DUT and stray admittance (Yo) is in parallel with DUT.
Figure 4-15. Typical error source model
4.7.3
Compensation method
As the error source model is different for the coaxial and non-coaxial sections of the test fixture, the compensation
method is also different.
Electrical length compensation eliminates measurement errors induced by the phase shift in the coaxial section.
Keysight RF impedance analyzers and RF LCR meters facilitate the electrical length compensation by allowing you to
choose the model number of the desired test fixture from among the displayed list, instead of entering the specified
electrical length of that test fixture to the instrument. (It is also possible to input the specified electrical length value.)
Open/short compensation is effective for residuals in the non-coaxial section. It is based on the same compensation
theory as described for low frequency measurements. (Refer to Section 4.3.2 for details.) The Yo and Zs can be
determined by measuring with the contact terminals opened and shorted, respectively.
As the test fixture is configured with the coaxial and non-coaxial sections, both compensations are required to
minimize combined errors. Load compensation is not required for normal measurements using Keysight-supplied test
fixtures.
When a test port extension or a user-fabricated test fixture is used, error sources will not match the model assumed
for the open/short compensation and they will affect measurement results. In such cases that measurement errors
cannot be removed sufficiently, consider attempting the open/short/load compensation. Actually, the open/short/
load compensation is substituted by the open/short/load calibration using working-standard devices because these
two functions are equivalent to each other. Note that when the open/short/load calibration is executed at
measurement terminals, the test port calibration data is invalidated (because the calibration plane is moved.)
Consequently, measurement accuracy depends on the calibrated accuracy of the short and load working-standard
devices (open calibration requires no device) as well as the proper contact when these standard devices are inserted
into the test fixture. It is important that special consideration be given to the precision of the standard values, contact
resistance, and positioning of the standard device on the test fixture.
4.7.4
Precautions for open and short measurements in RF region
To discuss calibration and compensation issues, we need to consider how residual parameters have large effects on
measurement results at high frequencies.
Assume that, for example, a residual inductance of 0.1 nH and a stray capacitance of 0.1 pF exist around the
measurement terminals of the test fixture. Notice how the effects of these small residuals differ depending on
frequency. Relationships of the residual parameter values to the typical impedance measurement range are
graphically shown in Figure 4-16. In the low frequency region, the residual parameter values are much smaller than
the values of normally measured devices. It is because the capacitors and inductors, which are designed for use in
low frequency electronic equipment, possess large values compared to small residuals. In the high frequency region,
however, devices such as those employed for higher frequency circuits and equipment have lower values. In the
frequency range typically above 100 MHz, the majority of the DUTs are low value devices (in the low nanohenries and
the low picofarads) and their values come close to the values of the residuals.
Accordingly, the residual parameters have greater effects on higher frequency measurements and become a primary
factor of measurement errors. The accuracy of measurement results after compensation depends on how precisely
the open/short measurements have been performed.
Figure 4-16. Relationship of residual parameter values to the typical impedance measurement range of the RF I-V method
To perform optimum compensation, observe the precautions for open/short measurements described in Section
4.3.2. In the high frequency region, the method of open/short compensation dominates the measurement correlation.
To obtain measurement results with a good correlation and repeatability, the compensation must be performed with
the same conditions. A difference in the compensation method will result in a difference in measured values, leading
to correlation problems on measurement results. Short measurement is more critical in terms of increasing the need
for low inductance measurements.
4.7.5
Consideration for short compensation
To make the short measurement at the contact terminals of a test fixture or of a component handler, a short bar
(chip) is usually employed. When measuring very low impedance (inductance), the following problems arise from the
short bar:
- Different residual impedance is dependent on size and shape
- Method of defining the residual impedance
If a different size or shape of the short bar is used, it is difficult to obtain a good correlation of the measurement
results. The residual impedance of the short bar is different if the size differs. Hence, the same size of short bar must
be used when making the short measurement.
If the definition of the short bar’s impedance is different, it causes a difference in measured values. To have a good
correlation, it is desirable to determine the short bar’s residuals. However, it cannot be determined only from the
inherent impedance of the short bar itself. The actual impedance depends on surrounding conditions such as contact
terminals, thickness of the closely located conductors, permittivity of insulators, ground conditions, etc.
Conceptually, there are two methods for defining the short bar’s impedance. One is to assume the impedance is zero.
This has been a primordial method of defining the short impedance. In this definition method, the measurement
result is a relative value of the DUT to the short bar. The other method is to define the short bar’s inductance as xx H.
(Residual resistance is negligible for a small short bar.) In this method, the measurement result is deemed as the
absolute value of the DUT. The residual inductance of the short bar is estimated from physical parameters (size and
shape) and is used as a reference. To estimate the inductance, the short bar needs to meet conditions, where
theoretical derivation is possible.
The measurement results from both definition methods are correct. The difference in the measurement result is
attributable to the difference in the definition. Practically, because of these incompatible definitions, a problem will
emerge when yielding correlation. To avoid this type of problem, it is necessary to establish an agreement on the
short bar’s size, shape, and the definition method of the residual inductance.
Note: Each of the Keysight 16196A/B/C/D coaxial test fixtures has a short device whose value is theoretically
definable. Since a 50 Ω coaxial configuration is established for the whole signal flow path, including the short
device placed in the fixture, the theoretical inductance value of the short device can be calculated from the
length and physical constants by using a transmission line formula. Its reference value is documented;
however, the use of the 16196A/B/C/D is not subject to the execution of the compensation based on the
reference value. You need to select the definition method of short inductance that agrees with your
measurement needs.
The chip-type short devices and load devices are readily available from the working-standard set supplied for
Keysight RF I-V measurement instruments. Otherwise, you can substitute appropriate devices for the short and load
chips by accurately determining (or properly defining) their characteristics.
4.7.6
Calibrating load device
To determine the values of a load device, you can use the same instrument that will be used to measure the DUTs.
The appropriate procedure for calibrating the load device is described below:
(1)
(2)
(3)
(4)
Perform open/short/load calibration at the instrument’s test port. In addition, for a capacitive or an inductive
load device, it is recommended that low loss capacitor calibration be performed.
Connect a direct-mounting type test fixture to the test port. It is recommended that the 16196A/B/C/D coaxial
test fixtures be used to insure the best measurement accuracy.
Perform open and short compensation. For short measurement, the method of minimizing short impedance
must be employed. (To do this, contact the terminals directly together if possible.) When the 16196A/B/C/D is
used, consider inputting the reference value of the residual inductance of the furnished short device to the
instrument. (Using the reference value is contingent upon how the reference of short inductance needs to be
defined for your measurement. Keysight chooses to take the historic approach to let Short = 0 H, but the actual
user of the test fixture can choose either approach.)
Connect the load device to the test fixture, select the parameters available for the instrument’s load calibration
function (typically R-X, L-Q, L-Rs, and C-D) and measure the device. Set the measurement time, test signal
level, and averaging so that the instrument can measure the load with maximum accuracy (or use the specified
test signal level of the device if required.)
4.7.7
Electrical length compensation
In the lower frequency region, using the open/short compensation function can minimize most of test fixture
residuals. In the RF region, however, this is not enough to reduce the effect of the test fixture residuals. The
wavelength of RF frequencies is short and is not negligible compared to the physical transmission line length of the
test fixture. So, a phase shift induced error will occur as a result of the test fixture, and this error cannot be reduced
by using open/short compensation. The phase shift can be compensated if the electrical length of the transmission
line is known. As shown in Figure 4-17, both the electrical length compensation and open/short compensation
should be performed after calibrating at the test port.
The electrical length compensation corrects phase error only and ignores propagation loss induced error. This is only
effective when transmission line (test port extension) is short enough to neglect the propagation loss.
Note: Theoretical explanation for the effects of the electrical length and the compensation is given in Appendix D.
Figure 4-17. Complete calibration and compensation procedure
4.7.8
Practical compensation technique
The calibration and compensation methods suitable for measurement are different depending on how the test cable
or fixture is connected to the test port. The following is a typical guideline for selecting appropriate calibration and
compensation methods.
Measurements using an Keysight test fixture without a test port extension
To make measurements using a test fixture connected directly to the test port, first perform calibration at the test
port. After calibration is completed, connect the test fixture to the test port and then perform electrical length
compensation (for the test fixture’s electrical length) and open/short compensation.
Measurement using a test port extension
When the measurement needs to be performed using a test port extension or a non-Keysight test fixture, it is
recommended that the open/short/load calibration be performed at the measurement terminals of the test fixture.
Typically, this method is applied when unknown devices are measured using a component handler. Because coaxial
terminations do not match geometrically with the contact terminals of the test fixture or of the component handler,
short and load devices whose values are defined or accurately known are required as substitution standards. (Open
calibration requires no device.) Compensation is not required because measurements are made at the calibration
plane.
4.8
Measurement correlation and repeatability
It is possible for different measurement results to be obtained for the same device when the same instrument and
test fixture is used. There are many possible causes for the measurement discrepancies, as well as residuals. Typical
factors for measurement discrepancies in RF impedance measurements are listed below.
-
Variance in residual parameter value
A difference in contact condition
A difference in open/short compensation conditions
Electromagnetic coupling with a conductor near the DUT
Variance in environmental temperature
4.8.1
Variance in residual parameter value
Effective residual impedance and stray capacitance vary depending on the position of the DUT
connected to the measurement terminals. Connecting the DUT to the tip of the terminals increases residual
inductance compared to when the DUT is at the bottom. Stray capacitance also varies with the position of the DUT
(see Figure 4-18.)
Figure 4-18. Difference in residual parameter values due to DUT positioning
4.8.2
A difference in contact condition
A change in the contact condition of the device also causes measurement discrepancies. When the device is
contacted straight across the measurement terminals, the distance of current flow between the contact points is
minimum, thus providing the lowest impedance measurement value. If the DUT tilts or slants, the distance of the
current flow increases, yielding an additional inductance between the contact points (see Figure 4-19.) Residual
resistance will also change depending on the contact points and produce a difference in measured D, Q, or R values.
The positioning error affects the measurement of low value inductors and worsens the repeatability of measured
values.
Figure 4-19. Measurement error caused by improper DUT positioning
4.8.3
A difference in open/short compensation conditions
Improper open/short measurements deteriorate the accuracy of compensated measurement results. If the open/
short measurement conditions are not always the same, inconsistent measurement values will result. Each short
device has its inherent impedance (inductance) value and, if not defined as zero or an appropriate value, the
difference of the short device used will produce resultant measurement discrepancies. Effective impedance of the
short device will vary depending on how it contacts to the measurement terminals. When the bottom-electrode test
fixture is used, contact points on the measurement terminals will be different from the case of the parallel-electrode
test fixture, as shown in Figure 4-20. If the short device is not straight (slightly curved), the measured impedance will
be different depending on which side of the device comes upside. These effects are usually small, but should be
taken into considerations especially when performing a very low inductance measurement, typically below 10 nH.
Figure 4-20. Difference in short impedance by test fixture types
4.8.4
Electromagnetic coupling with a conductor near the DUT
Electromagnetic coupling between the DUT and a metallic object near the DUT varies with mutual distance and
causes variance in measured values. Leakage flux generated around inductive DUT induces an eddy current in a
closely located metallic object. The eddy current suppresses the flux, decreasing the measured inductance and Q
factor values. The distance of the metallic object from the DUT is a factor of the eddy current strength as shown in
Figure 4-21 (a). As test fixtures contain metallic objects, this is an important cause of measurement discrepancies.
Open-flux-path inductors usually have directivity in generated leakage flux. As a result, measured values will vary
depending on the direction of the DUT. The difference in the eddy current due to the leakage flux directivity is
illustrated in Figures 4-21 (b), (c), and (d).
If a parasitic capacitance exists between the DUT and an external conductor, it is difficult to remove the effect on
measurement because the guarding technique is invalid. Thus, the DUT should be separated from the conductor
with enough distance to minimize measurement errors.
a) Metallic object near the leakage flux
of inductor will cause an eddy cur
rent effect. This effect increases as
the distance of the metallic object
from the inductor decreases.
c) Metallic object in parallel with the
leakage flux causes less eddy
current.
b) Inductor with leakage flux
directivity. Turning the inductor by
90 degrees reduces the eddy
current as shown in figure (c).
d) Inductor with less leakage flux
directivity. The eddy current effect is
almost independent from the
direction of inductor’s side faces.
Figure 4-21. Eddy current effect and magnetic flux directivity of device
4.8.5
Variance in environmental temperature
Temperature influences the electrical properties of materials used for the test fixtures and cables. When the test port
is extended using a coaxial cable, the dielectric constant of the insulation layer (between the inner and outer
conductors) of the cable, as well as physical cable length, will vary depending on the temperature. The effective
electrical length of the cable varies with the dielectric constants, thus resulting in measurement errors. Bending the
cable will also cause its effective electrical length to change. Keep the extension cable in the same position as it was
when calibration was performed.
5.0 Impedance Measurement Applications and Enhancements
Impedance measurement instruments are used for a wide variety of applications. In this section we present
fundamental measurement methods and techniques used to make accurate and consistent measurements for
various devices. Special measurement techniques, including the methods of enhancing the test signal level or DC
bias level, are also covered to expand the range of impedance measurement applications.
Capacitor measurement
Capacitors are one of the primary components used in electronic circuits. The basic structure of a capacitor is a
dielectric material sandwiched between two electrodes. The many available types of capacitors are classed according
to their dielectric types. Figure 5-1 shows the typical capacitance value ranges by the dielectric types of capacitors.
Table 5-1 lists the popular applications and features of the capacitors according to their dielectric classification.
Capacitance (C), dissipation factor (D), and equivalent series resistance, ESR, are the parameters generally
measured.
0.1 µF
1 nF
Dielectric type
5.1
4 0 0 µF
Plastic films
0.1 pF
10 0 µF
Ceramic
Mica
1 pF
6 pF
Air-dielectric
1 pF
1 nF
10 µF
2 5 nF
1 µF
Capacitance
Figure 5-1. Capacitance value by dielectric type
8 mF
Tantalum electrolytic
20 pF
0.5 F
Aluminum electrolytic
1 mF
1F
Table 5-1. Capacitor types
Type
Application
Advantage
Disadvantage
Film
– Blocking, buffering
bypass, coupling, and
iltering to medium
frequency
– Tuning and timing
– Wide range of capacitance and voltage
values
– High IR, low D, good Q
– Stable
– Low TC
– High voltage
– Medium cost
Mica
– Filtering, coupling,
and bypassing at high
frequencies
– Resonant circuit,
tuning
– High-voltage circuits
– Padding of larger
capacitors
– Low dielectric losses
and good temperature,
frequency, and aging
characteristics
– Low AC loss, high
frequency
– High IR
– Low cost
– Extensive test data,
reliable
– Low capacitance-to-volume ratio
Ceramic
– Bypassing, coupling,
and iltering to high
frequency
– High capacitance-to-volume ratio
– Chip style available
– Low D (low k type)
– Low cost
– Extensive test data,
reliable
– Poor temperature
coeficients and time
stability
– Large voltage dependency and susceptible
to pressure (high k
type)
Tantalum electrolytic
– Blocking, bypassing
coupling, and iltering
in low–frequency
circuits, timing, color
convergence circuits
squib iring, photolash
iring
– High capacitance-to-volume ratio
– Good temperature
coeficients
– Extensive test data
– Voltage limitation
– Leakage current
– Poor RF characteristics
– Medium cost
– Failure mode: short
Aluminum electrolytic
– Blocking, bypassing
coupling, and low
frequency iltering
Photolash
– Highest capacitance-to-volume ratio
of electrolytics
– Highest voltage of
electrolytics
– Highest capacitance
– Lowest cost per CV unit
for commercial types
– High ripple capability
– Affected by chlorinated
hydrocarbons
– High leakage current
– Requires reforming
after period of storage
– Poor RF characteristics
– Poor reliability
– Short life
5.1.1
Parasitics of a capacitor
A typical equivalent circuit for a capacitor is shown in Figure 5-2. In this circuit model, C denotes the main element of
the capacitor. Rs and L are the residual resistance and inductance existing in the lead wires and electrodes. Rp is a
parasitic resistance which represents the dielectric loss of the dielectric material.
C
L
Rs
Z = Rs +
Rp
Rp
+j
wL - w R p 2 C+ w3 R p 2 L C 2
1 + w2 R p 2 C 2
1 + w2 R p 2 C 2
Real part (R)
Imaginary part (X)
Figure 5-2. Capacitor equivalent circuit
Since real-world capacitors have complicated parasitics, when an impedance measuring instrument measures a
capacitor in either the series mode (Cs – D or Cs – Rs) or the parallel mode (Cp – D, Cp – G, or Cp – Rp), the
displayed capacitance value, Cs or Cp, is not always equal to the real capacitance value, C, of the capacitor. For
example, when the capacitor circuit shown in Figure 5-2 is measured using the Cs – Rs mode, the displayed
capacitance value, Cs, is expressed using the complicated equation shown in Figure 5-3. The Cs value is equal to the
C value only when the Rp value is sufficiently high (Rp >> 1/ωC) and the reactance of L is negligible (ωL << 1/ωC.)
Generally, the effects of L are seen in the higher frequency region where its inductive reactance, ωL, is not negligible.
The Rp is usually insignificant and can be disregarded in the cases of high-value capacitors (because Rp >> 1/ωC.)
For low-value capacitors, the Rp itself has an extremely high value. Therefore, most capacitors can be represented by
using a series C-R-L circuit model as shown in Figure 5-4. Figures 5-5 (a) and (b) show the typical impedance (|Z| _
θ) and Cs – D characteristics of ceramic capacitors, respectively. The existence of L can be recognized from the
resonance point seen in the higher frequency region.
Note: The relationship between typical capacitor frequency response and equivalent circuit model
is explained in Section 1.5.
C
L
–1
Rs
Cs =
Cs - Rs mode
Figure 5-3. Effects of parasitics in actual capacitance measurement
Figure 5-4. Practical capacitor equivalent circuit
–1 =
wX
Rp
1
C +
w2
R p2
C2
w2 L - w2 R p2 C + w4 R p2 L C 2
w2
=
1-
L
Rp2 C
CRp2
- w2 LC
(a) |Z| - θ characteristics
(b) C - D characteristics
Figure 5-5. Typical capacitor frequency response
5.1.2 Measurement techniques for high/low capacitance
Depending on the capacitance value of the DUT and the measurement frequency, you need to employ suitable
measurement techniques, as well as take necessary precautions against different measurement error sources.
High-value capacitance measurement
The high-value capacitance measurement is categorized in the low impedance measurement. Therefore, contact
resistance and residual impedance in the test fixture and cables must be minimized. Use a 4T, 5T, or 4TP
configuration to interconnect the DUT with the measurement instrument. When the 4T or 5T configuration is used,
the effects of electromagnetic field coupling due to a high test signal current flow through the current leads should
be taken into considerations. To minimize the coupling, twist the current leads together and the potential leads
together, as shown in Figure 5-6. Form a right angle (90°) between the current leads and potential leads connected
to DUT terminals.
Magnetic fields generated around
the test cables are canceled by
twisting the cables.
Figure 5-6. High-value capacitor measurement
Also, for an accurate measurement, open/short compensation should be properly performed. During the open/short
measurements (in the 4T or 5T configuration), maintain the same distance between the test cables as when the DUT
will be measured. For electrolytic capacitors, which require a DC bias voltage to be applied, the open/short compensation should be performed with the DC bias function set to ON (0 V bias output.)
The component dependency factors discussed in Section 1 should be taken into account, especially when measuring
high-value ceramic capacitors. The high-value ceramic capacitors exhibit a large dependence on frequency, test signal voltage (AC), DC bias, temperature, and time.
Low-value capacitance measurement
The low-value capacitance measurement is categorized in the high impedance measurement. Stray capacitance
between the contact electrodes of a test fixture is a significant error factor compared to the residual impedance. To
make interconnections with the DUT, use a 3T (shielded 2T), 5T (shielded 4T), or 4TP configuration. Proper guarding
techniques and the open/short compensation can minimize the effects of stray capacitance (refer to Section 3.4.)
Figure 5-7 shows the typical procedure for performing the open/short compensation when measuring SMD (chiptype) capacitors with the Keysight 16034E/G test fixtures.
Figure 5-7. Low-value chip capacitor measurement
Other than capacitance, important capacitor parameters are the dissipation factor, D, and the ESR. Special
precautions must be taken in the low D or low ESR measurements. Contact resistance and residual impedance in the
test fixture and cables will affect the measurement results even when the 4T configuration is used (refer to Section 4.)
DC biased capacitance measurement
The DC biased capacitance measurement can be performed using the internal DC bias function of an impedance
measuring instrument, or an external bias fixture for applying a bias voltage from an external DC source. When the
DC bias voltage is changed, a bias settling time needs to be taken until the capacitor is charged by the applied bias
voltage. The required bias settling time increases in proportion to the capacitance of the DUT. Accordingly, to
perform an accurate bias sweep measurement for a high-value capacitor, it is necessary to insert a delay time
between the step-up (or the step-down) of bias voltage and measurement trigger for each sweep measurement
point. The required bias settling time can be obtained from DC bias performance data of the instrument or bias
fixture used.
5.1.3
Causes of negative D problem
When measuring the dissipation factor (D) of a low loss capacitor, the impedance measuring instrument may
sometimes display a negative D value despite the fact that the real dissipation factor must be a positive value. A
negative D measurement value arises from a measurement error for a small resistance component of the measured
impedance. In this section, we discuss the causes of negative D and the methods for minimizing the measurement
errors that lead to the negative D problem. Five typical causes of negative D problem are:
Note:
Instrument inaccuracy
Contact resistance in the 4TP or 5T configuration
Improper short compensation
Improper cable length correction
Complicated residuals
The following discussion also applies to a negative Q problem because the Q factor is the
reciprocal of D.
D measurement error due to instrument inaccuracy
If a DUT has a low D value compared with the D measurement accuracy (allowable D measurement error) of the
instrument, a measured dissipation factor may become a negative value. Figure 5-8 shows how the D measurement
accuracy of instrument impacts a negative D value. For example, when D measurement accuracy (of instrument A) is
±0.001 for a low-loss capacitor that has a dissipation factor of 0.0008, the impedance measurement error is
represented by a dotted circle on the vector plane as shown in Figure 5-8. The shaded area of the dotted circle exists
on the left side of reactance axis (X axis.) This shaded area represents the negative D area in which the resistance
component of the measured impedance is a negative value. The allowable D value range is from –0.0002 to 0.0018.
In this case, there is possibility that a negative D value is displayed. If the D measurement accuracy (of instrument B)
is ±0.0005, the measured impedance vector is within the solid circle as shown in Figure 5-8. The negative D value is
not displayed because the allowable D value range is from 0.0003 to 0.0013. Accordingly, an impedance measuring
instrument with the best possible accuracy is required for avoiding negative D display in low dissipation factor
measurements.
R
Example:
Impedance
vector
D = 0.0008
(at specific measurement conditions)
X
Instrument
0.0005 accuracy
Negative D
0.001 accuracy
Figure 5-8. Negative D measurement value due to measurement inaccuracy
D accuracy
Possible readout
A
± 0.001
– 0.0002 to 0.0018
B
± 0.0005
0.0003 to 0.0013
Contact resistance
As described in Section 4.4, contact resistance between the DUT’s electrodes and the contact electrodes of the test
fixture causes D measurement error. While the contact resistance of the 2T test fixture directly adds to the measured
impedance as a positive D error, the contact resistance at the Hp and Lc electrodes of a 4T test fixture cause a
negative D error (see Figure 4-10.) When a capacitor that has a very low D is measured using a 4T test fixture, a
negative D value is displayed depending on the magnitude of the D measurement error due to a contact resistance.
Improper short compensation
When short compensation is performed based on an improper short measurement value, a negative D value may be
displayed. Major causes of an improper short measurement are a contact resistance at the test fixture’s electrodes
and a residual resistance of the shorting bar. As described in
Section 4.3, the resistance (Rs) and reactance (Xs) values obtained by short measurement are stored in the
instrument and removed from the measured impedance of the DUT by performing the short compensation. If the Rs
value is greater than the resistance component (Rxm) of the DUT’s impedance, the corrected resistance (Rxm – Rs)
becomes a negative value and, as a result, a negative D value is displayed. To avoid this problem, clean the test
fixture’s electrodes to minimize the contact resistance and use a shorting bar with the lowest possible residual
resistance.
Improper cable length correction
When cable length correction is not properly performed for the test cables used, a negative D value may be displayed
at high frequencies because a phase angle measurement error is caused by the cables. The error increases in
proportion to the square of the measurement frequency. After the cable length correction is performed, a small phase
error may remain and cause a negative D value because the characteristics of test cables are slightly different for the
respective cables. The open/short/load compensation can minimize the measurement error due to the differences
between the cables.
Complicated residuals
Using a long cable, a component scanner, or a component handler has the propensity to cause a negative D display
due to complicated residuals. When complex residual impedance and stray admittance exist in the connection circuit
between the DUT and the calibration plane of the impedance measuring instrument, the characteristics of the
connection circuit do not match the open/short compensation circuit model (see Figure 4-4.) Since the open/short
compensation cannot effectively remove the measurement error due to the complex residuals and strays, a D
measurement error causes a negative D display. The open/short/load compensation is an effective method for
eliminating measurement errors caused by complicated residuals.
5.2 Inductor measurement
5.2.1
Paracitics of an inductor
An inductor consists of wire wound around a core and is characterized by the core material used. Air is the simplest
core material for making inductors, but for volumetric efficiency of the inductor, magnetic materials such as iron,
permalloy, and ferrites are commonly used. A typical equivalent circuit for an inductor is shown in Figure 5-9 (a). In
this figure, Rp represents the magnetic loss (which is called iron loss) of the inductor core, and Rs represents the
copper loss (resistance) of the wire. C is the distributed capacitance between the turns of wire. For small inductors
the equivalent circuit shown in Figure 5-9 (b) can be used. This is because the value of L is small and the stray
capacitance between the lead wires (or between the electrodes) becomes a significant factor.
Figure 5-9. Inductor equivalent circuit
Generally, inductors have many parasitics resulting from the complexity of the structure (coil) and the property of the
magnetic core materials. Since a complex equivalent circuit is required for representing the characteristics, which
include the effects of many parasitics, a simplified model for approximation is used for practical applications. In this
section, we discuss the frequency response of a low-value inductor, which is represented by equivalent circuit model
shown in Figure 5-9 (b). This model will fit for many SMD (chip) type RF inductors.
When the inductor circuit shown in Figure 5-10 is measured using the Ls-Rs mode, the measured
Ls value is expressed by the equation shown in Figure 5-11. The measured Ls value is equal to the
L value only when the inductor has low Rs value (Rs << ωL) and low C value (1/ωC >> ωL). Typical frequency
characteristics of impedance (|Z|_ θ) for a low-value inductor are shown in Figure 5-12 (a). Since the reactance (ωL)
decreases at lower frequencies, the minimum impedance is determined by the resistance (Rs) of winding. The stray
capacitance Cp is the prime cause of the inductor frequency response at high frequencies. The existence of Cp can
be recognized from the resonance point, SRF, in the higher frequency region. At the SRF, the inductor exhibits
maximum impedance because of parallel resonance (ωL = 1/ωCp) due to the Cp. After the resonance frequency, the
phase angle of impedance is a negative value around –90° because the capacitive reactance of Cp is dominant. The
inductor frequency response in Ls – Rs measurement mode is shown in Figure 5-12 (b). The measured inductance
(Lm) rapidly increases as the frequency approaches the SRF because of the effect of resonance. The maximum Lm
value becomes greater as the device has a higher Q factor. At frequencies above the SRF, a negative inductance
value is displayed because the Lm value is calculated from a capacitive reactance vector, which is opposite to
inductive vector.
C
L
Rs
L (1 –
Rs
Z=
(1 -
2 LC) 2
–
+
2C 2R 2
s
+ j
2C 2R 2
s
+
2 LC) 2
(1 -
Real part (R)
CR s2
)
L
2 LC
Imaginary part (X)
Figure 5-10. Inductor equivalent circuit
C
L
Rs
X
Ls =
w
Ls - R s mode
C R s2
L (1 – w2 LC –
L
=
)
(1 – w2 LC ) 2 + w2 C 2 R s 2
When w2 C 2 R s 2 << 1 and
C R s2
L
L
<< 1,
Ls `
1 – w2 LC
Figure 5-11. Effects of parasitics in actual inductor measurement
Cp
+L
Ls Rs
Inductive
Capacitive
90º
Log | Z |
θ
(θ)
Impedance/
phase angle
(Negative L m value )
Lm =
Inductance
0º
|Z |
Ls
1 - ω2 C p L s
Effe ctive range of
0
Equiva lent C =
–90º
SRF
Frequenc y
(a) |Z| - θ frequency response
Figure 5-12. Typical inductor frequency response
Log f
–L
SRF
Log f
Frequenc y
(b) Inductance frequency response
–1
1
= C p (1 )
ω2 L m
ω2 C p L s
5.2.2
Causes of measurement discrepancies for inductors
Inductance measurement sometimes gives different results when a DUT is measured using different instruments.
There are some factors of measurement discrepancies as described below:
Test signal current
Inductors with a magnetic core exhibit a test signal current dependency due to the nonlinear magnetization
characteristics of the core material as shown in Figure 5-13 (a). The level of test signal current depends on the
impedance measurement instrument because many of the instruments output a voltage-driven test signal. Even
when two different instruments are set to output the same test signal (OSC) voltage, their output currents are
different if their source resistance, Rs, is not the same as shown in Figure 5-13 (b).
To avoid the measurement discrepancies, the OSC level should be adjusted for a defined test current by using the
auto level control (ALC) function or by determining the appropriate test voltage setting from the equation shown in
Figure 5-13 (b).
Figure 5-13. Inductor test signal current
Test fixture used
When a metal object is located close to an inductor, leakage flux from the inductor will induce eddy currents in the
metal object. The magnitudes of the induced eddy currents are dependent on the dimensions and physical geometry
of metal object, as shown in Figure 5-14 (a), causing differences in the measured values. The eddy current effect is
especially important for measuring open-flux-path inductors. Figure 5-14 (b) shows an example of the difference in
Ls – Q measurement values due to the eddy current effect. When a 40 mm x 40 mm square and 1.0 mm thick brass
plate is placed closely to a 100 µH RF inductor, the measured Ls – Q values decrease according to the approach of
the plate from # (sufficient distance) to 10 mm and 1 mm. The eddy current effect due to the leakage flux causes
discrepancies in measurement results between different types of test fixtures because the test fixtures are also metal
objects. To obtain consistent measurement results, it is necessary to define the test fixture used for inductor
measurements. Additionally, the DUT should be connected at the same position of the same test fixture.
Figure 5-14. Test fixture effects
Q measurement accuracy
Generally, the Q-factor measurement accuracy in the impedance measurement is not high enough to measure the
high Q device. Figure 5-15 shows the relationship of Q accuracy and measured Q values. Because the Q value is the
reciprocal of D, (Q = 1/D), the Q accuracy is related to the specified D measurement accuracy as shown in Figure
5-15. The Q measurement error increases with the DUT’s Q value and, therefore, the practically measurable Q range
is limited by the allowable Q measurement error. (For example, if the allowable Q error is ten percent and if the
instrument’s D accuracy is ±0.001, the maximum measurable Q value is 90.9. (See Appendix E for the Q
measurement accuracy calculation equation.)
Figure 5-15. Q measurement accuracy
Figure 5-16 shows the measured vector of a high Q inductor. Except for the resonant method, the impedance
measurement instrument calculates the Q value by Q = X/R. The impedance measurement error is represented by a
small circle enclosing the error vector (Δ). The R value of a high Q (low loss) inductor is very small relative to the
X value. Small changes in R results in large Q value changes (Q = X/R). Therefore, error in the R measurement can
cause significant error in the Q factor, especially in high Q devices. A negative Q problem also arises from the Q (D)
measurement error as described in Section 5.1.3.
Figure 5-16. Q measurement error
The following methods deliver improvement to Q measurement accuracy:
(1) Use the instrument with better accuracy
(2) Perform optimum compensation for residual resistance and cable length
(3) Use an equivalent circuit analysis function and calculate the Q value from the equivalent circuit parameter values
obtained for the DUT (refer to Section 5.15.)
Furthermore, the following phenomena may occur when a cored inductor is measured using an auto-balancing
bridge type instrument.
When a high level test signal is applied to an inductor, measurement may be impossible for a certain
frequency range. This is because the nonlinearity of the core material causes harmonic distortion of the test signal
current, which hinders measurements. If excessive distortion current flows into the Lpot terminal of the instrument, it
causes the bridge unbalance status (see Figure 5-17 (a).) To reduce the effects of core material nonlinearity,
decrease the test signal level. If the measurement frequency is fixed, it is possible to reduce the distortion current
flow into the Lpot terminal by connecting a low-pass filter (LPF) at the Lpot terminal as shown in Figure 5-17 (b).
When a high level DC bias current is applied to an inductor, measurement may be impossible for a certain frequency
range. This is because test signal distortion is caused by the magnetic saturation of the inductor core under the
applied bias magnetic field. To reduce the effects of core material nonlinearity, take the same precautions as those
for measurement at a high test signal level.
When a test cable is used to measure low-value inductors, measurement may be impossible for certain
values of inductance at higher frequencies. This is caused by resonance resulting from the DUT’s inductance and the
capacitance of Hp and Hc cables. In this case, the capacitance of the cables should be changed so that the resonant
frequency shifts to a much higher frequency than the maximum test frequency required. Reduce the length of the Hc
and Hp cables or use another type of cable to decrease the capacitance.
Figure 5-17. Harmonic distortion caused by inductors
5.3
Transformer measurement
A transformer is one end-product of an inductor so, the measurement techniques are the same as those used for
inductor measurement. Figure 5-18 shows a schematic with the key measurement parameters of a transformer. This
section describes how to measure these parameters, including L, C, R, and M.
C
R1
R2
L 1: Primary inductance
L 2: Secondary inductance
C 1 , C 2 : Distributed capacitance of windings
C1
L1
L2
C2
R 1 , R2 : DC resistance of windings
C: Inter-winding capacitance
M: Mutual inductance
M
Figure 5-18. Transformer parameters
5.3.1
Primary inductance (L1) and secondary inductance (L2)
L1 and L2 can be measured directly by connecting the instrument as shown in Figure 5-19. All other windings should
be left open. Note that the inductance measurement result includes the effects of capacitance. If the equivalent
circuit analysis function of Keysight’s impedance analyzer is used, the individual values for inductance, resistance,
and capacitance can be obtained.
Leakage inductance is a self-inductance due to imperfect coupling of the transformer windings and resultant
creation of leakage flux. Obtain leakage inductance by shorting the secondary with the lowest possible impedance
and measuring the inductance of the primary as shown in Figure 5-20.
Figure 5-19. Primary inductance measurement
Figure 5-20. Leakage inductance measurement
5.3.2
Inter-winding capacitance (C)
The inter-winding capacitance between the primary and the secondary is measured by connecting one side of each
winding to the instrument as shown in Figure 5-21.
5.3.3
Mutual inductance (M)
Mutual inductance (M) can be obtained by using either of two measurement methods:
(1) The mutual inductance can be derived from the measured inductance in the series aiding and
the series opposing configurations (see Figure 5-22 (a).) Since the combined inductance (La) in
the series aiding connection is La = L1 + L2 + 2M and that Lo in the series opposing connection is
Lo = L1 + L2 – 2M, the mutual inductance is calculated as M = (La – Lo)/4.
(2) By connecting the transformer windings as shown in Figure 5-22 (b), the mutual inductance value is directly
obtained from inductance measurement. When test current (I) flows through the primary winding, the secondary
voltage is given by V = jωM x I. Therefore, the mutual inductance can be calculated from the ratio between the
secondary voltage (V) and the primary current (I.) However, the applicable frequency range of both measurement
techniques is limited by the type and the parameter values of the transformer being measured. These methods
assume that the stray capacitance effect, including the distributed capacitance of windings, inter-winding
capacitance, and test lead capacitance, is sufficiently small. To minimize the cable capacitance effect for the method
shown in Figure 5-22 (b), the Hp test lead length should be made as short as possible. It is recommend to use both
techniques and to cross-check the results.
Hc
Hp
Lp
Lc
C
Figure 5-21. Inter-winding capacitance measurement
Hc
M
Hp
L1
L2
L1
L2
Lp
La
L o = L 1 + L 2 - 2M
M=
La -Lo
(a ) Series aiding and series opposing
Figure 5-22. Mutual inductance measurement
Lc
Lo
L a = L 1 + L 2 + 2M
I
V
4
V = jωM I
M=
V
jωI
(b) Direct connection technique
5.3.4
Turns ratio (N)
Turns ratio (N) measurement technique, which can be used with general impedance measuring instruments,
approximates the turns ratio (N:1) by connecting a resistor to the secondary as shown in Figure 5-23 (a). From the
impedance value measured at the primary, the approximate turns ratio can then be calculated. Direct turns ratio
measurement can be made with a network analyzer or built-in transformer measurement function (option) of the
Keysight 4263B LCR meter. The turns ratio can be determined from the voltage ratio measurements for the primary
and the secondary, as shown in Figure 5-23 (b). The voltmeter (V2) should have high input impedance to avoid
affecting the secondary voltage. The properties of magnetic core and the effects of stray capacitance limit the
applicable frequency range of the turns ratio measurement methods.
N
:
1
Z
N
R
:
1
Z = N2 R
V1
N=
V2
Z
N=
R
(a) Approximation by measuring
impedance ratio
Figure 5-23. Turns ratio measurement
(b) Direct measurement by using voltage
ratio measurement
V1
V2
The 4263B’s transformer measurement function enables the measurement of the N, M, L1, and the DC resistance of
the primary by changing measurement circuit connections with an internal switch. Figure 5-24 shows a simplified
schematic block diagram for the transformer measurement function of the 4263B. A test signal is applied to the
primary and L1 is calculated from the measured values of V1 and I1. M is calculated from V2 and I1. N is obtained from
the ratio of V1 and V2. In the DC resistance measurement, the applied voltage at the Hcur terminal is DC. The DC
resistance value is calculated from measured DC voltage V1 and current I1.
Using the Keysight 16060A transformer test fixture with the 4263B permits the L2 and DC resistance measurement
for the secondary, along with all the parameters for the primary. The circuit connection diagram of the 16060A is
shown in Figure 5-25.
Figure 5-24. 4263B transformer measurement function schematic block diagram
Figure 5-25. 16060A circuit connection diagram
5.4
Diode measurement
The junction capacitance of a switching diode determines its switching speed and is dependent on the reverse DC
voltage applied to it. An internal bias source of the measurement instrument is used to reverse-bias the diode. The
junction capacitance is measured at the same time. Figure 5-26 shows the measurement setup.
For variable capacitance diodes (varactor diode) that use capacitance-bias characteristics, it is important to measure
capacitance accurately while applying an accurate DC bias voltage. Figure 5-27 shows an example of measuring the
C-V characteristics of a varactor diode. Use a low test signal level (typically 20 mV rms) to precisely trace the
relationship of the capacitance to the DC bias voltage.
The varactors for high frequency applications require Q factor or ESR measurement along with capacitance at a
frequency above 100 MHz. The RF I-V measurement instrument is adequate for this measurement. It is possible to
measure Q or ESR with the same setup as for the C-V measurement by merely selecting the desired parameter.
Figure 5-26. Reverse biased diode measurement setup
Figure 5-27. Varactor C-V characteristics
5.5
MOS FET measurement
Evaluating the capacitances between the source, drain, and gate of an MOS FET is important in the design of high
frequency and switching circuits. Generally, these capacitances are measured while a variable DC voltage source is
connected to the drain terminal referenced to the source, and the gate is held at zero DC potential (Figure 5-28).
When an instrument is equipped with a guard terminal and an internal DC bias source, capacitances Cds, Cgd, and
Cgs can be measured individually. Figures 5-29 (a) through (c) show the connection diagrams for an instrument’s
High, Low, and Guard terminals. The guard is the outer conductors of BNC connectors of the UNKNOWN terminals.
The E4980A, with Option E4980A-001 has an independent DC source in addition to an internal DC bias and allows
the Cgs measurement set up to be simplified as sown in Figure 5-29 (d).
Figure 5-28. Capacitance of MOS FET
Guard
L
DC source
Hc
High
Hp
Lp
C
Cgs
Low
Lc
Typical values (for 1 MHz measurement):
C: 1 F
L: 100 H
(d) Cgs measurement using the E4980A
with Option E4980A-001
Figure 5-29. MOS capacitance measurement
5.6
Silicon wafer C-V measurement
The C-V (capacitance versus DC bias voltage) characteristic of a MOS structure is an important measurement
parameter for evaluating silicon wafers. To evaluate the capacitance that varies with applied DC bias voltage,
capacitance is measured at a low AC signal level while sweeping a number of bias voltage points. Because the device
usually exhibits a low capacitance (typically in the low picofarads), the instrument must be able to measure low
capacitance accurately with a high resolution at a low test signal level. Precise bias voltage output is also required for
accurate C-V measurement. Typical C-V measurement conditions are listed in Table 5-2. Auto-balancing bridge
instruments are usually employed to satisfy the required performance.
Figures 5-30 and 5-31 show measurement setup examples using the auto-balancing bridge instrument (Keysight
E4990A, E4980A/AL, etc.) with a wafer prober station. Since the Low terminal of the auto-balancing bridge
instrument is sensitive to incoming noise, it is important that the Low terminal not be connected to the substrate that
is electrically connected to the prober’s noisy ground. If the wafer chuck (stage) of the prober is isolated from the
ground and effectively guarded, the shielding conductor of the 4TP cable can be connected to the prober’s guard
terminal to minimize stray capacitance around the probes.
When a device with low resistivity is measured, applied DC voltage decreases due to DC leakage current through the
device, and this may cause C-V measurement error. Using the DC bias auto level control (ALC) function helps to
lessen this problem.
Table 5-2. Typical C-V measurement conditions
Frequency
10 kHz to 1 MHz
(10 kHz to 100 MHz for a thin gate oxide layer measurement
Capacitance range
0.0001 to 1000 pF
Capacitance accuracy
±0.1%
Test signal level
20 or 30 mVrms typical
DC bias voltage
0 to ± 40 V
Bias voltage resolution
≤ 10 mV
Bias voltage accuracy
≤ 10 mV
Impedance analyzer/LCR meter
Keysight test cables
Lp
Lc
Hp
In order to extend the 4TP configuration up
to the position of probe heads, don’t
electrically interconnect the outer
shielding conductors of the cables here.
Hc
Interconnecting the outer shielding
conductors of four cables at “a” is
absolutely required to properly
terminate the 4TP.
a
b
Chuck
Interconnecting the outer shielding
conductors of probes at “b” is
recommended to make the 2TP
configuration.
Wafer prober
Figure 5-30. C-V measurement setup using 4TP cable extension
Impedance analyzer/LCR meter
Keysight test cables
Lc
Hp
Lp
Hc
a
Interconnecting the outer shielding
conductors of two BNC connectors at
“a” is absolutely required to properly
terminate the 4TP.
b
Chuck
Wa fe r prober
Figure 5-31. C-V measurement setup using 2TP cable extension
Interconnecting the outer shielding
conductors of probes at “b” is
recommended to make the 2TP
configuration.
As a result of extremely high integration of logic LSIs using MOS FETs, the thickness of the MOS FETs’ gate oxide
layer is becoming thinner (less than 2.0 nm), and such MOS FETs have been produced recently. In evaluating these
kinds of MOS FETs, leakage current becomes larger by the tunneling effect. Since the MOS gate capacitance has
high impedance, most of the test signal's current flows as leakage current. Consequently, the C-V characteristic of
MOS FET with a thin gate oxide layer cannot be measured accurately. To solve this problem, the test frequency
should be set higher (1 MHz or more) than usual to reduce the capacitive impedance across the thin gate oxide layer
to as low as possible. It is also important to simplify the measurement configuration to reduce residuals that exist in
the measurement path. If you perform high-frequency C-V measurement using the 4TP configuration, the
measurement error increases due to the residual inductance of the cable that is connected between the guard
electrodes of probe heads. Also, the compensation does not work properly because the distance between probes
easily varies. To solve this problem, a simplified 2T configuration with the 42941A impedance probe, as shown in
Figure 5-32, is highly recommended for accurate high-frequency C-V measurement.
Note: Keysight offers an advanced C-V measurement solution for the ultra-thin gate oxide layer that uses the
Keysight 4294A LF impedance analyzer. To eliminate the effects of tunneling leakage current, the MOS gate
capacitance is calculated from the result of swept frequency impedance measurement (|Z| – θ) at multiple DC
bias points. (Refer to Application Note 4294-3, Evaluation of MOS Capacitor Oxide C-V Characteristics Using
the Keysight 4294A, literature number 5988-5102EN.)
Figure 5-32. Example of high-frequency C-V measurement system configuration
5.7
High-frequency impedance measurement using the probe
As shown in Table 5-3, an RF I-V instrument can be used for a wafer’s L, C, and R measurements, which are
measurements in RF frequencies. Figure 5-33 shows an example of a measurement configuration when using the RF I-V
instrument. This figure illustrates a measurement system configuration for using the E4991B with a probe. Option
E4991B-010, the probe station connection kit, makes it easier to establish a probing system that can perform on-wafer
measurements from 1 MHz to 3 GHz. This kit contains a small test head and an extension cable.
The E4991B has calibration, compensation, and DC bias functions, and compared to a network analyzer, the E4991B
provides a wider impedance measurement range and stable measurement performance (refer to Section 2.6.)
Table 5-3. Application examples of high-frequency impedance measurements
using probe
Application
Parameters
DUT
Frequency
Measurement requirement
Spiral inductor
L, Q
RFIC for mobile phone
GHz
- Low inductance (nH range)
- High Q
Transistor, Diode
C, D
CMOS FET, PIN diode for
Transistor/diode optical use
MHz/GHz
- Low inductance (nH range)
- Low capacitance (pF range)
Disk head
C, D
GMR head, magnetic head
MHz/GHz
- Low inductance
IC package
C, L
IC package
GHz
- Low inductance (nH range)
- Low capacitance (pF range)
Memory
C, D
FRAM, DRAM, SRAM
MHz/GHz
- Low capacitance (pF range)
Dielectric material
C, D
Thin ilm layer,
PC board
MHz/GHz
- Wide impedance range
- Low-loss
Chip
inductor/capacitor
L, Q
C, D
Chip inductor
Chip capacitor
MHz/GHz
- Stable contact to small electrodes
- Wide impedance range
- High Q/Low D
Figure 5-33. Impedance measurement configuration when using the RF I-V instrument
5.8
Resonator measurement
The resonator is the key component in an oscillator circuit. Crystal and ceramic resonators are commonly used in the
kHz and MHz range. Figures 5-34 (a) and (b) show typical equivalent circuit and frequency response for a resonator.
A resonator has four primary elements: C, L, R, and Co. C and L determine the series resonant frequency (fr) and Co
and L determine the parallel resonant frequency (fa.) Qm (mechanical Q) is another measurement parameter used to
describe the performance of resonators. An impedance analyzer or network analyzer is used to measure the
resonator characteristics.
Figure 5-34. Typical resonator characteristics
In the following methods, note the impedance analyzer has an advantage in the accuracy of the
measurement results.
Impedance analyzer advantages
-
-
The impedance value at resonant frequency can be read directly.
(Network analyzers generally read in units of dB.)
Measurement accuracy for low impedance at series resonance and for high impedance at parallel resonance are
better than in network analysis.
Measurement is made by simply connecting the resonator to the test fixture, and residuals can be removed
using the compensation function. (Network analyzers require a π network circuit to be connected and cannot
compensate for all the residuals.)
By using the equivalent circuit analysis function, all resonator parameters are easily known.
(Network analyzers require complicated calculation or special analysis software to be used.)
Network analyzer advantages
-
Faster measurement speed.
Higher measurement frequency range.
Keysight’s impedance analyzers are suitable for testing resonators. With their equivalent circuit analysis function,
each resonator parameter can be determined individually. Also the I-BASIC
programming function facilitates the calculation of Qm and the extraction of other parameters. Figure 5-35 shows a
resonator measurement setup using an auto-balancing bridge instrument for a frequency range up to approximately
100 MHz. For higher frequency measurement, the same setup can be used with RF I-V measurement instrument.
Take the following precautions to ensure
accurate measurements:
1.
It is often assumed that the series resonant frequency, fr, is coincident with the minimum impedance point. This
is practical for an approximate measurement, but it is not the true value of fr. The true value of fr is given at θ =
0 and is typically 1 to 2 Hz above the minimum impedance point. Search the 0°-phase angle point for fr
measurement.
2.
It is important to properly set the oscillator output level; resonators are test-signal dependent. The minimum
impedance value and the series resonant frequency may vary depending on the applied test signal level.
Decrease the test signal level while monitoring the test current (I-monitor function) until the specified test level
is obtained.
3.
Perform an open/short compensation. Use All Point compensation mode instead of the interpolation mode
because the resonator measurements are narrowband. Also, pay special attention to the short compensation
procedure. Improper short compensation will result in measurement error for fr and the minimum impedance
value.
4.
Keep the measurement temperature constant. Resonators are temperature sensitive. Place a resonator into the
test fixture with your hand and wait until the series resonant frequency becomes constant.
Figure 5-35. Resonator measurement setup
Figures 5-36 (a) and (b) show an example of an impedance-phase characteristic measurement and equivalent circuit
analysis results for a crystal resonator. Equivalent circuit mode (E) serves to obtain the four-element equivalent circuit parameter values for a crystal resonator.
(a)
(b)
Figure 5-36. Resonator equivalent circuit mode on 4294A
5.9
Cable measurements
The characteristic impedance (Z(Ω)) capacitance per unit length (C (pF/m)), and the propagation constants α (dB/m)
and β (rad/m) are parameters commonly measured when evaluating cables. Figure 5-37 shows a measurement setup
for coaxial cable using an auto-balancing bridge type impedance analyzer and the 16047E test fixture. Note that the
High terminal of the test fixture is connected to the outer conductor of the cable. This measurement setup avoids the
effects of noise picked up by the outer conductor of the cable and is important to regard when the cable length is
long. The characteristic impedance and propagation constants are determined by measuring the impedance of the
cable with its other end opened and shorted (open-short method), and calculating using the equations shown in
Figure 5-37. The I-BASIC programming function of the impedance analyzer facilitates the calculations required.
Figure 5-38 demonstrates an example of measured characteristic impedance versus frequency.
Figure 5-37. Coaxial cable measurement setup and parameter calculation
Figure 5-38. Measurement result
5.9.1
Balanced cable measurement
A balun transformer is required for measuring balanced cable because the instrument’s UNKNOWN terminal is
unbalanced (refer to Section 5.10.) Figure 5-39 shows the measurement setup for a
balanced cable. A balanced/unbalanced 4T converter (Keysight part number 16314-60011) can be used to measure
balanced cables from 100 Hz to 10 MHz using an auto-balancing bridge instrument. For measurement using a
network analyzer, 16315-60011, 16316A, and 16317A are available. These converters have different characteristic
impedance to allow impedance matching with DUT (cable) impedance of 50, 100 and 600 Ω, respectively, as shown
in Table 5-4.
Table 5-4. 16314-60011, 16315-60011, 16316A, and 16317A
Characteristic impedance
Converter
Unbalanced side
Balanced side
16314-60011
50 Ω
50 Ω
16315-60011
50 Ω
50 Ω
16316A
50 Ω
100 Ω
16317A
50 Ω
600 Ω
Figure 5-39. Balanced cable measurement setup
Applicable instrument
Auto-balancing bridge instrument
Network analyzer
5.10 Balanced device measurement
When a balanced DUT (such as balanced cable or the balanced input impedance of a differential amplifier) is
measured, it is necessary to connect a “balun” (balance-unbalance) transformer between the instrument and the
DUT. Looking from the DUT side, the UNKNOWN terminals of the impedance measurement instrument are in an
“unbalanced” configuration. Figure 5-40 (a) shows an example of an auto-balancing bridge instrument. Its Low
terminal is considered a virtual ground because it is held at approximately a 0 V potential. When a 1:1 balun
transformer is connected as shown in Figure 5-40 (b), the instrument can measure a balanced DUT directly.
Figure 5-40. Balanced device measurement
An actual balun transformer has a limited frequency range. The measurement must be made within its frequency
range. In addition to Keysight’s balanced/unbalanced converters, various types of commercial balun transformers are
available for various frequency ranges. To select the appropriate balun transformer, check the frequency range and
the impedance of the transformer’s balanced (DUT) side. Its impedance should be close to the characteristic
impedance of the DUT. The impedance of the unbalanced side should be 50 or 75 Ω as appropriate for the
measurement instrument. Open/short/load compensation for the balun transformer is required when the turns ratio
of the balun transformer used is not 1:1, or when an accurate measurement is needed. Open/short compensation is
not adequate because the balun transformer will produce both magnitude (|Z|) and phase errors due to its transfer
function characteristic. The terminal connectors of the balanced side should be connectable for both the standard
devices used for open/short/load compensation and the DUT. Figures 5-41 (a) through (d) show an example of an
actual balun configuration and compensation.
Figure 5-41. Measurement setup
5.11 Battery measurement
The internal resistance of a battery is generally measured using a 1 kHz AC signal. It is not allowed to directly
connect a battery to the auto-balancing bridge type impedance measurement instrument. If a battery is connected
directly, the instrument becomes the DC load, typically 100 Ω for the battery. The instrument may be damaged by a
discharge current flow from the battery. Figure 5-42 shows the recommended setup for this measurement. C1 and C2
block DC current from flowing into the instrument. The value of C1 should be calculated using the minimum
measurement frequency. For example, when the measurement is made at 1 kHz and above, C1 should be larger than
32 µF. The voltage rating of C1 and C2 must be higher than the output voltage of the battery.
Note: The Keysight 4338B milliohm meter can measure the internal resistance of a battery up to 40 V DC directly
connected to the measurement terminals because the DC blocking capacitors have been installed in the 1 kHz
bridge circuit.
Figure 5-42. Battery measurement setup
5.12 Test signal voltage enhancement
When measuring the impedance of test signal level dependent devices, such as liquid crystals, inductors, and high
value ceramic capacitors, it is necessary to vary the test signal voltage. Many of the auto-balancing bridge
instruments employ a test signal source whose output is variable, typically from 5 mV to 1V rms. Particularly, the
E4980A with Option E4980A-001 can output a test signal voltage of up to 20 V rms and is the most suitable for this
application.
In some cases, measurement needs exist for evaluating impedance characteristics at large test signal voltages
beyond the maximum oscillator output level of the instrument. For auto-balancing bridge instruments, output
voltage enhancement is possible if the test signal is amplified as shown in Figure 5-43. A voltage divider is also
required so that the input voltage of the Hp terminal is the same as the output voltage of the Hc terminal. The DUT’s
impedance is a concern. Because the current flowing through the DUT is also amplified and flows directly into the Rr
circuit, it should not exceed the maximum allowable input current of the Lc terminal. Typically, this is 10 mA. For
example, when a 10 V rms signal is applied to the DUT, the minimum measurable impedance is
10 V/10 mA = 1 kΩ. Also, it should be noted that measured impedance is 1/A (gain of amplifier) of an actual DUT’s
impedance. For example, when a 10 pF capacitor is measured using ×10 amplifier, displayed value will be 100 pF.
Note: For RF I-V instrument, it is impossible to amplify the test signal because at the test port the signal source
output is not separate from the voltmeter and current meter inputs.
Figure 5-43. Schematic diagram of test signal voltage enhancement circuit
Figure 5-44 shows a measurement setup example to boost the test signal voltage by factor of 10 (A = 10). The
amplifier used in this application should have constant gain in the measurement frequency range and output
impedance less than 100 Ω. R3 in Figure 5-44 needs to be adjusted to compensate for the magnitude error in
measured impedance and C2 needs to be adjusted for flat frequency response. This can be accomplished by
comparing the measured values with known values of a reference device. For better accuracy, perform the
open/short/load compensation at a test signal level below 1 V rms (not to cause an excessive current to flow in
short condition.) The required circuit constants of the divider are different depending on the input impedance of
the Hp terminal of the instrument.
Figure 5-44. Connection diagram of test signal voltage enhancement circuit
5.13 DC bias voltage enhancement
DC biased impedance measurement is popularly used to evaluate the characteristics of the device under conditions
where the device actually operates in circuits. The internal DC bias function of impedance measurement instruments
is normally designed to apply a bias voltage to capacitor DUTs. It is suited to DC biased capacitor measurements.
Maximum applicable bias voltage is different for instruments. The internal bias source can typically output a variable
bias voltage of up to ±40 V through the Hc terminal. An external DC voltage source is required to apply a DC bias
voltage that must exceed the limits of the internal DC bias function.
Some instruments have a DC bias input terminal for connecting an external DC voltage source. Use an external bias
fixture or adapter for other instruments with no internal DC bias and for DC bias requirements that exceed the
maximum voltage of the bias input terminal. Table 5-5 lists the available bias fixture and adapters.
Table 5-5. External bias fixture and adapters
Applicable
instrument
Maximum bias voltage
and current
Usable frequency
range
Applicable
DUT type*
Bias ixture
Auto-balancing
bridge
± 200 V,
2 mA
50 Hz to
2 MHz
Leaded
16065C
Bias adapter
Auto-balancing
bridge
± 40 V,
20 mA
50 Hz to
2 MHz
Leaded and
SMD
16200B
Bias adapter
RF I-V
E4990A+42942A,
4294A+42942A
± 40 V, 5 A
1 MHz to
1 GHz
Leaded and
SMD
Model number
Product type
16065A
Note:
Applicable DUT types for the 16065C and 16200A depend on the test ixture connected.
Use the 16065A external voltage bias fixture, which has a built-in protection circuit, for leaded devices and high
voltage DC bias of up to ±200 V. Figure 5-45 shows the setup for a +200 V DC biased measurement. Since the
16065A is equipped with a bias monitor output, a digital voltmeter is used to monitor the DC bias voltage actually
applied to the DUT. The 16065C external voltage bias adapter is designed to apply a bias voltage of up to ±40 V from
an external voltage source. This adapter can be connected between any 4TP test fixture and the instrument’s
UNKNOWN terminals, thus allowing the use of an appropriate test fixture that accommodates the DUT. The 16200B
external DC bias adapter operates specifically with the RF I-V measurement instruments and the E4990A/4294A with
the 42942A. This adapter resolves both voltage bias and current bias needs. When used for capacitor measurements,
it allows a bias voltage of up to 40 V DC across the DUT by using an external DC voltage source.
Figure 5-45. External DC bias measurement setup
5.13.1 External DC voltage bias protection in 4TP configuration
If the measurement frequency is above 2 MHz or the type of DUT is not suitable for these external bias fixtures, it is
recommended that a protective circuit, shown in Figure 5-46, is used. This circuit is usable with bias voltage up to
±200 V. To reduce the effects of this additional circuit, perform the open/short compensation with no bias voltage
applied.
Figure 5-46. External DC voltage bias protection circuit
5.14 DC bias current enhancement
DC current biasing is used for inductor and transformer measurement. In the low frequency region, the E4980A with
the 42841A bias current source are both suitable for this application because they can apply up to 20 A of bias
current. (This can be extended up to 40 A if two 42841As are connected in parallel.)
To deliver a bias current in RF impedance measurement, the 16200B external DC bias adapter can be used with the
RF I-V measurement instrument. The 16200B allows you to supply a bias current of up to 5 A across the DUT by
using an external DC current source. The 16200B is directly attached to the 7-mm test port and the test fixture onto
the 16200B as shown in Figure 5-47. To minimize the bias adapter-induced errors, perform open/short/load
calibration at the test fixture terminals with no bias voltage/current applied.
Figure 5-47. External DC bias measurement using the RF I-V measurement instrument
5.14.1 External current bias circuit in 4TP configuration
For external current bias measurement using other auto-balancing bridge instruments, an external DC current source
and a protection circuit are required. The following describes a protection circuit that can be used for DC bias current
measurements up to 10 A. Figure 5-48 shows the protection circuit schematic diagram.
Figure 5-48. External current bias protection circuit
Take caution of electrical shock hazards when using the external DC bias circuit.
A large energy is charged in L1 and L2, as well as the DUT (Lx), by a bias current delivered from an external power
supply and when the DUT is disconnected from the measurement circuit, the DUT generates a very high spike
voltage (kick-back voltage) to discharge the energy. To ensure operator safety, decrease the bias current to zero
before disconnecting the DUT.
L1 and L2 discharge through the protection circuit the instant the DUT is disconnected from the measurement circuit
or when the bias current is turned off. To prevent the instrument from being damaged by harmful discharge, the
protection circuit must be designed carefully for the withstanding voltage/current rating of each circuit component.
Refer to Application Note 346 A Guideline for Designing External DC Bias Circuit for more information.
5.15 Equivalent circuit analysis and its application
Keysight’s impedance analyzers are equipped with an equivalent circuit analysis function. The purpose of this
function is to model the various kinds of components as three- of four-element circuits. The values of the
component’s main elements and the dominant parasitics can be individually determined with this function.
Many impedance measurement instruments can measure the real (resistive) and the imaginary (inductive or
capacitive reactance) components of impedance in both the series and parallel modes. This models the component
as a two-element circuit. The equivalent circuit analysis function enhances this to apply to a three- or four-element
circuit model using the component’s frequency response characteristics. It can also simulate the frequency response
curve when the values of the three- or four-element circuit are input.
Impedance measurement at only one frequency is enough to determine the values of each element in a two-element
circuit. For three- or four-element circuits, however, impedance measurements at multiple frequencies are necessary.
This is because three (four) equations must be set up to obtain three (four) unknown values. Since two equations are
set up using one frequency (for the real and imaginary), one more frequency is necessary for one or two more
unknowns. The equivalent circuit analysis function automatically selects two frequencies where the maximum
—
—
measurement accuracy is obtained. (This is at the frequency where the √ 2 × minimum value or 1/√ 2 × maximum value
is obtained.) If the equivalent circuit model (described later) is properly selected, accuracy for obtained values of a
three- or four-element circuit is comparable to the measurement accuracy of the instrument.
The equivalent circuit analysis function has five circuit modes as shown in Figure 5-49, which also lists their
applications. The following procedure describes how to use the equivalent circuit analysis function.
1.
Perform a swept frequency measurement for the unknown DUT using the |Z| – θ or |Y| – θ function. The sweep
mode can be either linear or logarithmic.
2.
Observe the frequency response curve. See the typical frequency response curve given in
Figure 5-49. Choose the circuit mode that is most similar to the measured curve.
3.
Calculate the equivalent parameters by pressing the “Calculate Parameter” key (or the key with the same
function.) Three or four values for selected circuit mode are calculated and displayed.
4.
Check the simulated frequency response curve. The simulated curve is calculated from the obtained equivalent
parameters. If the fitting quality between the simulated curve and the actual measurement results is high, the
proper circuit mode was selected. If not, try one of other circuit modes.
Figure 5-49. Equivalent circuit models on 4294A
If the simulated frequency response curve partially fits the measurement results, it can be said that the selected
circuit mode is proper only for that part of the frequency range that it fits. Figure 5-50 (a) shows an example
measurement for a low value inductor. As shown in Figures 5-50 (b) and (c), the measurement result does not agree
with the simulated curves over the full frequency range. The higher frequency region is well simulated by circuit
mode A and the lower frequency region by circuit mode B. In other words, the circuit mode for the inductor is like the
circuit mode A at the higher frequencies and like circuit mode B at lower frequencies. At the higher frequencies C and
R in parallel with L are the dominant elements and circuit mode A describes the response curve best. At the lower
frequencies L and series R are the dominant circuit elements and circuit B describes the response curve best. From
these facts, we can determine that the real circuit mode should be the combination of circuit modes A and B, and is
like Figure 5-51 (a). Figure 5-51 (b) lists an I-BASIC program to simulate the frequency response for the circuit given
in Figure 5-51 (a). The value of Rs should be keyed in from the front panel and entered into the internal register, so
that the calculation can be executed and the simulated curve obtained. In this example, the simulated curve agreed
with the actual result as shown in Figure 5-51 (c) when the value of Rs is 1 Ω.
(a)
(b)
Circuit mode A
(c)
Circuit mode B
Figure 5-50. Frequency response simulation for a low-value inductor
(a)
(c)
Figure 5-51. Equivalent circuit enhancement
(b)
10
20
30
40
50
60
70
80
90
100
110
120
130
140
150
160
170
180
190
200
210
220
230
240
250
260
270
280
290
300
310
320
330
340
350
360
370
380
390
400
410
420
430
440
450
460
470
480
490
500
510
DIM Ztrc(1:201,1:2),Fmta$[9],Fmtb$[9]
DIM Ttrc(1:201,1:2)
DIM R(201),R1(201)
DIM X(201)
DIM Zdat(1:201,1:2)
DIM Tdat(1:201,1:2)
!
DEG
!
IF SYSTEM$(“SYSTEM ID”)=”HP4294A” THEN
ASSIGN @Agt4294a TO 800
ELSE
ASSIGN @Agt4294a TO 717
END IF
!
OUTPUT @Agt4294a;”FORM4”
!
OUTPUT @Agt4294a;”TRAC A”
OUTPUT @Agt4294a;”FMT?”
ENTER @Agt4294a;Fmta$
OUTPUT @Agt4294a;”OUTPMTRC?”
ENTER @Agt4294a;Ztrc(*)
!
OUTPUT @Agt4294a;”TRAC B”
OUTPUT @Agt4294a;”FMT?”
ENTER @Agt4294a;Fmtb$
OUTPUT @Agt4294a;”OUTPMTRC?”
ENTER @Agt4294a;Ttrc(*)
!
OUTPUT @Agt4294a;”DATMEM”
!
INPUT “Rs=”,Rs
!
FOR I=1 TO 201
R(I)=Ztrc(I,1)*COS(Ttrc(I,1))
X(I)=Ztrc(I,1)*SIN(Ttrc(I,1))
R1(I)=R(I)+Rs
Zdat(I,1)=SQR(R1(I)^2+X(I)^2)
Tdat(I,1)=ATN(X(I)/R1(I))
NEXT I
!
OUTPUT @Agt4294a;”TRAC A”
OUTPUT @Agt4294a;”FMT “&Fmta$
OUTPUT @Agt4294a;”INPUDTRC “;Zdat(*)
!
OUTPUT @Agt4294a;”TRAC B”
OUTPUT @Agt4294a;”FMT “&Fmtb$
OUTPUT @Agt4294a;”INPUDTRC “;Tdat(*)
!
GOTO 320
END
Measurement accuracy can be improved by taking advantage of the equivalent circuit analysis. Figure 5-52 (a) shows
an Ls-Q measurement example for an inductor. In this example, an impedance analyzer measures the Q value at 10
MHz. Measured data read by MARKER is Ls = 4.78 µH and Q = 49.6. The Q measurement accuracy for this
impedance at 10 MHz is calculated from the instrument’s specified D measurement accuracy of ±0.011, and the true
Q value will be between 32 and 109. The reason that the uncertainty of the Q value is so high is that the small
resistive component relative to reactance cannot be measured accurately. It is possible to measure the resistive
component accurately if the inductive component is canceled by the capacitance connected in series with the
inductor. When a loss-less capacitor of 1/(ω2L) = 53 pF is connected, the inductor will resonate at 10 MHz. (In this
example, a 46 pF capacitor is used for resonance.) Figure 5-52 (b) shows the |Z| - θ measurement results when a
46 pF capacitor is connected. This result can be modeled using circuit mode D, and the value of R is calculated to be
8.51 Ω. The value of L is obtained as 4.93 µH. Since the equivalent circuit analysis function uses approximately 8.51 ×
—
√2 Ω data to calculate the R value, the specified measurement accuracy for a 12 Ω resistance measurement can be
used and is ±1.3 percent. Therefore, the Q value can be calculated from Q = ωLs/R = 36.4 with an accuracy of ±2.4%
(sum of the L accuracy and R accuracy.) In this measurement, the capacitance value does not have to be exactly the
caculated value but the loss of the capacitor should be very small because it will affect the calculated Q value.
(a)
(b)
Figure 5-52. Q measurement accuracy improvement
Appendix A. The Concept of a Test Fixture’s Additional Error
A.1
System coniguration for impedance measurement
Frequently the system configured for impedance measurements uses the following components
(see Figure A-1.)
1.
2.
3.
Impedance measurement instrument
Cables and adapter interfaces
Test fixture
Figure A-1. System configuration for impedance measurement
The impedance measurement instrument’s accuracy is defined at the measurement port of the instrument. This
means that the accuracy of the measurement values at the measurement port is guaranteed and has calibration
traceability.
In an actual measurement, there can be an extension of the measurement port with a cable or an adapter conversion
to match the test fixture’s terminal configuration. For this reason, cables and conversion adapters are provided for
connectivity with the measurement port. These cables (and adapters) are designed to maintain high accuracy of the
measurement instrument while extending the measurement port. Most of the time, the measurement accuracy of the
instrument and the cable (or adapter) are specified together as a whole.
A test fixture is an accessory used to connect the DUT to the measurement instrument. Many test fixtures are provided to adapt to various shapes and sizes of DUTs. A test fixture is either connected directly to the measurement port of
the instrument, or to the port of the extension cable or conversion adapter, as described earlier. The test fixture’s
structure determines the applicable frequency and impedance ranges. Hence, it is necessary to use the appropriate
test fixture for the desired measurement conditions. In addition, each test fixture has its own inherent characteristic
error, which is detailed in its operation manual.
A.2
Measurement system accuracy
The equation for the accuracy of a measurement system is:
(Measurement accuracy) = (Instrument’s accuracy) + (Test fixture’s error)
The measurement instrument’s accuracy is determined by an equation with terms that are dependent on frequency,
measured impedance, signal level, and measurement time mode. By substituting the respective measurement conditions into the equation, the measurement accuracy is calculated. If a cable or a conversion adapter is used, then the
specified measurement accuracy is the accuracy of the measurement instrument with the cable or adapter. This combined measurement accuracy is shown in the instrument’s operation manual.
Typical equations for determining the test fixture's error are:
Ze = ±{ A + (Zs/Zx + Yo × Zx) × 100} (%)
De = Ze/100 (D ≤ 0.1)
Ze:
De:
A:
Zs/Zx × 100:
Yo × Zx × 100:
Zs:
Yo:
Zx:
Additional error for impedance (%)
Additional error for dissipation factor
Test fixture’s proportional error (%)
Short offset error (%)
Open offset error (%)
Test fixture’s short repeatability (Ω)
Test fixture’s open repeatability (S)
Measured impedance value of DUT (Ω)
Proportional error, open repeatability, and short repeatability are mentioned in the test fixture’s operation manual and
in the accessory guide. By inputting the measurement impedance and frequency (proportional error, open repeatability, and short repeatability are usually a function of frequency) into the above equation, the fixture’s additional error
can be calculated.
A.2.1
Proportional error
The term, proportional error, A, is derived from the error factor, which causes the absolute impedance error to be proportional to the impedance being measured. If only the first term is taken out of the above equation and multiplied by
Zx, then ∆Z = A × Zx (Ω). This means that the absolute value of the impedance error will always be A times the measured impedance. The magnitude of proportional error is dependent upon how precisely the test fixture is constructed
to obtain electrically and mechanically optimum matching with both the DUT and instrument. Conceptually, it is
dependent upon the simplicity of the fixture’s equivalent circuit model and the stability of residuals. Empirically, proportional error is proportional to the frequency squared.
A.2.2
Short offset error
The term, Zs/Zx x 100, is called short offset error. If Zx is multiplied to this term, then ∆Z = Zs (Ω). Therefore, this term
affects the absolute impedance error, by adding an offset. Short repeatability, Zs, is determined from the variations in
multiple impedance measurements of the test fixture in short condition. After performing short compensation, the
measured values of the short condition will distribute around 0 Ω in the complex impedance plane. The maximum
value of the impedance vector is defined as short repeatability. This is shown in Figure A-2. The larger short repeatability is, the more difficult it is to measure small impedance values. For example, if the test fixture’s short repeatability
is ±100 mΩ, then the additional error of an impedance measurement under 100 mΩ will be more than 100 percent. In
essence, short repeatability is made up of a residual resistance and a residual inductance part, which become larger
as the frequency becomes higher.
Figure A-2. Definition of short repeatability
A.2.3
Open offset error
The term, Yo x Zx x 100 is called open offset error. If Zx is multiplied to this term, then ∆Y = Yo. This term affects the
absolute admittance error, by adding an offset. Open repeatability, Yo, is determined from the variations in multiple
admittance measurements of the test fixture in the open condition. After performing open compensation, the measured values of the open condition will distribute around 0 S in the complex admittance plane. As shown in Figure A-3,
the maximum value of the admittance vector in the complex admittance plane is defined as open repeatability. The
larger open repeatability is, the more difficult it is to measure large impedance values. Open repeatability is made up
of a stray conductance and a stray capacitance part, which become larger as the frequency becomes higher.
Figure A-3. Definition of open repeatability
A.3 New market trends and the additional error for test fixtures
A.3.1
New devices
Recently, the debut of extremely low ESR capacitors, and the trend to use capacitors at much higher frequencies,
have increased demand for low impedance measurements. As a result, the test fixture’s short repeatability has
become increasingly important. In Figure A-4, the relationship between
proportional error, short offset error, and frequency are shown when measuring low impedance of 100 mΩ and 10 Ω.
Notice that when the measured impedance is less than 100 mΩ, short offset error influences the entirety of the test
fixture’s inherent error. As shown in the Figure A-4, when the DUT's impedance is 100 mΩ and the test fixture’s short
repeatability is 10 mΩ, the short offset error will be 10 percent. Since the proportional error is minimal in low frequencies, the additional error will also be 10 percent.
Figure A-4. Relationship between proportional error, short offset error, and frequency when measuring low impedance
Until recently, to allow for additional error in test fixtures it was common to just specify the proportional error (A.) As
shown in the 10 Ω measurement case, if the measured impedance is large in comparison to the test fixture’s short
repeatability, then the short offset error can be ignored completely. This is the reason why open and short offset error
was not previously specified. This is the reason for test fixtures that are only specified with proportional error. On the
contrary, for measured impedance from 1 Ω to 10 kΩ, proportional error (A) alone is sufficient to express the test
fixture’s additional error.
A.3.2
DUT connection configuration
In order to make short repeatability small, there are test fixtures that use the 4T connection configuration (for
example, Keysight 16044A). By employing this technique, the effect of contact resistance is reduced and short
repeatability is significantly improved. As a result, the range of accurate low impedance measurements is expanded
down to a low milliohm region.
Figure A-5 shows the difference between the 2T connection and the 4T connection. In a 2T connection, the contact
resistance that exists between the fixture’s contact electrodes and the DUT, is measured together with the DUT’s
impedance. Contact resistance cannot be eliminated by compensation because the contact resistance value changes
each time the DUT is contacted.
Fixture
Measurement
instrument
Fixture
Measurement
instrument
Hc
Hc
Hp
Hp
V
DUT
Lp
A
Contact
point
Contact
point
V
DUT
Shielding plate
Lp
Shielding plate
Lc
A
(a) Two terminal test fixture
Lc
( b) Four terminal test fixture
Figure A-5. Two-terminal and four-terminal connection techniques
In a 4T connection, the voltage and current terminals are separate. Since the voltmeter has high input impedance, no
current flows into the voltage terminals. Hence, the voltage that is applied across the DUT can be accurately detected
without being affected by the contact resistance. Also, the current that flows through the DUT flows directly into the
current terminal and is accurately detected without being affected by the contact resistance. As a result, the 4T
connection method can eliminate the effect of contact resistance and realize a small short repeatability. By using a 4T
test fixture, it is possible to measure low impedance with better accuracy than that which can be measured with a 2T
test fixture.
The 2T test fixture can be used up to a higher frequency than the 4T test fixture. Since the 2T test fixture has a simple
DUT connection configuration, the effects of residuals and mutual coupling (jωM), which cause measurement error to
increase with frequency, are smaller than those of the 4T test fixture and can be effectively reduced by compensation.
Thus, the 2T connection is incorporated in test fixtures designed for use in the higher frequency region (typically up to
120 MHz.)
A.3.3
Test fixture’s adaptability for a particular measurement
In order to make use of what has been discussed previously, the test fixture’s adaptability for a particular
measurement will be discussed. To see whether a test fixture is adaptable, it is important to think about the test
fixture’s additional error (proportional error, short repeatability, and open repeatability), measurement impedance, and
the test frequency range.
If the measurement impedance is in the 1 Ω to 10 kΩ range, use only proportional error to calculate the additional
error of the test fixture. It is fine to assume that this is a close approximation to the fixture’s additional error.
If the measurement impedance is not in this range, use proportional error, short repeatability, and open repeatability
to calculate the test fixture’s additional error. Recent test fixtures have all three terms specified in their operation
manual, so use these values for the calculation.
Some of the recent test fixtures (16044A), due to their structure, have different performance characteristics with
different measurement instruments. For these test fixtures, refer to their operation manual for more details about the
specifications.
If the test fixture is not specified with short and open repeatability, how can the fixture’s adaptability be determined? To
measure a test fixture’s short repeatability, measure the impedance of the short condition after performing short
compensation. Take the shorting plate out of the fixture and then replace it. Measure the short condition again. By
repeating this process at least 50 times, it will show the variations in the measured impedance of short condition (see
Figure A-6.) The final step to determine an approximation of short repeatability is to add a margin to the values
obtained. For open repeatability, measure the admittance of the test fixture’s open condition. In the same way,
determine open repeatability by measuring at least 50 times.
Specifications of
short repeatability
Actual measurement
of short repeatability
Figure A-6. Measurement of short repeatability (16034G)
Measurement settings
Measurement instrument:
Measurement frequency:
Measurement parameter:
Compensation:
Bandwidth:
Measurement method:
Display method:
4294A
40 Hz to 10 MHz
Z-θ
Performed short compensation
3
Inserted the shorting plate, measure the short condition, and remove
the shorting plate. Repeated this 50 times
Overlaying traces by using the accumulate mode
Lastly, a method of visually analyzing the accurate measurement range of a test fixture is introduced. This method is
only appropriate when all three error-terms (proportional error, open repeatability, and short repeatability) are known.
Table A-1 shows the additional error of the 16034G. The whole equation, with all three terms, can be solved for
measurement impedance rather than additional error, for example when additional error is equal to 0.5 percent. If the
obtained impedance values are plotted with measurement impedance (y-axis) versus frequency (x-axis), a graph
similar to the one shown in Figure A-7 can be obtained. The area inside the plotted curve shows the range of
impedance that can be measured with an additional error better than 0.5 percent. In the same way, other graphs can
be drawn with other additional error values to better visualize the accuracy for a given impedance and frequency
range. The operation manuals of recent test fixtures present such graphs.
Table A-1. Additional error of 16034G
Type of error
Impedance
Proportional error
– 0.5 × (f/10)2 [%]
Open repeatability
– 5 + 500 × (f/10) [nS]
Short repeatability
– 10 + 13 × (f/10) [mΩ]
Figure A-7. Range of impedance measurable with additional error ≤ 0.5 percent
Appendix B: Open/Short Compensation
The open/short compensation used in Keysight’s instrument models the residuals of a test fixture or test leads as a
linear four-terminal network (a two-terminal pair network) represented by parameters A, B, C, and D (shown in
Figure B-1.) This circuit model is basically same as that used in open/short/load compensation.
I1
Measurement
instrument
V1
I2
AB
C D
V2
Z du t
DU T
Unknown 4-terminal
circuit
Figure B-1. Four-terminal network circuit model of a test fixture or test cables
The difference between open/short and open/short/load compensation is that the open/short
compensation assumes the unknown network as a “symmetrical network.” From this restriction, the open/short compensation does not require the load measurement.
The circuit model shown in Figure B-1 is expressed by using the following matrix equation:
( ) ( )( )
V1
I1
=
A B
C D
V2
I2
(1)
The relationships between V1, I1, V2, and I2 are given by the following equations:
{
V1 = AV2 + BI2
I1 = CV2 + DI2
The measured impedance of the DUT, Zxm, is expressed as:
Zxm =
V1
AV2 + BI2
=
I1
CV2 + DI2
(2)
On the other hand, the true value of the DUT, Zdut, is expressed as:
Zdut =
V2
I2
(3)
From equations (2) and (3), the equation that expresses the relationship between Zxm and Zdut is derived as follows:
AV2 + BI2
Zxm =
=
CV2 + DI2
V2
+B
I2
AZdut + B
=
V
CZdut + D
C 2+D
I2
A
(4)
Open measurement
When nothing is connected to the measurement terminals (open condition), I2 is 0. Therefore, equation (5) is derived
by substituting I2 = 0 for I2 in the equation (2). Here, Zo means the impedance measured with measurement terminals
opened.
Zo =
AV2
A
=
CV2
C
cC=
A
Zo
(5)
Short measurement
When the measurement terminals are shorted, V2 is 0. Therefore, equation (6) is derived by substituting V2 = 0 for V2
in the equation (2). Here, Zs means the impedance measured with measurement terminals shorted.
Zs =
BI2
B
=
DI2
D
(6)
c B =DZs
By substituting B = DZs and C = A/Zo (of equations 6 and 5) for the parameters B and C, respectively, of equation (4),
the following equation is derived:
Zdut =
B – DZxm
CZxm – A
=
B – DZxm
(
=
D(Zs – Zxm)
) (
Zxm
–1 A
Zo
)
=
Zxm
–1 A
Zo
D(Zs – Zxm)
(Zxm – Zo)A
Z
(7)
Since the open/short compensation assumes that the unknown network circuit is a symmetrical network, the parameters A and D are equal:
A=D
(8)
Thus, equation (7) can be simplified as follows:
Zdut =
Zs – Zxm
Zo
Zxm – Zo
(9)
The definitions of the parameters used in this equation are:
Zdut
Corrected impedance of the DUT
Zxm
Measured impedance of the DUT
Zo
Measured impedance when the measurement terminals are open
Zs
Measured impedance when the measurement terminals are shorted
Note: These parameters are complex values that have real and imaginary components.
Appendix C: Open, Short, and Load Compensation
Since a non-symmetrical network circuit is assumed, equation (8) in Appendix B is not applied. Therefore, the
relationship between A and D parameters must be determined. The measurement of a reference DUT (load device) is
required to determine A and D.
When the applied voltage across a load device is V2’ and the current flow through it is I2’, the impedance of the load
device, Zstd, is expressed as:
V2’
I2’
Zstd =
(10)
The measured value of the load device, Zsm, is expressed by using matrix parameters like equation (2) of open/short
compensation, as follow:
AV2’ + BI2’
CV2’ + DI2’
Zsm =
(11)
By substituting Zstd for V2’ / I2’ in equation (11), the following equation is derived:
V’
A 2 +B
AV2’ + BI2’
I2’
AZstd + B
Zsm =
=
=
CV2’ + DI2’
V’
CZstd + D
C 2 +D
I2’
(12)
Using equation (5) of open measurement and equation (6) of short measurement, the relationship between the
parameters A and D is expressed by the following equation:
Zsm =
cD=
AZstd + B
AZstd + DZs
AZstd + DZs
=
= Zo
CZstd + D Zstd
AZstd + DZo
A+D
Zo
ZstdZsm – ZstdZo
ZoZs – ZsmZo A
(13)
By substituting equation (13) for the parameter D of equation (7), the equation for calculating the corrected
impedance of the DUT is derived as follows:
Zdut =
D(Zs – Zxm)
ZstdZsm – ZstdZo
(Zs – Zxm)
Zo =
Ax
Zo
(Zxm – Zo)A
ZoZs – ZsmZo
(Zxm – Zo)A
Zdut =
(Zs – Zxm)(Zsm – Zo)
Zstd
(Zxm – Zo)(Zs – Zsm)
(14)
The definitions of the parameters used in this equation are:
Zdut
Zxm
Zo
Zs
Zsm
Zstd
Corrected impedance of the DUT
Measured impedance of the DUT
Measured impedance when the measurement terminals are open
Measured impedance when the measurement terminals are short
Measured impedance of the load device
True value of the load device
Note: These parameters are complex values which have real and imaginary components.
Appendix D: Electrical Length Compensation
A test port extension can be modeled using a coaxial transmission line as shown in Figure D-1. When an impedance
element ZL is connected to the tip of the line, the measured impedance value Zi at the other end of the line (that is,
the test port) is given by the following equation:
ZL + Zo tan h γ e
Zi = Zo ———————————————————
ZL tan h γ e + Zo
γ = α + jβ = √ZY = √(R+jωL)(G+jωC)
Where,
γ:
α:
β:
e:
Zo:
Propagation constant of the transmission line
Attenuation constant of the transmission line
Phase constant of the transmission line
Transmission line length
Characteristic impedance of the transmission line
Figure D-1. Transmission line model of test port extension
The DUT impedance value is therefore calculated as:
Zo tan h γ e - Zi
ZL = Zo ———————————————————
Zi tan h γ e - Zo
–––
If the transmission line has no propagation loss (α = 0, β = ω√LC ), the equation for ZL is simplified as follows:
Zi - jZo tan β e
ZL = Zo ———————————
Zo - jZi tan β e
The true ZL value can be calculated if the phase shift quantity, β e, is known. Here, the phase constant β is related to
the test signal wavelength λ in the transmission line as follows:
2π
β = ———
λ
When a (virtual) transmission line in which the signal wavelength is equal to the wavelength in a vacuum is assumed,
the virtual line length ( ee) that causes the same phase shift (βe) as in the actual line is given by the following
equation:
λo
2πe 2πe e
e = ——— e (because β e = ———— = —————— )
λ
λ
λo
Where,
λo is a wavelength in vacuum
λ is a wavelength in transmission line
Therefore, the phase shift quantity, β e, can also be expressed by using the phase constant βo in vacuum and the
virtual line length e e (because β e = βo e e.) Since the βo value is derived from physical constants (βo = 2π/λo = ω/c,
c: velocity of light), it is possible to represent the phase shift by using only the virtual line length e e.
This virtual line length is specified as the electrical length of the test fixtures and airline extensions. Accordingly, the
compensation procedure to derive the impedance ZL can be simplified by using the electrical length value. In case of
the coaxial line, since the β value is proportional to √C(C: distributed capacitance of the line), the electrical length is
proportional to the square root of the dielectric constant of the insulation layer between the inner and outer
conductors.
Ihr Spezialist für
Mess- und Prüfgeräte
Appendix E: Q Measurement Accuracy Calculation
Q measurement accuracy for auto-balancing bridge type instruments is not specified directly as ±%. Q accuracy
should be calculated using the following equation giving the possible Q value tolerance.
1
1
± D
Qm
Qt =
Where,
Qt is the possible Q value tolerance
Qm is measured Q value
ΔD is D measurement accuracy
For example, when the unknown device is measured by an instrument which has D measurement accuracy of 0.001,
and the displayed Q value is 200, the Q tolerance is calculated as:
Qt =
1
1
=
1
0.005
± 0.001
± 0.001
200
This result means that the true Q value will be between 167 and 250.
Note: The following equation may be used to calculate the Q value tolerance. (The result is the same at that from
the above equation.)
Qm 2 × D
1 (Qm × D )
±
Qt = ±
This information is subject to change without notice.
© Keysight Technologies, 2009 - 2015 Published
in USA, October 7, 2015
5950-3000
www.keysight.com
Distributed by:
dataTec ▪ Ferdinand-Lassalle-Str. 52 ▪ 72770 Reutlingen ▪ Tel. 07121 / 51 50 50 ▪ Fax 07121 / 51 50 10 ▪ info@datatec.de ▪ www.datatec.de
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