This white paper describes signal integrity (SI) mechanisms that cause system

This white paper describes signal integrity (SI) mechanisms that cause system
Modeling System Signal Integrity
Uncertainty Considerations
WP-01153-1.0
White Paper
This white paper describes signal integrity (SI) mechanisms that cause system-level
timing uncertainty and how these mechanisms are modeled in the Quartus® II
TimeQuest Timing Analyzer for timing closure for external memory interface designs.
By using the Quartus II development software version 9.1 and later to achieve timing
closure for external memory interfaces, a designer does not need to allocate a separate
SI timing budget to account for simultaneous switching output (SSO), simultaneous
switching input (SSI), intersymbol interference (ISI), and board-level crosstalk for
Altera® flip-chip device families such as Stratix® IV and Arria® II FPGAs for typical
user implementation of external memory interfaces following good board design
practices.
Introduction
The widening performance gap between FPGAs, microprocessors, and memory
devices, along with the growth of memory-intensive applications, are driving the
need for faster memory technologies. This push to higher bandwidths has been
accompanied by an increase in the signal count and the signaling rates of FPGAs and
memory devices. In order to attain faster bandwidths, device makers continue to
reduce the supply voltage.
Initially, industry-standard DIMMs operated at 5 V. However, due to improvements
in DRAM storage density, the operating voltage was decreased to 3.3 V (SDR), then to
2.5 V (DDR), 1.8 V (DDR2), 1.5 V (DDR3), and 1.35 V (DDR3) to allow the memory to
run faster and consume less power. Plans are currently underway for DDR4 chips,
which are expected to run at voltages between 1.2 V and 1.0 V.
Because of this reduction in operating voltage and timing budgets, there is a higher
probability that an error may occur if the designer does not pay sufficient attention to
the system design, as via breakout layers, board trace spacing, pin assignment, and
power delivery network design all have a direct impact on the amount of timing
uncertainty seen by the receiver.
Figure 1 illustrates the industry trend of increasing peak bandwidth while comparing
various SDRAM technologies.
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January 2011
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Page 2
Source Synchronous Timing
Figure 1. Peak Bandwidth Comparison of Various Memory Technologies
12000
10600
10000
Bandwidth (MB/s)
8500
8000
6400
6000
6400
667
MHz
5328
533
MHz
4256
4000
3200
2656
2000
528
0
66
800
1064
133 200
MHz MHz
SDRAM
1600
1200
400
300 MHz
MHz
2128
1600
266
133
100 MHz MHz
MHz
RDRAM
266
200 MHz
MHz
DDR
333
MHz
400
MHz
DDR2
400
MHz
DDR3
Source Synchronous Timing
In source synchronous interfaces, the source of the clock is the same device as the
source of the data. Mainstream memory interfaces, such as DDR, DDR2, DDR3,
RLDRAM II, and QDRII are all source synchronous. In DDR, DDR2, and DDR3, a
bidirectional clock, or data strobe (DQS/DQS#), is used for both read and write
operations, while in RLDRAM II and QDR II, unidirectional clocks such as
DQ/DQ#/QK/QK# and K/K#/CQ/CQ# (respectively) are used.
Figure 2 shows a block diagram of a basic source-synchronous interface.
Figure 2. Source Synchronous Interface
Transmitter
Receiver
Data
Clock
Clock Signal
In DDR applications, the data strobe is edge aligned during a read operation (a data
transfer from the memory device to the FPGA) and center aligned during a write
operation (a data transfer from the FPGA to the memory device). When a strobe is
edge aligned with the data, the receiving device shifts the strobe as necessary to
capture the data. However, in the center-aligned example, the receiving device
directly uses the shifted clock to capture the data.
Figure 3 shows both edge-aligned and center-aligned data transfers.
Modeling System Signal Integrity Uncertainty Considerations
January 2011 Altera Corporation
SI Mechanisms and Timing Uncertainties
Page 3
Figure 3. Edge Aligned and Center Aligned Data Transfers
Center Aligned
Edge Aligned
Clock
Clock
Data
Data
In source-synchronous SDR interfaces, one edge of the clock, typically the rising edge,
transfers the data. The time required to transmit one bit, known as the unit interval
(UI), is equal to the period of the clock. In source-synchronous DDR interfaces, data is
transferred on both edges of the clock, as shown in Figure 4. The UI is equal to half the
period of the clock, assuming a 50/50 duty cycle.
Figure 4. SDR and DDR UI Definitions
DDR
SDR
Clock
UI
UI
Clock
Data
Data
Timing margins for chip-to-chip data transfers are defined by Equation 1:
Equation 1.
<Margin> = <Bit Period (UI)> - <Transmitter Uncertainties> - <Receiver Requirements> - <tEXT>
Where:
(1) Transmitter uncertainties include the timing difference between the fastest and slowest output edges on data signals,
tCO variation, clock skew, and jitter. Transmitter channel-to-channel skew (TCCS) accounts for the transmitter
uncertainties.
(2) The receiver requirements consist of a period of time during which the data must be valid to capture it correctly. The
receiver sampling window (SW) accounts for all the receiver requirements.
(3) tEXT specifies the board level skew across the data and clock traces. This is the maximum board trace variation
allowed between any two signal traces.
SI Mechanisms and Timing Uncertainties
The amount of push-out and pull-in for a given design due to simultaneous switching
noise (SSN) on the outputs and inputs (SSO and SSI) depends on the choices made
during the layout of the PCB. The key parameters responsible for the SI timing
uncertainty include the following:
January 2011
■
PCB via length
■
PCB power distribution network (PDN) design
■
I/O buffer drive strength and slew rate
■
Board trace crosstalk
■
ISI
■
Voltage reference (VREF)/termination voltage (VTT) variations
■
Receiver I/O termination
Altera Corporation
Modeling System Signal Integrity Uncertainty Considerations
Page 4
SI Mechanisms and Timing Uncertainties
When describing SSN in a system, it is useful to define the following terms:
■
Victim pin is the pin of interest.
■
Aggressor pins are pins other than the pin of interest that are transitioning and
causing noise to be injected onto the victim pin.
■
SSN is a noise voltage induced onto a victim I/O pin due to the switching
behavior of other aggressor I/O pins in the device. The SSN results in both voltage
and timing noise on the victim signal.
Figure 5 shows the two types of timing variations caused by SSO noise. Timing
push-out is caused when the victim signal is switching in the same direction as the
aggressor signals (Case A and C). Timing pull-in is caused when the victim signal is
switching in the opposite direction as the aggressor signals (Case B and D).
Figure 5. Timing Push-Out and Pull-In Due to SSO and SSI
At Quiet Condition
At Noisy Condition
At Noisy Condition
Timing
Push-Out
Case A
At Quiet Condition
Timing
Pull-In
Case B
Case C
Victim Pin Signal
Aggressor Pin Signal
Case D
The SSN seen is due to two physical mechanisms:
■
Mutual inductive coupling
■
Delta-I noise in the PDN
Inductive coupling is often the dominant mechanism for SSN, and is governed by
Equation 2.
Equation 2.
V = M  di  dt
Where:
(1) M is mutual inductive coupling.
(2) di/dt is the derivative of current over time.
Inductive coupling occurs when current from one conductor (aggressor) generates a
magnetic field that is coupled to another conductor (victim) and generates a voltage
across it. This effect grows with the number of switching outputs as:
V1 = M12 × di2 / dt + M13 × di3 / dt + …
Therefore, the larger the number of simultaneously switching buffers, the larger the
SSN due to mutual inductance.
f For more information about SSN coupling, refer to the FPGA Design for Signal and
Power Integrity conference paper.
Modeling System Signal Integrity Uncertainty Considerations
January 2011 Altera Corporation
SI Mechanisms and Timing Uncertainties
Page 5
Most inductive crosstalk occurs in the vertical structures rather than in the horizontal
transmission line structures. Examples of vertical coupling structures include C4
solder bumps, package vias, solder balls (package pins), PCB vias, and pins in a
DIMM connecter. The magnitude of inductive coupling is proportional to the parallel
length of the aggressor and victim signals. All vertical structures contribute some
amount of inductive coupling. However, most of the coupling occurs at the interface
between the FPGA package and the PCB in the PCB break-out via field, where the
parallel path is the longest between aggressors and victims.
The other dominant source of inductive coupling is the via field region under the
DIMM or the discreet memory device. Noise is inductively coupled from the
aggressor to the victim conductors during the aggressor rise and fall time and is not
coupled at any other time.
The value of the mutual inductance, M, which affects the amount to which the
different vias are coupled, is a function of the self inductance (length), L, of each via
and the coupling, k, between the vias. The coupling is, among other things, a function
of the distance between the vias, thus causing vias that are closer together to have a
larger mutual inductance between them. The designer must pay attention to the via
break out during layout to minimize the amount of coupling.
Figure 6 shows the important components of the various coupling mechanisms on a
memory system topology.
Figure 6. Noise Coupling Mechanisms
VTT Uncertainty
Customer
Board
Mutual Inductive Coupling
VTT
Altera Die in Package
VCC_IO
VCC_IO
VCC_IO
OPD
50 Ω
GND
50 Ω
ODC
Trace Coupling
GND
Drivers and
Receivers
C4
Bumps
Package
Traces
GND
Balls and
Vias
PCB
Trace
Termination
Receiver
Delta - I Noise in PDN
Delta-I noise in the PDN is caused when multiple output drivers switch
simultaneously and induce voltage changes in the chip and package PDN. This noise
manifests as a voltage drop on the power rail and a voltage spike on local GND
relative to the system GND. These changes in voltage are related to the amount of
loop inductance present in the PDN and the amount of current sunk by each
switching output, determined by Equation 3.
Equation 3.
V = L  di  dt
January 2011
Altera Corporation
Modeling System Signal Integrity Uncertainty Considerations
Page 6
SI Mechanisms and Timing Uncertainties
Loop inductance in the PDN is comprised of the inductance of the on-chip PDN, the
inductance associated with the package plane, vias and balls, the inductance
associated with the PWR and GND vias in the PCB breakout region, and the loop
inductance of the PCB planes. The larger the inductance in the PDN, the larger the
change in voltage. Furthermore, the larger the number of outputs switching at the
same time, the larger the value of di/dt and therefore, a larger value of PDN noise.
Similar to signal vias, the longer the lengths of the PWR and GND vias, the higher the
PCB loop’s inductance contribution to the overall PDN inductance.
In addition, and similar to inductive coupling, delta-I noise only occurs during the
signal transition, as this is the only time where the current changes as a function of
time. Delta-I noise does not occur in time frames where the driver current is constant
because there is no di/dt to generate the noise.
The di/dt of a switching I/O depends on the I/O buffer’s drive strength and the slew
rate setting enabled by the buffer. Stratix IV and Arria II FPGAs offer a variety of drive
strengths for each supported I/O standard. The I/O buffer drive strength of a given
driver is a measurement of how much current the driver launches on a given load. It
can also determine the largest load that can be driven at a certain speed, without
affecting the integrity of the transmitted signal. In other words, a stronger driver is
able to drive larger loads and longer transmission lines.
However, it is not always a good idea to simply choose the strongest driver because it
is able to drive larger loads and longer transmission lines. Stronger drivers launch
larger currents, and larger currents imply larger crosstalk, timing pull-out and pull-in
due to SSN, and power consumption. A stronger driver might provide a larger noise
margin but also generates a larger noise that impacts timing. Because choosing the
right driver directly affects the quality of the signal, it is important to choose the
minimum drive strength able to drive the load connected to the output of the FPGA.
Figure 7 shows the drive strength effects on the output signal when using a
transistor-to-transistor logic (TTL) standard that toggles from rail to rail. SSTL and
HSTL I/O standards behave differently because of the presence of pull-up resistors.
Figure 7. I/O Drive Strength Impact on the Output Signal
Voltages (lin)
Drive Strength Effects
1
8 ma
500 m
4 ma
2 ma
0
20 n
30 n
Time (lin) (TIME)
Modeling System Signal Integrity Uncertainty Considerations
January 2011 Altera Corporation
SI Mechanisms and Timing Uncertainties
Page 7
The I/O buffer slew rate determines the maximum rate of change of the output signal.
In other words, it determines the speed of the rising and falling times of the output
signal. Stratix IV and Arria II FPGAs have different slew rate settings that allow the
designer to modify the duration of the rise and fall times.
The drive strength specifies how much current the driver sources and sinks; the slew
rate specifies how fast the buffer sources and sinks the current. Together, these two
settings determine the rise and fall times of the output signal. The rise and fall times
are set by the process technology.
Figure 8 shows the rising edge of the output signal under four different settings. The
designer can choose the one that is optimal for the design based on the timing noise
trade off.
Figure 8. I/O Slew Rate Impact on the Output Signal
Slow
Medium
Medfast
Fast
Trace-to-trace coupling can result in board-level crosstalk, causing a timing pull-in or
push-out on the victim signal. The crosstalk results in a change in the effective
characteristic impedance and the propagation velocity of the trace. Additionally, it can
induce noise voltage onto the victim trace. The amount of crosstalk seen on the victim
trace depends on the number of toggling aggressors, the aggressor data pattern, the
air gap separation between the victim and aggressor traces, and the toggling rate of
the aggressor signals.
Trace-to-trace coupling is caused by board real estate constraints when fanning out
traces to the routing layers underneath the PCB via breakout region. After breakout,
the air gap between the traces should be increased to minimize coupling. A good rule
is to have a 3H air gap between the traces, where H is the dielectric height between the
trace and the nearest GND plane. Minimize H so that the trace couples strongly to the
GND reference plane and less to the adjacent signals. During layout, route with short
parallel sections and minimize long coupled sections between nets.
The traces on a PCB are bandwidth limited and behave like a low-pass filter. The lowpass filtering smears the transmitted signal, over time causing the effect of a bit period
(UI) to spread across the adjacent bit periods when a sequence of data bits is
transmitted (ISI).
January 2011
Altera Corporation
Modeling System Signal Integrity Uncertainty Considerations
Page 8
Quartus II Version 9.1 Timing Model Assumptions
ISI is pattern dependent and can result in a timing uncertainty known as
pattern-dependent jitter or data-dependent jitter. The skin effect of a conductor and
the dielectric loss is responsible for ISI. Reflections from poorly terminated loads can
also be a source of ISI. As frequency increases, dielectric loss is the dominant factor in
high-frequency attenuation because its effect is proportional to the frequency, where
the skin effect is proportional to the square root of frequency.
All PCB laminate materials have a specific dielectric constant and a loss tangent value.
Materials with a high loss tangent often see a deterioration of the signal with
frequency. Low-cost materials such as FR-4 have a high loss tangent, which results in
a large attenuation of the signal at high frequency. To minimize ISI, design the PCB
using a dielectric material with a lower loss tangent value based on the application
requirements. Dielectric materials with a lower loss tangent cost more than materials
with higher loss tangent.
Timing uncertainty is also caused by noise on the VREF or VTT power rail, offset of
the VTT relative to the VREF, drift of VREF or VTT over voltage and temperature, and
an external component mismatch. Stratix IV and Arria II FPGAs have calibration
circuits to ensure that the strobe signal stays in the center of the data valid window by
calibrating for voltage (V) and temperature (T) over time.
f For more information about the various calibration techniques for high-bandwidth
source-synchronous interfaces, refer to the Calibrating Techniques for High-Bandwidth
Source-Synchronous Interfaces conference paper.
The choice of receiver I/O termination can also result in system uncertainty because
non-optimal receiver termination may result in the signal being reflected back and
forth onto the transmission line, which can cause degradation in the signal edge rate
seen at the receiver. Choose the optimal on-die termination (ODT) value based on the
characteristic impedance of the traces on the PCB.
Quartus II Version 9.1 Timing Model Assumptions
The Quartus II software accounts for the timing uncertainty from many of the SI
mechanisms when analyzing timing for external memory interfaces. This feature in
the timing model applies to designs using Stratix IV and Arria II FPGAs that use
flip-chip technology for the package in Quartus II software version 9.1 and later.
For these families, the timing model assigns a timing uncertainty parameter due to
SSO and SSI based on mechanisms that can influence timing push-out and pull-in.
The timing model makes certain assumptions for PCB via length, PDN design, I/O
buffer drive strength and slew rate, board trace crosstalk, ISI, VREF/VTT variations,
and receiver I/O termination to reflect a typical memory interface application for the
analysis. The timing uncertainty values are based on simulations and system-level
characterization for the assumed parameters.
On a typical mainstream memory interface, a data signal strobe is associated with a
number of data bits, usually eight, but can vary from four to 36 bits. When the FPGA
writes to the memory device, time uncertainties include contributions from the
numerous internal FPGA circuits including the following:
■
Location of the DQ and DQS output pins
■
Width of the DQ group
Modeling System Signal Integrity Uncertainty Considerations
January 2011 Altera Corporation
Conclusion
Page 9
■
PLL clock uncertainties, including phase jitter between different output taps used
to center-align the DQS with respect to the DQ pins
■
Clock skew across the DQ output pins and between the DQ and DQS output pins
■
Package skew on the DQ and DQS output pins
■
Push-out and pull-in on the output pins due to multiple DQ and DQs pins
switching simultaneously at the same time (SSO)
Conclusion
Though the Quartus II software takes into account the timing uncertainty due to
various SI effects, such as SSO, SSI, ISI, and crosstalk, for both read and write paths,
the amount of uncertainty that the Quartus II software assumes is based on a typical
user implementation for external memory interfaces following good board design
practices. Any variations, such as designing the PCB with very deep signal vias, very
deep power and GND vias, minimal trace-to-trace spacing, and using a high-loss
tangent dielectric material for board design, lead to a higher amount of uncertainty.
In situations where a PCB design may deviate significantly from best practices and
the typical application assumed in the Quartus II timing model, Altera recommends
that designers complete further analysis in simulation using the appropriate package,
PCB, and I/O models. In most cases, the assumptions and techniques Quartus II
timing model uses for timing closure for external memory interfaces lead to an
accurate assessment of the interface performance.
January 2011
Altera Corporation
Modeling System Signal Integrity Uncertainty Considerations
Page 10
Further Information
Further Information
■
Altera’s Signal Integrity Center:
www.altera.com/technology/signal/sgl-index.html
■
FPGA Design for Signal and Power Integrity, DesignCon 2007:
www.altera.com/literature/cp/cp-01023.pdf
■
Calibrating Techniques for High-Bandwidth Source-Synchronous Interfaces, DesignCon
2007:
www.altera.com/literature/cp/cp-01024.pdf
Acknowledgements
■
Ravindra Gali, High-Speed I/O Applications Engineering, Altera Corporation
■
Zhi Wong, High-Speed I/O Applications Engineering Manager, Altera
Corporation
■
Navid Azizi, Software Engineering, Altera Corporation
■
John Oh, High-Speed I/O Applications Engineering Manager, Altera Corporation
■
Arun VR, Memory I/O Applications Engineering, Altera Corporation
Document Revision History
Table 1 lists the revision history for this application note.
Table 1. Document Revision History
Date
January 2011
Version
1.0
Changes
Initial release.
Modeling System Signal Integrity Uncertainty Considerations
January 2011 Altera Corporation
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