A Bi-Directional Quasi-Optical Lens Amplifier

A Bi-Directional Quasi-Optical Lens Amplifier
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 45, NO. 12, DECEMBER 1997
A Bi-Directional Quasi-Optical Lens Amplifier
Stein Hollung, Amanda E. Cox, and Zoya Basta Popović, Member, IEEE
Abstract—A 24-element bi-directional quasi-optical lens amplifier array is presented. The lens amplifier array is designed for
-band, and operates in transmission mode. Single-pole doublethrow (SPDT) switches are used to switch between transmit and
receive amplifiers. The lens amplifier array demonstrates gains
of 5.5 dB at 10.1 GHz in receive mode, and 2 dB at 10.2 GHz in
transmit mode, with more than 15-dB ON/OFF isolation. Several
applications for the lens amplifier array are demonstrated: a
quasi-optical transceiver front-end, reduction of multipath fading, and a multiuser frequency reuse application.
X
I. INTRODUCTION
S
EVERAL transmission-mode plane-wave-fed quasioptical amplifiers for microwave and millimeter-wave
power combining have been presented to date [1]–[3]. Each
of these amplifiers is fed with a plane wave from a source in
the far field. In order to improve feed efficiency without using
external lenses, lens amplifier arrays were developed [4], [5].
In a transmission-mode lens amplifier, the feed is placed in
the near field along the focal surface, thereby minimizing
diffraction loss. In transmission, a lens amplifier can provide
high effective radiated power. A lens amplifier can also be used
in reception, offering high dynamic range because the noises
contributed from the individual amplifiers are uncorrelated [6].
In reception, a plane wave is received, amplified, and focused
onto a receiver [7]. The transmit and receive functions of a
lens amplifier can be combined to form a quasi-optical
module, as shown in Fig. 1. In this paper, we present a 24element bi-directional quasi-optical lens amplifier with a unit
cell, as shown in Fig. 2. Section II describes the unit cell,
Section III discusses the array, and Section IV describes some
applications for this lens array.
II. SINGLE ARRAY ELEMENT
A bi-directional transmission amplifier element, designed
to operate around 10 GHz, was fabricated on a 0.507-mmthick Duroid substrate with
. A circuit schematic of
the array element is shown in Fig. 3. Orthogonally polarized
antiresonant slot antennas were used at the input and output
because of their wide-bandwidth and ease of fabrication with
Manuscript received April 1, 1997; revised July 31, 1997. This work was
supported by the U.S. Army Research Office under grant DAAH 04-96-10343, by the National Science Foundation under a Presidential Faculty Fellow
Award, and by the DOD (ARO) DURID program under Grant DAAH 04-951-0444 and Grant DAAG 55-97-1-0027.
The authors are with the Department of Electrical and Computer Engineering, University of Colorado at Boulder, CO 80309 USA.
Publisher Item Identifier S 0018-9480(97)08331-2.
Fig. 1. A quasi-optical T =R module consisting of a bi-directional amplifier
and a grid oscillator/mixer.
Fig. 2. A bi-directional X -band quasi-optical array amplifier element. The
slot antennas are 2.5 cm long and 2 mm wide, and the unit cell dimensions
3.7 cm.
are 3.5 cm
2
microstrip feed lines. Fig. 4 shows the measured and simulated
return loss for a single slot antenna. A 2:1 VSWR bandwidth
of about 40% was measured. The measured - and
plane radiation patterns for a single slot antenna are shown
in Fig. 5. The -plane pattern shows a null in the boresight direction as expected for a center-fed second resonance
slot. Both the characteristic impedance of the microstrip and
the antenna input impedance are 65 , to avoid additional
matching sections for the antennas. Two single-pole doublethrow (SPDT) switches are used to switch between a generalpurpose MESFET amplifier stage for transmit mode and
0018–9480/97$10.00  1997 IEEE
HOLLUNG et al.: BI-DIRECTIONAL QUASI-OPTICAL LENS AMPLIFIER
2353
M
Fig. 3. Circuit schematic of the unit cell. Both amplifiers are matched for
gain with matching circuits g at the gates and Md at the drains.
Fig. 5. Measured radiation patterns for a single antiresonant slot antenna at
10 GHz. 0 represents the circuit side and 180 represents the ground plane.
E -plane (–) and H -plane (- -).
Fig. 4. Measured (–) and simulated (- -) return loss for a single slot antenna.
The simulations were performed with the computer-aided design (CAD)
package Ensemble.
a pseudomorphic high-electron-mobility transistor (PHEMT)
amplifier stage for receive mode. Both amplifiers are matched
for gain in this unit cell. The receive amplifier is stabilized
with a 200- chip resistor from gate to source. Two p-i-n
diodes are used for each switch. The dc bias for the diodes is
supplied through the slot antenna feed lines. Both transistors
share the same drain bias to reduce the number of bias lines.
Measurements for the unit cell are conducted using -plane
horns placed in the far field and connected to an HP 70 820A
Microwave Transition Analyzer. In transmission, an incoming
plane wave from a vertically polarized horn is amplified by
the transmit amplifier, reradiated as a horizontally polarized
plane wave, and received by a horizontally polarized horn.
In reception, an incoming plane wave from the horizontally
polarized horn is amplified by the receive amplifier, reradiated
as a vertically polarized plane wave, and received by the
vertically polarized horn. Polarizers are inserted at a quarterwavelength on each side of the unit cell to improve the gain.
The received power is measured for both the transmit and
receive amplifiers over a range of frequencies. The gains
contributed by the amplifiers are approximately calculated
Fig. 6. Measured gain of the transmit (–) and receive (- -) amplifier in a
unit cell.
from the Friis formula and plotted in Fig. 6. A gain of 5.5
dB for the slot antennas with polarizers was used for the
calculations, based on the measurement shown in Fig. 5. The
transmit amplifier has a maximum measured gain of 10 dB at
11 GHz. The receive amplifier has a measured gain of 7.5 dB
at 10.8 GHz. Measured ON/OFF isolation of approximately 10
and 20 dB are seen for the receive and transmit amplifiers,
respectively.
III. BI-DIRECTIONAL LENS AMPLIFIER ARRAY
The bi-directional lens amplifier array consists of 24 elements in a triangular lattice with four elements in the first
and fifth row, five elements in the second and fourth row, and
six elements in the third row, as shown in Fig. 7. Lensing
delay lines are incorporated between the antenna pairs in
each unit cell. The delay-line lengths were calculated using
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 45, NO. 12, DECEMBER 1997
(a)
Fig. 7. Photograph of the circuit side of the 24-element bi-directional
amplifier array. Orthogonally polarized slot antennas are located in the ground
plane, as indicated in Fig. 2.
the design equations for a lens with one degree of freedom
[8]. The focal distance of the array is 27.5 cm and the
corresponding -number is 1.5. Each element in the array
measures 3.5 cm 3.7 cm. The gate bias lines are horizontal,
while the drain and diode bias lines are diagonally distributed.
An identical passive array, with the amplifiers replaced by 65through lines between the two slots, is used for calibration.
For reception, a transmit horn antenna, located in the far
field of the array, provides a horizontally polarized incident
plane wave to the array. The array receives, amplifies, and
reradiates a vertically polarized wave to a receive horn located
at the focal point. For transmission, the signal path is reversed.
Two polarizers are inserted at a quarter-wavelength distance
on each side of the array. The passive array is measured to
provide a reference for the measurements. The measurements
are normalized to a through measurement with the transmit
26 cm aperture
and receive horns co-polarized. A 22 cm
cut out of absorbing material is used for this calibration.
Measurements were first performed with the ten central
elements populated. A measured ON/OFF isolation of 25 dB for
both receive and transmit modes with ten elements is seen. In
receive mode, a maximum power gain of about 10 dB relative
to the passive array with ten elements connected is measured
at 9.4 GHz. In transmit mode, the measured power gain is
about 5 dB at 10.2 GHz.
These measurements were repeated with the 24 elements
populated (the results are plotted in Fig. 8). In receive mode,
a maximum power gain of 5.5 dB relative to the passive array
with all 24 elements connected is measured at 10.1 GHz. In
transmit mode, the measured power gain is 2 dB at 10.2 GHz.
(b)
Fig. 8. (a) Measured ON/OFF ratio of the 24-element active and passive
array in transmit mode, with polarizers at input and output. (b) Measured
ON/OFF ratio of the 24-element active and passive array in receive mode, with
polarizers at input and output. The measurements are normalized to a through
measurement with a rectangular aperture surrounded by absorbing material.
The solid lines represents the biased on state, the dotted lines the biased off
state, and the dashed lines shows the measured results for the passive array.
A possible explanation for reduction in gain when increasing
the size of the array from 10 to 24 elements is increased
nonuniformity of the amplifiers across the array. This could be
due to fabrication, unmatched devices, and coupling between
unit cells, as well as a larger fraction of edge elements in the
24-element array as compared to the ten-element array. The
absolute power gain relative to the rectangular aperture is 3.9
dB in reception and 0.9 dB in transmission. Measured ON/OFF
isolation ratios of approximately 15 dB in receive mode, and
more than 20 dB in transmit mode are seen. Notice that the
physical area of the active antenna array covers only 55% of
the aperture, so the absolute power gains are expected to be
higher than those measured. Also note that due to the limited
performance bandwidth of the amplifiers, the active array is
lossy compared to the passive array outside the amplifier
gain range.
The 24-element lens amplifier has a measured beamwidth of
about 10 in both - and -plane, with sidelobe levels of less
HOLLUNG et al.: BI-DIRECTIONAL QUASI-OPTICAL LENS AMPLIFIER
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Fig. 10. A 10-kHz square-wave signal received by a quasi-optical receiver
front-end and demodulated. A transmitted amplitude modulated 10.1-GHz
carrier is received, mixed to a 290-MHz IF, and demodulated using a vector
signal analyzer.
(a)
2
Fig. 11. Test setup for the multipath experiment. A 45 cm 30 cm metallic
mirror located parallel to the optical axis in front of the 24-element lens
amplifier is translated in 3-mm steps from the axis. For each step, the mirror
was rotated through a set of angles.
in the grid lock [9]; and 3) self-oscillating fundamental or
harmonic mixing in reception [7].
B. Multipath Fading Reduction
(b)
H
Fig. 9. (a) Measured (- -) and simulated (—) -plane radiation pattern for
the receive amplifier at 10.1 GHz. (b) Measured (- -) and simulated (—)
E -plane radiation pattern for the receive amplifier at 10.1 GHz. The theoretical
patterns were calculated using the measured antenna pattern of a single slot
antenna.
than 10 dB, as shown in Fig. 9. The theoretical patterns were
calculated using the measured radiation pattern of a single slot
antenna.
IV. APPLICATIONS
A. A Quasi-Optical Receiver
A complete quasi-optical receiver front end can be formed
by cascading the lens amplifier with a 10.39-GHz grid oscillator/mixer, as shown in Fig. 1. A transmitted 10.10-GHz carrier,
amplitude modulated with a 10-kHz square wave, is amplified
and focused by the ten-element lens amplifier toward the selfoscillating mixer located at the focal point. The 290-MHz IF
present on the bias lines of the grid oscillator is demodulated
using an HP 89441A Vector Signal Analyzer. The recovered
10-kHz square wave is shown in Fig. 10. The quasi-optical
grid oscillator is just one example of a feed type. Possible
advantages of using a grid oscillator are: 1) power combining
at the source level; 2) reduced phase noise as more devices
Reduction of multipath fading using a lens amplifier array
can be demonstrated by a simple measurement. A 45 cm
30 cm metallic mirror located parallel to the optical axis
in front of the amplifier array was translated in 3-mm steps
perpendicular to the optical axis, as shown in Fig. 11. For
each step, the mirror was rotated through a set of angles. The
received power was measured for all mirror positions with
and without the lens amplifier.
The measured maximum fades of a 10.1-GHz carrier signal,
with and without the 24-element lens amplifier, and normalized
to the received signal without the mirror inserted, are shown
in Fig. 12. Maximum fading nulls of less than 4 dB and
greater than 50 dB were measured with and without the lens
amplifier, respectively. This simple measurement show that a
lens amplifier can also provide a significant improvement of
multipath fading effects due to the increased directivity of the
receiver [10].
C. A Multiuser Application
A multiuser frequency reuse experiment demonstrates how
two separate signals incident from different angles can be
received independently at different locations along the focal
surface of the 24-element lens amplifier array. Beamsteering
patterns are measured at 10.1 GHz for receive locations at
20 , 0 , and 20 along the focal surface in the -plane of
the array, as shown in Fig. 13. The normalizing power levels
of the main lobes are within 2 dB. The measured sidelobe
levels for the 0 and 20 receive locations are below 10 dB.
2356
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 45, NO. 12, DECEMBER 1997
Fig. 14. Test setup for the multiuser experiment. Square-wave signals at
fA 50 kHz and fB = 150 kHz are frequency modulated on a 10.1-GHz
carrier and input at 0 and 20 from the optical axis, respectively. The
two signals are received by horns located on the focal surface behind the
24-element lens amplifier array, and demodulated using a vector signal
analyzer.
=
(a)
(a)
(b)
Fig. 12. Measured maximum multipath fading nulls of a 10.1-GHz carrier
signal. The received power is normalized to through measurements without
the mirror inserted. (a) With and (b) without the 24-element lens amplifier
array inserted. Notice the different vertical scales.
(b)
Fig. 15. (a) Received and frequency shift-keying (FSK) demodulated 50-kHz
signal for a receiver located at 0 . (b) Received and FSK demodulated
150-kHz signal for a receiver located at 20 . The signals are demodulated
using a vector signal analyzer.
The interfering signal appears as an additional ripple in the
demodulated signal.
V. CONCLUSION
Fig. 13. Beamscanning in E -plane at 10.1 GHz for the 24-element lens
amplifier array in receive mode. Receive horn at (–) 0 , (- -) 20 , and ( )
20 locations are shown. The normalizing power levels of the main lobes are
within 2 dB.
0
111
A multiuser experiment, with two incident signals at 0 and
20 and with the same incident power levels is performed,
as shown in Fig. 14. The received power of an interfering
signal originating 20 from the desired signal has a measured
relative power of about 10 dB in this setup. Square-wave
signals at
kHz and
kHz are frequency
modulated on a 10.1-GHz carrier and incident from 0 and
20 , respectively. The demodulated signals received at the 0
and 20 positions on the focal surface are shown in Fig. 15.
A bi-directional quasi-optical lens amplifier array is presented. In transmit mode, the vertically polarized antiresonant
slot antennas receive an input wave from a focal point and
SPDT switches route it through transmit amplifiers to the horizontally polarized output slots. In reception, the horizontally
polarized slots couple the input signal through the receive
amplifiers to the vertically polarized slots. Amplifier gains
of 5.5 dB at 10.1 GHz in receive mode, and 2 dB at 10.2
GHz in transmit mode are measured. The functionality of
this quasi-optical approach is demonstrated by: 1) a quasioptical
module; 2) a quasi-optical AM receiver; 3)
significant multipath fading null reduction; and 4) a multiuser
with frequency reuse FSK data link.
ACKNOWLEDGMENT
The authors wish to thank Rogers Corporation, Chandler,
AZ, for substrate donations, and Boulder Microwave Technologies, Inc., Boulder, CO, for use of Ensemble. The authors
also thank the DOD (ARO) DURID program that enabled the
purchase of most of the equipment used for this work.
HOLLUNG et al.: BI-DIRECTIONAL QUASI-OPTICAL LENS AMPLIFIER
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Stein Hollung was born in Oslo, Norway, on March
7, 1970. He received the electrical engineering degree from Oslo College of Engineering, Oslo, Norway, in 1992, the B.S. and M.S. degrees in electrical
engineering from the University of Colorado at
Boulder, in 1994 and 1995, respectively, and is currently working toward the Ph.D. degree in electrical
engineering.
Amanda E. Cox received the B.S. degree in electrical engineering, the M.S.
degree in aerospace engineering sciences from the University of Colorado at
Boulder in 1983 and 1996, respectively, and is currently working toward the
Ph.D. degree in electrical engineering.
She served as a Project Engineer at both Teledyne MEC, Palo Alto,
CA, developing traveling-wave tube amplifiers for satellite communications
and radar applications, and at TIW Systems, Sunnyvale, CA, developing
microwave subsystems for satellite communications and radio astronomy. Her
research interests include remote sensing, radio astronomy, GPS, and satellite
systems.
Zoya Basta Popović (S’86–M’90), for a photograph and biography, see this
issue, p. 2174.
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