SpectreRF_LNA_MMSIM141
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SpectreRF Workshop
LNA Design Using SpectreRF
MMSIM 14.1
September 2014
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Contents
LNA Design Using SpectreRF............................................................................................ 3 Purpose............................................................................................................................ 3 Audience ......................................................................................................................... 3 Overview ......................................................................................................................... 3 Introduction to LNAs .......................................................................................................... 3 The Design Example: A Differential LNA ......................................................................... 4 Testbench ........................................................................................................................ 5 LNA Measurements and Design Specifications ................................................................. 7 Example Measurements Using SpectreRF........................................................................ 14 Lab 1: Small Signal Gain (SP) ...................................................................................... 15 Lab 2: Noise Simulation ( hb and hbnoise ) ................................................................. 29 Lab 3: Gain Compression and THD (Xdb and Swept hb) ............................................ 37 Lab 4: IP3 Measurement---hb/hbac analysis ................................................................ 48 Lab 5: IP3 Measurement---hb Analysis with Two Tones ............................................. 55 Lab 6: IP3 Measurement---Rapid IP3 using AC analysis............................................. 59 Conclusion ........................................................................................................................ 63 References ......................................................................................................................... 63 September 2014
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LNA Design Using SpectreRF
Note: The procedures described in this workshop are deliberately broad and generic. Your
specific design might require procedures that are slightly different from those described here.
Purpose
This workshop describes how to use new hb analysis in the Virtuoso Analog Design
Environment (ADE) to measure parameters that are important in design verification of low
noise amplifiers (LNAs).
The hb analysis is one new GUI for the harmonic balance analysis, and provides a simple,
usable periodic steady-state analysis for the users. In the hb analysis, one tone or multi-tones
may be listed in the Tones field, and users needn’t separate them as PSS and QPSS did before.
In addition, two small signal analyses, hbac and hbnoise are also included. The hbac and
hbnoise analyses should be same as the old PAC and PNOISE analyses.
Audience
Users of SpectreRF in the Virtuoso Analog Design Environment.
Overview
This workshop describes a basic set of the most useful measurements for LNAs.
Introduction to LNAs
The first stage of a receiver is typically a low-noise amplifier (LNA), whose main function is
to set the noise boundary as well as to provide enough gain to overcome the noise of
subsequent stages (for example, in the mixer or IF amplifier). Aside from providing enough
gain while adding as little noise as possible, an LNA should accommodate large signals
without distortion, offer a large dynamic range, and present good matching to its input and
output. Good matching is extremely important if a passive band-select filter and image-reject
filter precedes and succeeds the LNA, because the transfer characteristics of many filters are
quite sensitive to the quality of the termination.
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The Design Example: A Differential LNA
The LNA measurements described in this workshop are calculated using SpectreRF in ADE.
The design investigated is the differential low noise amplifier shown below:
The following table lists typically acceptable values for the performance metrics of LNAs
used in heterodyne architectures.
Measurement
Acceptable Value
NF
2 dB
IIP3
-10 dBm
Gain
15 dB
Input and Output Impedance
50 Ω
Input and Output Return Los
-15 dB
Reverse Isolation
20 dB
Stability Factor
>1
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Testbench
Figure 1-2 shows a generic two-port amplifier model. Its input and output are each
terminated by a resistive port, like an amplifier measurement using a network analyzer.
Figure 1-2 A Generic Two-Port LNA
The LNA is characterized by the scattering matrix in Equation 1-1.
⎡bS ⎤ ⎡ S11 S12 ⎤ ⎡a S ⎤
⎢b ⎥ = ⎢ S S ⎥ ⎢a ⎥
⎣ L ⎦ ⎣ 21 22 ⎦ ⎣ L ⎦
where bS and bL are the reflected waves from the input and output of the LNA, a S and a L
are the incident waves to the input and output of the LNA. They are defined in terms of the
terminal voltage and current as follows
(1-1)
aS =
bS =
aL =
bL =
Vin
2 Rs
Vin
2 Rs
Vout
2 RLs
Vout
2 RLs
+
−
Rs
I in
2
Rs
2
+
−
I in
RL
2
RL
2
I out
I out
Spectre normalizes the LNA scattering matrix with respect to the source and load port
resistance. Therefore, the source reflection coefficient ΓS and load reflection coefficient ΓL
are both zero.
From network theory, the input and output reflection coefficients are expressed in Equations
1-2 and 1-3.
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S 21 S12 ΤL
1 − S 22 ΓL
(1-2)
Γin = S11 +
(1-3)
Γout = S 22 +
S12 S 21ΤS
1 − S11ΓS
The LNA scattering matrix is normalized in terms of the source and load resistance in
Equation 1-4.
(1-4)
ΓS = ΓL = 0
Thus, the input and output reflection coefficients are simply expressed in Equations 1-5 and
1-6.
(1-5)
Γin = S11
(1-6)
Γout = S 22
The main challenge of LNA design lies in the design of the input/output matching network to
render Γin and Γout close to zero so that the LNA is matched to the source and load ports.
With the knowledge of a generic LNA model, Figure 1-3 shows the testbench for a
differential LNA. The baluns used in the testbench are three-port devices. The baluns convert
between single-ended and differential signals. Sometimes, they also perform the resistance
transformation.
Figure 1-3 Testbench for a Double-Ended LNA
LNA design is a compromise among power, noise, linearity, gain, stability, input and output
matching, and dynamic range. These factors are characterized by the design specifications in
the table on page 4.
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LNA Measurements and Design Specifications
Power Consumption and Supply Voltage
You must trade off gain, distortion, and noise performance against power dissipation. Total
power dissipation for an operating LNA circuit should be within its design budget. Because
most LNAs are operated in Class-A mode, power consumption is easily available by
multiplying the DC supply voltage by the DC operating point current. Selecting the operating
point is a critical stage of LNA design which affects the power consumption, noise
performance, IP3, and dynamic range.
Gain
Three power gain definitions appear in the literature and are commonly used in LNA design.
!
!
!
GT , transducer power gain
G P , operating power gain
G A , available power gain
Besides these three gain definitions, there are three additional gain definitions you can use
to evaluate the LNA design.
!
!
!
Gumx , maximum unilateral transducer power gain
Gmax , maximum transducer power gain
Gmsg , maximum stability gain
There are also two gain circles that are helpful to the design of input and output matching
networks.
!
!
GPC, power gain circle
GAC, available gain circle
Transducer Power Gain
Transducer power gain, GT , is defined as the ratio between the power delivered to the load
and the power available from the source.
1 − ΓS
(1-9)
GT =
(1-10)
GT = S 21
2
2
1 − ΓL
2
S 21
2
1 − S11ΓS
1 − S 22 ΓL
In the test environment, from Equation 1-4, you have
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Operating power gain
Operating power gain, GT , is defined as the ratio between the power delivered to the load
and the power input to the network.
(1-11)
GP =
1
1 − Γin
S 21
2
1 − ΓL
2
2
1 − S 22 ΓL
2
In the test environment, from Equations 1-4 and 1-5, you have
(1-12)
GP =
1
1 − S11
S 21
2
2
Available power gain
Available power gain, G A , is defined as the ratio between the power available from the
network and the power available from the source.
(1-13)
GA =
1 − ΓS
2
1 − S11ΓS
2
S 21
1
2
1 − Γout
2
In the test environment, from Equations 1-4 and 1-6, you have
(1-14)
G A = S 21
2
1
1 − S 22
2
As the power available from the source is greater than the power input to the LNA network,
G P > GT , the closer the two gains are, the better the input matching is. Similarly, because the
power available from the LNA network is greater than the power delivered to the load,
G A > GT . The closer the two gains are, the better the output matching is.
Maximum Unilateral Transducer Power Gain
Maximum unilateral transducer power gain, Gumx , is the transducer power gain when you
assume that the reverse coupling of the LNA, S12 , is zero, and the source and load
impedances are conjugately matched to the LNA. That is ΓS = S11 and ΓL = S 22 . If S12 = 0 ,
from Equations 1-2 and 1-3, the input and output reflection coefficients are Γin = S11and
Γout = S 22 . Thus from Equation 1-9, you get Equation 1-15.
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(1-15)
Gumx =
1
1 − S11
2
S 21
1
2
1 − S 22
2
Maximum Transducer Power Gain
Maximum transducer power gain, Gmax , is the simultaneous conjugate matching power gain
when both the input and output are conjugately matched. ΓS = Γin and ΓL = Γout .When the
reverse coupling, S12 , is small, Gumx is close to Gmax .
Gmax =
S 21
S12
(K −
K 2 −1
)
The stability factor, K, is defined in “Stability”.
Maximum Stability Gain
Maximum stability gain, Gmsg , is the maximum of Gmax when the stability condition, K > 1, is
still satisfied.
Gmsg =
S 21
S12
Power Gain Circle
Power gain circle, GPC. From Equations 1-2 and 1-11, you can see that G P is solely a
function of the load reflection ΓL. Thus you can draw power gain contours on the Smith chart
of ΓL . The location for the peak of the contour corresponds to ΓL producing the
maximum G P . You can move the peak location by changing the design of the output
matching network. The best location for the contour peak is at the center of the Smith chart,
where ΓL = 0 .
Available Gain Circle
Available gain circle, GAC. From Equations 1-3 and 1-13, you can see that G A is solely a
function of the source reflection ΓS . Thus you can draw available gain contours on the Smith
chart of ΓS . The location for the peak of the contour corresponds to ΓS producing the
maximum G A . You can move the peak location by changing the design of the input matching
network. The best location for the contour peak is at the center of the Smith chart, where
ΓS = 0 .
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Noise in LNAs
According to the Friis equation for cascaded stages, the overall noise figure is mainly
determined by the first amplification stage, provided that it has sufficient gain. You achieve
low noise performance by carefully selecting the low noise transistor, DC biasing point, and
noise-matching at the input.
The noise performance is characterized by noise factor, F, which is defined as the ratio
between the input signal-to-noise ratio and the output signal-to-noise ratio
⎛ S ⎞
⎜ ⎟
⎝ N ⎠ in G A N in
(1-16)
F=
=
N out
⎛ S ⎞
⎜ ⎟
⎝ N ⎠ out
where N in is the available noise power from the source, N in = kTΔf , and N out is the
available noise power to the load.
According to linear noise theory, you can model the noise of a noisy two-port system with
two equivalent input noise generators: a series voltage source and a shunt current source.
This is shown in Figure 1-4.
Figure 1-4 Two-Port Noise Theory
The two noise sources are related by the correlation admittance. The noise factor, F, is
described by Equation 1-17.
(1-17)
F = Fmin +
Rn
YS − Yopt
GS
2
where R n is the equivalent noise resistance of the noisy two-port system
2
Rn =
en
4kTΔf
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The source admittance is Ys = Gs + jBs , the optimum source admittance is Yopt = Gopt + jBopt ,
and the minimum noise factor is Fmin . The optimum source admittance Yopt , the minimum
noise factor Fmin , and R n are solely determined by the two-port circuit itself.
From Equation 1-17, the noise factor, F, is a function of source admittance, YS. Thus you can
plot the noise factor contour on the source admittance Smith chart. Where Ys = Yopt , the
center of the noise factor contour corresponds to Fmin . You can move the center of the source
admittance Smith chart, Yopt , by changing the input matching network design. The best
choice is to move the center of the noise circles to the center of the Smith chart so that
Yopt = Rs .
You perform noise-matching by designing the input-matching network so that the center of
the LNA’s noise circle (NC) moves to the center of the source admittance Smith chart.
However, as previously mentioned, to maximize the available gain, you should also move the
center of the available gain circle (GAC) to the center of the source admittance Smith chart.
These two goals might turn out to be contradictory, in which case you must compromise so
that the centers of the noise circles and the gain circle are both close to the Smith chart center.
Several design topologies are available to help you to balance noise and gain matching. The
topologies include shunt-series feedback, common-gate and inductively-degenerated
common-source [3] [4].
Input and Output Impedance Matching
The input and output are each connected to the LNA with filters whose performance relies
heavily on the terminal impedance. Furthermore, input and output matching to the source and
load can maximize the gain. Input and output impedance matching is characterized by the
input and output return loss.
20 log Γin = 20 log S11
20 log Γout = 20 log S 22
You can also characterize the LNA’s input and output impedance matching by the voltage
standing wave ratio (VSWR):
VSWRin =
1 + Γin
1 − Γin
VSWRout =
1 + Γout
1 − Γout
=
1 + S11
1 − S11
=
1 + S 22
1 − S 22
Your primary design goals are to minimize the return loss and make the VSWR close to 1
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Reverse Isolation
The reverse isolation of an LNA determines the amount of the LO signal that leaks from the
mixer to the antenna. LO signal leakage arises from capacitive paths, substrate coupling, and
bond wire coupling. In a heterodyne receiver, because the LO signal is ωif away from the RF
signal, the image-reject and band-select filters and the LNA can all work together to
significantly attenuate the LO signal leaked from the VCO.
Insufficient isolation can cause feedback and even instability. Reverse isolation is
2
characterized by the reverse transducer gain power, S12 . You should minimize the reverse
transducer gain power as much as possible.
Stability
In the presence of feedback paths from the output to the input, the circuit might become
unstable for certain combinations of source and load impedances. An LNA design that is
normally stable might oscillate at the extremes of the manufacturing or voltage variations,
and perhaps at unexpectedly high or low frequencies.
The Stern stability factor characterizes circuit stability as in Equation 1-18.
2
(1-18)
Kf =
2
1 + Δ − S11 − S22
2
2 S21 S12
where B1f ( Δ) = S11S22 − S12 S21
When Kf > 1 and B1f < 1, the circuit is unconditionally stable. That is, the circle does not
oscillate with any combination of source and load impedances. You should perform the
stability evaluation for the S parameters over a wide frequency range to ensure that K
remains greater than one at all frequencies.
As the coupling ( S12 ) decreases, that is as reverse isolation increases, stability improves. You
might use techniques such as resistive loading and neutralization to improve stability for an
LNA. [2].
Equation 1-18 is valid for small-signal stability. If the circuit is unconditionally stable under
small-signal conditions, the circuit is less likely to be unstable when the input signal is large.
Aside from the two metrics Kf and B1f, you can also use the source and load stability circles
to check for LNA stability. The input stability circle draws the circle Γout = 1 on the Smith
chart of ΓS . The output stability circle draws the circle Γin = 1 on the Smith chart of Γ .
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The non-stable regions of the two circles should be far away from the center of the Smith
chart. In fact, it is better if the non-stable regions are located outside the Smith chart circles.
Linearity
Nonlinear LNAs can corrupt the RF input signal and cause the types of distortion [3]
described in Table 1-2.
Table 1-2 RF Input Signal Distortion In Nonlinear LNAs
Harmonic
Distortion
A nonlinear LNA might generate output with high order harmonics
when the input is a pure sinusoid.
Cross
Modulation
A nonlinear LNA might transfer the modulation on one channel’s
carrier to another channel’s carrier.
Blocking
In a nonlinear LNA, one large signal on one channel might
desensitize the amplification of a small signal on neighboring
channels. Many RF receivers must be able to withstand blocking
signals 60 to 70 dB greater than the wanted signal.
Gain
Compression
In a nonlinear LNA, gain decreases as input power increases
because of transistor saturation.
Intermodulation
In a nonlinear LNA, two large signals (interferers) on two adjacent
channels might generate a 3rd-order intermodulation component
which falls into the bandwidth of neighboring channels.
LNA linearity is characterized by the 1 dB compression point (P1 dB) and the 3rd order
interception point (IP3).
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Example Measurements Using SpectreRF
To test an LNA, place it into the testbenches described on page 6. You can then perform
various analyses to determine the gain, noise, power, linearity, stability, and matching
performance for the LNA.
This section demonstrates how to set up the required SpectreRF analyses and to make
measurements on LNAs. It explains how to extract the design parameters from the data
generated by the analyses.
The workshop begins by bringing up the Cadence Design Framework II environment for a
full view of the reference design:
To prepare for the workshop,
Action P-1:
cd into the ./rfworkshop directory.
Action P-2:
Run the tool virtuoso&.
Action P-3:
In the CIW window, select Tools — Library Manager….
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Lab 1: Small Signal Gain (SP)
The S Parameter (SP) analysis is the most useful linear small signal analysis for LNAs. In the
following actions, you set up an SP analysis by specifying the input and output ports and the
range of sweep frequencies.
Action 1-1:
In the Library Manager window, open the schematic view of the
Diff_LNA_test in the library RFworkshop.
Action 1-2:
Select the PORTrf source by placing the mouse cursor over it and clicking
the left mouse button. Then in the Virtuoso Schematic Editor select Edit —
Properties — Objects…. After these actions, the Edit Object Properties
window for the port cell comes up. Set up the port properties as follows:
Parameter
Value
Resistance
50 ohm
Port Number
1
DC voltage
(blank)
Source type
dc
Action 1-3:
In the Virtuoso Schematic Editing window, select Launch — ADE L.
Action 1-4:
(Optional) Choose Session — Load State in the Virtuoso Analog Design
Environment window, select Cellview in Load State Option and load state
“Lab1_sp”, then skip to Action 1-8.
Action 1-5:
In the Virtuoso Analog Design Environment window, select Analyses —
Choose….
Action 1-6:
In the Choosing Analyses window, select sp in the Analysis field of the
window.
Action 1-7:
In the S-Parameter Analysis window, in the Ports field, click Select. Then, in
the Virtuoso Schematic Editing window, in order, select the port cells, rf
(input) and load (output). Then, while the cursor is in the schematic window,
click the ESC key.
In the Sweep Range field, select Start-Stop, set Start to 1.0 G and Stop to
4.0G, set Sweep Type to Linear, select Number of Steps and set that to 50. In
the Do Noise field, select yes, set Output port to /load and Input port to /rf.
After these actions, the form looks like this:
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Note: Selecting yes under Do Noise sets up the Noise analysis. You can obtain small signal
noise when the input power level is low and the circuits are considered linear.
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The Virtuoso Analog Design Environment window looks like this:
Action 1-8:
Choose Simulation — Netlist and Run to start the simulation or click the
Netlist and Run icon in the Virtuoso Analog Design Environment window.
Action 1-9:
In the Virtuoso Analog Design Environment window, select Results —
Direct Plot — Main Form….
A waveform window and a Direct Plot Form window open.
Action 1-10: In the Analysis field, select sp. In the Function field, select GT (for
Transducer Gain). In the Modifier field, select dB10. The form looks like
this:
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Action 1-11: Click Plot. In the Function field, select GA (for Available Power Gain).
Click Plot again. In the Function field, select GP (for Operating Power
Gain). Click Plot once more.
These actions plot GT, GA, and GP in one waveform window. GT is the
smallest gain. This is expected from the discussion about “Gain” on page 7.
The power gain G P is closer to the transducer gain GT than the available gain
G A which means the input matching network is properly designed. That is,
S11 is close to zero.
Action 1-12: Go back to the Direct Plot Form. Click New Subwindow. Select Gmax (for
maximum Transducer Power Gain) and click Plot. In the Direct Plot Form
window, set Plotting Mode to Append. In the Function field, select Gmsg
(for Maximum Stability Gain). Click Plot. Select Gumx (for maximum
Unilateral Transducer Power Gain), and click Plot again.
You get the following waveforms:
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In the above plot, Gumx is very close to G max which means the reverse
coupling, S12 , is small. Obviously Gmsg is the largest of the six gains
plotted.
Action 1-13: Close the waveform window, and go back to the Direct Plot Form. In the
Function field, select GAC (Available Gain Circles). In Plot Type field,
choose Z-Smith, Sweep Gain Level (dB) at Frequency 2.4GHz from 14 to
18dB with steps set to 0.25 dB.
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Action 1-14: Click plot.
Action 1-15: In the waveform window, click New Subwindow.
Action 1-16: Go back to the Direct Plot Form. In the Function field, select GPC (Power
Gain Circles). Click Plot.
The waveforms look like this:
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The contours in the above figure are plotted for freq=2.4GHZ. In the GPC
plot, GP ≈ 16.5at ΓL = 0 . In the GAC plot, GA ≈ 16.25 at ΓS = 0 . These
results match the results G P / G A on page 20. As has been discussed, the
centers of the two contours are located close to the centers of the Smith
charts.
Action 1-17: Close the waveform window, and go back to the Direct Plot Window. In the
Direct Plot Form window, set Plotting Mode to Append. In the Function
field, choose Kf. Click Plot.
Action 1-18: In the Function field, choose B1f. Click Plot.
The Stability Curves are plotted, when Kf > 1 and B1f < 1, the circuit is
unconditionally stable.
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Action 1-19: Close the waveform window; go back to the Direct Plot Form window. In the
Function field, choose LSB (Load Stability Circles). In Plot Type, choose ZSmith. Specify Frequency Range from 2G to 3G with the step set to 0.2G.
Click Plot.
Action 1-20: Go back to the Direct Plot form, click New Subwindow.
Action 1-21: In the Function field, select SSB (Source Stability Circles). Click Plot.
The Load Stability Circles and Source Stability Circles are plotted:
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Action 1-22: Close the waveform window, and go back to the Direct Plot Form window. In
the Direct Plot Form window, set Plotting Mode to Append. In the Direct
Plot Form window, select NF (Noise Figure) in the Function field. In the
Modifier field, select dB10. Click Plot.
Action 1-23: In the Direct Plot Form window, click New Subwindow.
Action 1-24: In the function field, choose NC (Noise Circles). In the Plot type field,
choose Z-Smith. Sweep Noise Level at Frequency 2.4G Hz starting from 1.5
to 2.5 dB with steps set to 0.2 dB.
!
!
!
!
Note: You can perform small signal noise simulation using either the SP
or the Noise analyses. The Noise analysis provides only the noise figure,
NF. The SP analysis provides:
NFmin , minimum noise figure
RS , noise resistance
Gmin , optimum noise reflection coefficient
Yopt , optimum source admittance which is related to Gmin as shown in the
equation.
G min =
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YS + Yopt
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Action 1-25: Click Plot.
You get the following plot:
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In the above figure, the noise circle, NC, draws the NF on the Smith chart
of the source reflection coefficient, ΓS . The result in the NC plot where
ΓS = 0 and NF = 1.9 dB matches the result in the NF plot. The center of
the NC corresponds to ΓS (that is, Gmin ) which generates NFmin . The
optimum location for the center of the noise circle is at the center of the
Smith chart. However it is hard to center both the available gain circle,
GAC, and the noise circle, NC, in the Smith chart.
When you design an LNA, plot NC, GAC, and the source stability circle,
SSB, together in the same plot. Use this plot to trade off the gain, noise,
and stability for the input matching network design.
Action 1-26: Close the waveform window and go back to the Direct Plot Form window. In
the Direct Plot Form window, set Plotting Mode to Append. In the function
field, choose VSWR (Voltage standing-wave ratio). Click VSWR1, then
VSWR2.
You get the following waveforms:
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Action 1-27: Close the waveform window and go back to the Direct Plot Form. In the
function field, choose SP. In the Plot Type field, choose Rectangular. In the
Modifier field, select dB20. Click S11, S12, S21, then S22.
After these actions, the form looks like this:
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You get the following waveforms:
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Action 1-28: Close the waveform window and click Cancel in the Direct Plot Form.
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Lab 2: Noise Simulation ( hb and hbnoise )
Use the hb and hbnoise analyses to calculate the noise. (Use the Noise and SP analyses for
small-signal and linear noise analyses, where the circuits are linearized around the DC
operating point.) As the input power level increases, the circuit becomes nonlinear,
harmonics are generated, and the noise spectrum is folded. Therefore, you should use the hb
and hbnoise analyses. When the input power level remains low, the NF calculated from the
hbnoise, hbsp, Noise, and SP analyses should all match. For most cases, LNAs work with
very low input power level, so SP or noise analysis is enough.
HB analysis has several new features starting from mmsim 12.1. There is a new selection
called Run transient, which has 3 choices.
Decide automatically forces selection of Detect Steady State. This will monitor the tstab
waveform, and when steady-state is detected, it will switch to harmonic balance without
finishing the tstab interval. Run automatically sets a short tstab, and then begins running.
If steady-state is not detected in the first interval, it adds more cycles of the input and
keeps running up to a maximum of 250 periods for a driven circuit and 500 periods for an
oscillator. Decide automatically also sets the number of harmonics to auto. Auto will
determine the number of harmonics needed based on the tstab waveform.
Run transient=Yes is equivalent to setting tstab in mmsim 11.1. In this mode, Detect
Steady State is selected by default, and can be turned off manually. In this mode, tstab
cannot be extended if steady-state is not detected.
Run Transient=No is equivalent to setting tstab to zero in mmsim 11.1. The HB
analysis starts from the DC operating point.
Starting from mmsim 14.1, shooting PSS also provides the same feature and has the same
use model. Also, in mmsim14.1, dynamic parameter in PSS/HB tstab is available which
lets you to accelerate the tstab interval by using loose tolerance at the initial phase and
tight tolerance gradually.
Action 2-1:
Open the schematic view of the Diff_LNA_test in the library RFworkshop
Action 2-2:
From the Diff_LNA_test schematic, start the Virtuoso Analog Design
Environment with the Launch — ADE L command.
Action 2-3:
(Optional) Choose Session — Load State, select Cellview in Load State
Option and load state “Lab2_hbnoise” and skip to Action 2-13.
Action 2-4:
In the Virtuoso Analog Design Environment window, choose Analyses —
Choose…
Action 2-5:
In the Choosing Analyses window, select hb in the Analysis
field of the window.
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Action 2-6:
In the hb analyses window, make sure the Enabled button is on.
The Choosing Analyses — hb window looks like:
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Note that Decide automatically is set by default, and that the Number of
Harmonics is set to auto. Stop Time (tstab) is also set to auto when
Decide Automatically is set. The only thing you need to do here is to add
fundamental frequency.
Also note that Dynamic Parameter is enabled, at the beginning, reltol is
set as 1e-2 and after 1ns, it is tightened as 1e-3.
Action 2-7:
Now you set up the hb analysis. Click hbnoise in the Choosing Analyses
form. You must specify the noise source. You specify the reference sideband
as 0 for an LNA because an LNA has no frequency conversion form input to
output. The form looks like this:
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Action 2-8:
Make sure Enabled is selected, and click OK in the Choosing Analyses form.
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Action 2-9:
In the Virtuoso Analog Design Environment window, double click prf in the
field of Design Variables. Change the input power to -20.
Action 2-10: Click Change. Click OK to close the Editing Design Variables window.
The Virtuoso Analog Environment looks like this:
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Action 2-11: In the Virtuoso Analog Design Environment window, choose Simulation —
Netlist and Run or click the Netlist and Run icon to start the simulation.
Note in the log file window, the number of harmonics is selected
automatically.
Also note that in the log file, reltol is changed according to our setting:
Action 2-12: In the Virtuoso Analog Design Environment window, choose Results —
Direct Plot — Main Form.
Action 2-13: In the Direct Plot Form, select hbnoise, and configure the form as follows:
Action 2-14: Click Plot.
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The waveform window displays the noise figure. Comparing the above figure
with that from the SP analysis, you will notice that the noise figure plot
matches very closely. The noise figure from Pnoise is slightly larger than the
noise figure from SP because at Pin = -20 dBm, the LNA demonstrates very
weak nonlinearity and noise as other high harmonics are convoluted.
Action 2-15: Close the waveform window. Click Cancel on the Direct Plot Form. Close the
Virtuoso Analog Design Environment window.
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Lab 3: Gain Compression and THD (Xdb and Swept hb)
hb analysis calculates the operating power gain, which is the ratio of power delivered to the
load divided by the power available from the source. This gain definition is the same as that
for G P , so the gain from hb should match G P when the input power level is low and
nonlinearity is weak.
Starting from MMSIM13.1, a dedicated Xdb compression option is integrated in the HB
analysis form. It calculates compression points and compression curves directly, without the
need for post processing or manual setup of power sweeps. It supports voltage and powerbased compression point calculation. It is extremely useful when a large number of
compression simulations are needed such as in the corner sweeps or MC analysis.
Action 3-1:
Open the schematic view of the Diff_LNA_test in the library RFworkshop.
Action 3-2:
From the Diff_LNA_test schematic, start the Virtuoso Analog Design
Environment by choosing Launch — ADE L.
Action 3-3:
(Optional) Choose Session — Load State, select Cellview in Load State
Option and load state “Lab3_Xdb”, and skip to Action 3-7.
Action 3-4:
In the Virtuoso Analog Design Environment window, choose Analyses —
Choose…
Action 3-5:
In the Choosing Analyses window, select hb in the Analysis field of the
window. Set fundamental frequency as 2.4GHz. Click the Compression
check box. Set the compression section as follows:
Action 3-6:
Make sure Enabled is selected. Click OK on the Choosing Analyses form.
The Virtuoso Analog Design Environment window looks like this:
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Action 3-7:
In the Virtuoso Analog Design Environment window, choose Simulation —
Netlist and Run or click the Netlist and Run icon to start the simulation.
Action 3-8:
In the Virtuoso Analog Design Environment window, choose Results —
Direct Plot — Main Form.
Action 3-9:
In the Direct Plot Form, select xdb:
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Action 3-10: Click Plot button. The power curve appears in the waveform window.
The compression point is a marker on the curve.
Action 3-11: Close the waveform window.
Action 3-12: Another way to calculate compression point is to use the swept hb
analysis. In the hb analysis form, click sweep check box, set the sweep as
follows:
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Action 3-13: Save the hb analysis setting and run the simulation.
Action 3-14: After the simulation finishes, choose Results — Direct Plot — Main Form.
Select hb and set the form as follows:
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Action 3-15: Select the port load on the schematic, the 1dB plot appears:
The result matches with that from Xdb analysis well.
After the hb analysis, you can observe the harmonic distortion of the LNA by plotting the
spectrum of any node voltage. Harmonic distortion is characterized as the ratio of the
power of the fundamental signal divided by the sum of the power at the harmonics. In the
following steps, you plot the spectrum of a load when the input power level is -20 dBm.
Action 3-16: In the Direct Plot Form, select hb, and configure the form as follows:
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Action 3-17: Select output net RFout on the schematic.
The plot shows that the DC and all the even modes at the output are
suppressed because the LNA is a differential LNA.
If you write the nonlinear response of one side amplification as
y ( x) = α 0 + α 1 x + α 2 x 2 + α 3 x 3 ......
the output is
y = y ( x 2) − y (− x 2) = α 1 x + α 3 x 3 / 4
For the differential LNA, the common mode disturbance is rejected.
Action 3-18: After viewing the waveforms, close the waveform window.
Action 3-19: In the Direct Plot Form, select the hb analysis and the THD function.
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Action 3-20: Select output net RFout on the schematic.
The THD plot appears in the waveform window.
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Action 3-21: Close the waveform window and click Cancel on the Direct Plot Form.
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IP3 measurements
IP3 is an important RF specification. The IP3 measurement is defined as the cross point
of the power for the 1st order tones, ω1 and ω2 , and the power for the 3rd order tones,
2ω1 − ω2 and 2ω2 − ω1 , on the load.
As shown in the above figure, when you assume the input signal, x, is
x = A1 cos ω1 t + A2 cos ω 2 t
and the nonlinear response, y, is
y = α1 x + α 2 x 2 + α 3 x 3
then, the linear and third order components at the output are
2
2
3α A A
3α 3 A1 A2
cos( 2ω 2 − ω1 )t ,
α1 A1 cosω1t, α1 A2 cos ω2t, 3 1 2 cos(2ω1 − ω 2 )t ,
4
4
When A1 = A2 , the two first-order components have the same amplitude and the two
third-order components also have the same amplitude.
As the first-order components grow linearly and the third-order components grow
cubically, they eventually intersect as the input power level A increases. The third-order
intercept point is the point where the two output power curves intersect.
SpectreRF offers several ways to simulate IP3. The following 3 labs, for example, illustrate
different methods that can be used to calculate IP3 for LNAs.
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Lab 4: IP3 Measurement---hb/hbac analysis
The first method treats one tone, for example ω1 , as a large signal and performs one hb
analysis on the signal. The method treats the other tone, for example ω2 , as a small signal and
performs one hbac analysis on the signal based on the linear time-varying systems obtained
after the hb analysis. The IP3 point is the intersect point between the power for the signal ω2
and the power for the signal 2ω1 − ω2 . Because the magnitude of this component
is 0.75α 3 A12 A2 , it has a linear relationship with the power level of the tone ω2 . Thus the ω2
component can be treated as a small signal. The power level of both tones must be set to the
same value.
Action 4-1:
If it is not already open, open the schematic view of the Diff_LNA_test in the
library RFworkshop.
Action 4-2:
From the Diff_LNA_test schematic, choose Launch — ADE L to start the
Virtuoso Analog Design Environment.
Action 4-3:
(Optional) Choose Session — Load State, select Cellview in Load State
Option and load state “Lab4_IP3_hbac” and skip to Action 4-12.
Action 4-4:
In the Virtuoso Analog Design Environment window, choose Analyses —
Choose.
Action 4-5:
In the Choosing Analyses window, select hb in the Analysis
field of the window and set up the form as follows:
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Action 4-6:
Select Sweep and set the sweep values as follows:
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Action 4-7:
In the Choosing Analyses window, select hbac in the Analysis
field of the window.
Action 4-8:
Set up the form as shown here:
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Action 4-9:
Click OK in the Choosing Analyses form.
The Virtuoso Analog Design Environment window looks like this:
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Action 4-10: In the Virtuoso Analog Design Environment window, choose Simulation —
Netlist and Run or click the Netlist and Run icon to start the simulation.
Action 4-11: After the simulation ends, in the Virtuoso Analog Design Environment
window, choose Results — Direct Plot — Main Form.
Action 4-12: Choose hbac and set up the form as follows:
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Note: As defined, the IP3 point is the intersection point between the power for
the signal ω2 and the power for the signal 2ω1 − ω2 . So here you choose ω2
as the 1st order harmonic, and 2ω1 − ω2 the 3rd order harmonic.
Action 4-13: Select port load in the Diff_LNA_test schematic.
The IP3 plot appears in the waveform window. On the left panel, right click
on the name of the waveform, change its type to Continuous line.
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The IP3 plot shows as follows:
Action 4-14: Click Cancel in the Direct Plot Form and close the waveform window.
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Lab 5: IP3 Measurement---hb Analysis with Two Tones
This second method treats both tones as large signals and uses an hb analysis with two tones.
The first and second methods are equivalent because of the linear dependence of the output
component's magnitude, 2ω1 − ω2 , on the input component's magnitude, ω2 . Cadence
recommends using the hb and hbac analyses for IP3 simulation because that method is more
efficient than the hb analysis with two tones, and because the calculated IP3 is theoretically
expected to be the same and is actually very close numerically.
Action 5-1:
If it is not already open, open the schematic view of the Diff_LNA_test in the
library RFworkshop
Action 5-2:
From the Diff_LNA_test schematic, choose Launch — ADE L to start the
Virtuoso Analog Design Environment.
Action 5-3:
(Optional) Choose Session — Load State, select Cellview in Load State
Option and load state “Lab5_IP3_hb2” and skip to Action 5-7.
Action 5-4:
In the Virtuoso Analog Design Environment window, choose Analyses —
Choose.
Action 5-5:
In the Choosing Analyses window, select hb in the Analysis field of the
window and set the tone sectoin as follows:
Action 5-6:
Make sure Enabled is selected. In the Choosing Analyses window, click OK.
The Virtuoso Analog Design Environment window looks like this:
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Action 5-7:
In the Virtuoso Analog Design Environment window, choose Simulation —
Netlist and Run or click the Netlist and Run icon to start the simulation.
Action 5-8:
In the Virtuoso Analog Design Environment window, choose Results —
Direct Plot — Main Form.
Action 5-9:
In the Direct Plot Form, select hb_mt, and configure the form as follows:
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Action 5-10: Select output port load on the schematic. The IP3 plot appears in the
waveform window.
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Action 5-11: Close the waveform window. Click Cancel on the Direct Plot Form.
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Lab 6: IP3 Measurement---Rapid IP3 using AC analysis
This lab shows you how to calculate the IP3 of LNAs using perturbation technology. For for
descriptions about this method, refer to Rapid IPN application note.
Action 6-1:
Open the schematic view of the Diff_LNA_test in the library Rfworkshop.
Action 6-2:
Select the PORTrf source. Choose Edit — Properties — Objects and set the
port properties as described below:
Parameter
Value
Resistance
50 ohm
Port Number
1
DC voltage
( blank )
Source type
dc
Note: The RF input source should be set to DC. The perturbation method
is a type of nonlinear small signal analysis that treats the RF signal as a
small signal. If the RF input source is set to sinusoidal (or some other type
of large signal), the hb and later small signal results are affected.
Action 6-3:
Click OK in the Edit Object Properties window to close it.
Action 6-4:
Choose Check- Current Cellview.
Action 6-5:
From the Diff_LNA_test schematic, choose Launch — ADE L to start the
Virtuoso Analog Design Environment.
Action 6-6:
(Optional) Choose Session — Load State, select Cellview in Load State
Option and load state “Lab6_IP3_rapid” and skip to Action 6-10.
Action 6-7:
In the Virtuoso Analog Design Environment window, choose Analyses —
Choose…
Action 6-8:
In the Choosing Analyses window, select ac in the Analysis and set the rapid
IP3 analysis as follows:
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Action 6-9:
Make sure Enabled is selected. In the Choosing Analyses window, click OK.
The Virtuoso Analog Design Environment window looks like this:
Action 6-10: In the Virtuoso Analog Design Environment window, choose Simulation —
Netlist and Run or click the Netlist and Run icon to start the simulation.
Action 6-11: In the Virtuoso Analog Design Environment window, choose Results —
Direct Plot — Main Form.
Action 6-12: In the Direct Plot Form, select the ac analysis. Choose Rapid IP3 in the
Function field.
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Action 6-13: Click Plot to get the IP3 calculation results:
Action 6-14: After viewing the waveforms, click Cancel in the Direct Plot Form.
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Conclusion
This application note discusses:
•
•
•
•
LNA testbench setup
LNA design parameters
How to use SpectreRF to simulate an LNA and extract design parameters
Useful SpectreRF analysis tools for LNA design, such as SP, hb, hbnoise, hbac, 2tone hb analyses.
The results from the analyses are interpreted.
References
[1]
The Designer's Guide to Spice & Spectre, Kenneth S. Kundert, Kluwer Academic
Publishers, 1995.
[2]
Microwave Transistor Amplifiers, Guillermo Gonzalez, Prentice Hall, 1984.
[3]
RF Microelectronics, Behzad Razavi. Prentice Hall, NJ, 1998.
[4]
The Design of CMOS Radio Frequency Integrated Circuits, Thomas H. Lee,
Cambridge University Press, 1998.
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