AD8346

AD8346
0.8 GHz to 2.5 GHz
Quadrature Modulator
AD8346
High accuracy
1 degree rms quadrature error @ 1.9 GHz
0.2 dB I/Q amplitude balance @ 1.9 GHz
Broad frequency range: 0.8 GHz to 2.5 GHz
Sideband suppression: −46 dBc @ 0.8 GHz
Sideband suppression: −36 dBc @ 1.9 GHz
Modulation bandwidth: dc to 70 MHz
0 dBm output compression level @ 0.8 GHz
Noise floor: −147 dBm/Hz
Single 2.7 V to 5.5 V supply
Quiescent operating current: 45 mA
Standby current: 1 μA
16-lead TSSOP
FUNCTIONAL BLOCK DIAGRAM
IBBP
1
16
QBBP
IBBN
2
15
QBBN
COM1
3
14
COM4
COM1
4
13
COM4
LOIN
5
12
VPS2
LOIP
6
VPS1
7
PHASE
SPLITTER
11 VOUT
10 COM3
AD8346
ENBL
8
BIAS
9
COM2
05335-001
FEATURES
Figure 1.
APPLICATIONS
Digital and spread spectrum communication systems
Cellular/PCS/ISM transceivers
Wireless LAN/wireless local loop
QPSK/GMSK/QAM modulators
Single-sideband (SSB) modulators
Frequency synthesizers
Image reject mixer
GENERAL DESCRIPTION
The AD8346 is a silicon RFIC I/Q modulator for use from
0.8 GHz to 2.5 GHz. Its excellent phase accuracy and amplitude
balance allow high performance direct modulation to RF.
The differential LO input is applied to a polyphase network
phase splitter that provides accurate phase quadrature from
0.8 GHz to 2.5 GHz. Buffer amplifiers are inserted between
two sections of the phase splitter to improve the signal-tonoise ratio. The I and Q outputs of the phase splitter drive the
LO inputs of two Gilbert-cell mixers. Two differential V-to-I
converters connected to the baseband inputs provide the
baseband modulation signals for the mixers. The outputs of
the two mixers are summed together at an amplifier which is
designed to drive a 50 Ω load.
This quadrature modulator can be used as the transmit modulator in digital systems such as PCS, DCS, GSM, CDMA, and
ISM transceivers. The baseband quadrature inputs are directly
modulated by the LO signal to produce various QPSK and
QAM formats at the RF output.
Additionally, this quadrature modulator can be used with direct
digital synthesizers in hybrid phase-locked loops to generate
signals over a wide frequency range with millihertz resolution.
The AD8346 comes in a 16-lead TSSOP package, measuring
6.5 mm × 5.1 mm × 1.1 mm. It is specified to operate over a
−40°C to +85°C temperature range and a 2.7 V to 5.5 V supply
voltage range. The device is fabricated on Analog Devices’ high
performance 25 GHz bipolar silicon process.
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective companies.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
www.analog.com
Tel: 781.329.4700
Fax: 781.461.3113
© 2005 Analog Devices, Inc. All rights reserved.
AD8346
TABLE OF CONTENTS
Specifications..................................................................................... 3
Bias ............................................................................................... 10
Absolute Maximum Ratings............................................................ 4
Basic Connections...................................................................... 11
ESD Caution.................................................................................. 4
LO Drive ...................................................................................... 11
Pin Configuration and Function Descriptions............................. 5
RF Output.................................................................................... 11
Equivalent Circuits ........................................................................... 6
Interface to AD9761 TXDAC® .................................................. 12
Typical Performance Characteristics ............................................. 7
AC-Coupled Interface ............................................................... 13
Circuit Description......................................................................... 10
Evaluation Board ............................................................................ 14
Overview...................................................................................... 10
Characterization Setups................................................................. 16
LO Interface................................................................................. 10
SSB Setup..................................................................................... 16
V-to-I Converter......................................................................... 10
CDMA Setup............................................................................... 17
Mixers .......................................................................................... 10
Outline Dimensions ....................................................................... 18
Differential-to-Single-Ended Converter ................................. 10
Ordering Guide .......................................................................... 18
REVISION HISTORY
6/05—Rev. 0 to Rev. A
Updated Format..................................................................Universal
Changes to Figures 30, 31, 32........................................................ 14
Update Outline Dimensions ......................................................... 18
Changes to Ordering Guide .......................................................... 18
3/99—Revision 0: Initial Version
Rev. A | Page 2 of 20
AD8346
SPECIFICATIONS
VS = 5 V; TA = 25°C; LO frequency = 1900 MHz; LO level = –10 dBm; BB frequency = 100 kHz; BB inputs are dc-biased to 1.2 V; BB input
level = 1.0 V p-p each pin for 2.0 V p-p differential drive; LO source and RF output load impedances are 50 Ω, dBm units are referenced
to 50 Ω unless otherwise noted.
Table 1.
Parameters
RF OUTPUT
Operating Frequency
Quadrature Phase Error
I/Q Amplitude Balance
Output Power
Output VSWR
Output P1 dB
Carrier Feedthrough
Sideband Suppression
IM3 Suppression
Equivalent Output IP3
Output Noise Floor
RESPONSE TO CDMA IS95 BASEBAND SIGNALS
ACPR (Adjacent Channel Power Ratio)
EVM (Error Vector Magnitude)
Rho (Waveform Quality Factor)
MODULATION INPUT
Input Resistance
Modulation Bandwidth
LO INPUT
LO Drive Level
Input VSWR
ENABLE
ENBL HI Threshold
ENBL LO Threshold
ENBL Turn-On Time
ENBL Turn-Off Time
Conditions
Min
Typ
0.8
See Figure 35 for setup
See Figure 35 for setup
I and Q channels in quadrature
Max
Unit
2.5
GHz
Degree rms
dB
dBm
20 MHz offset from LO
1
0.2
−10
1.25:1
−3
−42
−36
−60
20
−147
See Figure 35 for setup
See Figure 35 for setup
See Figure 35 for setup
−72
2.5
0.9974
dBc
%
−3 dB
12
70
kΩ
MHz
−13
−12
−6
−35
−25
dBm
dBm
dBc
dBc
dBm
dBm/Hz
−6
dBm
2.0
2.5
V
V
μs
12
μs
1.9:1
0.5
Settle to within 0.5 dB of final SSB
output power
Time for supply current to drop below
2 mA
POWER SUPPLIES
Voltage
Current Active (ENBL HI)
Current Standby (ENBL LO)
2.7
35
Rev. A | Page 3 of 20
45
1
5.5
55
20
V
mA
μA
AD8346
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage VPS1, VPS2
Input Power LOIP, LOIN (relative to 50 Ω)
Min Input Voltage IBBP, IBBN, QBBP, QBBN
Max Input Voltage IBBP, IBBN, QBBP, QBBN
Internal Power Dissipation
θJA
Operating Temperature Range
Storage Temperature Range
Lead Temperature (Soldering 60 sec)
Min Rating
5.5 V
10 dBm
0V
2.5 V
500 mW
125°C/W
−40°C to +85°C
−65°C to +150°C
300°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other condition s above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. A | Page 4 of 20
AD8346
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
IBBP 1
16 QBBP
IBBN
2
15 QBBN
COM1 3
14 COM4
AD8346 13 COM4
TOP VIEW
LOIN 5 (Not to Scale) 12 VPS2
LOIP 6
11 VOUT
VPS1 7
10 COM3
ENBL 8
9
COM2
05335-002
COM1
4
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
Mnemonic
IBBP
2
IBBN
3
4
5
COM1
COM1
LOIN
6
LOIP
7
VPS1
8
9
10
11
12
ENBL
COM2
COM3
VOUT
VPS2
13
14
15
COM4
COM4
QBBN
16
QBBP
Description
I Channel Baseband Positive Input Pin. Input should be dc-biased to approximately 1.2 V.
Nominal characterized ac swing is 1 V p-p (0.7 V to 1.7 V). This makes the differential input
2 V p-p when IBBN is 180 degrees out of phase from IBBP.
I Channel Baseband Negative Input Pin. Input should be dc-biased to approximately 1.2 V.
Nominal characterized ac swing is 1 V p-p (0.7 V to 1.7 V). This makes the differential input
2 V p-p when IBBN is 180 degrees out of phase from IBBP.
Ground Pin for the LO phase splitter and LO buffers.
Ground Pin for the LO phase splitter and LO buffers.
LO Negative Input Pin. Internal dc bias (approximately VPS1 to 800 mV) is supplied. This
pin must be ac coupled.
LO Positive Input Pin. Internal dc bias (approximately VPS1 to 800 mV) is supplied. This pin
must be ac-coupled.
Power Supply Pin for the bias cell and LO buffers. This pin should be decoupled using
local 100 pF and 0.01 μF capacitors.
Enable Pin. A high level enables the device; a low level puts the device in sleep mode.
Ground Pin for the input stage of output amplifier.
Ground Pin for the output stage of output amplifier.
50 Ω DC-Coupled RF Output. User must provide ac coupling on this pin.
Power Supply Pin for baseband input voltage to current converters and mixer core. This
pin should be decoupled using local 100 pF and 0.01 μF capacitors.
Ground Pin for baseband input voltage to current converters and mixer core.
Ground Pin for baseband input voltage to current converters and mixer core.
Q Channel Baseband Negative Input. Input should be dc biased to approximately 1.2 V.
Nominal characterized ac swing is 1 V p-p. This makes the differential input 2 V p-p when
QBBN is 180° out of phase from QBBP.
Q Channel Baseband Positive Input. Input should be dc-biased to approximately 1.2 V.
Nominal characterized ac swing is 1 V p-p. This makes the differential input 2 V p-p when
QBBN is 180° out of phase from QBBP.
Rev. A | Page 5 of 20
Equivalent
Circuit
Circuit A
Circuit A
Circuit B
Circuit B
Circuit C
Circuit D
Circuit A
Circuit A
AD8346
EQUIVALENT CIRCUITS
VPS1
VPS2
TO MIXER
CORE
75kΩ
TO BIAS FOR
STARTUP/
SHUTDOWN
75kΩ
9kΩ
INPUT
30kΩ
ENBL
05335-003
3kΩ
ACTIVE LOADS
40kΩ
780Ω
Figure 5. Circuit C
Figure 3. Circuit A
VPS2
VPS1
LOIN
PHASE
SPLITTER
CONTINUES
43Ω
05335-004
43Ω
Figure 4. Circuit B
Figure 6. Circuit D
Rev. A | Page 6 of 20
05335-006
VOUT
LOIP
05335-005
BUFFER
AD8346
TYPICAL PERFORMANCE CHARACTERISTICS
2
–6
T = 25°C
1
OUTPUT POWER VARIATION (dB)
VP = 5.5V
–8
–9
–10
VP = 3V
–11
VP = 2.7V
–12
–13
–15
800
05335-007
–14
1000
1200
1400 1600 1800 2000
LO FREQUENCY (MHz)
2200
–4
–5
–6
–7
1
2
LO = 800MHz, –6dBm
VP = 5V
T = +85°C
0
VP = 5V
T = –40°C
LO = 800MHz, –10dBm
–8
LO = 1900MHz, –6dBm
–9
–10
LO = 1900MHz, –10dBm
–11
100
10
Figure 10. I and Q Input Bandwidth. FLO =1900 MHz, I or Q inputs
driven with differential amplitude of 2.00 V p-p.
SSB OUTPUT P1dB (dBm)
–12
–2
–4
VP = 2.7V
T = –40°C
–6
–8
VP = 2.7V
T = +85°C
–10
–13
–40 –30 –20 –10
0
10 20 30 40
TEMPERATURE (°C)
50
60
70
80
05335-008
–12
–14
800
Figure 8. SSB POUT vs. Temperature. I and Q inputs driven in quadrature
with differential amplitude of 2.00 V p-p at FBB = 100 kHz.
05335-011
SSB OUTPUT POWER (dBm)
–3
BASEBAND FREQUENCY (MHz)
–7
1000
1200
1400 1600 1800 2000
LO FREQUENCY (MHz)
2200
2400
Figure 11. SSB Output 1 dB Compression Point (OP 1 dB) vs. FLO.
I and Q inputs driven in quadrature at FBB = 100 kHz.
30
–35
T = +85°C
T = –40°C
–37
25
VP = 5.5V
–39
PERCENTAGE
20
–41
VP = 5V
–43
VP = 3V
–45
–47
15
10
VP = 2.7V
5
–49
–51
–40 –30 –20 –10
0
10 20 30 40
TEMPERATURE (°C)
50
60
70
80
05335-009
CARRIER FEEDTHROUGH (dBm)
–2
–8
0.1
2400
Figure 7. Single Sideband (SSB) Output Power (POUT) vs. LO Frequency (FLO).
I and Q inputs driven in quadrature at baseband frequency
(FBB) = 100 kHz with differential amplitude of 2.00 V p-p.
–6
0
–1
0
–90
Figure 9. Carrier Feedthrough vs. Temperature.
FLO = 1900 MHz, LO input level = –10 dBm.
05335-012
SSB POWER (dBm)
VP = 5V
05335-010
–7
–86
–82
–78 –74 –70 –66 –62 –58 –54
CARRIER FEEDTHROUGH (dBm/
AFTER NULLING TO <–60dBm @ 25°C)
–50
–46
Figure 12. Histogram Showing Carrier Feedthrough Distributions
at the Temperature Extremes after Nulling at Ambient
at FLO = 1900 MHz, LO Input Level = –10 dBm.
Rev. A | Page 7 of 20
AD8346
–7
–30
–8
–32
VP = 5.5V
SB SUPPRESSION (dBc)
VP = 3V
VP = 5V
–11
–12
VP = 2.7V
–13
–15
–40 –30 –20 –10
0
10 20 30 40
TEMPERATURE (°C)
50
60
70
2
4
6
8
10
12
14
16
BASEBAND FREQUENCY (MHz)
18
20
Figure 16. Sideband Suppression vs. FBB. FLO = 1900 MHz, I and Q inputs
driven in quadrature with differential amplitude of 2.00 V p-p.
–40
VP = 3V
–42
–44
VP = 5V
–46
VP = 2.7V
–48
–50
1200
1400 1600 1800 2000
LO FREQUENCY (MHz)
2200
VP = 5V
VP = 2.7V
–50
VP = 3V
–55
–60
–65
05335-014
–54
–45
VP = 5.5V
–70
–40 –30 –20 –10
2400
0
05335-017
–40
INPUT THIRD HARMONIC
DISTORTION (dBc)
CARRIER FEEDTHROUGH (dBm)
VP = 2.7V
–40
0
VP = 5.5V
–52
10 20 30 40
TEMPERATURE (°C)
50
60
70
80
Figure 17. Third Harmonic Distortion vs. Temperature.
FLO =1900 MHz, I and Q inputs driven in quadrature with
differential amplitude of 2.00 V p-p at FBB = 100 kHz.
Figure 14. Carrier Feedthrough vs. FLO.
LO input level = –10 dBm.
0
–32
T = 25°C
VP = 5.5V
–2
–34
–4
–36
VP = 5V
RETURN LOSS (dB)
SIDEBAND SUPPRESSION (dBc)
VP = 5V
–35
–38
1000
–38
–44
T = 25°C
800
VP = 5.5V
–36
80
Figure 13. SSB POUT vs. Temperature. FLO = 1900 MHz, I and Q inputs driven in
quadrature with differential amplitude of 2.00 V p-p at FBB = 100 kHz.
–36
VP = 3V
–42
05335-013
–14
–34
05335-016
–10
–38
–40
VP = 3V
–42
–6
T = +25°C
T = –40°C
–8
–10
T = +85°C
–12
–14
–44
–16
VP = 2.7V
05335-015
–46
–48
900
1100
1300 1500 1700 1900
LO FREQUENCY (MHz)
2100
2300
05335-018
SSB OUTPUT POWER (dBm)
–9
–18
–20
800
2500
Figure 15. Sideband Suppression vs. FLO. VPOS = 2.7 V, I and Q inputs driven in
quadrature with differential amplitude of 2.00 V p-p at FBB = 100 kHz.
Rev. A | Page 8 of 20
1000
1200
1400 1600 1800 2000
FREQUENCY (MHz)
2200
Figure 18. Return Loss of LOIN Input vs. FLO.
VPOS = 5.0 V, LOIP pin ac-coupled to ground.
2400
AD8346
–30
–40
VP = 2.7V
–32
–45
VP = 5.5V
VP = 2.7V
VP = 5V
–38
–40
VP = 3V
–50
VP = 5.5V
–55
VP = 5V
–60
–44
–40 –30 –20 –10
05335-019
–42
0
10 20 30 40
TEMPERATURE (°C)
50
60
70
–65
80
0
–35
SSB POUT
2
4
6
8
10
12
14
16
BASEBAND FREQUENCY (MHz)
18
20
Figure 22. Third Harmonic Distortion vs. FBB.
FLO =1900 MHz, I and Q inputs driven in quadrature
with differential amplitude of 2.00 V p-p.
Figure 19. Sideband Suppression vs. Temperature.
FLO = 1900 MHz, I and Q inputs driven in quadrature
with differential amplitude of 2.00 V p-p at FBB = 100 kHz.
–30
05335-022
SB SUPPRESSION (dBc)
–36
INPUT THIRD HARMONIC
DISTORTION (dB)c
VP = 3V
–34
–6
52
–8
50
–10
48
–60
–16
–65
–18
3RD HARMONIC
–70
–22
1.0
1.5
2.0
2.5
BASEBAND DIFFERENTIAL INPUT
VOLTAGE (V p-p)
3.0
42
VP = 3V
40
38
36
–40
0
0
–5
–5
–10
20
40
TEMPERATURE (°C)
60
80
RETURN LOSS (dB)
–10
T = –40°C
–15
–20
–25
T = +25°C
–20
–25
T = +25°C
–30
–35
–35
T = +85°C
1000
1200
1400 1600 1800 2000
FREQUENCY (MHz)
2200
–40
800
2400
T = –40°C
–15
–30
05335-021
RETURN LOSS (dB)
0
Figure 23. Power Supply Current vs. Temperature
Figure 20. Third Harmonic Distortion and SSB Output
Power vs. Baseband Differential Input Voltage Level.
FLO = 1900 MHz, I and Q inputs driven in quadrature at FBB = 100 kHz.
–40
800
–20
T = +85°C
05335-024
–80
0.5
VP = 5V
44
VP = 2.7V
–20
–75
VP = 5.5V
46
05335-023
–14
–55
SUPPLY CURRENT (mA)
–12
–50
SSB OUTPUT POWER (dBm)
–45
05335-020
INPUT THIRD HARMONIC
DISTORTION (dBc)
–40
1000
1200
1400
1600
1800
2000
2200
FREQUENCY (MHz)
Figure 24. Return Loss of VOUT Output vs. FLO.
VPOS = 5.0 V.
Figure 21. Return Loss of VOUT Output vs. FLO.
VPOS = 2.7 V.
Rev. A | Page 9 of 20
2400
AD8346
CIRCUIT DESCRIPTION
OVERVIEW
V-TO-I CONVERTER
The AD8346 can be divided into the following sections: local
oscillator (LO) interface, mixer, voltage-to-current (V-to-I)
converter, differential-to-single-ended (D-to-S) converter, and
bias. A detailed block diagram of the part is shown in Figure 25.
Each baseband input pin is connected to an op amp driving an
emitter follower. Feedback at the emitter maintains a current
proportional to the input voltage through the transistor. This
current is fed to the two mixers in differential form.
The LO interface generates two LO signals, with 90° of phase
difference between them, to drive two mixers in quadrature.
Baseband voltage signals are converted into current form in
the V-to-I converters, feeding into two mixers. The output of
the mixers are combined to feed the D-to-S converter which
provides the 50 Ω output interface. Bias currents to each
section are controlled by the Enable (ENBL) signal. Detailed
descriptions of each section follows.
MIXERS
There are two double-balanced mixers, one for the in-phase
channel (I-channel) and one for the quadrature channel
(Q channel). Each mixer uses the gilbert cell design with four
cross-connected transistors. The bases of the transistors are
driven by the LO signal of the corresponding channel. The
output currents from the two mixers are summed together in
two resistors in series with two coupled on-chip inductors. The
signal developed across the R-L loads is sent to the D-to-S stage.
LO INTERFACE
DIFFERENTIAL-TO-SINGLE-ENDED CONVERTER
The differential LO inputs allow the user to drive the LO differentially in order to achieve maximum performance. The LO can
be driven single-endedly but the LO feedthrough performance
is degraded, especially towards the higher end of the frequency
range. The LO interface consists of interleaved stages of
polyphase network phase splitters and buffer amplifiers. The
phase-splitter contains resistors and capacitors connected in a
circular manner to split the LO signal into I and Q paths in
precise quadrature with each other. The signal on each path
goes through a buffer amplifier to make up for the loss and high
frequency roll-off. The two signals then go through another
polyphase network to enhance the quadrature accuracy. The
broad operating frequency range of 0.8 GHz to 2.5 GHz is
achieved by staggering the RC time constants in each stage of
the phase-splitters. The outputs of the second phase-splitter are
fed into the driver amplifiers for the mixers’ LO inputs.
The differential-to-single-ended converter consists of two
emitter followers driving a totem-pole output stage. Output
impedance is established by the emitter resistors in the output
transistors. The output of this stage is connected to the output
(VOUT) pin.
BIAS
A band gap reference circuit based on the Δ-VBE principle
generates the proportional-to-absolute-temperature (PTAT)
currents used by the different sections as references. The band
gap voltage is also used to generate a temperature-stable current
in the V-to-I converters to produce a temperature-independent
slew rate. When the band gap reference is disabled by pulling
down the ENBL pin, all other sections are shut off accordingly.
IBBP
IBBN
V-TO-I
V-TO-I
AD8346
MIXER
LOIN
LOIP
PHASE
SPLITTER
2
PHASE
SPLITTER
1
D-TO-S
VOUT
MIXER
BIAS CELL
V-TO-I
QBBP
Figure 25. Detailed Block Diagram
Rev. A | Page 10 of 20
05335-025
ENBL
V-TO-I
QBBN
AD8346
BASIC CONNECTIONS
The basic connections for operating the AD8346 are shown in
Figure 27. A single power supply of between 2.7 V and 5.5 V is
applied to pins VPS1 and VPS2. A pair of ESD protection
diodes are connected internally between VPS1 and VPS2 so
these must be tied to the same potential. Both pins should be
individually decoupled using 100 pF and 0.01 μF capacitors,
located as close as possible to the device. For normal operation,
the enable pin, ENBL, must be pulled high. The turn-on
threshold for ENBL is 2 V. To put the device in its power-down
mode, ENBL must be pulled below 0.5 V. Pins COM1 to COM4
should all be tied to a low impedance ground plane.
have a bias level about 800 mV below supply. An LO drive
level of between −6 dBm and −12 dBm is required. For optimal
performance, a drive level of −10 dBm is recommended,
although a level of −6 dBm results in more stable temperature
performance (see Figure 8). Higher levels degrade linearity
while lower levels tend to increase the noise floor.
100pF
LO
LOIP
05335-026
AD8346
LOIN
100pF
The I and Q ports should be driven differentially. This is convenient as most modern high speed DACs have differential
outputs. For optimal performance, the drive signal should be a
2 V p-p (differential) signal with a bias level of 1.2 V, that is,
each input swings from 0.7 V to 1.7 V. The I and Q inputs have
input impedances of 12 kΩ. By dc coupling the DAC to the
AD8346 and applying small offset voltages, the LO feedthrough
can be reduced to well below its nominal value of −42 dBm
(see Figure 12).
The LO terminal can be driven single-ended, as shown in
Figure 26 at the expense of slightly higher LO feedthrough.
LOIN is ac coupled to ground using a capacitor and LOIP is
driven through a coupling capacitor from a (single-ended)
50 Ω source (this scheme could also be reversed with LOIP
being ac-coupled to ground).
LO DRIVE
RF OUTPUT
The return loss of the LO port is shown in Figure 18. No additional matching circuitry is required to drive this port from a
50 Ω source. For maximum LO suppression at the output, a
differential LO drive is recommended. In Figure 27, this is
achieved using a balun (M/A-COM Part Number ETC1-1-13).
The output of the balun is ac-coupled to the LO inputs which
The RF output is designed to drive a 50 Ω load, but must be accoupled, as shown in Figure 27. If the I and Q inputs are driven
in quadrature by 2 V p-p signals, the resulting output power is
about −10 dBm (see Figure 7 for variations in output power
over frequency).
2 IBBN
C6
100pF
5
1
T1
2
ETC1-1-13
4
+VS
QBBN 15
AD8346
IN
LO
QP
QBBP 16
C4
0.01μF
C7
100pF
3
C3
100pF
QN
3 COM1
COM4 14
4 COM1
COM4 13
5 LOIN
VPS2 12
6 LOIP
VOUT 11
7 VPS1
COM3 10
8 ENBL
COM2 9
Figure 27. Basic Connections
Rev. A | Page 11 of 20
C1
100pF
C2
0.01μF
+VS
VOUT
C5
100pF
05335-027
1 IBBP
IP
Figure 26. Single-Ended LO Drive
AD8346
INTERFACE TO AD9761 TXDAC®
Figure 28 shows a dc-coupled current output DAC interface.
The use of dual-integrated DACs, such as the AD9761 with
specified ±0.02 dB and ±0.004 dB gain and offset matching
characteristics, ensures minimum error contribution (over
temperature) from this portion of the signal chain. The use of a
precision thin-film resistor network sets the bias levels precisely
to prevent the introduction of offset errors, which increase LO
feedthrough. For instance, selecting resistor networks with a
0.1% ratio matching characteristics maintains 0.03 dB gain and
offset matching performance.
10 mA, giving a voltage swing of 0 V to 1 V (at the DAC
output). This results in a 0.5 V p-p swing at the I and Q inputs
of the AD8346 (resulting in a 1 V p-p differential swing).
Note that the ratio matching characteristics of the resistive
network, as opposed to its absolute accuracy, is critical in
preserving the gain and offset balance between the I and Q
signal path.
By applying small dc offsets to the I and Q signals from the
DAC, the LO suppression can be reduced from its nominal
value of −42 dBm to as low as −60 dBm while holding to
approximately −50 dBm over temperature (see Figure 12 for
a plot of LO feedthrough over temperature for an offset
compensated circuit).
Using resistive division, the dc bias level at the I and Q inputs
to the AD8346 is set to approximately 1.2 V. Each of the four
current outputs of the DAC delivers a full-scale current of
5V
+5V
DVDD
634Ω
DCOM
AVDD
500Ω
100Ω
IOUTA
LATCH
I
2×
I
DAC
0.1μF
VPS1
VPS2
IBBP
CFILTER
IOUTB
500Ω
100Ω
DAC
DATA
INPUTS
500Ω
500Ω
Σ
IBBN
VOUT
AD9761
500Ω
100Ω
QOUTA
LATCH
Q
2×
Q
DAC
CLOCK
100Ω
MUX
CONTROL
SLEEP
FS ADJ
RSET
2kΩ
QBBP
LOIN
500Ω
0.5V p-p EACH PIN
WITH VCM = 1.2V
QBBN
AD8346
REFIO
05335-028
WRITE
LOIP
PHASE
SPLITTER
CFILTER
QOUTB
SELECT
500Ω
500Ω
0.1μF
Figure 28. AD8346 Interface to AD9761 TxDAC
Rev. A | Page 12 of 20
AD8346
AC-COUPLED INTERFACE
An ac-coupled interface can also be implemented, as shown in
Figure 29. This is an advantage because there is almost no
voltage loss due to the biasing network, allowing the AD8346
inputs to be driven by the full 2 V p-p differential signal from
the AD9761 (each of the DAC’s 4 outputs delivering 1 V p-p).
The network shown has a high-pass corner frequency of
approximately 14.3 kHz (note that the 12 kΩ input impedance
of the AD8346 has been factored into this calculation).
Increasing the resistors in the network or increasing the
coupling capacitance reduces the corner frequency further.
As in the dc-coupled case, the bias levels on the I and Q inputs
should be set to as precise a level as possible, relative to each
other. This prevents the introduction of additional input offset
voltages. In Figure 29, the bias level on each input is set to
approximately 1.2 V. The 2.43 kΩ resistors should have a ratio
tolerance of 0.1% or better.
Note that the LO suppression can be manually optimized by
replacing a portion of the four top 2.43 kΩ resistors with
potentiometers. In this case, the bottom four resistors in the
biasing network no longer need to be precision devices.
5V
5V
1kΩ
DVDD
DCOM
IOUTA
LATCH
I
2×
2.43kΩ
2.43kΩ
AVDD
I
DAC
IOUTB
100Ω
0.01μF
CFILTER
IBBP
VPS1
VPS2
2.43kΩ
100Ω 0.01μF
DAC
DATA
INPUTS
0.1μF
Σ
IBBN
2.43kΩ
VOUT
AD9761
LOIP
QOUTA
LATCH
Q
2×
Q
DAC
QOUTB
2.43kΩ
FS ADJ
RSET
2kΩ
QBBP
PHASE
SPLITTER
LOIN
2.43kΩ
QBBN
0.01μF
2.43kΩ
MUX
CONTROL
SLEEP
2.43kΩ
AD8346
1V p-p EACH PIN
WITH VCM = 1.2V
REFIO
05335-029
CLOCK
0.01μF
CFILTER
100Ω
SELECT
WRITE
100Ω
0.1μF
Figure 29. AC-Coupled DAC Interface
Rev. A | Page 13 of 20
AD8346
EVALUATION BOARD
All connectors are of the SMA type. The I and Q inputs are
provided with pads for implementing a simple RC filter
network. The local oscillator input is driven through a balun
(M/A-COM Part Number ETC1-1-13).
The schematic of the AD8346 evaluation board is shown in
Figure 30. This is a 4-layer FR4 board; the two center layers are
used as ground planes and the top and bottom layers are used
for signal and power. Figure 31 shows the layout and Figure 32
shows the silkscreen. The evaluation board circuit closely
follows the basic connections circuit shown in Figure 27.
Slide SW1 to the A position to connect the ENBL pin to +VS
via the 10 kΩ pull-up resistor REP. Slide SW1 to the B position
to disable the device by grounding the ENOP pin through the
49.9 Ω pull-down resistor REG. The device may be enabled via
an external voltage applied to the SMA connector ENOP or TP2.
CIP
OPEN
RIP
IP
1 IBBP
0Ω
0Ω
5
CLON
100pF
1
CLOP
100pF
T1
2
ETC1-1-13
RLOS
OPEN
4
3
RLOP
OPEN
COM4
4 COM1
COM4 13
5 LOIN
VPS2 12
14
6 LOIP
VOUT 11
7 VPS1
COM3 10
8 ENBL
COM2
TP2
ENOP
+VS
C1
0.01μF
R7
0Ω
QN
0Ω
CQN
OPEN
RLON
OPEN
LO
RQN
QBBN 15
3 COM1
QP
0Ω
RQS
OPEN
2 IBBN
CIN
OPEN
RQP
QBBP 16
RIS
OPEN
RIN
IN
CQP
OPEN
AD8346
R2
C4
100pF
0Ω
C3
0.01μF
+VS
CVO
100pF
VOUT
9
C2
100pF
REP
10kΩ
A
ENOP
B
REG
49.9kΩ
Figure 30. Evaluation Board Schematic
Rev. A | Page 14 of 20
05335-030
SW1
05335-031
AD8346
05335-032
Figure 31. Layout of Evaluation Board
Figure 32. Silkscreen of Evaluation Board
Rev. A | Page 15 of 20
AD8346
CHARACTERIZATION SETUPS
SSB SETUP
Two main setups were used to characterize this product. These
setups are shown in Figure 33 and Figure 35. Figure 33 shows
the setup used to evaluate the product as an SSB. The AD8346
motherboard had circuitry that converted the single-ended
I and Q inputs from the arbitrary function generator to differential inputs with a dc bias of approximately 1.2 V. In addition,
the motherboard also provided connections for power supply
routing. The HP34970A and its associated plug-in 34901 were
used to monitor power supply currents and voltages being
supplied to the AD8346 evaluation board (a full schematic of
IEEE
the AD8346 evaluation board can be found in Figure 30).
The two HP34907 plug-ins were used to provide additional
miscellaneous dc and control signals to the motherboard. The
LO was driven by an RF signal generator (through the balun on
the evaluation board to present a differential LO signal to the
device) and the output was measured with a spectrum analyzer.
With the I channel driven with a sine wave and the Q channel
driven with a cosine wave, the lower sideband is the single
sideband output. The typical SSB output spectrum is shown in
Figure 34.
HP34970A
D1
D2
D3
34901 34907 34907
D1
+15V MAX
COM
+25V MAX
–25V MAX
IEEE
TEKAFG2020
D3
AD8346
I IN
OUTPUT 1
Q IN
OUTPUT 2
VN MOTHERBOARD
GND
VP
HP3631
P1
IN
HP8648C
IEEE
D2
VPS1
IN
IP
QP
IP
QP
QN
AD8346
QN
HP8593E
SWEEP OUT
EVAL BOARD
LO
ENBL
RFOUT
IEEE
ARB FUNC. GEN
VOUT
P1
RF I/P
CAL OUT
28VOLT
IEEE
05335-033
SPECTRUM
ANALYZER
IEEE
PC CONTROLLER
Figure 33. Evaluation Board SSB Test Setup
0
–10
–20
–30
–40
–50
–60
–70
05335-034
–80
–90
–100
CENTER 1.9GHz
50kHz/
SPAN 500kHz
Figure 34. Typical SSB Output Spectrum
Rev. A | Page 16 of 20
AD8346
CDMA SETUP
For evaluating the AD8346 with CDMA waveforms, the setup
shown in Figure 35 was used. This is essentially the same setup
as that used for the single sideband characterization, except that
the AFG2020 was replaced with the AWG2021 for providing the
I and Q input signals, and the spectrum analyzer used to monitor
the output was changed to an FSEA30 Rohde & Schwarz analyzer
with vector demodulation capability. The I/Q input signals for
these measurements were IS95 baseband signals generated with
Tektronix I/Q SIM software and downloaded to the AWG2021.
IEEE
For measuring ACPR, the I/Q input signals used were generated
with Pilot (Walsh Code 00), Sync (WC 32), Paging (WC 01),
and 6 Traffic (WC 08, 09, 10, 11, 12, 13) channels active. The
I/Q SIM software was set for 32× oversampling and was using a
BS equifilter. Figure 36 shows the typical output spectrum for
this configuration. The ACPR was measured 885 kHz away
from the carrier frequency.
For performing EVM, Rho, phase, and amplitude balance
measurements, the I/Q input signals used were generated with
only the pilot channel (Walsh Code 00) active. The I/Q SIM
software was set for 32× oversampling using a CDMA equifilter.
HP34970A
D1
D2
D3
34901 34907 34907
D1
IEEE
HP3631
AD8346
IN
IEEE
RFOUT
TEKAFG2020
D3
I IN
OUTPUT 1
Q IN
OUTPUT 2
VN MOTHERBOARD
GND
VP
P1
HP8648C
D2
VPS1
IN
IP
QP
IP
QP
QN
AD8346
QN
FSEA30
EVAL BOARD
LO
ENBL
IEEE
ARB FUNC. GEN
VOUT
P1
RF I/P
IEEE
SPECTRUM
ANALYZER
05335-035
+15V MAX
COM
+25V MAX
–25V MAX
IEEE
PC CONTROLLER
Figure 35. Evaluation Board CDMA Test Setup
–20
–30
–40
–50
–60
CH PWR = –20.7dBm
ACP UPR = –71.8dBc
ACP LWR = –71.7dBc
–70
–80
–90
05335-036
–100
–110
–120
CENTER 1.9GHz
187.5kHz/
SPAN 1.875MHz
Figure 36. Typical CDMA Output Spectrum
Rev. A | Page 17 of 20
AD8346
OUTLINE DIMENSIONS
5.10
5.00
4.90
16
9
4.50
4.40
4.30
6.40
BSC
1
8
PIN 1
1.20
MAX
0.15
0.05
0.20
0.09
0.30
0.19
0.65
BSC
COPLANARITY
0.10
SEATING
PLANE
8°
0°
0.75
0.60
0.45
COMPLIANT TO JEDEC STANDARDS MO-153AB
Figure 37.16-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-16)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8346ARU
AD8346ARU-REEL
AD8346ARU-REEL7
AD8346ARUZ-REEL 1
AD8346ARUZ-REEL71
AD8346-EVAL
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
16-Lead Thin Shrink Small Outline Package (TSSOP)
16-Lead (TSSOP) 13" Tape and Reel
16-Lead (TSSOP) 7" Tape and Reel
16-Lead (TSSOP) 13" Tape and Reel
16-Lead (TSSOP) 7" Tape and Reel
Evaluation Board
Z = Pb-free part.
Rev. A | Page 18 of 20
Package Option
RU-16
RU-16
RU-16
RU-16
RU-16
AD8346
NOTES
Rev. A | Page 19 of 20
AD8346
NOTES
©2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective companies.
C05335–0–6/05(A)
Rev. A | Page 20 of 20
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