TS4871
TS4871
OUTPUT RAIL TO RAIL 1W AUDIO POWER AMPLIFIER
WITH STANDBY MODE
■ OPERATING FROM VCC = 2.5V to 5.5V
PIN CONNECTIONS (Top View)
■ 1W RAIL TO RAIL OUTPUT POWER @
Vcc=5V, THD=1%, f=1kHz, with 8Ω Load
TS4871IST - MiniSO8
■ ULTRA LOW CONSUMPTION IN STANDBY
MODE (10nA)
Standby
1
8
VOUT2
Bypass
2
7
GND
VIN+
3
6
VCC
VIN-
4
5
VOUT1
■ 75dB PSRR @ 217Hz from 5V to 2.6V
■ ULTRA LOW POP & CLICK
■ ULTRA LOW DISTORTION (0.1%)
■ UNITY GAIN STABLE
■ AVAILABLE IN SO8, MiniSO8 & DFN8 3x3mm
TS4871ID-TS4871IDT - SO8
DESCRIPTION
Standby
1
8
VOUT2
Bypass
2
7
GND
VIN+
3
6
VCC
VIN-
4
5
VOUT1
The TS4871 is an Audio Power Amplifier capable
of delivering 1W of continuous RMS Ouput Power
into 8Ω load @ 5V.
This Audio Amplifier is exhibiting 0.1% distortion
level (THD) from a 5V supply for a Pout = 250mW
RMS. An external standby mode control reduces
the supply current to less than 10nA. An internal
thermal shutdown protection is also provided.
TS4871IQT - DFN8
The TS4871 has been designed for high quality
audio applications such as mobile phones and to
minimize the number of external components.
STANDBY
1
8
VOUT 2
BYPASS
2
7
GND
VIN+
3
6
Vcc
VIN-
4
5
VOUT 1
The unity-gain stable amplifier can be configured
by external gain setting resistors.
APPLICATIONS
■ Mobile Phones (Cellular / Cordless)
■ Laptop / Notebook Computers
TYPICAL APPLICATION SCHEMATIC
Cfeed
■ PDAs
Rfeed
Audio
Input
ORDER CODE
-40, +85°C
Cin
4
3
VinVin+
-
Vout1 5
+
Package
Marking
D
S
Q
•
•
RL
8 Ohms
Vcc
•
4871I
4871
2
Bypass
1
Standby
Av=-1
+
Vout2
8
Rstb
Bias
GND
TS4871
Temperature
Range: I
Rin
Cs
Vcc
■ Portable Audio Devices
Part
Number
Vcc
6
Cb
TS4871
7
MiniSO & DFN only available in Tape & Reel with T suffix(IST & IQT)
D = Small Outline Package (SO) - also available in Tape & Reel (DT)
June 2003
1/28
TS4871
ABSOLUTE MAXIMUM RATINGS
Symbol
VCC
Vi
Parameter
Supply voltage
1)
2)
Unit
6
V
GND to VCC
V
Toper
Operating Free Air Temperature Range
-40 to + 85
°C
Tstg
Storage Temperature
Tj
Rthja
Pd
Input Voltage
-65 to +150
°C
Maximum Junction Temperature
150
°C
Thermal Resistance Junction to Ambient 3)
SO8
MiniSO8
QNF8
175
215
70
Internally Limited4)
2
200
Class A
260
Power Dissipation
ESD
Human Body Model
ESD
Machine Model
Latch-up Latch-up Immunity
Lead Temperature (soldering, 10sec)
1.
2.
3.
4.
Value
°C/W
kV
V
°C
All voltages values are measured with respect to the ground pin.
The magnitude of input signal must never exceed VCC + 0.3V / G ND - 0.3V
Device is protected in case of over temperature by a thermal shutdown active @ 150°C.
Exceeding the power derating curves during a long period, involves abnormal operating condition.
OPERATING CONDITIONS
Symbol
VCC
Parameter
Supply Voltage
VICM
Common Mode Input Voltage Range
VSTB
Standby Voltage Input :
Device ON
Device OFF
RL
Rthja
2.5 to 5.5
V
V
GND ≤ VSTB ≤ 0.5V
VCC - 0.5V ≤ VSTB ≤ VCC
V
4 - 32
Ω
1)
1. This thermal resistance can be reduced with a suitable PCB layout (see Power Derating Curves Fig. 20)
2. When mounted on a 4 layers PCB
2/28
Unit
GND to VCC - 1.2V
Load Resistor
Thermal Resistance Junction to Ambient
SO8
MiniSO8
DFN8 2)
Value
150
190
41
°C/W
TS4871
ELECTRICAL CHARACTERISTICS
VCC = +5V, GND = 0V, Tamb = 25°C (unless otherwise specified)
Symbol
Typ.
Max.
Unit
Supply Current
No input signal, no load
6
8
mA
Standby Current 1)
No input signal, Vstdby = Vcc, RL = 8Ω
10
1000
nA
Voo
Output Offset Voltage
No input signal, RL = 8Ω
5
20
mV
Po
Output Power
THD = 1% Max, f = 1kHz, RL = 8Ω
1
W
0.15
%
Power Supply Rejection Ratio2)
f = 217Hz, RL = 8Ω, RFeed = 22KΩ, Vripple = 200mV rms
75
dB
ΦM
Phase Margin at Unity Gain
RL = 8Ω, CL = 500pF
70
Degrees
GM
Gain Margin
RL = 8Ω, CL = 500pF
20
dB
GBP
Gain Bandwidth Product
RL = 8Ω
2
MHz
ICC
ISTANDBY
THD + N
PSRR
Parameter
Min.
Total Harmonic Distortion + Noise
Po = 250mW rms, Gv = 2, 20Hz < f < 20kHz, RL = 8Ω
1. Standby mode is actived when Vstdby is tied to Vcc
2. Dynamic measurements - 20*log(rms(Vout)/rms(Vripple)). Vripple is the surimposed sinus signal to Vcc @ f = 217Hz
VCC = +3.3V, GND = 0V, Tamb = 25°C (unless otherwise specified)3)
Symbol
Typ.
Max.
Unit
Supply Current
No input signal, no load
5.5
8
mA
Standby Current 1)
No input signal, Vstdby = Vcc, RL = 8Ω
10
1000
nA
Voo
Output Offset Voltage
No input signal, RL = 8Ω
5
20
mV
Po
Output Power
THD = 1% Max, f = 1kHz, RL = 8Ω
450
mW
Total Harmonic Distortion + Noise
Po = 250mW rms, Gv = 2, 20Hz < f < 20kHz, RL = 8Ω
0.15
%
Power Supply Rejection Ratio2)
f = 217Hz, RL = 8Ω, RFeed = 22KΩ, Vripple = 200mV rms
75
dB
ΦM
Phase Margin at Unity Gain
RL = 8Ω, CL = 500pF
70
Degrees
GM
Gain Margin
RL = 8Ω, CL = 500pF
20
dB
GBP
Gain Bandwidth Product
RL = 8Ω
2
MHz
ICC
ISTANDBY
THD + N
PSRR
Parameter
Min.
1. Standby mode is actived when Vstdby is tied to Vcc
2. Dynamic measurements - 20*log(rms(Vout)/rms(Vripple)). Vripple is the surimposed sinus signal to Vcc @ f = 217Hz
3. All electrical values are made by correlation between 2.6V and 5V measurements
3/28
TS4871
ELECTRICAL CHARACTERISTICS
VCC = 2.6V, GND = 0V, Tamb = 25°C (unless otherwise specified)
Symbol
Typ.
Max.
Unit
Supply Current
No input signal, no load
5.5
8
mA
Standby Current 1)
No input signal, Vstdby = Vcc, RL = 8Ω
10
1000
nA
Voo
Output Offset Voltage
No input signal, RL = 8Ω
5
20
mV
Po
Output Power
THD = 1% Max, f = 1kHz, RL = 8Ω
260
mW
Total Harmonic Distortion + Noise
Po = 200mW rms, Gv = 2, 20Hz < f < 20kHz, RL = 8Ω
0.15
%
Power Supply Rejection Ratio2)
f = 217Hz, RL = 8Ω, RFeed = 22KΩ, Vripple = 200mV rms
75
dB
ΦM
Phase Margin at Unity Gain
RL = 8Ω, CL = 500pF
70
Degrees
GM
Gain Margin
RL = 8Ω, CL = 500pF
20
dB
GBP
Gain Bandwidth Product
RL = 8Ω
2
MHz
ICC
ISTANDBY
THD + N
PSRR
Parameter
Min.
1. Standby mode is actived when Vstdby is tied to Vcc
2. Dynamic measurements - 20*log(rms(Vout)/rms(Vripple)). Vripple is the surimposed sinus signal to Vcc @ f = 217Hz
Components
Functional Description
Rin
Inverting input resistor which sets the closed loop gain in conjunction with Rfeed. This resistor also
forms a high pass filter with Cin (fc = 1 / (2 x Pi x Rin x Cin))
Cin
Input coupling capacitor which blocks the DC voltage at the amplifier input terminal
Rfeed
Feed back resistor which sets the closed loop gain in conjunction with Rin
Cs
Supply Bypass capacitor which provides power supply filtering
Cb
Bypass pin capacitor which provides half supply filtering
Cfeed
Rstb
Gv
Low pass filter capacitor allowing to cut the high frequency
(low pass filter cut-off frequency 1 / (2 x Pi x Rfeed x Cfeed))
Pull-up resistor which fixes the right supply level on the standby pin
Closed loop gain in BTL configuration = 2 x (Rfeed / Rin)
REMARKS
1. All measurements, except PSRR measurements, are made with a supply bypass capacitor Cs = 100µF.
2. External resistors are not needed for having better stability when supply @ Vcc down to 3V. By the way,
the quiescent current remains the same.
3. The standby response time is about 1µs.
4/28
TS4871
Fig. 1 : Open Loop Frequency Response
Fig. 2 : Open Loop Frequency Response
0
-60
40
-80
-100
-120
-140
-60
-80
Phase
-100
20
-120
-140
0
-160
-160
-180
-20
-180
-20
-200
-40
0.3
1
10
100
1000
-200
-220
10000
-40
0.3
1
10
Frequency (kHz)
Fig. 3 : Open Loop Frequency Response
60
Vcc = 3.3V
RL = 8Ω
Tamb = 25°C
-60
-100
-120
20
-140
-160
0
Phase (Deg)
Gain (dB)
Phase
Gain
60
-40
-80
40
0
80
-20
Gain (dB)
Gain
Vcc = 3.3V
ZL = 8Ω + 560pF
Tamb = 25°C
Phase
10
100
1000
Frequency (kHz)
10000
-140
-160
-180
-200
-20
-220
-40
0.3
-240
Fig. 5 : Open Loop Frequency Response
Gain
60
Vcc = 2.6V
RL = 8Ω
Tamb = 25°C
60
-40
-60
-120
20
-140
-160
0
10000
Vcc = 2.6V
ZL = 8Ω + 560pF
Tamb = 25°C
Phase
-200
1
10
100
1000
Frequency (kHz)
10000
-240
-40
-60
-120
-140
-160
0
-180
-200
-20
-220
-220
-40
0.3
-20
-100
20
-180
-20
-240
-80
40
Gain (dB)
-100
100
1000
Frequency (kHz)
0
Gain
Phase (Deg)
Gain (dB)
Phase
10
80
-20
-80
40
1
Fig. 6 : Open Loop Frequency Response
0
80
-60
-120
0
-200
1
-40
-100
20
-220
-40
0.3
-20
-80
40
-180
-20
-220
10000
Fig. 4 : Open Loop Frequency Response
0
80
100
1000
Frequency (kHz)
Phase (Deg)
0
-40
Phase (Deg)
20
-20
Vcc = 5V
ZL = 8Ω + 560pF
Tamb = 25°C
Phase (Deg)
Gain
-40
Phase
Gain (dB)
60
Gain (dB)
40
0
-20
Vcc = 5V
RL = 8Ω
Tamb = 25°C
Gain
Phase (Deg)
60
-40
0.3
1
10
100
1000
Frequency (kHz)
10000
-240
5/28
TS4871
Phase
60
100
-100
80
-120
60
Gain (dB)
Gain
-140
40
-160
20
0
-20
-40
0.3
-180
1
10
100
-40
0.3
-80
80
-100
Phase
Gain (dB)
Gain
-140
40
-160
20
-180
0
-40
0.3
6/28
-200
Vcc = 2.6V
CL = 560pF
Tamb = 25°C
1
10
-220
100
1000
Frequency (kHz)
10000
-240
Phase (Deg)
-120
60
-20
-180
-220
Fig. 9 : Open Loop Frequency Response
-140
-160
-20
10000
-120
20
-200
100
1000
Frequency (kHz)
-100
Phase
40
0
Vcc = 5V
CL = 560pF
Tamb = 25°C
-80
Gain
Gain (dB)
80
-80
Phase (Deg)
100
Fig. 8 : Open Loop Frequency Response
-200
Vcc = 3.3V
CL = 560pF
Tamb = 25°C
1
10
-220
100
1000
Frequency (kHz)
10000
-240
Phase (Deg)
Fig. 7 : Open Loop Frequency Response
TS4871
Fig. 10 : Power Supply Rejection Ratio (PSRR)
vs Power supply
Fig. 11 : Power Supply Rejection Ratio (PSRR)
vs Feedback Capacitor
-30
-10
Vripple = 200mVrms
Rfeed = 22Ω
Input = floating
RL = 8Ω
Tamb = 25°C
-50
-20
-30
PSRR (dB)
PSRR (dB)
-40
Vcc = 5V, 3.3V & 2.6V
Cb = 1µF & 0.1µF
-60
-40
Vcc = 5, 3.3 & 2.6V
Cb = 1µF & 0.1µF
Rfeed = 22kΩ
Vripple = 200mVrms
Input = floating
RL = 8Ω
Tamb = 25°C
Cfeed=0
Cfeed=150pF
Cfeed=330pF
-50
-60
-70
-70
-80
10
100
1000
10000
Frequency (Hz)
-80
10
100000
Fig. 12 : Power Supply Rejection Ratio (PSRR)
vs Bypass Capacitor
-10
Cb=10µF
PSRR (dB)
-30
-40
1000
10000
Frequency (Hz)
Cin=1µF
Cin=330nF
Vcc = 5, 3.3 & 2.6V
Rfeed = 22k
Rin = 22k, Cin = 1µF
Rg = 100Ω, RL = 8Ω
Tamb = 25°C
-20
PSRR (dB)
-20
100
Cb=47µF
-50
100000
Fig. 13 : Power Supply Rejection Ratio (PSRR)
vs Input Capacitor
-10
Cb=1µF
Cfeed=680pF
Cin=220nF
-30
Vcc = 5, 3.3 & 2.6V
Rfeed = 22kΩ, Rin = 22k
Cb = 1µF
Rg = 100Ω, RL = 8Ω
Tamb = 25°C
-40
Cin=100nF
-60
-70
-50
Cb=100µF
-80
10
-60
10
100
1000
10000
100000
Cin=22nF
100
1000
10000
100000
Frequency (Hz)
Frequency (Hz)
Fig. 14 : Power Supply Rejection Ratio (PSRR)
vs Feedback Resistor
-10
-20
PSRR (dB)
-30
-40
Vcc = 5, 3.3 & 2.6V
Cb = 1µF & 0.1µF
Vripple = 200mVrms
Input = floating
RL = 8Ω
Tamb = 25°C
Rfeed=110kΩ
Rfeed=47kΩ
-50
-60
Rfeed=22kΩ
-70
Rfeed=10kΩ
-80
10
100
1000
10000
Frequency (Hz)
100000
7/28
TS4871
Fig. 16 : Pout @ THD + N = 10% vs Supply
Voltage vs RL
Fig. 15 : Pout @ THD + N = 1% vs Supply
Voltage vs RL
2.0
8Ω
Gv = 2 & 10
Cb = 1µF
F = 1kHz
BW < 125kHz
Tamb = 25°C
1.2
1.0
Output power @ 10% THD + N (W)
Output power @ 1% THD + N (W)
1.4
6Ω
4Ω
0.8
16Ω
0.6
0.4
0.2
32Ω
0.0
2.5
3.0
3.5
4.0
4.5
Gv = 2 & 10
Cb = 1µF
F = 1kHz
BW < 125kHz
Tamb = 25°C
1.8
1.6
1.4
4Ω
1.2
1.0
16Ω
0.8
0.6
0.4
0.2
32Ω
0.0
2.5
5.0
8Ω
3.0
3.5
4.5
5.0
Fig. 18 : Power Dissipation vs Pout
1.4
0.6
Vcc=5V
1.2 F=1kHz
THD+N<1%
Vcc=3.3V
F=1kHz
0.5 THD+N<1%
RL=4Ω
Power Dissipation (W)
Power Dissipation (W)
Fig. 17 : Power Dissipation vs Pout
1.0
0.8
0.6
RL=8Ω
0.4
RL=4Ω
0.4
0.3
0.2
RL=8Ω
0.1
0.2
RL=16Ω
0.0
0.0
0.2
0.4
0.6
0.8
RL=16Ω
1.0
1.2
0.0
0.0
1.4
0.2
Output Power (W)
0.4
0.6
0.8
Output Power (W)
Fig. 19 : Power Dissipation vs Pout
Fig. 20 : Power Derating Curves
0.40
2.0
Vcc=2.6V
F=1kHz
THD+N<1%
1.8
1.6
RL=4Ω
0.30
Power Dissipation (W)
Power Dissipation (W)
4.0
Vcc (V)
Vcc (V)
0.35
6Ω
0.25
0.20
0.15
RL=8Ω
0.10
QFN8
1.4
1.2
1.0
SO8
0.8
0.6
0.4
0.05
0.00
0.0
RL=16Ω
0.0
0.1
0.2
Output Power (W)
8/28
MiniSO8
0.2
0.3
0.4
0
25
50
75
100
Ambiant Temperature (°C)
125
150
TS4871
Fig. 21 : THD + N vs Output Power
Fig. 22 : THD + N vs Output Power
10
10
RL = 4Ω, Vcc = 5V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz, Tamb = 25°C
THD + N (%)
THD + N (%)
Rl = 4Ω
Vcc = 5V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20kHz
20kHz
1
20Hz
1kHz
20Hz, 1kHz
0.1
1E-3
0.01
0.1
Output Power (W)
0.1
1E-3
1
Fig. 23 : THD + N vs Output Power
1
Fig. 24 : THD + N vs Output Power
10
10
RL = 4Ω, Vcc = 3.3V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
0.01
0.1
Output Power (W)
1
RL = 4Ω, Vcc = 3.3V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
20kHz
1
20kHz
0.1
20Hz
1kHz
20Hz, 1kHz
0.1
1E-3
0.01
0.1
Output Power (W)
1
Fig. 25 : THD + N vs Output Power
0.01
0.1
Output Power (W)
1
Fig. 26 : THD + N vs Output Power
10
RL = 4Ω, Vcc = 2.6V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
10
1E-3
1
RL = 4Ω, Vcc = 2.6V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20kHz
20kHz
20Hz
0.1
1kHz
20Hz, 1kHz
0.1
1E-3
0.01
Output Power (W)
0.1
1E-3
0.01
Output Power (W)
0.1
9/28
TS4871
Fig. 27 : THD + N vs Output Power
Fig. 28 : THD + N vs Output Power
10
RL = 8Ω
Vcc = 5V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20Hz, 1kHz
THD + N (%)
THD + N (%)
10
20kHz
0.1
RL = 8Ω
Vcc = 5V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20Hz
20kHz
0.1
1kHz
1E-3
0.01
0.1
Output Power (W)
1
1E-3
Fig. 29 : THD + N vs Output Power
1
Fig. 30 : THD + N vs Output Power
10
10
RL = 8Ω, Vcc = 3.3V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
0.01
0.1
Output Power (W)
1
RL = 8Ω, Vcc = 3.3V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20kHz
20Hz
20kHz
20Hz, 1kHz
0.1
0.1
1kHz
1E-3
0.01
0.1
Output Power (W)
1
Fig. 31 : THD + N vs Output Power
1
10
RL = 8Ω, Vcc = 2.6V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20Hz, 1kHz
1E-3
RL = 8Ω, Vcc = 2.6V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
20Hz
20kHz
20kHz
0.1
10/28
0.01
0.1
Output Power (W)
Fig. 32 : THD + N vs Output Power
THD + N (%)
THD + N (%)
10
1E-3
0.1
0.01
Output Power (W)
0.1
1E-3
1kHz
0.01
Output Power (W)
0.1
TS4871
Fig. 33 : THD + N vs Output Power
Fig. 34 : THD + N vs Output Power
10
RL = 8Ω
Vcc = 5V
Gv = 2
Cb = 0.1µF, Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
RL = 8Ω, Vcc = 5V, Gv = 10
Cb = 0.1µF, Cin = 1µF
BW < 125kHz, Tamb = 25°C
20Hz
THD + N (%)
THD + N (%)
10
20Hz
20kHz
1kHz
1
20kHz
1kHz
0.1
0.1
1E-3
0.01
0.1
Output Power (W)
1
1E-3
Fig. 35 : THD + N vs Output Power
10
RL = 8Ω, Vcc = 3.3V
Gv = 2
Cb = 0.1µF, Cin = 1µF
BW < 125kHz
Tamb = 25°C
THD + N (%)
RL = 8Ω, Vcc = 3.3V, Gv = 10
Cb = 0.1µF, Cin = 1µF
BW < 125kHz, Tamb = 25°C
1
20Hz
20kHz
1
20kHz
20Hz
1kHz
1kHz
0.1
0.1
1E-3
0.01
0.1
Output Power (W)
1
Fig. 37 : THD + N vs Output Power
10
1E-3
0.01
0.1
Output Power (W)
1
Fig. 38 : THD + N vs Output Power
10
RL = 8Ω, Vcc = 2.6V
Gv = 2
Cb = 0.1µF, Cin = 1µF
BW < 125kHz
Tamb = 25°C
RL = 8Ω, Vcc = 2.6V, Gv = 10
Cb = 0.1µF, Cin = 1µF
BW < 125kHz, Tamb = 25°C
THD + N (%)
THD + N (%)
1
Fig. 36 : THD + N vs Output Power
10
THD + N (%)
0.01
0.1
Output Power (W)
1
20Hz
20kHz
1
20kHz
1kHz
1kHz
0.1
1E-3
20Hz
0.1
0.01
Output Power (W)
0.1
1E-3
0.01
Output Power (W)
0.1
11/28
TS4871
Fig. 39 : THD + N vs Output Power
Fig. 40 : THD + N vs Output Power
10
10
20kHz
RL = 16Ω, Vcc = 5V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
1
THD + N (%)
THD + N (%)
1
RL = 16Ω, Vcc = 5V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
20kHz
0.1
0.1
1kHz
20Hz, 1kHz
0.01
1E-3
0.01
0.1
Output Power (W)
1
Fig. 41 : THD + N vs Output Power
0.01
1E-3
20kHz
0.1
1
RL = 16Ω
Vcc = 3.3V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
20kHz
0.1
1kHz
20Hz
20Hz, 1kHz
0.01
1E-3
0.01
Output Power (W)
0.01
1E-3
0.1
Fig. 43 : THD + N vs Output Power
THD + N (%)
THD + N (%)
0.1
10
RL = 16Ω
Vcc = 2.6V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
20kHz
0.1
1
RL = 16Ω
Vcc = 2.6V
Gv = 10
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
20Hz
20kHz
0.1
20Hz, 1kHz
0.01
1E-3
12/28
0.01
Output Power (W)
Fig. 44 : THD + N vs Output Power
10
1
1
10
RL = 16Ω, Vcc = 3.3V
Gv = 2
Cb = Cin = 1µF
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
0.01
0.1
Output Power (W)
Fig. 42 : THD + N vs Output Power
10
1
20Hz
0.01
Output Power (W)
1kHz
0.1
0.01
1E-3
0.01
Output Power (W)
0.1
TS4871
Fig. 45 : THD + N vs Frequency
Pout = 1.2W
RL = 4Ω, Vcc = 5V
Gv = 2
Cb = 1µF
BW < 125kHz
Tamb = 25°C
1
THD + N (%)
THD + N (%)
1
Fig. 46 : THD + N vs Frequency
Pout = 1.2W
RL = 4Ω, Vcc = 5V
Gv = 10
Cb = 1µF
BW < 125kHz
Tamb = 25°C
Pout = 600mW
0.1
20
100
1000
Frequency (Hz)
0.01
20
10000
Pout = 600mW
0.1
100
1000
Frequency (Hz)
Fig. 47 : THD + N vs Frequency
Fig. 48 : THD + N vs Frequency
RL = 4Ω, Vcc = 3.3V
Gv = 2
Cb = 1µF
BW < 125kHz
Tamb = 25°C
RL = 4Ω, Vcc = 3.3V
Gv = 10
Cb = 1µF
BW < 125kHz
Tamb = 25°C
1
THD + N (%)
THD + N (%)
1
Pout = 540mW
Pout = 540mW
Pout = 270mW
100
1000
Frequency (Hz)
Pout = 270mW
Fig. 49 : THD + N vs Frequency
THD + N (%)
1
0.1
20
10000
100
1000
Frequency (Hz)
10000
Fig. 50 : THD + N vs Frequency
RL = 4Ω, Vcc = 2.6V
Gv = 2
Cb = 1µF
BW < 125kHz
Tamb = 25°C
RL = 4Ω, Vcc = 2.6V
Gv = 10
Cb = 1µF
BW < 125kHz
Tamb = 25°C
1
Pout = 240mW
THD + N (%)
0.1
20
10000
Pout = 240 & 120mW
Pout = 120mW
0.1
20
100
1000
Frequency (Hz)
10000
0.1
20
100
1000
Frequency (Hz)
10000
13/28
TS4871
Fig. 51 : THD + N vs Frequency
Fig. 52 : THD + N vs Frequency
1
Cb = 0.1µF
Cb = 1µF
RL = 8Ω
Vcc = 5V
Gv = 2
Pout = 450mW
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
1
RL = 8Ω
Vcc = 5V
Gv = 2
Pout = 900mW
BW < 125kHz
Tamb = 25°C
Cb = 0.1µF
Cb = 1µF
0.1
100
1000
Frequency (Hz)
Fig. 53 : THD + N vs Frequency
THD + N (%)
Cb = 0.1µF
RL = 8Ω, Vcc = 5V
Gv = 10
Pout = 450mW
BW < 125kHz
Tamb = 25°C
Cb = 0.1µF
0.1
100
1000
Frequency (Hz)
10000
Fig. 55 : THD + N vs Frequency
20
100
1000
Frequency (Hz)
10000
Fig. 56 : THD + N vs Frequency
1
1
Cb = 0.1µF
Cb = 1µF
RL = 8Ω, Vcc = 3.3V
Gv = 2
Pout = 200mW
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
RL = 8Ω, Vcc = 3.3V
Gv = 2
Pout = 400mW
BW < 125kHz
Tamb = 25°C
0.1
14/28
10000
Cb = 1µF
0.1
20
1000
Frequency (Hz)
1
Cb = 1µF
20
100
Fig. 54 : THD + N vs Frequency
RL = 8Ω, Vcc = 5V
Gv = 10
Pout = 900mW
BW < 125kHz
Tamb = 25°C
1
0.1
20
10000
THD + N (%)
20
Cb = 0.1µF
Cb = 1µF
0.1
100
1000
Frequency (Hz)
10000
20
100
1000
Frequency (Hz)
10000
TS4871
RL = 8Ω, Vcc = 3.3V
Gv = 10
Pout = 400mW
BW < 125kHz
Tamb = 25°C
1
THD + N (%)
Fig. 58 : THD + N vs Frequency
Cb = 0.1µF
Cb = 1µF
RL = 8Ω, Vcc = 3.3V
Gv = 10
Pout = 200mW
BW < 125kHz
Tamb = 25°C
1
THD + N (%)
Fig. 57 : THD + N vs Frequency
Cb = 0.1µF
Cb = 1µF
0.1
0.1
20
100
1000
Frequency (Hz)
10000
Fig. 59 : THD + N vs Frequency
20
100
1000
Frequency (Hz)
10000
Fig. 60 : THD + N vs Frequency
1
THD + N (%)
Cb = 0.1µF
RL = 8Ω, Vcc = 2.6V
Gv = 2
Pout = 220mW
BW < 125kHz
Tamb = 25°C
Cb = 1µF
RL = 8Ω, Vcc = 2.6V
Gv = 2
Pout = 110mW
BW < 125kHz
Tamb = 25°C
THD + N (%)
1
Cb = 0.1µF
Cb = 1µF
0.1
100
1000
Frequency (Hz)
10000
Fig. 61 : THD + N vs Frequency
THD + N (%)
100
1000
Frequency (Hz)
10000
Fig. 62 : THD + N vs Frequency
RL = 8Ω, Vcc = 2.6V
Gv = 10
Pout = 220mW
BW < 125kHz
Tamb = 25°C
1
20
Cb = 0.1µF
1
Cb = 0.1µF
THD + N (%)
20
0.1
RL = 8Ω, Vcc = 2.6V
Gv = 10
Pout = 110mW
BW < 125kHz
Tamb = 25°C
Cb = 1µF
Cb = 1µF
0.1
0.1
20
100
1000
Frequency (Hz)
10000
20
100
1000
Frequency (Hz)
10000
15/28
TS4871
Fig. 63 : THD + N vs Frequency
Fig. 64 : THD + N vs Frequency
1
1
RL = 16Ω, Vcc = 5V
Gv = 10, Cb = 1µF
BW < 125kHz
Tamb = 25°C
THD + N (%)
THD + N (%)
RL = 16Ω, Vcc = 5V
Gv = 2, Cb = 1µF
BW < 125kHz
Tamb = 25°C
Pout = 310mW
0.1
Pout = 620mW
0.1
Pout = 310mW
Pout = 620mW
0.01
20
100
1000
Frequency (Hz)
0.01
20
10000
Fig. 65 : THD + N vs Frequency
100
1000
Frequency (Hz)
10000
Fig. 66 : THD + N vs Frequency
1
1
THD + N (%)
THD + N (%)
RL = 16Ω, Vcc = 3.3V
Gv = 2, Cb = 1µF
BW < 125kHz
Tamb = 25°C
Pout = 270mW
0.1
RL = 16Ω, Vcc = 3.3V
Gv = 10
Cb = 1µF
BW < 125kHz
Tamb = 25°C
Pout = 270mW
0.1
Pout = 135mW
Pout = 135mW
0.01
20
100
1000
Frequency (Hz)
20
10000
Fig. 67 : THD + N vs Frequency
THD + N (%)
RL = 16Ω, Vcc = 2.6V
Gv = 10, Cb = 1µF
BW < 125kHz
Tamb = 25°C
Pout = 80mW
0.1
100
1000
Frequency (Hz)
Pout = 160mW
0.1
Pout = 80mW
Pout = 160mW
16/28
10000
1
RL = 16Ω, Vcc = 2.6V
Gv = 2, Cb = 1µF
BW < 125kHz
Tamb = 25°C
0.01
20
1000
Frequency (Hz)
Fig. 68 : THD + N vs Frequency
1
THD + N (%)
100
10000
0.01
20
100
1000
Frequency (Hz)
10000
TS4871
Fig. 69 : Signal to Noise Ratio vs Power Supply
with Unweighted Filter (20Hz to 20kHz)
Fig. 70 : Signal to Noise Ratio vs Power Supply
with Weighted Filter Type A
100
100
90
90
RL=4Ω
RL=8Ω
80
SNR (dB)
SNR (dB)
RL=16Ω
70
Gv = 2
Cb = Cin = 1µF
THD+N < 0.4%
Tamb = 25°C
60
50
2.5
3.0
3.5
4.0
4.5
RL=8Ω
RL=4Ω
RL=16Ω
80
Gv = 10
Cb = Cin = 1µF
THD+N < 0.7%
Tamb = 25°C
70
60
2.5
5.0
3.0
3.5
Fig. 71 : Signal to Noise Ratio vs Power Supply
with Weighted Filter type A
5.0
7
Vstandby = 0V
Tamb = 25°C
6
100
RL=16Ω
5
RL=4Ω
RL=8Ω
90
Icc (mA)
SNR (dB)
4.5
Fig. 72 : Current Consumption vs Power
Supply Voltage
110
80
4
3
2
Gv = 2
Cb = Cin = 1µF
THD+N < 0.4%
Tamb = 25°C
70
60
2.5
4.0
Vcc (V)
Vcc (V)
3.0
3.5
4.0
4.5
1
0
5.0
0
1
2
3
4
5
Vcc (V)
Vcc (V)
Fig. 73 : Signal to Noise Ratio Vs Power Supply
with Unweighted Filter (20Hz to 20kHz)
Fig. 74 : Current Consumption vs Standby
Voltage @ Vcc = 5V
90
7
Vcc = 5V
Tamb = 25°C
6
80
RL=8Ω
70
RL=16Ω
Icc (mA)
SNR (dB)
5
RL=4Ω
4
3
2
Gv = 10
Cb = Cin = 1µF
THD+N < 0.7%
Tamb = 25°C
60
50
2.5
3.0
3.5
4.0
Vcc (V)
4.5
5.0
1
0
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
Vstandby (V)
17/28
TS4871
Fig. 75 : Current Consumption vs Standby
Voltage @ Vcc = 2.6V
Fig. 76 : Current Consumption vs Standby
Voltage @ Vcc = 3.3V
6
6
Vcc = 2.6V
Tamb = 25°C
5
4
Icc (mA)
Icc (mA)
4
3
3
2
2
1
1
0
0.0
0.5
1.0
1.5
Vstandby (V)
2.0
0
0.0
2.5
0.5
1.0
2.5
3.0
1.0
Tamb = 25°C
0.8
0.7
0.6
0.5
RL = 4Ω
RL = 8Ω
0.4
0.3
0.2
0.1
RL = 16Ω
0.0
2.5
Tamb = 25°C
0.9
Vout1 & Vout2
Clipping Voltage Low side (V)
0.9
3.0
3.5
4.0
4.5
0.8
0.7
0.6
RL = 4Ω
0.5
RL = 8Ω
0.4
0.3
0.2
0.1
RL = 16Ω
0.0
2.5
5.0
3.0
Power supply Voltage (V)
3.5
4.0
4.5
5.0
Power supply Voltage (V)
Fig. 79 : Vout1+Vout2 Unweighted Noise Floor
Fig. 80 : Vout1+Vout2 A-weighted Noise Floor
120
Vcc = 2.5V to 5V, Tamb = 25 C
Cb = Cin = 1 F
Input Grounded
BW = 20Hz to 20kHz (Unweighted)
100
Av = 10
80
60
40
Standby mode
Av = 2
Output Noise Voltage ( V)
120
Output Noise Voltage ( V)
2.0
Fig. 78 : Clipping Voltage vs Power Supply
Voltage and Load Resistor
1.0
Vcc = 2.5V to 5V, Tamb = 25 C
Cb = Cin = 1 F
Input Grounded
BW = 20Hz to 20kHz (A-Weighted)
100
80
Av = 10
60
40
Standby mode
Av = 2
20
20
0
0
20
18/28
1.5
Vstandby (V)
Fig. 77 : Clipping Voltage vs Power Supply
Voltage and Load Resistor
Vout1 & Vout2
Clipping Voltage High side (V)
Vcc = 3.3V
Tamb = 25°C
5
100
1000
Frequency (Hz)
10000
20
100
1000
Frequency (Hz)
10000
TS4871
APPLICATION INFORMATION
Fig. 81 : Demoboard Schematic
C1
R2
C2
R1
Vcc
S1
Vcc
Vcc
S2
GND
C6 +
100µ
R3
6
C3
C5
R4
C4
Pos input
P2
S6
Vcc
Neg. input
P1
C7
100n
4
R5
VinVin+
3
-
Vout1 5
+
C9
+
470µ
S5
Positive Input mode
Vcc
Av=-1
+
Vcc
R7
330k
2
Bypass
1
Standby
S8
Standby
Vout2
8
C10
+
470µ
Bias
GND
D1
PW ON
GND
S4
GND
S7
R6
R8
OUT1
S3
TS4871
7
C11
+ C12
1u
C8
Fig. 82 : SO8 & MiniSO8 Demoboard Components Side
19/28
TS4871
Fig. 83 : SO8 & MiniSO8 Demoboard Top
Solder Layer
The output power is:
Pout =
(2 Vout RMS ) 2
(W )
RL
For the same power supply voltage, the output
power in BTL configuration is four times higher
than the output power in single ended
configuration.
■ Gain In Typical Application Schematic
(see page 1)
In flat region (no effect of Cin), the output voltage
of the first stage is:
R fe ed
Vout1 = – Vin -------------------- (V)
Rin
For the second stage : Vout2 = -Vout1 (V)
Fig. 84 : SO8 & MiniSO8 Demoboard Bottom Solder
Layer
The differential output voltage is:
Rfee d
Vout2 – Vo ut1 = 2Vin -------------------- (V)
Rin
The differential gain named gain (Gv) for more
convenient usage is:
Vout2 – Vou t1
Rfee d
Gv = --------------------------------------- = 2 -------------------Vin
Rin
Remark : Vout2 is in phase with Vin and Vout1 is
180 phased with Vin. It means that the positive
terminal of the loudspeaker should be connected
to Vout2 and the negative to Vout1.
■ Low and high frequency response
■ BTL Configuration Principle
The TS4871 is a monolithic power amplifier with a
BTL output type. BTL (Bridge Tied Load) means
that each end of the load is connected to two
single ended output amplifiers. Thus, we have :
Single ended output 1 = Vout1 = Vout (V)
Single ended output 2 = Vout2 = -Vout (V)
And Vout1 - Vout2 = 2Vout (V)
20/28
In low frequency region, the effect of Cin starts.
Cin with Rin forms a high pass filter with a -3dB cut
off frequency.
1
F C L = -------------------------------- ( Hz )
2 π R in Cin
In high frequency region, you can limit the
bandwidth by adding a capacitor (Cfeed) in
parallel on Rfeed. Its form a low pass filter with a
-3dB cut off frequency.
1
F C H = ----------------------------------------------- ( Hz )
2π Rfe ed Cfeed
TS4871
■ Power dissipation and efficiency
Hypothesis :
• Voltage and current in the load are sinusoidal
(Vout and Iout)
• Supply voltage is a pure DC source (Vcc)
Regarding the load we have:
V O UT = V PEAK sin ωt (V)
and
V OU T
I OU T = ----------------- (A)
RL
and
VPEAK 2
P O U T = ---------------------- (W)
2 RL
Then, the average current delivered by the supply
voltage is:
I CC
AVG
VPEAK
= 2 -------------------- (A)
πR L
The power delivered by the supply voltage is
Psupply = Vcc IccAVG (W)
Then, the power dissipated by the amplifier is
Pdiss = Psupply - Pout (W)
2 2 Vcc
P di ss = ---------------------- P OU T – P O UT (W)
π RL
and the maximum value is obtained when:
∂Pdiss
---------------------- = 0
∂P OU T
2 Vcc 2
π2RL
π
----- = 78.5%
4
■ Decoupling of the circuit
Two capacitors are needed to bypass properly the
TS4871, a power supply bypass capacitor Cs and
a bias voltage bypass capacitor Cb.
Cs has especially an influence on the THD+N in
high frequency (above 7kHz) and indirectly on the
power supply disturbances.
With 100µF, you can expect similar THD+N
performances like shown in the datasheet.
If Cs is lower than 100µF, in high frequency
increases, THD+N and disturbances on the power
supply rail are less filtered.
To the contrary, if Cs is higher than 100µF, those
disturbances on the power supply rail are more
filtered.
Cb has an influence on THD+N in lower frequency,
but its function is critical on the final result of PSRR
with input grounded in lower frequency.
If Cb is lower than 1µF, THD+N increase in lower
frequency (see THD+N vs frequency curves) and
the PSRR worsens up
If Cb is higher than 1µF, the benefit on THD+N in
lower frequency is small but the benefit on PSRR
is substantial (see PSRR vs. Cb curve : fig.12).
Note that Cin has a non-negligible effect on PSRR
in lower frequency. Lower is its value, higher is the
PSRR (see fig. 13).
■ Pop and Click performance
Pop and Click performance is intimately linked
with the size of the input capacitor Cin and the bias
voltage bypass capacitor Cb.
and its value is:
Pdiss max =
The maximum theoretical value is reached when
Vpeak = Vcc, so
(W)
Remark : This maximum value is only depending
on power supply voltage and load values.
The efficiency is the ratio between the output
power and the power supply
πV P E A K
P O UT
η = ------------------------ = ----------------------Psup ply
4V C C
Size of Cin is due to the lower cut-off frequency
and PSRR value requested. Size of Cb is due to
THD+N and PSRR requested always in lower
frequency.
Moreover, Cb determines the speed that the
amplifier turns ON. The slower the speed is, the
softer the turn ON noise is.
The charge time of Cb is directly proportional to
21/28
TS4871
the internal generator resistance 50kΩ.
Then, the charge time constant for Cb is
τb = 50kΩxCb (s)
As Cb is directly connected to the non-inverting
input (pin 2 & 3) and if we want to minimize, in
amplitude and duration, the output spike on Vout1
(pin 5), Cin must be charged faster than Cb. The
charge time constant of Cin is
τin = (Rin+Rfeed)xCin (s)
Thus we have the relation
τin << τb (s)
5Cs
t D i s ch C s = -------------- = 83 ms
Icc
Now, we must consider the discharge time of Cb.
At power OFF or standby ON, Cb is discharged by
a 100kΩ resistor. So the discharge time is about
τbDisch ≈ 3xCbx100kΩ (s).
In the majority of application, Cb=1µF, then
τbDisch≈300ms >> tdischCs.
■ Power amplifier design examples
Given :
The respect of this relation permits to minimize the
pop and click noise.
Remark : Minimize Cin and Cb has a benefit on
pop and click phenomena but also on cost and
size of the application.
Example : your target for the -3dB cut off
frequency is 100 Hz. With Rin=Rfeed=22 kΩ,
Cin=72nF (in fact 82nF or 100nF).
With Cb=1µF, if you choose the one of the latest
two values of Cin, the pop and click phenomena at
power supply ON or standby function ON/OFF will
be very small
50 kΩx1µF >> 44kΩx100nF (50ms >> 4.4ms).
Increasing Cin value increases the pop and click
phenomena to an unpleasant sound at power
supply ON and standby function ON/OFF.
Why Cs is not important in pop and click
consideration ?
Hypothesis :
• Cs = 100µF
• Supply voltage = 5V
• Supply voltage internal resistor = 0.1Ω
• Supply current of the amplifier Icc = 6mA
•
•
•
•
•
•
•
Load impedance : 8Ω
Output power @ 1% THD+N : 0.5W
Input impedance : 10kΩ min.
Input voltage peak to peak : 1Vpp
Bandwidth frequency : 20Hz to 20kHz (0, -3dB)
Ambient temperature max = 50°C
SO8 package
First of all, we must calculate the minimum power
supply voltage to obtain 0.5W into 8Ω. With curves
in fig. 15, we can read 3.5V. Thus, the power
supply voltage value min. will be 3.5V.
Following
equation
the
maximum
Pdiss max =
power
2 Vcc 2
π2RL
dissipation
(W)
with 3.5V we have Pdissmax=0.31W.
Refer to power derating curves (fig. 20), with
0.31W the maximum ambient temperature will be
100°C. This last value could be higher if you follow
the example layout shown on the demoboard
(better dissipation).
The gain of the amplifier in flat region will be:
At power ON of the supply, the supply capacitor is
charged through the internal power supply
resistor. So, to reach 5V you need about five to ten
times the charging time constant of Cs (τs =
0.1xCs (s)).
Then, this time equal 50µs to 100µs << τb in the
majority of application.
At power OFF of the supply, Cs is discharged by a
constant current Icc. The discharge time from 5V
to 0V of Cs is:
22/28
V OUTP P 2 2 R L P OUT
G V = --------------------- = ------------------------------------ = 5.65
VINPP
VINPP
We have Rin > 10kΩ. Let's take Rin = 10kΩ, then
Rfeed = 28.25kΩ. We could use for Rfeed = 30kΩ
in normalized value and the gain will be Gv = 6.
In lower frequency we want 20 Hz (-3dB cut off
frequency). Then:
So, we could use for Cin a 1µF capacitor value
TS4871
1
C IN = ------------------------------ = 795nF
2π RinF C L
which gives 16Hz.
In Higher frequency we want 20kHz (-3dB cut off
frequency). The Gain Bandwidth Product of the
TS4871 is 2MHz typical and doesn’t change when
the amplifier delivers power into the load.
The first amplifier has a gain of:
Rfee d
----------------- = 3
R in
and the theoretical value of the -3dB cut-off higher
frequency is 2MHz/3 = 660kHz.
We can keep this value or limit the bandwidth by
adding a capacitor Cfeed, in parallel on Rfeed.
Then:
C FE E D
1
= --------------------------------------- = 265pF
2π R F E E D F C H
So, we could use for Cfeed a 220pF capacitor
value that gives 24kHz.
Now, we can calculate the value of Cb with the
formula τb = 50kΩxCb >> τin = (Rin+Rfeed)xCin
which permits to reduce the pop and click effects.
Then Cb >> 0.8µF.
We can choose for Cb a normalized value of 2.2µF
that gives good results in THD+N and PSRR.
In the following tables, you could find three
another examples with values required for the
demoboard.
Remark : components with (*) marking are
optional.
Application n°1 : 20Hz to 20kHz bandwidth and
6dB gain BTL power amplifier.
Components :
Designator
Part Type
R1
22k / 0.125W
R4
22k / 0.125W
R6
Short Cicuit
R7
330k / 0.125W
R8*
(Vcc-Vf_led)/If_led
C5
470nF
C6
100µF
C7
100nF
C9
Short Circuit
C10
Short Circuit
C12
1µF
S1, S2, S6, S7
2mm insulated Plug
10.16mm pitch
S8
3 pts connector 2.54mm
pitch
P1
PCB Phono Jack
D1*
Led 3mm
U1
TS4871ID or TS4871IS
Application n°2 : 20Hz to 20kHz bandwidth and
20dB gain BTL power amplifier.
Components :
Designator
Part Type
R1
110k / 0.125W
R4
22k / 0.125W
R6
Short Cicuit
R7
330k / 0.125W
R8*
(Vcc-Vf_led)/If_led
C5
470nF
C6
100µF
C7
100nF
23/28
TS4871
Designator
Part Type
Application n°4 : Differential inputs BTL power
amplifier.
C9
Short Circuit
C10
Short Circuit
C12
1µF
S1, S2, S6, S7
2mm insulated Plug
10.16mm pitch
We have also : R4 = R5, R1 = R6, C4 = C5.
S8
3 pts connector 2.54mm
pitch
The gain of the amplifier is:
P1
PCB Phono Jack
D1*
Led 3mm
U1
TS4871ID or TS4871IS
Application n°3 : 50Hz to 10kHz bandwidth and
10dB gain BTL power amplifier.
In this configuration, we need to place these
components : R1, R4, R5, R6, R7, C4, C5, C12.
R1
G V D I FF = 2 -------R4
For Vcc=5V, a 20Hz to 20kHz bandwidth and 20dB
gain BTL power amplifier you could follow the bill
of material below.
Components :
Components :
Designator
Designator
Part Type
Part Type
R1
33k / 0.125W
R1
110k / 0.125W
R2
Short Circuit
R4
22k / 0.125W
R4
22k / 0.125W
R5
22k / 0.125W
R6
Short Cicuit
R6
110k / 0.125W
R7
330k / 0.125W
R7
330k / 0.125W
R8*
(Vcc-Vf_led)/If_led
R8*
(Vcc-Vf_led)/If_led
C2
470pF
C4
470nF
C5
150nF
C5
470nF
C6
100µF
C6
100µF
C7
100nF
C7
100nF
C9
Short Circuit
C9
Short Circuit
C10
Short Circuit
C10
Short Circuit
C12
1µF
C12
1µF
2mm insulated Plug
10.16mm pitch
D1*
Led 3mm
S1, S2, S6, S7
S1, S2, S6, S7
S8
3 pts connector 2.54mm
pitch
2mm insulated Plug
10.16mm pitch
P1
PCB Phono Jack
D1*
U1
24/28
S8
3 pts connector 2.54mm
pitch
Led 3mm
P1, P2
PCB Phono Jack
TS4871ID or TS4871IS
U1
TS4871ID or TS4871IS
TS4871
■ Note on how to use the PSRR curves
Fig. 86 : PSRR measurement schematic
(page 7)
We have finished a design and we have chosen
the components values :
Rfeed
6
• Rin=Rfeed=22kΩ
• Cin=100nF
• Cb=1µF
Vcc
Vripple
Vcc
4
Rin
3
VinVin+
-
Vout1
5
Vs-
+
Cin
Rg
100 Ohms
2
Bypass
1
Standby
Vout2
8
Vs+
Bias
GND
Now, on fig. 13, we can see the PSRR (input
grounded) vs frequency curves. At 217Hz we have
a PSRR value of -36dB.
In reality we want a value about -70dB. So, we
need a gain of 34dB !
Now, on fig. 12 we can see the effect of Cb on the
PSRR (input grounded) vs. frequency. With
Cb=100µF, we can reach the -70dB value.
RL
Av=-1
+
Cb
TS4871
7
■ Principle of operation
The process to obtain the final curve (Cb=100µF,
Cin=100nF, Rin=Rfeed=22kΩ) is a simple transfer
point by point on each frequency of the curve on
fig. 13 to the curve on fig. 12. The measurement
result is shown on the next figure.
Fig. 85 : PSRR changes with Cb
PSRR (dB)
-40
Cin=100nF
Cb=1µF
Cin=100nF
Cb=100µF
■High/low cut-off frequencies
-70
10
100
R ms ( V r i p pl e )
--------------------------------------------Rms ( Vs + - Vs - )
Remark : The measure of the Rms voltage is not a
Rms selective measure but a full range (2 Hz to
125 kHz) Rms measure. It means that we
measure the effective Rms signal + the noise.
-50
-60
The PSRR value for each frequency is:
PSRR ( d B ) = 20 x Log 10
Vcc = 5, 3.3 & 2.6V
Rfeed = 22k, Rin = 22k
Rg = 100Ω, RL = 8Ω
Tamb = 25°C
-30
• We fixed the DC voltage supply (Vcc), the AC
sinusoidal ripple voltage (Vripple) and no supply
capacitor Cs is used
1000
10000
100000
For their calculation, please check this "Frequency
Response Gain vs Cin, & Cfeed" graph:
Frequency (Hz)
10
5
What is the PSRR ?
Gain (dB)
0
The PSRR is the Power Supply Rejection Ratio.
It's a kind of SVR in a determined frequency range.
The PSRR of a device, is the ratio between a
power supply disturbance and the result on the
output. We can say that the PSRR is the ability of
a device to minimize the impact of power supply
disturbances to the output.
Cfeed = 330pF
Cfeed = 680pF
-5
-10
-15
-20
-25
10
Cin = 470nF
Cfeed = 2.2nF
Cin = 22nF
Cin = 82nF
Rin = Rfeed = 22kΩ
Tamb = 25°C
100
1000
Frequency (Hz)
10000
How do we measure the PSRR ?
25/28
TS4871
PACKAGE MECHANICAL DATA
SO-8 MECHANICAL DATA
DIM.
mm.
MIN.
TYP
inch
MAX.
MIN.
TYP.
MAX.
A
1.35
1.75
0.053
0.069
A1
0.10
0.25
0.04
0.010
A2
1.10
1.65
0.043
0.065
B
0.33
0.51
0.013
0.020
C
0.19
0.25
0.007
0.010
D
4.80
5.00
0.189
0.197
E
3.80
4.00
0.150
0.157
e
1.27
0.050
H
5.80
6.20
0.228
0.244
h
0.25
0.50
0.010
0.020
L
0.40
1.27
0.016
0.050
k
ddd
8˚ (max.)
0.1
0.04
0016023/C
26/28
TS4871
PACKAGE MECHANICAL DATA
27/28
TS4871
PACKAGE MECHANICAL DATA
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consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from
its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications
mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information
previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or
systems without express written approval of STMicroelectronics.
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28/28
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