650 kHz/1.3 MHz, 4 A, Step-Up, PWM, DC-to-DC Switching Converter ADP1614 Data Sheet

650 kHz/1.3 MHz, 4 A, Step-Up, PWM, DC-to-DC Switching Converter ADP1614 Data Sheet

ADP1614

APPLICATIONS INFORMATION

ADIsimPOWER DESIGN TOOL

The ADP1614 is supported by the ADIsimPower ™ design toolset.

ADIsimPower is a collection of tools that produce complete power designs that are optimized for a specific design goal. The tools enable the user to generate a full schematic and bill of materials and to calculate performance in minutes. ADIsimPower can optimize designs for cost, area, efficiency, and parts count while taking into consideration the operating conditions and limitations of the IC and the external components. For more information about the ADIsimPower design tools, visit www.analog.com/ADIsimPower . The toolset is available from this website, and users can request an unpopulated board.

SETTING THE OUTPUT VOLTAGE

The ADP1614 features an adjustable output voltage range of V

IN to 20 V. The output voltage is set by the resistor voltage divider,

R1 and R2 (see Figure 32), from the output voltage (V

OUT

) to the

1.245 V feedback input at FB. Use the following equation to determine the output voltage:

V

OUT

= 1.245 × (1 + R1/R2)

Choose R1 based on the following equation:

(1)

R1

=

R2

×

V

OUT

1

1 .

245

.

245

(2)

INDUCTOR SELECTION

The inductor is an essential part of the step-up switching converter. It stores energy during the on time of the power switch and transfers that energy to the output through the output rectifier during the off time. To balance the trade-offs between small inductor current ripple and efficiency, inductance values in the range of 4.7 µH to 22 µH are recommended. In general, lower inductance values have higher saturation current and lower series resistance for a given physical size. However, lower inductance values result in higher peak current, which can lead to reduced efficiency and greater input and/or output ripple and noise. A peak-to-peak inductor ripple current close to 30% of the maximum dc input current typically yields an optimal compromise.

For determining the inductor ripple current in continuous operation, the input (V

IN

) and output (V

OUT

) voltages determine the switch duty cycle (D) as follows:

D

=

V

OUT

V

OUT

V

IN

(3)

Data Sheet

The duty cycle and switching frequency (f

SW

) can be used to determine the on time:

t

ON

=

D f

SW

(4)

The inductor ripple current (∆I

L

) in steady state is calculated by

I

L

=

V

IN

×

L t

ON

(5)

Solve for the inductance value (L) as follows:

L

=

V

IN

×

I

L t

ON

(6)

Ensure that the peak inductor current (the maximum input current plus half the inductor ripple current) is below the rated saturation current of the inductor. Likewise, make sure that the maximum rated rms current of the inductor is greater than the maximum dc input current to the regulator.

For continuous current-mode (CCM) duty cycles greater than

50% that occur with input voltages less than one-half the output voltage, slope compensation is required to maintain stability of the current-mode regulator. For stable current-mode operation, ensure that the selected inductance is equal to or greater than the minimum calculated inductance, L

MIN

, for the application parameters in the following equation:

L

>

L

MIN

=

(

V

OUT

8

×

2

×

f

SW

V

IN

) (7)

Inductors smaller than the 4.7 µH to 22 µH recommended range can be used as long as Equation 7 is satisfied for the given application. For input/output combinations that approach the

90% maximum duty cycle, doubling the inductor is recommended

to ensure stable operation. Table 5 suggests a series of inductors

for use with the ADP1614 .

Table 5. Suggested Inductors

Manufacturer Part Series

Coilcraft

TOKO Inc.

XAL40xx, XAL50xx, XAL6060, DO3316P

FDV06xx, DG6045C, FDSD0630, DEM8045C,

FDVE1040

Würth Elektronik WE-HCI, WE-TPC, WE-PD, WE-PD2, WE -PDF

Vishay Dale IHLP-2020, IHLP-2525, IHLP-3232, IHLP-4040

TDK Components SPM6530, VLP8040, VLF10040, VLF10045

Taiyo Yuden NRS8030, NRS8040

Rev. B | Page 14 of 18

Data Sheet

CHOOSING THE INPUT AND OUTPUT CAPACITORS

The ADP1614 requires input and output bypass capacitors to supply transient currents while maintaining constant input and output voltages. Use low equivalent series resistance (ESR) capacitors of 10 µF or greater to prevent noise at the ADP1614 input. Place the capacitor between VIN and GND, as close as possible to the ADP1614 . Ceramic capacitors are preferable because of their low ESR characteristics. Alternatively, use a high value, medium ESR capacitor in parallel with a 0.1 µF low

ESR capacitor, placed as close as possible to the ADP1614 .

The output capacitor maintains the output voltage and supplies current to the load while the ADP1614 switch is on. The value and characteristics of the output capacitor greatly affect the output voltage ripple and stability of the regulator. A low ESR ceramic dielectric capacitor is preferable. The output voltage ripple (∆V

OUT

) is calculated as follows:

V

OUT

=

Q

C

C

OUT

=

I

OUT

×

t

ON

C

OUT

C

OUT

f

I

OUT

SW

×

×

V

(

V

OUT

OUT

×

V

IN

V

OUT

)

(8)

I

where:

Q

C

is the charge removed from the capacitor.

C

OUT

is the output capacitance.

OUT

is the output load current.

t

ON

is the on time of the switch.

The on time of the switch is determined as follows:

t

ON

=

D f

SW

(9)

The input (V

IN

) and output (V

OUT

) voltages determine the switch duty cycle (D) as follows:

D

=

V

OUT

V

OUT

V

IN

(10)

Choose the output capacitor based on the following equation:

(11)

Multilayer ceramic capacitors are recommended for this application.

DIODE SELECTION

The output rectifier conducts the inductor current to the output capacitor and load while the switch is off. For high efficiency, minimize the forward voltage drop of the diode. For this reason, using Schottky rectifiers is recommended. However, for high voltage, high temperature applications, where the Schottky rectifier reverse leakage current becomes significant and can degrade efficiency, use an ultrafast junction diode.

Many diode manufacturers derate the current capability of the diode as a function of the duty cycle. Verify that the output

ADP1614

diode is rated to handle the average output load current with the minimum duty cycle. The minimum duty cycle in CCM of the ADP1614 is

D

MIN

=

V

OUT

V

IN

(

MAX

)

V

OUT

(12) where V

IN(MAX)

is the maximum input voltage.

The following are suggested Schottky diode manufacturers:

• ON Semiconductor

• Diodes, Inc.

• Toshiba

• ROHM Semiconductor

LOOP COMPENSATION

The ADP1614 uses external components to compensate the regulator loop, allowing optimization of the loop dynamics for a given application.

The step-up converter produces an undesirable right-half plane zero in the regulation feedback loop. This requires compensating the regulator such that the crossover frequency occurs well below the frequency of the right-half plane zero. The right-half plane zero is determined by the following equation:

F

Z

(

RHP

)

=



V

IN

V

OUT



2

×

R

2

π

LOAD

×

L

where:

F

Z

(RHP) is the right-half plane zero.

R

LOAD

is the equivalent load resistance or the output voltage divided by the load current.

To stabilize the regulator, ensure that the regulator crossover frequency is less than or equal to one-fifth of the right-half plane zero.

The regulator loop gain is

(13)

A

VL

=

V

FB

V

OUT

×

V

IN

V

OUT

×

G

MEA

×

R

OUT

Z

COMP

×

G

CS

×

Z

OUT

where:

A

VL

is the loop gain.

V

FB

is the feedback regulation voltage, 1.245 V.

V

OUT

is the regulated output voltage.

V

IN

is the input voltage.

G

MEA

is the error amplifier transconductance gain.

R

OUT

= 67 MΩ.

Z

COMP

is the impedance of the series RC network from COMP to GND.

G

CS

is the current sense transconductance gain (the inductor current divided by the voltage at COMP), which is internally set by the ADP1614 .

Z

OUT

is the impedance of the load in parallel with the output capacitor.

(14)

Rev. B | Page 15 of 18

ADP1614

To determine the crossover frequency, it is important to note that at the crossover frequency the compensation impedance (Z

COMP

) is dominated by a resistor, and the output impedance (Z

OUT

) is dominated by the impedance of an output capacitor. Therefore, when solving for the crossover frequency, the equation (by definition of the crossover frequency) is simplified to

A

VL

2

π ×

=

f

C

V

FB

V

OUT

1

×

×

C

OUT

V

IN

V

OUT

=

1

×

G

MEA

×

R

COMP

×

G

CS

×

(15) where:

R

COMP

is the compensation resistor.

f

C

is the crossover frequency.

Solve for R

COMP

as follows:

R

COMP

=

2

π ×

V

FB f

C

×

×

V

IN

C

OUT

×

G

×

(

MEA

V

OUT

×

G

)

CS

2 where:

V

FB

= 1.245 V.

G

MEA

G

CS

= 150 µA/V.

= 7 A/V.

Therefore,

(16)

R

COMP

=

4806

×

f

C

×

C

OUT

×

(

V

OUT

)

2

V

IN

(17)

After the compensation resistor is known, set the zero formed by the compensation capacitor and resistor to one-fourth of the crossover frequency, or

C

COMP

=

π ×

f

C

2

×

R

COMP

(18) where C

COMP

is the compensation capacitor.

ERROR

AMPLIFIER

FB 2 g m

COMP

1

V

BG

R

COMP

C2

C

COMP

Figure 34. Compensation Components

Data Sheet

Capacitor C2 is chosen to cancel the zero introduced by the ESR of the output capacitor.

Solve for C2 as follows:

C2

=

ESR

×

C

OUT

R

COMP

(19)

If a low ESR, ceramic output capacitor is used for C

OUT

, C2 is optional. For optimal transient performance, R

COMP

and C

COMP might need to be adjusted by observing the load transient response of the ADP1614 . For most applications, the compensation resistor should be within the range of 1 kΩ to 100 kΩ, and the compensation capacitor should be within the range of 100 pF to

10 nF.

SOFT START CAPACITOR

Upon startup (EN ≥ 1.6 V) or fault recovery, the voltage at SS ramps up slowly by charging the soft start capacitor (C

SS

) with an internal 5.5 µA current source (I

SS

). As the soft start capacitor charges, it limits the peak current allowed by the part to prevent excessive overshoot at startup. Use the following equation to determine the necessary value of the soft start capacitor (C

SS

) for a specific overshoot and start-up time when the part is at the current limit with maximum load:

C

SS

=

I

SS

t

V

SS

where:

I

SS

= 5.5 μA (typical).

Δt is the start-up time at the current limit.

V

SS

= 1.23 V (typical).

(20)

If the applied load does not place the part at the current limit, the value of C

SS

can be reduced. A 68 nF soft start capacitor results in negligible input current overshoot at startup and, therefore, is suitable for most applications. If an unusually large output capacitor is used, a longer soft start period is required to prevent input inrush current.

However, if fast startup is required, the soft start capacitor can be reduced or removed, which allows the ADP1614 to start quickly but with greater peak switch current.

Rev. B | Page 16 of 18

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