STMicroelectronics L6574, L6561 Electronic Ballast APPLICATION NOTE
விளம்பரம்
விளம்பரம்
AN993
APPLICATION NOTE
ELECTRONIC BALLAST WITH PFC
USING L6574 AND L6561
by F. Sandrini, U. Moriconi and I. Dal Santo
The advent of dedicate IC for lamp ballast applications is replacing the old solutions based on bipolar transistor driven by a saturable pulse transformer.
The L6574 is an innovative high performance ballast driver, designed in 600V BCD OFF-LINE technology, which ensures all the features needed to drive and control properly a fluorescent bulb. It is provided with a built-in VCO and an OP-AMP, useful to implement a closed loop control of the lamp current, therefore of the lamp power.
INTRODUCTION
Half bridge converter for electronic lamp ballast
Voltage fed series resonant half bridge inverters are currently used for fluorescent lamps (fig.1). This topology allows to easily operate in Zero Voltage Switching (ZVS) resonant mode, reducing the transistor switching losses and the electromagnetic interference. Moreover, by varying the switching frequency it is possible to modulate the current in the lamp, therefore the output power.
To design a cost effective, compact and smart electronic lamp ballast it could be used a dedicated IC able to drive directly the power MOSFETs of the half bridge. Such controllers require a high voltage capability for the high side floating transistor driver.
Figure 1. Half Bridge topology
HV
DRIVER
L
RES
C
RES
D98IN819
Lamp requirements
To provide long life and to insure an efficient ignition of the lamp the cathodes must be preheated. In fact the preheating of the filaments allows an easy strike of the lamp reducing ignition voltage. During preheating time the lamp is characterised by a high impedance and the current flows only in the filaments.
The resistance value of the filaments are strictly dependent on the lamp model. Typically these filaments present a initial low value (a few Ohms) that will increase by 5 times during the preheating.
After the preheating time the lamp must be ignited, by increasing the voltage across it. The ignition voltage value also depends on the lamp type, and it increases with the aging. For a typical TL 58W it is not much less than 1000V. Using a simple inverter, with a constant switching frequency, external circuitry must be used (e.g. PTC or discrete timer). Instead with the ST L6574 smart controller both the preheating and the ignition functions are achieved by using simple resistors and a capacitor, which set all the start-up procedure.
July 1999
1/15
AN993 APPLICATION NOTE
L6574 Ballast Driver
The L6574, whose internal block diagram is shown in fig.2, is an IC intended to drive two power MOS or
IGBT in half bridge topology, ensuring all the features needed to drive and control properly a fluorescent bulb. The device is available in DIP16 and SO16N packages.
The most significant features of the L6574 concern the following points:
• high voltage rail up to 600V;
• dV/dt immunity
±
50 V/ns in full temperature range;
• driver current capability (250 mA source and 450 mA sink);
• switching times 80/40 ns rise fall with 1 nF load;
•
CMOS shut down input;
• under voltage lock out;
• preheat and frequency shifting timing;
• sense OP AMP for closed loop control or protection features;
• high accuracy current controlled oscillator;
• integrated bootstrap diode;
• clamping on V
S
;
•
SO16, DIP16 package.
Figure 2. Internal Block diagram of the L6574
H.V.
R
IGN
OPOUT
5
OPIN-
OPIN+
6
7
OP AMP
+
-
Imin
V
REF
4
Imax
V
REF
2
R
PRE
3
VCO
Cf
V
S
12
CONTROL
LOGIC
UV
DETECTION
DEAD
TIME
Ifs Ipre
+
-
+
-
Vthpre
1
C
PRE
BOOTSTRAP
DRIVER
DRIVING
LOGIC
HVG
DRIVER
LEVEL
SHIFTER
LVG DRIVER
+
-
V
THE
+
-
V
THE
16
V
BOOT
15 HVG
14 OUT
V
S
11 LVG
10 GND
8
EN1
9
EN2
D97IN493B
C
BOOT
LOAD
2/15
AN993 APPLICATION NOTE
Device Pins Description
N.
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
Name
C
PRE
R
PRE
C
F
R
IGN
OP out
OP in-
OP in+
EN1
EN2
GND
LVG
V
S
N.C.
OUT
HVG
V
BOOT
Function
Preheat Timing Capacitor. The capacitor C time, according to the relations: t
K
FS
PRE
= K
PRE
PRE
⋅
C sets the preheating and the frequency shift
PRE and t
= 0.15s/
µ
F). This feature is obtained by charging C
SH
PRE
= K
FS
⋅
C
PRE
(typ. K
PRE
= 1.5s/
µ
F, with two different currents. During t
PRE this current is independent of the external components, so C
PRE
(preheat timing comparator threshold). During t
SH is charged up to 3.5 V the current depends on R
PRE value (i.e. on the difference between f
PRE and f
IGN the voltage at pin 1 is 5V (see fig.5).
). In this way t
SH is always set at 0.1t
PRE
. In steady state
Maximum Oscillation Frequency Setting. The resistance connected between this pin and ground sets the f
PRE value, fixing the difference between f
PRE voltage at this pin is fixed at V
REF
=2V.
and f
IGN
(f
PRE
> f
IGN
). The
Oscillator Frequency Setting. The capacitor C
F
, along with to R
PRE f
ING
. In normal operation this pin shows a triangular wave.
and R
IGN
, sets f
PRE and
Minimum Oscillation Frequency Setting. The resistance connected between this pin and ground sets the f
IGN value. The voltage at this pin is fixed at V
REF
=2V.
Out of the operational amplifier. To implement a feedback control loop this pin can be connected to the R
IGN pin by means an appropriate circuitry.
Inverting Input of the operational amplifier.
Non Inverting Input of the operational amplifier.
Enable 1. This pin (active high), forces the device in a latched shutdown state (like in the under voltage conditions). There are two ways to resume normal operation. The first is to reduce the supply voltage below the undervoltage threshold and then increase it again until the valid supply is recognised. The second is activating EN2 input. The enable 1 is especially designed for strong fault (e.g. in case of lamp disconnection).
Enable 2. EN2 input (active high) restarts the start-up procedure (preheating and ignition sequence). This features is useful if the lamp does not turn-on after the first ignition sequence.
Ground.
Low Side Driver Output. This pin must be connected to the low side power MOSFET gate of the half bridge. A resistor connected between this pin and the power MOS gate can be used to reduce the peak current.
Supply Voltage. This pin, connected to the supply filter capacitor, is internally clamped (15.6V
typical).
Non Connected. This pin set a distance between the pins related to the HV and those related to the LV side.
High Side Driver Floating Reference. This pin must be connected close to the source of the high side power MOS or IGBT.
High Side Driver Output. This pin must be connected to the high side power MOSFET gate of the half bridge. A resistor connected between this pin and the power MOS gate can be used to reduce the peak current.
Bootstrapped Supply Voltage. Between this pin and V
S must be connected the bootstrap capacitor. A patented integrated circuitry replaces the external bootstrap diode, by means of a high voltage DMOS, synchronously driven with the low side power MOSFET.
3/15
AN993 APPLICATION NOTE
DEVICE BLOCK DESCRIPTION
The preheating control section and the bootstrap section are tightly connected to application design.
Here below there are some details on their working and how they have to be used.
Preheating and ignition section
L6574 turn on sequence is divided into three phases: preheating, ignition and normal operation (fig.3).
Preheating phase is characterised by the highest oscillation frequency (f max
) for a period T pre
. During ignition phase the frequency shifts from f max to f min
(that is the normal operation frequency) in a period T sh
.
Figure 3.
preheating phase freq fmax ignition phase normal operation fmin
Tpre
Tsh time
All the parameters specified above are set choosing properly few external components.
T pre and T sh are set by means of the capacitor C pre that is connected at pin 1.
During the preheating time, T pre
, the capacitor C pre is charged by means of a constant current I pre internally generated, which doesn’t depend on the external components. The voltage across C pre increases linearly up to the ”preheating threshold”, in which the preheating phase finishes.
T pre
=
C pre
V th
I pre
, where V th
= 3.5V and I pre
= 2.3
µ
A
That is to say:
T pre
= 1.5s/
µ
F
⋅
C pre
Figure 5. Timing Oscillator block
Figure 4. Timing block
Ifs Ipre
TIMING Imax Imin gm
Iosc
OSC
Cpre
Cpre DISCHARGE
4/15
AN993 APPLICATION NOTE
After the preheating time the capacitor C pre is first quickly discharged, and then recharged by the current
I fs
, generating a second voltage ramp which fed a transconductance amplifier, as shown in fig.5 (the switch is closed). Thus this voltage signal is converted in a growing current which is subtracted to I max
, to produce the frequency shifting from f max to f min
. The current which drives the oscillator to set the frequency during the shifting is equal to:
I osc
=
I min
+
(
I max
− g m
V
Cpre
( t
−
T pre
)) =
I min
+
I max
− g m
I fs
C pre
( t
−
T pre
)
[a]
Where:
I min
=
V
REF
R ign
, I max
=
V
REF
R pre
, V
REF
=
2V
R ign and R pre are the resistors connected to pin 4 and pin 2.
At the end of preheating time (t = T pre
) L6574 oscillates at f max
, set by:
I osc
(
T pre
) =
I osc
(
0
) =
I min
=
I max
=
V
REF
1
R ign
+
1
R pre
That means that the preheating frequency depends on both R pre and R ign
.
At the end of the frequency shifting (t = T pre
+ T sh
), the second term of eq.[a] decreases to zero and the switching frequency is set only by I min
(i.e. R ign
):
I o sc
(
T sh
) =
I min
=
V
REF
R ign
Since the second term of eq. [a] is equal to zero, we have:
I max
− g m
I fs
C p re
T sh
=
0
→
T sh
=
C pre
I max g m
I fs
[b]
Note that there is not a fixed threshold of the voltage across C pre in which the ignition phase finishes (i.e.
the end of the frequency shifting): T sh depends on C pre
, I max
, g m
, and I fs
(eq. [b]). This fact can be seen also in fig.6. Making T sh independent of I max
, the I fs current has been designed to be a fraction of I max
, so:
I fs
=
I max
K
→
T sh
=
C pre
I max g m
I max
K
→
T sh
=
C pre g m
K
→
T sh
= k fs
C pre
In this way the frequency shift time depends only by the capacitor C pre
. The typical value of the k fs constant (Frequency Shift Timing Constant) is 0.15
s/
µ
F, that is: k fs
= k pre
/10.
Figure 6. C
pre
voltage and frequency shifting
V(Cpre)
So choosing Cpre both T pre and T sh are set.
The frequencies f min and f max depend on the resistors R pre and R ign
, but also on the capacitor C f
(Oscillator frequency setting -> capacitor at pin 3). f min is set choosing C f and R ign
, then with R pre the
∆ f = f max
- f min is set. Simplified equations can be used: freq preheating ignition time f min
=
1.41
R ign
⋅
C f f max
=
1.41
⋅
(
R pre
R pre
⋅
R ign
+
R ign
)
⋅
C f time
5/15
AN993 APPLICATION NOTE
These equation fit well with measured values, especiallyin the frequencies range 30 to 100kHz: in fig. 7 there is a comparison between f min measured and computed data (@C f
= 470pF).
Figure 7. Operating Frequency @ C
f
= 470pF
120 measured values computed values
100
To summarise:
parameter
T pre
T sh
∆ f = f max
- f min f min
set choosing
C pre
C pre
R pre
R ign
& C f
80
60
40
If R pre is not connected at pin 2, there is no
∆ f , so f max and f min have the same value. Moreover there is no I max
, that is to say there is no I fs
. In this case
C pre is charged only once by I pre
= 2.3
µ
A up to 3.5V
(fig 7: there is only the first voltage raising) . When
20
40 60
R (k W )
80 100 120
C pre is discharged there is no current I fs to charge it once more. If R pre has been connected when preheating and ignition phases are ended, pin 1(C pre
) is at 4.8-5V with a current capability of few
µ
A (1/6
I
Rpre
). If R pre is not connected, C pre is at GND level in normal operation phase.
Bootstrap section
A bootstrap circuitry is needed to supply the high voltage section. This function is normally accomplished by a high voltage fast recovery diode (fig. 8a). In the L6574 a patented integrated structure replaces the external diode. It is realised by a high voltage DMOS, driven synchronously with the low side driver
(LVG), with in series a diode, as shown in fig. 8b
An internal charge pump (fig. b) provides the DMOS driving voltage .
The diode connected in series to the DMOS has been added to avoid undesirable turn on of it.
C boot selection and charging:
To choose the proper C boot value the external MOS can be seen as an equivalent capacitor. This capacitor C ext is related to the MOS total gate charge :
C ext
=
Q gate
V gate
The ratio between the capacitors C ext and C boot is proportional to the cyclical voltage loss .
It has to be:
C boot
>>>C ext e.g.: if Q gate is 30nC and V gate is 10V, C ext is 3nF. With C boot
= 100nF the drop would be 300mV.
If HVG has to be supplied for a long time, the C boot selection has to take into account also the leakage losses.
The internal bootstrap driver gives great advantages: the external fast recovery diode can be avoided (it usually has great leakage current). This structure can work only if V out is close to GND (or lower) and in the meanwhile the LVG is on. The charging time (T charge
) of the C boot is the time in which both conditions are fulfilled and it has to be long enough to charge the capacitor.
The bootstrap driver introduces a voltage drop due to the DMOS R dson
(typical value @25
° is 150
Ohm). At low frequency this drop can be neglected. Anyway increasing the frequency it must be taken in to account.
The following equation is useful to compute the drop on the bootstrap DMOS:
6/15
AN993 APPLICATION NOTE
V drop
=
I charge
R dson
→
V drop
=
Q gate
T charge
R dson where Q gate is the gate charge of the external power MOS, R dson is the on resistance of the bootstrap
DMOS, and T charge is the charging time of the bootstrap capacitor.
For example: using a power MOS with a total gate charge of 30nC the drop on the bootstrap DMOS is about 1V, if the T charge is 5
µ s. In fact:
V drop
=
30nC
5
µ s
⋅
150
Ω
~0.9V
V drop has to be taken into account when the voltage drop on C boot is calculated: if this drop is too high, or there is not a sufficient charging time, an external diode can be used.
Figure 8. Bootstrap driver
D
BOOT
V
S
HVG
V
BOOT
H.V.
V
OUT
C
BOOT
TO LOAD
V
S
HVG
V
BOOT
H.V.
V
OUT
C
BOOT
TO LOAD a
LVG b
LVG
EX_D99IN1056
DEMO APPLICATION DESCRIPTION
The design has been developed to drive a TL fluorescent lamp up to 58W. It is composed of two sections: the PFC, using the L6561 controller, and the ballast, based on the L6574 (see fig.9 and fig.10).
The application is provided with a current feedback, that allows power control (and in case the dimming function) by varying the switching frequency during the normal lamp burning. The application is also provided with a safety circuitry, that acts in case of open load or faulty ignition of the lamp.
7/15
AN993 APPLICATION NOTE
Figure 9. Demo Application circuit
STP6NB50
Q2STP4NB50/
STP6NB50
Q3STP4NB50/
STP6NB50
Q1
8/15
Figure 10. PCB and components layout.
37mm
AN993 APPLICATION NOTE
220mm
C
O
M
P
O
N
E
N
T
L
A
Y
O
U
T
L
A
Y
E
R
C
O
M
P
L
A
Y
E
R
B
A
C
K
9/15
AN993 APPLICATION NOTE
Power factor section
Even if the PFC stage is not strictly necessary for electronic ballast application, in this design it has been introduced for the following reasons.
The front-end stage of conventional off-line converters, typically made up of a full wave rectifier bridge with a capacitor filter, gets an unregulated DC bus from the AC mains. Therefore the instantaneous line voltage is below the voltage on the capacitor most of the time, thus the rectifiers conduct only for a small portion of each line half-cycle. The current drawn from the mains is then a series of narrow pulses whose amplitude is 5-10 times higher than the resulting DC value.
Lots of drawbacks result from that: much higher peak and RMS current drawn from the line, distortion of the AC line voltage, overcurrents in the neutral line of the three-phase systems and, after all, a poor utilisation of the power system’s energy capability.
This can be measured in terms of either Total Harmonic Distortion (THD), as norms provides for, or
Power Factor (PF), intended as the ratio between the real power (the one transferred to the output) and the apparent power (RMS line voltage times RMS line current) drawn from the mains, which is more immediate. A traditional input stage with capacitive filter has a low PF (0.5-0.7) and a high THD (> 100%).
The new European norms and the International standard requirements have spurred the design of high power factor ballasts and they are starting to impose a limit on the input current harmonic content. For these reasons power factor corrector (PFC) is now diffusing in consumer and industrial lighting. With a high power factor switching preregulator, interposed between the input rectifier bridge and the bulk filter capacitor, the power factor will be improved (up to 0.99). The current capability is increased, the bulk capacitor peak current and the harmonic disturbances are reduced.
The L6561 is an IC intended to control PFC preregulators by using the transition mode technique and is optimised for lamp ballast applications.
The operation can be summarised in the following description (for more information, see AN966). The
AC mains voltage, that can range from 85V to 265V, is rectified by a diodes bridge and delivered to the boost converter. The input capacitor has been split in two parts (C
1 and C
2
) to increase the performance in terms of harmonic distortion (THD). In fact, due to the wide range supply and the possibility to change the output power, the minimisation of the capacitor connected after the bridge, allows to reduce the
THD. The boost converter consists of a boost inductor (T
1
), a controlled power switch (Q
1
), a catch diode (D
1
), an output capacitor (C
7
) and, obviously, a control circuitry (see fig.3).
The PFC section has been designed to supply a 400V DC and a power of 60W.
Ballast Section
The regulated voltage is delivered to the ballast section. The ballast is based on the high performances
L6574, which is an OFF-LINE half bridge driver designed in 600V BCD technology. It adds to the full integrated half bridge driver topology a built-in voltage controlled oscillator (VCO), a preheating start-up procedure and an operational amplifier dedicated to the feedback loop. To avoid cross conduction of the power MOSFETs or IGBTs, the internal logic ensures a minimum dead time.
The load consists of a series resonant circuit (L
1
-C
19
) with the lamp connected across the capacitor
(C
19
). This topology allows to operate in Zero Voltage Switching, to reduce the transistor switching losses and the electromagnetic interference generated by the output wiring of the lamp.
The blocking capacitor (C
17
//C
18
) allows to obtain a zero average lamp current. In steady state the voltage across these capacitors is as high as half the high voltage bus, that is about 200V.
Preheating and ignition sequence
The turn-on sequence can be divided in three phases: preheating, ignition and normal lamp burning.
The preheating of the lamp filaments is achieved by a high switching frequency f
PRE
, about 60 kHz, set by R
PRE
= R
13
+ R
14
+ R
15 and C
F
= C
13
, to ensure that a current flows in the filaments without lamp ignition. In fact the initial voltage applied across the lamp is below the strike potential. The duration of the preheating period t
PRE is set by the capacitor C
PRE
= C
14
. The choice of this time is strictly dependent on the lamp type. In the application t
PRE has been set at 1.5 sec.
The ignition sequence begins after t
PRE
. The switching frequency decreases towards the resonance point (L
1
-C
19
), increasing the voltage across the lamp, and causing the ignition. The time interval in which the frequency shifts, t
SH
, amounts to t
SH
= t
PRE
/10 = 150ms. At the end of t
SH the frequency reaches 31 kHz (R
19
-C
13
), and then the current feedback loop is activated.
10/15
AN993 APPLICATION NOTE
Current feedback loop
The current control is achieved by varying the switching frequency of the VCO. Since controlling the average current in the lamp means controlling the output power, it is quite easy to perform dimming function. The OP-AMP compares the low-pass filtered half-bridge current, shunted by R
25
, with a reference, achieved by a partition of the voltage at pin 2 (V
PIN2
= 2 V). This set-point could be changed by the trimmer R
14
, to perform the dimming function. The OP-AMP output is connected to R
ING pin by D
3 and R
18
.
The diode D
3 is necessary to avoid that the switching frequency decreases below the value set by R
19
.
At start-up the voltage across R
25
(fig.11) remains low until the lamp ignition. So the inverting input of
OP-AMP (pin 6) stays low too, while the non inverting input (pin 7) is set at a constant voltage (setpoint) by the divider R
13
, R
14 and R
15
.
Therefore the OP-AMP output (pin 5) remains high (5V) until the lamp ignition, and D
3 is off. In this condition the L6574 oscillates at f
PRE
.
As the lamp strikes on (after t
PRE and t
SH
), the average voltage across R
25 increases and the feedback is able to regulate the lamp current.
Figure 11. Current feedback loop
Figure 12. Cpre waveform (Ch1) and OP AMP output (Ch2)
R25
0.68
7
Q3
STP4NB50/
STP6NB50
R33 9.1K
C9 8.2nF
D3
1N4148
R18 100K R19 100K
6 5
-
+
RIGN
4
D98IN818A
Start-up and supply
The start-up procedure is very important in an application that contains two different sections.
The ballast section starts before the PFC, avoiding any extra-voltage at the PFC section output, and so the L6561 dynamic OVP activation (see AN966). This behaviour is guaranteed under all conditions because the V
S turn-on threshold of L6574 is lower than the L6561 one.
At start-up the L6574 is powered by the resistor (R
3
+ R
4
). This resistor must be chosen so as to ensure the ”before start-up current” of both the L6561 and L6574.
When the ballast section is running, the charge pump (C
11
, R
16
, D
2 and D
Z1
) allows to supply both the devices. The resistor R
16 allows to reduce the peak current.
Safety circuitry
In normal operation the inductive load ensures a zero voltage switching mode, but if the lamp is disconnected the switching losses in the power MOSFETs will increase considerably. To prevent this occurrence a safety circuitry has been designed. When the lamp is connected the EN1 input (pin 8) of the
L6574 is held close to ground by the series of R
27
, the lamp filament and R
25
. If the lamp is not present
EN1 is pulled up to V
S by R
26
, forcing the L6574 in a latched shutdown state. To resume normal operation it is necessary to turn off the ballast and then turn it on again.
A second alarm has been designed to protect the application against the extra voltages which would arise if the lamp did not strike after the ignition sequence, because of an old lamp. A partition of this ex-
11/15
AN993 APPLICATION NOTE
tra voltage is rectified and delivered to the EN2 input (pin 9) of the L6574, restarting the start-up procedure (preheat and ignition sequence).
Figure 13. Open load safety circuit Figure 14. Extra voltage safety circuit
+V
S
LAMP
R25
0.68
C19
8.2nF
1500V
R27
6.8K
R26
390K
EN1 (PIN8)
C16
1uF
R32
20K
D98IN816A
EN2 (PIN9)
R24
6.8K
L1 2.1mH
C17
100nF
250V
D4 1N4148
R28
750K
R29
750K
C15
330nF
R30
3.9K
C18
100nF
250V
LAMP
R25
0.68
D98IN817
C19
8.2nF
1500V
DESIGN HINTS
Inductance and capacitors evaluation
To design an application with L6574 a preliminary evaluation of the components can be done only fixing the lamp type and its electrical characteristics.
We can summarise this evaluation process in few marks, but this is an ”iterative” process, because we have to do some assumptions that have to be checked at the end of the process. A simplified schematic can well represent the load:
Figure 15. Lamp Simplified Schematic
L L r lamp
C rr
C r
Cb Cb
In fig. 15 r is the lamp filament resistance, rr is the operating lamp equivalent resistance ( when the lamp is off or during preheating rr is an open circuit).
First of all we have to evaluate a proper inductance value.
L has to give the right current value to the lamp when it is already ignited and is working (”choke” induc-
12/15
AN993 APPLICATION NOTE
tance). So it depends on the current required by the lamp , that is to say on the lamp operating wattage and voltage, on the operating frequency f min
, but also on the voltage across L and the lamp (V b
). The greater V b variation, the greater inductance in order to give a constant current to the lamp. V b variations are due to the High Voltage Bus variations and to the ripple on half battery capacitor (C b
). The ripple depends on C b size, and we have to do a proper hypothesis on it to estimate V b variation. Taking into account all these data and hypothesis, we can do some simplifying assumption: during operating condition the most of the current flows into the lamp, not into C, and all the power delivered to the system is delivered to the lamp. V lamp is the operating voltage across the lamp and P lamp is the operating lamp wattage, so a good approximation to conduct the choke inductance is:
L
=
I
L
⋅
V
L
2
⋅ π ⋅ f
→
V b
P lamp
⋅
(
V b
−
V l amp
2
⋅ π ⋅ f min
)
The second step is the evaluation of the capacitance across the lamp (C).
When the lamp is not yet ignited, C has to allow a sufficient current to flow into the lamp filament in order to have a proper preheating. The power to be delivered to the lamp filaments (P fil
), the preheating frequency, the lamp filament resistance r and the maximum voltage to be applied across C without causing lamp ignition are constraints that help to evaluate the capacitor size. Setting the current through the lamp filament and the max. voltage across the capacitor, you have a capacitor range of values.
C
=
I
C
⋅
2
⋅ π ⋅
1 f
⋅
V
C
→
√
r
⋅
1
2
⋅ π
⋅ f pre
⋅
V
Cmax
V
Cmax
< V ignition
Those values of L, C b and C have to be corrected in order to have standard and commercial components values. Using these values and lamp equivalent resistances the transfer functions during preheating and operating condition can be calculate.
The preheating transfer function allows to see if moving towards the resonant frequency (L-C) there is a frequency at which the voltage across the lamp enables its ignition, and this frequency has to be between f max and f min
. The transfer function gain depends also on r, but r changes greatly during preheating (also 3-4 times) and this has to be considered (see fig.16).
Figure 16. Preheating Transfer Function
V lamp
V in
F=F(L,C,r@25C)
F=F(L,C,r@Tpre)
1/2
∗π∗ sqrt(L
∗
C) f pre f min frequency shifting during T sh freq
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AN993 APPLICATION NOTE
The operating transfer function allows to check if we have an operating voltage across the lamp (at f = f min
) that is similar to the one used to evaluate L (fig. 17).
Figure 17. Operating Transfer Function
V lamp
V in freq f = f(L,C,rr) f min
If one of these condition is not verified, the evaluation process has to be restarted changing the initial hypothesis: i.e. changing the frequencies, or the assumption on Cb ripple. If this everything is OK, the values found can be used , and a preliminary stage of design can be concluded: to better set the components and the frequencies values experiments are needed, also to verify the initial assumptions, and in case to reiterate another step of evaluation process with better assumptions.
The following flow chart can help the iteration process:
Figure 18. Iterative Process
DATA: operating lamp elec.characteristic
HVB
HYPOTHESIS: fmin
Cb
L evaluation
DATA: ignition and preheating lamp elec.characteristic
HYPOTHESIS: fpre
C evaluation
Transfer preheating funct.
OK
Transfer operating funct.
OK check on board
OK
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AN993 APPLICATION NOTE
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