STMicroelectronics STEVAL-ILD005V1 Lighting Control Application Note
Below you will find brief information for STEVAL-ILD005V1. This evaluation board implements a cost-effective MOS-based reverse phase cut dimming solution, ideal for analog wall dimmers. It can effectively dim resistive or capacitive lighting loads, including halogen lamps and various dimmable CFL/LED lamps available for domestic or industrial use. The board offers on/off power switching and up/down dimming through a single linear rotary potentiometer.
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AN4717
STEVAL-ILD005V1: Trailing edge phase control rotary wall dimmer based on STF17N62K3 power MOSFET
Anselmo Liberti
Introduction
The STEVAL-ILD005V1 evaluation board implements a simple and cost-effective MOS-based reverse phase cut dimming solution for analog wall dimmers using a pair of STF17N62K3 power MOSFETs controlled by a triple 3-input NOR gate to effectively dim resistive or capacitive lighting loads available for domestic or industrial use (i.e. halogen lamps, electronic low voltage transformers and various dimmable CFL/LED lamps). The AC analog control dimming technique allows on/off power switching and up/down dimming functions through a single linear rotary potentiometer equipped with a mechanical switch which opens the circuit at the minimum setting. Control of the board is achieved with a triple 3input NOR gate (pin-to-pin compatible with ST's HCF4025BE 14-lead dual in-line CMOS gate logic device in a plastic SO-14 micropackage), which acts as a gate driving voltage signal source and performs zero crossing detection and timing through the use of a few other passive components and diodes.
The 15.5 A /620 V SuperMESH3 MOSFET STF17N62K3 in the TO-220 full package used for the lamp current commutations is the most suitable device for the requirements of this trailing edge phase cut dimmer switch application operating at line frequency (50/60 Hz) because its high total input capacitance and sufficiently low on-resistance at the operating currents provides the best trade-off between efficiency and EMI in a particularly simplified dimmer design solution.
The main features of this dimmer are:
operation for 2-wire wall dimmer
trailing-edge control only (compatible with all lamps commonly found on the market)
operation on 110 V rms
or 230 V rms
line voltage (±10 %) and 50 Hz or 60 Hz line frequency
dimmable power range (without heatsinks mounted on power MOSFETs)
40 W to 300 W for 230 V rms
line
15 W to 90 W for 110 V rms
line
maximum operating ambient temperature: 50 °C
power efficiency @ Max power - 230 V
AC
> 95%
power factor @ Max power > 0.98
control and regulation interface with a switched type single linear rotary potentiometer
compliance with EMC standards:
EN55015 (for European market): compliant with 300 W - 230 V
AC
halogen lamp
IEC 61000-4-5: criteria A for 2 kV surge tests
RoHS compliant
Other features include:
flicker-free dimming performance
smooth dimming operation and silent dimming control with no buzzing or audible noise
acceptable levels of EMI noise without the need for LC filters to reduce conducted emission disturbances
low inrush current generation
August 2015 DocID027977 Rev 1 1/48
www.st.com
Operation principle AN4717
This application note describes the operating principle and working conditions of the STEVAL-ILD005V1 board. An electrical evaluation of the dimmer performance is also given in terms of temperatures, turnoff energies and EMC standard testing.
Figure 1: STEVAL-ILD005V1 board: top and side views
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Contents
Contents
1 Operation principle ........................................................................... 5
Functional block diagram .................................................................... 5
Steady-state stage-wise circuit analysis ........................................... 11
2 Operating conditions and selected components ........................ 20
Triple 3-inputs CMOS NOR gate ...................................................... 20
Linear rotary potentiometer .............................................................. 21
3 Board performances ...................................................................... 22
Experimental test and results @ 230VAC ........................................ 22
Electromagnetic compatibility test .................................................... 26
EMC conducted noise compliance according to EN 55015 .............. 26
Surge voltage immunity test ............................................................. 27
Board connection .............................................................................. 29
Steps for board testing ..................................................................... 30
Safety instruction .............................................................................. 30
5 Dimmer schematic .......................................................................... 31
6 Board layout and silkscreen .......................................................... 32
8 Temperatures and turn-off energies ............................................. 34
9 Conduction times calculation ....................................................... 36
10 Regulation stage setting ................................................................ 40
11 Experimental test and results @ 110Vac ...................................... 43
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List of figures
List of figures
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Figure 17: Voltage signals VPIN3, VPIN9 and VPIN13 in steady state operation @ minimum input power
– top and bottom assembly ...................................................................... 32
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1
1.1
Operation principle
Operation principle
Functional block diagram
The simplified functional diagram of the board is shown in the figure below. In this dimmer design, a single source driving network is used to simultaneously activate and deactivate both gates of the two STF17N62K3 MOSFETs (Q1 and Q2) having sufficiently low R
DS(on) at the operating currents (R
DS(on) max
= 0.34
Ω @ V
GS
= 10 V and I
D
= 7.5 A at T
C
= 25 °C) and connected to each other in an anti-series configuration or common source electrical topology.
In steady state working conditions, the digital control logic stage supplies the gate driving network with a voltage signal V
DR
equal to 15 V and pulsed at twice the mains frequency in order to properly drive the MOSFETs with a PWM gate to source voltage signal V
GS
~ 14V.
The dimmer operates in trailing edge operation mode and each MOSFET is therefore always turned on instantanly at the beginning of the half line cycle in both positive and negative main envelopes and maintains the on-state condition up to a controllable time instant set internally to the same half cycle.
Figure 2: STEVAL-ILD005V1 functional block diagram
From the block diagram above, the following distinct functional stages interact with each other to perform the overall dimming functionality:
Zero crossing detection network: consists of two passive sub-networks (each consisting of resistors, capacitors and diodes) connected separately to both phase and neutral lines and symmetrically arranged with respect to the ground plan. It allows detection of the crossing point of the input main voltage with the zero level in the ascendant or descendant slope phases for the respective positive and negative portions of the main line envelope in order to properly activate the conduction phases of the two MOSFETs.
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Operation principle
AN4717
Supply stage: allows the generation of the dc voltage signal V
CC
clamped to around
15 V by a Zener diode in order to provide the V
DD
supply voltage necessary for the triple 3-input NOR gate logic activation.
Gate driving network: consists of resistances only with an appropriate setting to guarantee the proper driving condition for the MOSFET gates and to achieve the best trade-off between efficiency and EMI performance.
Timing control and regulation stage: consists of the potentiometer and other passive components (resistances and capacitances) opportunely set to allow the regulation and control of the conduction time of the MOSFETs in order to perform the dimming function.
Digital control logic stage: implemented with a triple 3-Input NOR Gate logic to detect the zero crossing point instant time and activate the switch on phases of the
MOSFETs with the application of the gate driving voltage signal for an interval set by the Timing control and regulation stage and depending on the specific operating condition for the dimming function.
No EMI filtering stage (capacitor + inductor) is required at the input of the dimmer because the gate driving network is set to slow the current falling edge down at the turn-off switching transients in order to reduce the conducted electromagnetic noise and meet the requirements of the EN55015 standard.
The electrical design is also optimized to limit the MOSFET power losses during conduction phases and maximize the overall efficiency for the best trade-off between power dissipation and EMI. Due to the alternating trend of both line mains and lamp current, as the two
MOSFETs are connected to each other in an anti-series configuration with a single gate driving network implemented to simultaneously control both devices, in every half cycle of the main voltage, the current always flows through one MOSFET in forward conduction mode (or in the first quadrant operation and crossed by a positive drain current) and simultaneously through the other MOSFET in reverse conduction mode (or in the third quadrant operation and crossed by a negative drain current). In this way, it is also possible to exploit the ohmic resistance reverse conduction capability of the MOSFET channel when conducting a reverse direction current (from source to drain) by maintaining the same
R
DS(on)
characteristic as the conventional first quadrant operation.
Obviously, the reverse drain current flows through the MOSFET channel resistance by shunting the parasitic body diode, provided that the V
DS(on)
value is small enough and lower than the forward drop voltage of the intrinsic reverse diode, due to the appropraite low onresistance value in that particular operating condition for the MOSFET current. External diodes (D5 and D6) connected in antiparallel to each MOSFET are also inserted to allow an additional path for the sharing of the lamp current with the MOSFET (channel or intrinsic diode) during the negative drain current conduction phase and in particular during the startup phases when the average lamp current levels can be higher than the steady state working condition as a consequence of the cold state of the board and lighting load.
At 230 V
AC
input main voltage conditions, the minimum and maximum duration of the settable duty cycles are approximately equal to 20% and 82% respectively of the whole half line cycle.
For a positive mains half cycle, the lamp current passes through:
Q1 working on the first-quadrant operation or forward conduction with positive current
Q2 working on the third-quadrant operation and reverse conduction current (through channel resistance or intrinsic diode)
D6 working on forward biasing condition
For negative mains half cycle, the lamp current passes through:
Q2 working on the first-quadrant operation or forward conduction with positive current;
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1.1.1
Operation principle
Q1 working on the third-quadrant operation and reverse conduction current (through channel resistance or intrinsic diode);
D5 working on forward biasing condition.
A dead time for the current across the MOSFETs also occurs when the gate driving voltage signal is zero and remains in this state until the end of the half line cycle with no current flowing in the lamp.
Digital control logic
The following digital control logic stage functional diagram shows the interconnection of the three 3-input NOR gate PINS and the relative PIN descriptions.
Figure 3: Digital control logic functional diagram
VDD
(PIN 14)
PIN 3
IN
PIN 4
F
PIN 6 PIN 1
IN
PIN 5
PIN 2
F
PIN 9
OUT
PIN 8 PIN 13
IN
OUT
PIN 10 ( PIN 8) PIN 10
PIN 12
PIN 11
F
F
(PIN 7)
VSS
F
GSPG07072015DI1610
Pin no.
Table 1: Pin description
Name and function
3, 5, 13
9, 10 (≡ 8)
1 ≡ 6
Data inputs
Data outputs
2, 4, 11, 12
7
14
Intermediate connection
Connected to GND
Negative supply voltage V
SS
to GND
Positive supply voltage to V
DD
The digital control is implemented by a logic circuit with 3 inputs and 2 outputs and the associated logic function (calculating the correlation between each output and the inputs) can be expressed via the following mathematical expressions (where I
x
and 0
y
indicate the respective input and output PINs to the logic circuit):
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Operation principle
Equation 1
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{
𝑶
9
= (𝑰
3
𝑶
10
+ 𝑰
5
= 𝑰
13
) ∙ 𝑰
13
All inputs are protected by a CMOS protection network as shown in the figure below, referring to the single triple input NOR gate where the numbers in parentheses are terminal numbers for the other gates (refer to the CMOS NOR gate CD4025B datasheet).
Figure 4: Single triple input NOR gate schematic
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Therefore, for all the logic inputs, the voltage range (with reference to V
SS
= 0 V pin voltage) is from -0.5 V to V
DD
+0.5 V, where the positive supply voltage V
DD
= V
CC
– VD7 and the maximum dc voltage signal V
CC
(generated by the supply stage) is clamped to
+15 V by a Zener diode between V
CC
and the negative supply voltage pin (V
SS
) of the logic.
The following truth table describes the Boolean function with all the possible operating conditions (or states) that the logic circuit can theoretically assume in the steady state working condition and expressed as correlations between inputs and outputs with binary symbols 0 and 1.
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Operation principle
Pin
Table 2: Truth Table of the logic states
Truth table
Inputs Outputs
3 5 13 9 10
State Working phase
Operating condition
0
0
0
0
0
1
0
1
0
0
0
0
1
0
1
1
2
3
Steady state (positive and negative main voltage envelope)
Steady state (positive and negative main voltage envelope)
X
Steady state (negative main voltage envelope)
MOSFET on state
MOSFET off state
(first phase)
Logic values
0 1 1 1 0 4
MOSFET off state
(second phase)
1
1
0
0
0
1
0
1
1
0
5
6
X
Steady state (positive main voltage envelope)
X
MOSFET off state
(second phase)
1 1 0 0 1 7
1 1 1 1 0 8 X
In the above table, the symbol X indicates that the input combinations referring respectively to STATES 3, 5, 7 and 8 never occur during regular dimmer operation and therefore can be ignored.
The real operational states are:
STATE 1: occurs during the steady on-state operation of the MOSFETs
STATES 2, 4 and 6: occur during the steady off-state operation of the MOSFETs
All the operating conditions for the above states can also occur during the dimmer startup phases when a low biasing condition for the driving voltage V
DR
at the output of the digital control logic circuit imposes an inadequate gate-to-source voltage V
GS
(lower than the nominal value of ~ 14 V) because the supply voltage V
DD
s till hasn’t reached the steady state value of 15 V during its rise in the startup transient.
The supply stage voltage signal and all the input/output data from/to the Triple 3-input NOR
gate logic circuit are detailed below and distinguished in
Figure 5: "Digital signals processed by the digital control logic"
as digital signals of the logic circuit functionality:
Supply stage: voltage signal V
CC
Input data to the logic circuit:
signals from cross detection stage: voltage signals V
PIN3
and V
PIN5
at PIN3 and
PIN5 respectively
signal from timing control stage: voltage signal V
PIN13
at PIN13
Output data from the logic circuit:
signal to gate driving stage: voltage signal V
PIN10
at PIN10
signal to timing control stage: voltage signal V
PIN9
at PIN9
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Operation principle
Figure 5: Digital signals processed by the digital control logic
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The theoretical waveforms and command sequences from/to the logic circuit are depicted
Figure 6: "Theoretical waveforms and command sequences"
voltage variation under steady state working conditions (STATE 2 is not included as it is a fast transitory in comparison to the others), with:
V
LINE-min
, V
LINE-max
= maximum and minimum values of the main voltage
VZ-D8 = D8 diode breakdown voltage (equal to 15 V with a tolerance of 20 %)
V
CC-max
= maximum value of the supply voltage V
CC
(with V
CC
= V
DD
+V
D7
~V
DD
)
V
REG
= Voltage drop on the whole resistive regulation stage with the resistive potentiometer and the other series and parallel resistances (R21, R22, R23 and R24)
V
IH-PIN3
, V
IH-PIN5
, V
IH-PIN13
= high voltage levels of the input data at PIN3, PIN5 and
PIN13 (with V
IH-PIN3
= V
IH-PIN5
~ V
CC max
by neglecting the voltage drops on the resistances R13/R14 and on the diodes D1/D2/D7; V
IH-PIN13
~ V
CC max
-V
REG
)
V
T
= the threshold value (set to around V
CC max
/2) of each NOR gate input
V
OH-PIN9
, V
OH-PIN10
= high voltage levels of the output data at PIN9 and PIN10 (with
V
OH-PIN9
= V
OH-PIN10
~ V
CC max
)
Moreover, for all the low voltage levels of the input (V
PIN3
, V
PIN5
and V
PIN13
) and output (V
PIN9 and V
PIN10
), the data is assumed to be:
Equation 2
V
IL-PIN3
= V
IL-PIN5
= V
IL-PIN13
= V
OL-PIN9
= V
OL-PIN10
= V
SS
= 0 V
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Operation principle
Figure 6: Theoretical waveforms and command sequences
1.2 Steady-state stage-wise circuit analysis
In a trailing edge phase-cut dimmer, the MOSFETs are turned on when the line voltage
(and therefore the current through the devices) passes through zero during the ascendant or descendent slopes of its respective positive or negative envelopes. The dimmer uses a timing control and regulation circuit to turn the current supply to the fixture on and off at
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Operation principle regular intervals, typically 100 or 120 times per second (for 50 Hz or 60 Hz line frequencies), too fast to be perceived by the human eye.
AN4717
The circuit implements two STF17N62K3 MOSFETs as switches: when the switches are simultaneously turned on, a significant current flows through the devices and the lamp, while when the switches are simultaneously turned off, negligible current flows through the circuit. The amount of time in which the current is on determines the brightness or dimming effect of the lamp. In fact, by turning the switch on and off twice every line cycle, the rms voltage applied to the lamp is reduced from 230 V
AC
to a lower value, so the rms current that flows through the lamp is also reduced and produces less light.
During the dimmer startup phases, the MOSFETs working in the on state may operate under a low biasing condition for the gate driving voltage signal because the supply voltage
V
CC
, such as all the input/output logic circuit PINs, hasn't properly reached the nominal voltage of ~15 V.
This section provides a comprehensive description of the trailing edge phase cut dimmer circuit operation through a stage-by-stage circuit analysis under steady state conditions, divided into sequential functional phases according to the variation over a period of the main voltage starting from the time instant at zero voltage value on the ascendant slope of the positive envelope. For simplification purposes, STATE 2 is not included (considered a fast transitory compared to the others during the normal operation) and the leakage currents flowing on the diodes working in reverse-bias operation are neglected.
Conduction phase on the positive main voltage envelope (STATE 1): the zero crossing detection network recognizes the instant when the main line passes through the zero value on the ascendant voltage slope because all the three input signals to the digital control logic circuit V
PIN3
, V
PIN5
and V
PIN13
are at the same low voltage detected as zero logic level conditions (respectively I3=0, I5=0 and I13=0). In this case, the turn on phase of the two MOSFETs is instantaneously activated by the logic circuit that applies a high logic level signal (O10=1, V
DR
=V
OH-PIN10
~V
CC max
) at the output of the PIN10 for the resistive driving network to the gates, maintaining a zero logic level output at PIN9 (09 = 0, V
PIN9
= 0 V).
Due to the slow discharging transient of the supply stage (i.e., paralleled capacitances
C3 and C4) through the regulation stage (with the resistive potentiometer and the other series and parallel resistances R21, R22, R23 and R24), the voltage V
PIN13 increases during the whole conduction phase via the charging current I
REG-ON
on the capacitive node corresponding to PIN13. Therefore, the conduction phase lasts for the time required for the voltage on PIN13 (V
PIN13
) to reach the threshold value of around
V
CC max
/2 starting from the zero value condition.
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Operation principle
Figure 7: Conduction phase on the positive main voltage envelope
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Operation principle
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Turn-off phase on the positive main voltage envelope (STATE 2 and STATE 6):
STATE 2: when the voltage V
PIN13
of the input signal at PIN13 reaches the threshold value V
T
~V ccmax
/2, the following changes occur in sequence: a. input signal V
PIN13
changes its status detected by the digital control logic from low logic level (I13=0) to high logic level (I13=1); b. output signal V
PIN10
changes its status from high logic level (O10=1,
V
PIN10
~V ccmax
) to zero logic level (O10=0, V
PIN10
= 0 V).
Under this operating condition, the voltage signal V
DR
applied to the resistive driving network for the MOSFET gates is turned off instantly.
STATE 6: As a consequence of the disabled condition for driving the MOSFETs, the charge of the capacitance C1 placed on the cross detection network with a current I
DET-3
becomes involved and the following changes occur in sequence: a. input signal V
PIN3
changes its status from low logic level (I3=0) to high logic level (I3=1, V
PIN3
~ V
CC max
) b. output signal V
PIN9
changes its status from zero logic level to high logic level
(O9=1, V
PIN9
~V
CC max
) c. input signal V
PIN13
instantly rises to V
IH-PIN13
= V
CC max
– V reg
and a very low discharge current I
REG-OFF
continues to flow from the supply stage (paralleled capacitances C3 and C4) towards the node associated with PIN13 involving full discharge and voltage polarity inversion phases sequentially on paralleled capacitances C5 and C6.
When the main voltage reaches zero during its positive descendant slope, the following changes occur in sequence:
1. capacitance C1 is fully discharged and the input signal V
PIN3
changes status from high logic level (I3=1 associated with V
PIN3
~ V
CC max
) to zero logic level (I3=0)
2. output signal V
PIN9
changes status to zero logic level (O9=0), as the input signal V
PIN5 is at low logic level (I5=0)
3. input signal V
PIN13
instantly passes from high to zero logic level condition.
This last operating (STATE 1) condition, with all the three V
PIN3
, V
PIN5
and V
PIN13
input signals at the same zero logic level condition (I3=0, I5=0 and I13=0), sets the digital control logic circuit for the new conduction phase in the next half main cycle.
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Operation principle
Figure 8: Turn-off phase on the positive main voltage envelope
Conduction phase on the negative main voltage envelope (STATE 1): the zero crossing detection network recognizes the instant when the main line passes through the zero value on the descendant voltage slope because all the three input signals to the digital control logic circuit V
PIN3
, V
PIN5
and V
PIN13
are at the same low voltage detected as zero logic level conditions (respectively I3=0, I5=0 and I13=0). In this case, the turn-on phase of the two MOSFETs is instantaneously activated by the logic circuit that applies a high logic level signal to the output of PIN10 (O10=1,
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Operation principle
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V
DR
=V
PIN10
~V
CC max
) for the resistive driving network to the gates, maintaining the output at PIN9 at zero (09=0,V
PIN9
= 0 V).
Due to the slow discharging transient of the supply stage (paralleled capacitances C3 and C4) through the regulation stage (resistive potentiometer and the series and parallel resistances R21, R22, R23 and R24), the voltage V
PIN13
rises during the whole conduction phase via the charging current I
REG-ON
on the capacitive node associated with PIN13. Therefore, the conduction phase lasts for the period required for V
PIN13
to reach the threshold value of around V ccmax
/2 starting from the zero value condition.
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Operation principle
Figure 9: Conduction phase on the negative main voltage envelope
Turn-off phase on the negative main voltage envelope (STATE 2 and STATE 4):
STATE 2: when the input signal voltage V
PIN13
reaches the threshold value
V
T
~V ccmax
/2, the following changes occur in sequence: a. input signal V
PIN13
changes its status detected by the digital control logic from low logic level (I13=0) to high logic level (I13=1) b. output signal V
PIN10
changes its status from high logic level (O10=1, V
PIN10
~V ccmax
) to zero logic level (O10=0, V
PIN10
= 0 V)
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Operation principle
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Under this operating condition, the voltage signal V
DR
applied to the resistive driving network for the MOSFET gates is instantly turned off.
STATE 4: as a consequence of the disabled condition for driving the MOSFETs, the charge of capacitance C2 placed on the cross detection network with current I
DET-5 becomes involved and the following changes occur in sequence: a. input signal V
PIN5
changes its status from the low logic level (I5=0) to the high logic level condition (I5=1, V
PIN5
~ V ccmax
) b. output signal V
PIN9
changes from zero to high logic level (O9=1, V
PIN9
~V ccmax
) c. input signal V
PIN13
voltage instantly rises to V
IH-PIN13
= V ccmax
– V reg
and a very low discharge current I
REG-OFF
continues to flow from the supply stage (paralleled capacitances C3 and C4) towards the node relating to PIN13 involving the full discharge and voltage polarity inversion phases sequentially on the paralleled capacitances C5 and C6.
When the main voltage reaches zero during its negative ascendant slope, the following changes occur in sequence:
1. capacitance C2 is fully discharged and input signal V
PIN5
changes from high logic level
(I5=1, V
PIN5
~ V ccmax
) to zero logic level (I5=0)
2. output signal V
PIN9
changes to zero logic level (O9=0), as the input signal V
PIN3
is in the low logic level condition (I3=0)
3. input signal V
PIN13
instantly passes from the high to zero logic level.
This last operating (STATE 1) condition with all the three input signals V
PIN3
, V
PIN5
and
V
PIN13
at the same zero logic level (respectively I3=0, I5=0 and I13=0) sets the digital control logic circuit for the new conduction phase in the next half main cycle.
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Operation principle
Figure 10: Turn-off phase on the negative main voltage envelope
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Operating conditions and selected components
2
2.1
2.2
2.3
Operating conditions and selected components
AN4717
Basic features
This board is designed to dim resistive or capacitive lighting loads available for domestic or industrial use. The main features of this dimmer are:
operation for 2-wire wall dimmer
trailing-edge control only (compatible with all lamps commonly found on the market)
operation on 110 V rms
or 230 V rms
line voltage (±10 %) and 50 Hz or 60 Hz line frequency
dimmable power range (without heatsinks mounted on MOSFETs)
40 W to 300 W for 230 V rms
line
15 W to 90 W for 110 V rms
line
maximum operating ambient temperature: 50 °C
power efficiency @ Max power - 230 V
AC
> 95%
power factor @ Max power > 0.98
control and regulation interface with a switched type single linear rotary potentiometer
supported loads
dimmable CFL/LED lamps
incandescent bulbs with halogen technology
electronic low voltage transformers dimmable by trailing edge phase control
MOSFET
The STF17N62K3 15.5 A, 620 V SuperMESH3 MOSFET in a TO-220 full package is the most suitable device for the requirements of this trailing edge phase cut dimmer switch application operating at line frequency (50/60 Hz) thanks to its high total input capacitance
(C iss
=3100 pF @ V
DS
= 50 V) and sufficiently low on-resistance at operating currents.
The main features of STF17N62K3 are:
extremely low on-resistance: R
DS(on) max
= 0.34
Ω with V
GS
= 10 V and I
D
= 7.5 A at
T
C
= 25 °C high avalanche capability with I
zener-protected.
AR
= 15.5 A and E
AS
= 260 mJ
extremely high dv/dt capability: dv/dt = 9 V/ns with I
SD
≤ 15.5 A, di/dt ≤ 400 A/µs
Triple 3-inputs CMOS NOR gate
The CMOS NOR gate CD4025B (pin-to-pin compatible with ST's HCF4025BE 14-lead dual in-line CMOS gate logic device in a plastic SO-14 micropackage) fits very well with the light dimmer application requirements by performing the control of the board.
Its main features are:
propagation delay time: t
PD
= 60 ns (typ.) at V
DD
= 10 V, C
L
= 50 pF
buffered inputs and outputs
standardized symmetrical output characteristic
100 % tested for maximum quiescent current at 20 V
5 V, 10 V and 15 V parametric ratings
maximum input current of 1 μA at 18 V over full package temperature range; 100 nA at
18 V and 25 °C
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2.4
Operating conditions and selected components
noise margin (over full package temperature range): 1 V at V
DD
=5 V, 2 V at V
DD
=10 V and 2.5 V at V
DD
=15 V.
Linear rotary potentiometer
A single linear rotary potentiometer equipped with a mechanical switch which opens the circuit at minimum setting is used to implement lamp current dimming and power regulation with on/off power switch and up/down dimming functions.
The main characteristics include:
resistance range (LAW): 1 kΩ to 1 MΩ
resistance tolerance: ±20 % (±10 % per selection)
rated dissipation at 40 °C: 0.4 W
limiting element voltage: 500 V DC or AC RMS
terminal resistance: 5 Ω maximum
noise (ENR): 2 % maximum
mechanical rotation: 300°
operating temperature range: -25 to +70 °C
DocID027977 Rev 1 21/48
Board performances
3
3.1
Board performances
AN4717
Experimental test and results @ 230VAC
The MOS-based dimmer solution includes two STF17N62K3 MOSFETs connected in antiseries configuration and a single linear potentiometer to set the turn-on time (or duty cycle) of the devices as in the following three operating conditions representing the minimum, medium and maximum levels of lamp current dimming and input power absorbed by the board:
minimum power: ~2.0 ms (or ~20 % of duty cycle in the semiperiod)
medium power: ~4.5 ms (or ~45 % of duty cycle in the semiperiod)
maximum power: ~8.2 ms (or ~82 % of duty cycle in the semiperiod)
The following figures show the waveforms acquired on the steady state operation with
300 W halogen technology incandescent bulbs at 230 V
AC
input mains for the working conditions described above for the MOSFETs.
The waveforms were acquired by scope with the signal color codes:
Q1 drain current: signal ID1 = yellow
Q2 drain current: signal ID2 = blue
Q1 drain-source voltage: signal VDS1 = red
Q2 drain-source voltage: signal VDS2 = green
Figure 11: Steady state operation with ~2.0 ms turn-on time @ minimum input power
22/48 DocID027977 Rev 1
AN4717
Board performances
Figure 12: Steady state operation with ~4.5 ms turn-on time @ medium input power
Figure 13: Steady state operation with ~ 8.2 ms turn-on time @ maximum input power
The following figures show the waveforms acquired for the input signals to the digital logic circuit, respectively the voltages on the pins PIN3 (V
PIN3
) and PIN5 (V
PIN5
) together with the
230 V
AC
input mains, during the steady state operation with halogen technology incandescent bulbs at the minimum, medium and maximum input power conditions.
Waveforms were acquired by scope with the following signal color codes:
input signal on the PIN3: voltage V
IN3
input signal on the PIN5: voltage V
IN5
input main voltage: signal V
Line
= red
= green
= yellow
DocID027977 Rev 1 23/48
Board performances
AN4717
Figure 14: Voltage signals VPIN3 and VPIN5 in steady state operation @ minimum input power
Figure 15: Voltage signals VPIN3 and VPIN5 in steady state operation @ medium input power
24/48
Figure 16: Voltage signals VPIN3 and VPIN5 in steady state operation @ maximum input power
DocID027977 Rev 1
AN4717
Board performances
The figure below shows the waveforms acquired for the digital logic circuit input and output signals, respectively the voltages on the pins PIN3 (V
PIN3
), PIN13 (V
PIN13
) and PIN9 (V
PIN9
) together with the 230 V
AC
input mains during the steady state operation with halogen technology incandescent bulbs at the minimum input power condition and 230 V
AC
input mains.
Waveforms were acquired by scope with the following signal color codes:
input signal on the PIN3: voltage V
IN3
input signal on the PIN13: voltage V
output signal on the PIN9: voltage V
input main voltage: signal V
Line
= red
= green
IN13
= blue
OUT9
= yellow
Figure 17: Voltage signals VPIN3, VPIN9 and VPIN13 in steady state operation @ minimum input power
The following figure shows the waveforms acquired respectively for the input digital logic circuit signal of the voltage on PIN13 (V
PIN13
), the gate-to-source voltage signal V
GS
and the dimmer input voltage signal V in
during the steady state operation with halogen technology incandescent bulbs at the minimum input power condition and 230 V
AC
input mains.
Waveforms were acquired by scope with the following signal color codes:
input signal on the PIN13: voltage V
gate-source voltage: signal V
IN13
GS
= blue
= yellow
dimmer input voltage: signal V
IN
= red
DocID027977 Rev 1 25/48
Board performances
AN4717
Figure 18: Voltage signals VPIN13 and VGS in steady state operation @ minimum input power
3.2
3.2.1
As can be seen, the MOSFET gate-source voltage is set to approx. 14 V during the conduction phases in order to keep the channel of the devices activated while crossed by the lamp current.
Electromagnetic compatibility test
EMC conducted noise compliance according to EN 55015
Conducted emission noise was tested according to standard EN 55015 under the following conditions:
230 V
RMS
- 50 Hz mains voltage
300 W/230 V
AC
incandescent bulbs with halogen technology
tests were carried out with the rotary potentiometer set to different rotation angles corresponding to different conduction angles/times of the MOSFETs and input power levels absorbed by the dimmer
The following figure gives the measurement of the maximum conducted noise level reached for the minimum input power level condition (40 W).
26/48 DocID027977 Rev 1
AN4717
Board performances
Figure 19: Conducted noise @ minimum input power
3.2.2
The above figure demonstrates that the conducted emission noise measured at the minimum input power condition (i.e., worst case for the generation of the EMI disturbances) was compliant with the EN55015 electronic standard specifications, with the QP and AV detectors always below the corresponding QP and AV limits in the frequency range from
9 kHz to 30 MHz. As can be seen, thanks to the appropriate setting of the resistances on the gate driving network to the MOSFETs, no LC noise input filter is needed to meet the
European EMC standard regulation in force for the conducted disturbances.
Surge voltage immunity test
Surge immunity tests were carried out on the MOS-based dimmer in order to analyze the board performance when sequences of high energy but relatively slow transient overvoltages are applied on the power lines. The tests were performed according to the standard IEC 61000-4-5 relating to the immunity requirements by applying voltage surges of amplitudes equal to 2 kV in both positive and negative polarities with different phase angles shifted in relation to the input main voltage.
A 300 W/230 V
AC
halogen technology incandescent bulbs is connected in series to the board running under normal operating conditions at 230V
AC
and 50Hz frequency.
The figure below shows the STF17N62K3 MOSFET response during the tests performed when a 2 kV surge pulse with positive polarity is applied to the AC power port in differential mode (symmetrical configuration as line-to-line between the phase and neutral lines) with
0º and 90º shifted phase angles respectively under medium input power working conditions for the dimmer. The waveforms acquired by scope for the MOSFET Q1 are the drainsource voltage (VDS1 signal in red), the drain-source current (IDS1 signal in yellow) and the input main voltage (V
Line
signal in green).
DocID027977 Rev 1 27/48
Board performances
Figure 20: Surge voltage pulse @ medium input power
AN4717
The MOSFET STF17N62K3 was able to withstand the surge pulses without any failure condition and the dimmer continued to operate normally without any temporary degradation or loss of functionality and with automatic return to normal operation after the disturbance ceased.
28/48 DocID027977 Rev 1
AN4717
4
4.1
Getting started
Getting started
Board connection
The figure below shows the connection diagram of the STEVAL-ILD005V1 to the lamp to be dimmed.
Please unplug the circuit from the line before setting the different connections.
Figure 21: Connection diagram
4.2
1. header J1 is used to connect the two-wire dimmer to the mains in series with the lamp
2. terminals T1 or T2 of the Header J1 can either be connected to the lamp or directly to the mains
Appropriate connection of the lamp to mains neutral or line depends on the applicable electrical safety regulations, even if the lamp must usually be connected to neutral.
User interface
The user interface consists of a rotative potentiometer with integrated ON/OFF switch:
1. the ON/OFF switch is used to turn the dimmer board and lamp on and off
2. the clockwise rotation of the potentiometer up to the maximum rotational angle
(representing the maximum resistive value) allows continuous power absorption and light emission modification by increasing lamp brightness up to the maximum intensity level
3. the anticlockwise rotation of the potentiometer up to the minimum rotational angle
(representing the minimum resistive value) allows continuous power absorption and light emission modification by decreasing lamp brightness down to the minimum intensity level
DocID027977 Rev 1 29/48
Getting started
4.3 Steps for board testing
Follow this procedure to use the STEVAL-ILD005V1 board:
AN4717
1.
connect one lamp terminal to one J1 header terminal (T1 or T2) as shown in
2. connect the other lamp terminal to a plug of the 230 Vac AC power source switched off
3. connect the other J1 terminal to the other plug of the 230Vac AC power source switched off
4. switch the 230Vac AC power source on
5. if the potentiometer switch opens the circuit in the OFF position, close the switch by rotating the potentiometer clockwise to turn the dimmer board and lighting load on
At this point, the STEVAL-ILD005V1 is working correctly if:
the lamp brightness increases by rotating the potentiometer clockwise
the lamp brightness decreases by rotating the potentiometer in anticlockwise
the lamp brightness remains stable when no rotation is performed
4.4 Safety instruction
Follow this procedure to use the STEVAL-ILD005V1 board:
Waring: The high voltage levels used to operate the MOS-based dimmer evaluation board could present a serious electrical shock hazard. This evaluation board must be used in a suitable laboratory by qualified personnel only, familiar with the
installation, use, and maintenance of electrical power systems.
The STEVAL-ILD005V1 evaluation board is designed for demonstration purposes only, and shall not be used either for domestic installation or for industrial installation.
30/48 DocID027977 Rev 1
AN4717
5 Dimmer schematic
Figure 22: Dimmer schematic
Dimmer schematic
A 5 A/250 V
AC
fuse F1 is mounted to protect MOSFETs in case of lamp flash-overs or accidental short-circuits.
DocID027977 Rev 1 31/48
Board layout and silkscreen
6 Board layout and silkscreen
Figure 23: Component layout
– top and bottom assembly
AN4717
Figure 24: PCB layout
– top and bottom view
32/48 DocID027977 Rev 1
AN4717
7
Bill of material
Bill of material
Reference
U1
Q1, Q2
P1A/P1B
R19, R20
R21
R22
R23
R24
VR1
R26
R27, R28
R1, R2, R3, R4, R5, R6, R7, R8, R9 R10, R11, R12
R13, R14
R15, R16
R17, R18
Table 3: Bill of material
Part / value
CD4025BM
STF17N62K3
1 MΩ/0.4 W
12 kΩ / 2 W
2.2 kΩ / 0.25 W
100 kΩ / 0.125 W
10 kΩ / 0.25 W
1 kΩ / 0.25 W
1
MΩ / 0.25 W
560 kΩ / 0.25 W
150 kΩ / 0.25 W
120 kΩ / 0.25 W
Varistor / 420 V
0 Ω / 0.25 W
22 kΩ / 0.25 W
C4
C3
C5
C6
D1, D2, D3, D4, D7
D5, D6
D8
C1, C2
Rfuse
FUSE HOLDER
J1
S1M
N.M.
1N4007
BZV55B15
5 A / 250 V
2.2 nF / 0.25 W
47 µF / 25V
100 nF / 0.25 W
22 nF / 100V
AC
6.3 A / 250 V
CC
AC
2-pin PCB screw terminal block
DocID027977 Rev 1 33/48
Temperatures and turn-off energies
8 Temperatures and turn-off energies
AN4717
The figure below shows the different temperatures and turn-off switching energies measured for the STF17N62K3 MOSFETs working under the three different operating conditions for minimum, medium and maximum levels of lamp current dimming and input power absorbtion by the board connected to 300 W/230 V
AC
halogen technology incandescent bulbs.
Figure 25: Temperatures and turn-off energies @ 230 VAC
34/48
The maximum case temperatures (approx. 76 °C without heatsinks) are reached in free air conditions (at 25 °C room temperature) for the medium input power level operation and can be reduced by mounting appropriate heatsinks in contact with the MOSFETs. Switching losses can be reduced by decreasing the original value of the two gate drive resistances
R2
7 and R28 (set to 22 kΩ) to speed up the turn-off switching transients. This modification implies a new noise emission condition for the conducted disturbances introduced by the dimmer and may require the use of an appropriate EMI input filter to guarantee compliance with the EN55015 standard.
The following figure depicts the turn-off transient detail for MOSFET Q2 switching under the medium input power operating condition.
DocID027977 Rev 1
AN4717
Temperatures and turn-off energies
Figure 26: Turn-off energy @ medium input power
DocID027977 Rev 1 35/48
Conduction times calculation
9 Conduction times calculation
AN4717
Assuming a sinusoidal envelope over the period T = 1/freq for the current lamp waveform under normal steady state conditions, as in:
Equation 3
𝐼 𝑙𝑎𝑚𝑝
(𝑡) = 𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
∙ sin(2𝜋 ∙ 𝑓𝑟𝑒𝑞 ∙ 𝑡) for 0<t<T
COND
and T/2<t<T/2+T
COND by neglecting the voltage drops on the R
DS(on)
channel resistances and on the forward biased antiparallel diodes during the MOSFET conduction phases, we can consider the mains voltage entirely applied on the lamp and therefore express the power P
IN
for
𝐷 =
𝑇
𝐶𝑂𝑁𝐷
and conduction time T
COND
thus:
𝑇
Equation 4
𝑃
𝐼𝑁
=
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
∙ 𝑉 𝑚𝑎𝑖𝑛_𝑀𝐴𝑋 𝜋
For a 90° MOSFET conduction angle,
[𝜋 ∙ 𝐷 − (
1
2
∙ 𝑠𝑖𝑛(2𝜋 ∙ 𝐷) ∙ 𝑐𝑜𝑠(2𝜋 ∙ 𝐷))]
𝐷 =
𝑇
𝐶𝑂𝑁𝐷
𝑇
=
1
4
, so:
Equation 5
𝑃
𝐼𝑁_90𝑑𝑒𝑔
=
1
4
∙ 𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
∙ 𝑉 𝑚𝑎𝑖𝑛_𝑀𝐴𝑋
The max value I lamp_MAX
for the current lamp signal (equalt to the maximum dimmer input current I
IN_MAX
) can be expressed with the following formula where P
IN_90deg
is the input power level at 90° MOSFET conduction angle and V main_MAX
is the maximum mains voltage value:
Equation 6
𝐼 𝑙𝑎𝑚𝑝
𝑀𝐴𝑋
=
4 ∙ 𝑃
𝐼𝑁_90𝑑𝑒𝑔
𝑉 𝑚𝑎𝑖𝑛_𝑀𝐴𝑋
By inserting:
𝑃
𝐼𝑁_90𝑑𝑒𝑔
= 𝑃
𝐼𝑁_𝑀𝐸𝐷
+ ∆% ∙ 𝑃
𝐼𝑁_𝑀𝐸𝐷
≅ 158𝑊 with P
IN_MED
= 150 W and ∆ % = 5 % and
𝑉 𝑚𝑎𝑖𝑛_𝑀𝐴𝑋
= 230 ∙ √2𝑉
We obtain:
Equation 7
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
=
4 ∙ 158
230 ∙ √2
≅ 1.94𝐴
The rms value of the lamp current I lamp_RMS
(equal to the rms value of the dimmer input current I
IN_RMS
) can be expressed with the following formula where P
IN_MAX
is the maximum input power, V main_RMS
is the rms value of the mains voltage and PF
MAX
is the power factor at the maximum input power:
Equation 8
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
= 𝐼
𝐼𝑁_𝑅𝑀𝑆
=
𝑉
𝑃
𝐼𝑁_𝑀𝐴𝑋 𝑚𝑎𝑖𝑛_𝑅𝑀𝑆
∙ 𝑃𝐹
𝑀𝐴𝑋
36/48 DocID027977 Rev 1
AN4717
Conduction times calculation
By imposing the following conditions for maximum input power operation:
P
IN_MAX
= 300 W
V main_RMS
= 230 V
PF
MAX
= 0.98
We obtain:
Equation 9
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
=
300
230 ∙ 0.98
= 1.33𝐴
The following formula correlates I lamp_MAX
with I lamp_RMS
in the phase cut dimming principle with a lamp current I lamp
having sinusoidal profile where
𝐷 =
𝑇
𝐶𝑂𝑁𝐷
𝑇
for MOSFET conduction time T
COND
:
Equation 10
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
=
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
Which can be reformulated thus:
Equation 11
√𝐷 − 𝑠𝑖𝑛(4𝜋𝐷)
4𝜋
= (
𝐼
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
)
Where:
Equation 12
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
=
1.33
1.94
= 0.686
With an appropriate choice for T
COND
= 8.2 ms:
Equation 13
𝐷 =
𝑇
𝐶𝑂𝑁𝐷
𝑇
=
8.2 𝑚𝑠
20 𝑚𝑠
= 0.41 and by inserting this value in √
𝐷 − 𝑠𝑖𝑛(4𝜋𝐷)
4𝜋
Equation 14
√𝐷 − 𝑠𝑖𝑛(4𝜋𝐷)
4𝜋
− (
𝐼
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
) ≅ 0.694 − 0.686 = 8 ∙ 10
−3
Therefore, from the result obtained in the above equation, the time T
COND
= 8.2 ms is a good approximation of the right value of the MOSFET conduction time during maximum input power conditions.
Assuming a linear envelope in the period T=1/freq for the current and voltage lamp waveforms during normal steady state operation at minimum input power, as in:
Equation 15
𝐼 𝑙𝑎𝑚𝑝
(𝑡) =
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
𝑇
𝐶𝑂𝑁𝐷
∙ 𝑡 for 0<t<T
COND
and T/2<t<T/2+T
COND
DocID027977 Rev 1 37/48
Conduction times calculation
Equation 16
𝑉 𝑙𝑎𝑚𝑝
(𝑡) =
𝑉 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
𝑇
𝐶𝑂𝑁𝐷
∙ 𝑡 for 0<t<T
COND
and T/2<t<T/2+T
COND
AN4717
In the above equation, by neglecting the voltage drops on the R
DS(on)
channel resistances and on the forward biased antiparallel diodes during the MOSFET conduction phases, we can express V lamp_MAX
for the voltage lamp signal as a function of the mains voltage thus:
Equation 17
𝑉 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
= 230 ∙ √2 ∙ sin(2𝜋 ∙ 𝑓𝑟𝑒𝑞 ∙ 𝑇
𝐶𝑂𝑁𝐷
)
As D = T
COND
/T, the lamp power for the minimum input power condition can be expressed as:
Equation 18
𝑃
𝐼𝑁_𝑀𝐼𝑁
=
2
3
∙ 𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
∙ 𝑉 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
∙ 𝐷
By imposing the following minimum input power working conditions:
P
IN_MIN
= 40 W
V main_RMS
= 230 V
PF
MIN
= 0.31 we calculate the rms value of the lamp current I lamp_RMS
(the rms value of the dimmer input current I
IN_RMS
Equation 19
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
= 𝐼
𝐼𝑁_𝑅𝑀𝑆
=
𝑃
𝐼𝑁_𝑀𝐼𝑁
𝑉 𝑚𝑎𝑖𝑛_𝑅𝑀𝑆
∙ 𝑃𝐹
𝑀𝐼𝑁
=
40
230 ∙ 0.31
= 0.561
The following formula correlates I lamp_MAX
with I lamp_RMS
in the phase cut dimming principle, with lamp current I lamp
having a linear variation profile where D=T
COND
/T for MOSFET conduction time T
COND
:
Equation 20
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
=
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
Equation 21
𝑉 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
∙ 𝐷 =
3
2
∙
𝑃
𝐼𝑁_𝑀𝐼𝑁
𝐼 𝑙𝑎𝑚𝑝_𝑀𝐴𝑋
⇒ √𝐷 ∙ 𝑠𝑖𝑛(2𝜋 ∙ 𝐷) =
√3
2
∙
𝐼
𝑃
𝐼𝑁_𝑀𝐼𝑁 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
∙ 𝑉 𝑚𝑎𝑖𝑛_𝑅𝑀𝑆 where:
Equation 22
√3
2
∙
𝑃
𝐼𝑁_𝑀𝐼𝑁
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
∙ 𝑉 𝑚𝑎𝑖𝑛_𝑅𝑀𝑆
=
√3
2
∙
40
0.561 ∙ 230
= 0.268
38/48 DocID027977 Rev 1
AN4717
Conduction times calculation
With an appropriate choice for T
COND
= 2.0 ms:
Equation 23
𝐷 =
𝑇
𝐶𝑂𝑁𝐷
𝑇
=
2 𝑚𝑠
20 𝑚𝑠
= 0.1 and by substituting this value into
Equation 24
√𝐷 ∙ 𝑠𝑖𝑛(2𝜋 ∙ 𝐷) −
√3
2
∙
𝑃
𝐼𝑁_𝑀𝐼𝑁
𝐼 𝑙𝑎𝑚𝑝_𝑅𝑀𝑆
∙ 𝑉 𝑚𝑎𝑖𝑛_𝑅𝑀𝑆
≅ 0.185 − 0.268 = −8 ∙ 10
−2
Therefore, from the above equation, the time T
COND
= 2.0 ms is a good approximation of the right value of the MOSFET conduction time under minimum input power conditions.
DocID027977 Rev 1 39/48
Regulation stage setting
10 Regulation stage setting
AN4717
To calculate the potentiometer resistance variation range and magnitude of the other passive components (resistances and capacitances) of the dimmer regulation stage, we can analyze the conduction phases of the MOSFETS at minimum and maximum levels of power absorbed in input.
Figure 27: Regulation stage
40/48
In both phases, the supply stage electrolytic capacitance C4 (set to 47μF) discharges on the capacitive section of the regulation stage (C5 and C6 paralleled) through the resistive section with the potentiometer and resistances R21, R22, R23 and R24. During this discharge transient of capacitance C4, the two capacitances C5 and C6 are charging and increase the voltage level up to the NOR gate input threshold voltage V
T
, set to around
V
CCmax
/2.
The condition of reaching the threshold V
T
corresponds with the duration time of the conduction phase set by the potentiometer for that specific lamp dimming functionality before the beginning of the MOSFET turn-off phase.The law governing the charging transient of the regulation stage capacitive section during the MOSFET conduction phases at both minimum and maximum input power conditions is given by the formula for the charge of a capacitor in an RC series circuit supplied by a constant voltage generator, generally expressed as:
Equation 25
𝑉
𝐶
(𝑡) = (𝑉
𝐶
(0) − 𝑉
𝐶𝐶
) ∙ 𝑒 + 𝑉
𝐶𝐶 where:
V
C
(t) is the voltage on the two paralleled capacitances C5 and C6 of the regulation stage
V
C
(0) is the voltage at time t = 0 on the two paralleled capacitances C5 and C6 of the regulation stage
V
CC
τ=R
is the voltage on the supply stage electrolytic capacitance C4 reg
.C reg
is the time constant of the circuit, where R reg
is the equivalent resistance of the resistive section with the potentiometer and resistances R21, R22, R23, R24 and
C reg
= C
5
||C
6
DocID027977 Rev 1
AN4717
Figure 28: RC series circuit
Regulation stage setting
Due to the high value of the electrolytic capacitance C4 (set to 47μF) and the low current levels flowing on the regulation stage during the C4 discharge phase, we can assume the supply stage voltage V
CC
to be constant during the charging phase of the capacitive section
C reg
= C
5
||C
6
.
So, by inserting Vc(0) = 0, Equation 25 is reduced to:
Equation 26
𝑉
𝐶
(𝑡) = 𝑉
𝐶𝐶
∙ (1 − 𝑒 )
We can reformulate in terms of the τ parameter thus:
Equation 27 𝜏 = 𝑙𝑛 (
𝑉
𝐶𝐶 𝑡
𝑉
𝐶𝐶
− 𝑉
𝐶
(𝑡))
Moreover, as the voltage level on C5 and C6 always rises during the MOSFET conduction phase up to the NOR gate input threshold voltage V
T
(set to around V
CCmax
/2), we can assume:
V
C
(T
ON
) = V
CC
/2 for the MOSFET conduction time t = T
ON
so Equation 27 becomes:
Equation 28 𝜏 = 𝑅 𝑟𝑒𝑔
∙ 𝐶 𝑟𝑒𝑔
=
𝑇
𝑂𝑁 𝑙𝑛(2)
For the minimum input power condition, T
ON
= 2 ms and so:
Equation 29
𝑅 𝑟𝑒𝑔_𝑚𝑖𝑛
∙ 𝐶 𝑟𝑒𝑔
=
2 ∙ 10
−3 𝑙𝑛(2)
= 2.88𝑚𝑠𝑒𝑐
By selecting R reg_min
= 120 kΩ for the equivalent resistance of the regulation stage resistive section at minimum input power:
Equation 30
𝐶 𝑟𝑒𝑔
=
2.88 ∙ 10
−3
120 ∙ 10 3
≅ 24 𝑛𝐹
And therefore capacitance C reg
= C
5
||C
6
= 22 nF is chosen for the design at the minimum input power condition.
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Regulation stage setting
AN4717
At the maximum input power condition, due to the long time constant for the charging transient, we consider a linear variation for the voltage on the two paralleled capacitances
C5 and C6 of the regulation stage; therefore, Equation 26 can be transformed into the following linearized form:
Equation 31
𝑉
𝐶
(𝑡) =
𝐼
𝑅𝑒𝑔−𝑚𝑎𝑥
𝐶
∙ 𝑡 where for t = T
ON
:
V
C
(T
ON
) = V
CC
/2
I reg_max
can be chosen as the current at the t = 0 to equal I reg_max
= V
CC
/R.
By substituting these two conditions in Equation 31, for T
ON
= 8.2 ms we obtain:
Equation 32
𝑅 𝑟𝑒𝑔_𝑚𝑎𝑥
∙ 𝐶 𝑟𝑒𝑔
= 2 ∙ 𝑇
𝑂𝑁
⇒ 𝑅 𝑟𝑒𝑔_𝑚𝑎𝑥
=
2 ∙ 𝑇
𝑂𝑁
𝐶 𝑟𝑒𝑔
=
2 ∙ 8.2 ∙ 10
−3
22 ∙ 10 −9
= 745 𝑘𝛺
The variability range of the equivalent resistance R reg
is therefore:
Equation 33
120 𝑘𝛺 ≤ 𝑅 𝑟𝑒𝑔
≤ 745 𝑘𝛺
The following values were therefore chosen for the potentiometer and resistances R21,
R22, R23, R24 of the regulation stage resistive section:
potentiometer = 1
MΩ
R21 = 1
MΩ, R22 = 560 kΩ, R23 = 150 kΩ, R24 = 120 kΩ
Using these resistances, the actual variation range for the equivalent resistance R reg becomes:
Equation 34
120 𝑘𝛺 ≤ 𝑅 𝑟𝑒𝑔
≤ 751 𝑘𝛺
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11
Experimental test and results @ 110Vac
Experimental test and results @ 110Vac
The following figures show the waveforms acquired under steady state operation with 90 W halogen technology incandescent bulbs at 110 V
AC
input mains for the above mentioned operating conditions for the MOSFETs.
Waveforms were acquired by scope with the following signal color codes:
Q1 drain current: signal ID1= yellow
Q2 drain current: signal ID2 = blue
Q1 drain-source voltage: signal VDS1 = red
Q2 drain-source voltage: signal VDS2 = green
Figure 29: Steady state operation @ 110 VAC and minimum input power
Figure 30: Steady state operation @ 110 VAC and medium input power
DocID027977 Rev 1 43/48
Experimental test and results @ 110Vac
Figure 31: Steady state operation @ 110 VAC and maximum input power
AN4717
The figure below shows the different temperatures and turn-off switching energies measured for the STF17N62K3 MOSFETs working under minimum, medium and maximum operating conditions for lamp current dimming and input powers absorbed by the board connected to 90 W/110 V
AC
halogen technology incandescent bulbs.
Figure 32: Temperatures and turn-off energies @ 110 VAC
44/48
The maximum case temperatures (~48 °C without heatsinks) are reached in free air conditions (25 °C room temperature) for the medium input power level operation with the original setting of 22 kΩ.
DocID027977 Rev 1
AN4717
Experimental test and results @ 110Vac
The following figure details the turn-off transient for MOSFET Q2 switching during medium input power operation.
Figure 33: Turn-off energy@ medium input power
DocID027977 Rev 1 45/48
Conclusion
12
AN4717
Conclusion
The STEVAL-ILD005V1 evaluation board implements a trailing edge operation mode for 2wire analog wall dimmers with flicker-free dimming and no audible noise performance for dimmable R-C lighting loads in the power range from 40 W to 300 W at 230 V
AC
input mains, by using a triple 3-input NOR gate to perform gate driving, zero crossing and timing control functions and a switched type single linear rotary potentiometer as the control interface. Thanks to the electrical characteristics of the STF17N62K3 MOSFET with high
C iss
values and low R
DS(on)
values at the operating currents, this MOS-based dimmer solution is able to meet the European EMC standard for conducted emission noise
(EN55015), without requiring any bulky LC input filters, relying instead on appropriate setting for the resistances of the gate driving network to the active devices. At the same time, the MOSFETs show satisfactory thermo-electrical performance without any heatsink components in contact with the case and the 620 V breakdown voltage ensures good margins for passing surge immunity tests with up to 2 kV voltage pulses, in compliance with the IEC61000-4-5 standards.
46/48 DocID027977 Rev 1
AN4717
13
Revision history
Revision history
Date
12-Aug-2015
Table 4: Document revision history
Version
1
Changes
Initial release.
DocID027977 Rev 1 47/48
AN4717
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– PLEASE READ CAREFULLY
STMicroelectronics NV and its subsidiaries (“ST”) reserve the right to make changes, corrections, enhancements, modifications, and improvements to ST products and/or to this document at any time without notice. Purchasers should obtain the latest relevant information on ST products before placing orders. ST products are s old pursuant to ST’s terms and conditions of sale in place at the time of order acknowledgement.
Purchasers are solely responsible for the choice, selection, and use of ST products and ST assumes no liability for application assistance or the design of
Purchasers’ products.
No license, express or implied, to any intellectual property right is granted by ST herein.
Resale of ST products with provisions different from the information set forth herein shall void any warranty granted by ST for such product.
ST and the ST logo are trademarks of ST. All other product or service names are the property of their respective owners.
Information in this document supersedes and replaces information previously supplied in any prior versions of this document.
© 2015 STMicroelectronics
– All rights reserved
48/48 DocID027977 Rev 1
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Key features
Operation for 2-wire wall dimmer
Trailing-edge control (compatible with all common lamps)
Operates on 110 Vrms or 230 Vrms line voltage (±10%) and 50 Hz or 60 Hz line frequency
Dimmable power range: 40-300W (230Vrms) / 15-90W (110Vrms) without heatsinks
High power efficiency (>95% @ Max power - 230 VAC)
High power factor (>0.98 @ Max power)
Control and regulation via a switched linear rotary potentiometer
Compliance with EMC standards (EN55015, IEC 61000-4-5)
RoHS compliant
Flicker-free and smooth dimming performance
Frequently asked questions
The STEVAL-ILD005V1 evaluation board implements a simple and cost-effective MOS-based reverse phase cut dimming solution for analog wall dimmers, operating in a trailing edge control mode.
This dimmer is compatible with resistive or capacitive lighting loads, including halogen lamps, electronic low voltage transformers, and various dimmable CFL/LED lamps, suitable for domestic or industrial use.
Control of the board and dimming functions are achieved using a single linear rotary potentiometer equipped with a mechanical switch, which also provides on/off power switching.
The STF17N62K3 15.5 A / 620 V SuperMESH3 MOSFET in the TO-220 full package is used for lamp current commutations in this dimmer switch application.
No, the design of the STEVAL-ILD005V1 is optimized to achieve acceptable levels of EMI noise without the need for LC filters to reduce conducted emission disturbances.
