STMicroelectronics R5972D Datasheet
Advertisement
Advertisement
R5972D
Up to 1.5 A step-down switching regulator for aerospace applications
Datasheet
-
preliminary data
– Advanced mold and frame designs for superior resilience in harsh environments
(acceleration, EMI, thermal, humidity)
– Extended screening capability on request
SO-8
Features
General features
– 1.5 A DC output current
– Operating input voltage from 4 V to 36 V
– 3.3 V / (± 2%) reference voltage
– Large ambient temperature range:
-40 °C to 125 °C
– Output voltage adjustable from 1.235 V to
V
IN
– Low dropout operation: 100% duty cycle
– 250 kHz internally fixed frequency
– Voltage feed-forward
– Zero load current operation
– Internal current limiting
– Inhibit for zero current consumption
– Protection against feedback disconnection
– Thermal shutdown
Aerospace and defense features
– Suitable for use in aerospace and defense applications
– Dedicated traceability and part marking
– Production parts approval documents available
– Adapted extended life time and obsolescence management
– Extended product change notification process
– Designed and manufactured to meet subppm quality goals
Application
Dedicated to aerospace applications
Description
The R5972D is a step-down monolithic power switching regulator with a minimum switch current limit of 1.8 A so it is able to deliver up to 1.5 A DC current to the load depending on the application conditions. The output voltage can be set from
1.235 V to V
IN
.
The device uses an internal P-channel DMOS transistor (with a typical R
DS(on)
of 250 m ) as a switching element to minimize the size of the external components.
An internal oscillator fixes the switching frequency at 250 kHz. The large ambient temperature range makes it ideal for aerospace and defense applications. A pulse-by-pulse current limit with the internal frequency modulation offers an effective constant current short-circuit protection.
October 2014 DocID026873 Rev 1
This is preliminary information on a new product now in development or undergoing evaluation. Details are subject to change without notice.
1/35 www.st.com
Contents
Contents
R5972D
Datasheet parameters over the temperature range . . . . . . . . . . . . . . . . 7
PWM comparator and power stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Additional features and protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Output overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Error amplifier and compensation network . . . . . . . . . . . . . . . . . . . . . . . . 13
2/35 DocID026873 Rev 1
R5972D
Contents
Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Positive buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Negative buck-boost regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Compensation network with MLCC at the output . . . . . . . . . . . . . . . . . . . 28
DocID026873 Rev 1 3/35
35
Pin settings
Figure 1. Pin connection (top view)
R5972D
No.
1
2, 3, 6, 7
4
5
8
Pin
OUT
GND
COMP
FB
VCC
Table 1. Pin description
Description
Regulator output.
Ground.
E/A output for frequency compensation.
Feedback input. Connecting directly to this pin results in an output voltage of 1.23 V. An external resistive divider is required for higher output voltages.
Unregulated DC input voltage.
4/35 DocID026873 Rev 1
R5972D Electrical data
Symbol
V
8
V
1
I
1
V
4
, V
5
P
TOT
T
J
T
STG
Table 2. Absolute maximum ratings
Parameter
Input voltage
OUT pin DC voltage
OUT pin peak voltage at t = 0.1 s
Maximum output current
Analog pins
Power dissipation at T
A
70 °C
Operating junction temperature range
Storage temperature range
Value
40
-1 to 40
-5 to 40
Int. limit.
4
1.2
-40 to 150
-55 to 150
Unit
V
V
V
V
W
°C
°C
Symbol
Table 3. Thermal data
Parameter
R thJA
Maximum thermal resistance junction ambient
1. Package mounted on evaluation board.
Value
65
(1)
Unit
°C/W
DocID026873 Rev 1 5/35
35
Electrical characteristics R5972D
T
J
= -40 °C to 125 °C, V
CC
= 12 V, unless otherwise specified.
Symbol
R
V
CC
DS(on) f
I
L
SW
V
OH
V
OL
I o source
Parameter
Table 4. Electrical characteristics
Test condition
Operating input voltage range
MOSFET on resistance
Maximum limiting current
Switching frequency
Duty cycle
High level output voltage
Low level output voltage
Source output current
(1)
Dynamic characteristics (see test circuit)
V
5
Voltage feedback
Efficiency
DC characteristics
I qop
I q
Total operating quiescent current
Quiescent current
Error amplifier
V
V
CC
V
CC
0
= 5 V
= 5 V, T
J
4.4 V < V
CC
= 5 V, V
CC
= 25 °C
< 36 V
= 12 V
Duty cycle = 0; V
FB
= 1.5 V
V
FB
= 1 V
V
FB
= 1.5 V
V
COMP
= 1.9 V; V
FB
= 1 V
Min.
4
I o sink
I b
Sink output current
Source bias current
V
COMP
= 1.9 V; V
FB
= 1.5 V 1 gm
DC open loop gain
Transconductance
R
L
=
I
COMP
= -0.1 mA to 0.1 mA;
V
COMP
= 1.9 V
50
1. With T
J
= 85 °C, I lim_min
= 2 A, assured by design, characterization and statistical correlation.
1.8
2
212
0
1.198
3.5
190
Typ.
Max.
Unit
0.250
2.5
2.5
250
1.235
90
3
300
1.5
2.5
65
2.3
36
0.5
3
3
280
100
1.272
5
2.5
0.4
4
V
A kHz
%
V
% mA mA
V
V
A mA
A dB mS
6/35 DocID026873 Rev 1
R5972D
4
Datasheet parameters over the temperature range
Datasheet parameters over the temperature range
The 100% of the population in the production flow is tested at three different ambient temperatures (-40 C, +25 C, +125 C) to guarantee the datasheet parameters inside the junction temperature range (-40 C, +125 C).
The device operation is so guaranteed when the junction temperature is inside the (-40 C,
+150 C) temperature range. The designer can estimate the silicon temperature increase respect to the ambient temperature evaluating the internal power losses generated during
the device operation (please refer to the Section 2.2
).
However the embedded thermal protection disables the switching activity to protect the device in case the junction temperature reaches the T
SHTDWN temperature.
(+150 C ± 10 C)
All the datasheet parameters can be guaranteed to a maximum junction temperature of
+125 C to avoid triggering the thermal shutdown protection during the testing phase because of self-heating.
DocID026873 Rev 1 7/35
35
Functional description R5972D
The main internal blocks are shown in the device block diagram in
A voltage monitor circuit which checks the input and the internal voltages.
A fully integrated sawtooth oscillator with a frequency of 250 kHz 15%, including also the voltage feed-forward function and an input/output synchronization pin.
Two embedded current limitation circuits which control the current that flows through the power switch. The pulse-by-pulse current limit forces the power switch OFF cycleby-cycle if the current reaches an internal threshold, while the frequency shifter reduces the switching frequency in order to significantly reduce the duty cycle.
A transconductance error amplifier.
A pulse width modulator (PWM) comparator and the relative logic circuitry necessary to drive the internal power.
A high-side driver for the internal P-MOS switch.
A circuit to implement the thermal protection function.
Figure 2. Block diagram
8/35
The internal regulator circuit (shown in Figure 3
) consists of a start-up circuit, an internal voltage pre-regulator, the bandgap voltage reference and the bias block that provides current to all the blocks. The starter supplies the start-up currents to the entire device when the input voltage goes high and the device is enabled (inhibit pin connected to ground). The pre-regulator block supplies the bandgap cell with a pre-regulated voltage V
REG a very low supply voltage noise sensitivity.
that has
DocID026873 Rev 1
R5972D Functional description
An internal block continuously senses the V cc
, V ref
. If the voltages go higher than their thresholds, the regulator begins operating. There is also a hysteresis on the V
CC
(UVLO).
and V bg
Figure 3. Internal circuit
The R5972D features two types of current limit protection: pulse-by-pulse and frequency foldback.
The schematic of the current limitation circuitry for the pulse-by-pulse protection is shown in
Figure 4 . The output power PDMOS transistor is split into two parallel PDMOS transistors.
The smallest one includes a resistor in series, R
SENSE
R
SENSE
. The current is sensed through
and if it reaches the threshold, the mirror becomes unbalanced and the PDMOS is switched off until the next falling edge of the internal clock pulse. Due to this reduction of the
ON time, the output voltage decreases. Since the minimum switch ON time necessary to sense the current in order to avoid a false overcurrent signal is too short to obtain a sufficiently low duty cycle at 250 kHz (see
Section 8.4 on page 23 ), the output current in
strong overcurrent or short-circuit conditions could be not properly limited. For this reason the switching frequency is also reduced, thus keeping the inductor current under its maximum threshold. The frequency shifter (
Figure 4 ) functions based on the feedback
voltage. As the feedback voltage decreases (due to the reduced duty cycle), the switching frequency decreases also.
DocID026873 Rev 1 9/35
35
Functional description
Figure 4. Current limitation circuitry
R5972D
5.5
10/35
The voltage error amplifier is the core of the loop regulation. It is a transconductance operational amplifier whose non inverting input is connected to the internal voltage reference (1.235 V), while the inverting input (FB) is connected to the external divider or directly to the output voltage. The output (COMP) is connected to the external compensation network. The uncompensated error amplifier has the following characteristics:
Table 5. Uncompensated error amplifier characteristics
Description Values
Transconductance
Low frequency gain
Minimum sink/source voltage
Output voltage swing
Input bias current
2300 µS
65 dB
1500 µA/300 µA
0.4 V/3.65 V
2.5 µA
The error amplifier output is compared to the oscillator sawtooth to perform PWM control.
PWM comparator and power stage
This block compares the oscillator sawtooth and the error amplifier output signals to generate the PWM signal for the driving stage.
The power stage is a highly critical block, as it functions to guarantee a correct turn ON and turn OFF of the PDMOS. The turn ON of the power element, or more accurately, the rise time of the current at turn ON, is a very critical parameter. At a first approach, it appears that the faster the rise time, the lower the turn ON losses.
However, there is a limit introduced by the recovery time of the recirculation diode.
In fact, when the current of the power element is equal to the inductor current, the diode turns OFF and the drain of the power is able to go high. But during its recovery time, the
DocID026873 Rev 1
R5972D Functional description diode can be considered a high value capacitor and this produces a very high peak current, responsible for numerous problems:
Spikes on the device supply voltage that cause oscillations (and thus noise) due to the board parasites.
Turn ON overcurrent leads to a decrease in the efficiency and system reliability.
Major EMI problems.
Shorter freewheeling diode life.
The fall time of the current during turn OFF is also critical, as it produces voltage spikes (due to the parasites elements of the board) that increase the voltage drop across the PDMOS.
In order to minimize these problems, a new driving circuit topology has been used and the block diagram is shown in
Figure 5 . The basic idea is to change the current levels used to
turn the power switch ON and OFF, based on the PDMOS and the gate clamp status.
This circuitry allows the power switch to be turned OFF and ON quickly and addresses the freewheeling diode recovery time problem. The gate clamp is necessary to ensure that V
GS of the internal switch does not go higher than V
GS max. The ON/OFF control block protects against any cross conduction between the supply line and ground.
Figure 5. Driving circuitry
The shutdown block generates a signal that turns OFF the power stage if the temperature of the chip goes higher than a fixed internal threshold (150 ± 10 °C). The sensing element of the chip is very close to the PDMOS area, ensuring fast and accurate temperature detection. A hysteresis of approximately 20 °C keeps the device from turning ON and OFF continuously.
DocID026873 Rev 1 11/35
35
Additional features and protection
6 Additional features and protection
R5972D
6.2
If the feedback is disconnected, the duty cycle increases towards the maximum allowed value, bringing the output voltage close to the input supply. This condition could destroy the load.
To avoid this hazardous condition, the device is turned OFF if the feedback pin is left floating.
Output overvoltage protection
Overvoltage protection, or OVP, is achieved by using an internal comparator connected to the feedback, which turns OFF the power stage when the OVP threshold is reached. This threshold is typically 30% higher than the feedback voltage.
When a voltage divider is required to adjust the output voltage (
OVP intervention will be set at:
Equation 1
V
OVP
= 1.3
R
1
+
R
2
R
------------------- V
FB
Where R
1
R
2
is the resistor connected between the output voltage and the feedback pin, and is between the feedback pin and ground.
Due to the fact that the internal power is a PDMOS, no boostrap capacitor is required and so the device works properly even with no load at the output. In this case it works in burst mode, with a random burst repetition rate.
12/35 DocID026873 Rev 1
R5972D
7 Closing the loop
Figure 6. Block diagram of the loop
Closing the loop
7.1 Error amplifier and compensation network
The output L-C filter of a step-down converter contributes with 180 degrees phase shift in the control loop. For this reason a compensation network between the COMP pin and
GROUND is added. The simplest compensation network together with the equivalent circuit
of the error amplifier are shown in Figure 7
. R
C
and C
C
introduce a pole and a zero in the open loop gain. CP does not significantly affect system stability but it is useful to reduce the noise of the COMP pin.
The transfer function of the error amplifier and its compensation network is:
Equation 2
A
0
=
2
A s
R
0
C
--------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------s R
0
C
0
+ C p
R c
C c
+
1 s
R
C c
+ R
0
C
0
+ C p
+ R c
C c
+ 1
Where A vo
= G m
· R o
DocID026873 Rev 1 13/35
35
Closing the loop
Figure 7. Error amplifier equivalent circuit and compensation network
R5972D
14/35
The poles of this transfer function are (if C c
>> C
0
+ C
P
):
Equation 3
F
P1
=
2
1
R
0
C c
Equation 4
F
P2
=
2 R c
1
C
0
+ C p
whereas the zero is defined as:
Equation 5
F
Z1
=
2 R c
C c
F
P1
is the low frequency which sets the bandwidth, while the zero F
Z1
is usually put near to the frequency of the double pole of the L-C filter (see
). F
P2 a very high frequency.
is usually at
DocID026873 Rev 1
R5972D Closing the loop
The transfer function of the L-C filter is given by:
Equation 6
A
LC
= s
2
L C
OUT
R
LOAD
LOAD
1 ESR C
+ s
OUT
ESR C
s
OUT
R
LOAD
+ L + R
LOAD where R
LOAD
is defined as the ratio between V
OUT
and I
OUT
.
If R
LOAD
>> ESR, the previous expression of A
LC
can be simplified and becomes:
Equation 7
A
LC s =
1 ESR C
s
2
+
OUT
s
----------------------------------------------------------------------------------------------
OUT
ESR C
OUT
+
The zero of this transfer function is given by:
Equation 8
F
O
=
2
1
ESR C
OUT
F
0
is the zero introduced by the ESR of the output capacitor and it is very important to increase the phase margin of the loop.
The poles of the transfer function can be calculated through the following expression:
Equation 9
=
– ESR COUT
2
– COUT
COUT
In the denominator of A
LC
the typical second order system equation can be recognized:
Equation 10 s
2
+ 2 n
s +
2 n
If the damping coefficient is very close to zero, the roots of the equation become a double root whose value is n
.
Similarly for A
LC
the poles can usually be defined as a double pole whose value is:
Equation 11
F
PLC
=
2
OUT
DocID026873 Rev 1 15/35
35
Closing the loop R5972D
The PWM gain is given by the following formula:
Equation 12
G
PWM
=
V cc
OSCMAX
– V
OSCMIN
where V
OSCMAX
is the maximum value of a sawtooth waveform and V
OSCMIN
is the minimum value. A voltage feed-forward is implemented to ensure a constant GPWM. This is obtained by generating a sawtooth waveform directly proportional to the input voltage V
CC
.
Equation 13
V
OSCMAX
– V
OSCMIN
=
CC
Where K is equal to 0.076. Therefore the PWM gain is also equal to:
Equation 14
G
PWM
=
1
K
= const
This means that even if the input voltage changes, the error amplifier does not change its value to keep the loop in regulation, thus ensuring a better line regulation and line transient response.
In summary, the open loop gain can be expressed as:
Equation 15
G s = G
PWM
R
1
R
2
R
2
A
O
A
LC
Example 1
Considering R
C
= 4.7 k , C
C
= 22 nF and C
P
= 220 pF, the poles and zeroes of A
0
are:
F
P1
= 9 Hz
F
P2
= 150 kHz
F
Z1
= 1.5 kHz
If L = 33 µH, C
OUT
= 100 µF and ESR = 80 m , the poles and zeroes of A
LC
become:
F
PLC
= 3.3 kHz
F
ZESR
= 19.89 kHz
Finally R
1
= 5.6 k and R
2
= 3.3 k .
The gain and phase bode diagrams are plotted respectively in
and
.
16/35 DocID026873 Rev 1
R5972D
Figure 8. Module plot
Closing the loop
Figure 9. Phase plot
The cut-off frequency and the phase margin are:
Equation 16
F
C
= 33KHz phase margin = 46°
DocID026873 Rev 1 17/35
35
Application information
Figure 10. Application schematic
R5972D
18/35
Input capacitor
The input capacitor must be able to support the maximum input operating voltage and the maximum RMS input current.
Since step-down converters draw current from the input in pulses, the input current is squared and the height of each pulse is equal to the output current. The input capacitor has to absorb all this switching current, which can be up to the load current divided by two (worst case, with duty cycle of 50%). For this reason, the quality of these capacitors has to be very high to minimize the power dissipation generated by the internal ESR, thereby improving system reliability and efficiency. The critical parameter is usually the RMS current rating, which must be higher than the RMS input current. The maximum RMS input current (flowing through the input capacitor) is:
Equation 17
I
RMS
= I
O
D –
2
+
D
2
-------
2
Where is the expected system efficiency, is equal to I
O
D is the duty cycle and I
O
is the output DC current. This function reaches its maximum value at D = 0.5 and the equivalent RMS current
divided by 2 (considering = 1). The maximum and minimum duty cycles are:
Equation 18
D
MAX
=
V
V
INMIN
+
–
V
V
F
SW
DocID026873 Rev 1
R5972D Application information and
Equation 19
D
MIN
=
V
V
INMAX
+
–
V
V
SW
I
Where V
F
RMS
is the freewheeling diode forward voltage and internal PDMOS. Considering the range D
MIN
going through the input capacitor.
to D
MAX
V
SW
the voltage drop across the
, it is possible to determine the max.
Capacitors that can be considered are:
Electrolytic capacitors:
These are widely used due to their low price and their availability in a wide range of
RMS current ratings.
The only drawback is that, considering ripple current rating requirements, they are physically larger than other capacitors.
Ceramic capacitors:
If available for the required value and voltage rating, these capacitors usually have a higher RMS current rating for a given physical dimension (due to very low ESR).
The drawback is the considerably high cost.
Tantalum capacitors:
Very good, small tantalum capacitors with very low ESR are becoming more available.
However, they can occasionally burn if subjected to very high current during charge.
Therefore, it is better to avoid this type of capacitor for the input filter of the device. They can, however, be subjected to high surge current when connected to the power supply.
Manufacturer
Table 6. List of ceramic capacitors for the R5972D
Series Capacitor value (µF) Rated voltage (V)
TAIYO YUDEN
MURATA
UMK325BJ106MM-T
GRM42-2 X7R 475K 50
10
4.7
50
50
High dv/dt voltage spikes on the input side can be critical for DC/DC converters. A good power layout and input voltage filtering help to minimize this issue. In addition to the above considerations, a 1 µF/50 V ceramic capacitor as close as possible to the VCC and GND pins is always suggested to adequately filter VCC spikes.
Output capacitor
The output capacitor is very important to meet the output voltage ripple requirement.
Using a small inductor value is useful to reduce the size of the choke but it increases the current ripple. So, to reduce the output voltage ripple, a low ESR capacitor is required.
Nevertheless, the ESR of the output capacitor introduces a zero in the open loop gain, which helps to increase the phase margin of the system. If the zero goes to a very high frequency, its effect is negligible. For this reason, ceramic capacitors and very low ESR capacitors in general should be avoided.
Tantalum and electrolytic capacitors are usually a good choice for this purpose. A list of
some tantalum capacitor manufacturers is provided in Table 7 .
DocID026873 Rev 1 19/35
35
Application information R5972D
Manufacturer
Sanyo POSCAP
(1)
AVX
KEMET
Sprague
Table 7. Output capacitor selection
Series
TAE
THB/C/E
TPS
T494/5
595D
Cap value ( µ F) Rated voltage (V)
100 to 470
100 to 470
100 to 470
100 to 470
220 to 390
4 to 16
4 to 16
4 to 35
4 to 20
4 to 20
ESR (m )
25 to 35
25 to 55
50 to 200
30 to 200
160 to 650
1. POSCAP capacitors have some characteristics which are very similar to tantalum.
Inductor
The inductor value is very important as it fixes the ripple current flowing through the output
I capacitor. The ripple current is usually fixed at 20 - 40% of I omax
O
, which is 0.3 - 0.6 A with max = 1.5 A. The approximate inductor value is obtained using the following formula:
Equation 20
L =
V
IN
–
V
I
OUT
T
ON where T
ON
V
OUT
is the ON time of the internal switch, given by D · T. For example, with
= 3.3 V, V
IN
= 12 V and I
O
= 0.45 A, the inductor value is about 22 µH. The peak current through the inductor is given by:
Equation 21
I
PK
= I
O
+
I
2 and it can be observed that if the inductor value decreases, the peak current (which must be lower than the current limit of the device) increases. So, when the peak current is fixed, a higher inductor value allows a higher value for the output current. In
some inductor manufacturers are listed.
Manufacturer
Coilcraft
Coiltronics
BI
Epcos
Wurth Elektronik
Table 8. Inductor selection
Series Inductor value ( µ H) Saturation current (A)
DO3316T
UP1B
HM76-3
B82476
74456115
15 to 33
22 to 33
15 to 33
15 to 33
15 to 33
2.0 to 3.0
2.0 to 2.4
2.5 to 3.3
2 to 3
2.5 to 3
20/35 DocID026873 Rev 1
R5972D Application information
The layout of switching DC-DC converters is very important to minimize noise and interference. Power-generating portions of the layout are the main cause of noise and so high switching current loop areas should be kept as small as possible and lead lengths as short as possible.
High impedance paths (in particular the feedback connections) are susceptible to interference, so they should be as far as possible from the high current paths. A layout
example is provided in Figure 11
.
The input and output loops are minimized to avoid radiation and high frequency resonance problems. The feedback pin connections to the external divider are very close to the device to avoid pickup noise.
Figure 11. Layout example
The dissipated power of the device is tied to three different sources:
Conduction losses due to the not insignificant R
DSON
, which are equal to:
Equation 22
P
ON
= R
DSON
I
OUT
2
D
Where D is the duty cycle of the application. Note that the duty cycle is theoretically given by the ratio between V
OUT
and V
IN
, but in practice it is substantially higher than this value to
DocID026873 Rev 1 21/35
35
Application information R5972D compensate for the losses in the overall application. For this reason, the switching losses related to the R
DSON
increase compared to an ideal case.
Switching losses due to turning ON and OFF. These are derived using the following equation:
Equation 23
P
SW
= V
IN
I
OUT
T
ON
+
2
T
F
SW
= V
IN
I
OUT
T
SW
F
SW
Where T
RISE
and T
FALL
represent the switching times of the power element that cause the switching losses when driving an inductive load (see
). T
SW
is the equivalent switching time.
Figure 12. Switching losses
22/35
Quiescent current losses.
Equation 24
P
Q
= V
IN
I
Q
Where I
Q
is the quiescent current.
Example 2
V
IN
= 12 V
V
OUT
= 3.3 V
I
OUT
= 1.5 A
R
DS(on)
has a typical value of 0.25 at 25 °C and increases up to a maximum value of 0.5. at
150 °C. We can consider a value of 0.4 .
T
SW
is approximately 70 ns.
I
Q
has a typical value of 2.5 mA at V
IN
= 12 V.
DocID026873 Rev 1
R5972D Application information
The overall losses are:
Equation 25
=
P
TOT
=
2
R
DSON
+
I
OUT
2
70
+
IN
10
– 9
I
OUT
250
T
SW
10
3
+
F
SW
+
V
IN
10
– 3
I
Q
=
0.615W
The junction temperature of device will be:
Equation 26
T
J
= T
A
+ Rth P
TOT
Where T
A ambient. is the ambient temperature and Rth
J-A
is the thermal resistance junction to
Considering that the device is mounted on the board with a good ground plane, that it has a thermal resistance junction to ambient (Rth
J-A temperature of about 70 °C:
) of about 65 °C/W, and an ambient
Equation 27
T
J
= + C
In overcurrent protection mode, when the peak current reaches the current limit, the device reduces the T
ON
down to its minimum value (approximately 250 nsec) and the switching frequency to approximately one third of its nominal value even when synchronized to an external signal (see
Section 5.3: Current protection on page 9 ). In these conditions, the duty
cycle is strongly reduced and, in most applications, this is enough to limit the current to ILIM.
In any event, in case of heavy short-circuit at the output (V
O application conditions (V
CC
= 0 V) and depending on the
value and parasitic effect of external components) the current peak could reach values higher than ILIM. This can be understood considering the inductor current ripple during the ON and OFF phases:
ON phase
Equation 28
I
L TON
=
V
IN
– V – DCR
L
L
+ R
DSON
I
T
ON
OFF phase
Equation 29
I
L TOFF
=
– V
D
+ V out
L
+ DCR
L
I
---------------------------------------------------------------
T
OFF
where V
D
is the voltage drop across the diode, DCR
L
is the series resistance of the inductor.
In short-circuit conditions V
OUT
is negligible so during T
OFF
the voltage across the inductor is very small as equal to the voltage drop across parasitic components (typically the DCR of the inductor and the V
FW
of the freewheeling diode) while during T
ON inductor is instead maximized as approximately equal to V
IN
.
the voltage applied the
DocID026873 Rev 1 23/35
35
Application information R5972D
So Equation 28 and Equation 29 in overcurrent conditions can be simplified to:
Equation 30
I
L TON
=
V
IN
– DCR
L
L
+ R
DSON
I
T
ON MIN
V
L
250ns considering T
ON
that has been already reduced to its minimum.
Equation 31
I
L TOFF
=
– V
D
+ V
L
+ DCR
L
I
SW
– V
D
+ V out
L
+ DCR
L
I
12 s considering that f
SW
has been already reduced to one third of the nominal.
In case a short-circuit at the output is applied and V
IN
= 12 V the inductor current is controlled in most of the applications (see
). When the application must sustain the short-circuit condition for an extended period, the external components (mainly the inductor and diode) must be selected based on this value.
In case the V
IN
is very high, it could occur that the ripple current during T does not compensate the current increase during T
ON example of a power-up phase with V
IN
= V
IN MAX
OFF
( Equation 30 ).
= 36 V where
IL TON
>
( Equation 31 )
IL TOFF
, so the current escalates and the balance between Equation 30 and Equation 31 occurs at a current slightly higher than the current limit. This must be taken into account in particular to avoid the risk of an abrupt inductor saturation.
Figure 13. Short-circuit current V
IN
= 12 V
24/35 DocID026873 Rev 1
R5972D
Figure 14. Short-circuit current V
IN
= 24 V
Application information
Figure 15. Short-circuit current V
IN
= 36 V
DocID026873 Rev 1 25/35
35
Application information
8.5 Positive buck-boost regulator
R5972D
The device can be used to implement a step-up/down converter with a positive output voltage.
The output voltage is given by:
Equation 32
V
OUT
= V
IN
where the ideal duty cycle D for the buck boost converter is:
Equation 33
D =
IN
V
OUT
+ V
OUT
However, due to power losses in the passive elements, the real duty cycle is always higher than this. The real value (that can be measured in the application) should be used in the following formulas.
The peak current flowing in the embedded switch is:
Equation 34
I
SW
=
I
LOAD
+
I
2
=
I
--------------+
V
IN f
SW while its average current is equal to:
Equation 35
I
SW
=
I
LOAD
This is due to the fact that the current flowing through the internal power switch is delivered to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
protection (see Table 4 on page 6 for details) while the average current must be lower than
the rated DC current of the device.
As a consequence, the maximum output current is:
Equation 36
I
OUT MAX
I
SW MAX
where I
SW MAX
represents the rated current of the device.
The current capability is reduced by the term (1 - D) and so, for example, with a duty cycle of
0.5, and considering an average current through the switch of 1.5 A, the maximum output current deliverable to the load is 0.75 A.
shows the schematic circuit of this topology for a 12 V output voltage and 5 V input.
26/35 DocID026873 Rev 1
R5972D
Figure 16. Positive buck-boost regulator
Application information
In
Figure 17 the schematic circuit for a standard buck-boost topology is shown. The output
voltage is:
Equation 37
V
OUT
= – V
IN
where the ideal duty cycle D for the buck boost converter is:
Equation 38
D =
V
–
IN
V
–
OUT
V
OUT
The considerations given in
for the real duty cycle are still valid here.
Also the Equation 34 till Equation 36 can be used to calculate the maximum output current.
So, as an example, considering the conversion V
IN
= 12 V to V
OUT
= -5 V, I
LOAD
= 0.2 A:
Equation 39
D =
5
= 0.706
Equation 40
I
SW
=
I
LOAD
=
0.2
= 0.7A
An important thing to take into account is that the ground pin of the device is connected to the negative output voltage. Therefore, the device is subjected to a voltage equal to
V
IN
- V
O
, which must be lower than 36 V (the maximum operating input voltage).
DocID026873 Rev 1 27/35
35
Application information
Figure 17. Negative buck-boost regulator
R5972D
28/35
MLCCs (multiple layer ceramic capacitor) with values in the range of 10 µF - 22 µF and rated voltages in the range of 10 V - 25 V are available today at relatively low cost from many manufacturers.
These capacitors have very low ESR values (a few m ) and thus are occasionally used for the output filter in order to reduce the voltage ripple and the overall size of the application.
However, a very low ESR value affects the compensation of the loop (see
13 ) and in order to keep the system stable, a more complicated compensation network may
be required. However, due to the architecture of the internal error amplifier the bandwidth with this compensation is limited.
That is why output capacitors with a not negligible ESR are suggested. The selection of the output capacitor have to guarantee that the zero introduced by this component is inside the designed system bandwidth and close to the frequency of the double pole introduced by the
LC filter. A general rule for the selection of this compound for the system stability is provided in Equation 41 .
Equation 41 f
Z ESR
=
2
1
f
LC
f
Z ESR
OUT
bandwidth
LC
shows an example of a compensation network stabilizing the system with ceramic capacitors at the output (the optimum component value depends on the application).
DocID026873 Rev 1
R5972D
Figure 18. MLCC compensation network example
Application information
DocID026873 Rev 1 29/35
35
Typical characteristics R5972D
Figure 19. Line regulation
Figure 21. Output voltage vs. junction temperature
Figure 20. Load regulation
Vo (V)
3.312
3.308
3.304
3.3
3.296
3.292
3.288
3.284
3.28
3.276
0
Vcc = 12V
Vo = 3.3V
Tj = 25°C
Tj = 125°C
0.5
Io (A)
1 1.5
Figure 22. Switching frequency vs. junction temperature
Figure 23. Quiescent current vs. junction temperature
Figure 24. Shutdown current vs. junction temperature
30/35 DocID026873 Rev 1
R5972D Typical characteristics
Figure 25. Junction temperature vs. output current - V
IN
= 5 V
Figure 26. Junction temperature vs. output current - V
IN
= 12 V
Figure 27. Efficiency vs output current -
V
IN
= 5 V
Figure 28. Efficiency vs output current -
V
IN
= 12 V
DocID026873 Rev 1 31/35
35
Package information R5972D
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK ® packages, depending on their level of environmental compliance. ECOPACK specifications, grade definitions and product status are available at: www.st.com
.
ECOPACK is an ST trademark.
32/35 DocID026873 Rev 1
R5972D
Figure 29. SO-8 package outline
Package information
Table 9. SO-8 package mechanical data
Dimensions (mm) Dimensions (inch)
Symbol
Min.
Typ.
A 1.35
Max.
Min.
Typ.
Max.
0.069
A1 0.10
A2 1.10
B 0.33
C 0.19
0.010
0.065
0.020
0.010
D (1)
E 3.80 e 1.27
H 5.80 h 0.25
0.157
0.050
0.244
0.020
L 0.40 k 0° (min.), 8° (max.)
0.050 ddd 0.10 0.004
1. Dimensions D does not include mold flash, protrusions or gate burrs. Mold flash, protrusions or gate burrs shall not exceed 0.15 mm (0.006 inch) in total (both sides).
DocID026873 Rev 1 33/35
35
Revision history
Date
01-Oct-2014
Table 10. Document revision history
Revision Changes
1 Initial release.
R5972D
34/35 DocID026873 Rev 1
R5972D
IMPORTANT NOTICE – PLEASE READ CAREFULLY
STMicroelectronics NV and its subsidiaries (“ST”) reserve the right to make changes, corrections, enhancements, modifications, and improvements to ST products and/or to this document at any time without notice. Purchasers should obtain the latest relevant information on
ST products before placing orders. ST products are sold pursuant to ST’s terms and conditions of sale in place at the time of order acknowledgement.
Purchasers are solely responsible for the choice, selection, and use of ST products and ST assumes no liability for application assistance or the design of Purchasers’ products.
No license, express or implied, to any intellectual property right is granted by ST herein.
Resale of ST products with provisions different from the information set forth herein shall void any warranty granted by ST for such product.
ST and the ST logo are trademarks of ST. All other product or service names are the property of their respective owners.
Information in this document supersedes and replaces information previously supplied in any prior versions of this document.
© 2014 STMicroelectronics – All rights reserved
DocID026873 Rev 1 35/35
35
Download
Public link updated
The public link to your chat has been updated. You may view and manage your chat links after registration here.
Document public link
Advertisement