STMicroelectronics LED5000 Datasheet
광고
광고
LED5000
3 A monolithic step-down current source with dimming capability
Datasheet - production data
Features
5.5 V to 48 V operating input voltage range
850 kHz fixed switching frequency
200 mV typ. current sense voltage drop
Buck / buck-boost / floating boost topologies
PWM dimming
± 3% output current accuracy overtemperature
200 mΩ typical R
DSON
Peak current mode architecture
Short-circuit protection
Compliant with ceramic output capacitors
Inhibit for zero current consumption
Thermal shutdown
Applications
High brightness LED driving
Street lighting
Signage
Halogen bulb replacement
General lighting
Description
The LED5000 device is an 850 kHz fixed switching frequency monolithic step-down DC-DC converter designed to operate as a precise constant current source with an adjustable current capability up to 3 A DC. The embedded
PWM dimming circuitry features LED brightness control. The regulated output current level is set by connecting a sensing resistor to the feedback pin. The 200 mV typical R
SENSE
voltage drop enhances performance in terms of efficiency. The size of the overall application is minimized thanks to the high switching frequency and its compatibility with ceramic output capacitors. The device is fully protected against overheating, overcurrent and output short-circuit. The
LED5000 is available in an HSOP8 package.
Figure 1: Typical application circuit
July 2017 DocID023951 Rev 6
This is information on a product in full production.
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Contents
Contents
LED5000
Pin connection ................................................................................... 4
2 Maximum ratings ............................................................................. 5
Maximum ratings ............................................................................... 5
Thermal data ..................................................................................... 5
ESD protection .................................................................................. 5
3 Electrical characteristics ................................................................ 6
4 Functional description .................................................................... 8
Power supply and voltage reference ................................................. 8
Voltage monitor ................................................................................. 9
Dimming block ................................................................................. 10
Thermal shutdown ........................................................................... 10
5 Application notes - buck conversion ........................................... 11
Closing the loop .............................................................................. 11
(s) control to output transfer function ........................................ 11
Error amplifier compensation network ............................................. 13
LED small signal model ................................................................... 14
Total loop gain ................................................................................. 16
Compensation network design ........................................................ 16
Example of system design .............................................................. 17
Dimming operation .......................................................................... 19
Dimming frequency vs. dimming depth ............................................ 21
Component selection ...................................................................... 23
Sensing resistor ................................................................................ 23
Inductor and output capacitor selection ............................................ 23
Layout considerations ..................................................................... 27
Thermal considerations ................................................................... 28
Short-circuit protection .................................................................... 30
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LED5000 Contents
Application circuit ............................................................................ 33
6 Application notes - alternative topologies ................................... 36
Inverting buck-boost ........................................................................ 36
Positive buck-boost ......................................................................... 40
Floating boost .................................................................................. 43
Compensation network design for alternative topologies ................ 48
7 Package information ..................................................................... 51
HSOP8 package information ........................................................... 51
8 Ordering information ..................................................................... 53
9 Revision history ............................................................................ 54
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Pin settings
1
1.1
Pin settings
Pin connection
Figure 2: Pin connection (top view)
LED5000
1.2 Pin description
Table 1: Pin description
Type Description
1 BOOT Analog circuitry power supply connection
2 DIM
Dimming control input. Logic low prevents the switching activity, logic high enables it. A square wave on this pin implements LEDs current PWM dimming. Connect to VIN if not
used (see Section 5.8: "Dimming operation" )
3 INH
5 FB
Inhibit pin. Connect to GND if not used
4 COMP Analog circuitry
Feedback input. Connect a proper sensing resistor to set the LED current
6 GND Ground connection
7 VIN
Power input voltage
8 SW
Switching node
- e.p.
Exposed pad to be connected to GND to increase the package thermal performance and the device noise immunity
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LED5000
2
2.1
2.2
2.3
Maximum ratings
Maximum ratings
Symbol
V
IN
V
INH
V
DIM
V
COMP
BOOT
SW
V
FB
T
J
T
STG
T
LEAD
Maximum ratings
Table 2: Absolute maximum ratings
Parameter
Power supply input voltage
Inhibit input
Dimming input
Comp output
Bootstrap pin
Switching node
Feedback voltage
Operating junction temperature range
Storage temperature range
Lead temperature (soldering 10 sec.)
Value
-0.3 to 52
-0.3 to 7
-0.3 to (V
IN
+ 0.3)
-0.3 to 3
-0.3 to 55
-1 to (V
IN
+ 0.3)
-0.3 to 3
-40 to 150
-65 to 150
260
Thermal data
Symbol
R
Table 3: Thermal data
Parameter
Thermal resistance junction ambient
Value
40
Notes:
(1) Device soldered to the STEVAL-ILL056V1 demonstration board.
ESD protection
Symbol
Table 4: ESD protection
Test condition
ESD
HBM
MM
Value
4
500
Unit
°C/W
Unit
KV
V
Unit
V
V
°C
°C
°C
V
V
V
V
V
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Electrical characteristics LED5000
3 Electrical characteristics
All tests performed at T
J
= 25 °C, V
CC
= 12 V, V
INH
= 0 V unless otherwise specified. The specification is guaranteed from -40 to +125 °C - T
J
temperature range by design, characterization and statistical correlation.
Table 5: Electrical characteristics
Test condition Symbol Parameter
V
IN
Operating input voltage range
R
DS(on)
MOSFET on resistance
I
SW
Maximum limiting current t
HICCUP
Hiccup time f
SW
Switching frequency
Duty cycle
I
SW
= 1 A
Min. Typ. Max. Unit
5.5 48
0.2 0.4
V
Ω
3.7 4.5 5.2 A
16 ms
600 850 1000 kHz
90 %
T
ON MIN
90 ns
T
OFF MIN
Minimum conduction time of the external
DC characteristics
V
FB
Voltage feedback
I
FB
FB biasing current
I q
Quiescent current
I qst-by
Standby quiescent current
Inhibit
V
DIM
> 1.5 V
V
DIM
> 1.5 V, V
IN
= 48 V
V
INH
> 1.5 V
75 90 120 ns
194 200 206 mV
50
1.3 2 nA mA
1.7 2.4 mA
12 16 34 µA
V
INH
Inhibit levels
I
INH
Dimming
Inhibit biasing current
Device ON
V
IN
= 5.5 V to 48 V
Device OFF
V
IN
= 5.5 V to 48 V
V
INH
= 5 V
0.5 V
1.5 V
0.7 1.6 2.5 µA
V
DIM
Dimming levels
Switching activity
V
IN
= 5.5 V to 48 V
Switching activity prevented
V
IN
= 5.5 V to 48 V
2.2
0.5
V
V
Error amplifier
V
OH
V
OL
High level output voltage
Low level output voltage
I o source
Source output current
V
FB
= 0 V
V
FB
= 400 mV
V
COMP
= 1.5 V; V
FB
= 0 V
3
16 23
V
150 mV
30 µA
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LED5000
Symbol Parameter
I o sink
Sink output current
I b
Source bias current
DC open loop gain gm Transconductance
Thermal shutdown
T
SHDWN
Thermal shutdown temperature
T
HYS
Thermal shutdown hysteresis
Notes:
(1) Parameter guaranteed by design.
Test condition
V
COMP
V
= 1.5 V; V
FB
= 0.4
V
FB
= 250 mV
RL = ∞
I
COMP
= TBD;
V
COMP
= TBD
Electrical characteristics
Min. Typ. Max. Unit
30 16 23
50
90
µA nA dB
220 µS
140 150 160 °C
15 °C
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Functional description
4
LED5000
Functional description
The LED5000 is based on a “peak current mode” architec ture with fixed frequency control.
As a consequence the intersection between the error amplifier output and the sensed inductor current generates the control signal to drive the power switch.
The main internal blocks shown in the block diagram in
Figure 3: "LED5000 block diagram"
are:
A fully integrated sawtooth oscillator with a typical frequency of 850 kHz
A transconductance error amplifier
A high side current sense amplifier to track the inductor current
A pulse width modulator (PWM) comparator and the circuitry necessary to drive the internal power element
The soft-start circuitry to decrease the inrush current at power-up
The dimming block to implement PWM dimming
The inhibit block for standby operation
The current limitation circuit based on the pulse-by-pulse current and the HICCUP protection
The bootstrap circuitry to drive the embedded NMOS switch
A circuit to implement the thermal protection function
Figure 3: LED5000 block diagram
4.1
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Power supply and voltage reference
The internal regulator circuit consists of a startup circuit, an internal voltage pre-regulator, the bandgap voltage reference and the bias block that provides current to all the blocks.
The starter supplies the startup current to the entire device when the input voltage goes high and the device is enabled (inhibit pin connected to ground). The pre-regulator block supplies the bandgap cell with a pre-regulated voltage that has a very low supply voltage noise sensitivity.
DocID023951 Rev 6
LED5000
4.2
Functional description
Voltage monitor
An internal block continuously senses the V cc
, V ref
and V bg
. If the monitored voltages are good, the regulator begins operating. There is also a hysteresis on the V cc
(UVLO).
Figure 4: Internal circuit
4.3 Soft-start
The startup phase is implemented ramping the reference of the embedded error amplifier in
1 msec typ. time. It minimizes the inrush current and decreases the stress of the power components at power up.
Figure 5: Soft-start open
During normal operation a new soft-start cycle takes place in case of:
thermal shutdown event
UVLO event
The soft-start is disabled during the dimming operation to maximize the dimming performance.
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Functional description
4.4 Dimming block
LED5000
The DIM input features the LED brightness control with the PWM dimming operation (see
Section 5.8: "Dimming operation"
4.5 Inhibit block
The inhibit block features the standby mode accordingly with
Table 5: "Electrical characteristics "
. The INH pin high level disables the device so the power consumption is
reduced to less than 40 µA. The INH pin is 5 V tolerant.
4.6
4.7
Error amplifier
The voltage error amplifier is the core of the loop regulation. It is a transconductance operational amplifier whose non inverting input is connected to the internal voltage reference (200 mV), while the inverting input (FB) is connected to the output current sensing resistor.
Table 6: Uncompensated error amplifier characteristics
Description Values
Transconductance 220 µS
Low frequency gain 90 dB
The error amplifier output is compared with the inductor current sense information to perform PWM control.
Thermal shutdown
The shutdown block generates a signal that disables the power stage if the temperature of the chip goes higher than a fixed internal threshold (150 ± 10 °C typical). The sensing element of the chip is close to the PDMOS area, ensuring fast and accurate temperature detection. A 15 °C typical hysteresis prevents the device from turning ON and OFF continuously during the protection operation.
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LED5000
5
5.1
Application notes
- buck conversion
Application notes - buck conversion
Closing the loop
Figure 6: Block diagram of the loop
5.2 G
CO
(s) control to output transfer function
The accurate control to output transfer function for a buck peak current mode converter can be written as:
Equation 1 where R
LOAD
represents the load resistance (see
Section 5.4: "LED small signal model"
R
CS
the equivalent sensing resistor of the current sense circuitry equal to 0.38, ω p
the single pole introduced by the LC filter and ω z capacitor.
the zero given by the ESR of the output
F
H
(s) accounts the sampling effect performed by the PWM comparator on the output of the error amplifier that introduces a double pole at one half of the switching frequency.
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Application notes
- buck conversion
Equation 2 where ESR is the equivalent series resistor to the output capacitor.
Equation 3 where:
Equation
4
LED5000
12/56
S n
represents the slope of the sensed inductor current, S e
the slope of the external ramp
(V
PP
peak to peak amplitude equal to 1.2 V) that implements the slope compensation to avoid sub-harmonic oscillations at duty cycle over 50%.
The sampling effect contribution F
H
(s) is:
Equation 5 where:
Equation 6 and
DocID023951 Rev 6
LED5000
Equation 7
5.3
Application notes
- buck conversion
Error amplifier compensation network
The external compensation network connected at the output of the error amplifier is dimensioned to stabilize the system depending on the application conditions.
Figure 7: Transconductance embedded error amplifier
R
C
and C
C
introduce a pole and a zero in the open loop gain. C
P
does not significantly affect system stability but it can be useful to reduce the noise at the output of the error amplifier.
The transfer function of the error amplifier and its compensation network is:
Equation 8
Where A
V0
= G m
· R
0
(R
O
= output resistor of OTA = 200 * 10 ^ 6 Ω).
The poles of this transfer function are (if C c
>> C
0
+ C
P
):
Equation 9
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Application notes
- buck conversion
Equation 10
5.4
LED5000 whereas the zero is defined as:
Equation 11
LED small signal model
Once the system reaches the working condition the LEDs composing the row are biased and their equivalent circuit can be considered as a resistor for frequencies << 1 MHz.
The LED manufacturer typically provides the equivalent dynamic resistance of the LED biased at a different DC current. This parameter is required to study the behavior of the system in the small signal analysis.
For instance, the equivalent dynamic resistance of Luxeon III Star from Lumiled measured with different biasing current levels is reported below:
In case the LED datasheet does not provide the equivalent resistor value, it can be easily derived as the tangent to the diode I - V characteristic in the present working point (see
Figure 8: "Equivalent series resistor"
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LED5000 Application notes
- buck conversion
Figure 8: Equivalent series resistor
Figure 9: "Load equivalent circuit"
shows the equivalent circuit of the LED constant current
generator.
The equivalent loading resistor in the LEDs working point is:
Equation 12 where R
S
is the resistor put in series to the LED string.
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Application notes
- buck conversion
Figure 9: Load equivalent circuit
LED5000
5.5
5.6
16/56
As a consequence the LED equivalent circuit gives the α
LED
(s) term correlating the output voltage with the high impedance FB input:
Equation 13
Total loop gain
In summary, the open loop gain can be expressed as:
Equation 14
Compensation network design
The maximum bandwidth of the system can be designed up to f
SW
/6 to guarantee a valid small signal model.
Equation 15
DocID023951 Rev 6
LED5000 Application notes
- buck conversion where ω
P
) is the pole introduced by the power components. The following
calculations are valid in the hypothesis that BW > ω
P
which is the typical condition.
With the power components selected in accordance with
Section 5.9: "Component selection"
and given the BW specification, the components composing the compensation
network can be calculated as:
Equation 16
5.7 where the term m
C
LOAD
the equivalent loading resistor
S
the resistor put in series to the LED string, G m
the error amplifier transconductance and R
CS
the equivalent sensing resistor of the current sense circuitry
Table 5: "Electrical characteristics "
Equation 17 where K represents the leading position of the F
Z
bandwidth. In general, a value of 2 gives enough phase margin to the overall small loop transfer function.
Example of system design
Design specification:
V
IN
= 48 V, V
FW_LED
= 3.7 V, n
LED
= 10, r
LED
= 1.1 Ω, I
LED
= 1 A, I
LED RIPPLE
= 2%.
The inductor and capacitor value are dimensioned to meet the I
LED RIPPLE
specification
(see
Section 5.9.2: "Inductor and output capacitor selection"
for output capacitor and inductor
selection guidelines):
L = 22 µH, C
OUT
= 1.0 µF mlcc (negligible ESR).
Section 5.9.1: "Sensing resistor"
the sensing resistor value is:
Equation 18
Assuming a system bandwidth of:
Equation 19
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Application notes
- buck conversion
The ideal values of the components making up the compensation network are:
Equation 20
LED5000
Final component selection is based on commercial values and a small capacitor C
P
is added to reduce noise at the error amplifier output. C
P
slightly decreases the BW and phase margin.
Equation 21
The gain and phase margin bode diagrams are plotted, respectively, in
Figure 10: Module plot
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LED5000 Application notes
- buck conversion
Figure 11: Phase plot
5.8
The cut-off frequency and the phase margin are:
Equation 22
Dimming operation
The dimming input disables the switching activity, masking the PWM comparator output.
The inductor current dynamic performance when dimming input goes high depends on the designed system response. The best dimming performance is obtained by maximizing the bandwidth and phase margin, when possible.
As a general rule, the output capacitor minimization improves dimming performance.
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Application notes
- buck conversion
Figure 12: Dimming operation example
LED5000
In fact, when dimming enables the switching activity, a small capacitor value is fast charged with low inductor value. As a consequence, the LEDs current rising edge time is improved and the inductor current oscillation reduced. An oversized output capacitor value requires extra current for fast charge so generating an inductor current overshoot and oscillations.
The switching activity is prevented as soon as the dimming signal goes low. Nevertheless, the LED current drops to zero only when the voltage stored in the output capacitor goes below a minimum voltage determined by the selected LEDs. As a consequence, a big capacitor value makes the LED current falling time worse than a smaller one.
The LED5000 device embeds dedicated circuitry to improve LED current rising edge time.
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LED5000 Application notes
- buck conversion
Figure 13: LED rising edge operation
Figure 14: LED rising edge operation (zoom)
5.8.1 Dimming frequency vs. dimming depth
Section 5.8: "Dimming operation"
the LEDs current rising and falling edge time
mainly depends on the system bandwidth (T
RISE
) and the selected output capacitor value
(T
RISE
and T
FALL
).
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Application notes
- buck conversion
LED5000
The dimming performance depends on the minimum current pulse shape specification of the final application. The ideal minimum current pulse has rectangular shape, in any case it degenerates into a trapezoid or, at worst, into a triangle, depending on the ratio (T
RISE
+
T
FALL
)/ T
DIM
.
Equation 23
Figure 14: "LED rising edge operation (zoom)"
as an example.
Equation 24
Assuming the minimum current pulse (T
MIN_PULSE
) shape specification as:
Equation 25 where T
DIMMING
represents the dimming period and D
MIN
the minimum duty cycle which gives the T
MIN_PULSE
charge. In the given example T
MIN_PULSE
= 9 µs.
Figure 15: dimming signal
22/56
Given T
MIN_PULSE it is possible to calculate the maximum dimming depth given the dimming frequency or vice versa.
For example, assuming a 10 KHz dimming frequency the maximum dimming depth is 9% or given a 5% dimming depth it follows a 5.5 KHz maximum f
DIM
.
The LED5000 dimming performance is strictly dependent on the system small signal response. As a consequence, an optimized compensation network (good phase margin
DocID023951 Rev 6
LED5000 Application notes
- buck conversion and bandwidth maximized) and minimized C
OUT
value are crucial for best performance.
Once the external power components and the compensation network are selected, a direct measurement to determine T
RISE
, T
FALL
achieved dimming performance.
5.9
5.9.1
Component selection
Sensing resistor
In closed loop operation the LED5000 feedback pin voltage is 200 mV, so the sensing resistor calculation is expressed as:
Equation 26
5.9.2
Section 5.1: "Closing the loop"
) regulates the sensing resistor
voltage drop, the average current is regulated into the LEDs. The integration period is at minimum
5 * T
SW
since the system bandwidth can be dimensioned up to f
SW
/5 at maximum.
A system loop based on a peak current mode architecture features consistent advantages in comparison with simpler closed loop regulation schemes like the hysteretic or the constant ON/OFF control.
The system performs the output current regulation over a period which is at least five times longer than the switching frequency. The output current regulation neglects the ripple current contribution and its reliance on external parameters like input voltage and output voltage variations (line transient and LED forward voltage spread). This performance cannot be achieved with simpler regulation loops like hysteretic control.
For the same reason, the switching frequency is constant over the application conditions, that helps to tune the EMI filtering and to guarantee the maximum LED current ripple specification in the application range. This performance cannot be achieved using constant
ON/OFF time architectures.
Inductor and output capacitor selection
The output capacitor filters the inductor current ripple that, given the application condition, depends on the inductor value. As a consequence the LED current ripple, that is the main specification for a switching current source, depends on the inductor and output capacitor selection.
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Application notes
- buck conversion
Figure 16: Equivalent circuit
LED5000
The LED ripple current can be calculated as the inductor ripple current ratio flowing into the
output impedance using the Laplace transform (see
Equation 27
24/56 where the term 8/π 2 represents the main harmonic of the inductor current ripple (which has a triangular shape) and ∆I
L
is the inductor current ripple.
Equation 28 so L value can be calculated as:
Equation 29 where T
OFF
is the OFF time of the embedded high switch, given by 1 - D.
As a consequence the lower is the inductor value (so higher the current ripple), the higher would be the C
OUT
value to meet the specification.
A general rule to dimension L value is:
Equation 30
DocID023951 Rev 6
LED5000 Application notes
- buck conversion
Finally the required output capacitor value can be calculated equalizing the LED current
ripple specification with the module of the Fourier transformer (see
at f
SW
frequency.
Equation 31
Example 1
Section 5.6: "Compensation network design"
V
IN
= 48 V, I
LED
= 700 mA, Δ
ILED
/I
LED
= 2%, V
FW_LED
= 3.7 V, n
LED
= 10.
A lower inductor value maximizes the inductor current slew rate for better dimming
Equation 32
5.9.3
which is satisfied selecting a 10 µH inductor value.
The output capacitor value has to be dimensioned according to
Finally, given the selected inductor value, a 1 µF ceramic capacitor value kesvg the LED current ripple ratio lower than the 2% of the nominal current. An output ceramic capacitor type (negligible ESR) is suggested to minimize the ripple contribution given a fixed capacitor value.
Manufacturer
Wurth Elektronik
Coilcraft
Table 7: Inductor selection
Series
WE-HCI 7040
WE-HCI 7050
XPL 7030
Inductor value (µH)
1 to 4.7
4.9 to 10
2.2 to 10
Saturation current (A)
20 to 7
20 to 4.0
29 to 7.2
Input capacitor
The input capacitor must be able to support the maximum input operating voltage and the maximum RMS input current.
Since step-down converters draw current from the input in pulses, the input current is squared and the height of each pulse is equal to the output current. The input capacitor has to absorb all this switching current, whose RMS value can be up to the load current divided by two (worst case, with duty cycle of 50%). For this reason, the quality of these capacitors has to be very high to minimize the power dissipation generated by the internal ESR, thereby improving system reliability and efficiency. The critical parameter is usually the
RMS current rating, which must be higher than the RMS current flowing through the capacitor. The maximum RMS input current (flowing through the input capacitor) is:
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Application notes
- buck conversion
Equation 33
LED5000
Where η is the expected system efficiency, D is the duty cycle and I
O
is the output DC current. Considering η = 1 this function reaches its maximum valu e at D = 0.5 and the equivalent RMS current is equal to I
O
divided by 2. The maximum and minimum duty cycles are:
Equation 34 and
Equation 35
Where V
F
is the freewheeling diode forward voltage and V
SW
the voltage drop across the internal PDMOS. Considering the range D
MIN
to D
MAX
, it is possible to determine the max.
I
RMS
going through the input capacitor.
Capacitors that can be considered are:
Electrolytic capacitors:
These are widely used due to their low price and their availability in a wide range of RMS current ratings.
The only drawback is that, considering ripple current rating requirements, they are physically larger than other capacitors.
Ceramic capacitors:
If available for the required value and voltage rating, these capacitors usually have a higher
RMS current rating for a given physical dimension (due to very low ESR).
The drawback is the considerably high cost.
Tantalum capacitors:
Small tantalum capacitors with very low ESR are becoming more available. However, they can occasionally burn if subjected to very high current during charge.
Therefore, it is suggested to avoid this type of capacitor for the input filter of the device as they could be stressed by a high surge current when connected to the power supply.
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LED5000 Application notes
- buck conversion
Table 8: List of ceramic capacitors for the LED5000
Manufacturer Series Capacitor value (µF) Rated voltage (V)
Taiyo Yuden UMK325BJ106MM-T 10 50
Murata GRM42-2 X7R 475K 50 4.7 50
In case the selected capacitor is ceramic (so neglecting the ESR contribution), the input voltage ripple can be calculated as:
Equation 36
5.10 Layout considerations
The layout of switching DC-DC converters is very important to minimize noise and interference. Power-generating portions of the layout are the main cause of noise and so high switching current loop areas should be kept as small as possible and lead lengths as short as possible.
High impedance paths (in particular the feedback connections) are susceptible to interference, so they should be as far as possible from the high current paths. A layout
The input and output loops are minimized to avoid radiation and high frequency resonance problems. The feedback pin to the sensing resistor path must be designed as short as possible to avoid pick-up noise. Another important issue is the ground plane of the board.
Since the package has an exposed pad, it is very important to connect it to an extended ground plane in order to reduce the thermal resistance junction to ambient and increase the noise immunity during the switching operation.
In addition, to increase the design noise immunity, different signal and power grounds
should be designed in the layout (see
Section 5.13: "Application circuit"
serves the small signal components, the device analog ground pin, the exposed pad and a small filtering capacitor connected to the VCC pin. The power ground serves the device ground pin and the input filter. The different grounds are connected underneath the output capacitor. Neglecting the current ripple contribution, the current flowing through this component is constant during the switching activity and so this is the cleanest ground point of the buck application circuit.
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Application notes
- buck conversion
Figure 17: Layout example
LED5000
5.11 Thermal considerations
The dissipated power of the device is tied to three different sources:
Conduction losses due to the R
DSON
, which are equal to:
Equation 37
Where D is the duty cycle of the application. Note that the duty cycle is theoretically given by the ratio between V
OUT
(n
LED
* V
LED
+ 200 mV) and V
IN
, but in practice it is substantially higher than this value to compensate for the losses in the overall application.
For this reason, the conduction losses related to the R
DSON
increase compared to an ideal case.
Switching losses due to turning ON and OFF. These are derived using
Equation 38
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LED5000 Application notes
- buck conversion
Where T
RISE
and T
FALL
represent the switching times of the power element that cause the
switching losses when driving an inductive load (see
SW
is the equivalent switching time.
Figure 18: Switching losses
Quiescent current losses.
Equation 39
Example 2
Section 6.4: "Compensation network design for alternative topologies"
V
IN
= 42 V, V
FW_LED
= 3.7 V, n
LED
= 8, I
LED
= 1500 mA.
The typical output voltage is:
Equation 40
R
DSON_HS
has a typical value of 200 Ω at 25 °C.
For the calculation we can estimate R
DSON_HS
= 300 mΩ as a consequence of T
J
increase during the operation.
T
SW_EQ
is approximately 12 ns.
I
Q
has a typical value of 2.4 mA at V
IN
= 48 V.
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Application notes
- buck conversion
The overall internal losses are:
Equation 41
LED5000 where T
SW_EQ
= (T
RISE
+ T
FALL
)/2 = 12 nS.
Equation 42
The junction temperature of the device will be:
Equation 43
Where T
A
is the ambient temperature and Rth
J-A
is the thermal resistance junction to ambient. The junction to ambient (Rth
J-A
) thermal resistance of the device assembled in the
HSOP8 package and mounted on the evaluation is about 40 °C/W.
Assuming the ambient temperature around 40 °C, the estimated junction temperature is:
5.12 Short-circuit protection
In overcurrent protection mode, when the peak current reaches the current limit threshold, the device disables the power element and it is able to reduce the conduction time down to the minimum value (approximately 100 nsec typical) to keep the inductor current limited.
This is the pulse-by-pulse current limitation to implement constant current protection feature.
In overcurrent condition, the duty cycle is strongly reduced and, in most applications, this is enough to limit the switch current to the current threshold.
The inductor current ripple during ON and OFF phases can be written as:
ON phase
Equation 44
30/56 DocID023951 Rev 6
LED5000
OFF phase
Equation 45
Application notes
- buck conversion where DCR
L
is the series resistance of the inductor and V
FWDIODE
is the forward voltage drop across the external rectifying diode.
The pulse-by-pulse current limitation is effective to implement constant current protection when:
Equation 46
we can gather that the implementation of the constant
current protection becomes more critical the lower is the V
OUT
and the higher is V
IN
.
In fact, in short-circuit condition the voltage applied to the inductor during the OFF time becomes equal to the voltage drop across parasitic components (typically the DCR of the inductor and the forward voltage of the diode) since V
OUT
is negligible, while during T
ON
the voltage applied the inductor is maximized and it is approximately equal to V
IN
.
In general the worst case scenario is a heavy short-circuit at the output with maximum input
in overcurrent conditions can be simplified to:
Equation 47 considering T
ON
that has been already reduced to its minimum.
Equation 48 where T
SW
= 1/f
SW
considering the nominal f
SW
.
At high input voltage ΔI
L TON
could be higher than ΔI
L TOFF
and so the inductor current could
escalate. As a consequence, the system typically meets
higher than the nominal value thanks to the increased voltage drop across stray components. In most application conditions the pulse-by-pulse current limitation is effective to limit the inductor current. Whenever the current escalates, a second level current protection called “hiccup mode” is enab led. The hiccup protection offers an additional protection against heavy short-circuit condition at very high input voltage even considering
DocID023951 Rev 6 31/56
Application notes
- buck conversion
LED5000 the spread of the minimum conduction time of the power element. In case the hiccup current level (6.2 A typical) is triggered the switching activity is prevented for 16 msec typ.
Table 5: "Electrical characteristics "
Figure 19: "Constant current protection triggering hiccup mode"
constant current protection when a short-circuit is applied at the output at the maximum input voltage.
Figure 19: Constant current protection triggering hiccup mode
32/56 DocID023951 Rev 6
LED5000
5.13 Application circuit
Application notes
- buck conversion
Figure 20: Evaluation board application circuit
The network D3, R4, RS implements an inexpensive overvoltage protection. R4 effect can be neglected during normal operation since the FB biasing current is negligible (tens of nA,
Table 5: "Electrical characteristics "
) but it limits the current flowing in the Zener diode
D3. In case the load is disconnected or in case of open LED:
Equation 49
R1 must be dimensioned to limit the D1 rated power so it is an inexpensive small signal
Zener diode.
The overvoltage limits the output voltage in case of LED disconnection so protecting LEDs when the string is reconnected with the device enabled. In case the OVP is not implemented, a large amount of non-controlled current could flow through the LEDs during the output capacitor discharging phase, thereby damaging the devices.
DocID023951 Rev 6 33/56
Application notes
- buck conversion
Reference
C1, C2
C3, C6
C4
C5
C7
C8
D1
D2
D3
Table 9: Component list
Part number
C3225X7S1H106M
C3216X7R1H105K
Description
10 µF 50 V (size 1210)
100 nF 50 V (size 0805)
470 pF 50 V (size 00603)
22 pF 50 V (size 0603)
1 µF 50 V (size 1206)
Not mounted
3 A 60 V STPS3L60U
BZX384-C4V7
BZX384-C39
L1
R
S
R1
R2
R3
R4
XFL6060-473ME
ERJ14BSFR27U
47 µH
I
I
SAT
= 1.8 A (10% drop)
RMS
= 3.7 A (40 °C rise)
(size 7.3 x 7.3 x 4.1 mm)
0.27 Ω 1%
(size 1206)
10 kΩ 1% (size 0603)
Not mounted
24.9 kΩ 1% (size 0603)
47 kΩ 1% (size 0603)
Figure 21: PCB layout (component side)
LED5000
Manufacturer
TDK
TDK
ST
Coilcraft
Panasonic
34/56 DocID023951 Rev 6
LED5000 Application notes
- buck conversion
Figure 22: PCB layout (bottom side)
DocID023951 Rev 6 35/56
Application notes
- alternative topologies
6
6.1
LED5000
Application notes - alternative topologies
Thanks to the wide input voltage range, the adjustable external compensation network and enhanced dimming capability, the LED5000 device is suitable to implement boost and buck-boost topologies.
Inverting buck-boost
The buck-boost topology fits the application with an input voltage range that overlaps the output voltage, which is the voltage drop across the LEDs and the sensing resistor.
Figure 23: "Inverting buck-boost"
component count as the buck conversion and it is more efficient than the positive buckboost. A current generator based on this topology implies two main application constraints:
the output voltage is negative so the LEDs must be reversed
the device GND floats with the negative output voltage. The device is supplied between V
IN
and V
OUT
(< 0). As a consequence:
Equation 50 so:
Equation 51 where V
OUT
< 0.
Figure 23: Inverting buck-boost
36/56
Example 3
V
IN RANGE
= 12 - 24 V, V
FW_LED
= 3.7 V, n
LED
= 5 so V
OUT
= 18.7 V.
DocID023951 Rev 6
LED5000 Application notes
- alternative topologies
Since the maximum operating voltage of the LED5000 is 48 V, according to
the maximum input voltage of the application is 48 - 18.7 = 29.3 V.
The output voltage is given by:
Equation 52 where the ideal duty cycle D
IDEAL
for the buck-boost converter is:
Equation 53
However, due to power losses (mainly switching and conduction losses), the real duty cycle is always higher than this. The real value (which can be measured in the application) should be used in the following formulas.
The peak current flowing in the embedded switch is:
Equation 54 while its average current level is equal to:
Equation 55
This is due to the fact that the current flowing through the internal power switch is delivered to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
Table 5: "Electrical characteristics "
for details) while the average current
must be lower than the rated DC current of the device.
As a consequence, the maximum output current is:
Equation 56
DocID023951 Rev 6 37/56
Application notes
- alternative topologies where I
SW MAX
represents the rated current of the device.
LED5000
The current capability is reduced by the term (1 - D
REAL
) and so, for example, with a duty cycle of 0.5, and considering an average current through the switch of 3 A, the maximum output current deliverable to the load is 1.5 A.
Figure 24: "LED current source based on inverting BB topology"
circuit for an LED current source based on inverting buck-boost topology. The input voltage ranges from 10 to 26 V and it can drive a string composed of 5 LEDs with1 A DC.
Figure 24: LED current source based on inverting BB topology
The circuitry Q1, R2, R3, R4 implements a level shifter to convert the dimming signal voltage levels (referred to GND) to the device rails, since the LED5000 local ground is referred to the negative output voltage (given by the voltage drop across the LEDs and the
Figure 25: "Inverting BB dimming operation"
operation: the light blue trace represents the DIM pin, the yellow the SW (high level is V
IN
, low level is -V
OUT
), the green trace the inductor current (see
the output voltage.
38/56 DocID023951 Rev 6
LED5000 Application notes
- alternative topologies
Figure 25: Inverting BB dimming operation
The network D1, R1, RS implements an inexpensive overvoltage protection. R1 effect can be neglected during normal operation since the FB biasing current is negligible (tens of nA,
Table 5: "Electrical characteristics "
) but it limits the current flowing in the Zener diode
D1. In case the load is disconnected or in case of an open LED:
Equation 57
R1 must be dimensioned to limit the D1 rated power so it is an inexpensive small signal
Zener diode.
The overvoltage protection plays an important role for the inverting buck-boost topology. In fact, in case of open row, the output voltage tends to diverge thus exceeding the input
voltage absolute maximum rate and the device would be damaged (see
overvoltage protection limits V
OUT
and thereby it protects the device in case of load disconnection.
To design the compensation network for the inverting buck-boost topology please refer to
Section 5.6: "Compensation network design"
DocID023951 Rev 6 39/56
Application notes
- alternative topologies
Figure 26: Inverting BB PCB layout (component side)
LED5000
Figure 27: Inverting BB PCB layout (bottom side)
6.2 Positive buck-boost
Positive buck-boost fits those applications that require a buck-boost topology (i.e.: the input voltage range crosses the output voltage value) and where the inverting buck-boost is not
suitable because of the main constraints for the final application (refer to
As a consequence the inverting buck-boost is the preferred option because it requires less components and it has higher efficiency compared to the positive buck-boost topology.
40/56 DocID023951 Rev 6
LED5000 Application notes
- alternative topologies
Figure 28: Positive buck-boost
The positive buck-boost implementation (
Figure 28: "Positive buck-boost"
more diode and an external power switch than inverting buck-boost. The device is not floating, referred to GND, and it is supplied with the input voltage of the application (the input voltage in inverting buck-boost topology is instead V
IN
- V
OUT, r
for details). The LED5000 device does not see the output voltage
during the switching activity so V
OUT
can be higher than the maximum input voltage.
The equations for the positive buck-boost are similar to those seen for the inverting.
Equation 58 where the ideal duty cycle D
IDEAL
for the buck-boost converter is:
Equation 59
However, due to power losses (mainly switching and conduction losses), the real duty cycle is always higher than this. The real value (which can be measured in the application) should be used in the following formulas.
The peak current flowing in the embedded switch is:
Equation 60 while its average current level is equal to:
DocID023951 Rev 6 41/56
Application notes
- alternative topologies
Equation 61
LED5000
This is due to the fact that the current flowing through the internal power switch is delivered to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
Table 5: "Electrical characteristics "
for details) while the average current
must be lower than the rated DC current of the device.
As a consequence, the maximum output current is:
Equation 62 where I
SW MAX
represents the rated current of the device.
The current capability is reduced by the term (1 - D
REAL
) and so, for example, with a duty cycle of 0.5, and considering an average current through the switch of 3 A, the maximum output current deliverable to the load is 1.5 A.
Figure 29: "LED current source based on positive BB + topology"
schematic for an LED current source based on positive buck-boost topology. The input voltage ranges from 18 to 30 V and it can drive a string composed of 7 LEDs with 0.7 A DC
(V
FW_LED
= 3.75 V so V
OUT
= 26.4 V).
The network D5, R4, R
S
implements an inexpensive overvoltage protection. R4 effect can be neglected during the normal operation since the FB biasing current is negligible (tens of
Table 5: "Electrical characteristics "
) but it limits the current flowing in the Zener
diode D5. In case the load is disconnected or in case of open LED:
Equation 63
R4 must be dimensioned to limit the D5 rated power so it is an inexpensive small signal
Zener diode.
42/56 DocID023951 Rev 6
LED5000 Application notes
- alternative topologies
Figure 29: LED current source based on positive BB + topology
6.3
In case of open row, the positive output voltage tends to diverge, exceeding the D3 maximum reverse voltage and so the diode would be damaged. The overvoltage protection limits V
OUT
and it protects the power components in case of load disconnection.
The network D4, R8 implements a level shifter to drive the gate of the transistor Q1. The voltage at Q1 is:
Equation 64
Considering the V
IN
range 18 to 30 V:
Equation 65
The gate is driven inside the component specification. R8 can be dimensioned to discharge the gate when V
SW
is low.
In case the input voltage range of the application is not suitable to implement a level shifter to drive Q1, a dissipative clamping network (like R5, D6) must be used.
To design the compensation network for the positive buck-boost topology please refer to
Section 6.4: "Compensation network design for alternative topologies"
Floating boost
The floating boost topology (see ) serves those applications with an input voltage range narrower than the output voltage, that is the voltage drop across the LEDs and the sensing
DocID023951 Rev 6 43/56
Application notes
- alternative topologies
LED5000 resistor (i.e.: V
IN
< V
OUT
). The topology is called floating since the output voltage is referred to V
IN
and not GND, but this is typically suitable for a floating load like a string of LEDs.
Figure 30: Floating boost
44/56
The device is supplied by the output voltage so the maximum voltage drop across the
LEDs string is 48 V. The direct path of the boost conversion (C
OUT
, V
DIODE
, L) guarantees the proper startup when the input voltage is:
Equation 66 where V
OP_MIN
is the minimum operating voltage.
The equations for the floating boost are:
Equation 67
The ideal duty cycle D
IDEAL
for the boost converter is:
Equation 68
As seen for the buck-boost topologies (
Section 6.1: "Inverting buck-boost"
Section
"@NA short description"
), due to power losses the real duty cycle is always higher than the ideal. The real value (that can be measured in the application) should be used in the following formulas to estimate the switch current.
The peak current flowing in the embedded switch is:
DocID023951 Rev 6
LED5000
Equation 69
Application notes
- alternative topologies while its average current level is equal to:
Equation 70
This is due to the fact that the current flowing through the internal power switch is delivered to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
Table 5: "Electrical characteristics "
for details) while the average current
must be lower than the rated DC current of the device.
As a consequence, the maximum output current depends on the application conditions:
Equation 71 where I
SW MAX
represents the rated current of the device.
The current capability is reduced by the term (1-D
REAL
) and so, for example, with a duty cycle of 0.5, and considering an average current through the switch of 3 A, the maximum output current deliverable to the load is 1.5 A.
Figure 31: "LED current source based on floating boost topology"
schematic for an LED current source based on the floating boost topology. The input voltage ranges from 12 to 36 V and it can drive a string composed of 11 LEDs with 0.7 A
DC (V
FW_LED
= 3.74 V so V
OUT
= 41 V).
DocID023951 Rev 6 45/56
Application notes
- alternative topologies
Figure 31: LED current source based on floating boost topology
LED5000
The network D1, R1, R
S
implements an inexpensive overvoltage protection. R1 effect can be neglected during the normal operation since the FB biasing current is negligible (tens of
Table 5: "Electrical characteristics "
) but it limits the current flowing in the Zener
diode D1. In case the load is disconnected or in case of open LED:
Equation 72
46/56
R1 must be dimensioned to limit the D1 rated power so it is an inexpensive small signal
Zener diode.
The circuitry Q1, R3, R4, R5 implements a level shifter to convert the dimming signal voltage levels (referred to GND) to the device rails, since the LED5000 local ground is floating. The LED5000 local GND level is:
Equation 73 where V
LGND
represent the local GND value.
Figure 25: "Inverting BB dimming operation"
shows the dimming operation: the light blue
trace represents the DIM pin, the yellow the SW (high level is V
IN
, low level is V
IN
- V
OUT
),
the green trace the inductor current (see
) and the purple is the output voltage.
DocID023951 Rev 6
LED5000 Application notes
- alternative topologies
Figure 32: Floating BB dimming operation
To design the compensation network for the boost topology please refer to
Figure 33: Floating boost PCB layout (component side)
Figure 34: Floating boost PCB layout (bottom side)
DocID023951 Rev 6 47/56
Application notes
- alternative topologies
6.4
LED5000
Compensation network design for alternative topologies
The small signal analysis for the alternative topologies can be written as:
Equation 74
48/56
that shares similar terms with
which is valid for the buck (see
addition K
DX
depends on the topology (different for boost and buck-boost) and w
Z_RHP
) is a zero in the right half plane:
Equation 75
The RHP (right half plane) zero has the same 20 dB/dec rising gain magnitude as a conventional zero, but with 90 degree phase drop instead of lead. This characteristic cannot be compensated with the error amplifier network so the loop gain is designed to roll off at lower frequency in order to keep its contribution outside the small signal analysis.
ω
Z_RHP
) depends on the equivalent output resistance, inductor value and
the duty cycle. As a consequence the minimum ω determines the maximum system bandwidth:
Z_RHP over the input voltage range
Equation 76 the system phase margin depends on K.
This paragraph provides the equations to calculate the components of the compensation network once selected the power components and given the BW specification.
Table 10: "BB and boost parameters"
D
, K m
, K parameters useful for the next calculations of the compensation network.
The DC gain of the total small loop is:
Equation 77
DocID023951 Rev 6
LED5000 Application notes
- alternative topologies where G m
is the error amplifier transconductance, R
EA
the equivalent output resistance of the error amplifier, R
CS
the internal current sense gain (for these parameters refer to
5: "Electrical characteristics "
S
the sensing resistor value, and K
D
can be calculated
Table 10: "BB and boost parameters"
The calculation of the components composing the compensation network depends on the relative position of the pole f p
) and the designed bandwidth BW.
Equation 78
Table 10: BB and boost parameters
Boost Buck-boost
K
D
K m
6.4.1
k
f
p
< BW
In case the pole f p
is inside the system bandwidth BW, the component values composing the compensation network can be calculated as:
Equation 79 and
Equation 80
Where K represents the leading position of the F
Z
bandwidth. In general a decade (K = 10) gives enough phase margin to the overall small loop transfer function.
DocID023951 Rev 6 49/56
Application notes
- alternative topologies
6.4.2 fp > BW
LED5000
In case the pole f p
is outside the system bandwidth BW, the component values composing the compensation network can be calculated as:
Equation 81 and
Equation 82 where K represents the leading position of the F
Z
p
.
50/56 DocID023951 Rev 6
LED5000
7
7.1
Package information
Package information
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK ® packages, depending on their level of environmental compliance. ECOPACK ® specifications, grade definitions and product status are available at: www.st.com
.
ECOPACK ® is an ST trademark.
HSOP8 package information
Figure 35: HSOP8 package outline
DocID023951 Rev 6 51/56
Package information
Symbol
E2 e h
D
D1
E
E1
L k
A
A1
A2 b c
Table 11: HSOP8 package mechanical data
Dimensions (mm)
Min. Typ.
1.25
0.38
0.17
4.80
3.10
5.80
3.80
2.20
4.90
3.30
6.00
3.90
2.40
1.27
0.30
0.45
0°
LED5000
Max.
1.75
0.15
0.51
0.25
5.00
3.50
6.20
4.00
2.60
0.50
0.80
8°
52/56 DocID023951 Rev 6
LED5000
8 Ordering information
Order code
LED5000PHR
Table 12: Order code
Package
HSOP8
Ordering information
Packing
Tape & reel
DocID023951 Rev 6 53/56
Revision history LED5000
9 Revision history
Date
31-Jan-2013
Revision
1
Table 12: Document revision history
Changes
Initial release.
27-Feb-2014 2
Updated package name in package photo on page 1 (replaced “HPSO8” by
“HSOP8”).
Updated Section : Features on page 1 (replaced “200 mW” by “200 mΩ” in
“typical R
DSON
”).
Updated Table 3 on page 7 (added note 1 below Table 3).
Updated Figure 3 on page 10 (replaced by a new block diagram).
Updated Table 6 on page 13 (changed “transconductance“ from 2200 µS to
220 µS).
Updated Section 5.2 on page 14 [updated text below Equation 1 - added
“equal to 0.38”. Added “where: ESR is the equivalent series resistor to the output capacitor.” below Equation 2 on page 15. Added “equal to 1.2 V” below Equation 4 on page 15 to “(V
PP
peak to peak amplitude)” ].
Updated Section 5.3 (updated text below Equation 8 on page 16 - replaced
“Where
Avo = Gm · Ro” by “Where A
V0
= G m
· R
0
(R
O
= output resistor of OTA = 200
* 10 ^ 6 Ω)”.
Updated Section 5.4 (updated Equation 12 on page 18 - replaced “R
SENSE
” by “R
S
”, added “where R below Equation 12).
S
is the resistor put in series to the LED string”
Updated Section 5.6 (updated text below Equation 16 on page 20 - replaced text “R
S
the sensing resistor value” by “R
S
the resistor put in series to the
LED string” and “R
CS
the current sense gain” by “R
CS
the equivalent sensing resistor of the current sense circuitry equal to 0.38”).
Updated Section 5.7 on page 20 (replaced “r
LED
“Design specification”) .
= 1.1 W” by “r
LED
= 1.1 Ω“ in
Updated Equation 29 on page 26 (added equations).
Updated Equation 41 on page 31 (replaced “TSW“ by “TSW_EQ“, added text
“where: TSW_EQ = (TRISE + TFALL )/2 = 12 nS” below Equation 41).
Updated text below Equation 43 on page 31 (updated package name - replaced “HPSO8” by “HSOP8”, replaced “60” of “the estimated junction temperature” by “40”).
Updated Figure 20 on page 34 (replaced by a new application circuit - replaced “R56” by “R27”).
Updated Table 9 on page 35 [replaced “MSS7341”-473MLD” by
“XFL6060 -473ME”, “I
SAT
= 1.0 A (30% drop)” by “I
SAT
= 1.8 A (10% drop)” and
“I
RMS
= 1.85 A (40 °C rise)” by “I
RMS
= 3.7 A (40 °C rise)”.
of “L1“ component ].
Updated Figure 24 on page 39, Figure 29 on page 43 and Figure 31 on page 46 (replaced by new figures).
Updated Section 7 on page 51 (updated titles, reversed order of Figure 35 and Table 11, minor modifications).
Updated package name in Table 12 on page 53 (replaced “HPSO8” by
“HSOP8”).
Updated cross-references throughout document.
Minor modifications throughout document.
54/56 DocID023951 Rev 6
LED5000
Date
28-Apr-2014
30-Sep-2014
04-Apr-2016
26-Jul-2017
Revision
3
4
5
6
Revision history
Changes
Updated Section 6.1: Inverting buck-boost on page 37 (replaced 10 LEDs by
5 LEDs in text above Figure 24 on page 39).
Updated Figure 24 on page 39 (replaced by new figure).
Updated Section 8: "Ordering information".
Minor text changes.
Minor text changes.
DocID023951 Rev 6 55/56
LED5000
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Purchasers are solely responsible for the choice, selection, and use of ST products and ST assumes no liability for application assistance or the design of Purchasers’ products.
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Resale of ST products with provisions different from the information set forth herein shall void any warranty granted by ST for such product.
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Information in this document supersedes and replaces information previously supplied in any prior versions of this document.
© 2017 STMicroelectronics – All rights reserved
56/56 DocID023951 Rev 6
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