The HVLED805 is a high-voltage primary switcher intended for operating directly from the rectified mains with minimum external parts to provide an efficient, compact and cost effective solution for LED driving. It combines a high-performance low-voltage PWM controller chip and an 800V, avalanche-rugged power MOSFET, in the same package.

The PWM is a current-mode controller IC specifically designed for ZVS (zero voltage switching) fly-back LED drivers, with constant output current (CC) regulation using primarysensing feedback. This eliminates the need for the opto-coupler, the secondary voltage reference, as well as the current sense on the secondary side, still maintaining a good LED current accuracy. Moreover it guarantees a safe operation when short circuit of one or more

LEDs occurs.

In addition, the device can also provide a constant output voltage regulation (CV): it makes the application able to work safely when the LED string opens due to a failure.

Quasi-resonant operation is achieved by means of a transformer demagnetization sensing input that triggers MOSFET’s turn-on. This input serves also as both output voltage monitor, to perform CV regulation, and input voltage monitor, to achieve mains-independent CC regulation (line voltage feed forward).

The maximum switching frequency is top-limited below 166 kHz, so that at medium-light load a special function automatically lowers the operating frequency still maintaining the operation as close to ZVS as possible. At very light load, the device enters a controlled burst-mode operation that, along with the built-in high-voltage start-up circuit and the low operating current of the device, helps minimize the residual input consumption.

Although an auxiliary winding is required in the transformer to correctly perform CV/CC regulation, the chip is able to power itself directly from the rectified mains. This is useful especially during CC regulation, where the fly-back voltage generated by the winding drops.

In addition to these functions that optimize power handling under different operating conditions, the device offers protection features that considerably increase end-product’s safety and reliability: auxiliary winding disconnection or brownout detection and shorted secondary rectifier or transformer’s saturation detection. All of them are auto restart mode.

Doc ID 18077 Rev 1 3/29

Maximum ratings HVLED805

Table 2.

Symbol

Absolute maximum ratings

Pin Parameter

E av

V cc

DMG

V comp

P tot

T stg

1,2, 13-16 Drain-to-source (ground) voltage

1,2, 13-16 Drain current

(1)

1,2, 13-16 Single pulse avalanche energy (T j

= 25°C, I

= 0.7A)

3 Supply voltage (Icc < 25mA)

Zero current detector current

Analog input

Power dissipation @T

= 50°C

Junction temperature range

Storage temperature

Limited by maximum temperature allowed.

Value Unit

-1 to 800

A mJ

Self limiting V

±2 mA

-0.3 to 3.6

0.9

-40 to 150 °C

-55 to 150 °C

Table 3.

Symbol

R thJP

R thJA

Thermal data

Parameter

Thermal resistance, junction-to-pin

Thermal resistance, junction-to-ambient

Max. value Unit

110

°C/W

4/29 Doc ID 18077 Rev 1

HVLED805 Electrical characteristics

= -25 to 125 °C, Vcc=14 V; unless otherwise specified.

Table 4.

Symbol

Electrical characteristics

Parameter Test condition Min.

Typ. Max. Unit

Start-up timer

RESTART

Start timer period

START

Restart timer period during burst mode

Demagnetization detector

DMGb

Input bias current V

DMG

= 0.1 to 3V

Doc ID 18077 Rev 1

105 140 175 µs

420 500 700 µs

0.1

1 µA

5/29

Electrical characteristics HVLED805

Table 4.

Symbol

Electrical characteristics (continued)

Parameter

Equivalent feedforward resistor

Transconductance error amplifier

Test condition

DMGH

DMGL

DMGA

DMGT

DMGON

BLANK

Upper clamp voltage

Lower clamp voltage

Arming voltage

Triggering voltage

DMG

= 1 mA

= - 1 mA positive-going edge negative-going edge

Min. source current during MOSFET ON-time

Trigger blanking time after MOSFET’s turn-off

COMP

≥

1.3V

COMP

= 0.9V

Line feedforward

DMG

= 1mA

REF gm

Voltage reference

Transconductance

COMP

Voltage gain

Gain-bandwidth product

Source current

Sink current

COMPH

COMPL

Upper COMP voltage

Lower COMP voltage

COMPBM

Burst-mode threshold

Hys Burst-mode hysteresis

Min.

Typ. Max. Unit

3.0

3.3

3.6

-90 -60 -30 mV

100 110 120 mV

50 60 70 mV

-25 -50 -75 µA

µs

Tj = 25 °C

(1)

Tj = -25 to 125°C and

Vcc=12V to 23V

(1)

COMP

= ±10 µA

COMP

= 1.65 V

Open loop

2.45

2.4

1.3

2.51

2.2

500

DMG

= 2.3V, V

COMP

= 1.65V

70 100

DMG

= 2.7V, V

COMP

= 1.65V

400 750

DMG

= 2.3V

2.7

DMG

= 2.7V

0.7

2.57

2.6

3.2

V mS dB kHz

µA

V mV

Current reference

ILEDx

CLED

Maximum value

Current reference voltage

Current sense

COMP

= V

COMPL

(1)

1.5

1.6

1.7

0.192

0.2

0.208

V t

LEB

Leading-edge blanking t d(H-L) Delay-to-output

CSx

CSdis

Max. clamp value

Hiccup-mode OCP level

Parameters tracking each other

(1) dVcs/dt = 200 mV/µs

(1)

200 250 300 ns

300 ns

0.7

0.75

0.8

0.92

1 1.08

6/29 Doc ID 18077 Rev 1

HVLED805 Pin connection

Figure 2.

Pin connection (top view)

SOURCE 1

2 SOURCE

VCC

GND

ILED

DMG

COMP

N.A.

DRAIN

N.C.

Note: The copper area for heat dissipation has to be designed under the drain pins

Doc ID 18077 Rev 1 7/29

Pin connection HVLED805

Table 5.

Pin functions

Name Function

1, 2 SOURCE

Power section source and input to the PWM comparator. The current flowing in the MOSFET is sensed through a resistor connected between the pin and GND. The resulting voltage is compared with an internal reference (0.75V typ.) to determine MOSFET’s turn-off. The pin is equipped with 250 ns blanking time after the gate-drive output goes high for improved noise immunity. If a second comparison level located at 1V is exceeded the IC is stopped and restarted after Vcc has dropped below 5V.

VCC

GND

Supply Voltage of the device. An electrolytic capacitor, connected between this pin and ground, is initially charged by the internal high-voltage start-up generator; when the device is running the same generator will keep it charged in case the voltage supplied by the auxiliary winding is not sufficient. This feature is disabled in case a protection is tripped. Sometimes a small bypass capacitor (100nF typ.) to GND might be useful to get a clean bias voltage for the signal part of the IC.

Ground. Current return for both the signal part of the IC and the gate drive. All of the ground connections of the bias components should be tied to a trace going to this pin and kept separate from any pulsed current return.

8-11

ILED

DMG

COMP

N.A

N.C

CC regulation loop reference voltage. An external capacitor will be connected between this pin and GND. An internal circuit develops a voltage on this capacitor that is used as the reference for the MOSFET’s peak drain current during CC regulation. The voltage is automatically adjusted to keep the average output current constant.

Transformer’s demagnetization sensing for quasi-resonant operation. Input/output voltage monitor. A negative-going edge triggers MOSFET’s turn-on. The current sourced by the pin during MOSFET’s ON-time is monitored to get an image of the input voltage to the converter, in order to compensate the internal delay of the current sensing circuit and achieve a CC regulation independent of the mains voltage. If this current does not exceed 50µA, either a floating pin or an abnormally low input voltage is assumed, the device is stopped and restarted after Vcc has dropped below 5V. Still, the pin voltage is sampled-and-held right at the end of transformer’s demagnetization to get an accurate image of the output voltage to be fed to the inverting input of the internal, transconductance-type, error amplifier, whose noninverting input is referenced to 2.5V. Please note that the maximum I

DMG

sunk/sourced current has to not exceed ±2 mA (AMR) in all the Vin range conditions. No capacitor is allowed between the pin and the auxiliary transformer.

Output of the internal transconductance error amplifier. The compensation network will be placed between this pin and GND to achieve stability and good dynamic performance of the voltage control loop.

Not available. These pins must be left not connected

13 to 16 DRAIN

Not internally connected. Provision for clearance on the PCB to meet safety requirements.

Drain connection of the internal power section. The internal high-voltage start-up generator sinks current from this pin as well. Pins connected to the internal metal frame to facilitate heat dissipation.

8/29 Doc ID 18077 Rev 1

HVLED805

Figure 3.

OSS

output capacitance variation

500

OSS

(pF)

400

300

200

100

0 25 50

(V)

100 125 150

Pin connection

Figure 4.

Off state drain and source current test circuit

1 4V

Idss

2.5V

D MG

COM P IL ED

V CC

CUR RE NT

CON TR OL

D RA IN

G ND S OUR CE

V in

75 0V

Note: The measured I

DSS

is the sum between the current across the 12 M Ω start-up resistor (62.5

µA typ. @ 750 V) and the effective MOSFET’s off state drain current

Doc ID 18077 Rev 1 9/29

Pin connection

Figure 5.

Start-up current test circuit

Icc sta rt-u p

2.5V

D MG

1 1.8 V

COM P IL ED

V CC

CUR RE NT

CON TR OL

D RA IN

G ND

S OUR CE

HVLED805

10/29

Figure 6.

Quiescent current test circuit

Iq _m ea s

14 V

33 k

2 .5V

DM G

C OMP ILE D

VC C

C UR REN T

C ONTRO L

DR AIN

GN D SO URC E

1 0k

0 .2V

0.8 V

Iq = Iq_meas -

3.3k

Ω

Doc ID 18077 Rev 1

HVLED805

Figure 7.

Operating supply current test circuit

Pin connection

10 k

27 k

2 20 k

2.5 V

D MG

10 k

1 0

5 0kHz

-5 V

Icc

15 V

V CC

CU RRE NT

CO NTR OL

DRA IN

1 .5k

2 W

15 0V

CO MP IL ED

GND

S OU RCE

5 .6

+ 2 .8V

Note: The circuit across the DMG pin is used for switch-on synchronization

Figure 8.

Quiescent current during fault test circuit

Iq (fa ult)

1 4V

2.5V

D MG

COM P IL ED

V CC

CUR RE NT

CON TR OL

D RA IN

G ND

S OUR CE

Doc ID 18077 Rev 1 11/29

Application information HVLED805

The HVLED805 is an off-line all-primary sensing switching regulator, specific for offline LED drivers based on quasi-resonant ZVS (zero voltage switching at switch turn-on) flyback topology.

Depending on converter’s load condition, the device is able to work in different modes

(

Figure 9 for constant voltage operation):

QR mode at heavy load. Quasi-resonant operation lies in synchronizing MOSFET's turn-on to the transformer’s demagnetization by detecting the resulting negative-going edge of the voltage across any winding of the transformer. Then the system works close to the boundary between discontinuous (DCM) and continuous conduction

(CCM) of the transformer. As a result, the switching frequency will be different for

different line/load conditions (see the hyperbolic-like portion of the curves in Figure 9

Minimum turn-on losses, low EMI emission and safe behavior in short circuit are the main benefits of this kind of operation. The resulting constant current mode fixes the average current also in case of a short-circuit failure of one or more LEDs.

2. Valley-skipping mode at medium/ light load. Depending on voltage on COMP pin, the device defines the maximum operating frequency of the converter. As the load is reduced MOSFET’s turn-on will not any more occur on the first valley but on the second one, the third one and so on. In this way the switching frequency will no longer increase

(piecewise linear portion in

Figure 9 ).

3. Burst-mode with no or very light load. When the load is extremely light or disconnected, the converter will enter a controlled on/off operation with constant peak current.

Decreasing the load will then result in frequency reduction, which can go down even to few hundred hertz, thus minimizing all frequency-related losses and making it easier to comply with energy saving regulations or recommendations. Being the peak current very low, no issue of audible noise arises. Thanks to this feature, the application is able to safely manage the open circuit caused by an LED failure.

Figure 9.

Multi-mode operation of HVLED805 (Constant voltage operation) f f f f f f

12/29

Doc ID 18077 Rev 1

HVLED805

5.1

Application information

Power section and gate driver

The power section guarantees safe avalanche operation within the specified energy rating as well as high dv/dt capability. The Power MOSFET has a V(BR)DSS of 800V min. and a typical R

DSon of 11

Ω .

The gate driver of the power MOSFET is designed to supply a controlled gate current during both turn-on and turn-off in order to minimize common mode EMI. Under UVLO conditions an internal pull-down circuit holds the gate low in order to ensure that the power MOSFET cannot be turned on accidentally.

5.2 High voltage startup generator

Figure 10

shows the internal schematic of the high-voltage start-up generator (HV generator). It includes an 800 V-rated N-channel MOSFET, whose gate is biased through the series of a 12 M

resistor and a 14 V zener diode, with a controlled, temperaturecompensated current generator connected to its source. The HV generator input is in common with the DRAIN pin, while its output is the supply pin of the device (Vcc). A mains

“UVLO” circuit (separated from the UVLO of the device that sense Vcc) keeps the HV generator off if the drain voltage is below V

START

(50 V typical value).

Figure 10.

High-voltage start-up generator: internal schematic

DRAIN

Vc c _O K

14 V 12M

Mai ns UV LO

H V_ EN

CO NTRO L

Ic ha rge

S OURCE

IHV

Vcc

With reference to the timing diagram of

Figure 11

, when power is applied to the circuit and the voltage on the input bulk capacitor is high enough, the HV generator is sufficiently biased to start operating, thus it will draw about 5.5 mA (typical) from the bulk capacitor.

Doc ID 18077 Rev 1 13/29

Application information HVLED805

Most of this current will charge the bypass capacitor connected between the Vcc pin and ground and make its voltage rise linearly.

As the Vcc voltage reaches the start-up threshold (13 V typ.) the chip starts operating, the internal power MOSFET is enabled to switch and the HV generator is cut off by the Vcc_OK signal asserted high. The IC is powered by the energy stored in the Vcc capacitor.

The chip is able to power itself directly from the rectified mains: when the voltage on the V

CC pin falls below Vcc restart

(10.5V typ.) , during each MOSFET’s off-time the HV current generator is turned on and charges the supply capacitor until it reaches the V

CCOn threshold.

In this way, the self-supply circuit develops a voltage high enough to sustain the operation of the device. This feature is useful especially during CC regulation, when the flyback voltage generated by the auxiliary winding alone may not be able to keep Vcc above V

CCrestart

At converter power-down the system will lose regulation as soon as the input voltage falls below V

Start

. This prevents converter’s restart attempts and ensures monotonic output voltage decay at system power-down.

Figure 11.

Timing diagram: normal power-up and power-down sequences

V Start

Vcc

Vcc restart t t t t

I I charge

Normal operation

CV mode

Normal operation

CC mode t tt t tt t t

14/29 Doc ID 18077 Rev 1

HVLED805 Application information

5.3 Secondary demagnetization detection and triggering block

The demagnetization detection (DMG) and Triggering blocks switch on the power MOSFET if a negative-going edge falling below 50 mV is applied to the DMG pin. To do so, the triggering block must be previously armed by a positive-going edge exceeding 100 mV.

This feature is used to detect transformer demagnetization for QR operation, where the signal for the DMG input is obtained from the transformer’s auxiliary winding used also to supply the IC.

Figure 12.

DMG block, triggering block

Aux

R dmg

DMG

Rfb

D MG

CLAMP

110mV

60mV

BLAN KIN G

TIME

STAR TER

From CC/C V Block

TUR N-ON

LOGIC

LEB R

From OCP

To Driver

The triggering block is blanked after MOSFET’s turn-off to prevent any negative-going edge that follows leakage inductance demagnetization from triggering the DMG circuit erroneously.

This blanking time is dependent on the voltage on COMP pin: it is T

BLANK

= 30 µs for V

COMP

= 0.9 V, and decreases almost linearly down to T

BLANK

= 6 µs for V

COMP

= 1.3 V

The voltage on the pin is both top and bottom limited by a double clamp, as illustrated in the internal diagram of the DMG block of

Figure 12

. The upper clamp is typically located at 3.3

V, while the lower clamp is located at -60mV. The interface between the pin and the auxiliary winding will be a resistor divider. Its resistance ratio as well as the individual resistance

values will be properly chosen (see “ Section 5.5: Constant current operation on page 18

”

and “ Section 5.6: Voltage feedforward block on page 20

”.

Please note that the maximum I

DMG

sunk/sourced current has to not exceed ±2 mA (AMR) in all the Vin range conditions. No capacitor is allowed between DMG pin and the auxiliary transformer.

The switching frequency is top-limited below 166 kHz, as the converter’s operating frequency tends to increase excessively at light load and high input voltage.

A Starter block is also used to start-up the system, that is, to turn on the MOSFET during converter power-up, when no or a too small signal is available on the DMG pin.

The starter frequency is 2 kHz if COMP pin is below burst mode threshold, i.e. 1 V, while it becomes 8 kHz if this voltage exceed this value.

Doc ID 18077 Rev 1 15/29

Application information HVLED805

After the first few cycles initiated by the starter, as the voltage developed across the auxiliary winding becomes large enough to arm the DMG circuit, MOSFET’s turn-on will start to be locked to transformer demagnetization, hence setting up QR operation.

The starter is activated also when the IC is in CC regulation and the output voltage is not high enough to allow the DMG triggering.

If the demagnetization completes – hence a negative-going edge appears on the DMG pin – after a time exceeding time T

BLANK

from the previous turn-on, the MOSFET will be turned on again, with some delay to ensure minimum voltage at turn-on. If, instead, the negativegoing edge appears before T

BLANK

has elapsed, it will be ignored and only the first negativegoing edge after T

BLANK

will turn-on the MOSFET. In this way one or more drain ringing cycles will be skipped (“valley-skipping mode”,

Figure 13

) and the switching frequency will be prevented from exceeding 1/T

BLANK

Figure 13.

Drain ringing cycle skipping as the load is progressively reduced

DS t

T osc

P in

= P in''

< P in' t

T osc

P in

= P in'''

< P in'' t

T osc

P in

= P in'

(limit condition)

Note: That when the system operates in valley skipping-mode, uneven switching cycles may be observed under some line/load conditions, due to the fact that the OFF-time of the MOSFET is allowed to change with discrete steps of one ringing cycle, while the OFF-time needed for cycle-by-cycle energy balance may fall in between. Thus one or more longer switching cycles will be compensated by one or more shorter cycles and vice versa. However, this mechanism is absolutely normal and there is no appreciable effect on the performance of the converter or on its output voltage.

The IC is specifically designed to work in primary regulation and the output voltage is sensed through a voltage partition of the auxiliary winding, just before the auxiliary rectifier diode.

Figure 14

shows the internal schematic of the constant voltage mode and the external connections.

16/29 Doc ID 18077 Rev 1

HVLED805 Application information

Figure 14.

Voltage control principle: internal schematic

Rdmg

DMG

Aux

Rfb

S/H

DEMAG

LOGIC

2.5V

To PWM Logic

From Rsense

COMP

Due to the parasitic wires resistance, the auxiliary voltage is representative of the output just when the secondary current becomes zero. For this purpose, the signal on DMG pin is sampled-and-held at the end of transformer’s demagnetization to get an accurate image of the output voltage and it is compared with the error amplifier internal reference.

During the MOSFET’s OFF-time the leakage inductance resonates with the drain capacitance and a damped oscillation is superimposed on the reflected voltage. The S/H logic is able to discriminate such oscillations from the real transformer’s demagnetization.

When the DMG logic detects the transformer’s demagnetization, the sampling process stops, the information is frozen and compared with the error amplifier internal reference.

The internal error amplifier is a transconductance type and delivers an output current proportional to the voltage unbalance of the two outputs: the output generates the control voltage that is compared with the voltage across the sense resistor, thus modulating the cycle-by-cycle peak drain current.

The COMP pin is used for the frequency compensation: usually, an RC network, which stabilizes the overall voltage control loop, is connected between this pin and ground.

The output voltage can be defined according the formula:

Equation 1

REF n

AUX n

SEC

⋅

OUT

−

REF

⋅

DMG

Where n

SEC

and n

AUX

are the secondary and auxiliary turn’s number respectively.

The R

DMG

value can be defined depending on the application parameters (see “ Section 5.6:

Voltage feedforward block on page 20 ” section).

Doc ID 18077 Rev 1 17/29

Application information HVLED805

Figure 15

presents the principle used for controlling the average output current of the flyback converter.

The output voltage of the auxiliary winding is used by the demagnetization block to generate the control signal for the mosfet switch Q1. A resistor R in series with it absorbs a current

/R, where V

is the voltage developed across the capacitor C.

The flip-flop’s output is high as long as the transformer delivers current on secondary side.

This is shown in

Figure 16 .

The capacitor C has to be chosen so that its voltage V

can be considered as a constant.

Since it is charged and discharged by currents in the range of some ten µA (I

CLED

is typically 20 µA) at the switching frequency rate, a capacitance value in the range 4.7-10 nF is suited for switching frequencies in the ten kHz.

The average output current can be expressed as:

Equation 2

OUT

⋅

⎛

⎝

ONSEC

⎞

⎠

Where I

is the secondary peak current, T

ONSEC

is the conduction time of the secondary side and T is the switching period.

Taking into account the transformer ratio n between primary and secondary side, I

can also be expressed is a function of the primary peak current I

Equation 3

= n

⋅

As in steady state the average current I

Equation 4

CLED

⋅

(

−

ONSEC

)

⎛

⎝

CLED

−

⎞

⎠

⋅

ONSEC

Which can be solved for V

Equation 5

CLED

⋅

ONSEC

Where V

CLED

•

LED

and is internally defined.

As V

is fed to the CC comparator, the primary peak current can be expressed as:

18/29 Doc ID 18077 Rev 1

HVLED805 Application information

Equation 6

SENSE

Combining (2), (3) (5) and (6):

Equation 7

OUT

= n

⋅

CLED

SENSE

This formula shows that the average output current does not depend anymore on the input or the output voltage, neither on transformer inductance values. The external parameters defining the output current are the transformer ratio n and the sense resistor R

SENSE

Figure 15.

Current control principle

Iref

To PWM Logic

From R sense

R dmg

DMG

D EMAG

LOGIC

Rfb

Aux

ILED

CLED

Doc ID 18077 Rev 1 19/29

Application information

Figure 16.

Constant current operation: Switching cycle waveforms

P t

I s t

Q t

I CLED t

CLED

= −

HVLED805

5.6 Voltage feedforward block

The current control structure uses the voltage V

to define the output current, according to

(7). Actually, the CC comparator will be affected by an internal propagation delay Td, which will switch off the MOSFET with a peak current than higher the foreseen value.

This current overshoot will be equal to:

Equation 8

Δ I

⋅

P d

Will introduce an error on the calculated CC setpoint, depending on the input voltage.

The HVLED805 implements a Line Feedforward function, which solves the issue by introducing an input voltage dependent offset on the current sense signal, in order to adjust the cycle-by-cycle current limitation.

The internal schematic is shown in Figure 17 .

20/29 Doc ID 18077 Rev 1

HVLED805 Application information

Figure 17.

Feedforward compensation: internal schematic

Aux

Rdmg DMG

Rfb

Feedforward

Logic

IFF

Block

PWM

LOGIC

DRAIN

RFF

SO URCE

Rsense

During MOSFET’s ON-time the current sourced from DMG pin is mirrored inside the

“ Feedforward Logic ” block in order to provide a feedforward current, I

Such “feedforward current” is proportional to the input voltage according to the formula:

Equation 9

= m

⋅

R dmg

Where m is the primary-to-auxiliary turns ratio.

According to the schematic, the voltage on the non-inverting comparator will be:

Equation 10

⋅ I +I

⋅ (

R +R

SENSE

)

The offset introduced by feedforward compensation will be:

Equation 11

OFFSET

= m

⋅

R dmg

⋅

(

SENSE

)

As R

>>R

SENSE

, the previous one can be simplified as:

Equation 12

OFFSET

IN m

⋅

R dmg

Doc ID 18077 Rev 1 21/29

Application information HVLED805

This offset is proportional to V

and is used to compensate the current overshoot, according to the formula:

Equation 13

⋅

T d

L p

⋅

SENSE

IN m

⋅

R dmg

Finally, the R dmg

resistor can be calculated as follows:

Equation 14

R dmg

AUX

PRI

⋅

T d

L p

⋅

SENSE

In this case the peak drain current does not depend on input voltage anymore.

One more consideration concerns the R dmg

value: during MOSFET’s ON-time, the current sourced by the DMG pin, I

DMG

, is compared with an internal reference current I

DMGON

(-50

µA typical).

If I

DMG

< I

DMGON

, the brownout function is activated and the IC is shut-down.

This feature is especially important when the auxiliary winding is accidentally disconnected and considerably increases the end-product’s safety and reliability.

5.7 Burst-mode operation at no load or very light load

When the voltage at the COMP pin falls 65 mV below a threshold fixed internally at a value,

COMPBM

, the IC is disabled with the MOSFET kept in OFF state and its consumption reduced at a lower value to minimize Vcc capacitor discharge.

In this condition the converter operates in burst-mode (one pulse train every T

START

=500

µs), with minimum energy transfer.

As a result of the energy delivery stop, the output voltage decreases: after 500 µs the controller switches-on the MOSFET again and the sampled voltage on the DMG pin is compared with the internal reference. If the voltage on the EA output, as a result of the comparison, exceeds the V

COMPL

threshold, the device restarts switching, otherwise it stays

OFF for another 500 µs period.

In this way the converter will work in burst-mode with a nearly constant peak current defined by the internal disable level. A load decrease will then cause a frequency reduction, which can go down even to few hundred hertz, thus minimizing all frequency-related losses and making it easier to comply with energy saving regulations. This kind of operation, shown in

the timing diagrams of Figure 19 along with the others previously described, is noise-free

since the peak current is low

22/29 Doc ID 18077 Rev 1

HVLED805 Application information

Figure 18.

Load-dependent operating modes: timing diagrams

COMP

COMPL

65 mV hyster.

Normal-mode

START

Burst-mode

START

Normal-mode

5.8

5.9 Soft-start and starter block

The soft start feature is automatically implemented by the constant current block, as the primary peak current will be limited from the voltage on the C

LED

capacitor.

During start-up, as the output voltage is zero, the IC will start in CC mode with no high peak current operations. In this way the voltage on the output capacitor will increase slowly and the soft-start feature will be ensured.

Actually the C

LED

value is not important to define the soft-start time, as its duration depends on others circuit parameters, like transformer ratio, sense resistor, output capacitors and load. The user will define the best appropriate value by experiments.

Hiccup mode OCP

The device is also protected against short circuit of the secondary rectifier, short circuit on the secondary winding or a hard-saturated flyback transformer. A comparator monitors continuously the voltage on the R

SENSE

and activates a protection circuitry if this voltage exceeds 1 V.

To distinguish an actual malfunction from a disturbance (e.g. induced during ESD tests), the first time the comparator is tripped the protection circuit enters a “warning state”. If in the subsequent switching cycle the comparator is not tripped, a temporary disturbance is assumed and the protection logic will be reset in its idle state; if the comparator will be tripped again a real malfunction is assumed and the device will be stopped.

This condition is latched as long as the device is supplied. While it is disabled, however, no energy is coming from the self-supply circuit; hence the voltage on the V

capacitor will decay and cross the UVLO threshold after some time, which clears the latch. The internal start-up generator is still off, then the V

voltage still needs to go below its restart voltage

Doc ID 18077 Rev 1 23/29

Application information HVLED805 before the V

capacitor is charged again and the device restarted. Ultimately, this will result in a low-frequency intermittent operation (Hiccup-mode operation), with very low stress on the power circuit. This special condition is illustrated in the timing diagram of

Figure 18

Figure 19.

Hiccup-mode OCP: timing diagram

Secondary diode is shorted here

Vcc

OFF

Vcc rest

SOURCE

Vcs dis

1 V t t

V DS

Two switching cycles t

A proper printed circuit board layout is essential for correct operation of any switch-mode converter and this is true for the HVLED805 as well. Careful component placing, correct traces routing, appropriate traces widths and compliance with isolation distances are the major issues. In particular:

● The compensation network should be connected as close as possible to the COMP pin, maintaining the trace for the GND as short as possible

● Signal ground should be routed separately from power ground, as well from the sense resistor trace.

24/29 Doc ID 18077 Rev 1

HVLED805

Figure 20.

Suggested routing for converter

ACIN

VDD DRAIN

DMG

HVLED805

COMP GND ILED SOURCE

Application information

LED

...

Doc ID 18077 Rev 1 25/29

Package mechanical data

6 Package mechanical data

HVLED805

In order to meet environmental requirements, ST offers these devices in different grades of

ECOPACK

packages, depending on their level of environmental compliance. ECOPACK

® specifications, grade definitions and product status are available at: www.st.com

ECOPACK

is an ST trademark.

Table 6.

Dim.

SO16N mechanical data mm inch

Min Typ Max Min Typ Max a1 0.1 0.009 b b1 0.19

C 0.5

0.010

0.020

D (1) 9.8

E 5.8

10 0.386 0.394

0.244 e 1.27 0.050 e3 8.89 0.350

F (1)

G 4.60

0.208

26/29 Doc ID 18077 Rev 1

HVLED805

Figure 21.

Package dimensions

Package mechanical data

Doc ID 18077 Rev 1 27/29

Revision history

Table 7.

Date

Document revision history

Revision

14-Oct-2010 1 Initial release

Changes

HVLED805

28/29 Doc ID 18077 Rev 1

HVLED805

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Doc ID 18077 Rev 1 29/29

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